LTC3406B-2 2.25MHz, 600mA Synchronous Step-Down Regulator in ThinSOTTM U FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ DESCRIPTIO The LTC ®3406B-2 is a high efficiency monolithic synchronous buck regulator using a constant frequency, current mode architecture. Supply current with no load is 350µA, dropping to <1µA in shutdown. The 2.5V to 5.5V input voltage range makes the LTC3406B-2 ideally suited for single Li-Ion battery-powered applications. 100% duty cycle capability provides low dropout operation, extending battery life in portable systems. PWM pulse skipping mode operation provides very low output ripple voltage for noise sensitive applications. High Efficiency: Up to 96% 600mA Output Current at VIN = 3V 2.5V to 5.5V Input Voltage Range 2.25MHz Constant Frequency Operation No Schottky Diode Required Low Dropout Operation: 100% Duty Cycle Low Quiescent Current: 350µA 0.6V Reference Allows Low Output Voltages Shutdown Mode Draws < 1µA Supply Current Current Mode Operation for Excellent Line and Load Transient Response Overtemperature Protected Low Profile (1mm) SOT-23 Package The switching frequency is internally set at 2.25MHz, allowing the use of tiny surface mount inductors and capacitors. The internal synchronous switch increases efficiency and eliminates the need for an external Schottky diode. Low output voltages are easily supported with the 0.6V feedback reference voltage. The LTC3406B-2 is available in a low profile (1mm) SOT-23 package. Refer to LTC3406 for applications that require Burst Mode® operation. U APPLICATIO S ■ ■ ■ ■ ■ Cellular Telephones Personal Information Appliances Wireless and DSL Modems Digital Still Cameras MP3 Players Portable Instruments , LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode is a registered trademark of Linear Technology Corporation. ThinSOT is a trademark of Linear Technology Corporation. Protected by U.S. Patents, including 6580258, 5481178. U ■ TYPICAL APPLICATIO High Efficiency Step-Down Converter Efficiency vs Load Current 100 2.2µH* VIN 4.7µF CER SW LTC3406B-2 10µF CER 22pF 1M VFB RUN GND 499k 3406B TA01a VOUT 1.8V 600mA 90 VOUT = 1.8V TA = 25°C 80 EFFICIENCY (%) VIN 2.7V TO 5.5V 70 60 VIN = 3.6V VIN = 2.7V 50 40 30 20 10 0.1 VIN = 4.2V 1 100 10 OUTPUT CURRENT (mA) 1000 3406B TA01b sn3406b2 3406b2fs 1 LTC3406B-2 W W W AXI U U ABSOLUTE RATI GS U U W PACKAGE/ORDER I FOR ATIO (Note 1) Input Supply Voltage .................................. – 0.3V to 6V RUN, VFB Voltages ..................................... – 0.3V to VIN SW Voltage (DC) ......................... – 0.3V to (VIN + 0.3V) P-Channel Switch Source Current (DC) ............. 800mA N-Channel Switch Sink Current (DC) ................. 800mA Peak SW Sink and Source Current ........................ 1.3A Operating Temperature Range (Note 2) .. – 40°C to 85°C Maximum Junction Temperature (Notes 3, 6) ..... 125°C Storage Temperature Range ................ – 65°C to 150°C Lead Temperature (Soldering, 10 sec)................. 300°C ORDER PART NUMBER TOP VIEW 5 VFB RUN 1 LTC3406B-2ES5 GND 2 SW 3 4 VIN S5 PART MARKING S5 PACKAGE 5-LEAD PLASTIC TSOT-23 LTAGH TJMAX = 125°C, θJA = 250°C/ W, θJC = 90°C/ W Consult LTC Marketing for parts specified with wider operating temperature ranges. ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C. VIN = 3.6V unless otherwise specified. SYMBOL PARAMETER IVFB Feedback Current VFB Regulated Feedback Voltage CONDITIONS (Note 4) TA = 25°C (Note 4) 0°C ≤ TA ≤ 85°C (Note 4) –40°C ≤ TA ≤ 85°C ● ∆VFB Reference Voltage Line Regulation VIN = 2.5V to 5.5V (Note 4) ● ∆VOVL ∆Output Overvoltage Lockout ∆VOVL = VOVL – VFB, LTC3406B ● IPK Peak Inductor Current VIN = 3V, VFB = 0.5V or VOUT = 90%, Duty Cycle < 35% VLOADREG Output Voltage Load Regulation VIN Input Voltage Range IS Input DC Bias Current Shutdown MIN TYP MAX UNITS ±30 nA 0.5880 0.5865 0.5850 0.6 0.6 0.6 0.6120 0.6135 0.6150 V V V 0.04 0.4 %/V 20 50 80 mV 0.75 1 1.25 0.5 ● 2.5 (Note 5) VFB = 0.5V or VOUT = 90% VRUN = 0V, VIN = 4.2V ● % 5.5 V 350 0.1 500 1 µA µA 2.25 310 2.7 MHz kHz 0.4 0.5 Ω fOSC Oscillator Frequency VFB = 0.6V or VOUT = 100% VFB = 0V or VOUT = 0V RPFET RDS(ON) of P-Channel FET ISW = 100mA RNFET RDS(ON) of N-Channel FET ISW = –100mA 0.35 0.45 Ω ILSW SW Leakage VRUN = 0V, VSW = 0V or 5V, VIN = 5V ±0.01 ±1 µA VRUN RUN Threshold ● 1 1.5 V IRUN RUN Leakage Current ● ±0.01 ±1 µA Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: The LTC3406B-2ES5 is guaranteed to meet performance specifications from 0°C to 70°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: LTC3406B-2ES5: TJ = TA + (PD)(250°C/W) 1.8 A 0.3 Note 4: The LTC3406B-2ES5 is tested in a proprietary test mode that connects VFB to the output of the error amplifier. Note 5: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. Note 6: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125°C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. sn3406b2 3406b2fs 2 LTC3406B-2 U W TYPICAL PERFOR A CE CHARACTERISTICS (From Figure 1a Except for the Resistive Divider Resistor Values) Efficiency vs Input Voltage TA = 25°C IOUT = 100mA IOUT = 600mA 80 IOUT = 10mA 75 70 65 3 2 VOUT = 1.5V 90 TA = 25°C 80 80 EFFICIENCY (%) EFFICIENCY (%) 85 100 VOUT = 1.2V TA = 25°C 90 95 90 5 4 INPUT VOLTAGE (V) 70 VIN = 3.6V 60 VIN = 2.7V 50 VIN = 4.2V 40 VIN = 4.2V 40 20 10 0.1 1000 3406B G03 Reference Voltage vs Temperature Oscillator Frequency vs Temperature 0.614 VOUT = 2.5V TA = 25°C 1000 1 100 10 OUTPUT CURRENT (mA) 3406B G02 Efficiency vs Output Current 90 50 20 1 100 10 OUTPUT CURRENT (mA) VIN = 3.6V VIN = 2.7V 60 30 10 0.1 6 70 30 3406B G01 100 Efficiency vs Output Current Efficiency vs Output Current 100 EFFICIENCY (%) 100 2.55 VIN = 3.6V VIN = 3.6V VIN = 2.7V VIN = 4.2V 70 60 VIN = 3.6V 50 40 30 2.40 FREQUENCY (MHz) EFFICIENCY (%) 80 REFERENCE VOLTAGE (V) 0.609 0.604 0.599 0.594 2.25 2.10 0.589 20 10 0.1 1 100 10 OUTPUT CURRENT (mA) 0.584 –50 –25 1000 Oscillator Frequency vs Supply Voltage 2.25 2.10 1.95 3 4 5 SUPPLY VOLTAGE (V) 6 3406B G07 100 125 RDS(ON) vs Input Voltage 0.7 TA = 25°C 0.6 1.824 0.5 RDS(ON) (Ω) 2.40 50 25 75 0 TEMPERATURE (°C) 3406B G06 VIN = 3.6V TA = 25°C 1.834 OUTPUT VOLTAGE (V) OSCILLATOR FREQUENCY (MHz) 1.844 TA = 25°C 2 1.95 –50 –25 125 Output Voltage vs Load Current 2.55 1.80 100 3406B G05 3406B G04 2.70 50 25 75 0 TEMPERATURE (°C) 1.814 1.804 0.3 1.794 0.2 1.784 0.1 1.774 MAIN SWITCH 0.4 SYNCHRONOUS SWITCH 0 0 100 200 300 400 500 600 700 800 900 LOAD CURRENT (mA) 3406B G08 0 1 5 4 2 3 INPUT VOLTAGE (V) 6 7 3406B G09 sn3406b2 3406b2fs 3 LTC3406B-2 U W TYPICAL PERFOR A CE CHARACTERISTICS (From Figure 1a Except for the Resistive Divider Resistor Values) Dynamic Supply Current vs Supply Voltage RDS(ON) vs Temperature 0.7 390 450 VIN = 4.2V VIN = 3.6V 0.5 0.4 0.3 0.2 0.1 50 25 75 0 TEMPERATURE (°C) 390 370 350 330 310 290 270 MAIN SWITCH SYNCHRONOUS SWITCH 0 –50 –25 VOUT = 1.8V 430 ILOAD = 0A TA = 25°C 410 DYNAMIC SUPPLY CURRENT (µA) 0.6 DYNAMIC SUPPLY CURRENT (µA) VIN = 2.7V RDS(ON) (Ω) Dynamic Supply Current vs Temperature 250 100 125 2 2.5 3 3.5 4 4.5 5 SUPPLY VOLTAGE (V) 5.5 3406B G10 Switch Leakage vs Temperature 330 310 290 270 250 –50 SWITCH LEAKAGE (pA) 250 150 100 MAIN SWITCH 60 VOUT 10mV/DIV AC COUPLED MAIN SWITCH IL 100mA/DIV 40 20 VIN = 3.6V VOUT = 1.8V ILOAD = 50mA SYNCHRONOUS SWITCH 0 –50 –25 50 25 75 0 TEMPERATURE (°C) 100 125 0 0 1 2 3 4 INPUT VOLTAGE (V) 3406B G13 5 6 Load Step Load Step RUN 5V/DIV VOUT 1V/DIV AC COUPLED VOUT 100mV/DIV AC COUPLED VOUT 100mV/DIV AC COUPLED IL 500mA/DIV IL 500mA/DIV IL 500mA/DIV ILOAD 500mA/DIV ILOAD 500mA/DIV 3406B G16 3406B G15 1µs/DIV 3406B G14 Start-Up from Shutdown VIN = 3.6V 40µs/DIV VOUT = 1.8V ILOAD = 600mA (LOAD: 3Ω RESISTOR) 125 SW 2V/DIV SYNCHRONOUS SWITCH 80 100 Discontinuous Operation RUN = 0V TA = 25°C 100 200 50 0 TEMPERATURE (°C) 3406B G12 Switch Leakage vs Input Voltage 120 VIN = 5.5V RUN = 0V SWITCH LEAKAGE (nA) 350 3406B G11 300 50 6 VIN = 3.6V VOUT = 1.8V 370 I LOAD = 0A VIN = 3.6V 20µs/DIV VOUT = 1.8V ILOAD = 0mA TO 600mA 3406B G17 VIN = 3.6V 20µs/DIV VOUT = 1.8V ILOAD = 50mA TO 600mA 3406B G18 sn3406b2 3406b2fs 4 LTC3406B-2 U W TYPICAL PERFOR A CE CHARACTERISTICS (From Figure 1a Except for the Resistive Divider Resistor Values) Load Step Load Step VOUT 100mV/DIV AC COUPLED VOUT 100mV/DIV AC COUPLED IL 500mA/DIV IL 500mA/DIV ILOAD 500mA/DIV ILOAD 500mA/DIV VIN = 3.6V 20µs/DIV VOUT = 1.8V ILOAD = 100mA TO 600mA 3406B G19 VIN = 3.6V 20µs/DIV VOUT = 1.8V ILOAD = 200mA TO 600mA 3406B G20 U U U PI FU CTIO S RUN (Pin 1): Run Control Input. Forcing this pin above 1.5V enables the part. Forcing this pin below 0.3V shuts down the device. In shutdown, all functions are disabled drawing <1µA supply current. Do not leave RUN floating. VIN (Pin 4): Main Supply Pin. Must be closely decoupled to GND, Pin 2, with a 2.2µF or greater ceramic capacitor. VFB (Pin 5): Feedback Pin. Receives the feedback voltage from an external resistive divider across the output. GND (Pin 2): Ground Pin. SW (Pin 3): Switch Node Connection to Inductor. This pin connects to the drains of the internal main and synchronous power MOSFET switches. sn3406b2 3406b2fs 5 LTC3406B-2 W FU CTIO AL DIAGRA U U SLOPE COMP OSC 4 VIN OSC FREQ SHIFT – – 5Ω + ICOMP + 0.6V VFB + – EA 5 Q R Q RS LATCH VIN – OVDET RUN 0.65V OV SWITCHING LOGIC AND BLANKING CIRCUIT ANTISHOOTTHRU 3 SW + 0.6V REF SHUTDOWN + 1 S IRCMP 2 GND – 3406B BD U OPERATIO (Refer to Functional Diagram) Main Control Loop The LTC3406B-2 uses a constant frequency, current mode step-down architecture. Both the main (P-channel MOSFET) and synchronous (N-channel MOSFET) switches are internal. During normal operation, the internal top power MOSFET is turned on each cycle when the oscillator sets the RS latch, and turned off when the current comparator, ICOMP, resets the RS latch. The peak inductor current at which ICOMP resets the RS latch, is controlled by the output of error amplifier EA. When the load current increases, it causes a slight decrease in the feedback voltage, FB, relative to the 0.6V reference, which in turn, causes the EA amplifier’s output voltage to increase until the average inductor current matches the new load current. While the top MOSFET is off, the bottom MOSFET is turned on until either the inductor current starts to reverse, as indicated by the current reversal comparator IRCMP, or the beginning of the next clock cycle. The comparator OVDET guards against transient overshoots >6.25% by turning the main switch off and keeping it off until the fault is removed. Pulse Skipping Mode Operation At light loads, the inductor current may reach zero or reverse on each pulse. The bottom MOSFET is turned off by the current reversal comparator, IRCMP, and the switch voltage will ring. This is discontinuous mode operation, and is normal behavior for the switching regulator. At very light loads, the LTC3406B-2 will automatically skip pulses in pulse skipping mode operation to maintain output regulation. Refer to LTC3406 data sheet if Burst Mode operation is preferred. Short-Circuit Protection When the output is shorted to ground, the frequency of the oscillator is reduced to about 310kHz, 1/7 the nominal sn3406b2 3406b2fs 6 LTC3406B-2 U OPERATIO (Refer to Functional Diagram) frequency. This frequency foldback ensures that the inductor current has more time to decay, thereby preventing runaway. The oscillator’s frequency will progressively increase to 2.25MHz when VFB rises above 0V. Dropout Operation As the input supply voltage decreases to a value approaching the output voltage, the duty cycle increases toward the maximum on-time. Further reduction of the supply voltage forces the main switch to remain on for more than one cycle until it reaches 100% duty cycle. The output voltage will then be determined by the input voltage minus the voltage drop across the P-channel MOSFET and the inductor. 2.2µH* VIN 2.7V TO 5.5V VIN CIN** 4.7µF CER SW LTC3406B-2 COUT 10µF CER 22pF 1M VFB RUN GND 499k 3406B F01a *MURATA LQH32CN2R2M33 **TAIYO YUDEN JMK212BJ475MG † TAIYO YUDEN JMK316BJ106ML Figure 1a. High Efficiency Step-Down Converter 90 The LTC3406B-2 will operate with input supply voltages as low as 2.5V, but the maximum allowable output current is reduced at this low voltage. Figure 2 shows the reduction in the maximum output current as a function of input voltage for various output voltages. Slope Compensation and Inductor Peak Current Slope compensation provides stability in constant frequency architectures by preventing subharmonic oscillations at high duty cycles. It is accomplished internally by adding a compensating ramp to the inductor current signal at duty cycles in excess of 40%. Normally, this results in a reduction of maximum inductor peak current for duty cycles > 40%. However, the LTC3406B-2 uses a patent-pending scheme that counteracts this compensating ramp, which allows the maximum inductor peak current to remain unaffected throughout all duty cycles. 1200 VOUT = 1.8V TA = 25°C 1000 80 EFFICIENCY (%) VOUT 1.8V 600mA Low Supply Operation MAXIMUM OUTPUT CURRENT (mA) 100 † An important detail to remember is that at low input supply voltages, the RDS(ON) of the P-channel switch increases (see Typical Performance Characteristics). Therefore, the user should calculate the power dissipation when the LTC3406B-2 is used at 100% duty cycle with low input voltage (See Thermal Considerations in the Applications Information section). 70 60 VIN = 3.6V VIN = 2.7V 50 40 30 20 10 0.1 VIN = 4.2V 800 600 1000 3406B F01b Figure 1b. Efficiency vs Load Current VOUT = 2.5V VOUT = 1.5V 400 200 0 1 100 10 OUTPUT CURRENT (mA) VOUT = 1.8V 2.5 3.0 3.5 4.0 4.5 SUPPLY VOLTAGE (V) 5.0 5.5 3406B F02 Figure 2. Maximum Output Current vs Input Voltage sn3406b2 3406b2fs 7 LTC3406B-2 U W U U APPLICATIO S I FOR ATIO The basic LTC3406B-2 application circuit is shown in Figure 1. External component selection is driven by the load requirement and begins with the selection of L followed by CIN and COUT. Table 1. Representative Surface Mount Inductors Part Number Value (µH) DCR (ΩMAX) MAX DC Current (A) Size WxLxH (mm3) Sumida CDRH2D11 1.5 2.2 3.3 0.068 0.098 0.123 0.90 0.78 0.60 3.2 x 3.2 x 1.2 Sumida CDRH2D18/LD 2.2 3.3 4.7 0.041 0.054 0.078 0.85 0.75 0.63 3.2 x 3.2 x 2.0 Sumida CMD4D06 2.2 3.3 4.7 0.116 0.174 0.216 0.95 0.77 0.75 3.5 x 4.1 x 0.8 Murata LQH32C 1.0 2.2 4.7 0.060 0.097 0.150 1.00 0.79 0.65 2.5 x 3.2 x 2.0 Taiyo Yuden LQLBC2518 1.0 1.5 2.2 0.080 0.110 0.130 0.78 0.66 0.60 1.8 x 2.5 x 1.8 Toko D412F 2.2 3.3 4.7 0.14 0.20 0.22 1.14 0.90 0.80 4.6 x 4.6 x 1.2 Inductor Selection For most applications, the value of the inductor will fall in the range of 1µH to 4.7µH. Its value is chosen based on the desired ripple current. Large value inductors lower ripple current and small value inductors result in higher ripple currents. Higher VIN or VOUT also increases the ripple current as shown in equation 1. A reasonable starting point for setting ripple current is ∆IL = 240mA (40% of 600mA). ∆IL = ⎛ V ⎞ VOUT ⎜ 1 − OUT ⎟ VIN ⎠ ⎝ f L 1 ( )( ) (1) The DC current rating of the inductor should be at least equal to the maximum load current plus half the ripple current to prevent core saturation. Thus, a 720mA rated inductor should be enough for most applications (600mA + 120mA). For better efficiency, choose a low DC-resistance inductor. Inductor Core Selection Different core materials and shapes will change the size/ current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and don’t radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. The choice of which style inductor to use often depends more on the price vs size requirements and any radiated field/EMI requirements than on what the LTC3406B-2 requires to operate. Table 1 shows some typical surface mount inductors that work well in LTC3406B-2 applications. CIN and COUT Selection In continuous mode, the source current of the top MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: CIN required IRMS ≅ IOMAX 8 [ ( VOUT VIN − VOUT VIN 1/ 2 )] This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that the capacitor manufacturer’s ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Always consult the manufacturer if there is any question. The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. The output ripple ∆VOUT is determined by: ⎛ 1 ⎞ ∆VOUT ≅ ∆IL ⎜ ESR + ⎟ ⎝ 8 fCOUT ⎠ where f = operating frequency, COUT = output capacitance and ∆IL = ripple current in the inductor. For a fixed output voltage, the output ripple is highest at maximum input voltage since ∆IL increases with input voltage. sn3406b2 3406b2fs LTC3406B-2 U W U U APPLICATIO S I FOR ATIO Using Ceramic Input and Output Capacitors Higher values, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple current, high voltage rating and low ESR make them ideal for switching regulator applications. Because the LTC3406B-2’s control loop does not depend on the output capacitor’s ESR for stable operation, ceramic capacitors can be used freely to achieve very low output ripple and small circuit size. However, care must be taken when ceramic capacitors are used at the input and the output. When a ceramic capacitor is used at the input and the power is supplied by a wall adapter through long wires, a load step at the output can induce ringing at the input, VIN. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, a sudden inrush of current through the long wires can potentially cause a voltage spike at VIN, large enough to damage the part. When choosing the input and output ceramic capacitors, choose the X5R or X7R dielectric formulations. These dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size. Output Voltage Programming The output voltage is set by a resistive divider according to the following formula: ⎛ R2 ⎞ VOUT = 0.6⎜ 1 + ⎟ ⎝ R1⎠ (2) The external resistive divider is connected to the output, allowing remote voltage sensing as shown in Figure 3. 0.6V ≤ VOUT ≤ 5.5V R2 VFB LTC3406B-2 R1 GND 3406B F03 Figure 3. Setting the LTC3406B-2 Output Voltage Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, two main sources usually account for most of the losses in LTC3406B-2 circuits: VIN quiescent current and I2R losses. The VIN quiescent current loss dominates the efficiency loss at very low load currents whereas the I2R loss dominates the efficiency loss at medium to high load currents. In a typical efficiency plot, the efficiency curve at very low load currents can be misleading since the actual power lost is of no consequence as illustrated in Figure 4. 1 VIN = 3.6V 0.1 POWER LOSS (W) Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalum. These are specially constructed and tested for low ESR so they give the lowest ESR for a given volume. Other capacitor types include Sanyo POSCAP, Kemet T510 and T495 series, and Sprague 593D and 595D series. Consult the manufacturer for other specific recommendations. 0.01 VOUT = 2.5V VOUT = 1.8V VOUT = 1.2V 0.001 VOUT = 1.5V 0.0001 0.1 1 10 100 LOAD CURRENT (mA) 1000 3406B F04 Figure 4. Power Lost vs Load Current sn3406b2 3406b2fs 9 LTC3406B-2 U W U U APPLICATIO S I FOR ATIO 1. The VIN quiescent current is due to two components: the DC bias current as given in the electrical characteristics and the internal main switch and synchronous switch gate charge currents. The gate charge current results from switching the gate capacitance of the internal power MOSFET switches. Each time the gate is switched from high to low to high again, a packet of charge, dQ, moves from VIN to ground. The resulting dQ/dt is the current out of VIN that is typically larger than the DC bias current. In continuous mode, IGATECHG = f(QT + QB) where QT and QB are the gate charges of the internal top and bottom switches. Both the DC bias and gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages. 2. I2R losses are calculated from the resistances of the internal switches, RSW, and external inductor RL. In continuous mode, the average output current flowing through inductor L is “chopped” between the main switch and the synchronous switch. Thus, the series resistance looking into the SW pin is a function of both top and bottom MOSFET RDS(ON) and the duty cycle (DC) as follows: RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 – DC) The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Charateristics curves. Thus, to obtain I2R losses, simply add RSW to RL and multiply the result by the square of the average output current. Other losses including CIN and COUT ESR dissipative losses and inductor core losses generally account for less than 2% total additional loss. Thermal Considerations In most applications the LTC3406B-2 does not dissipate much heat due to its high efficiency. But, in applications where the LTC3406B-2 is running at high ambient temperature with low supply voltage and high duty cycles, such as in dropout, the heat dissipated may exceed the maximum junction temperature of the part. If the junction temperature reaches approximately 150°C, both power switches will be turned off and the SW node will become high impedance. To avoid the LTC3406B-2 from exceeding the maximum junction temperature, the user will need to do some thermal analysis. The goal of the thermal analysis is to determine whether the power dissipated exceeds the maximum junction temperature of the part. The temperature rise is given by: TR = (PD)(θJA) where PD is the power dissipated by the regulator and θJA is the thermal resistance from the junction of the die to the ambient temperature. The junction temperature, TJ, is given by: TJ = TA + TR where TA is the ambient temperature. As an example, consider the LTC3406B-2 in dropout at an input voltage of 2.7V, a load current of 600mA and an ambient temperature of 70°C. From the typical performance graph of switch resistance, the RDS(ON) of the P-channel switch at 70°C is approximately 0.52Ω. Therefore, power dissipated by the part is: PD = ILOAD2 • RDS(ON) = 187.2mW For the SOT-23 package, the θJA is 250°C/ W. Thus, the junction temperature of the regulator is: TJ = 70°C + (0.1872)(250) = 116.8°C which is below the maximum junction temperature of 125°C. Note that at higher supply voltages, the junction temperature is lower due to reduced switch resistance (RDS(ON)). Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to (∆ILOAD • ESR), where ESR is the effective series resistance of COUT. ∆ILOAD also begins to charge or discharge COUT, which generates a feedback error signal. The regulator loop then acts to return VOUT to its steadystate value. During this recovery time VOUT can be monitored for overshoot or ringing that would indicate a stability problem. For a detailed explanation of switching control loop theory, see Application Note 76. sn3406b2 3406b2fs 10 LTC3406B-2 U W U U APPLICATIO S I FOR ATIO A second, more severe transient is caused by switching in loads with large (>1µF) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the load switch resistance is low and it is driven quickly. The only solution is to limit the rise time of the switch drive so that the load rise time is limited to approximately (25 • CLOAD). Thus, a 10µF capacitor charging to 3.3V would require a 250µs rise time, limiting the charging current to about 130mA. PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the 1 RUN LTC3406B-2. These items are also illustrated graphically in Figures 5 and 6. Check the following in your layout: 1. The power traces, consisting of the GND trace, the SW trace and the VIN trace should be kept short, direct and wide. 2. Does the VFB pin connect directly to the feedback resistors? The resistive divider R1/R2 must be connected between the (+) plate of COUT and ground. 3. Does the (+) plate of CIN connect to VIN as closely as possible? This capacitor provides the AC current to the internal power MOSFETs. 4. Keep the switching node, SW, away from the sensitive VFB node. 5. Keep the (–) plates of CIN and COUT as close as possible. VFB 5 LTC3406B-2 2 – R2 R1 GND COUT VOUT 3 + L1 SW VIN CFWD 4 CIN + VIN – 3406B F05 BOLD LINES INDICATE HIGH CURRENT PATHS Figure 5. LTC3406B-2 Layout Diagram VIA TO GND R1 VIN VIA TO VIN CFWD LTC3406B-2 VOUT L1 VIA TO VOUT R2 PIN 1 SW COUT CIN GND 3406B F06 Figure 6. LTC3406B-2 Suggested Layout sn3406b2 3406b2fs 11 LTC3406B-2 U W U U APPLICATIO S I FOR ATIO Design Example A 2.2µH inductor works well for this application. For best efficiency choose a 720mA or greater inductor with less than 0.2Ω series resistance. CIN will require an RMS current rating of at least 0.3A ≅ ILOAD(MAX)/2 at temperature and COUT will require an ESR of less than 0.25Ω. In most cases, a ceramic capacitor will satisfy this requirement. As a design example, assume the LTC3406B-2 is used in a single lithium-ion battery-powered cellular phone application. The VIN will be operating from a maximum of 4.2V down to about 2.7V. The load current requirement is a maximum of 0.6A but most of the time it will be in standby mode, requiring only 2mA. Efficiency at both low and high load currents is important. Output voltage is 2.5V. With this information we can calculate L using equation (1), ⎛ V ⎞ L= VOUT ⎜ 1 − OUT ⎟ VIN ⎠ ⎝ f ∆IL For the feedback resistors, choose R1 = 316k. R2 can then be calculated from equation (2) to be: ⎛V ⎞ R2 = ⎜ OUT − 1⎟ R1 = 1000k ⎝ 0.6 ⎠ 1 ( )( ) (3) Figure 7 shows the complete circuit along with its efficiency curve. Substituting VOUT = 2.5V, VIN = 4.2V, ∆IL = 240mA and f = 2.25MHz in equation (3) gives: L= ⎛ 2.5V ⎞ 2.5V ⎜1 − ⎟ = 1.87µH 1.5MHz(240mA) ⎝ 4.2V ⎠ VIN 2.7V TO 4.2V 4 CIN† 4.7µF CER VIN SW 2.2µH* 3 22pF COUT** 10µF CER LTC3406B-2 1 VFB RUN VOUT 2.5V 5 1M GND 316k 2 3406B F07a *MURATA LQH32CN2R2M33 ** TAIYO YUDEN JHK316BJ106ML † TAIYO YUDEN JMK212BJ475MG Figure 7a 100 90 VOUT = 2.5V TA = 25°C EFFICIENCY (%) 80 VIN = 2.7V VIN = 4.2V 70 60 50 VIN = 3.6V 40 30 20 10 0.1 1 100 10 OUTPUT CURRENT (mA) 1000 3406B G04 Figure 7b sn3406b2 3406b2fs 12 LTC3406B-2 U TYPICAL APPLICATIO S Single Li-Ion 1.2V/600mA Regulator for High Efficiency and Small Footprint VIN 2.7V TO 4.2V 4 CIN† 4.7µF CER VIN SW 3 2.2µH* 22pF VFB RUN GND 2 VOUT 1.2V COUT** 10µF CER LTC3406B-2 1 Load Step 5 301k VOUT 100mV/DIV AC COUPLED IL 500mA/DIV 301k 3406B TA09 *MURATA LQH32CN2R2M33 ** TAIYO YUDEN JHK316BJ106ML † TAIYO YUDEN JMK212BJ475MG ILOAD 500mA/DIV VIN = 3.6V 20µs/DIV VOUT = 1.2V ILOAD = 0mA TO 600mA Efficiency vs Output Current 100 90 Load Step VOUT = 1.2V TA = 25°C VOUT 100mV/DIV AC COUPLED EFFICIENCY (%) 80 70 IL 500mA/DIV VIN = 3.6V 60 50 40 3406B TA11 VIN = 2.7V ILOAD 500mA/DIV VIN = 4.2V 30 VIN = 3.6V 20µs/DIV VOUT = 1.2V ILOAD = 100mA TO 600mA 20 10 0.1 1 100 10 OUTPUT CURRENT (mA) 3406B TA12 1000 3406B G02 sn3406b2 3406b2fs 13 LTC3406B-2 U TYPICAL APPLICATIO S 5V Input to 3.3V/0.6A Regulator 4 VIN 5V † CIN 4.7µF CER VIN SW 3 2.2µH* VOUT 3.3V 22pF COUT** 10µF CER LTC3406B-2 1 RUN VFB GND 2 Load Step 5 VOUT 100mV/DIV AC COUPLED IL 500mA/DIV 1M 221k ILOAD 500mA/DIV 3406B TA13 *MURATA LQH32CN2R2M33 ** TAIYO YUDEN JHK316BJ106ML † TAIYO YUDEN JMK212BJ475MG VIN = 3.6V 20µs/DIV VOUT = 3.3V ILOAD = 0mA TO 600mA Efficiency vs Output Current 100 90 Load Step VOUT = 3.3V VIN = 5V VOUT 100mV/DIV AC COUPLED 80 EFFICIENCY (%) 3406B TA15 70 IL 500mA/DIV 60 50 ILOAD 500mA/DIV 40 30 VIN = 3.6V 20µs/DIV VOUT = 3.3V ILOAD = 100mA TO 600mA 20 10 0.1 1 100 10 OUTPUT CURRENT (mA) 3406B TA16 1000 3406B TA14 sn3406b2 3406b2fs 14 LTC3406B-2 U PACKAGE DESCRIPTIO S5 Package 5-Lead Plastic TSOT-23 (Reference LTC DWG # 05-08-1635) 0.62 MAX 0.95 REF 2.90 BSC (NOTE 4) 1.22 REF 1.4 MIN 3.85 MAX 2.62 REF 2.80 BSC 1.50 – 1.75 (NOTE 4) PIN ONE RECOMMENDED SOLDER PAD LAYOUT PER IPC CALCULATOR 0.30 – 0.45 TYP 5 PLCS (NOTE 3) 0.95 BSC 0.80 – 0.90 0.20 BSC 0.01 – 0.10 1.00 MAX DATUM ‘A’ 0.30 – 0.50 REF 0.09 – 0.20 (NOTE 3) NOTE: 1. DIMENSIONS ARE IN MILLIMETERS 2. DRAWING NOT TO SCALE 3. DIMENSIONS ARE INCLUSIVE OF PLATING 4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 5. MOLD FLASH SHALL NOT EXCEED 0.254mm 6. JEDEC PACKAGE REFERENCE IS MO-193 1.90 BSC S5 TSOT-23 0302 sn3406b2 3406b2fs Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 15 LTC3406B-2 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1616 500mA (IOUT), 1.4MHz, High Efficiency Step-Down DC/DC Converter 90% Efficiency, VIN: 3.6V to 25V, VOUT(MIN) = 1.25V, IQ = 1.9mA, ISD < 1µA, ThinSOT Package LT1676 450mA (IOUT), 100kHz, High Efficiency Step-Down DC/DC Converter 90% Efficiency, VIN: 7.4V to 60V, VOUT(MIN) = 1.24V, IQ = 3.2mA, ISD = 2.5µA, S8 Package LTC1877 600mA (IOUT), 550kHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.7V to 10V, VOUT(MIN) = 0.8V, IQ = 10µA, ISD < 1µA, MS8 Package LTC1878 600mA (IOUT), 550kHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.7V to 6V, VOUT(MIN) = 0.8V, IQ = 10µA, ISD < 1µA, MS8 Package LTC1879 1.2A (IOUT), 550kHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.7V to 10V, VOUT(MIN) = 0.8V, IQ = 15µA, ISD < 1µA, TSSOP-16 Package LTC3403 600mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter with Bypass Transistor 96% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = Dynamically Adjustable, IQ = 20µA, ISD < 1µA, DFN Package LTC3404 600mA (IOUT), 1.4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.7V to 6V, VOUT(MIN) = 0.8V, IQ = 10µA, ISD < 1µA, MS8 Package LTC3405/LTC3405A 300mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter 96% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 20µA, ISD < 1µA, ThinSOT Package LTC3406/LTC3406B 600mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter 96% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 20µA, ISD < 1µA, ThinSOT Package LTC3407 Dual Output (600mA × 2) 1.5MHz Synchronous Step-Down DC/DC Converter 95% Efficiency VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40µA, MS10E Package LTC3408 600mA (IOUT), 1.5MHz Synchronous Step-Down DC/DC Converter with 0.08Ω Bypass Transistor 96% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = Dynamically Adjustable, IQ = 1.5mA, ISD < 1µA, DFN Package LTC3411 1.25A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 60µA, ISD < 1µA, MS Package LTC3412 2.5A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 60µA, ISD < 1µA, TSSOP-16E Package LTC3414 4A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.25V to 5.5V, VOUT(MIN) = 0.8V, IQ = 64µA, TSSOP-20E Package LTC3440 600mA (IOUT), 2MHz, Synchronous Buck-Boost DC/DC Converter 95% Efficiency, VIN: 2.5V to 5.5V, VOUT: 2.5V to 5.5V, IQ = 25µA, ISD < 1µA, MS Package LTC3441 1A (IOUT), 1MHz, Synchronous Buck-Boost DC/DC Converter 95% Efficiency, VIN: 2.5V to 5.5V, VOUT: 2.4V to 5.25V, IQ = 25µA, DFN Package sn3406b2 3406b2fs 16 Linear Technology Corporation LT/TP 0204 1K • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2003