FAIRCHILD RV4391N

PRODUCT SPECIFICATION
RC4391
Pin Descriptions
Pin Assignments
LBR
1
8
VFB
Pin
Number
Pin Function Description
LBD
2
7
VREF
1
Low Battery Resistor (LBR)
CX
3
6
+VS
2
Low Battery Detector (LBD)
3
Timing Capacitor (CX)
4
Ground
5
External Inductor (LX)
6
+Supply Voltage (+VS)
GND
4
LX
5
65-3471-02
7
+1.25V Reference Voltage (VREF)
8
Feedback Voltage (VFB)
Absolute Maximum Ratings
Parameter
Conditions
Min
Typ
Internal Power Dissipation
Max
Unit
500
mW
+30
V
0
70
°C
RV4391
-25
85
°C
RM4391
-55
125
°C
-65
Supply Voltage1
(Pin 6 to Pin 4 or Pin 6 to Pin 5)
Operating Temperature
RC4391
150
°C
PDIP, SOIC
125
°C
CerDIP
175
°C
Switch Current (IMAX)
Peak
375
mA
PD TA <50˚C
PDIP
468
mW
CerDIP
833
mW
SOIC
300
mW
(10 seconds)
300
°C
Storage Temperature
Junction Temperature
Lead Soldering Temperature
Note:
1. The maximum allowable supply voltage (+VS) in inverting applications will be reduced by the value of the negative output
voltage, unless an external power transistor is used in place of Q1.
Thermal Characteristics
Therm. Res qJC
Therm. Res. qJA
For TA >50˚C Derate at
2
8-Lead Plastic DIP
8-Lead Ceramic DIP
Small Outline SO-8
—
45°C/W
—
160°C/W
150°C/W
240°C/W
6.25 mW/°C
8.33 mW/°C
4.17 mW/°C
RC4391
PRODUCT SPECIFICATION
Electrical Characteristics
(VS = +6.0V, over the full operating temperature range unless otherwise noted)
Symbol
Parameters
Condition
Min
+VS
Supply Voltage
(Note 1)
4.0
ISY
Supply Current
VS = +25V
VREF
Reference Voltage
VOUT
Output Voltage
LI1
Line Regulation
Typ
Max
Units
30
V
300
500
mA
1.13
1.25
1.36
V
VOUT nom = -5.0V
-5.5
-5.0
-4.5
V
VOUT nom = -15V
-16.5
-15.0
-13.5
VOUT nom = -5.0V,
%VOUT
CX = 150pF
VS = +5.8V to +15V
2.0
4.0
1.5
3.0
VOUT nom = -15V,
CX = 150pF
VS = +5.8V to +15
L01
Load Regulation
VOUT nom = -5.0V,
%VOUT
CX = 350pF, VS = +4.5V,
PLOAD = 0mW to 75mW
0.2
0.5
PLOAD = 0mW to 75mW
0.2
0.3
Pin 5 = -20V
0.1
30
VOUT nom = -15V,
CX = 350pF, VS = +4.5V,
ICO
Switch Leakage Current
mA
Note:
1. The maximum allowable supply voltage (+VS) in inverting applications will be reduced by the value of the negative output
voltage, unless an external power transistor is used.
3
PRODUCT SPECIFICATION
RC4391
Electrical Characteristics
(VS = +6.0V, TA = +25°C unless otherwise noted)
Symbol
Parameters
Condition
ISY
Supply Voltage
VS = +4.0V,
Min
Typ
Max
Units
170
250
mA
300
500
No External Loads
VS = +25V
No External Loads
VOUT
LI1
Output Voltage
Line Regulation
VOUT nom = -5.0V
-5.35
-5.0
-4.65
VOUT nom = -15V
-15.85
-15.0
-14.15
VOUT nom = -5.0V
V
%VOUT
CX = 150pF,
1.5
3.0
1.0
2.0
VS = +5.8V to +15V
VOUT nom = -15V,
CX = 150pF
VS = +5.8V to +15V
L01
Load Regulation
VOUT nom = -5.0V,
%VOUT
CX = 350pF, VS = +4.5V,
0.2
0.4
0.07
0.14
1.25
1.32
PLOAD = 0mW to 75mW
VOUT nom = -15V,
CX = 350pF, VS = +4.5V,
PLOAD = 0mW to 75mW
4
VREF
Reference Voltage
1.18
ISW
Switch Current
Pin 5 = 5.5V
ICO
Switch Leakage Current
Pin 5 = -24V
ICX
Cap. Charging Current
Pin 3 = 0V
ILBDL
LBD Leakage Current
Pin 1 = 1.5V, Pin 2 = 6.0V
ILBD0
LBD On Current
Pin 1 = 1.1V, Pin 2 = 0.4V
ILBRB
LBR Bias Current
Pin 1 = 1.5V
75
6.0
210
100
V
mA
0.01
5.0
mA
10
14
mA
0.01
5.0
mA
600
mA
0.7
mA
RC4391
PRODUCT SPECIFICATION
6.5
8
6.0
7
5.5
6
FO (kHz)
5.0
4.5
4.0
5
4
3
2
65-3268
3.5
3.0
0
5
10
15
20
65-3272
FO (kHz)
Typical Performance Characteristics
1
0
25
-55
0
70
125
TA (¡C)
+VS (V)
Figure 1. Oscillator Frequency vs. Supply Voltage
Figure 2. Oscillator Frequency vs. Temperature
1.260
1.260
1.255
1.255
VREF (V)
1.250
1.250
1.245
65-3269
1.245
1.240
-55
0
25
70
125
65-3273
VREF (V)
25
1.240
4
6
TA (¡C)
10
20
30
+VS (V)
Figure 3. Reference Voltage vs. Temperature
Figure 4. Reference Voltage vs. Supply Voltage
4
600
500
3
+VS (V)
300
2
200
0
1
2
3
4
5
6
7
8
VCE (SAT) (V)
Figure 5. Collector Current vs. Q1 Saturation Voltage
0
65-3271
1
100
20
10
65-3270
IC (mA)
400
-55
0
25
70
125
TA (¡C)
Figure 6. Minimum Supply Voltage vs. Temperature
5
PRODUCT SPECIFICATION
RC4391
Principles of Operation
The basic switching inverter circuit is the building block on
which the complete inverting application is based.
A simplified diagram of the voltage inverter circuit with
ideal components and no feedback circuitry is shown in
Figure 7. When the switch S is closed, charging current from
the battery flows through the inductor L, which builds up a
magnetic field, increasing as the switch is held closed. When
the switch is opened, the magnetic field collapses, and the
energy stored in the magnetic field is converted into a current
which flows through the inductor in the same direction as the
changing current. Because there is no path for this current to
flow through the switch, the current must flow through the
diode to charge the capacitor C. The key to the inversion is
the ability of the inductor to become a source when the
charging current is removed.
reference. Because CF is initially discharged a positive
voltage is applied to the comparator, and the output of the
comparator gates the squarewave oscillator. This gated
squarewave signal turns on, then off, the PNP output transistor. This turning on and off of the output transistor performs
the same function as opening and closing the ideal switch in
the simplified diagram; i.e., it stores energy in the inductor
during the on time and releases it into the capacitor during
the off time.
The comparator will continue to allow the oscillator to turn
the switch transistor on and off until enough energy has been
stored in the output capacitor to make the comparator input
voltage decrease to less than 0V. The voltage applied to the
comparator is set by the output voltage, the reference voltage, and the ratio of R1 to R2.
The equation V = L (di/dt) gives the maximum possible
voltage across the inductor; in the actual application, feedback circuitry and the output capacitor will decrease the
output voltage to a regulated fixed value.
S
+VS
L
A complete schematic for the standard inverting application
is shown in Figure 8. The ideal switch in the simplified
diagram is replaced by the PNP transistor switch between
pins 5 and 6. CF functions as the output filter capacitor, and
D1 and LX replace D and L.
65-1601
Figure 7. Simple Inverting Regulator
C F*
33µF
R3
260K
-V OUT
LBR
R6
100K
Parts
List
R1 =
R1
C2
R4
590K
Q2
C1
VFB
R2 =
Cx =
Lx =
R2
VREF
+1.25V
REF/Bias
LBD
VREF
C1
0.1µF
LBD
Output
D1
+VS
B
OSC
900 k W
75 k W
300 k W
75 k W
150 pF
150 pF
1.0 mH Dale TE3 Q4 TA
= Optional
+Vs
-VOUT = (1.25V) (
A
Cx
-15V
Output
-5.0V
Output
F
1N914
CX
VOUT
C
(+)
When power is first applied, the ground sensing comparator
(pin 8) compares the output voltage to the +1.25V voltage
To
+Vs
(–)
D
R1
)
R2
D
Q1
GND
LX
RC4391
Lx
E
65-1602
*Caution: Use current limiting protection circuit for high values of CF (Figure 13)
Figure 8. Inverting Regulator – Standard Circuit
6
RC4391
PRODUCT SPECIFICATION
1.78V
CX
A
0.62V
O SC
(Internal)
B
IL
I LOAD
0 mA
C
+V S
(Internal)
D
+V S - 0.7V
VBAT
LX
V OUT
LX
V BEQ1
Max
I LX
0 mA
I MAX
E
ID
0 mA
F
+V S - V SW
V LX
Ground
G
-V OUT - V D
65-2472
Figure 9. Inverting Regulator Waveforms
This feedback system will vary the duration of the on time in
response to changes in load current or battery voltage (see
Figure 9). If the load current increases (waveform C), then
the transistor will remain on (waveform D) for a longer portion of the oscillator cycle, (waveform B) to build up to a
higher peak value. The duty cycle of the switch transistor
varies in response to changes in load and line.
Step-Down Regulator
The step-down circuit function is similar to inversion; it uses
the same components (switch, inductor, diode, filter capacitor), and charges and discharges the inductor by closing and
opening the switch. The great difference is that the inductor
is in series with the load; therefore, both the charging current
and the discharge current flow into the load. In the inverting
circuit only the discharge current flows into the load. Refer
to Figure 10.
When the switch S is closed, current flows from the battery,
through the inductor, and through the load resistor to ground.
After the switch is opened, stored energy in the inductor
causes current to keep flowing through the load, the circuit
being completed by the catch diode D. Since current flows to
the load during charge and discharge, the average load cur-
L
S
+VS
D
(+)
C
RL
VOUT
(-)
65-2473
Figure 10. Simple Step-Down Regulator
rent will be greater than in an inverting circuit. The significance of that is that for equal load currents the step-down
circuit will require less peak inductor current than an inverting circuit. Therefore, the inductor will not require as large
of a core, and the switch transistor will not be stressed as
heavily for equal load currents.
Figure 11 depicts a complete schematic for a step-down circuit using the RC4391. Observe that the ground lead of the
4391 is not connected to circuit ground; instead, it is tied to
the output voltage. It is by this rearrangement that the feedback system, which senses voltages more negative than the
ground lead, can be used to regulate a non-negative output
voltage.
7
PRODUCT SPECIFICATION
RC4391
When power is first applied, the output filter capacitor is discharged so the ground lead potential starts at 0V. The reference voltage is forced to +1.25V above the ground lead and
pulls the feedback input (pin 8) more positive than the
ground lead. This positive voltage forces the control network
to begin pulsing the switch transistor. As the switching
action pumps up the output voltage, the ground lead rises
with the output until the voltage on the ground lead is equal
to the feedback voltage. At that point, the control network
reduces the time on time of the switch to maintain a constant
output.
This control network will vary the on time of the switch in
response to changes in load current or battery voltage (see
Figure 12). If the load current increases (waveform C), then
the transistor will remain on (waveformD) for a longer portion of the oscillator cycle, (waveform B), thus allowing the
inductor current (waveform E) to build up to a higher peak
value. The duty cycle of the switch transistor varies in
response to changes in load and line.
Design Equations
The inductor value and timing capacitor (CX) value must be
carefully tailored to the input voltage, input voltage range,
output voltage, and load current requirements of the application. The key to the problem is to select the correct inductor
value for a given oscillator frequency, such that the inductor
current rises to a high enough peak value (IMAX) to meet the
average load current drain. The selection of this inductor
value must take into account the variation of oscillator
frequency from unit to unit and the drift of frequency over
temperature. Use ±30% as a maximum variation of oscillator
frequency.
R1
LBR
VFB
C2
Q2
C1
R2
LBD
CX
VREF
+1.25V
REF/Bias VREF
+VS
B
OSC
+Vs
A
Cx
D
Q1
GND
LX
+V OUT
RC4391
E
Lx
CF
+VOUT
R1
= (1.25V) (
)
R2
F
D1
1N914
65-2475
Important Note: This circuit must have a minimum load ³ 1 mA always connected.
Figure 11. Step-Down Regulator – Standard Circuit
8
RC4391
PRODUCT SPECIFICATION
1.78V
A
CX
0.62V
(Internal)
B
OSC
IL
I LOAD
C
0 mA
+VS
D
(Internal)
+V - 0.7V
VBAT
LX
V OUT - V BAT
LX
VBEQ1
I MAX
I LX
E
0 mA
+V S - VSW
VOUT
F
VLX
VS ( -0.7V)
65-2474
Figure 12. Step-Down Regulator Waveforms
The oscillator creates a squarewave using a method similar
to the 555 timer IC, with a current steering flip-flop controlled by two voltage sensing comparators. The oscillator
frequency is set by the timing capacitor (CX) according to
the following equation.
–6
4.1x10
F O ( Hz ) = ----------------------C ( pF )
x
The squarewave output of the oscillator is internal and
cannot be directly measured, but is equal in frequency to the
triangle waveform measurable at pin 3. The switch transistor
is normally on when the triangle waveform is ramping up
and off when ramping down. Capacitor selection depends on
the application; higher operating frequencies will reduce the
output voltage ripple and will allow the use of an inductor
with a physically smaller inductor core, but excessively
high frequencies will reduce load driving capability and
efficiency.
Inverting Design Procedure
1. Select an operating frequency and timing capacitor value
as shown above (frequencies from 10kHz to 50kHz are
typical).
2. Find the maximum on time TON (add 3mS for the turn off
base recombination delay of Q1):
1
T ON = ---------- + 3mS
2F O
3. Calculate the peak inductor current IMAX (if this value
is greater than 375mA then an external power transistor
must be used in place of Q1):
( V OUT + V D )2I L
I MAX = --------------------------------------------------------( F O ) ( T ON ) ( V S – V SW )
Where:
VS = Supply Voltage
VSW = Saturation Voltage of Q1 (typically 0.5V)
VD = Diode Forward Voltage (typically 0.7V)
IL = DC Load Current
4. Find an inductance value for LX:
V S – V SW
L X ( Henries ) = æ -------------------------ö ( T ON )
è I MAX ø
The inductor chosen must exhibit this value of inductance
and have a current rating equal to IMAX.
9
PRODUCT SPECIFICATION
RC4391
Step-Down Design Procedure
Compensation
1. Select an operating frequency.
When large values (> 50 kW) are used for the voltage setting
resistors (R1 and R2 of Figure 8) stray capacitance at the
VFB input can add lag to the feedback response, destabilizing the regulator, increasing low frequency ripple, and lowering efficiency. This can often be avoided by minimizing the
stray capacitance at the VFB node. It can also be remedied by
adding a lead compensation capacitor of 100 pF to 10 nF. In
inverting applications, the capacitor connects between
-VOUT and VFB; for step-down circuits it connects between
ground and VFB. Most applications do not require this
capacitor.
2. Determine the maximum on time TON as in the inverting
design procedure.
3. Calculate IMAX:
2I L
I MAX = ---------------------------------------------------------------------------( V S – V OUT )
( F O ) ( T ON ) --------------------------------- + 1
( V OUT – V D )
4. Calculate LX:
V S – V SW
L X (Henries) = æ -------------------------ö ( T ON )
è I MAX ø
Alternate Design Procedure
The design equations above will not work for certain input/
output voltage ratios, and for these circuits another method
of defining component values must be used. If the slope of
the current discharge waveform is much less than the slope
of the current charging waveform, then the inductor current
will become continuous (never discharging completely), and
the equations will become extremely complex. So, if the
voltage applied across the inductor during the charge time is
greater than during the discharge time, use the design procedure below. For example, a step-down circuit with 20V input
and 5V output will have approximately 15V across the
inductor when charging, and approximately 5V when discharging. So in this example the inductor current will be continuous and the alternate procedure will be necessary. The
alternate procedure may also be used for discontinuous circuits.
1. Select an operating frequency based on efficiency and
component size requirements (a value between 10kHz
and 50kHz is typical).
2. Build the circuit and apply the worst case conditions to
it, i.e., the lowest battery voltage and the highest load
current at the desired output voltage.
3. Adjust the inductor value down until the desired output
voltage is achieved, then decrease its value by 30% to
cover manufacturing tolerances.
4. Check the output voltage with an oscilloscope for ripple,
at high supply voltages, at voltages as high as are
expected. Also check for efficiency by monitoring supply
and output voltages and currents:
( V OUT ) ( I OUT ) ö
æ eff = ----------------------------------------è
( +V S ) ( I SY )x100ø
5. If the efficiency is poor, go back to Step 1 and start over.
If the ripple is excessive, then increase the output filter
capacitor value or start over.
Inductors
Efficiency and load regulation will improve if a quality high
Q inductor is used. A ferrite pot core is recommended; the
wind-yourself type with an air gap adjustable by washers or
spacers is very useful for bread-boarding prototypes. Care
must be taken to choose a core with enough permeability to
handle the magnetic flux produced at IMAX. If the core saturates, then efficiency and output current capability are
severely degraded and excessive current will flow through
the switch transistor. A pot core inductor design section is
provided later in this datasheet.
An isolated AC current probe for an oscilloscope (example:
Tektronix P6042) is an excellent tool for saturation problems; with it the inductor current can be monitored for nonlinearity at the peaks (a sign of saturation).
Low Battery Detector
An open collector signal transistor Q2 with comparator C2
provides the designer with a method of signaling a display or
computer whenever the battery voltage falls below a programmed level (see Figure 13). This level is determined by
the +1.25V reference level and by the selection of two external resistors according to the equation:
R4
V TH = V REF æ ------- + 1ö
è R5
ø
When the battery drops below this threshold Q2 will turn on
and sink typically 600mA. The low battery detection circuit
can also be used for other less conventional applications such
as the voltage dependent oscillator circuit of Figure 18.
+Vs
R4
LBD
1 LBR
C2
2
Q2
R5
V REF
1.25V
65-1651A
Figure 13. Low Battery Detector
10
I LBD
RC4391
PRODUCT SPECIFICATION
Device Shutdown
The entire device may be shut down to an extremely low current non-operating condition by disconnecting the ground
(pin 4). This can be easily done by putting an NPN transistor
in series with ground pin and switching it with an external
signal. This switch will not affect the efficiency of operation,
but will add to and increase the reference voltage by an
amount equal to the saturation voltage of the transistor used.
A mechanical switch can also be used in series between
circuit ground and pin 4, without introducing any reference
offset.
Power Transistor Interfaces
The most important consideration in selecting an external
power transistor is the saturation voltage at IC = IMAX.
The lower the saturation voltage is, the better the efficiency
will be. Also, a higher beta transistor requires less base drive
and therefore less power will be.
Also, a higher beta transistor requires less base drive and
therefore less power will be consumed in driving it, improving efficiency losses in the interface. The part numbers given
in the following applications are recommended, but other
types may be more appropriate depending on voltage and
power levels.
When troubleshooting external power transistor circuits,
ensure that clean, sharp-edged waveforms are driving the
interface and power transistors. Monitor these waveforms
with an oscilloscop—disconnect the inductor, and tie the
VFB input (pin 8) high through a 10K resistor. This will
cause the regulator to pulse at maximum duty cycle without
drawing excessive inductor currents. Check for expected on
time and off time, and look for slow rise times that might
cause the power transistor to enter its linear operating region.
The following external power transistor circuits may demand
some adjustment to resistor values to satisfy various power
levels and input/output voltages. CX and LX values must be
selected according to the design equations (pages 2-213 and
2-214).
Inverting Medium Power Application
Figure 8 is a schematic of an inverting medium power supply
(250mW to 1W) using an external PNP switch transistor.
Supply voltage is applied to the IC via R3: when the internal
switch transistor is turned on current through R4 is also
drawn through R3; creating a voltage drop from base to
emitter of the external switch transistor. This drop turns on
the external transistor.
Voltage pulses on the supply lead (pin 6) do not affect circuit
operation because the internal reference and bias circuitry
have good supply rejection capabilities. A power Schottky
diode is used for higher efficiency.
Inverting High Power Application
For higher power applications (500mW to 5W), refer to
Figure 9. This circuit uses an extra external transistor to provide well controlled drive current in the correct phase to the
power switch transistor. The value of R3 sets the drive
current to the switch by making the interface transistor act as
a current source. R4 and R5 must be selected such that the
RC time constant of R4 and the base capacitance of Q2 do
not slow the response time (and affect duty cycle), but not so
low in value that excess power is consumed and efficiency
suffers. The resistor values chosen should be proportional to
the supply voltage (values shown are for +5V).
Step-Down Power Applications
Figures 16 and 17 show medium and high power interfaces
modified to perform step-down functioning. The design
+5V
C1
0.1µF
R3
1k½
R2
62 k½
Q1
2N3635
-24V
7
6
5
VFB VREF +Vs
Lx
5
4391
Cx
R4
50½
CF
100µF
220µH
GND
4
3
150 pF
Motorola
MBR030
R1
1.2 M½
Cx
65-2476
Figure 14. Inverting Medium Power Application
11
PRODUCT SPECIFICATION
RC4391
+Vs
R6
1K
C1
0.1µF
R5
2K
R2
Q1
TIP116
Q2
2N33904
MBR140P
5
VFB
7
6
5
VREF +Vs L x
4391
Cx
-V OUT
R4
4.7K
R3
750½
R1
GND
4
3
CF
Lx
Cx
65-2478
Figure 9. Inverting High Power Application
equations and suggestions for the circuits of Figures 14 and
15 also apply to these circuits. For a certain range of load
power, the RC4193 can be used for step-sown applications.
A load range from 400mW to 2W can be sustained with
fewer components (especially when stepping down greater
than 30V) than the comparable RC4391 circuit. Refer to
Fairchild Semiconductor's RC4191/4192/4193 data sheet for
a schematic of this medium power step-down application.
Voltage Dependent Oscillator
The RC4391's ability to supply load current at low battery
voltages depends on the inductor value and the oscillator frequency. Low values of inductance or a low oscillator frequency will cause a higher peak inductor current and
therefore increase the load current capability. A large inductor current is not necessarily best , however, because the
large amount of energy delivered with each cycle will cause
a large voltage ripple at the output, especially at high input
voltages. This trade-off between load current capability and
output ripple can be improved with the circuit connection
shown in Figure 18. This circuit uses the low battery detector
to sense for a low battery voltage condition and will
decrease the oscillator frequency after a pre- programmed
threshold is reached.
12
The threshold is programmed exactly as the normal low battery detector connection:
R4
V TH = V REF æ ------- + 1ö
è R5
ø
When the battery voltage reaches this threshold the comparator will turn on the open collector transistor at pin 2, effectively pulling CY in parallel with CX. This added
capacitance will reduce the oscillator frequency, according to
the following equation:
–6
4.1x10
F O ( Hz ) = ------------------------------------------------C X ( pF ) + C Y ( pF )
Current Limiting
The oscillator (CX) pin can be used to add short circuit protection and to protect against over current at start-up (when
using large values for the output filter capacitor —greater
than 100 mF). A transistor VBE is used as a current sensing
comparator which resets the oscillator upon sensing an over
current condition, thus providing cycle-by-cycle current limiting. Figure 19 shows how this is applied.
RC4391
PRODUCT SPECIFICATION
+Vs
C1
0.1 µF
R3
1K
R2
2N3635
8
VFB
R1
7
6
VREF +Vs
5
Lx
MBR030
R4
30 - 100½
4391
Cx
Lx
GND
4
3
Cx
+V OUT
CF
Note: A minimum load ³1mA must be connected.
65-2479
Figure 16. Step-Down Medium Power Application
TIP116
*
MBR140P
500½
6
+1.3V
VBAT
7
+VS
VREF
R2
5K
8 V
FB
R1
5K
470 pF
LX
4391
CX
5
2N3904
GND
3
CX
R3
1K
R4
20K
4
470 pF
250µH
(+)
CF
470µF
V OUT
(+5V at 1A as shown)
(-)
Note: A minimum load ³1mA must be connected.
*Optional — Extends supply voltage range.
65-2077
Figure 17. Step-Down High Power Application
To
+VS
OSC
CX
+VS
3
+1.25V
LBD
2
CV
1W
CX
+VS
R4
1 LBR
C2
2N3906
or Equivalent
Q2
3
CX
4391
CX
R5
65-2053
Figure 18. Voltage Dependent Oscillator
65-2159
Figure 18. Current Limiting
13
14
Q46
R1
540K
Q7
4X
Q1
R4
76K
R3
8.2K
Q8
C1
25 pF
Q4
0.6
Q5
Q45
0.3
R9
60K
+VS
(6)
R10
50K
R2
3K
Q6
Q9
R8
150K
Q44
Q2
(7)
VREF
R11
160K
Q4B
D7
Q14
Q11
Q10
Q3A
0.5
D8
Q12
Q3B
0.5
LBR
(1)
Q16
2X
Q15
Q22
D9
Q20
Q24
D18
(8)
VFB
D16
Q25
D17
Q23
Q21
Q17
Q18
Q19
0.1
0.1
0.1
D15
LBD
(2)
D10
Q26
40X
R6
20K
Q27
10X
R5
10K
Q3
Q35 Q36
Q13
Q41
Q29
50X
(5)
LX
R7
10K
Q28
CX
(3)
Q42
2X
Q37
Q31
0.9
D12
Q38
Q32
0.9
Q43
2X
Q40
Q39
Q33
0.9
65-6364
(4)
GND
D14
D13
Q34
1.1
PRODUCT SPECIFICATION
RC4391
Simplified Schematic Diagram
RC4391
PRODUCT SPECIFICATION
Troubleshooting Chart
Symptom
Draws excessive supply current on star-up.
Possible Problems
Inductance value too low.
Output frequency (FO) too low.
Combination of low resistance inductor and high
value filter capacitor — needs current limiting circuit
(Figure 13).
Output voltage is low.
Inductance value too high for FO or core saturating.
Inductor "sings" with audible hum.
Not potted well or bolted loosely.
LX pin appears noisy — scope will not synchronize.
Normal operating condition.
Inductor is saturating:
-IMAX
ILX
1. Core too small.
2. Core too hot.
3. Operating frequency too low.
Time
Inductor current shows nonlinear waveform.
Waveform has resistive component:
-IMAX
ILX
1. Wire size too small.
2. Power transistor lacks base drive.
3. Components not rated high enough.
Time
4. Battery has high series resistance.
Inductor current shows nonlinear waveform.
-IMAX
External transistor lacks base drive or beta is too
low.
ILX
Time
Inductor current is linear until high current is reached.
Poor efficiency.
Core saturating.
Diode or transistor:
1. Not fast enough.
2. Not rated for current level (high VCESAT).
High series resistance.
Operating frequency too high.
Motorboating (erratic current pulses).
Loop stability problem — needs feedback from
VOUT to VFB (pin 8), 100pF to 1000pF
15
PRODUCT SPECIFICATION
RC4391
Pot Core Inductor Design
Electrical Circuit
Magentic Circuit
I
E=I*R
E
H =B •
R
1
U
North
South
Flux
65-3464-07
Figure 20. Electricity vs. Magnetism
Electricity Versus Magnetism
Electrically the inductor must meet just one requirement, but
that requirement can be hard to satisfy. The inductor must
exhibit the correct value of inductance (L, in Henrys) as the
inductor current rises to its highest operating value (IMAX).
This requirement can be met most simply by choosing a very
large core and winding it until it reaches the correct inductance value, but that brute force technique wastes size,
weight and money. A more efficient design technique must
be used.
the point where the permeability decreases, the magnetic
field has realigned all of the magnetic domains in the core
material. Once all of the domains have been aligned the core
will then carry no more flux than just air, it becomes as if
there were no core at all. This phenomenon is called saturation. Because the inductance value, L, is dependent on the
amount of flux, core saturation will cause the value of L to
decrease dramatically, in turn causing excessive and possibly
destructive inductor current.
6000
First Difference: Permeability instead of being analogous to
resistance, is actually more like conductance (1/R). As permeability increases, flux increases.
Second Difference: Resistance is a linear function. As voltage increases, current increases proportionally, and the resistance value stays the same. In a magnetic circuit the value of
permeability varies as the applied magnetic force varies. This
nonlinear characteristic is usually shown in graph form in
ferrite core manufacturer's data sheet.
As the applied magnetizing force increases, at some point the
permeability will start decreasing, and therefore the amount
of magnetic flux will not increase any further, even as the
magnetizing force increases. The physical reality is that, at
16
B Gauss
First, one must understand how the inductor's magnetic field
works. The magnetic circuit in the inductor is very similar to
a simple resistive electrical circuit. There is a magnetizing
force (H, in oersteds), a flow of magnetism, or flux density
(B, in Gauss), and a resistance to the flux, called permeability
(U, in Gauss per oersted). H is equivalent to voltage in the
electrical model, flux density is like current flow, and permeability is like resistance (except for two important differences
discussed to the right).
+25¡C
5000
+85¡C
4000
+125¡C
3000
2000
1000
Stackpole Ceramag 24B
Hysteresis Loop vs. Temperature
0
-0.5 0 0.5 1
2 2.5 3
5
7
65-2170
Question: What happens if too small a core is used?
9
H Oersteds
Figure 21. Typical Manufacturer’s Curve Showing
Saturation Effects
Pot Cores for RC4391
Pot core inductors are best suited for the RC4391 switching
regulator for several reasons:
1. They are available in a wide range of sizes. RC4391
applications are usually low power with relatively low
peak currents (less than 500mA). A small inexpensive pot
core can be chosen to meet the circuit requirements.
2. Pot cores are easily mounted. They can be bolted
directly to the PC card adjacent to the regulator IC.
RC4391
PRODUCT SPECIFICATION
Use of the Design Aid Graph
3. Pot cores can be easily air-gapped. The length of the
gap is simply adjusted using different washer
thicknesses. cores are also available with predetermined
air gaps.
1. From the application requirement, determine the
inductor value (L) and the required peak current (IMAX).
2. Observe the curves of the design aid graph and determine
the smallest core that meets both the L and I
requirements.
4. Electromagnetic interference (EMI) is kept to a
minimum. the completely enclosed design of a pot core
reduces stray electromagnetic radiation—an important
consideration if the regulator circuit is built on a PC card
with other circuitry.
3. Note the approximate air gap at IMAX for the selected
core, and order the core with the gap. (If the gapping is
done by the user, remember that a washer lspacer results
in an air gap of twice the washer thickness, because two
gaps will be created, one at the center post and one at the
rim, like taking two bites from a doughnut.)
Not quite. Core size is dependent on the amount of energy
stored, not on load power. Raising the operating frequency
allows smaller cores and windings. Reduction of the size of
the magnetics is the main reason switching regulator design
tends toward higher operating frequency. Designs with the
RC4391 should use 75 kHz as a maximum running frequency, because the turn off delay of the power transistor
and stray capacitive coupling begin to interfere. Most applications are in the 10 to 50 kHz range, for efficiency and EMI
reasons.
4. If the required inductance is equal to the indicated value
on the graph, then wind the core with the number of turns
shown in the table of sizes. The turns given are the
maximum number for that gauge of wire that can be
easily wound in cores winding area.
5. If the required inductance is less than the value indicated
on the graph, a simple calculation must be done to find
the adjusted number of turns. Find AL (inductance index)
for a specific air gap.
The peak inductor current (IMAX) must reach a high enough
value to meet the load current and simultaneously the inductor value is decreased, then the core can be made smaller.
For a given core size and winding, an increase in air gap
spacing (an air gap is a break in the material in the magnetic
path, like a section broken off a doughnut) will cause the
inductance to decrease and IMAX (the usable peak current
before saturation )to increase.
inHenriesö
L ( indicated )
--------------------------------- = AL æ ------------------------2 ø
2
è
Turn
Turns
Then divide the required inductance value by AL to give the
actual turns squared, and take the square root to find the
actual turns needed.
The curves shown are typical of the ferrite manufacturer's
power HF material, such as Siemens N27 or Stackpole 24B,
which are usually offered in standard millimeter sizes
including the sizes shown.
#1
22X
13 mm
24 Gauge
70 Turns
DCW = 0.5W
L ( required )
ActualTurns = ------------------------------AL
Air Gap = 0.02"
3A
#3
#4
14X
8 mm
28 Gauge
60 Turns
DCW = 0.6W
11X
7 mm
30 Gauge
50 Turns
DCW = 1W
2A
Air Gap = 0.006"
#1
1A
#2
No Air Gap
#3
0
65-2171
#2
18X
11 mm
26 Gauge
70 Turns
DCW = 0.7W
IMAX (Amperes*)
Air Gap = 0.012"
#4
1 mH
2 mH
3 mH
Inductor Value (Henries)
*Includes safety margin (25%) to ensure nonsaturation
Figure 22. Inductor Design Aid
17
PRODUCT SPECIFICATION
RC4391
If the actual number of turns is significantly less than the
number from the table then the wire size can be increased to
use up the leftover winding area and reduce resistive losses.
6. Wind and gap the core as per calculations, and measure
the value with an inductance meter. Some adjustment of
the number of turns may be necessary.
Where:
N = number of turns
Ae = core area from data sheet (in cm2)
le = magnetic path length from data sheet (in cm)
ue =permeability of core from manufacturer's graph
g = center post air gap (in cm)
The saturation characteristics may be checked with the
inductor wired into the switching regulator application
circuit. To do so, build and power up the circuit. Then clamp
an oscilloscope current probe (recommend Tektronix P6042
or equivalent) around the inductor lead and monitor the current in the inductor. Draw the maximum load current from
the application circuit so that the regulator is running at close
to full duty cycle. Compare the waveform you see to those
pictured.
Manufacturers
Check for saturation at the highest expected ambient
temperature.
Siemens Company
186 Wood Avenue South
Iselin, NJ 08830
7. After the operation in circuit has been checked,
reassemble and pot the core using a potting compound
recommended by the manufacturer.
If the core material differs greatly in magnetic
characteristics from the standard power material shown
in Figure 16, then the following general equation can be
used to help in winding and gapping. This equation can
be used for any core geometry, such as an E-E core.
2
Below is a list of several pot core manufacturers:
Ferroxcube Company
5083 Kings Highway
Saugerties, NY 12477
Indiana General Electronics
Keasley, NJ 08832
Stackpole Company
201 Stackpole Street
St. Mary, PA 15857
TDK Electronics
13-1, 1-Chrome
Nihonbaski, Chuo-ku, Tokyo
8
( 1.26 ) ( N ) ( Ae ) ( 10 )
L X = ----------------------------------------------------g = ( le/ue )
Proper Operation
(Waveform is Fairly Linear)
Improper Operation
(Waveform is Nonlinear, Inductor
Is Saturating)
IMAX
0
IMAX
0
65-3464-08
Figure 23. Inductor Current Waveforms
18
RC4391
PRODUCT SPECIFICATION
Mechanical Dimensions
8-Lead Ceramic DIP Package
Inches
Symbol
Min.
A
b1
b2
c1
D
E
e
eA
L
Q
s1
a
Millimeters
Max.
Min.
—
.200
.014
.023
.045
.065
.008
.015
—
.405
.220
.310
.100 BSC
.300 BSC
.125
.200
.015
.060
.005
—
90¡
105¡
Notes:
Notes
Max.
—
5.08
.36
.58
1.14
1.65
.20
.38
—
10.29
5.59
7.87
2.54 BSC
7.62 BSC
3.18
5.08
.38
1.52
.13
—
90¡
105¡
1. Index area: a notch or a pin one identification mark shall be located
adjacent to pin one. The manufacturer's identification shall not be
used as pin one identification mark.
8
2, 8
2. The minimum limit for dimension "b2" may be .023 (.58mm) for leads
number 1, 4, 5 and 8 only.
8
4
4
5, 9
7
3. Dimension "Q" shall be measured from the seating plane to the base
plane.
3
6
4. This dimension allows for off-center lid, meniscus and glass overrun.
5. The basic pin spacing is .100 (2.54mm) between centerlines. Each
pin centerline shall be located within ±.010 (.25mm) of its exact
longitudinal position relative to pins 1 and 8.
6. Applies to all four corners (leads number 1, 4, 5, and 8).
7. "eA" shall be measured at the center of the lead bends or at the
centerline of the leads when "a" is 90¡.
8. All leads – Increase maximum limit by .003 (.08mm) measured at the
center of the flat, when lead finish applied.
9. Six spaces.
D
4
1
Note 1
E
5
8
s1
eA
e
A
Q
a
c1
L
b2
b1
19
PRODUCT SPECIFICATION
RC4391
Mechanical Dimensions (continued)
8-Lead Plastic DIP Package
Inches
Symbol
A
A1
A2
B
B1
C
D
D1
E
E1
e
eB
L
Millimeters
Min.
Max.
Min.
Max.
—
.015
.115
.014
.045
.008
.348
.005
.300
.240
.210
—
.195
.022
.070
.015
.430
—
.325
.280
—
.38
2.93
.36
1.14
.20
8.84
.13
7.62
6.10
5.33
—
4.95
.56
1.78
.38
10.92
—
8.26
7.11
.100 BSC
—
.430
.115
.160
2.54 BSC
—
10.92
2.92
4.06
8¡
8¡
N
Notes:
Notes
1. Dimensioning and tolerancing per ANSI Y14.5M-1982.
2. "D" and "E1" do not include mold flashing. Mold flash or protrusions
shall not exceed .010 inch (0.25mm).
3. Terminal numbers are for reference only.
4. "C" dimension does not include solder finish thickness.
5. Symbol "N" is the maximum number of terminals.
4
2
2
5
D
4
1
5
8
E1
D1
E
e
A2
A
A1
C
L
B1
20
B
eB
RC4391
PRODUCT SPECIFICATION
Mechanical Dimensions (continued)
8-Lead SOIC Package
Inches
Symbol
Min.
A
A1
B
C
D
E
e
H
h
L
N
a
ccc
Millimeters
Max.
Min.
Max.
.053
.069
.004
.010
.013
.020
.008
.010
.189
.197
.150
.158
.050 BSC
1.35
1.75
0.10
0.25
0.33
0.51
0.20
0.25
4.80
5.00
3.81
4.01
1.27 BSC
.228
.010
.016
5.79
0.25
0.40
.244
.020
.050
8
6.20
0.50
1.27
8
0¡
8¡
0¡
8¡
—
.004
—
0.10
8
Notes:
Notes
1. Dimensioning and tolerancing per ANSI Y14.5M-1982.
2. "D" and "E" do not include mold flash. Mold flash or
protrusions shall not exceed .010 inch (0.25mm).
3. "L" is the length of terminal for soldering to a substrate.
4. Terminal numbers are shown for reference only.
5
2
2
5. "C" dimension does not include solder finish thickness.
6. Symbol "N" is the maximum number of terminals.
3
6
5
E
1
H
4
h x 45¡
D
C
A1
A
SEATING
PLANE
e
B
–C–
LEAD COPLANARITY
a
L
ccc C
21
PRODUCT SPECIFICATION
RC4391
Ordering Information
Part Number
Package
Operating Temperature Range
RC4391N
8 Lead Plastic DIP
0˚C to +70°C
RC4391M
8 Lead Plastic SOIC
0˚C to +70°C
RV4391N
8 Lead Plastic DIP
-25°C to +85°C
RM4391D
8 Lead Ceramic DIP
-55˚C to +125°C
LIFE SUPPORT POLICY
FAIRCHILD’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES
OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF FAIRCHILD SEMICONDUCTOR
CORPORATION. As used herein:
1. Life support devices or systems are devices or systems
which, (a) are intended for surgical implant into the body,
or (b) support or sustain life, and (c) whose failure to
perform when properly used in accordance with
instructions for use provided in the labeling, can be
reasonably expected to result in a significant injury of the
user.
2. A critical component in any component of a life support
device or system whose failure to perform can be
reasonably expected to cause the failure of the life support
device or system, or to affect its safety or effectiveness.
www.fairchildsemi.com
5/20/98 0.0m 001
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Ó 1998 Fairchild Semiconductor Corporation