LT1511 Constant-Current/ Constant-Voltage 3A Battery Charger with Input Current Limiting DESCRIPTIO U FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ The LT®1511 current mode PWM battery charger is the simplest, most efficient solution to fast charge modern rechargeable batteries including lithium-ion (Li-Ion), nickelmetal-hydride (NiMH) and nickel-cadmium (NiCd) that require constant-current and/or constant-voltage charging. The internal switch is capable of delivering 3A* DC current (4A peak current). Full-charging current can be programmed by resistors or a DAC to within 5%. With 0.5% reference voltage accuracy, the LT1511 meets the critical constant-voltage charging requirement for Li-Ion cells. Simple Design to Charge NiCd, NiMH and Lithium Rechargeable Batteries—Charging Current Programmed by Resistors or DAC Adapter Current Loop Allows Maximum Possible Charging Current During Computer Use Precision 0.5% Accuracy for Voltage Mode Charging High Efficiency Current Mode PWM with 4A Internal Switch 5% Charging Current Accuracy Adjustable Undervoltage Lockout Automatic Shutdown When AC Adapter is Removed Low Reverse Battery Drain Current: 3µA Current Sensing Can Be at Either Terminal of the Battery Charging Current Soft-Start Shutdown Control A third control loop is provided to regulate the current drawn from the AC adapter. This allows simultaneous operation of the equipment and battery charging without overloading the adapter. Charging current is reduced to keep the adapter current within specified levels. U APPLICATIO S ■ ■ The LT1511 can charge batteries ranging from 1V to 20V. Ground sensing of current is not required and the battery’s negative terminal can be tied directly to ground. A saturating switch running at 200kHz gives high charging efficiency and small inductor size. A blocking diode is not required between the chip and the battery because the chip goes into sleep mode and drains only 3µA when the wall adapter is unplugged. Chargers for NiCd, NiMH, Lead-Acid, Lithium Rechargeable Batteries Switching Regulators with Precision Current Limit , LTC and LT are registered trademarks of Linear Technology Corporation. *See LT1510 for 1.5A Charger U TYPICAL APPLICATIO R7† 500Ω BOOST LT1511 COMP1 L1** 20µH D2 MBR0540T CLN VCC SW C2 0.47µF D1 MBRD340 CLP GND 200pF + + 10µF RS4† ADAPTER CURRENT SENSE CIN* 10µF UV VC SPIN RS3 200Ω 1% BAT RS2 200Ω 1% RS1 0.033Ω BATTERY CURRENT SENSE 50pF VIN (ADAPTER INPUT) 11V TO 28V TO MAIN SYSTEM POWER R5† UNDERVOLTAGE LOCKOUT R6 5k PROG OVP SENSE NOTE: COMPLETE LITHIUM-ION CHARGER, NO TERMINATION REQUIRED. RS4, R7 AND C1 ARE OPTIONAL FOR IIN LIMITING *TOKIN OR UNITED CHEMI-CON/MARCON CERAMIC SURFACE MOUNT **20µH COILTRONICS CTX20-4 †SEE APPLICATIONS INFORMATION FOR INPUT CURRENT LIMIT AND UNDERVOLTAGE LOCKOUT C1 1µF D3 MBRD340 300Ω 1k CPROG 1µF RPROG 4.93k 1% 0.33µF R3 390k 0.25% BATTERY VOLTAGE SENSE + COUT 22µF TANT + 4.2V + VBAT 2 Li-Ion 4.2V 1511 • F01 R4 162k 0.25% Figure 1. 3A Lithium-Ion Battery Charger 1 LT1511 U W U U W W W ABSOLUTE MAXIMUM RATINGS PACKAGE/ORDER INFORMATION (Note 1) Supply Voltage (VMAX, CLP and CLN Pin Voltage) ...................... 30V Switch Voltage with Respect to GND ...................... – 3V Boost Pin Voltage with Respect to VCC ................... 25V Boost Pin Voltage with Respect to GND ................. 57V Boost Pin Voltage with Respect to SW Pin .............. 30V VC, PROG, OVP Pin Voltage ...................................... 8V IBAT (Average)........................................................... 3A Switch Current (Peak) .............................................. 4A Operating Junction Temperature Range Commercial ........................................... 0°C to 125°C Industrial ......................................... – 40°C to 125°C Operating Ambient Temperature Commercial ............................................ 0°C to 70°C Industrial ........................................... – 40°C to 85°C Storage Temperature Range ................. – 65°C to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C ORDER PART NUMBER TOP VIEW GND** 1 24 GND** SW 2 23 GND** BOOST 3 22 VCC1* GND** 4 21 VCC2* GND** 5 20 VCC3* UV 6 19 PROG GND** 7 LT1511CSW LT1511ISW *ALL VCC PINS SHOULD BE CONNECTED TOGETHER CLOSE TO THE PINS ** ALL GND PINS ARE FUSED TO INTERNAL DIE ATTACH PADDLE FOR HEAT SINKING. CONNECT THESE PINS TO EXPANDED PC LANDS FOR PROPER HEAT SINKING. 30°C/W THERMAL RESISTANCE ASSUMES AN INTERNAL GROUND PLANE DOUBLING AS A HEAT SPREADER 18 VC OVP 8 17 UVOUT CLP 9 16 GND** CLN 10 15 COMP2 COMP1 11 14 BAT SENSE 12 13 SPIN SW PACKAGE 24-LEAD PLASTIC SO WIDE TJMAX = 125°C, θJA = 30°C/ W** Consult factory for Military grade parts. ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VCC = 16V, VBAT = 8V, VMAX (maximum operating VCC) = 28V, RS2 = RS3 = 200Ω (see Block Diagram), VCLN = VCC. No load on any outputs unless otherwise noted. PARAMETER CONDITIONS MIN TYP MAX UNITS 4.5 4.6 6.8 7.0 mA mA 100 10 105 12 13 mV mV mV 110 13 14 mV mV mV Overall Supply Current VPROG = 2.7V, VCC ≤ 20V VPROG = 2.7V, 20V < VCC ≤ 25V Sense Amplifier CA1 Gain and Input Offset Voltage (With RS2 = 200Ω, RS3 = 200Ω) (Measured across RS1)(Note 2) 8V ≤ VCC ≤ 25V , 0V ≤ VBAT ≤ 20V RPROG = 4.93k RPROG = 49.3k TJ < 0°C VCC = 28V, VBAT = 20V RPROG = 4.93k RPROG = 49.3k TJ < 0°C ● ● ● ● 95 8 7 ● ● 90 7 6 6 VCC Undervoltage Lockout (Switch OFF) Threshold Measured at UV Pin ● 7 8 V UV Pin Input Current 0.2V ≤ VUV ≤ 8V ● 0.1 5 µA UV Output Voltage at UVOUT Pin In Undervoltage State, IUVOUT = 70µA ● 0.1 0.5 V UV Output Leakage Current at UVOUT Pin 8V ≤ VUV, VUVOUT = 5V ● 0.1 3 µA Reverse Current from Battery (When VCC Is Not Connected, VSW Is Floating) VBAT ≤ 20V, VUV ≤ 0.4V 3 15 µA 2 LT1511 ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VCC = 16V, VBAT = 8V, VMAX (maximum operating VCC) = 28V, RS2 = RS3 = 200Ω (see Block Diagram), VCLN = VCC. No load on any outputs unless otherwise noted. PARAMETER CONDITIONS MIN TYP MAX UNITS 0.1 0.25 6 8 10 20 9 12 µA µA mA mA 0.15 0.25 Ω 25 35 mA/A 2 4 100 200 µA µA 20 µA Overall Boost Pin Current VCC = 20V, VBOOST = 0V VCC = 28V, VBOOST = 0V 2V ≤ VBOOST – VCC < 8V (Switch ON) 8V ≤ VBOOST – VCC ≤ 25V (Switch ON) Switch Switch ON Resistance 8V ≤ VCC ≤ VMAX, ISW = 3A, VBOOST – VSW ≥ 2V ∆IBOOST/∆ISW During Switch ON VBOOST = 24V, ISW ≤ 3A Switch OFF Leakage Current VSW = 0V, VCC ≤ 20V 20V < VCC ≤ 28V ● ● ● Minimum IPROG for Switch ON ● 2 4 Minimum IPROG for Switch OFF at VPROG ≤ 1V ● 1 2.4 Maximum VBAT for Switch ON ● mA VCC – 2 V – 125 µA Current Sense Amplifier CA1 Inputs (Sense, BAT) Input Bias Current ● Input Common Mode Low ● Input Common Mode High ● – 50 – 0.25 SPIN Input Current V VCC – 2 V – 100 – 200 µA 2.465 2.477 V 2.489 2.489 V V Reference Reference Voltage (Note 3) Reference Voltage RPROG = 4.93k, Measured at OVP with VA Supplying IPROG and Switch OFF All Conditions of VCC, TJ > 0°C TJ < 0°C (Note 4) 2.453 ● ● 2.441 2.43 Oscillator Switching Frequency Switching Frequency 180 200 220 kHz ● ● 170 160 200 230 230 kHz kHz ● TA = 25°C 85 90 93 VC = 1V, IVC = ±1µA 150 250 All Conditions of VCC, TJ > 0°C TJ < 0°C Maximum Duty Cycle % % Current Amplifier CA2 Transconductance Maximum VC for Switch OFF IVC Current (Out of Pin) ● VC ≥ 0.6V VC < 0.45V 550 µmho 0.6 V 100 3 µA mA 3 LT1511 ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VCC = 16V, VBAT = 8V, VMAX (maximum operating VCC) = 28V. No load on any outputs unless otherwise noted. PARAMETER CONDITIONS MIN TYP MAX UNITS Transconductance (Note 3) Output Current from 50µA to 500µA 0.25 0.6 1.3 mho Output Source Current VOVP = VREF + 10mV, VPROG = VREF + 10mV 1.1 OVP Input Bias Current At 0.75mA VA Output Current At 0.75mA VA Output Current, TJ > 90°C – 15 Turn-On Threshold 0.75mA Output Current Transconductance Output Current from 50µA to 500µA CLP Input Current CLN Input Current Voltage Amplifier VA mA ±3 ±10 25 nA nA 93 100 107 mV 0.5 1 2 mho 0.75mA Output Current, VUV ≥ 0.4V 0.3 1 µA 0.75mA Output Current VUV ≥ 0.4V 0.8 2 mA Current Limit Amplifier CL1, 8V ≤ Input Common Mode Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: Tested with Test Circuit 1. Note 3: Tested with Test Circuit 2. Note 4: A linear interpolation can be used for reference voltage specification between 0°C and – 40°C. U W TYPICAL PERFORMANCE CHARACTERISTICS Thermally Limited Maximum Charging Current 8 2.8 96 16.8V BATTERY 92 CHARGER EFFICIENCY 90 88 (θJA =30°C/W) TAMAX =60°C TJMAX =125°C 5 10 20 15 INPUT VOLTAGE (V) INCLUDES LOSS IN DIODE D3 84 25 30 1511 • TPC01 NOTE: FOR 4.2V AND 8.4V BATTERIES MAXIMUM CHARGING CURRENT IS 3A FOR VIN – VBAT ≥ 3V 0°C 125°C 4 25°C 2 1 82 2.0 5 3 86 2.2 ICC (mA) 94 4.2V BATTERY VIN ≥ 8V 2.4 7 6 8.4V BATTERY VIN ≥ 11V 2.6 VCC = 16V VIN = 16.5 VBAT = 8.4V 98 EFFICIENCY (%) MAXIMUM CHARGING CURRENT (A) 100 12.6V BATTERY 4 ICC vs Duty Cycle Efficiency of Figure 1 Circuit 3.0 0 80 0.2 0.6 1.0 1.4 1.8 IBAT (A) 2.2 2.6 3.0 1511 • TPC02 0 10 20 30 40 50 60 DUTY CYCLE (%) 70 80 1511 • TPC03 LT1511 U W TYPICAL PERFORMANCE CHARACTERISTICS Switching Frequency vs Temperature VREF Line Regulation ICC vs VCC 7.0 210 0.003 MAXIMUM DUTY CYCLE 0.002 0°C 6.5 25°C 200 195 0.001 6.0 ∆VREF (V) ICC (mA) FREQUENCY (kHz) 205 125°C 5.5 ALL TEMPERATURES 0 –0.001 190 5.0 185 180 –20 –0.002 4.5 0 20 40 60 80 100 120 140 TEMPERATURE (°C) 0 10 5 15 VCC (V) 20 25 30 –0.003 0 5 10 15 VCC (V) 20 1511 • TPC05 1511 • TPC04 IVA vs ∆VOVP (Voltage Amplifier) 30 1511 • TPC06 Maximum Duty Cycle 4 25 VC Pin Characteristics 98 –1.20 –1.08 97 –0.96 2 125°C 1 96 –0.84 95 –0.72 IVC (mA) DUTY CYCLE (%) 94 –0.48 –0.36 92 –0.24 –0.12 25°C 91 0 90 0 0 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 IVA (mA) 20 40 60 80 100 120 0.12 140 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 VC (V) 1511 • TPC08 1511 • TPC09 Reference Voltage vs Temperature Switch Current vs Boost Current vs Boost Voltage PROG Pin Characteristics 50 6 45 BOOST CURRENT (mA) 40 125°C 25°C 0 0 TEMPERATURE (°C) 1511• TPC07 IPROG (mA) –0.60 93 35 2.470 VCC = 16V 2.468 VBOOST = 38V 28V 18V REFERENCE VOLTAGE (V) ∆VOVP (mV) 3 30 25 20 15 10 2.466 2.464 2.462 2.460 5 –6 0 1 2 3 VPROG (V) 4 5 1511 • TPC10 0 2.458 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 SWITCH CURRENT (A) 1511 • TPC11 0 25 50 75 100 TEMPERATURE 125 150 LT1511 • TPC12 5 LT1511 U U U PIN FUNCTIONS GND (Pins 1, 4, 5, 7, 16, 23, 24): Ground Pin. SW (Pin 2): Switch Output. The Schottky catch diode must be placed with very short lead length in close proximity to SW pin and GND. BOOST (Pin 3): This pin is used to bootstrap and drive the switch power NPN transistor to a low on-voltage for low power dissipation. In normal operation, VBOOST = VCC + VBAT when switch is on. Maximum allowable VBOOST is 55V. UV (Pin 6): Undervoltage Lockout Input. The rising threshold is at 6.7V with a hysteresis of 0.5V. Switching stops in undervoltage lockout. When the supply (normally the wall adapter output) to the chip is removed, the UV pin has to be pulled down to below 0.7V (a 5k resistor from adapter output to GND is required) otherwise the reverse battery current drained by the chip will be approximately 200µA instead of 3µA. Do not leave UV pin floating. If it is connected to VIN with no resistor divider, the built-in 6.7V undervoltage lockout will be effective. OVP (Pin 8): This is the input to the amplifier VA with a threshold of 2.465V. Typical input current is about 3nA out of pin. For charging lithium-ion batteries, VA monitors the battery voltage and reduces charging when battery voltage reaches the preset value. If it is not used, the OVP pin should be grounded. CLP (Pin 9): This is the positive input to the supply current limit amplifier CL1. The threshold is set at 100mV. When used to limit supply current, a filter is needed to filter out the 200kHz switching noise. CLN (Pin 10): This is the negative input to the amplifier CL1. COMP1 (Pin 11): This is the compensation node for the amplifier CL1. A 200pF capacitor is required from this pin to GND if input current amplifier CL1 is used. At input adapter current limit, this node rises to 1V. By forcing COMP1 low with an external transistor, amplifier CL1 will be defeated (no adapter current limit). COMP1 can source 200µA. 6 SENSE (Pin 12): Current Amplifier CA1 Input. Sensing can be at either terminal of the battery. SPIN (Pin 13): This pin is for the internal amplifier CA1 bias. It has to be connected to RS1 as shown in the 3A Lithium Battery Charger (Figure 1). BAT (Pin 14): Current Amplifier CA1 Input. COMP2 (Pin 15): This is also a compensation node for the amplifier CL1. It gets up to 2.8V at input adapter current limit and/or at constant-voltage charging. UVOUT (Pin 17): This is an open collector output for undervoltage lockout status. It stays low in undervoltage state. With an external pull-up resistor , it goes high at valid VCC. Note that the base drive of the open collector NPN comes from CLN pin. UVOUT stays low only when CLN is higher than 2V. Pull-up current should be kept under 100µA. VC (Pin 18): This is the control signal of the inner loop of the current mode PWM. Switching starts at 0.7V. Higher VC corresponds to higher charging current in normal operation. A capacitor of at least 0.33µF to GND filters out noise and controls the rate of soft-start. To shut down switching, pull this pin low. Typical output current is 30µA. PROG (Pin 19): This pin is for programming the charging current and for system loop compensation. During normal operation, VPROG stays close to 2.465V. If it is shorted to GND the switching will stop. When a microprocessor controlled DAC is used to program charging current, it must be capable of sinking current at a compliance up to 2.465V. VCC (Pins 20, 21, 22): This is the supply of the chip. For good bypass, a low ESR capacitor of 20µF or higher is required, with the lead length kept to a minimum. VCC should be between 8V and 28V and at least 3V higher than VBAT. Undervoltage lockout starts and switching stops when VCC goes below 7V. Note that there is a parasitic diode inside from SW pin to VCC pin. Do not force VCC below SW by more than 0.7V with battery present. All three VCC pins should be shorted together close to the pins. LT1511 W BLOCK DIAGRAM – UV UVOUT + + 6.7V 200kHz OSCILLATOR + SHUTDOWN 0.7V + VCC – VSW S BOOST – VCC R R + + SLOPE COMPENSATION SW 1.5V VCC SPIN VBAT – PWM C1 – + B1 + GND QSW R R2 + SENSE RS3 – BAT RS2 CA1 IPROG R1 1k R3 0VP – VA CA2 + 75k RS1 BAT + VC IBAT gm = 0.64 VREF Ω – VREF 2.465V 100mV + CLP + CL1 – CLN COMP1 COMP2 PROG CPROG IPROG RPROG (I )(R ) IBAT = PROG S2 RS1 RS2 = 2.465V RPROG RS1 1511 BD ( )( ) (RS3 = RS2) 7 LT1511 TEST CIRCUITS Test Circuit 1 SPIN LT1511 CA1 CA2 1k + 0.047µF + – VC 60k – SENSE RS3 200Ω BAT RS2 200Ω RS1 10Ω + VBAT VREF PROG 1µF RPROG 300Ω + LT1006 1k + 1511 • TC01 – ≈ 0.65V 20k Test Circuit 2 LT1511 OVP + VA – VREF PROG 10k IPROG 10k 0.47µF RPROG – + + LT1013 2.465V 1511 • TC02 U OPERATION The LT1511 is a current mode PWM step-down (buck) switcher. The battery DC charging current is programmed by a resistor RPROG (or a DAC output current) at the PROG pin (see Block Diagram). Amplifier CA1 converts the charging current through RS1 to a much lower current IPROG fed into the PROG pin. Amplifier CA2 compares the output of CA1 with the programmed current and drives the PWM loop to force them to be equal. High DC accuracy is achieved with averaging capacitor CPROG. Note that IPROG has both AC and DC components. IPROG goes through R1 and generates a ramp signal that is fed to the PWM control comparator C1 through buffer B1 and level shift resistors 8 R2 and R3, forming the current mode inner loop. The Boost pin drives the switch NPN QSW into saturation and reduces power loss. For batteries like lithium-ion that require both constant-current and constant-voltage charging, the 0.5%, 2.465V reference and the amplifier VA reduce the charging current when battery voltage reaches the preset level. For NiMH and NiCd, VA can be used for overvoltage protection. When input voltage is not present, the charger goes into low current (3µA typically) sleep mode as input drops down to 0.7V below battery voltage. To shut down the charger, simply pull the VC pin low with a transistor. LT1511 U U W U APPLICATIONS INFORMATION Input and Output Capacitors Soft-Start In the 3A Lithium Battery Charger (Figure 1), the input capacitor (CIN) is assumed to absorb all input switching ripple current in the converter, so it must have adequate ripple current rating. Worst-case RMS ripple current will be equal to one half of output charging current. Actual capacitance value is not critical. Solid tantalum capacitors such as the AVX TPS and Sprague 593D series have high ripple current rating in a relatively small surface mount package, but caution must be used when tantalum capacitors are used for input bypass. High input surge currents can be created when the adapter is hot-plugged to the charger and solid tantalum capacitors have a known failure mechanism when subjected to very high turn-on surge currents. Highest possible voltage rating on the capacitor will minimize problems. Consult with the manufacturer before use. Alternatives include new high capacity ceramic (5µF to 20µF) from Tokin or United Chemi-Con/ Marcon, et al., and the old standby, aluminum electrolytic, which will require more microfarads to achieve adequate ripple rating. Sanyo OS-CON can also be used. The LT1511 is soft started by the 0.33µF capacitor on the VC pin. On start-up, VC pin voltage will rise quickly to 0.5V, then ramp at a rate set by the internal 45µA pull-up current and the external capacitor. Battery charging current starts ramping up when VC voltage reaches 0.7V and full current is achieved with VC at 1.1V. With a 0.33µF capacitor, time to reach full charge current is about 10ms and it is assumed that input voltage to the charger will reach full value in less than 10ms. The capacitor can be increased up to 1µF if longer input start-up times are needed. The output capacitor (COUT) is also assumed to absorb output switching current ripple. The general formula for capacitor current is: ( ) V 0.29 (VBAT) 1 – BAT VCC IRMS = (L1)(f) For example, VCC = 16V, VBAT = 8.4V, L1 = 20µH, and f = 200kHz, IRMS = 0.3A. EMI considerations usually make it desirable to minimize ripple current in the battery leads, and beads or inductors may be added to increase battery impedance at the 200kHz switching frequency. Switching ripple current splits between the battery and the output capacitor depending on the ESR of the output capacitor and the battery impedance. If the ESR of COUT is 0.2Ω and the battery impedance is rased to 4Ω with a bead or inductor, only 5% of the current ripple will flow in the battery. In any switching regulator, conventional timer-based soft starting can be defeated if the input voltage rises much slower than the time out period. This happens because the switching regulators in the battery charger and the computer power supply are typically supplying a fixed amount of power to the load. If input voltage comes up slowly compared to the soft start time, the regulators will try to deliver full power to the load when the input voltage is still well below its final value. If the adapter is current limited, it cannot deliver full power at reduced output voltages and the possibility exists for a quasi “latch” state where the adapter output stays in a current limited state at reduced output voltage. For instance, if maximum charger plus computer load power is 30W, a 15V adapter might be current limited at 2.5A. If adapter voltage is less than (30W/2.5A = 12V) when full power is drawn, the adapter voltage will be sucked down by the constant 30W load until it reaches a lower stable state where the switching regulators can no longer supply full load. This situation can be prevented by utilizing undervoltage lockout, set higher than the minimum adapter voltage where full power can be achieved. A fixed undervoltage lockout of 7V is built into the VCC pin, but an additional adjustable lockout is also available on the UV pin. Internal lockout is performed by clamping the VC pin low. The VC pin is released from its clamped state when the UV pin rises above 6.7V and is pulled low when the UV pin drops below 6.2V (0.5V hysteresis). At the same time UVOUT goes high with an external pull-up resistor. This signal can be used to alert the system that charging is about to start. The charger will start delivering current about 4ms after VC is released, as set by the 0.33µF 9 LT1511 U U W U APPLICATIONS INFORMATION capacitor. A resistor divider is used to set the desired VCC lockout voltage as shown in Figure 2. A typical value for R6 is 5k and R5 is found from: R5 = R6(VIN – VUV ) VUV VUV = Rising lockout threshold on the UV pin VIN = Charger input voltage that will sustain full load power Example: With R6 = 5k, VUV = 6.7V and setting VIN at 12V; R5 = 5k (12V – 6.7V)/6.7V = 4k The resistor divider should be connected directly to the adapter output as shown, not to the VCC pin to prevent battery drain with no adapter voltage. If the UV pin is not used, connect it to the adapter output (not VCC) and connect a resistor no greater than 5k to ground. Floating the pin will cause reverse battery current to increase from 3µA to 200µA. If connecting the unused UV pin to the adapter output is not possible for some reason, it can be grounded. Although it would seem that grounding the pin creates a permanent lockout state, the UV circuitry is arranged for phase reversal with low voltages on the UV pin to allow the grounding technique to work. 100mV + + CLP 1µF CL1 CLN – 500Ω AC ADAPTER OUTPUT RS4* VCC VIN + LT1511 R5 being charged without complex load management algorithms. Additionally, batteries will automatically be charged at the maximum possible rate of which the adapter is capable. This feature is created by sensing total adapter output current and adjusting charging current downward if a preset adapter current limit is exceeded. True analog control is used, with closed loop feedback ensuring that adapter load current remains within limits. Amplifier CL1 in Figure 2 senses the voltage across RS4, connected between the CLP and CLN pins. When this voltage exceeds 100mV, the amplifier will override programmed charging current to limit adapter current to 100mV/RS4. A lowpass filter formed by 500Ω and 1µF is required to eliminate switching noise. If the current limit is not used, both CLP and CLN pins should be connected to VCC. Charging Current Programming The basic formula for charging current is (see Block Diagram): IBAT = IPROG ( )( RS2 2.465V = RPROG RS1 )( ) RS2 RS1 where RPROG is the total resistance from PROG pin to ground. For the sense amplifier CA1 biasing purpose, RS3 should have the same value as RS2 and SPIN should be connected directly to the sense resistor (RS1) as shown in the Block Diagram. For example, 3A charging current is needed. To have low power dissipation on RS1 and enough signal to drive the amplifier CA1, let RS1 = 100mV/3A = 0.033Ω. This limits RS1 power to 0.3W. Let RPROG = 5k, then: UV *RS4 = 100mV ADAPTER CURRENT LIMIT R6 1511 • F02 Figure 2. Adapter Current Limiting Adapter Limiting An important feature of the LT1511 is the ability to automatically adjust charging current to a level which avoids overloading the wall adapter. This allows the product to operate at the same time that batteries are 10 )(R ) (I )(R RS2 = RS3 = BAT PROG S1 2.465V (3A)(5k)(0.033) = = 200Ω 2.465V Charging current can also be programmed by pulse width modulating IPROG with a switch Q1 to RPROG at a frequency higher than a few kHz (Figure 3). Charging current will be proportional to the duty cycle of the switch with full current at 100% duty cycle. LT1511 U U W U APPLICATIONS INFORMATION LT1511 PROG 300Ω RPROG 4.7k 5V 0V Q1 VN2222 PWM CPROG 1µF 1511 • F03 IBAT = (DC)(3A) Figure 3. PWM Current Programming When power is on, there is about 200µA of current flowing out of the BAT and Sense pins. If the battery is removed during charging, and total load including R3 and R4 is less than the 200µA, VBAT could float up to VCC even though the loop has turned switching off. To keep VBAT regulated to the battery voltage in this condition, R3 and R4 can be chosen to draw 0.5mA and Q3 can be added to disconnect them when power is off (Figure 4). R5 isolates the OVP pin from any high frequency noise on VIN. An alternative way is to use a Zener diode with a breakdown voltage two or three volts higher than battery voltage to clamp the VBAT voltage. Lithium-Ion Charging The 3A Lithium Battery Charger (Figure 1) charges lithiumion batteries at a constant 3A until battery voltage reaches a limit set by R3 and R4. The charger will then automatically go into a constant-voltage mode with current decreasing to zero over time as the battery reaches full charge. This is the normal regimen for lithium-ion charging, with the charger holding the battery at “float” voltage indefinitely. In this case no external sensing of full charge is needed. + R3 12k 0.25% LT1511 OVP Q3 VN2222 – + VIN R5 220k – VBAT 4.2V 4.2V R4 4.99k 0.25% LT1511 • F04 Figure 4. Disconnecting Voltage Divider Battery Voltage Sense Resistors Selection To minimize battery drain when the charger is off, current through the R3/R4 divider is set at 15µA. The input current to the OVP pin is 3nA and the error can be neglected. With divider current set at 15µA, R4 = 2.465/15µA = 162k and, R3 = (R4)(VBAT − 2.465) = 162k (8.4 − 2.465) 2.465 2.465 = 390k Li-Ion batteries typically require float voltage accuracy of 1% to 2%. Accuracy of the LT1511 OVP voltage is ±0.5% at 25°C and ±1% over full temperature. This leads to the possibility that very accurate (0.1%) resistors might be needed for R3 and R4. Actually, the temperature of the LT1511 will rarely exceed 50°C in float mode because charging currents have tapered off to a low level, so 0.25% resistors will normally provide the required level of overall accuracy. Some battery manufacturers recommend termination of constant-voltage float mode after charging current has dropped below a specified level (typically around 10% of the full current) and a further time out period of 30 minutes to 90 minutes has elapsed. This may extend the life of the battery, so check with manufacturers for details. The circuit in Figure 5 will detect when charging current has dropped below 400mA. This logic signal is used to initiate a timeout period, after which the LT1511 can be shut down by pulling the VC pin low with an open collector or drain. Some external means must be used to detect the need for additional charging or the charger may be turned on periodically to complete a short float-voltage cycle. Current trip level is determined by the battery voltage, R1 through R3 and the sense resistor (RS1). D2 generates hysteresis in the trip level to avoid multiple comparator transitions. 11 LT1511 U U W U APPLICATIONS INFORMATION IBAT R1 = RS3 200Ω SENSE RS1 0.033Ω LT1511 RS2 200Ω BAT VBAT ADAPTER OUTPUT BAT R1* C1 1.6k 0.1µF 3.3V OR 5V D1 1N4148 3 8 – 7 LT1011 2 R2 560k + R4 470k NEGATIVE EDGE TO TIMER 4 1 D2 1N4148 R3 430k * TRIP CURRENT = = R1(VBAT) (R2 + R3)(RS1) (1.6k)(8.4V) ≈ 400mA (560k + 430k)(0.033Ω) 1511 • F04 Figure 5. Current Comparator for Initiating Float Time Out Nickel-Cadmium and Nickel-Metal-Hydride Charging The circuit in the 3A Lithium Battery Charger (Figure 1) can be modified to charge NiCd or NiMH batteries. For example, 2-level charging is needed; 2A when Q1 is on and 200mA when Q1 is off. PROG 0.33µF R1 49.3k R2 5.49k Q1 1511 • F05 Figure 6. 2-Level Charging For 2A full current, the current sense resistor (RS1) should be increased to 0.05Ω so that enough signal (10mV) will be across RS1 at 0.2A trickle charge to keep charging current accurate. For a 2-level charger, R1 and R2 are found from; 12 ILOW R2 = (2.465)(4000 ) IHI − ILOW All battery chargers with fast charge rates require some means to detect full charge state in the battery to terminate the high charging current. NiCd batteries are typically charged at high current until temperature rise or battery voltage decrease is detected as an indication of near full charge. The charging current is then reduced to a much lower value and maintained as a constant trickle charge. An intermediate “top off” current may be used for a fixed time period to reduce 100% charge time. NiMH batteries are similar in chemistry to NiCd but have two differences related to charging. First, the inflection characteristic in battery voltage as full charge is approached is not nearly as pronounced. This makes it more difficult to use dV/dt as an indicator of full charge, and change of temperature is more often used with a temperature sensor in the battery pack. Secondly, constant trickle charge may not be recommended. Instead, a moderate level of current is used on a pulse basis (≈ 1% to 5% duty cycle) with the time-averaged value substituting for a constant low trickle. Please contact the Linear Technology Applications Department about charge termination circuits. If overvoltage protection is needed, R3 and R4 should be calculated according to the procedure described in LithiumIon Charging section. The OVP pin should be grounded if not used. LT1511 1k (2.465)(4000) When a microprocessor DAC output is used to control charging current, it must be capable of sinking current at a compliance up to 2.5V if connected directly to the PROG pin. Thermal Calculations If the LT1511 is used for charging currents above 1.5A, a thermal calculation should be done to ensure that junction temperature will not exceed 125°C. Power dissipation in the IC is caused by bias and driver current, switch resistance and switch transition losses. The SO wide package, with a thermal resistance of 30°C/W, can provide a full 3A charging current in many situations. A graph is shown in the Typical Performance Characteristics section. LT1511 U W U U APPLICATIONS INFORMATION PBIAS = (3.5mA )(VIN) + 1.5mA(VBAT ) 2 VBAT ) ( + 7.5mA + (0.012)(I ) VIN [ SW BAT ] (IBAT )(VBAT ) 1+ VBAT 30 PDRIVER = 55(VIN) C2 BOOST L1 D2 2 PSW 2 IBAT ) (RSW )(VBAT ) ( = + VIN ( tOL)(VIN)(IBAT )( f) RSW = Switch ON resistance ≈ 0.16Ω tOL = Effective switch overlap time ≈ 10ns f = 200kHz Example: VIN = 15V, VBAT = 8.4V, IBAT = 3A; PBIAS = (3.5mA )(15) + 1.5mA(8.4) + (8.4)2 15 [7.5mA + (0.012)(3)] = 0.27W (3)(8.4)2 1+ 830.4 PDRIVER = = 0.33W 55(15) 2 3) (0.16)(8.4) ( PSW = + 10−9 (15)(3)(200kHz ) 15 = 0.81 + 0.09 = 0.9W Total Power in the IC is: 0.27 + 0.33 + 0.9 = 1.5W Temperature rise will be (1.5W)(30°C/W) = 45°C. This assumes that the LT1511 is properly heat sunk by connecting the seven fused ground pins to expanded traces and that the PC board has a backside or internal plane for heat spreading. The PDRIVER term can be reduced by connecting the boost diode D2 (see Figure 1) to a lower system voltage (lower than VBAT) instead of VBAT. (IBAT )(VBAT )(VX ) 1+ V30X Then PDRIVER = 55(VIN ) For example, VX = 3.3V then: LT1511 SPIN VX IVX + 1511 • F07 10µF Figure 7. Lower VBOOST .3V (3A)(8.4V)(3.3V) 1+ 330 PDRIVER = = 0.11W 55(15V ) The average IVX required is: PDRIVER 0.11W = = 34mA 3.3V VX Fused-lead packages conduct most of their heat out the leads. This makes it very important to provide as much PC board copper around the leads as is practical. Total thermal resistance of the package-board combination is dominated by the characteristics of the board in the immediate area of the package. This means both lateral thermal resistance across the board and vertical thermal resistance through the board to other copper layers. Each layer acts as a thermal heat spreader that increases the heat sinking effectiveness of extended areas of the board. Total board area becomes an important factor when the area of the board drops below about 20 square inches. The graph in Figure 8 shows thermal resistance vs board area for 2-layer and 4-layer boards with continuous copper planes. Note that 4-layer boards have significantly lower thermal resistance, but both types show a rapid increase for reduced board areas. Figure 9 shows actual measured lead temperatures for chargers operating at full current. Battery voltage and input voltage will affect device power dissipation, so the data sheet power calculations must be used to extrapolate these readings to other situations. Vias should be used to connect board layers together. Planes under the charger area can be cut away from the rest of the board and connected with vias to form both a 13 LT1511 U W U U APPLICATIONS INFORMATION low thermal resistance system and to act as a ground plane for reduced EMI. Glue-on, chip-mounted heat sinks are effective only in moderate power applications where the PC board copper cannot be used, or where the board size is small. They offer very little improvement in a properly laid out multilayer board of reasonable size. Higher Duty Cycle for the LT1511 Battery Charger Maximum duty cycle for the LT1511 is typically 90%, but this may be too low for some applications. For example, if an 18V ±3% adapter is used to charge ten NiMH cells, the charger must put out 15V maximum. A total of 1.6V is lost in the input diode, switch resistance, inductor resistance and parasitics, so the required duty cycle is 15/16.4 = 91.4%. As it turn out, duty cycle can be extended to 93% THERMAL RESISTANCE (°C/W) 45 by restricting boost voltage to 5V instead of using VBAT as is normally done. This lower boost voltage also reduces power dissipation in the LT1511, so it is a win-win decision. Connect an external source of 3V to 6V at VX node in Figure 10 with a 10µF CX bypass capacitor. Even Lower Dropout For even lower dropout and/or reducing heat on the board, the input diode D3 should be replaced with a FET (see Figure 11). It is pretty straightforward to connect a P-channel FET across the input diode and connect its gate to the battery so that the FET commutates off when the input goes low. The problem is that the gate must be pumped low so that the FET is fully turned on even when the input is only a volt or two above the battery voltage. Also there is a turn-off speed issue. The FET should turn STANDARD CONNECTION 40 35 SW C3 0.47µF 2-LAYER BOARD SW C3 0.47µF BOOST LT1511 D2 30 HIGH DUTY CYCLE CONNECTION D2 SPIN 4-LAYER BOARD 25 BOOST LT1511 SPIN SENSE VX 3V TO 6V BAT SENSE BAT CX 10µF 20 VBAT MEASURED FROM AIR AMBIENT TO DIE USING COPPER LANDS AS SHOWN ON DATA SHEET 15 VBAT + + 10 0 5 20 15 25 10 BOARD AREA (IN2) 30 35 1511 F10 LT1511 • F08 Figure 10. High Duty Cycle Figure 8. LT1511 Thermal Resistance VIN 110 + Q1 NOTE: PEAK DIE TEMPERATURE WILL BE ABOUT 10°C HIGHER THAN LEAD TEMPERATURE AT 3A CHARGING CURRENT 100 LEAD TEMPERATURE (°C) HIGH DUTY CYCLE CONNECTION Q2 90 2-LAYER BOARD RX 50k 80 D1 C2 0.47µF D2 4-LAYER BOARD 70 60 VIN = 16V VBAT = 8.4V ICHRG = 3A TA = 25°C 50 40 0 5 VCC BOOST LT1511 SPIN Q1 = Si4435DY Q2 = TP0610L 4-LAYER BOARD WITH VBOOST = 3.3V 20 15 25 10 BOARD AREA (IN2) SW VX 3V TO 6V SENSE BAT CX 10µF VBAT 30 + 35 LT1511 • F09 1511 F11 Figure 9. LT1511 Lead Temperature 14 Figure 11. Replacing the Input Diode LT1511 U U W U APPLICATIONS INFORMATION off instantly when the input is dead shorted to avoid large current surges from the battery back through the charger into the FET. Gate capacitance slows turn-off, so a small P-channel (Q2) is to discharge the gate capacitance quickly in the event of an input short. The body diode of Q2 creates the necessary pumping action to keep the gate of Q1 low during normal operation. Note that Q1 and Q2 have a VGS spec limit of 20V. This restricts VIN to a maximum of 20V. For low dropout operation with VIN > 20V consult factory. Optional Connection of Input Diode and Current Sense Resistor D3 ADAPTER IN CLP + LT1511 C1 1µF CLN L1 RS4 TO SYSTEM POWER VCC SW + PARASITIC INTERNAL DIODE The circuit in Figure 12b allows system power to go to 0V without drawing battery current by adding an additional diode, D4. To ensure proper operation, the LT1511 current sense amplifier inputs (CLP and CLN) were designed to work above VCC and not to draw current from VCC when the inputs are pulled to ground by a powered-down adapter. Layout Considerations The typical application shown in Figure 1 on the first page of this data sheet shows a single diode to isolate the VCC pin from the adapter input. This simple connection may be unacceptable in situations where the main system power must be disconnected from both the battery and the adapter under some conditons. In particular, if the adapter is disconnected or turned off and it is desired to also R7 500Ω disconnect the system load from the battery, the system will remain powered through the parasitic diode from the SW pin to the VCC pin. CIN RS1 1511 F12a Switch rise and fall times are under 10ns for maximum efficiency. To prevent radiation, the catch diode, SW pin and input bypass capacitor leads should be kept as short as possible. A ground plane should be used under the switching circuitry to prevent interplane coupling and to act as a thermal spreading path. All ground pins should be connected to expanded traces for low thermal resistance. The fast-switching high current ground path, including the switch, catch diode and input capacitor, should be kept very short. Catch diode and input capacitor should be close to the chip and terminated to the same point. This path contains nanosecond rise and fall times with several amps of current. The other paths contain only DC and/or 200kHz tri-wave and are less critical. Figure 13 indicates the high speed, high current switching path. Figure 14 shows critical path layout. Contact Linear Technology for an actual LT1511 circuit PCB layout or Gerber file. Figure 12a. Standard Connection SWITCH NODE L1 R7 500Ω CLP LT1511 SW L1 + CLN VCC PARASITIC INTERNAL DIODE C1 1µF + CIN VBAT ADAPTER IN RS4 VIN D3 D4 TO SYSTEM POWER CIN HIGH FREQUENCY CIRCULATING PATH D1 COUT BAT LT1511 • F13 RS1 1511 F12b Figure 13. High Speed Switching Path Figure 12b. Modified Input Diode Connection Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 15 LT1511 U U W U APPLICATIONS INFORMATION GND D1 GND SW GND GND GND GND VCC1 CIN CIN GND L1 GND TO GND TO GND RS1 COUT GND NOTE: CONNECT ALL GND PINS TO EXPANDED PC LANDS FOR PROPER HEAT SINKING LT1511 • F14 Figure 14. Critical Electrical and Thermal Path Layout U PACKAGE DESCRIPTION Dimensions in inches (millimeters) unless otherwise noted. SW Package 24-Lead Plastic Small Outline (Wide 0.300) (LTC DWG # 05-08-1620) 0.598 – 0.614* (15.190 – 15.600) 0.291 – 0.299** (7.391 – 7.595) 0.010 – 0.029 × 45° (0.254 – 0.737) 0.037 – 0.045 (0.940 – 1.143) 0.093 – 0.104 (2.362 – 2.642) 24 23 22 21 20 19 18 17 16 15 14 13 0° – 8° TYP 0.009 – 0.013 (0.229 – 0.330) NOTE 1 0.050 (1.270) TYP 0.394 – 0.419 (10.007 – 10.643) NOTE 1 0.004 – 0.012 (0.102 – 0.305) 0.014 – 0.019 0.016 – 0.050 (0.356 – 0.482) (0.406 – 1.270) TYP NOTE: 1. PIN 1 IDENT, NOTCH ON TOP AND CAVITIES ON THE BOTTOM OF PACKAGES ARE THE MANUFACTURING OPTIONS. THE PART MAY BE SUPPLIED WITH OR WITHOUT ANY OF THE OPTIONS *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE 1 2 3 4 5 6 7 8 9 10 11 12 S24 (WIDE) 0996 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC 1325 Microprocessor-Controlled Battery Management System Can Charge, Discharge and Gas Gauge NiCd and Lead-Acid Batteries with Software Charging Profiles LT1372/LT1377 500kHz/1MHz Step-Up Switching Regulators High Frequency, Small Inductor, High Efficiency Switchers, 1.5A Switch LT1376 500kHz Step-Down Switching Regulator High Frequency, Small Inductor, High Efficiency Switcher, 1.5A Switch LT1505 High Current, High Efficiency Battery Charger 94% Efficiency, Synchronous Current Mode PWM LT1510 Constant-Voltage/Constant-Current Battery Charger Up to 1.5A Charge Current for Lithium-Ion, NiCd and NiMH Batteries LT1512 SEPIC Battery Charger VIN Can Be Higher or Lower Than Battery Voltage LT1769 Constant-Voltage/Constant-Current Battery Charger Up to 2A Charge Current for Lithium-Ion, NiCd and NiMH Batteries ® 16 Linear Technology Corporation 1511fb LT/TP 0399 REV B 2K • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com LINEAR TECHNOLOGY CORPORATION 1995