TS615 DUAL WIDE BAND OPERATIONAL AMPLIFIER WITH HIGH OUTPUT CURRENT ■ LOW NOISE : 2.5nV/√Hz ■ HIGH OUTPUT CURRENT : 420mA ■ VERY LOW HARMONIC AND INTERMODULATION DISTORTION ■ HIGH SLEW RATE : 410V/µs ■ -3dB BANDWIDTH : 40MHz@gain=12dB on 25Ω load single ended. ■ 21.2Vp-p DIFFERENTIAL OUTPUT SWING on 50Ω load, 12V power supply ■ CURRENT FEEDBACK STRUCTURE ■ 5V to 12V POWER SUPPLY ■ SPECIFIED FOR 20Ω and 50Ω DIFFEREN- P TSSOP14 Exposed-Pad (Plastic Micro package) TIAL LOAD ■ POWER DOWN FUNCTION WITH A SHORT CIRCUITED OUTPUT to keep the matching with the line in sleep mode DESCRIPTION The TS615 is a dual operational amplifier featuring a high output current 410mA. These drivers can be configured differentially for driving signals in telecommunication systems using multiple carriers. The TS615 is ideally suited for xDSL (High Speed Asymmetrical Digital Subscriber Line) applications. This circuit is capable of driving a 10Ω or 25Ω load at ±2.5V, 5V, ±6V or +12V power supply. The TS615 will be able to reach a -3dB bandwidth of 40MHz on 25Ω load with a 12dB gain. This device is designed for the high slew rates to support low harmonic distortion and intermodulation. The TS615 is fitted out with Power Down function to decrease the consumption. During this sleep state the device displays a short circuit output in order to keep the impedance matching with the line. The TS615 is housed in TSSOP14 Exposed-Pad plastic package for a very low thermal resistance. APPLICATION ORDER CODE Part Number Temperature Range Package -40, +85°C PW TS615IPWT PW= Thin Shrink Small Outline Package with Exposed-Pad (TSSOP Exposed-Pad) only available in Tape & Reel (PWT) PIN CONNECTIONS (top view) -VCC1 1 14 -VCC2 Output1 2 +VCC1 3 13 Output2 + - - + 12 +VCC2 11 Non Inverting Input2 Non Inverting Input1 4 Inverting Input1 5 10 Inverting Input2 PowerDown 6 9 NC NC 7 8 NC Top View Cross Section View Showing Exposed-Pad This pad can be connected to a (-Vcc) copper area on the PCB ■ Line driver for xDSL ■ Multiple Video Line Driver December 2002 1/27 TS615 ABSOLUTE MAXIMUM RATINGS Symbol Parameter VCC Supply voltage 1) Vid Differential Input Voltage 2) Vin 3) Value Unit ±7 V ±2 V ±6 V Toper Operating Free Air Temperature Range -40 to + 85 °C Tstd Storage Temperature -65 to +150 °C Tj Input Voltage Range Maximum Junction Temperature 150 °C 4 °C/W Thermal Resistance Junction to Ambient Area 40 °C/W Maximum Power Dissipation (@25°C) 3.1 W CDM : Charged Device Model 1.5 kV 2 kV 200 V 1 kV 1 kV 100 V Rthjc Thermal Resistance Junction to Case Rthja Pmax. ESD except pins 4, 5, 10, 11 ESD HBM : Human Body Model MM : Machine Model CDM : Charged Device Model only pins 4, HBM : Human Body Model 5, 10, 11 MM : Machine Model 4) Output Short Circuit 1. 2. 3. 4. All voltage values, except differential voltage are with respect to network terminal. Differential voltage are non-inverting input terminal with respect to the inverting input terminal. The magnitude of input and output voltage must never exceed VCC +0.3V. An output current limitation protects the circuit from transient currents. Short-circuits can cause excessive heating. Destructive dissipation can result from short circuit on amplifiers. OPERATING CONDITIONS Symbol VCC Vicm Parameter Value Unit ±2.5 to ±6 V -VCC+1.5V to +VCC-1.5V V Power Supply Voltage Common Mode Input Voltage TYPICAL APPLICATION: Differential Line Driver for xDSL Applications 11 12 + +Vcc 1/2TS615 10 _ 14 12.5Ω 13 -Vcc Vi Vo R2 1:2 R1 25Ω GND R4 R3 Vi 5 4 3 _ 6 Pw-Dwn 2/27 +Vcc 1/2TS615 + 1 Vo -Vcc 2 12.5Ω 100Ω TS615 ELECTRICAL CHARACTERISTICS VCC = ±6Volts, Rfb=910Ω,Tamb = 25°C (unless otherwise specified) Note: as described on page 24 (table 71), the TS615 requires a 620Ω feedback resistor for an optimised bandwidth with a gain of 12B for a 12V power supply. Nevertheless, due to production test constraints, the TS615 is tested with the same feedback resistor for 12V and 5V power supplies (910Ω). Symbol Parameter Test Condition Min. Typ. Max. 1.25 2.1 3.5 Unit DC PERFORMANCE Vio ∆Vio Input Offset Voltage Differential Input Offset Voltage Iib+ Positive Input Bias Current Iib- Negative Input Bias Current ZIN+ ZINCIN+ CMR SVR Input(+) Impedance Input(-) Impedance Input(+) Capacitance Common Mode Rejection Ratio 20 log (∆Vic/∆Vio) Supply Voltage Rejection Ratio Tamb Tmin. < Tamb < Tmax. Tamb = 25°C Tamb Tmin. < Tamb < Tmax. Tamb Tmin. < Tamb < Tmax. ∆Vic = ±4.5V Tmin. < Tamb < Tmax. ∆Vcc=±2.5V to ±6V Tmin. < Tamb < Tmax. 20 log (∆Vcc/∆Vio) ICC Total Supply Current per Operator No load DYNAMIC PERFORMANCE and OUTPUT CHARACTERISTIC Vout = 7Vp-p, RL = 25Ω ROL Open Loop Transimpedance Tmin. < Tamb. < Tmax. Small Signal Vout<20mVp -3dB Bandwidth AV = 12dB, RL = 25Ω Large Signal Vout=3Vp Full Power Bandwidth BW AV = 12dB, RL = 25Ω Small Signal Vout<20mVp Gain Flatness @ 0.1dB AV = 12dB, RL = 25Ω Vout = 6Vp-p, AV = 12dB, RL Tr Rise Time = 25Ω Vout = 6Vp-p, AV = 12dB, RL Tf Fall Time = 25Ω Vout = 6Vp-p, AV = 12dB, RL Ts Settling Time = 25Ω Vout = 6Vp-p, AV = 12dB, RL SR Slew Rate = 25Ω RL=25Ω Connected to GND VOH High Level Output Voltage RL=25Ω Connected to GND VOL Low Level Output Voltage Vout = -4Vp Output Sink Current Tmin. < Tamb < Tmax. Iout Vout = +4Vp Output Source Current Tmin. < Tamb < Tmax. 58 72 6 7.8 3 3.2 82 54 1 63 61 2.5 30 mV 15 µA 5 21 8.9 25 40 µA kΩ Ω pF dB 79 78 14 mV dB 17 mA MΩ MHz 26 7 MHz 10.6 ns 12.2 ns 50 ns 330 410 V/µs 4.8 5.1 -5.5 -530 -440 420 365 -350 330 -5.2 V V mA 3/27 TS615 Note: as described on page 24 (table 71), the TS615 requires a 620Ω feedback resistor for an optimised bandwidth with a gain of 12B for a 12V power supply. Nevertheless, due to production test constraints, the TS615 is tested with the same feedback resistor for 12V and 5V power supplies (910Ω). Symbol Parameter NOISE AND DISTORTION eN Equivalent Input Noise Voltage iNp Equivalent Input Noise Current (+) iNn Equivalent Input Noise Current (-) 2nd Harmonic distortion HD2 (differential configuration) HD3 3rd Harmonic distortion (differential configuration) IM2 2nd Order Intermodulation Product (differential configuration) IM3 4/27 3rd Order Intermodulation Product (differential configuration) Test Condition F = 100kHz F = 100kHz F = 100kHz Vout = 14Vp-p, AV = 12dB F= 110kHz, RL = 50Ω diff. Vout = 14Vp-p, AV = 12dB F= 110kHz, RL = 50Ω diff. F1= 100kHz, F2 = 110kHz Vout = 16Vp-p, AV = 12dB RL = 50Ω diff. F1= 370kHz, F2 = 400kHz Vout = 16Vp-p, AV = 12dB RL = 50Ω diff. F1 = 100kHz, F2 = 110kHz Vout = 16Vp-p, AV = 12dB RL = 50Ω diff. F1 = 370kHz, F2 = 400kHz Vout = 16Vp-p, AV = 12dB RL = 50Ω diff. Min. Typ. Max. Unit 2.5 15 21 nV/√Hz pA/√Hz pA/√Hz -87 dBc -83 dBc -76 dBc -75 -88 dBc -87 TS615 ELECTRICAL CHARACTERISTICS VCC = ±2.5Volts, Rfb=910Ω,Tamb = 25°C (unless otherwise specified) Symbol Parameter Test Condition Min. Typ. Max. Tamb 0.5 2.5 Tmin. < Tamb < Tmax. 1.2 Unit DC PERFORMANCE Vio ∆Vio Input Offset Voltage Differential Input Offset Voltage Iib+ Positive Input Bias Current Iib- Negative Input Bias Current Tamb = 25°C Tamb 5 Tmin. < Tamb < Tmax. 8 Tamb 0.8 Tmin. < Tamb < Tmax. 1.24 mV 2.5 mV 30 µA 11 µA ZIN+ Input(+) Impedance 71 ZIN- Input(-) Impedance 62 Ω CIN+ Input(+) Capacitance Common Mode Rejection Ratio 1.5 pF ∆Vic = ±1V 20 log (∆Vic/∆Vio) Tmin. < Tamb. < Tmax. Supply Voltage Rejection Ratio ∆Vcc=±2V to ±2.5V 20 log (∆Vcc/∆Vio) Tmin. < Tamb. < Tmax. CMR SVR 55 60 dB 58 63 77 dB 76 ICC Total Supply Current per Operator No load DYNAMIC PERFORMANCE and OUTPUT CHARACTERISTICS Vout = 2Vp-p, RL = 10Ω ROL Open Loop Transimpedance Tmin. < Tamb. < Tmax. kΩ 11.9 2 15 5.4 mA MΩ 2.1 -3dB Bandwidth Small Signal Vout<20mVp AV = 12dB, RL = 10Ω Full Power Bandwidth Large Signal Vout = 1.4Vp AV = 12dB, RL = 10Ω 20 Gain Flatness @ 0.1dB Small Signal Vout<20mVp AV = 12dB, RL = 10Ω 5.7 MHz Tr Rise Time Vout = 2.8Vp-p, AV = 12dB RL = 10Ω 11 ns Tf Fall Time Vout = 2.8Vp-p, AV = 12dB RL = 10Ω 11.5 ns Ts Settling Time Vout = 2.2Vp-p, AV = 12dB RL = 10Ω 39 ns SR Slew Rate Vout = 2.2Vp-p, AV = 12dB RL = 10Ω 100 130 V/µs VOH High Level Output Voltage RL=10Ω Connected to GND 1.5 1.75 V Low Level Output Voltage RL=10Ω Connected to GND BW VOL Iout Output Sink Current Output Source Current Vout = -1.25Vp 20 MHz -2.05 -350 Tmin. < Tamb < Tmax. Vout = +1.25Vp Tmin. < Tamb < Tmax. 30 -450 200 -1.8 V -470 270 mA 245 5/27 TS615 Symbol Parameter Test Condition NOISE AND DISTORTION eN Equivalent Input Noise Voltage iNp Equivalent Input Noise Current (+) iNn Equivalent Input Noise Current (-) Min. Typ. HD2 2nd Harmonic distortion (differential configuration) F = 100kHz F = 100kHz F = 100kHz Vout = 6Vp-p, AV = 12dB F= 110kHz, RL = 20Ω diff. HD3 3rd Harmonic distortion (differential configuration) Vout = 6Vp-p, AV = 12dB F= 110kHz, RL = 20Ω diff. -98 F1= 100kHz, F2 = 110kHz Vout = 6Vp-p, AV = 12dB RL = 20Ω diff. -86 F1= 370kHz, F2 = 400kHz Vout = 6Vp-p, AV = 12dB RL = 20Ω diff. -88 F1 = 100kHz, F2 = 110kHz Vout = 6Vp-p, AV = 12dB RL = 20Ω diff. -90 F1 = 370kHz, F2 = 400kHz Vout = 6Vp-p, AV = 12dB RL = 20Ω diff. -85 IM2 IM3 2nd Order Intermodulation Product (differential configuration) 3rd Order Intermodulation Product (differential configuration) Max. Unit 2.5 15 21 nV/√Hz pA/√Hz pA/√Hz -97 dBc dBc dBc dBc POWER DOWN MODE FEATURES (The Power Down command is a MOS input featuring a high input impedance) VCC = ±2.5Volts, 5Volts, ±6Volts or 12Volts, Tamb = 25°C Symbol Parameter Min. Typ. Max. Unit -VCC+0.8 V Pin (6) Threshold Voltage for Power Down Mode Vpdw Iccpdw Rpdw Cpdw Low Level -VCC High Level -VCC+2 69 80 µA Power Down Mode Total Current Consumption@ VCC=12V 148 180 µA Power Down Mode Output Impedance @ VCC=5V 19 23 Ω Power Down Mode Output Impedance @ VCC=12V 15.3 19 Power Down Mode Output Capacitance 63 POWER DOWN CONTROL CIRCUIT STATUS Vpdw=Low Level Active Vpdw=High Level Standby 6/27 +VCC Power Down Mode Total Current Consumption@ VCC=5V Ω pF TS615 Figure 1 : Load Configuration Figure 4 : Load Configuration Load: RL=25Ω, VCC=±6V Load: RL=10Ω, VCC=±2.5V +6V + TS615 _ 50Ω cable 49.9Ω TS615 25Ω 33Ω 1W -6V +2.5V + 50Ω cable 10Ω _ 50Ω 49.9Ω 50Ω 11Ω 0.5W -2.5V Figure 2 : Closed Loop Gain vs. Frequency Figure 5 : Closed Loop Gain vs. Frequency AV=+1 AV=-1 2 2 20 0 -160 (Vcc=±2.5V) -2 0 -4 phase (Vcc=±6V) -180 (Vcc=±2.5V) -40 -8 (Vcc=±6V) -10 -60 -12 -80 (Vcc=±2.5V, Rfb=1.1kΩ, Rload=10Ω) (Vcc=±6V, Rfb=750Ω, Rload=25Ω) -14 -200 (Vcc=±2.5V) -6 -220 -8 (Vcc=±6V) -240 -10 -12 -260 (Vcc=±2.5V, Rfb=1kΩ, Rin=1kΩ, Rload=10Ω) (Vcc=±6V, Rfb=680Ω, Rin=680Ω, Rload=25Ω) -14 -100 -16 -280 -16 -120 100 1k 10k 100k 1M 10M -300 100M 100 1k Frequency (Hz) 10k 100k 1M 10M 100M Frequency (Hz) Figure 3 : Closed Loop Gain vs. Frequency Figure 6 : Closed Loop Gain vs. Frequency AV=+2 AV=-2 8 40 8 (Vcc=±6V) gain 6 20 (Vcc=±2.5V) phase 4 -140 gain 6 -160 0 (Vcc=±2.5V) phase 4 2 (Vcc=±6V) -180 -40 -2 (Vcc=±6V) -60 -4 -6 -80 -8 -100 -10 -200 (Vcc=±2.5V) 0 -220 -2 (Vcc=±6V) Phase (°) -20 (Vcc=±2.5V) 0 (gain (dB)) 2 Phase (°) (gain (dB)) Phase (°) -20 -6 (gain (dB)) -4 Phase (°) (gain (dB) -140 gain (Vcc=±2.5V) phase -2 40 (Vcc=±6V) gain 0 -240 -4 -6 -260 (Vcc=±2.5V, Rfb=1kΩ, Rin=510Ω, Rload=10Ω) (Vcc=±6V, Rfb=680Ω, Rin=750//620Ω, Rload=25Ω) -8 -280 -10 -120 100 1k 10k 100k 1M Frequency (Hz) 10M 100M -300 100 1k 10k 100k 1M 10M 100M Frequency (Hz) 7/27 TS615 Figure 7 : Closed Loop Gain vs. Frequency Figure 10 : Closed Loop Gain vs. Frequency AV=+4 AV=-4 14 14 40 -140 gain gain 12 12 20 phase 10 (Vcc=±6V) phase 0 (Vcc=±6V) -180 -40 4 (Vcc=±6V) 2 -60 0 -80 (Vcc=±2.5V, Rfb=910Ω, Rg=300Ω, Rload=10Ω) (Vcc=±6V, Rfb=620Ω, Rg=560//330Ω, Rload=25Ω) -2 (gain (dB)) Phase (°) -20 (Vcc=±2.5V) 6 -200 (Vcc=±2.5V) 6 -220 4 (Vcc=±6V) 2 -240 0 -260 (Vcc=±2.5V, Rfb=1kΩ, Rin=320//360Ω, Rload=10Ω) (Vcc=±6V, Rfb=620Ω, Rin=360//270Ω, Rload=25Ω) -2 -100 -280 -4 -4 -300 -120 100 1k 10k 100k 1M 10M 100 100M 1k 10k 100k 1M 10M 100M Frequency (Hz) Frequency (Hz) Figure 8 : Closed Loop Gain vs. Frequency Figure 11 : Closed Loop Gain vs. Frequency AV=+8 AV=-8 20 20 40 -140 gain gain 18 18 20 phase -160 (Vcc=±2.5V) (Vcc=±2.5V) 16 16 (Vcc=±6V) phase 0 -180 (Vcc=±6V) -40 10 (Vcc=±6V) -60 8 6 (gain (dB)) Phase (°) -20 (Vcc=±2.5V) 12 -220 10 (Vcc=±6V) -240 8 6 -80 (Vcc=±2.5V, Rfb=680Ω, Rg=240//160Ω, Rload=10Ω) (Vcc=±6V, Rfb=510Ω, Rg=270//100Ω, Rload=25Ω) 4 -200 (Vcc=±2.5V) 12 Phase (°) 14 14 (gain (dB)) Phase (°) 8 8 (gain (dB)) -160 (Vcc=±2.5V) (Vcc=±2.5V) 10 -260 (Vcc=±2.5V, Rfb=680Ω, Rin=160//180Ω, Rload=10Ω) (Vcc=±6V, Rfb=510Ω, Rin=150//110Ω, Rload=25Ω) 4 -100 -280 2 2 -300 -120 100 1k 10k 100k 1M 10M 100 100M 1k 10k 100k 1M 10M 100M Frequency (Hz) Frequency (Hz) Figure 9 : Bandwidth vs. Temperature Figure 12 : Positive Slew Rate AV=+4, Rfb=620Ω, VCC=±6V, RL=25Ω AV=+4, Rfb=910Ω 4 50 Vcc=±6V Load=25Ω 45 2 VOUT (V) Bw (MHz) 40 35 0 30 -2 Vcc=±2.5V Load=10Ω 25 20 -40 -20 0 20 40 Temperature (°C) 8/27 60 80 -4 0.0 10.0n 20.0n 30.0n Time (s) 40.0n 50.0n TS615 Figure 16 : Positive Slew Rate AV= - 4, Rfb=620Ω, VCC=±6V, R L=25Ω 2 4 1 2 VOUT (V) VOUT (V) Figure 13 : Positive Slew Rate AV=+4, Rfb=910Ω, VCC=±2.5V, RL=10Ω 0 -2 -1 -2 0.0 0 10.0n 20.0n 30.0n 40.0n -4 0.0 50.0n 10.0n Figure 14 : Negative Slew Rate 2 2 1 VOUT (V) VOUT (V) 4 0 -2 50.0n 40.0n 50.0n -1 10.0n 20.0n 30.0n 40.0n -2 0.0 50.0n 10.0n 20.0n 30.0n Time (s) Figure 15 : Negative Slew Rate AV=+4, Rfb=910Ω, VCC=±2.5V, RL=10Ω Figure 18 : Negative Slew Rate AV= - 4, Rfb=620Ω, VCC=±6V, RL=25Ω 2 4 1 2 VOUT (V) VOUT (V) 40.0n 0 Time (s) 0 -1 -2 0.0 30.0n Figure 17 : Positive Slew Rate AV= - 4, Rfb=910Ω, VCC=±2.5V, RL=10Ω AV=+4, Rfb=620Ω, VCC=±6V, RL=25Ω -4 0.0 20.0n Time (s) Time (s) 0 -2 10.0n 20.0n 30.0n Time (s) 40.0n 50.0n -4 0.0 10.0n 20.0n 30.0n 40.0n 50.0n Time (s) 9/27 TS615 Figure 19 : Negative Slew Rate AV= - 4, Rfb=910Ω, VCC=±2.5V, RL=10Ω Figure 22 : Input Voltage Noise Level AV=+92, Rfb=910Ω, Input+ connected to Gnd via 10Ω 2 5.0 Input Voltage Noise (nV/√Hz) VOUT (V) + 0 -2 0.0 10.0n 20.0n 30.0n 40.0n 4.5 _ 4.0 10Ω Output - 6V Ω 910 910Ω 3.5 3.0 2.5 2.0 100 50.0n + 6V 1k 10k Time (s) 100k 1M (Frequency (Hz) Figure 20 : Slew Rate vs. Temperature AV=+4, Rfb=910Ω, VCC=±2.5V, RL=10Ω Figure 23 : Transimpedance vs. Temperature Open Loop 30 200 25 150 Vcc=±6V 20 Positive SR 50 ROL (MΩ) Slew Rate (V/µs) 100 0 −50 Negative SR 15 10 −100 Vcc=±2.5V −150 −200 −40 5 −20 0 20 40 60 80 0 -40 Temperature (°C) -20 0 20 40 60 80 Temperature (°C) Figure 21 : Slew Rate vs. Temperature AV=+4, Rfb=910Ω, VCC=±6V, RL=25Ω Figure 24 : Icc vs. Power Supply Open loop, no load 16 600 14 500 12 400 10 8 Icc(+) 6 200 100 0 4 Positive&Negative SR Rfb=620Ω ICC (mA) Slew Rate (V/µs) 300 Positive&Negative SR Rfb=910Ω −100 2 0 -2 -4 −200 -6 −300 -10 −500 −600 −40 Icc(-) -8 −400 -12 -14 −20 0 20 40 Temperature (°C) 60 80 -16 5 6 7 8 VCC (V) 10/27 9 10 11 12 TS615 Figure 25 : Iib vs. Power Supply Figure 28 : Iib(+) vs. Temperature Open loop, no load Open loop, no load 7 8 IB+ 6 7 Vcc=±6V 6 5 IIB(+) (µA) IB (µA) 5 4 3 IB - 2 4 3 2 Vcc=±2.5V 1 1 0 0 5 6 7 8 9 10 11 12 -1 -40 -20 0 Vcc (V) 20 40 60 80 Temperature (°C) Figure 26 : Iib(-) vs. Temperature Figure 29 : Voh & Vol vs. Power Supply Open loop, no load Open loop, RL=25Ω 6 5 5 VOH 4 4 3 VOH & VOL (V) Vcc=±6V IIB(-) (µA) 3 2 2 1 0 VOL -1 -2 Vcc=±2.5V -3 1 -4 -5 0 -40 -6 -20 0 20 40 60 5 80 6 7 8 Temperature (°C) 9 10 11 12 Vcc (V) Figure 27 : Icc vs. Temperature Figure 30 : Voh vs. Temperature Open loop, no load Open loop 6 14 12 10 5 Icc(+) for Vcc=±2.5V 8 6 Icc(+) for Vcc=±6V 4 2 VOH (V) ICC (mA) 4 0 -2 -4 -6 -8 Vcc=±6vV Load=25Ω 3 2 Icc(-) for Vcc=±6V Icc(-) for Vcc=±2.5V 1 -10 Vcc=±2.5V Load=10Ω -12 -14 -40 -20 0 20 40 Temperature (°C) 60 80 0 -40 -20 0 20 40 60 80 Temperature (°C) 11/27 TS615 Figure 31 : Vol vs. Temperature Figure 34 : CMR vs. Temperature Open loop Open loop, no load 0 70 Vcc=±2.5V Load=10Ω -1 68 66 CMR (dB) -2 VOL (V) Vcc=±6V 64 -3 Vcc=±6V Load=25Ω -4 62 60 58 56 Vcc=±2.5V 54 -5 52 -6 -40 -20 0 20 40 60 50 -40 80 -20 0 Temperature (°C) 20 40 60 80 Temperature (°C) Figure 32 : Differential V io vs. Temperature Figure 35 : SVR vs. Temperature Open loop, no load Open loop, no load 450 84 400 82 350 SVR (dB) ∆VIO (µV) Vcc=±2.5V 300 Vcc=±6V 80 78 Vcc=±6V 250 76 200 -40 -20 0 20 40 60 80 -40 Vcc=±2.5V -20 0 Temperature (°C) 20 40 60 80 60 80 Temperature (°C) Figure 33 : Vio vs. Temperature Figure 36 : Iout vs. Temperature Open loop, no load Open loop, VCC=±6V, RL=10Ω 300 2.0 250 Vcc=±6V 200 150 1.5 100 Isource 1.0 Iout (mA) VIO (mV) 50 0.5 0 -50 -100 -150 -200 -250 Isink -300 0.0 -350 Vcc=±2.5V -0.5 -40 -20 0 20 -400 40 Temperature (°C) 12/27 60 80 -450 -40 -20 0 20 40 Temperature (°C) TS615 Figure 37 : Iout vs. Temperature Figure 40 : Isource vs. Output Amplitude. Open loop, VCC=±2.5V, RL=25Ω VCC=±2.5V, Open Loop, no Load 700 300 250 600 200 150 Isource 50 Iout (mA) Isource (mA) 100 0 -50 -100 -150 -200 -250 500 400 300 200 Isink -300 100 -350 -400 -450 -40 -20 0 20 40 60 0 0.0 80 0.5 1.0 Temperature (°C) 1.5 2.0 2.5 Vout (V) Figure 38 : Maximum Output Amplitude vs. Load Figure 41 : Isink vs. Output Amplitude AV=+4, Rfb=620Ω, VCC=±6V VCC=±6V, Open Loop, no Load 0 12 10 -100 Vcc=±6V Isink (mA) VOUT-MAX (VP-P) -200 8 6 -300 -400 4 -500 Vcc=±2.5V 2 -600 -700 0 0 50 100 150 -6 200 -5 -4 RLOAD (Ω) -3 -2 -1 Figure 42 : Isource vs. Output Amplitude VCC=±2.5V, Open Loop, no Load VCC=±6V, Open Loop, no Load 0 700 -100 600 -200 500 Isource (mA) Isink (mA) Figure 39 : Isink vs. Output Amplitude. -300 -400 400 300 -500 200 -600 100 -700 -2.5 0 Vout (V) 0 -2.0 -1.5 -1.0 Vout (V) -0.5 0.0 0 1 2 3 4 5 6 Vout (V) 13/27 TS615 Figure 44 : Group Delay No load, Open Loop VCC=±6V, VCC=±2.5V 200 100 150 90 100 80 Av=4 Vcc=±6V, Rfb=620Ω, Load=25Ω Vcc=±2.5V, Rfb=910Ω, Load=10Ω IF Bw = 10Hz Smoothing=19.247MHz on 10ns/div scale 70 50 Delay (ns) ICC pdw (µA) Figure 43 : Icc (Power Down) vs. Temperature Vcc=±6V 0 Vcc=±2.5V -50 60 50 40 -100 30 -150 -200 -40 20 -20 0 20 40 Temperature (°C) 14/27 60 80 10 300k 1M 10M Frequency (Hz) 50M TS615 2 n Vout = C 0 + C 1 V in + C 2 V in + …C n V in In this expression, we recognize the second order intermodulation IM2 by the frequencies (ω1-ω2) and (ω1+ω2) and the third order intermodulation IM3 by the frequencies (2ω1-ω2), (2ω1+ω2), (−ω1+2ω2) and (ω1+2ω2). due to a non-linearity in the input-output amplitude transfer. In the case of the input is Vin=Asinωt, C0 is the DC component, C1(Vin) is the fundamental, Cn is the amplitude of the harmonics of the output signal Vout. The measurement of the intermodulation product of the driver is achieved by using the driver as a mixer by a summing amplifier configuration. By this way, the non-linearity problem of an external mixing device is avoided. INTERMODULATION DISTORTION PRODUCT A non-ideal output of the amplifier can be described by the following development : A one-frequency (one-tone) input signal contributes to a harmonic distortion. A two-tones input signal contributes to a harmonic distortion and intermodulation product. This intermodulation product or intermodulation distortion study of a two-tones input signal is the first step of the amplifier characterization of driving capability in the case of a multi-tone signal. Figure 45 : Non-inverting Summing Amplifier 1kΩ 49.9Ω 11 Vin1 + +Vcc 1/2TS615 1:√2 49.9Ω 10 Rfb1 33Ω Rg1 Vin2 Vout diff. 1:√2 + C ( A sin ω t + B sin ω t ) 2 1 2 2 … + C ( A sin ω t + B sin ω t ) n 1 2 V in 49.9Ω 13 _ 400Ω 50Ω No rth Hills 0 315PB In this case : 1kΩ 400Ω 50Ω √2:1 100Ω 50Ω 33Ω Rg2 North Hills 0315PB Rfb2 No rth Hills 0 315PB 49.9Ω n = A sin ω t + B sin ω t 2 1 1kΩ _ 49.9Ω 1/2TS615 + -Vcc 1kΩ 49.9Ω V o ut = C 0 + C 1 ( A sin ω 1 t + B sin ω 2 t ) and : + C1 ( A sin ω 1 t + B sin ω 2 t ) C2 2 2 – ------- A cos 2ω 1 t + B cos 2ω 2 t 2 + 2 C2 AB ( cos ( ω 1 – ω 2 )t – cos ( ω 1 – ω 2 ) t ) The following graphs show the IM2 and the IM3 of the amplifier in different configuration. The two-tones input signal is achieved by the multisource generator Marconi 2026. Each tone has the same amplitude. The measurement is achieved by the spectrum analyzer HP3585A. C 3 + 3 ------- ¥ 4 3 3 + C A sin 3ω t + B sin 3ω t 1 2 3 2 3C A B 3 1 + ------------------------ sin ( 2ω 1 – ω 2 )t – --- sin ( 2ω 1 + ω )t 2 2 2 2 3C 3 A B 1 + ------------------------ sin ( – ω + 2ω ) t – --- sin ( ω 1 + 2ω )t 1 2 2 2 2 … + C n ( V in ) n A 2 + B2 V out = C 0 + C 2 --------------------- 2 A 3 sin ω t + B 3 sin ω t + 2A2 B sin ω t + 2AB 2 sin ω t 2 1 2 1 15/27 TS615 Figure 49 : Intermodulation vs. Load 370kHz & 400kHz, AV=+1.5, Rfb=1kΩ, RL=14Ω diff.,VCC=±2.5V 370kHz & 400kHz, AV=+1.5, Rfb=1kΩ, Vout=6.5Vpp,V CC=±2.5V -30 -30 -40 -40 IM3 340kHz, 430kHz, 1140kHz, 1170kHz -50 -50 IM2 30kHz IM2 770kHz -60 IM2 and IM3 (dBc) IM2 and IM3 (dBc) Figure 46 : Intermodulation vs. Output Amplitude IM3 340kHz, 430kHz -70 -80 -60 IM2 30kHz IM2 770kHz -70 -80 -90 -90 IM3 1140kHz, 1170kHz -100 -100 0 1 2 3 4 5 6 7 -110 8 0 20 40 60 80 100 120 140 160 180 200 Differential Load (Ω) Differential Output Voltage (Vp-p) Figure 50 : Intermodulation vs. Output Amplitude 370kHz & 400kHz, AV=+1.5, Rfb=1kΩ, RL=28Ω diff.,VCC=±2.5V 100kHz & 110kHz, AV=+4, Rfb=620Ω, RL=200Ω diff.,VCC=±6V -30 -30 -40 -40 -50 -50 -60 IM3 340kHz, 430kHz -70 IM2 and IM3 (dBc) IM2 and IM3 (dBc) Figure 47 : Intermodulation vs. Output Amplitude IM2 770kHz IM2 30kHz -80 IM3 90kHz, 120kHz -60 IM2 210kHz IM3 310kHz -70 IM3 320kHz -80 -90 -90 -100 IM3 1140kHz, 1170kHz -100 -110 0 1 2 3 4 5 6 7 8 2 4 Differential Output Voltage (Vp-p) 6 8 10 12 14 16 18 20 22 Differential Output Voltage (Vp-p) Figure 48 : Intermodulation vs. Gain Figure 51 : Intermodulation vs. Output Amplitude 370kHz & 400kHz, RL=20Ω diff., Vout=6Vpp, VCC=±2.5V 100kHz & 110kHz, AV=+4, Rfb=620Ω, RL=50Ω diff., VCC=±6V -30 -30 -40 -40 -60 IM2 30kHz IM2 770kHz -70 -80 -70 -80 -90 -100 -100 -110 1.5 2.0 2.5 3.0 Closed Loop Gain (Linear) 16/27 IM2 210kHz -60 -90 -110 1.0 IM3 90kHz, 120kHz, 310kHz, 320kHz -50 IM2 and IM3 (dBc) IM2 and IM3 (dBc) -50 IM3 340kHz, 430kHz, 1140kHz, 1170kHz 3.5 4.0 2 4 6 8 10 12 14 16 18 Differential Output Voltage (Vp-p) 20 22 TS615 Figure 52 : Intermodulation vs. Frequency Range Figure 54 : Intermodulation vs. Output Amplitude AV=+4, Rfb=620Ω, RL=50Ω diff., Vout=16Vpp, VCC=±6V 370kHz & 400kHz, AV=+4, Rfb=620Ω, RL=50Ω diff., VCC=±6V -30 -60 Quadratic Summation of all IM2 and IM3 components generated by each two-tones signal -65 IM3 1140kHz, 1170kHz -50 f1=100kHz f2=110kHz -75 f1=1MHz f2=1.1MHz f1=400kHz f2=430kHz f1=200kHz f2=230kHz -80 IM2 and IM3 (dBc) -70 (dB) IM2 30kHz -40 -85 -60 IM3 340kHz, 430kHz -70 -80 -90 -90 -95 -100 -100 100k IM2 770kHz -110 200k 300k 400k 500k 600k 700k 800k 900k 1M 1.1M 1M 0 2 4 6 8 10 12 14 16 18 20 22 Differential Output Voltage (Vp-p) Frequency (Hz) Figure 53 : Intermodulation vs. Output Amplitude 370kHz & 400kHz, AV=+4, Rfb=620Ω, RL=200Ω diff.,VCC=±6V -30 -40 IM2 and IM3 (dBc) -50 IM2 770kHz IM2 30kHz -60 IM3 1140kHz, 1170kHz -70 IM3 340kHz, 430kHz -80 -90 -100 -110 0 2 4 6 8 10 12 14 16 18 20 22 Differential Output Voltage (Vp-p) 17/27 TS615 PRINTED CIRCUIT BOARD LAYOUT CONSIDERATIONS Figure 55 : Exposed-Pad Package 1 The implementation of a proper ground plane in both sides of the PCB is mandatory to provide low inductance and low resistance common return. Most important for controlling the gain flatness and the bandwidth are stray capacitances at the output and inverting input. For minimizing the coupling, the space between signal lines and ground plane will be increased. Connections of the feedback components must be as short as possible in order to decrease the associated inductance which affect high frequency gain errors. It is very important to choose external components as small as possible such as surface mounted devices, SMD, in order to minimize the size of all the DC and AC connections. THERMAL INFORMATION The TS615 is housed in an Exposed-Pad plastic package. As described on the figure 56, this package uses a lead frame upon which the dice is mounted. This lead frame is exposed as a thermal pad on the underside of the package. The thermal contact is direct with the dice. This thermal path provide an excellent thermal performance. The thermal pad is electrically isolated from all pins in the package. It should be soldered to a copper area of the PCB underneath the package. Through these thermal paths within this copper area, heat can be conducted away from the package. In this case, the copper area should be connected to (-VCC). 18/27 DICE In this range of frequency, printed circuit board parasites can affect the closed-loop performance. Side View Bottom View DICE Cross Section View Figure 56 : Evaluation Board TS615 Figure 57 : Schematic Diagram J106 R107 1/2TS615 J110 R111 R112 R103 R118 2 _ R114 R111 R114 J110 5 R120 R116 R102 2 _ 5 R118 + R102 + 1/2 TS615 4 R107 J106 4 R120 R106 R116 R101 Non-Inverting Amplifier J105 J108 R109 Inverting Amplifier R115 10 R115 _ R119 R117 11 J111 _ 10 R109 J108 1/2TS615 R104 13 R121 R104 1/2TS615 + R119 13 + 11 J111 R121 R108 R117 J107 R110 R113 R105 R113 J109 Differential Amplifier 4 R107 + R118 2 Non-Inverting Summing Amplifier R107 _ 1/2TS615 R119 4 + 1/2TS615 13 R118 2 _ 5 J111 R114 J110 + R111 R121 R117 11 R106 R102 R115 10 J105 J106 R120 R112 R101 R111 R114 J110 R120 R116 R102 1/2TS615 _ 5 R116 J106 Power down J112 Differential Amplifier 4 R107 J106 100nF + 1/2TS615 5 4 J102 GND 5 J103 -Vcc C106 R114 R111 2 -Vcc +Vcc C107 R112 -Vcc Exposed-Pad 100uF C104 C103 100nF 100nF -Vcc R115 10 100nF 11 12 1/2TS615 3 11 + 14 C108 -Vcc + R119 13 J111 13 R110 J109 R113 2 _ R105 J104 _ 1/2TS615 +Vcc R121 _ 1 10 J110 6 + 1/2 TS615 1 R118 2 _ 3 J101 +Vcc R120 R122 +Vcc R117 100uF C101 C102 100nF C105 -Vcc +Vcc R116 Power Supply R102 R105 R113 R110 J109 100nF -Vcc 19/27 TS615 Figure 58 : Component Locations - Top Side Figure 60 : Top Side Board Layout Figure 59 : Component Locations - Bottom Side Figure 61 : Bottom Side Board Layout 20/27 TS615 NOISE MEASUREMENT 2 2 2 2 2 2 2 = eN × g + iNn × R2 + iNp × R3 × g R2 2 R2 2 … + ------- × 4kTR1 + 4kTR2 + 1 + ------- × 4kTR3, ( eq2 ) R1 R 1 eNo Figure 62 : Noise Model The input noise of the instrumentation must be extracted from the measured noise value. The real output noise value of the driver is: + iN+ R3 output TS615 HP3577 Input noise: 8nV/√Hz _ N3 eN iN- The input noise is called the Equivalent Input Noise as it is not directly measured but it is evaluated from the measurement of the output divided by the closed loop gain (eNo/g). After simplification of the fourth and the fifth term of (eq2) we obtain: R1 N1 eNo eN : input voltage noise of the amplifier iNn : negative input current noise of the amplifier iNp : positive input current noise of the amplifier The closed loop gain is : R fb A V = g = 1 + ---------R g 4 kTR2 R2 V1 = eN × 1 + ------- R1 R2 V3 = iNp × R3 × 1 + ------- R1 R2 V4 = – ------- × 4kTR1 R1 R2 V6 = 1 + ------- 4kTR3 R1 Assuming the thermal noise of a resistance R as: 4kTR ∆F with ∆F the specified bandwidth. On 1Hz bandwidth the thermal noise is reduced to 4kTR k is the Boltzmann’s constant equals 1,374.10-23J/°K. T is the temperature (°K). to The output noise eNo is calculated using the Superposition Theorem. But it is not the sum of all noise sources. The output noise is the square root of the sum of the square of each noise source. eNo = 2 2 2 2 2 2 2 2 2 2 2 = eN × g + iNn × R2 + iNp × R3 × g R2 2 … + g × 4kTR2 + 1 + ------- × 4kTR3, ( eq4 ) R1 Measurement of eN: We assume a short-circuit on the non-inverting input (R3=0). (eq4) comes: 2 2 2 2 eN × g + iNn × R2 + g × 4kTR2, ( eq5 ) In order to easily extract the value of eN, the resistance R2 will be chosen as low as possible. In the other hand, the gain must be large enough. R1=10Ω, R2=910Ω, R3=0, Gain=92 Equivalent Input Noise: 2.57nV/√Hz Input Voltage Noise: eN=2.5nV/√Hz V2 = iNn × R 2 2 2 eNo = The six noise sources are : V5 = 2 2 ( Measured ) – ( instrumentation ) , ( eq3 ) eNo = R2 N2 2 2 V1 + V2 + V3 + V4 + V5 + V6 ,( eq1 ) Measurement of iNn: R3=0 and the output noise equation is still the (eq5). This time the gain must be decreased to decrease the thermal noise contribution. R1=100Ω, R2=910Ω, R3=0, Gain=10.1 Equivalent Input Noise: 3.40nV/√Hz Negative Input Current Noise: iNn =21pA/√Hz Measurement of iNp: To extract iNp from (eq3), a resistance R3 is connected to the non-inverting input. The value of R3 must be chosen in order to keep its thermal noise contribution as low as possible against the iNp contribution. R1=100Ω, R2=910Ω, R3=100Ω, Gain=10.1 Equivalent Input Noise: 3.93nV/√Hz Positive Input Current Noise: iNp=15pA/√Hz Conditions: frequency=100kHz, VCC=±2.5V Instrumentation: Spectrum Analyzer HP3585A (input noise of the HP3585A: 8nV/√Hz) 21/27 TS615 POWER SUPPLY BYPASSING A proper power supply bypassing comes very important for optimizing the performance in high frequency range. Bypass capacitors should be placed as close as possible to the IC pins to improve high frequency bypassing. A capacitor greater than 1µF is necessary to minimize the distortion. For a better quality bypassing a capacitor of 10nF is added following the same condition of implementation. These bypass capacitors must be incorporated for the negative and the positive supply. Figure 63 : Circuit for Power Supply Bypassing +VCC 10µF + 10nF + TS615 - 10nF 10µF + -VCC SINGLE POWER SUPPLY The following figure show the case of a 5V single power supply configuration Figure 64 : Circuit for +5V single supply +5V + +5V Rin 1kΩ The amplifier must be biased with a mid supply (nominally +VCC/2), in order to maintain the DC component of the signal at this value. Several options are possible to provide this bias supply (such as a virtual ground using an operational amplifier), or a two-resistance divider which is the cheapest solution. A high resistance value is required to limit the current consumption. On the other hand, the current must be high enough to bias the non-inverting input of the amplifier. If we consider this bias current (30µA max.) as the 1% of the current through the resistance divider to keep a stable mid supply, two resistances of 2.2kΩ can be used in the case of a 12V power supply and two resistances of 820Ω can be used in the case of a 5V power supply. The input provides a high pass filter with a break frequency below 10Hz which is necessary to remove the original 0 volt DC component of the input signal, and to fix it at +VCC/2. CHANNEL SEPARATION - CROSSTALK The following figure show the crosstalk from an amplifier to a second amplifier. This phenomenon, accented in high frequencies, is unavoidable and intrinsic of the circuit. Nevertheless, the PCB layout has also an effect on the crosstalk level. Capacitive coupling between signal wires, distance between critical signal nodes, power supply bypassing, are the most significant points. Figure 65 : Crosstalk vs. Frequency 10µF IN necessary to assume a positive output dynamic range between 0V and +VCC supply rails. Considering the values of VOH and VOL, the amplifier will provide an output dynamic from +0.5V to 10.6V on 25Ω load for a 12V supplying, from 0.45V to 3.8V on 10Ω load for a 5V supplying. 100µF OUT ½ TS615 _ AV=+4, Rfb=620Ω, VCC=±6V, Vout=2Vp Rs -50 Rload R1 820Ω RG + 1µF 10nF + Rfb 910Ω CG CrossTalk (dB) R2 820Ω -60 -70 -80 -90 -100 -110 The TS615 operates from 12V down to 5V power supplies. This is achieved with a dual power supply of ±6V and ±2.5V or a single power supply of 12V and 5V referenced to the ground. In the case of this asymmetrical supplying, a new biasing is 22/27 -120 -130 10k 100k Frequency (Hz) 1M 10M TS615 Figure 68 : Standby Mode. Time On>Off POWER DOWN MODE BEHAVIOUR Figure 66 : Equivalent Schematic 0 Vcc + Enabled Output + 5 _ .. . A1 Vout -2 (Volts) .. 3 4 -1 POWER DOWN pin6 2 Ouput 1 -3 Rpdw -4 Vcc Vcc - -5 14 10 -Vcc .. . + _ A2 .. Vpdw -6 0 Rpdw 10 20 30 40 50 Time (µs) 13 Ouput 2 POWER DOWN pin6 12 Vcc + Please note that the short circuited output in power down mode is referenced to (-VCC). No problem appears when used in differential mode. Nevertheless, when used in single ended on a load referenced to GND, the (-VCC) level contributes to a current consumption through the load. As described on the Figure 68, the interest of featuring an output short circuit in power down mode is to keep the best impedance matching between the system and the twisted pair telephone line when the modem is in sleep mode. By this way, the modem can be waked-up with a signal from the line without any damage of this signal. This concept is particularly intended for the ADSL over voice modems, where the modem in sleep mode, must be waked-up by the phone call. Figure 69 : Standby Mode. Time Off>On 1 Vout 0 Disabled Output −1 (Volts) 11 Disabled Output -Vcc 1 Enabled Output −2 −3 −4 Vpdw −5 −6 0 1 2 3 4 5 Time (µs) Figure 70 : Standby Mode. Input/Output Isolation vs. Frequency AV=+4, Rfb=620Ω, VCC=±6V, Vout=3Vp Figure 67 : Matching in Sleep Mode 0 -10 Consumption=80µA -20 Matching 25Ω 5Ω max. 12.5Ω POWER DOWN Line (100Ω) Isolation (dB) 1:2 TS615 -30 Transformer 12.5Ω -40 -50 -60 -70 -80 -90 -100 -110 The system can be waked-up from the line -120 -130 10k 100k 1M 10M Frequency (Hz) 23/27 TS615 CHOICE OF THE FEEDBACK CIRCUIT ACTIVE FILTERING Table 71 : Closed-Loop Gain - Feedback Components Figure 73 : Low-Pass Active Filtering. Sallen-Key VCC (V) ±6 ±2.5 Gain Rfb (Ω) +1 +2 +4 +8 -1 -2 -4 -8 +1 +2 +4 +8 -1 -2 -4 -8 750 680 620 510 680 680 620 510 1.1k 1k 910 680 1k 1k 910 680 C1 R1 R2 + IN OUT C2 TS615 _ 25Ω RG INVERTING AMPLIFIER BIASING In this case a resistance is necessary to achieve a good input biasing, as R on (fig.30). This resistance is calculated by assuming the negative and positive input bias current. The aim is to make the compensation of the offset bias current which could affect the input offset voltage and the output DC component. Assuming Ib-, Ib+, Rin, Rfb and a zero volt output, the resistance R comes: R = Rin // R fb . Figure 72 : Compensation of the Input Bias Current Rfb 910Ω The resistors Rfb and R G give directly the gain of the filter as a classical non-inverting amplification configuration : R fb A V = g = 1 + ---------Rg Assuming the following expression as the response of the system: Vout jω g T jω = ------------------- = --------------------------------------------Vinjω 2 jω ( jω ) 1 + 2ζ ------- + -------------2 ωc ω c Rfb Ib- Rin _ the cutoff frequency is not gain dependent and it comes: Vcc+ Output TS615 + Vcc- Ib+ R Load 1 ω c = -------------------------------------R1R2C 1C2 the damping factor comes: 1 ζ = --- ω c ( C1 R 1 + C1 R 2 + C2 R 1 – C1 R 1 g ) 2 The higher the gain the more sensitive the damping factor is. When the gain is higher than 1 it is preferable to use some very stable resistors and capacitors values. In the case of R1=R2: Rfb 2C – C ---------2 1R g ζ = -----------------------------------2 C 1 C2 24/27 TS615 INCREASING THE LINE LEVEL BY USING AN ACTIVE IMPEDANCE MATCHING With a passive matching, the output signal amplitude of the driver must be twice the amplitude on the load. To go beyond this limitation an active matching impedance can be used. With this technique, it is possible to keep a good impedance matching with an amplitude on the load higher than the half of the output driver amplitude. This concept is shown in figure 74 for a differential line. Figure 74 : TS615 as a differential line driver with an active impedance matching 1µ + _ Vcc+ 1k Vo° 1:n Vo R3 1/2 R1 RL Vcc/2 1/2 R1 10µ Vi 1k Hybrid & Transformer 100Ω R5 100n GND + 2R2 R2 1 + ----------- + -------Vo ( noload ) R1 R3 G = --------------------------------- = -----------------------------------Vi R2 1 – -------R3 The gain, for the loaded system will be (eq1): GND R2 Vi 10n Rs1 As Vo° equals Vo without load, the gain in this case becomes : 2 R2 R2 1 + ----------- + -------Vo ( with load ) 1 R1 R3 GL = -------------------------------------- = --- ------------------------------------ ,( eq1 ) Vi 2 R2 1 – -------R3 100n Vcc+ 2Vi ( Vi – Vo° ) ( Vi + Vo ) ---------, --------------------------- and -----------------------R1 R2 R3 _ Vo Vo° R4 Vcc+ As shown in figure76, this system is an ideal generator with a synthesized impedance as the internal impedance of the system. From this, the output voltage becomes: Rs2 Vo = ( ViG ) – ( RoIout ) ,( eq2 ) GND 100n with Ro the synthesized impedance and Iout the output current. On the other hand Vo can be expressed as: Component Calculation Let us consider the equivalent circuit for a single ended configuration, Figure75. Figure 75 : Single ended equivalent circuit + 2R2 R2 Vi 1 + ----------- + -------- R1 R3 Rs1Iout Vo = ------------------------------------------------ – ----------------------- ,( eq3 ) R2 R2 1 – -------1 – -------R3 R3 By identification of both equations (eq2) and (eq3), the synthesized impedance is, with Rs1=Rs2=Rs: Rs1 Vi _ Rs Ro = ----------------- ,( eq4 ) R2 1 – -------R3 Vo° Vo R2 Figure 76 : Equivalent schematic. Ro is the synthesized impedance -1 R3 1/2R1 1/2RL Ro Vi.Gi Iout 1/2RL Let us consider the unloaded system. Assuming the currents through R1, R2 and R3 as respectively: 25/27 TS615 Unlike the level Vo° required for a passive impedance, Vo° will be smaller than 2Vo in our case. Let us write Vo°=kVo with k the matching factor varying between 1 and 2. Assuming that the current through R3 is negligible, it comes the following resistance divider: kV oRL Ro = -----------------------------RL + 2R s1 By fixing an arbitrary value of R2, (eq6) gives: R2 R3 = --------------------2Rs 1 – ----------RL Finally, the values of R2 and R3 allow us to extract R1 from (eq1), and it comes: 2R2 R1 = ----------------------------------------------------------- ,( eq7 ) R2 R2 2 1 – -------- GL – 1 – ------- R3 R3 After choosing the k factor, Rs will equal to 1/2RL(k-1). A good impedance matching assumes: 1 R o = --- RL ,( eq5 ) 2 with GL the required gain. Figure 77 : Components Calculation for Impedance Matching Implementation GL (gain for the loaded system) From (eq4) and (eq5) it becomes: R2 2Rs -------- = 1 – ----------- ,( eq6 ) R3 RL 26/27 R1 GL is fixed for the application requirements GL=Vo/Vi=0.5(1+2R2/R1+R2/R3)/(1-R2/R3) 2R2/[2(1-R2/R3)GL-1-R2/R3] R2 (=R4) Abritrary fixed R3 (=R5) Rs R2/(1-Rs/0.5RL) 0.5RL(k-1) Load viewed by each driver kRL/2 TS615 PACKAGE MECHANICAL DATA 14 PINS - THIN SHRINK SMALL OUTLINE PACKAGE (TSSOP Exposed-Pad) c k E1 L1 C SEATING PLANE L 0,25 mm GAUGE PLANE E A E2 A2 9 1 14 aaa C D D1 8 b e A1 PIN 1 IDENTIFICATION Millimeters Inches Dimensions Min. A A1 A2 b c D D1 E E1 E2 e L L1 k aaa 0.800 0.190 0.090 4.900 6.200 4.300 0.450 0d Typ. 1.000 5.000 3.000 6.400 4.400 3.000 0.650 0.600 1.000 Max. Min. 1.200 0.150 1.050 0.300 0.200 5.100 0.031 0.007 0.004 0.193 6.600 4.500 0.244 0.169 0.750 0.018 8d 0.100 0d Typ. 0.039 0.197 1.18 0.252 0.173 1.18 0.026 0.024 0.039 Max. 0.047 0.006 0.041 0.012 0.008 0.201 0.260 0.177 0.030 8d 0.004 Information furnished is believed to be accurate and reliable. However, STMicroelectronics assumes no responsibility for the consequences of use of such information nor for any infringement of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of STMicroelectronics. Specifications mentioned in this publication are subject to change without notice. This publication supersedes and replaces all information previously supplied. STMicroelectronics products are not authorized for use as critical components in life support devices or systems without express written approval of STMicroelectronics. © The ST logo is a registered trademark of STMicroelectronics © 2002 STMicroelectronics - Printed in Italy - All Rights Reserved STMicroelectronics GROUP OF COMPANIES Australia - Brazil - China - Finland - France - Germany - Hong Kong - India - Italy - Japan - Malaysia - Malta - Morocco Singapore - Spain - Sweden - Switzerland - United Kingdom © http://www.st.com 27/27