MIC2588/MIC2594 Micrel MIC2588/MIC2594 Single-Channel, Negative High-Voltage Hot Swap Power Controllers General Description Features The MIC2588 and the MIC2594 are single-channel, negative-voltage hot swap controllers designed to address the need for safe insertion and removal of circuit boards into “live” high-voltage system backplanes, while using very few external components. The MIC2588 and the MIC2594 are each available in an 8-pin SOIC package and work in conjunction with an external N-Channel MOSFET for which the gate drive is controlled to provide inrush current limiting and output voltage slew-rate control. Overcurrent fault protection is also provided and includes a programmable overcurrent threshold. During an output overload condition, a constantcurrent regulation loop is engaged to ensure that the system power supply maintains regulation. If a fault condition exceeds a built-in 400µs nuisance-trip delay, the MIC2588 and the MIC2594 will latch the circuit breaker’s output off and will remain in the off state until reset by cycling either the UV/OFF pin or the power to the IC. A master Power-Good signal is provided to indicate that the output voltage of the soft-start circuit is within its valid output range. This signal can be used to enable one or more DC-DC converter modules. All support documentation can be found on Micrel’s web site at www.micrel.com. • MIC2588: Pin-for-pin functional equivalent to the LT1640/LT1640A/LT4250 • Provides safe insertion and removal from live –48V (nominal) backplanes • Operates from –19V to –80V • Electronic circuit breaker function • Built-in 400µs “nuisance-trip” delay (tFLT) • Regulated maximum output current into faults • Programmable inrush current limiting • Fast response to short circuit conditions (< 1µs) • Programmable undervoltage and overvoltage lockouts (MIC2588-xBM) • Programmable UVLO hysteresis (MIC2594-xBM) • Fault reporting: Active-HIGH (-1BM) and Active-LOW (-2BM) Power-Good output signal Applications • • • • Central office switching –48V power distribution Distributed power systems Server Networks Typical Application -48V RTN (Long Pin) -48V RTN DC-DC CONVERTER 1 *D1 SMAT70A 100V -48V RTN (Short Pin) 2 R1 698kΩ 1% R3 12.4kΩ 1% C1 1uF R2 11.8kΩ 1% /PWRGD VDD OV DRAIN 8 7 C4 0.1uF 4 UV ON/OFF OUT- +2.5V RTN GATE SENSE VEE +2.5VOUT # IN- MIC2588-2BM 3 OUT+ IN+ C5 100uF 6 5 RFDBK 15kΩ *C2 22nF CFDBK 6.8nF 100V R4 10Ω C3 0.22uF VDD *C6 0.33uF -48VIN (Long Pin) -48VOUT RSENSE 0.01Ω 5% M1 SUM110N10-09 Nominal Undervoltage and Overvoltage Thresholds: VUV = 36.5V VOV = 71.2V * Optional components (See Applications Information for more details) # An external pull-up resistor for the power-good signal is necessary for DC-DC supplies (and all other load modules) not equipped with internal pull-up impedence Micrel, Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel + 1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com September 2005 1 M9999-083005 MIC2588/MIC2594 Micrel Ordering Information Part Number PWRGD Polarity Lockout Functions Circuit Breaker Function Package Standard Pb-Free MIC2588-1BM MIC2588-1YM Active-High Undervoltage and Overvoltage Latched Off 8-pin SOIC MIC2588-2BM MIC2588-2YM Active-Low Undervoltage and Overvoltage Latched Off 8-pin SOIC MIC2594-1BM MIC2594-1YM Active-High Programmable UVLO Hysteresis Latched Off 8-pin SOIC MIC2594-2BM MIC2594-2YM Active-Low Programmable UVLO Hysteresis Latched Off 8-pin SOIC Pin Configuration PWRGD 1 8 VDD /PWRGD 1 8 VDD OV 2 7 DRAIN OV 2 7 DRAIN UV 3 6 GATE UV 3 6 GATE VEE 4 5 SENSE VEE 4 8-Pin SOIC (M) MIC2588-1BM PWRGD 1 5 SENSE 8-Pin SOIC (M) MIC2588-2BM 8 VDD /PWRGD 1 8 VDD ON 2 7 DRAIN ON 2 7 DRAIN OFF 3 6 GATE OFF 3 6 GATE VEE 4 5 SENSE VEE 4 5 SENSE 8-Pin SOIC (M) MIC2594-1BM M9999-083005 8-Pin SOIC (M) MIC2594-2BM 2 September 2005 MIC2588/MIC2594 Micrel Pin Description Pin Number Pin Name Pin Function 1 PWRGD /PWRGD Power-Good Output: Open-drain. Asserted when the voltage on the DRAIN pin (VDRAIN) is within VPGTH of VEE, indicating that the output voltage is within proper specifications. 1 MIC25XX-1 PWRGD Active-High 1 MIC25XX-2 /PWRGD Active-Low MIC2588-1 and MIC2594-1: PWRGD will be high-impedance when VDRAIN is less than VPGTH, and will pull-down to VDRAIN when VDRAIN is greater than VPGTH. Asserted State: Open-Drain. MIC2588-2 and MIC2594-2: /PWRGD will pull-down to VDRAIN when VDRAIN is less than VPGTH, and will be high impedance when VDRAIN is greater than VPGTH. Asserted State: Active-Low. 2 OV Threshold 2 ON Turn-On Threshold 3 UV Threshold MIC2588: Undervoltage Threshold Input. When the voltage at the UV pin is less than the VUVL threshold, the GATE pin is immediately pulled low by an internal 100µA current pull-down. The UV pin is also used to cycle the device off and on to reset the circuit breaker. Taken together, the OV and UV pins form a window comparator which defines the limits of VEE within which the load may safely be powered. 3 OFF Turn-Off Threshold MIC2594: Turn-Off Threshold. When the voltage at the OFF pin is less than the VOFFL threshold, the GATE pin is immediately pulled low by an internal 100µA current pull-down. The OFF pin is also used to cycle the device off and on to reset the circuit breaker. Taken together, the ON and OFF pins provide programmable hysteresis for the turn-on command voltage. 4 VEE Negative Supply Voltage Input. Connect to the negative, or low side, terminal of the input power supply. 5 SENSE Circuit Breaker Sense Input: The current-limit threshold is set by connecting a resistor between this pin and VEE. When the current-limit threshold of IR = 50mV is exceeded for an internal delay tFLT (400µs), the circuit breaker is tripped and the GATE pin is immediately pulled low by IGATEOFF. Toggling the UV/OFF pin will reset the circuit breaker. To disable the circuit breaker, externally connect SENSE and VEE together. 6 GATE Gate Drive Output: Connect to the gate of an external N-Channel MOSFET. 7 DRAIN Drain Sense Input: Connect to the drain of an external N-Channel MOSFET. 8 VDD September 2005 MIC2588: Overvoltage Threshold Input. When the voltage at the OV pin is greater than the VOVH threshold, the GATE pin is immediately pulled low by an internal 100µA current pull-down. MIC2594: Turn-On Threshold. At initial system power-up or after the device has been shut off by the OFF pin, the voltage on the ON pin must exceed the VONH threshold in order for the MIC2594 to be enabled. Positive Supply Input. Connect to the positive, or high side, terminal of the input power supply. 3 M9999-083005 MIC2588/MIC2594 Micrel Absolute Maximum Ratings(1) Operating Ratings(2) (All voltages are referred to VEE) Supply Voltage (VDD – VEE) ...........................–0.3V to 100V DRAIN, PWRGD pins ....................................–0.3V to 100V GATE pin ......................................................–0.3V to 12.5V SENSE, OV, UV, ON, OFF pins.........................–0.3V to 6V Lead Temperature (soldering) Standard package (-xBM) (IR Reflow, Peak Temperature)........ 240°C +0°C/–5°C Pb-Free Package-(xYM) (IR Reflow, Peak Temperature)........ 260°C +0°C/–5°C ESD Ratings(3) Human Body Model ................................................... 2kV Machine Model ........................................................ 100V Supply Voltage (VDD – VEE) .......................... +19V to +80V Ambient Temperature Range (TA) ................ –40°C to 85°C Junction Temperature (TJ) ......................................... 125°C Package Thermal Resistance SOIC (θJA) ........................................................ 152°C/W DC Electrical Characteristics(4) VDD = 48V, VEE = 0V, TA = 25°C, unless otherwise noted. Bold indicates specifications apply over the full operating temperature range of –40°C to +85°C. Symbol Parameter VDD – VEE Supply Voltage IDD VTRIP Min Circuit Breaker Trip Voltage GATE Pin Pull-up Current IGATEOFF GATE Pin Sink Current VGATE GATE Drive Voltage, (VGATE – VEE) SENSE Pin Current Typ 19 Supply Current IGATEON ISENSE Condition Max Units 80 V 3 5 mA VTRIP = VSENSE – VEE 40 50 60 mV 30 45 60 µA (VSENSE – VEE) = 100mV VGATE = 2V 100 230 VGATE = VEE to 8V 19V ≤ (VDD – VEE) ≤ 80V 15V ≤ (VDD – VEE) ≤ 80V VSENSE = 50mV 9 10 mA 11 0.2 V µA UV Pin High Threshold Voltage Low-to-High transition 1.213 1.243 1.272 V VUVL UV Pin Low Threshold Voltage High-to-Low transition 1.198 1.223 1.247 V VOVH OV Pin High Threshold Voltage Low-to-High transition 1.198 1.223 1.247 OV Pin Low Threshold Voltage High-to-Low transition 1.165 1.203 1.232 VOVHYS OV Pin Hysteresis VUVH VUVHYS VOVL UV Pin Hysteresis 20 mV 20 V V mV VONH ANSI ON Pin High Threshold Voltage Low-to-High transition 1.198 1.223 1.247 V VOFFH ANSI OFF Pin Low Threshold Voltage High-to-Low transition 1.198 1.223 1.247 V ICNTRL Input Bias Current (OV, UV, ON, OFF Pins) VUV = 1.25V 0.5 µA VPGTH Power-Good Threshold 1.40 V VOLPG High-to-Low transition (VDRAIN – VEE) PWRGD Output Voltage VOLPG – VDRAIN (relative to voltage at the DRAIN pin) 0mA ≤ IPG(LOW) ≤ 1mA –0.25 0.8 V –0.25 0.8 V 1 µA MIC25XX-1 MIC25XX-2 ILKG(PG) PWRGD Output Leakage Current (VDRAIN – VEE) < VPGTH (VDRAIN – VEE) > VPGTH VPWRGD = VDD = 80V 1.1 1.26 Notes: 1. Exceeding the “Absolute Maximum Ratings” may damage the devices. 2. The devices are not guaranteed to function outside the specified operating conditions. 3. Devices are ESD sensitive. Handling precautions recommended. Human body model: 1.5kΩ in series with 100pF. Machine model: 200pF, no series resistance. 4. Specification for packaged product only. M9999-083005 4 September 2005 MIC2588/MIC2594 Micrel AC Electrical Characteristics(5) Symbol Parameter Condition tFLT Built-in Overcurrent Nuisance Trip Time Delay(6) (Figure 1) tOCSENSE Overcurrent Sense to GATE Low (Figure 2) tOVPHL OV to GATE Low(6) (Figure 3) High(6) Min Typ Max 400 VSENSE – VEE = 100mV µs 3.5 1 Units µs µs tOVPLH OV to GATE tUVPLH UV to GATE High(6) (Figure 4) 1 µs tPGL(1) DRAIN High to PWRGD Output Low(6) (-1 Version parts only) RPULLUP = 100kΩ, CLOAD on PWRGD = 50pF 1 µs tPGL(2) DRAIN Low to /PWRGD Output Low(6) (-2 Version parts only) RPULLUP = 100kΩ, CLOAD on /PWRGD = 50pF 1 µs tPGH(1) DRAIN Low to PWRGD Output High(6) (-1 Version parts only) RPULLUP = 100kΩ, CLOAD on PWRGD = 50pF 2 µs tPGH(2) DRAIN High to /PWRGD Output High(6) RPULLUP = 100kΩ, CLOAD on /PWRGD = 50pF (-2 Version parts only) 2 µs tUVPHL (Figure 3) 1 µs UV to GATE Low(6) (Figure 4) 1 µs Notes: 5. Specification for packaged product only. 6. Not 100% production tested. Parameters are guaranteed by design. Timing Diagrams OVERCURRENT EVENT t < tFLT ILIMIT t tFLT ILOAD 0A Load current is regulated at ILIMIT = 50mV/RSENSE VDRAIN Output OFF (at VDD) (at VEE) (at VEE) VGATE (VEE +10V) Reduction in VDRAIN to support ILIMIT = 50mV/RSENSE (at VEE) Figure 1. Overcurrent Response 100mV VSENSE - VEE tOCSENSE VGATE 1V Figure 2. SENSE to GATE LOW Timing Response September 2005 5 M9999-083005 MIC2588/MIC2594 Micrel 1.223V 1.203V VOV tOVPLH tOVPHL VGATE 1V 1V Figure 3. Overvoltage Response VUV 1.223V 1.243V tUVPHL tUVPLH VGATE 1V 1V Figure 4. Undervoltage Response MIC2588/94-1 VDRAIN VEE VPGTH VPGTH tPGL1 tPGH1 PWRGD not asserted PWRGD asserted - High Impedance VPWRGD Ð VDRAIN = 0V PWRGD not asserted VPWRGD Ð VDRAIN = 0V PWRGD VEE MIC2588/94-2 VDRAIN VEE VPGTH VPGTH tPGL2 tPGH2 /PWRGD VEE Figure 5. DRAIN to Power-Good Response September 2005 6 M9999-083005 MIC2588/MIC2594 Micrel Typical Characteristics [Section under construction] 6 SUPPLY CURRENT (mA) SUPPLY CURRENT (mA) 6 5 4 3 2 1 0 -40 -20 0 20 40 60 80 100 TEMPERATURE (°C) 5 1 25 35 45 55 65 75 SUPPLY VOLTAGE (V) 40 6 4 GATE Pull-Up Current vs. Temperature 1.26 1.25 V UVH 1.24 1.23 VUVL 1.22 1.21 OV PIN THRESHOLD (V) 1.27 1.2 -40 -20 0 20 40 60 80 100 TEMPERATURE (°C) 0.4 Power-Good Low Voltage vs. Temperature 30 20 200 150 100 0.25 0.2 0.15 0.1 0.05 0 -40 -20 0 20 40 60 80 100 TEMPERATURE (°C) M9999-083005 VONH (V) 0.3 50 0 -40 -20 0 20 40 60 80 100 TEMPERATURE (°C) 0 -40 -20 0 20 40 60 80 100 TEMPERATURE (°C) OV Pin Threshold vs. Temperature Power Good Threshold vs. Temperature 1.5 1.45 1.4 1.35 1.3 1.25 1.2 1.15 1.1 V OVH V OVL 1.05 1 -40 -20 0 20 40 60 80 100 TEMPERATURE (°C) 1.5 0.35 VOLPG (V) 250 1.5 1.45 1.4 1.35 1.3 1.25 1.2 1.15 1.1 POWER GOOD THRESHOLD (V) 80 UV Pin Threshold vs. Temperature 1.28 UV PIN THRESHOLD (V) 20 30 40 50 60 70 SUPPLY VOLTAGE (V) GATE Sink Current vs. Temperature 300 10 0 10 4 350 ON Pin Threshold vs. Temperature 1.5 1.45 1.45 1.4 1.35 1.3 1.4 1.35 1.3 1.25 1.2 1.15 1.1 1.05 1 -40 -20 0 20 40 60 80 100 TEMPERATURE (°C) 7 VPGTH+ VPGTH– 1.05 1 -40 -20 0 20 40 60 80 100 TEMPERATURE (°C) VOFFL (V) 2 6 0 -40 -20 0 20 40 60 80 100 TEMPERATURE (°C) 85 IGATEOFF (mA) 8 IGATEON (µA) VGATE (V) 50 8 2 A 10 vs. Temperature 10 2 60 T = 25°C A 3 GATE Drive (V GATE - VEE) 12 12 T = 25°C 4 0 15 vs. Supply Voltage GATE Drive (V GATE - VEE) Supply Current vs. Supply Voltage VGATE (V) Supply Current vs. Temperature OFF Pin Threshold vs. Temperature 1.25 1.2 1.15 1.1 1.05 1 -40 -20 0 20 40 60 80 100 TEMPERATURE (°C) September 2005 MIC2588/MIC2594 Micrel Circuit Breaker Trip Voltage vs. Temperature 60 58 VTRIP (mV) 56 54 52 50 48 46 44 42 40 -40 -20 0 20 40 60 80 100 TEMPERATURE (°C) Test Circuit -48V RTN (Long Pin) -48V RTN R5 47kΩ 1 R1 698kΩ 1% 2 R2 11.8kΩ 1% *D1 SMAT70A C1 1uF VDD OV DRAIN C5 100uF 8 C4 0.1uF 7 MIC2588-2BM 3 4 R3 12.4kΩ 1% /PWRGD UV GATE VEE SENSE 6 5 RFDBK 10kΩ CFDBK C2 0.22uF R4 10Ω C3 0.1uF -48VIN (Long Pin) RSENSE (see photos) M1 IRF540 -48VOUT ILOAD MIC2588/MIC2594 Test Circuit M9999-083005 8 September 2005 MIC2588/MIC2594 Micrel Functional Characteristics Hot-Plug Turn-On Response ILOAD (1A/div) ILOAD DRAIN /PWRGD (500mA/div) (20V/div) (20V/div) VGATE (10V/div) DRAIN VGATE (50V/div) (5V/div) Turn-On Response /PWRGD (50V/div) RSENSE = 0.05Ω CFDBK = 22nF RSENSE = 0.05Ω CFDBK = 22nF C3 = 0.47µF RFDBK = 1.2kΩ TIME (25ms/div.) Turn-On (Circuit Breaker Trip) Into Large CLOAD Overcurrent Response (Short Circuit) VGATE (5V/div) DRAIN (50V/div) TIME (5ms/div.) Short Circuit Applied VGATE (5V/div) VSENSE (50mV/div) Circuit Breaker Trips ILOAD (2A/div) ILOAD (2A/div) RSENSE = 0.01Ω C5 = 1500µF CFDBK = 6.8nF C3 = 0.22µF TIME (100µs/div.) M9999-083005 RSENSE = 0.01Ω CFDBK = 22nF C2 = C3 = 0.22µF TIME (100µs/div.) 9 September 2005 MIC2588/MIC2594 Micrel Functional Diagram VDD1 Internal VDD and Reference Generator VDD VDD1 45A VREF1 GATE SENSE + – VEE 100A Current Limit State 50mV VEE VDD1 VEE PWRGD Nuisance Trip Filter (400s) VEE /PWRGD Logic + Circuit Breaker – + UV VTH(UV/OV) EN VEE – OV + – Internal PG + 6V Clamp DRAIN VPGTH For Power Good circuitry only denotes -2 option MIC2588 Block Diagram September 2005 10 M9999-083005 MIC2588/MIC2594 Micrel Functional Description The maximum voltage on C3 at turn-on must be less than VTHRESHOLD of M1. 1. For a standard 10V enhancement N-Channel MOSFET, VTHRESHOLD is about 4.25V. 2. Choose 2V as the maximum voltage to avoid turnon transients. Hot Swap Insertion When circuit boards are inserted into systems carrying live supply voltages (“hot swapped”), high inrush currents often result due to the charging of bulk capacitance that resides across the circuit board’s supply pins. These current spikes can cause the system’s supply voltages to temporarily go out of regulation, causing data loss or system lock-up. In more extreme cases, the transients occurring during a hot swap event may cause permanent damage to connectors or on-board components. The MIC2588 and the MIC2594 are designed to address these issues by limiting the magnitude of the transient or inrush current during hot swap events. This is achieved by controlling the rate at which power is applied to the circuit board (di/dt and dv/dt management). Additionally, the MIC2588 and the MIC2594 incorporate input voltage supervisory functions and current limiting, thereby providing robust protection for both the system and the circuit board. (M1) × C3 = V VGS dt CLOAD dv M1DRAIN C 3 6.8nF 750pF CFDBK IGATEON (1) The presence of C3 and RFDBK prevent turn-on of the external pass device by limiting the hot swap current surges induced by AC coupled transients from the drain to the gate of M1 (i.e., CFDBK + CGD(M1)). An appropriate value for C3 may be determined using the formula for a capacitive voltage divider. M9999-083005 75V – 2V 0.275 F 2V For C3, the nearest standard 5% value is 0.22µF. While the value for RFDBK is not critical, it should be chosen to allow a maximum of a few milliamperes to flow in the gate-drain circuit of M1 during turn-on. While the final value for RFDBK is determined empirically, initial values between RFDBK = 15kΩ to 27kΩ for systems with a maximum value of VIN(max) = 75V are appropriate. Resistor R4, in series with the MOSFET's gate, minimizes the potential for parasitic high frequency oscillations from occurring in M1. While the exact value of R4 is not critical, commonly used values for R4 range from 10Ω to 33Ω. Power-Good (PWRGD or /PWRGD) Output For the MIC2588-1 and the MIC2594-1, the Power-Good output signal (PWRGD) will be high impedance when VDRAIN drops below VPGTH, and will pull down to VDRAIN when VDRAIN is above VPGTH. For the MIC2588-2 and the MIC2594-2, /PWRGD will pull down to the potential of the VDRAIN pin when VDRAIN drops below VPGTH, and will be high impedance when VDRAIN is above VPGTH. Hence, the -1 parts have an active-high PWRGD signal and the -2 parts have an active-low /PWRGD output. Either PWRGD or /PWRGD may be used as an enable signal for one or more dt CLOAD (2) 220F 60 A 6.6nF 2A where the nearest standard 5% value is 6.8nF. Substituting 6.8nF into Equation 2 from above yields: I IINRUSH CLOAD – GATEON C FDBK | I INRUSH | VGS(M1) CFDBK dt Relating the above to the maximum transient (or inrush) current charging the load capacitance upon hot swap or power-up involves an extension of the same formula: IINRUSH VIN(max) – VGS(M1) 220F 45 A 4.95nF 2A Good engineering practice suggests the use of the worstcase parameter values for IGATEON from the “DC Electrical Characteristics” section: CFDBK dv M1DRAIN – VGS(M1) CFDBK + CGD(M1) CFDBK where IGATE(+) = Gate Charging Current = IGATEON; IGATE(–) ≅ –IGATE(+), due to the extremely high transconductance values of power MOSFETs; and IGATE(–) CFDBK where VIN(max) = VDD – VEE(min). For example, we can determine appropriate capacitor values given a hot swap controller that is required to maintain the inrush current into a 220µF load capacitance at 2A maximum and an input supply voltage as high as VIN(max) = 75V. One of the suggested MOSFETs to be used with the MIC2588/ MIC2594 is an SUM110N10-09,a 100V D2PAK device which has a typical CGD of 750pF. Calculating a value for CFBDK using Equation 1 yields: IGATE(–) – IGATEON CFDBK IN(max) C3 = CFDBK + CGD(M1) × Start-Up Cycle When the input voltage to the controller is between the overvoltage and undervoltage thresholds (MIC2588) or is greater than VON (MIC2594), a start cycle is initiated to deliver power to the load. At this time, the GATE pin of the controller applies a constant charging current (IGATEON) to the gate of the external MOSFET (M1). CFDBK creates a Miller integrator out of the MOSFET circuit, which limits the slew-rate of the voltage at the drain of M1. The drain voltage rate-of-change (dv/dt) of M1 is: dv M1DRAIN VGS(M1) × [C3 + CFDBK + CGD(M1) ] = VIN(max) × CFDBK + CGD(M1) 11 September 2005 MIC2588/MIC2594 Micrel subsequent DC/DC converter modules or for other system uses as desired. When used as an enable signal, the time necessary for the PWRGD (or /PWRGD) signal to pull-up (when in high impedance state) will depend upon the load (RC) that is present on this output. Circuit Breaker Function The MIC2588 and the MIC2594 employ an electronic circuit breaker that protects the MOSFET and other system components against faults such as short circuits. The current limit threshold is set via an external resistor, RSENSE, connected between the VEE and SENSE pins and is determined by: ILIM VTRIP RSENSE VUV VUVL (typ) (4) R1+R2 +R3 (5) VOV VOVH (typ) R3 Given VUV, VOV, and any one resistor value, the remaining two resistor values can be found. A suggested value for R3 is that which will provide a minimum of 100µA of current through the voltage divider chain at VDD = VUV. This yields the following as a starting point: R3 (3) VOVH (typ) 100A 12.23k The closest standard 1% value for R3 = 12.4kΩ. Using Equations 3 and 4 above, solving for R2 and R1 yields: where VTRIP is the circuit breaker trip threshold specified in the Electrical Characteristics Table. An internal 400µs timer limits the length of time (tFLT) for which the circuit can draw current in excess of its programmed threshold before the circuit breaker is tripped. This short delay prevents nuisance tripping of the circuit breaker due to system transients while providing rapid protection against large-scale transient faults. Whenever the voltage across RSENSE exceeds 50mV, two things happen: 1. A constant-current regulation loop is engaged and is designed to hold the voltage across RSENSE equal to 50mV. This protects both the load and the MIC2588 circuit from excessively high currents. This loop will engage in less than 1µs from the time at which the overvoltage condition on RSENSE occurs. 2. The internal 400µs timer is started. If the 400µs timeout period expires, the circuit breaker trips and the GATE pin is immediately pulled low by an internal current pull-down. This operation turns off the MOSFET quickly and disconnects the input from the load. Undervoltage/Overvoltage Detection—MIC2588 The MIC2588 has “UV” and “OV” input pins. These pins can be used to detect input supply rail undervoltage and overvoltage conditions. Undervoltage lockout prevents energizing the load until the supply input is stable and within tolerance. In a similar fashion, overvoltage turn-off prevents damage to sensitive circuit components should the input voltage exceed normal operational limits. Each of these pins is internally connected to an analog comparator with 20mV of hysteresis. When the UV pin falls below its VUVL threshold or the OV pin is above its VOVH threshold, the GATE pin is immediately pulled low. The GATE pin will be held low until UV exceeds its VUVH threshold or OV drops below its VOVL threshold. The UV and OV circuits’ threshold trip points are programmed using the resistor divider R1, R2, and R3 as shown in the “Typical Application.” The equations to set the trip points are shown below. For the following example, the circuit’s nominal UV threshold is set to VUV = 37V and the nominal OV threshold is placed at VOV = 72V, values commonly used in Central Office power distribution applications. September 2005 R1+R2 +R3 R2 +R3 V R2 R3 OV – 1 VUV 72V R2 12.4k – 1 37V R2 11.729k The closest standard 1% value for R2 = 11.8kΩ. Next, the value for R1 is calculated: –1.223V V R1 R3 OV – R2 1.223V 72V – 1.223V R1 12.4k – 11.8kΩ 1.223V R1 705.808k The closest standard 1% value for R1 = 698kΩ. Using standard 1% resistor values, the circuit’s nominal UV and OV thresholds are: VUV = 36.5V VOV = 71.2V Good general engineering design practices must consider the tolerances associated with these parameters, including but not limited to, power supply tolerance, undervoltgae and overvoltage tolerances, and the tolerances of the external passive components. Programmable UVLO Hysteresis—MIC2594 The MIC2594 has user-programmable hysteresis by means of the ON and OFF pins. This allows setting the part to turn on at a voltage V1, and not turn off until a second voltage V2, where V2 < V1. This can significantly simplify dealing with source impedances in the supply bus while at the same time increasing the amount of available operating time from a loosely regulated power supply (for example, a battery supply). Similarly to the MIC2588, each of these pins is internally connected to an analog comparator with 20mV of hysteresis. The MIC2594 holds the output off until the voltage at the ON pin exceeds its VONH threshold value given in the “Electrical Characteristics” table. Once the output has been enabled by the ON pin, it will remain on until the voltage at 12 M9999-083005 MIC2588/MIC2594 Micrel the OFF pin falls below its VOFFL threshold value, or the part turns off due to a fault. Should either event occur, the GATE pin is immediately pulled low and will remain low until the ON pin once again exceeds its VONH threshold. The circuit’s turn-on and turn-off points are set using the resistor divider R1, R2, and R3 as shown in the “Typical Application.” The equations to establish the trip points are shown below. In the following example, the circuit’s nominal ON threshold is set to VON = 40V and the circuit’s nominal OFF threshold is VOFF = 35V. VON = VONH (typ) V R2 = R3 ON – 1 VOFF 40V R2 = 12.4k – 1 35V R2 = 1.771k The closest standard 1% value for R2 = 1.78kΩ. R1 R2 R3 VOFF = VOFFL (typ) R1= R3 R3 R1 R2 R3 R2 R3 R1= 12.4k Given VOFF, VON, and any one resistor value, the remaining two resistor values can be readily found. A suggested value for R3 is that which will provide a minimum of 100µA of current through the voltage divider chain at VDD = VOFF. This yields the following as a starting point: R3 = VOFFL (typ) 100A 1.223V 40V – 1.223V – 1.78kΩ 1.223V R1= 391.380k The closest standard 1% value for R1 = 392kΩ. Using standard 1% resistor values, the circuit’s nominal ON and OFF thresholds are: VON = 40.1V VOFF = 35V Good general engineering design practices must consider the tolerances associated with these parameters, including but not limited to, power supply tolerance, undervoltgae and overvoltage tolerances, and the tolerances of the external passive components. 12.23k The closest standard 1% value for R3 = 12.4kΩ. solving for R2 and R1 yields: M9999-083005 VON –1.223V – R2 13 September 2005 MIC2588/MIC2594 Micrel Applications Information is connected to the DRAIN pin of the MIC2588/ MIC2594, and the resulting transient does have enough voltage and energy to damage this, or any, high-voltage hot swap controller. 2. If the load’s bypass capacitance (for example, the input filter capacitors for DC-DC converter module(s)) are on a board from which the board with the MIC2588/MIC2594 and the MOSFET can be unplugged, the same type of inductive transient damage can occur to the MIC2588/MIC2594. For many applications, the use of additional circuit components can be implemented for optimum system performance and/or protection. The circuit, shown in Figure 6, includes several components to address some the following system (dynamic) responses and/or functions: 1) suppression of transient voltage spikes, 2) elimination of false “tripping” of the circuit breaker due to undervoltage and overcurrent glitches, and 3) the implementation of an external reset circuit. It is not mandatory that these techniques be utilized, however, the application environment will dictate suitability. For protection against sudden on-card load dumps at the DRAIN pin of the MIC2588/MIC2594 controller, a 68V, 1W, 5% Zener diode clamp (D2) connected from the DRAIN to the VEE of the controller can be implemented, as shown. To protect the controller from large-scale transients at the card input, a 100V clamp diode (D1, SMAT70A or equivalent) can be used. In either case, very short lead lengths and compact layout design is strongly recommended to prevent unwanted transients in the protection circuitry. Power buss inductance often produces localized (plug-in card) high-voltage transients during a turn-off event. Managing these repeated voltage stresses with sufficient input bulk capacitance and/or transient suppressing diode clamps is highly recommended for maximizing the life of the hot swap controller(s). Optional External Circuits for Added Protection/Performance In many telecom applications, it is very common for circuit boards to encounter large-scale supply-voltage transients in backplane environments. Because backplanes present a complex impedance environment, these transients can be as high as 2.5 times steady-state levels, or 120V in worstcase situations. In addition, a sudden load dump anywhere on the circuit card can generate a very high voltage spike at the drain of the output MOSFET which, in turn, will appear at the DRAIN pin of the MIC2588/MIC2594. In both cases, it is good engineering practice to include protective measures to avoid damaging sensitive ICs or the hot swap controller from these large-scale transients. Two typical scenarios in which large-scale transients occur are described below: 1. An output current load dump with no bypass (charge bucket or bulk) capacitance to VEE. For example, if LLOAD = 5µH, VIN = 56V and tOFF = 0.7µs, the resulting peak short-circuit current prior to the MOSFET turning off would reach: 56V 0.7 s 7.8A 5 H If there is no other path for this current to take when the MOSFET turns off, it will avalanche the drain-source junction of the MOSFET. Since the total energy represented is small relative to the sturdiness of modern power MOSFETs, it’s unlikely that this will damage the transistor. However, the actual avalanche voltage is unknown; all that can be guaranteed is that it will be greater than the VBD(D-S) of the MOSFET. The drain of the transistor -48V RTN -48V RTN *R5 47kΩ R1 698kΩ 1% 1 2 *D1 100V C1 0.47µF System Reset R2 11.8kΩ 1% R3 12.4kΩ 1% -48VIN OV DRAIN C5 47uF 8 C4 0.1uF 7 3 UV GATE SENSE VEE 6 *D2 68V 5 RFDBK 10kΩ C2 0.22uF *R7 10kΩ VDD MIC2588-2BM 4 *M2 /PWRGD *C6 0.47uF *R6 2.7kΩ CFDBK 10nF 100V R4 10Ω C3 0.33uF RSENSE 0.01Ω 5% M1 SUM110N10-09 -48VOUT * Optional components (See Applications Information for more details) An SOT-363 is recommended for M2. D2 is a 68V, 1W Zener diode. Figure 6 Optional Components for Added Performance/Protection September 2005 14 M9999-083005 MIC2588/MIC2594 Micrel For systems that experience known load current surges exceeding the 400µs internal overcurrent filter (tFLT), the RC circuit consisting of R6 and C6 provides a means for additional overcurrent filtering to eliminate false “tripping” of the circuit breaker due to these transient load current surges. It is highly recommended to limit the increase of the overcurrent filter to approximately 2x the internal filter to allow the MOSFET to operate within its thermal specifications and SOA. R6 and C6 act as a low-pass filter to reduce the slew rate of the SENSE pin voltage. The SENSE pin current is nominally 200nA, resulting in a slight voltage drop across R6 that will combine in series with the voltage across RSENSE to produce an effective circuit breaker trip voltage of VTRIP – (R6 × ISENSE). The following equation can be used to select component values for a given overcurrent filter delay. ┌ VTRIP – V(tO) ┌ tOCDLY – (R × C) ln │ 1 – │ V(t) – V(tO) └ └ variations over time and temperature) and circuit breaker threshold voltages, a slightly more detailed calculation must be used to determine the minimum and maximum hot swap load currents. As the MIC2588’s minimum current limit threshold voltage is 40mV, the minimum hot swap load current is determined where the sense resistor is 3% high: IHOT_SWAP(min) Keep in mind that the minimum hot swap load current should be greater than the application circuit’s upper steady-state load current boundary. Once the lower value of RSENSE has been calculated, it is good practice to check the maximum hot swap load current (IHOT_SWAP(MAX)) that the circuit may let pass in the case of tolerance build-up in the opposite direction. Here, the worse case maximum is found using a VTRIP(MAX) threshold of 60mV and a sense resistor 3% low in value: (8) where VTRIP is the typical circuit breaker trip voltage specified in the electrical specifications, V(t0) is the voltage drop across the sense resister before the short or overcurrent condition occurs, and V(t) is the voltage drop across the sense resistor when the short or overcurrent is applied. The following example sets an overcurrent delay of 1ms for a 7.5A load current surge with a 2A steady-state load current and 5A current limit (RSENSE = 10mΩ). VTRIP = 50mV V(t0) = 2A × 10mΩ = 20mV V(t) = 7.5A × 10mΩ = 75mV Using Equation 8, for R6 = 2.7kΩ, C6 is 0.47µF. The capacitor (C2) connected from UV to reference (VEE) is used as a glitch filter for the input undervoltage monitor. C2 combines with the resistive network at the UV pin to form an RC time constant to slow the UV pin voltage fall time whenever the input voltage experiences a negative (magnitude) transient. During start-up, the UV rise time will also be affected by a longer RC time constant due to R1, therefore, the output start cycle will be delayed until the UV pin crosses its threshold. The circuit in Figure 6 consisting of M2, R7, R8, and a digital control signal, can be used to reset the controller after the GATE (and output) turns off. Once the output has been latched off, applying a low-high-low pulse on the GATE of M2 via the System Enable control can toggle the UV pin. System Enable is a user defined signal referenced to VEE. Sense Resistor Selection The sense resistor is nominally valued at: RSENSE(nom) VTRIP(typ) IHOT_SWAP(nom) IHOT_SWAP(max) 60mV 61.9mV 0.97 × RSENSE(nom) RSENSE(nom) In this case, the application circuit must be sturdy enough to operate up to approximately 1.5x the steady-state hot swap load current. For example, if an MIC2588 circuit must pass a minimum hot swap load current of 4A without nuisance trips, RSENSE should be set to: RSENSE(nom) 40mV 10mΩ 4A where the nearest 1% standard value is 10.0mΩ. At the other tolerance extremes, IHOT_SWAP(MAX) for the circuit in question is then simply: IHOT_SWAP(max) 61.9mV 6.19A 10mΩ With a knowledge of the application circuit’s maximum hot swap load current, the power dissipation rating of the sense resistor can be determined using P = I2 × R. Here, the current is IHOT_SWAP(max) = 6.19A and the resistance RSENSE(max) = (1.03)(RSENSE(nom)) = 10.3mΩ. Thus, the sense resistor’s maximum power dissipation is: PMAX = (6.19A)2 × (10.3mΩ) = 0.395W A 0.5W sense resistor is a good choice in this application. Power MOSFET Selection Selecting the proper external MOSFET for use with theMIC2588/MIC2594 involves three straightforward tasks: •Choice of a MOSFET which meets minimum voltage requirements. •Selection of a device to handle the maximum continuous current (steady-state thermal issues). •Verify the selected part’s ability to withstand any peak currents (transient thermal issues). Power MOSFET Operating Voltage Requirements The first voltage requirement for the MOSFET is easily stated: (9) where VTRIP(TYP) is the typical (or nominal) circuit breaker threshold voltage (50mV) and IHOT_SWAP(NOM) is the nominal load current level necessary to trip the internal circuit breaker. To accommodate worse-case tolerances in the sense resistor (for a ±1% initial tolerance, allow ±3% tolerance for September 2005 40mV 38.8mV 1.03 × RSENSE(nom) RSENSE(nom) 15 M9999-083005 MIC2588/MIC2594 Micrel the drain-source breakdown voltage of the MOSFET must be greater than VIN(MAX), or VDD – VEE(min). The second breakdown voltage criterion that must be met is the gate-source voltage. For the MIC2588/MIC2594, the gate of the external MOSFET is driven up to a maximum of 11V above VEE. This means that the external MOSFET must be chosen to have a gate-source breakdown voltage of 12V or more; 20V is recommended. Most power MOSFETs with a 20V gate-source voltage rating have a 30V drain-source breakdown rating or higher. For many 48V telecom applications, transient voltage spikes can approach, and sometimes exceed, 100V. The absolute maximum input voltage rating of the MIC2588/MIC2594 is 100V; therefore, a drain-source breakdown voltage of 100V is suggested for the external MOSFET. Additionally, an external input voltage clamp is strongly recommended for applications that do not utilize conditioned power supplies. Power MOSFET Steady-State Thermal Issues The selection of a MOSFET to meet the maximum continuous current is a fairly straightforward exercise. First, arm yourself with the following data: •The value of ILOAD(CONT, MAX.) for the output in question (see Sense Resistor Selection). •The manufacturer’s datasheet for the candidate MOSFET. •The maximum ambient temperature in which the device will be required to operate. •Any knowledge you can get about the heat sinking available to the device (e.g., can heat be dissipated into the ground plane or power plane, if using a surface-mount part? Is any airflow available?). The datasheet will almost always give a value of on resistance for a given MOSFET at a gate-source voltage of 4.5V and 10V. For MIC2588/MIC2594 applications, choose the gatesource ON resistance at 10V and call this value RON. Since a heavily enhanced MOSFET acts as an ohmic (resistive) device, almost all that’s required to determine steady-state power dissipation is to calculate I2R. The one addendum to this is that MOSFETs have a slight increase in RON with increasing die temperature. A good approximation for this value is 0.5% increase in RON per °C rise in junction temperature above the point at which RON was initially specified by the manufacturer. For instance, if the selected MOSFET has a calculated RON of 10mΩ at aTJ = 25°C, and the actual junction temperature ends up at 110°C, a good first cut at the operating value for RON would be: RON ≈ 10mΩ[1 + (110 – 25)(0.005)] ≈14.3mΩ The final step is to make sure that the heat sinking available to the MOSFET is capable of dissipating at least as much power (rated in °C/W) as that with which the MOSFET’s performance was specified by the manufacturer. Here are a few practical tips: 1. The heat from a TO-263 power MOSFET flows almost entirely out of the drain tab. If the drain tab can be soldered down to one square inch or more, the copper will act as the heat sink for the part. This copper must be on the same layer of the board as September 2005 the MOSFET drain. 2. Airflow works. Even a few LFM (linear feet perminute) of air will cool a MOSFET down substantially. If you can, position the MOSFET(s) near the inlet of a power supply’s fan, or the outlet of a processor’s cooling fan. 3. The best test of a candidate MOSFET for an application (assuming the above tips show it to be a likely fit) is an empirical one. Check the MOSFET’s temperature in the actual layout of the expected final circuit, at full operating current. The use of a thermocouple on the drain leads, or infrared pyrometer on the package, will then give a reasonable idea of the device’s junction temperature. Power MOSFET Transient Thermal Issues If the prospecitve MOSFET has been shown to withstand the environmental voltage stresses and the worse-case steadystate power dissipation is addressed, the remaining task is to verify if the MOSFET is capable of handling extreme overcurrent load faults, such as a short circuit, without overheating. A power MOSFET can handle a much higher pulsed power without damage than its continuos power dissipation ratings imply due to an inherent trait, thermal inertia. With respect to the specification and use of power MOSFETs, the parameter of interest is the “Transient Thermal Impedence”, or Zθ, which is a real number (variable factor) used as a multiplier of the thermal resistance (Rθ). The multiplier is determined using the given “Transient Thermal Imepedence Graph”, normalized to Rθ, that displays curves for the thermal impedence versus power pulse duration and duty cycle. The single-pulse curve is appropriate for most hot swap applications. Zθ is specified from junction-to-case for power MOSFETs typically used in telecom applications. The following example provides a method for estimating the peak junction temperature of a power MOSFET in determining if the MOSFET is suitable for a particular application. VIN (VDD – VEE) = 48V, ILIM = 4.2A, and the power MOSFET is SUM110N10-09 (TO-263 package) from Vishay-Siliconix. This MOSFET has an RON of 9.5mΩ (TJ = 25°C), the junction-to-case thermal resistance (Rθ(J-C)) is 0.4°C/W, junction-to-ambient thermal resistance (Rθ(J-A)) is 40°C/W, and the Transient Thermal Impedence Curve is shown in Figure 7. Consider, say, the MOSFET is switched on at time t1 and the steady-state load current passing through the MOSFET is 3A. At some point in time after t1, at time t2, there is an unexpected short-circuit applied to the load, causing the MIC2588/MIC2594 controller to adjust the GATE output voltage and regulate the load current for 400µs at the programmed current limit value, 4.2A in this example. During this short-circuit load condition, the dissipation in the MOSFET is calculated by: PD(short) = VDS × ILIM VDS = 0V – (-48V) = 48V PD(short) = 48V × 4.2A = 201.6W for 400µs. At first glance, it would appear that a very hefty MOSFET is required to withstand this extreme overload condition. Upon further examination, the calculation to approximate the peak junction temperature is not a difficult task. The first step is to determine the maximum steady-state junction temperature, 16 M9999-083005 MIC2588/MIC2594 Micrel then add the rise in temperature due to the maximum power dissipated during a transient overload caused by a short circuit condtion. The equation to estimate the maximum steady-state junction temperature is given by: TJ(steady-state) ≅ TC(max) + ΔTJ (10) TC(max) is the highest anticipated case temperaure, prior to an overcurrent condition, at which the MOSFET will operate and is estimated from the following equation based on the highest ambient temperature of the system environment. TC(max) = TA(max) + PD × (Rθ(J-A) – Rθ(J-C)) (11) Let’s assume a maximum ambient of 60°C. The power dissipation of the MOSFET is determined by the current through the MOSFET and the on-resistance (I2R), which we will estimate at 17mΩ (specification given at TJ = 125°C). Using our example information and substituting into Equation 11, TC(max) = 60°C + [((3A)2 × 17mΩ) × (40 – 0.4)°C/W] = 66.06°C Substituting the variables into Equation 10, TJ is determined by: Another iteration shows that the result (73.63°C) is converging quickly, so we’ll estimate the maximum TJ(steady-state) at 74°C. The use of the Transient Thermal Impedence Curves is necessary to determine the increase in junction temperature associated with a worst-case transient condition. From our previous calculation of the maximum power dissipated during a short circuit event for the MIC2588/MIC2594, we calculate the transient junction temperature increase as: TJ(transient) = PD(short) × Rθ(J-C) × Multiplier (12) Assume the MOSFET has been on for a long time – several minutes or more – and delivering the steady-state load current of 3A to the load when the load is short circuited. The controller will regulate the GATE output voltage to limit the current to the programmed value of 4.2A for approximately 400µs before immediately shutting off the output. For this situation and almost all hot swap applications, this can be considered a single pulse event as there is no significant duty cycle. From Figure 7, find the point on the X-axis (“Square-Wave Pulse Duration”) for 1ms, allowing for a healthy margin of the 400µs tFLT, and read up the Y-axis scale to find the intersection of the Single Pulse curve. This point is the normalized transient thermal impedence (Zθ(J-C)), and the effective transient thermal impedence is the product of Rθ(J-C) and the multiplier, 0.45 in this example. Solving Equation 12, TJ(transient) = (201.6W) × (0.4°C/W) × 0.45 = 36.3°C Finally, add this result to the maximum steady state junction temperature calculated previously to determine the estimated maximum transient junction temperature of the MOSFET: TJ(max.transient) = 74°C + 36.3°C = 110.3°C, which is safely under the specified maximum junction temperature of 200°C for the SUM110N10-09. TJ(steady-state) ≅ TC(max)+[RON+(TC(max)–TC)(0.005) × (RON)][I2×(Rθ(J-A)–Rθ(J-C))] ≅ 66.06°C+[17mΩ+(66.06°C–25°C)(0.005/°C) × (17mΩ)][(3A)2×(40–0.4)°C/W] ≅ 66.06°C + 7.30°C ≅ 73.36°C Since this is not a closed-form equation, getting a close apporoximation may take one or two iterations. On the second iteration, start with TJ equal to the value calculated above. Doing so in this example yields; TJ(steady-state) ≅ 66.06°C+[17mΩ+(73.36°C–25°C)×(0.005/°C) ×(17mΩ)][(3A)2×(40–0.4)]°C/W ≅ 73.62°C FIgure 7. Transient Thermal Impedance - SUM110N10-09 September 2005 17 M9999-083005 MIC2588/MIC2594 Micrel PCB Layout Considerations to flow while the rise in temperature for a given copper plate (e.g., 1oz. or 2oz.) is kept to a maximum of 10°C to 25°C. The return (or power ground) trace should be the same width as the positive voltage power traces (input/load) and isolated from any ground and signal planes so that the controller’s power is common mode. Also, these traces should be as short as possible in order to minimize the IR drops between the input and the load. Finally, the use of plated-through vias will be necessary to make circuit connections to the power, ground and signal planes of multi-layer PCBs. 4-Wire Kelvin Sensing Because of the low value typically required for the sense resistor, special care must be used to measure accurately the voltage drop across it. Specifically, the measurement technique across each RSENSE must employ 4-wire Kelvin sensing. This is simply a means of making sure that any voltage drops in the power traces connecting to the resistors are not picked up by the signal conductors measuring the voltages across the sense resistors. Figure 8 illustrates how to implement 4-wire Kelvin sensing. As the figure shows, all the high current in the circuit (from VEE through RSENSE, and then to the source of the output MOSFET) flows directly through the power PCB traces and RSENSE. The voltage drop resulting across RSENSE is sampled in such a way that the high currents through the power traces will not introduce any parasitic voltage drops in the sense leads. It is recommended to connect the hot swap controller’s sense leads directly to the sense resistor’s metalized contact pads. RSENSE metalized contact pads Power Trace From VEE RSENSE PCB Track Width: 0.03" per Ampere using 1oz Cu Signal Trace to MIC2588/MIC2594 VEE Pin Power Trace To MOSFET Source Signal Trace to MIC2588/MIC2594 SENSE Pin Note: Each SENSE lead trace shall be balanced for best performance with equal length/equal aspect ratio. Other Layout Considerations Figure 9 is a suggested PCB layout diagram for the MIC2588/ MIC2594. Many hot swap applications will require load currents of several amperes. Therefore, the power (VEE and Return) trace widths (W) need to be wide enough to allow the current Figure 8. 4-Wire Kelvin Sense Connections for RSENSE Current Flow to the Load W GROUND PAD Via to the Return (VDD) plane MIC2588-2BM ^R1 /PWRGD R3 OV DRAIN UV GATE R2 Via to the bottom side Via to the Return (VDD) plane Via to the power (VEE output) plane VDD VEE SENSE D1 C1 RFDBK Current Flow from the Load C3 CFDBK Via to the power (VEE output) plane R4 W W Via to the ground plane *SENSE RESISTOR (WSR-2 or WSL2512) *POWER MOSFET (TO-263) - DRAWING IS NOT TO SCALE*See Table 1 for part numbers and vendors ^R1 placed on bottom side Power Plane -------- (red) Ground Plane ------- (black) Trace width (W) guidelines and additional information given in "PCB Layout Recommendations" section of the datasheet Figure 9. Recommended PCB Layout for Sense Resistor, Power MOSFET, Overvoltage/Undervoltage Resistive Divider Network, and Timer Capacitors September 2005 18 M9999-083005 MIC2588/MIC2594 Micrel MOSFET and Sense Resistor Vendors Device types, part numbers, and manufacturer contacts for power MOSFTETS and sense resistors are provided in Table 1. MOSFET Vendors Key MOSFET Type(s) Breakdown Voltage (VDSS) SUM75N06-09L (TO-263) SUM70N06-11 (TO-263) SUM50N06-16L (TO-263) 60V 60V 60V SUP85N10-10 (TO-220AB) SUB85N10-10 (TO-263) SUM110N10-09 (TO-263) SUM60N10-17 (TO-263) 100V 100V 100V 100V www.siliconix.com (203) 452-5664 International Rectifier IRF530 (TO-220AB) IRF540N (TO-220AB) 100V 100V www.irf.com (310) 322-3331 Renesas 2SK1298 (TO-3PFM) 2SK1302 (TO-220AB) 2SK1304 (TO-3P) 60V 100V 100V www.renesas.com (408) 433-1990 Vishay - Siliconix Contact Information www.siliconix.com (203) 452-5664 Resistor Vendors Sense Resistors Contact Information Vishay - Dale “WSL” and “WSR” Series www.vishay.com/docswsl_30100.pdf (203) 452-5664 IRC “OARS” Series “LR” Series second source to “WSL” www.irctt.com/pdf_files/OARS.pdf www.irctt.com/pdf_files/LRC.pdf (828) 264-8861 Table 1. MOSFET and Sense Resistor Vendors September 2005 19 M9999-083005 MIC2588/MIC2594 Micrel /PWRGD Signal Drive Capability The /PWRGD signal can be used to drive an optoisolator or an LED. The use of an optoisolator is sometimes needed to protect I/O signals (e.g., /PWRGD, RESET, ENABLE) of both the controller and downstream DC-DC converter(s) from damage caused by common mode transients. Such is the case when an EMI filter is utilized to prevent DC-DC converter switching noise from being injected back onto the power supply. The circuit of Figure 10 shows how to configure an optoisolator driven by the /PWRGD signal of the MIC2588 controller. -48V RTN R5 43kΩ 1 -48V RTN (Long Pin) 6 MOC207-M 2 5 DC-DC CONVERTER 1 *D1 SMAT70A 100V 2 -48V RTN (Short Pin) R1 698kΩ 1% R3 12.4kΩ 1% C1 1uF R2 11.8 kΩ 1% /PWRGD VDD OV DRAIN 8 IN+ C5 100uF 7 C4 0.1uF IN- MIC2588-2BM 3 4 UV +2.5VOUT OUT+2.5V RTN GATE SENSE VEE OUT+ ON/OFF# 6 5 RFDBK 15kΩ *C2 22nF CFDBK 6.8nF 100V R4 10Ω C3 0.22uF VDD *C6 0.33uF -48VIN (Long Pin) -48VOUT RSENSE 0.01Ω 5% M1 SUM110N10-09 Nominal Undervoltage and Overvoltage Thresholds: VUV = 36.5V VOV = 71.2V * Optional components (See Applications Information for more details) # An external pull-up resistor for the power-good signal is necessary for DC-DC supplies (and all other load modules) not equipped with internal pull-up impedence Figure 10. Optoisolator Driven by /PWRGD Signal September 2005 20 M9999-083005 MIC2588/MIC2594 Micrel Package Information 0.026 (0.65) MAX) PIN 1 0.157 (3.99) 0.150 (3.81) DIMENSIONS: INCHES (MM) 0.020 (0.51) 0.013 (0.33) 0.050 (1.27) TYP 0.064 (1.63) 0.045 (1.14) 45 0.0098 (0.249) 0.0040 (0.102) 0.197 (5.0) 0.189 (4.8) 0–8 0.010 (0.25) 0.007 (0.18) 0.050 (1.27) 0.016 (0.40) SEATING PLANE 0.244 (6.20) 0.228 (5.79) 8-Pin SOIC (M) MICREL, INC. TEL 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA + 1 (408) 944-0800 FAX + 1 (408) 474-1000 WEB http://www.micrel.com The information furnished by Micrel in this datasheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer. Micrel Products are not reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is at Purchaser’s own risk and Purchaser agrees to fully indemnify Micrel for any damages resulting from such use or sale. © 2005 Micrel, Incorporated. M9999-083005 21 September 2005