STMICROELECTRONICS L6590A

L6590A
FULLY INTEGRATED POWER SUPPLY
■
WIDE-RANGE MAINS OPERATION
■
"ON-CHIP" 700V V(BR)DSS POWER MOS
■
65 kHz INTERNAL OSCILLATOR
■
STANDBY MODE FOR HIGH EFFICIENCY AT
LIGHT LOAD
■
OVERCURRENT AND LATCHED
OVERVOLTAGE PROTECTION
■
NON DISSIPATIVE BUILT-IN START-UP
CIRCUIT
■
THERMAL SHUTDOWN WITH HYSTERESIS
■
BROWNOUT PROTECTION
MINIDIP
ORDERING NUMBER: L6590AN
- HOME APPLIANCES/LIGHTING
■
MAIN APPLICATIONS
■ WALL PLUG POWER SUPPLIES UP TO 15W
■
AC-DC ADAPTERS
■
AUXILIARY POWER SUPPLIES FOR:
- CRT AND LCD MONITOR (BLUE ANGEL)
- DESKTOP PC/SERVER
- FAX, TV, LASER PRINTER
LINE CARD, DC-DC CONVERTERS
DESCRIPTION
The L6590A is a monolithic switching regulator designed in BCD OFF-LINE technology, able to operate
with wide range input voltage and to deliver up to
15W output power. The internal power switch is a lateral power MOSFET with a typical RDS(on) of 13Ω
and a V(BR)DSS of 700V minimum.
TYPICAL APPLICATION CIRCUIT
AC line
88 to 264 Vac
Pout
up to 15W
DRAIN
1
L6590A
BOK
Vcc
5
3
6, 7, 8
GND
October 2000
4
COMP
1/19
L6590A
efficiency (Pin < 1W @ Pout = 0.5W with wide range
mains).
DESCRIPTION (continued)
The MOSFET is source-grounded, thus it is possible
to build flyback, boost and forward converters.
Internal protections like cycle-by-cycle current limiting, latched output overvoltage protection, mains undervoltage protection and thermal shutdown
generate a 'robust' design solution.
The device is meant to work with secondary feedback for tight tolerance of the regulated output voltage.
The IC uses a special leadframe with the ground pins
(6, 7 and 8) internally connected in order for heat to
be easily removed from the silicon die. An heatsink
can then be realized by simply making provision of
few cm2 of copper on the PCB. Furthermore, the pin
close to the high-voltage one is not connected to
ease compliance with safety distances on the PCB.
The internal fixed oscillator frequency and the integrated non dissipative start-up generator minimize
the external component count and power consumption.
The device is equipped with a standby function that
automatically reduces the oscillator frequency from
65 to 22 kHz under light load conditions to enhance
BLOCK DIAGRAM
DRAIN
(1)
START-UP
VCC
(3)
OVER
VOLTAGE
SUPPLY
& UVLO
THERMAL
S.DOWN
+
-
VREF
BROWNOUT
GND
(6,7,8)
-
BOK
(5)
+
+
-
OVER
CURRENT
2.5V
PWM
STANDBY
OSC
65/22 kHz
1 mA
COMP
(4)
PIN CONNECTIONS (Top view)
2/19
DRAIN
GND
N.C.
GND
Vcc
GND
COMP
BOK
L6590A
PIN FUNCTIONS
N°
Pin
Description
1
DRAIN
2
N.C.
Not internally connected. Provision for clearance on the PCB.
3
Vcc
Supply pin of the IC. An electrolytic capacitor is connected between this pin and ground. The
internal start-up generator charges the capacitor until the voltage reaches the start-up threshold.
The PWM is stopped if the voltage at the pin exceeds a certain value.
4
COMP
PWM Control Input. The voltage on this pin (VCOMP) controls the PWM modulator: the higher
VCOMP, the higher the duty cycle. The pin will be driven by a current sink (usually the transistor of
an optocoupler) able to modulate VCOMP by modulating the current.
5
BOK
Brownout Protection. If the voltage applied to this pin is lower than 2.5V the PWM is disabled.
This pin is typically used for sensing the input voltage of the converter through a resistor divider.
If not used, the pin can be either left floating or connected to Vcc through a 15 kΩ resistor.
6 to 8
GND
Connection of both the source of the internal MOSFET and the return of the bias current of the
IC. Pins connected to the metal frame to facilitate heat dissipation.
Drain connection of the internal power MOSFET. The internal high voltage start-up generator
sinks current from this pin.
THERMAL DATA
Symbol
Parameter
Value
Unit
35 to 60
°C/W
15
°C/W
Value
Unit
-0.3 to 700
V
Drain Current
0.7
A
Vcc
IC Supply Voltage
18
V
Iclamp
Vcc Zener Current
20
mA
PWM Control Input Sink Current
3
mA
BOK pin Sink Current
1
mA
1.5
W
Operating Junction Temperature
-40 to 150
°C
Storage Temperature
-40 to 150
°C
Rthj-amb
Thermal Resistance Junction-ambient (*)
Rthj-pins
Thermal Resistance Junction-pins
(*) Value depending on PCB copper area and thickness.
ABSOLUTE MAXIMUM RATINGS
Symbol
Vds
Id
Ptot
Tj
Tstg
Parameter
Drain Source Voltage
Power Dissipation at Tamb < 50°C
3 cm2, 2 oz copper dissipating area on PCB
3/19
L6590A
ELECTRICAL CHARACTERISTCS (Tj = -25 to 125°C, Vcc = 10V; unless otherwise specified))
Symbol
Parameter
Test Condition
Min.
Typ.
Max.
Unit
POWER SECTION
V(BR)DSS Drain Source Voltage
Idss
RDS(on)
Id < 200 µA; Tj = 25 °C
Off state drain current
Vds = 560V; Tj = 125 °C
Drain-to-Source on resistance
RDS(on) vs. Tj: see fig. 17
Id = 120mA; Tj = 25 °C
Id = 120mA; Tj = 125 °C
700
V
200
µA
13
16
Ω
23
28
PWM CONTROL INPUT
Vout High
Isource = -0.5mA
3.8
4.5
ICOMP
Source Current
1.5V < VCOMP < 3.5V
-0.5
-1
RCOMP
Dynamic Resistance
1.5V < VCOMP < 3.5V
VCOMPH
V
-2.5
9
mA
kΩ
OSCILLATOR SECTION
Fosc
Oscillator Frequency
Tj = 25 °C
Dmin
Min. Duty Cycle
VCOMP = 1V
Dmax
Max. Duty Cycle
VCOMP = 4V
58
65
72
52
65
74
67
kHz
0
%
70
73
%
DEVICE OPERATION SECTION
Iop
Operating Supply Current
fsw = Fosc
4.5
7
mA
IQ
Quiescent Current
MOS disabled
3.5
6
mA
VCC charge Current
Vcc = 0V to Vccon - 0.5V;
Vds = 100 to 400V; Tj = 25°C
-3
-4.5
-7
mA
Vcc = 0V to Vccon - 0.5V;
Vds = 100 to 400V
-2.5
-4.5
-7.5
mA
Iclamp = 10mA (*)
15.5
16.5
17.5
V
Icharge
VCCclamp VCC Clamp Voltage
Vccon
Start Threshold
voltage
(*)
13.5
14.5
15.5
V
Vccoff
Min operating voltage after Turn
on
(*)
6
6.6
7.2
V
Vdsmin
Drain start voltage
40
V
CIRCUIT PROTECTIONS
Ipklim
VccOVP
LEB
Pulse-by-pulse Current Limit
di/dt = 120 mA/ µs
550
625
700
mA
Overvoltage Protection
Icc = 10 mA (*)
15
16
17
V
Masking Time
After MOSFET turn-on (**)
120
ns
STANDBY SECTION
FSB
4/19
Oscillator Frequency
19
22
25
kHz
L6590A
ELECTRICAL CHARACTERISTICS (continued)
Symbol
Parameter
Test Condition
Min.
Typ.
Max.
Unit
Ipksb
Peak switch current for Standby
Operation
Transition from Fosc to FSB
80
mA
Ipkno
Peak switch current for Normal
Operation
Transition from FSB to Fosc
190
mA
BROWNOUT PROTECTION
Vth
Threshold Voltage
Voltage either rising or falling
2.325
2.5
2.675
V
IHys
Current Hysteresis
Vpin = 3V
-30
-50
-70
µA
VCL
Clamp Voltage
Ipin = 0.5 mA
5.6
6.4
7.2
V
150
165
°C
40
°C
THERMAL SHUTDOWN (***)
Threshold
Hysteresis
(*) Parameters tracking one the other
(**) Parameter guaranteed by design, not tested in production
(***) Parameters guaranteed by design, functionality tested in production
5/19
L6590A
Figure 1. Start-up & UVLO Thresholds
Figure 4. IC Consumption Before Start-up
Vcc [V]
Icc [µA]
15
700
Tj = -25 °C
14
600
13
12
500
11
Tj = 25 °C
400
10
9
Tj = 125 °C
300
8
200
7
6
-50
0
50
100
150
100
7
8
9
10
Tj [°C]
11
12
Figure 2. Start-up Current Generator
Figure 5. IC Quiescent Current
Icc [mA]
Icc [mA]
5.5
14
15
4
Tj = 25 °C
MOSFET disabled
Vdrain = 40 V
5
Tj = -25 °C
3.8
4.5
Tj = 25 °C
3.6
4
3.4 Tj = 125 °C
3.5
3
13
Vcc [V]
Tj = 125 °C
0
2
4
6
8
10
12
Tj = -25 °C
3.2
3
6
8
10
Vcc [V]
12
14
16
18
Vcc [V]
Figure 3. Start-up Current Generator
Figure 6. IC Operating Current
Icc [mA]
Icc [mA]
5.5
5
Vdrain = 60 V
MOSFET switching @ 65 kHz
Tj = -25 °C
Tj = 125 °C
5
4.5
Tj = 25 °C
Tj = 25 °C
4.5
4
Tj = -25 °C
4
Tj = 125 °C
3.5
3.5
3
0
2
4
6
Vcc [V]
6/19
8
10
12
3
7
8
9
10
11
Vcc [V]
12
13
14
15
L6590A
Figure 7. IC Operating Current
Figure 10. OVP Threshold vs. Temperature
Icc [mA]
Vth [V]
16
4.4
MOSFET switching @ 22 kHz
4.2
Tj = 125 °C
15.8
4
Tj = 25 °C
15.6
3.8
Tj = -25 °C
3.6
15.4
3.4
15.2
3.2
3
7
8
9
10
11
12
13
14
15
15
-50
0
50
Vcc [V]
100
150
Tj [°C]
Figure 11. OCP Threshold vs. Current Slope
Figure 8. Switching Frequency vs.
Temperature
Ipklim / (Ipklim @ di/dt = 120 mA/µs)
fsw [kHz]
1.06
80
Normal operation
70
Tj = 25 °C
1.04
60
1.02
50
1
40
Standby
30
0.98
20
10
-50
0
50
100
150
0.96
50
100
150
200
250
dI/dt [mA/µs]
Tj [°C]
Figure 9. Vcc clamp vs. Temperature
Figure 12. OCP threshold vs. Temperature
VCCclamp [V]
Ipklim / (Ipklim @ Tj = 25°C)
1.1
18
di/dt = 120 mA/µs
1.08
17.8
1.06
17.6
Iclamp = 20 mA
1.04
17.4
Iclamp = 10 mA
17.2
17
-50
1.02
1
0
50
Tj [°C]
100
150
0.98
-50
0
50
100
150
Tj [°C]
7/19
L6590A
Figure 13. COMP pin Characteristic
Figure 16. Drain Leakage vs. Drain Voltage
VCOMP [V]
Idrain [µA]
6
50
Tj = 125 °C
Tj = 25 °C
5
40
Tj = 25 °C
30
Tj = -25 °C
4
3
2
20
1
0
0
0.2
0.4
0.6
0.8
1
1.2
1.4
10
100
200
300
ICOMP [mA]
400
500
600
700
Vdrain [V]
Figure 14. COMP pin Dynamic Resistance vs.
Temperature
Figure 17. Rds(ON) vs. Temperature
Rds(ON) / (Rds(ON) @ Tj=25°C)
RCOMP [kOhm]
1.8
10.5
1.6
Idrain = 120 mA
10
1.4
9.5
1.2
9
1
8.5
0.8
8
-50
0
50
100
150
0.6
-50
0
50
100
150
Tj [°C]
Tj [°C]
Figure 15. Breakdown Voltage vs. Temperature
Figure 18. Rds(ON) vs. Idrain
BVDSS / (BVDSS @ Tj = 25°C)
Rds(ON) / (Rds(ON) @ Idrain=120 mA)
1.08
1.3
1.06
Tj = 25 °C
Idrain = 200 µA
1.2
1.04
1.02
1.1
1
0.98
1
0.96
0.94
0.92
-50
0
50
Tj [°C]
8/19
100
150
0.9
0
100
200
300
Idrain [mA]
400
500
600
L6590A
Figure 19. Coss vs. Drain Voltage
Figure 20. Standby Function Thresholds
Coss [pF]
Drain Peak Current [mA]
250
220
22 kHz → 65 kHz
200
Tj = 25 °C
200
180
160
150
140
100
120
65 kHz → 22 kHz
100
50
80
0
0
100
200
300
400
500
600
700
60
-50
0
50
Vdrain [V]
100
150
Tj [°C]
Figure 21. Test Board electrical schematic
F1
2A/250V
Vin
88 to 264 Vac
CxA
100 nF
L
22 mH
BD1
DF06M
T1
C1
22 µF
400 V
CxB
100 nF
L1
4.7 µH
D4 1N5821
D1
BZW06-154
C8
220 µF
10V
Rubycon
ZL
C5, C6, C7
470 µF
16V
Rubycon ZL
D2
STTA106
5 Vdc / 2 A
R1 10 Ω
R5 1.8 MΩ
1
5
3
IC1
L6590A
R6 39 kΩ
C2
22 µF
25V
6, 7, 8
D3
1N4148
R2
560 Ω
4
C3
22 nF
OP1
PC817
1
4
2
3
R6
6.8 kΩ
R3
2.43 kΩ
C9
100 nF
1
2
R5
2 kΩ
3
C4
2.2 nF
Y1 class
IC2
TL431
R4
2.43 kΩ
T1 specification
Core E20/10/6, ferrite 3C85 or N67 or equivalent
≈0.6 mm gap for a primary inductance of 1.4 mH
Lleakage <30 µH
Primary : 128 T, 2 series windings 64T each, AWG32 (∅ 0.22 mm)
Sec : 6 T, 4xAWG32
Aux : 14 T, AWG32
9/19
L6590A
Figure 22. Test Board evaluation data
Test Board Efficiency
Test board Load & Line regulation
Efficiency [%]
Output Voltage [V]
80
5
264 VAC
4.98
88 VAC
70
60
4.96
50
20
88 VAC
4.9
0.003
0.01
0.03
0.1
0.3
1
10
0.003
3
0.01
0.03
Test Board Light-load Consumption
0.3
1
3
Device Power Dissipation
Pdiss [W]
Input Power [mW]
5
1,200
Rthj-amb= 58 °C/W @ 1.5W
Pout
1,000
0.5W
800
2
88 VAC
264 VAC
1
600
0.25W
0.5
400
0.1W
0.05W
0.2
200
100
150
200
250
300
350
400
Figure 23. Test Board EMI characterization
220 VAC
110 VAC
0.1
0W
DC Input Voltage [V]
10/19
0.1
Load Current [A]
Load Current [A]
0
50
220 VAC
30
110 VAC
4.92
264 VAC
40
220 VAC
4.94
110 VAC
450
0.05
0.003
0.01
0.03
0.1
0.3
Load Current [A]
1
3
L6590A
Figure 24. Test Board main waveforms
Ch1: Vdrain
A1: Idrain
Vin = 400 VDC
Iout = 2 A
Vin = 100 VDC
Iout = 2 A
A1: Idrain
Ch1: Vdrain
A1: Idrain
A1: Idrain
Vin = 100 VDC
Iout = 50 mA
Ch1: Vdrain
Vin = 400 VDC
Iout = 50 mA
Ch1: Vdrain
Figure 25. Test Board load transient response
Vout
Vout
Vin = 200 VDC
Iout = 0.2 ↔ 0.4 A
Iout
Standby Function
is not tripped
transition
65 22 kHz
⇒
transition
22 65 kHz
⇒
Vin = 200 VDC
Iout = 0.1 ↔ 0.3 A
Iout
Standby Function
is tripped
11/19
L6590A
APPLICATION INFORMATION
In the following sections the functional blocks as well as the most important internal functions of the device will
be described.
Start-up circuit
When power is first applied to the circuit and the voltage on the bulk capacitor is sufficiently high, an internal
high-voltage current generator is sufficiently biased to start operating and drawing about 4.5 mA through the
primary winding of the transformer and the drain pin. Most of this current charges the bypass capacitor connected between pin Vcc (3) and ground and makes its voltage rise linearly.
As the Vcc voltage reaches the start-up threshold (14.5V typ.) the chip, after resetting all its internal logic, starts
operating, the internal power MOSFET is enabled to switch and the internal high-voltage generator is disconnected. The IC is powered by the energy stored in the Vcc capacitor until the self-supply circuit (typically an
auxiliary winding of the transformer) develops a voltage high enough to sustain the operation.
As the IC is running, the supply voltage, typically generated by a self-supply winding, can range between 16 V
(Overvoltage protection limit, see the relevant section) and 7 V, threshold of the Undervoltage Lockout. Below
this value the device is switched off (and the internal start-up generator is activated). The two thresholds are in
tracking.
The voltage on the Vcc pin is limited at safe values by a clamp circuit. Its 17V threshold tracks the Overvoltage
protection threshold.
Figure 26. Start-up circuit internal schematic
DRAIN
POWER
MOSFET
15 MΩ
UVLO
150 Ω
Vcc
17 V
L6590A
GND
Power MOSFET and Gate Driver
The power switch is implemented with a lateral N-channel MOSFET having a V(BR)DSS of 700V min. and a typical RDS(on) of 13Ω. It has a SenseFET structure to allow a virtually lossless current sensing (used only for
protection).
During operation in Discontinuous Conduction Mode at low mains the drain voltage is likely to go below ground.
Any risk of injecting the substrate of the IC is prevented by an internal structure surrounding the switch.
The gate driver of the power MOSFET is designed to supply a controlled gate current during both turn-on and
turn-off in order to minimize common mode EMI.
Under UVLO conditions an internal pull-down circuit holds the gate low in order to ensure that the power MOSFET cannot be turned on accidentally.
12/19
L6590A
Figure 27. PWM Control internal schematic
Max. Duty cycle
S
OSCILLATOR
Clock
R
+
PWM
-
Q
to gate
driver
from OCP
comparator
L6590A
1 mA
COMP
Oscillator and PWM Control
PWM regulation is accomplished by implementing voltage mode control. As shown in fig. 27, this block includes
an oscillator, a PWM comparator, a PWM latch and a PWM control input.
The oscillator operates at a frequency internally fixed at 65 kHz with a precision of ± 10%. This value has been
selected so that the second harmonic falls below 150 kHz, beyond which some international standards envisage
more severe limits. The maximum duty cycle is limited at 70% typ.
The PWM latch (reset dominant) is set by the clock pulses of the oscillator and is reset by either the PWM comparator or the Overcurrent comparator. The inverting input of the PWM comparator is externally available (pin
4, COMP) in order for an optocoupler-based circuit to close the control loop that regulates the converter's output
voltage.
In case of overcurrent the voltage at pin COMP saturates high and the conduction of the power MOSFET is
stopped by the OCP comparator instead of the PWM comparator.
Under zero load conditions the COMP pin is close to its low saturation and the gate drive delivers as short pulses
as it can, limited by internal delays. They are however too long to maintain the long-term energy balance, thus
from time to time some cycles need being skipped and the operation becomes asynchronous. This is automatically done by the control loop.
Standby Function
The standby function, optimized for flyback topology, automatically detects a light load condition for the converter and decreases the oscillator frequency. The normal oscillation frequency is automatically resumed when the
output load builds up and exceeds a defined threshold.
This function allows to minimize power losses related to switching frequency, which represent the majority of losses
in a lightly loaded flyback, without giving up the advantages of a higher switching frequency at heavy load.
The Standby function is realized by monitoring the peak current in the power switch. If the load is low that it does
not reach a threshold (80 mA typ.), the oscillator frequency will be set at 22 kHz typ.
When the load demands more power and the peak primary current exceeds a second threshold (190 mA typ.)
the oscillator frequency is reset at 65 kHz. This 110 mA hysteresis prevents undesired frequency change when
power is such that the peak current is close to either threshold.
The signal coming from the sense circuit is digitally filtered to avoid false triggering of this function as a result of
large load changes or noise.
13/19
L6590A
Figure 28. Standby Function timing diagram
Pout
Peak
Primary
Current
00000000000000000000000000000000000000000000
80 mA
190 mA
00000000000000000000000000000000000000000000
00000000000000000000000000000000000000000000
00000000000000000000000000000000000000000000
00000000000000000000000000000000000000000000
00000000000000000000000000000000000000000000
00000000000000000000000000000000000000000000
00000000000000000000000000000000000000000000
00000000000000000000000000000000000000000000
00000000000000000000000000000000000000000000
00000000000000000000000000000000000000000000
Load
regulation
Vout
small glitch
STANDBY
(before filter)
2 ms
1 ms
STANDBY
(filtered)
65 kHz
fsw
22 kHz
Brownout Protection
Brownout Protection is basically a not-latched device shutdown functionality. It will typically be used to detect a
mains undervoltage (brownout). This condition may cause overheating of the primary power section due to an
excess of RMS current.
Figure 29. Brownout Function internal schematic and timing diagram
HV Input bus
VON
VOFF
HV Input bus
Vcc
VinOK
50 µA
BOK
Vcc
+
6.4 V
VinOK
-
2.5 V
L6590A
PWM
Vout
14/19
00000000000000000000
00000000000000000000
00000000000000000000
00000000000000000000
00000000000000000000
00000000000000000000
00000000000000000000
00000000000000000000
L6590A
Another problem is the spurious restarts that are likely to occur during converter power down if the input voltage
decays slowly (e.g. with a large input bulk capacitor) and that cause the output voltage not to decay to zero
monothonically.
Converter shutdown can be accomplished with the L6590A by means of an internal comparator that can be used
to sense the voltage across the input bulk capacitor. This comparator is internally referenced to 2.5V and disables the PWM if the voltage applied at its non-inverting input, externally available, is below the reference. PWM
operation is re-enabled as the voltage at the pin is more than 2.5V.
The brownout comparator is provided with current hysteresis instead of a more usual voltage hysteresis: an internal 50 µA current generator is ON as long as the voltage applied at the non-inverting input exceeds 2.5V and
is OFF if the voltage is below 2.5V. This approach provides an additional degree of freedom: it is possible to set
the ON threshold and the OFF threshold separately by properly choosing the resistors of the external divider,
which is not possible with voltage hysteresis.
Overvoltage Protection
The IC incorporates an Overvoltage Protection (OVP) that can be particularly useful to protect the converter and
the load against voltage feedback loop failures. This kind of failure causes the output voltage to rise with no
control and easily leads to the destruction of the load and of the converter itself if not properly handled.
If such an event occurs, the voltage generated by the auxiliary winding that supplies the IC will fly up tracking
the output voltage. This will activate an internal clamp circuit and, as the current sunk by this clamp exceeds
about 10 mA, the operation of the IC will be stopped. This condition is latched and maintained until the Vcc voltage falls below the UVLO threshold. The converter will then operate intermittently.
Figure 30. OVP internal schematic
Vcc
DRAIN
to
MOSFET
CLAMP
+
to OVP
latch
OVP
-
GND
L6590A
Overcurrent Protection
The device uses pulse-by-pulse current limiting for Overcurrent Protection (OCP), in order to prevent overstress
of the internal MOSFET: its current during the ON-time is monitored and, if it exceeds a determined value, the
conduction is terminated immediately. The MOSFET will be turned on again in the subsequent switching cycle.
As previously mentioned, the internal powerMOSFET has a SenseFET structure: the source of a few cells are
connected together and kept separate from the other source connections so as to realize a 1:100 current divider.
The "sense" portion is connected to a ground referenced, sense resistor having a low thermal coefficient. The
OCP comparator senses the voltage drop across the sense resistor and resets the PWM latch if the drop exceeds a threshold, thus turning off the MOSFET. In this way the overcurrent threshold is set at about 0.65 A
(typical value).
15/19
L6590A
At turn-on, there are large current spikes due to the discharge of parasitic capacitances and, in case of Continuos Conduction Mode operation, to secondary diode reverse recovery as well, which could falsely trigger the
OCP comparator. To increase noise immunity the output of the OCP comparator is blanked for a short time
(about 120 ns) just after the MOSFET is turned on, so that any disturbance within this time slot is rejected (Leading Edge Blanking).
Figure 31. OCP internal schematic
DRAIN
Max. Duty cycle
S
OSCILLATOR
Driver
Clock
R
Q
1
1/100
+
PWM
-
+
OCP
-
Clock
LEB
Rsense
0.5 V
GND
Thermal Shutdown
Overheating of the device due to an excessive power throughput or insufficient heatsinking is avoided by the
Thermal Shutdown function. A thermal sensor monitors the junction temperature close to the power MOSFET
and, when the temperature exceeds 150 °C (min.), sets an alarm signal that stops the operation of the device.
This is a not-latched funtion and the power MOSFET is re-enabled as the temperature falls about 40 °C.
16/19
L6590A
APPLICATION IDEAS
Figure 32. 15W Auxiliary SMPS for PC
T1
Vin = 200 to 375 Vdc
L1
4.7 µH
D4 STPS10L25D
D1
BZW06-154
C8
100 µF
10V
C5, C6, C7
470 µF
10 V
D2
STTA106
5 Vdc / 3 A
R1 10 Ω
R2
1.8 MΩ
1
D3
1N4148
C2
22 µF
25 V
3
R4
560 Ω
IC1
5
C1
10 nF
4
L6590A
C3
47 nF
6, 7, 8
4
1
3
2
R5
2.43 kΩ
OP1
PC817
R3
20 kΩ
1
C4
2.2 nF
Y
R7
240 Ω
C9
470 nF
3
R6
2.43 kΩ
IC2
TL431
2
T1 specification
Core E20/10/6, ferrite 3C85 or N67 or equivalent
≈ 0.9 mm gap for a primary inductance of 2 mH
Lleakage <50 µH
Primary : 200 T, 2 series windings 100T each, AWG33 (∅ 0.22 mm)
Sec : 9 T, 2 x AWG23 (∅ 0.64 mm)
Aux : 21 T, AWG33
Figure 33. 10W AC-DC Adapter
F1
2A/250V
Vin
88 to 264 Vac
CxA
100 nF
L
22 mH
BD1
DF06M
T1
CxB
100 nF
C1
22 µF
400 V
L1
4.7 µH
D4 STPS3L60S
D1
BZW06-154
C8
100 µF
16 V
C6, C7
330 µF
16 V
D2
STTA106
12 Vdc / 0.8 A
R5
1 kΩ
R1 10 Ω
R2
1.8 MΩ
1
C2
22 µF
25 V
3
D3
1N4148
R4
100 Ω
IC1
5
C3
10 nF
L6590A
6, 7, 8
4
C4
47 nF
R3
39 kΩ
T1 specification
Core E20/10/6, ferrite 3C85 or N67 or equivalent
≈0.5 mm gap for a primary inductance of 1.6 mH
Lleakage <30 µH
Primary : 130 T, 2 series windings 65T each, AWG33 (∅ 0.22 mm)
Sec : 14 T, AWG26 (∅ 0.4 mm)
Aux : 14 T, AWG33
4
1
3
2
OP1
PC817
C5
2.2 nF
Y
D5
BZX79C10
REFERENCES
[1] “Getting Familiar with the L6590 Family, High-voltage Fully Integrated Power Supply” (AN1261)
[2] “Offline Flyback Converters Design Methodology with the L6590 Family” (AN1262)
17/19
L6590A
18/19
L6590A
Information furnished is believed to be accurate and reliable. However, STMicroelectronics assumes no responsibility for the consequences
of use of such information nor for any infringement of patents or other rights of third parties which may result from its use. No license is granted
by implication or otherwise under any patent or patent rights of STMicroelectronics. Specifications mentioned in this publication are subject
to change without notice. This publication supersedes and replaces all information previously supplied. STMicroelectronics products are not
authorized for use as critical components in life support devices or systems without express written approval of STMicroelectronics.
The ST logo is a registered trademark of STMicroelectronics
 2000 STMicroelectronics - All Rights Reserved
STMicroelectronics GROUP OF COMPANIES
Australia - Brazil - China - Finland - France - Germany - Hong Kong - India - Italy - Japan - Malaysia - Malta - Morocco - Singapore - Spain
- Sweden - Switzerland - United Kingdom - U.S.A.
http://www.st.com
19/19