ISL6260, ISL6260B ® Data Sheet January 3, 2006 Features Ordering Information PART NUMBER (Note) TEMP. RANGE (°C) PART MARKING • Multiple Current Sensing Approaches Supported - Lossless DCR Current Sensing - Precision Resistive Current Sensing • Supports PSI# and Narrow VDC for Enhanced Battery Life (EBL) Initiatives • Superior Noise Immunity and Transient Response • Thermal Monitor • Differential Remote Voltage Sensing • High Efficiency Across Entire Load Range • Programmable 1, 2 or 3 Power Channels • Balanced Channel Loading Including Transients • Small Footprint QFN 40 Lead 6x6 Package • Pb-Free Plus Anneal Available (RoHS Compliant) Applications • Mobile laptop computers Pinout DPRSLPVR VR_ON VID6 VID5 VID4 VID3 ISL6260CRZ, ISL6260BCRZ (QFN) TOP VIEW DPRSTP# The ISL6260 and ISL6260B have several other key features. Current sensing can be done using either DCR sensing or discrete precision resistor sensing. A single NTC thermistor thermally compensates both the gain and time constant of the DCR variation. A unity gain, differential amplifier is provided for remote CPU die sensing. This allows the voltage on the CPU die to be accurately measured and regulated per Intel IMVP-6 specifications. • Microprocessor Voltage Identification Input - 7-Bit VID Input - 0.300V to 1.500V in 12.5mV Steps - Supports VID Changes On-The-Fly CLK_EN# Intel Mobile Voltage Positioning is a smart voltage regulation technology, which effectively reduces power dissipation in Intel Pentium processors. The ISL6260 and ISL6260B support the IMVP-6 mobile processor voltage regulation specifications. ISL6260 and ISL6260B are pin-to-pin compatible. ISL6260B responds to PSI# signal by adding or dropping PWM2 and adjusting overcurrent protection accordingly. To improve audible noise, the DPRSLPVR signal can be used to reduce slew rates entering and exiting Deeper Sleep. • Precision Multiphase Core Voltage Regulation - 0.5% System Accuracy Over Temperature - Enhanced Load Line Accuracy 3V3 The ISL6260 and ISL6260B provide microprocessor core voltage regulation by driving up to 3 channels in parallel. The multiphase buck converter architecture uses interleaved channels to multiply the output voltage ripple frequency and reduce output channel currents. The reduction in ripple results in fewer components, lower component cost, reduced power dissipation, and smaller implementation area. The ISL6260, ISL6260B multiphase controller together with the ISL6208 gate drivers form the basis for a portable power supply solution to power Intel's next generation mobile microprocessors. The modulator at the heart of this power system is derived from Intersil's Robust Ripple Regulator technology, (R3) Compared with the traditional multiphase buck regulator, the R3 technology multiphase converter has faster transient response. This is due to the R3 modulator commanding variable switching frequency during load transients. PGOOD Multiphase Core Regulator for IMVP-6 Mobile CPUs FN9162.1 40 39 38 37 36 35 34 33 32 31 PSI# 1 30 VID2 PGD_IN 2 29 VID1 RBIAS 3 28 VID0 VR_TT# 4 PACKAGE (Pb-FREE) PKG. DWG. # NTC 5 27 PWM1 26 PWM2 GND PAD (BOTTOM) 25 PWM3 ISL6260BCRZ -10 to 100 40 Ld 6x6 QFN L40.6x6 VW 8 23 ISEN1 ISL6260BCRZ-T ISL6260BCRZ -10 to 100 40 Ld 6x6 QFN L40.6x6 COMP 9 22 ISEN2 NOTE: Intersil Pb-free plus anneal products employ special Pb-free material sets; molding compounds/die attach materials and 100% matte tin plate termination finish, which are RoHS compliant and compatible with both SnPb and Pb-free soldering operations. Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020. FB 10 21 ISEN3 1 11 12 13 14 15 16 17 18 19 20 VDD ISL6260BCRZ VSS 24 FCCM VIN OCSET 7 VSUM -10 to 100 40 Ld 6x6 QFN L40.6x6 VO ISL6260CRZ DFB ISL6260CRZ-T DROOP SOFT 6 RTN -10 to 100 40 Ld 6x6 QFN L40.6x6 VSEN ISL6260CRZ VDIFF ISL6260CRZ CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc. Copyright © Intersil Americas Inc. 2005, 2006. All Rights Reserved All other trademarks mentioned are the property of their respective owners. ISL6260, ISL6260B Functional Pin Description PGOOD 3V3 CLK_EN# DPRSTP# DPRSLPVR VR_ON VID6 VID5 VID4 VID3 VW 40 39 38 37 36 35 34 33 32 31 A resistor from this pin to COMP programs the switching frequency. (7kΩ gives approximately 300kHz). VW pin sources current. COMP This pin is the output of the error amplifier. PSI# 1 30 VID2 PGD_IN 2 29 VID1 RBIAS 3 28 VID0 FB This pin is the inverting input of error amplifier. VR_TT# 4 27 PWM1 NTC 5 This pin is the output of the differential amplifier. 26 PWM2 GND PAD (BOTTOM) SOFT 6 VDIFF VSEN 25 PWM3 OCSET 7 24 FCCM Remote core voltage sense input. Connect to microprocessor die. VW 8 23 ISEN1 RTN COMP 9 22 ISEN2 FB 10 21 ISEN3 Remote voltage sensing return. Connect to ground at microprocessor die. 12 13 14 15 16 17 18 19 20 VDIFF VSEN RTN DROOP DFB VO VSUM VIN VSS VDD DROOP 11 Output of droop amplifier. Output = VO + DROOP. DFB Inverting input to droop amplifier. PSI# VO Low load current indicator input. When asserted low, indicates a reduced load-current condition. For ISL6260B, when PSI# is asserted low, PWM2 will be disabled. An input to the IC that reports the local output voltage. VSUM PGD_IN This pin is connected to the current summation junction. Digital Input. When asserted high, indicates VCCP and VCC_MCH voltages are within regulation. PGD_IN signal high is needed for the CLK_EN# to be low and PGOOD to be high. VIN Battery supply voltage, used for feed forward. VSS Signal ground; Connect to local controller ground. RBIAS 147K Resistor to VSS sets internal current reference. VDD 5V bias power. VR_TT# Thermal overload output indicator. ISEN3 Individual current sensing for channel 3. NTC Thermistor input to VRTT# circuit. ISEN2 Individual current sensing for channel 2. SOFT A capacitor from this pin to Vss sets the maximum slew rate of the output voltage. Soft pin is the non-inverting input of the error amplifier. ISEN1 OCSET Forced Continuous Conduction Mode (FCCM) enable pin to MOSFET drivers. It will disable diode emulation. Overcurrent set input. A resistor from this pin to VO sets DROOP voltage limit for OC trip. A 10µA current source is connected internally to this pin. Individual current sensing for channel 1. FCCM PWM3 PWM output for channel 3. 2 FN9162.1 January 3, 2006 ISL6260, ISL6260B PWM2 VR_ON PWM output for channel 2. For ISL6260B, PSI# low will make this output tri-state. Voltage Regulator enable input. A high level logic signal on this pin enables the regulator. PWM1 DPRSLPVR PWM output for channel 1. Deeper Sleep Enable signal. A high level logic signal on this pin indicates that the micro-processor is in Deeper Sleep Mode and indicates that slow entry and exit from C4 should occur. DPRSLPVR low indicates large charging or discharging soft pin current, and therefore fast output voltage transitions. VID0, VID1, VID2, VID3, VID4, VID5, VID6 VID input with VID0 = LSB. CLK_EN# Digital output to enable System PLL Clock; Goes active 10µs after PG_IN is active and Vcore is within 10% of Boot Voltage. PGOOD Power Good open-drain output. Will be pulled up externally by a 680Ω resistor to VCCP or 1.9kΩ to 3.3V. DPRSTP# Deeper Sleep Enable signal. A low level logic signal on this pin indicates that the micro-processor is in Deeper Sleep Mode. 3V3 3.3V supply voltage for CLK_EN# logic, such an implementation will improve power consumption from 3.3V compared to open drain circuit other wise. 3 FN9162.1 January 3, 2006 ISL6260, ISL6260B Absolute Maximum Ratings Thermal Information Supply Voltage, VDD . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3 - +7V Battery Voltage, VIN. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +25V Open Drain Outputs, PGOOD, VR_TT# . . . . . . . . . . . . . -0.3 - +7V ALL OTHER PINS . . . . . . . . . . . . . . . . . . . . . -0.3V to (VDD + 0.3V) Thermal Resistance θJA (°C/W) θJC (°C/W) QFN Package (Notes 1, 2) . . . . . . . . . 30 5.5 Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . 150°C Storage Temperature . . . . . . . . . . . . . . . . . . . . . . . .-65°C to +150°C Maximum Lead Temperature (Soldering, 10s). . . . . . . . . . . . +300°C Operating Conditions Temperature Range . . . . . . . . . . . . . . . . . . . . . . . . . .-10°C to 100°C Supply Voltage Range (Typical). . . . . . . . . . . . . . . . . . . . . +5V ±5% CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. NOTES: 1. θJA is measured with the component mounted on a low effective thermal conductivity test board in free air. See Tech Brief TB379 for details. 2. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside. Electrical Specifications Operating Conditions: VDD = 5V, TA = -10°C to +100°C, unless otherwise noted. PARAMETER SYMBOL TEST CONDITIONS MIN TYP MAX UNITS 2.9 3.5 mA VR_ON = 0V 1 µA INPUT POWER SUPPLY +5V Supply Current IVDD VR_ON = 3.3V +3.3V Supply Current I3V3 No load on CLK_EN# 1 µA Battery Supply Current IVIN VR_ON = 0V 1 µA VIN Input Resistance RVIN VR_ON = 3.3V 900 Power-On-Reset Threshold PORr VDD rising 4.35 PORf VDD falling 4.00 No load; closed loop, active mode range VID = 0.75V - 1.50V -0.5 +0.5 % VID = 0.5V - 0.7375V -8 +8 mV VID = 0.3 - 0.4875V -15 +15 mV 1.224 V kΩ 4.5 V 4.15 V SYSTEM AND REFERENCES System Accuracy %Error (VCC_CORE) 1.176 VBOOT 1.200 Maximum Output Voltage VCC_CORE(max) VID = [0000000] 1.500 V Minimum Output Voltage VCC_CORE(min) VID = [1100000] 0.300 V 0.0 V VID Off State VID = [1111111] RBIAS Voltage RBIAS = 147kΩ 1.45 1.47 1.49 V Rfset = 7kΩ, 3 channel operation, Vcomp = 2V 285 300 315 kHz See Equation 4 Rfset selection 200 500 kHz -0.3 +0.3 mV CHANNEL FREQUENCY Nominal Channel Frequency fSW(nom) Adjustment Range AMPLIFIERS Droop Amplifier Offset Error Amp DC Gain Av0 Error Amp Gain-Bandwidth Product GBW FB Input Current IIN(FB) 4 CL= 20pF 90 dB 18 MHz 10 150 nA FN9162.1 January 3, 2006 ISL6260, ISL6260B Electrical Specifications Operating Conditions: VDD = 5V, TA = -10°C to +100°C, unless otherwise noted. (Continued) PARAMETER SYMBOL TEST CONDITIONS MIN TYP MAX UNITS 2 mV ISEN Imbalance Voltage Maximum of ISENs - Minimum of ISENs Input Bias Current 20 nA SOFT CURRENT -46 -41 -36 μA ±175 ±200 ±225 μA DPRSLPVR = 3.3V -46 -41 -36 μA IC4EA DPRSLPVR = 3.3V 36 41 46 μA IC4EB DPRSLPVR = 0V 175 200 225 μA 0.26 0.4 V 1 μA Soft-start current ISS SOFT Geyserville Current IGV |SOFT-VDAC| >100mV SOFT Deeper Sleep Entry Current IC4 SOFT Deeper Sleep Exit Current SOFT Deeper Sleep Exit Current POWER GOOD AND PROTECTION MONITORS PGOOD Low Voltage VOL IPGOOD= 4mA PGOOD Leakage Current IOH PGOOD = 3.3V -1 PGOOD Delay tpgd CLK_ENABLE# LOW to PGOOD HIGH 5.5 6.8 8.1 ms Overvoltage Threshold OVH VO rising above setpoint for >1ms 160 200 240 mV Severe Overvoltage Threshold OVHS VO rising for >2µs 1.675 1.7 1.725 V 10 10.2 μA 4 mV OCSET Reference Current I(Rbias) = 10µA 9.8 OC Threshold Offset DROOP rising above OCSET for >150μs -2 Current Imbalance Threshold One ISEN above another ISEN for >1.2ms Undervoltage Threshold (VDIFF/SOFT) UVf VO falling below setpoint for >1.2ms 9 -355 -295 mV -235 mV 1.0 V LOGIC THRESHOLDS VR_ON, DPRSLPVR and PGD_IN Input Low VIL(3.3V) VR_ON, DPRSLPVR and PGD_IN Input High VIH(3.3V) VID0-VID6, PSI#, DPRSTP# Input Low VIL(1.0V) VID0-VID6, PSI#, DPRSTP# Input High VIH(1.0V) 2.3 V 0.3 0.7 V V PWM PWM (PWM1-PWM3) Output Low VOL(5.0V) Sinking 5mA 1.0 V FCCM Output Low VOL_FCCM Sinking 3mA 1.0 V PWM (PWM1-PWM3) and FCCM Output High VOH(5.0V) Sourcing 5mA 3.5 PWM = 2.5V -1 NTC Source Current NTC = 1.3V 53 Over-Temperature Threshold V (NTC) falling 1.165 PWM Tri-State Leakage V 1 μA 60 67 μA 1.18 1.2 V 6.5 9 Ω THERMAL MONITOR VR_TT# Low Output Resistance RTT I = 20mA CLK_EN# High Output Voltage VOH 3V3 = 3.3V, I = -4mA CLK_EN# Low Output Voltage VOL I = 4mA CLK_EN# OUTPUT LEVELS 5 2.9 3.1 0.26 V 0.4 V FN9162.1 January 3, 2006 ISL6260, ISL6260B Typical Operating Performance 3 Phase, DCR Sense, (1) 7821, (2) 7832 per phase, 300kHz, 0.5µH 1.46 90 VOUT (V) Vout(V) Efficiency (%) Efficiency(%) 100 80 70 Vin = 8.0V Vin = 12.6V Vin = 19.0V 60 10 10 Vin = 8.0V 1.42 Vin = 12.6V 1.4 Vin = 19.0V 1.38 1.36 1.34 50 11 1.44 1.32 100 100 0 IOUT (A) 10 20 Iout(A) 40 50 Iout(A) FIGURE 1. ACTIVE MODE EFFICIENCY, 3 PHASE, CCM, PSI# = HIGH, VID = 1.4375V FIGURE 2. ACTIVE MODE LOAD LINE, 3 PHASE, CCM, PSI# = HIGH VID= 1.435V 100 1.44 90 80 70 Vout (V) Efficiency (%) Efficiency(%) 30 IOUT (A) Vin = 8.0V Vin = 12.6V 60 1.43 Vin = 8.0V 1.42 Vin = 12.6V Vin = 19.0V 1.41 1.4 1.39 1.38 Vin = 19.0V 1.37 50 1.36 0.1 1 10 0 10 IOUT (A) Iout(A) 20 FIGURE 3. DEEPER SLEEP MODE EFFICIENCY, 3 PHASE, DCM OPERATION, PSI# = LOW, VID = 1.4375V FIGURE 4. DEEPER SLEEP MODE LOAD LINE, 3 PHASE, CCM, PSI# = LOW VID= 1.435V 0.76 100 Vin = 8.0V 0.75 Vin = 12.6V 0.74 80 Vout(V) Efficiency (%) 90 70 Vin = 19.0V 0.73 0.72 0.71 0.7 V in = 8 .0 V 60 30 Iout (A) V in = 12 .6 V 0.69 V in = 19 .0 V 50 0.68 0.1 1 Iout (A) 10 0 10 20 30 Iout(A) FIGURE 5. DEEPER SLEEP MODE EFFICIENCY, 3 PHASE, DCM OPERATION, PSI# = LOW, VID = 0.75V 6 FIGURE 6. DEEPER SLEEP MODE LOAD LINE, 3 PHASE, DCM OPERATION, PSI# = LOW, VID = 0.75V FN9162.1 January 3, 2006 ISL6260, ISL6260B 100 100 90 90 Efficiency (%) Efficiency (%) Typical Operating Performance 3 Phase, DCR Sense, (1) 7821, (2) 7832 per phase, 300kHz, 0.5µH (Continued) 80 70 Vin = 8.0V Vin = 12.6V 60 Vin = 12.6V 70 Vin = 19.0V 50 1 10 Iout (A) 100 FIGURE 7. ACTIVE MODE EFFICIENCY, 2 PHASE, CCM, PSI# = HIGH, VID = 1.4375V 0.1 1 Iout (A) 10 FIGURE 8. DEEPER SLEEP MODE EFFICIENCY, 2 PHASE, DCM OPERATION, PSI# = LOW, VID = 1.4375V 100 1.44 Vin = 8.0V 1.42 90 Vin = 12.6V Vin = 19.0V 1.4 Vout (V) Efficiency (%) Vin = 8.0V 60 Vin = 19.0V 50 80 70 Vin = 8.0V Vin = 12.6V Vin = 19.0V 60 1.38 1.36 1.34 1.32 50 0.1 1 Iout (A) 0 10 10 20 30 40 50 Iout (A) FIGURE 9. DEEPER SLEEP MODE EFFICIENCY, 2 PHASE, DCM OPERATION, PSI# = LOW, VID = 0.75V FIGURE 10. ACTIVE MODE LOAD LINE, 2 PHASE, CCM, PSI# = HIGH, VID = 1.435V 0.76 1.44 1.43 1.42 Vin = 8.0V 0.75 Vin = 12.6V 0.74 Vin = 19.0V 1.41 Vout (V) Vout (V) 80 1.4 1.39 Vin = 8.0V Vin = 12.6V 0.73 Vin = 19.0V 0.72 0.71 1.38 0.7 1.37 0.69 0.68 1.36 0 10 20 30 Iout (A) FIGURE 11. DEEPER SLEEP MODE LOAD LINE, 2 PHASE, DCM OPERATION, PSI# = LOW, VID = 1.4375V 7 0 10 20 30 Iout (A) FIGURE 12. DEEPER SLEEP MODE LOAD LINE, 2 PHASE, DCM OPERATION, PSI# = LOW, VID = 0.75V FN9162.1 January 3, 2006 ISL6260, ISL6260B Typical Operating Performance [email protected] Vsoft [email protected] Vout VR_ON PGD_IN IMVP-6 Pgood CLK_EN# FIGURE 13. SOFT-START WAVEFORM SHOWING SLEW RATE OF 2mV/µs, 0V TO 1.2V (BOOT VOLTAGE) FIGURE 14. SOFT-START WAVEFORM SHOWING CLK_EN# AND IMVP-6 PGOOD Vout Vin Vout FIGURE 15. 12V-18V INPUT LINE TRANSIENT RESPONSE FIGURE 16. SOFT-START INRUSH CURRENT, VIN = 8V FIGURE 17. 3 PHASE CURRENT BALANCE, FULL LOAD = 50A FIGURE 18. 2 PHASE CURRENT BALANCE, FULL LOAD = 50A 8 FN9162.1 January 3, 2006 ISL6260, ISL6260B Typical Operating Performance (Continued) Vout COMP PIN FIGURE 19. TRANSIENT LOAD RESPONSE, 40A LOAD STEP @ 200A/µs, 3 PHASE FIGURE 20. TRANSIENT LOAD 3 PHASE OPERATION CURRENT BALANCE FIGURE 21. TRANSIENT LOAD 3 PHASE OPERATION, ZOOM OF RISING EDGE CURRENT BALANCE FIGURE 22. TRANSIENT LOAD 3 PHASE OPERATION, ZOOM OF FALLING EDGE CURRENT BALANCE VID MSB VID MSB Vout FIGURE 23. ISL6260, VID MSB BIT CHANGE FROM 1.4375V TO 0.65V SHOWING 9mV/µs SLEW RATE, DPRSLPVR = 0, DPRSTP# = 1 9 Vout FIGURE 24. SLEW RATE ENTERING C4, VID MSB BIT CHANGE FROM 1.4375V TO 0.65V SHOWING 2mV/µs SLEW RATE, DPRSLPVR = 1, DPRSTP# = 0 FN9162.1 January 3, 2006 ISL6260, ISL6260B Typical Operating Performance (Continued) Vout Vout @ 1.7V PWM DPRSTP# AND PSI# Vout @ 0.85V DPRSLPVR AND MSB FIGURE 25. C4 ENTRY AND EXIT SLEW RATES WITH DPRSLPVR AND DPRSTP# (ISL6260) FIGURE 26. 1.7V OVP SHOWING OUTPUT PULLED LOW TO 0.85V AND PWM TRI_STATE PWM PWM Vout Iphase Pgood Vout FIGURE 27. UNDERVOLTAGE RESPONSE SHOWING PWM TRI-STATE, VOUT < VID - 300mV Pgood FIGURE 28. OCP - RESPONSE PWM PSI# CLK_EN# Iphase Vout Vout Pgood FIGURE 29. OCP - SHORT CIRCUIT PROTECTION 10 Phase 2 FIGURE 30. ISL6260B, PHASE ADDING AND DROPPING IN ACTIVE MODE, LOAD CURRENT = 15A FN9162.1 January 3, 2006 ISL6260, ISL6260B Typical Operating Performance (Continued) Phase 3 current PSI# CLK_EN# Vout PHASE 1 CURRENT PHASE 2 CURRENT PHASE 2 PHASE 2 FIGURE 31. ISL6260B PHASE ADDING AND DROPPING IN DEEPER SLEEP MODE, LOAD CURRENT = 4.35A FIGURE 32. ISL6260B, INDUCTOR CURRENT WAVEFORM WITH PHASE ADDING AND DROPPING IN DCM OR DEEPER SLEEP MODE PHASE 3 CURRENT PHASE 1 CURRENT PHASE 1 CURRENT PHASE 2 CURRENT PHASE 2 CURRENT Pgood PHASE 2 FIGURE 33. ISL6260B, INDUCTOR CURRENT WAVEFORM WITH PHASE ADDING AND DROPPING IN CCM OR ACTIVE MODE 11 FIGURE 34. ISL6260B, OVERCURRENT DUE TO PHASE DROPPING FN9162.1 January 3, 2006 ISL6260, ISL6260B Simplified Application Circuit for DCR Current Sensing disabled, allows the regulator to operate in Diode Emulation for improved light load efficiency. Figure 35 shows a simplified application circuit for the ISL6260 or ISL6260B converter. Both the MOSFET driver IC and main control IC are shown. The driver has a forcecontinuous-conduction-mode (FCCM) input, that when V+5 VIN V+3.3 VIN VDD VIN 3V3 V+5 RBIAS VCC NTC PWM1 ISEN1 VR_TT# VR_TT# PWM BOOT LO UGATE ISL6208 PHASE SOFT 7 VID<0:6> VIDs DPRSTP# RL CL FCCM LGATE GND ISEN1 DPRSTP# VO' ISL6260 DPRSLPVR VSUM DPRSLPVR PSI# VIN VO PSI# V+5 PGD_IN MCHOK CLK_ENABLE# VCC PWM PWM2 CLK_EN# ISEN2 VR_ON VR_ON IMVP6_PWRGD CO BOOT LO UGATE ISL6208 PHASE PGOOD RL CL FCCM Remote Sense at CPU CORE VSEN LGATE GND ISEN2 RTN Ri VSUM FCCM C3 R3 V+5 VCC FB C1 VO' VIN VDIFF BOOT R1 PWM PWM3 ISEN3 COMP C2 LO UGATE ISL6208 RFSET VSUM VSUM VW PHASE FCCM LGATE GND OCSET GND DFB DROOP VO RL CL ISEN3 VSUM RN VO' CC S VO' FIGURE 35. TYPICAL APPLICATION CIRCUIT FOR DCR SENSING 12 FN9162.1 January 3, 2006 ISL6260, ISL6260B Simplified Application Circuit for Resistive Current Sensing continuous-conduction-mode (FCCM) input, that when disabled, allows the regulator to operate in Diode Emulation for improved light load efficiency. Figure 36 shows a simplified application circuit for the ISL6260 or ISL6260B converter. Both the MOSFET drivers and main control IC are shown. The driver has a force V+5 V+3.3 VIN VIN VDD VIN 3V3 V+5 RBIAS VCC NTC PWM1 ISEN1 VR_TT# VR_TT# PWM BOOT LO UGATE RSEN ISL6208 SOFT 7 VID<0:6> PHASE VIDs LGATE GND DPRSTP# DPRSTP# VSUM FCCM CL VO' ISL6260 DPRSLPVR DPRSLPVR PSI# RL ISEN1 VIN VO PSI# V+5 PGD_IN MCHOK CLK_ENABLE# CLK_EN# VCC PWM2 ISEN2 PWM VR_ON VR_ON IMVP6_PWRGD CO BOOT LO UGATE RSEN ISL6208 PHASE PGOOD VSUM FCCM Remote Sense at CPU CORE VSEN LGATE GND ISEN2 RTN FCCM VIN VDIFF R3 C1 R1 CL VO' Ri C3 RL V+5 VCC FB BOOT PWM3 ISEN3 COMP PWM C2 UGATE LO RSEN ISL6208 RFSET PHASE VW VSUM VSUM FCCM LGATE GND OCSET GND DFB DROOP VSUM ISEN3 VO RL CL VO' VO' FIGURE 36. TYPICAL APPLICATION CIRCUIT FOR DISCRETE RESISTOR CURRENT SENSING 13 FN9162.1 January 3, 2006 Functional Block Diagram RBIAS PGOOD PGD_IN 3V3 CLK_EN# VIN VDD ISEN1 ISEN2 ISEN3 VIN VO VID0 PROTECTION VID2 14 VID3 54µA POWER VID1 IBAL CLK_EN# LOGIC GOOD MONITOR BALANCE DAC DACOUT VID4 IBAL VDIFF OC NTC FLT VID5 VR_TT# VID6 - FAST_OC OR WAY-OC + 1.18V OC MODE CONTROL FCCM VIN VO 1.2V + 2X VO VR_ON FLT MODULATOR MODE PWM1 SOFT CONTROL DPRSLPVR DPRSTP# OC NUMBER OF PHASES 10µA VIN VO GAIN SELECT OCSET FLT MODULATOR (ISL6260B ONLY) - (1, 2/3, or 1/2) PWM2 OC + VSUM DFB OC + DROOP - DROOP + - VO PWM3 E/A + - VO NUMBER OF PHASES + 1 - + CLOCK MODE CONTROL VIN VO V O VSEN RTN FLT MODULATOR + 1 VIN VDIFF SOFT FB COMP VW FN9162.1 January 3, 2006 FIGURE 37. SIMPLIFIED BLOCK DIAGRAM GND CHANNEL SELECT ISL6260, ISL6260B PSI# 6µA CURRENT ISL6260, ISL6260B Theory of Operation VDD Operational Description The ISL6260 and ISL6260B are multiphase regulators implementing Intel® IMVP-6 protocol. They can be programmed for one-, two- or three-channel operation for microprocessor core applications up to 70A. With their mating gate driver, the ISL6208, the SL6260 and ISL6260B give optimum steady-state and transient performance. At the heart of the ISL6260 and ISL6260B is the patented R3 (Robust Ripple Regulator®) modulator. The R3® modulator combines the best features of fixed frequency PWM and hysteretic PWM while eliminating many of their shortcomings. The ISL6260 and ISL6260B modulator internally synthesize an analog of the inductor ripple currents and use hysteretic comparators on those signals to determine PWM pulse widths. Operating on these largeamplitude, noise-free synthesized signals allows the ISL6260 and ISL6260B to achieve lower output ripple and lower phase jitter than conventional hysteretic and fixed PWM controllers. Unlike conventional hysteretic converters, the ISL6260 and ISL6260B have an error amplifier that allows the controller to maintain a 0.5% voltage tolerance. The hysteresis window voltage rides on the error amplifier output such that a load current transient results in an increase in switching frequency to give the R3 regulator a faster response than conventional fixed frequency PWM controllers. The sharing of the hysteretic window voltage also inherently shares the transient load current between the phases. The individual average phase voltages are monitored and controlled to equally share the static current among the phases. ISL6260B disables PWM2 when PSI# is asserted low. ISL6260 does not drop phase with PSI# signal. VR_ON 100µs 10mV/µs 2mV/µs VBOOT VID COMMANDED VOLTAGE SOFT & VO 20µs PGD_IN CLK_EN# IMVP-6 PGOOD ~7ms FIGURE 38. SOFT-START WAVEFORMS USING A 20nF SOFT CAPACITOR Start-up Timing With the controller's +5V VDD voltage above the POR threshold, the start-up sequence begins when VR_ON exceeds the 3.3V logic HIGH threshold. Approximately 100µs later SOFT and VOUT start ramping up to the boot voltage of 1.2V. During this interval, the SOFT capacitor is charged with approximately 40µA. Therefore, if the SOFT capacitor is selected to be 20nF, the SOFT ramp will be at about 2mV/µs for a soft-start time of 600µs. Once VOUT is within 10% of the boot voltage and PGD_IN is HIGH for six PWM cycles (20µs for frequency = 300kHz), then CLK_EN# is pulled LOW and the SOFT capacitor is charged up with approximately 200µA. Therefore, VOUT slews at +10mV/µs to the voltage set by the VID pins. Approximately seven milliseconds later, PGOOD is asserted HIGH. A typical startup timing is shown in Figure 38. Similar results occur if VR_ON or PGD_IN or both are tied to VDD, with the softstart sequence starting 120µs after VDD crosses the POR threshold. PGD_IN - LATCH It should be noted that PGD_IN going low will cause the converter to latch off. This state will be cleared when VR_ON is toggled. This feature allows the converter to respond to other system voltage outages immediately. For ISL6260B only, PGD_IN de-assertion (0) during normal operation will make CLK_EN# go high. 15 FN9162.1 January 3, 2006 ISL6260, ISL6260B Static Operation After the start sequence, the output voltage will be regulated to the value set by the VID inputs per Table 1. This Table is presented in its entirety in the Intel IMVP-6™ specification. The ISL6260, ISL6260B will control the no-load output voltage to an accuracy of ±0.5% over the range of 0.75V to 1.5V. TABLE 1. TRUNCATED VID TABLE FOR INTEL IMVP-6™ SPECIFICATION VID6 VID5 VID4 VID3 VID2 VID1 VID0 VOUT 0 0 0 0 0 0 0 1.500V 0 0 0 0 0 0 1 1.4875 0 0 0 0 1 0 1 1.4375 0 0 0 0 1 1 1 1.4125 0 0 0 1 0 0 0 1.4000 0 0 1 0 0 0 1 1.2875 0 0 1 1 0 0 0 1.2000 0 0 1 1 1 0 0 1.1500 0 1 0 1 0 0 0 1.0000 0 1 0 1 0 1 1 0.9625 0 1 1 1 1 0 0 0.7500 1 0 0 0 1 0 0 0.6500 1 0 1 0 0 0 0 0.5000 1 1 0 0 0 0 0 0.300 1 1 0 0 0 0 1 Off 1 1 0 0 0 1 0 Off ... Off 1 1 1 1 1 1 0 Off 1 1 1 1 1 1 1 Off A fully-differential amplifier allows CPU die voltage sensing for precise voltage control at the microprocessor die. The inputs to the amplifier are the VSEN and RTN pins. As the load current increases from zero, the output voltage will droop from the VID table value by an amount proportional to current to achieve the IMVP-6 load line. The ISL6260 and ISL6260B provide for current to be measured using resistors in series with the channel inductors as shown in the application circuit of Figure 36 or using the intrinsic series resistance of the inductors as shown in the application circuit of Figure 35. In both cases, signals representing the inductor currents are summed at VSUM which is the noninverting input to the DROOP amplifier shown in the block diagram of Figure 37. The voltage at the DROOP pin minus the output voltage, VO´ is thus a high-bandwidth analog of the total inductor current. This value is used as an input to the differential amplifier to achieve the IMVP-6 load line as well as the input to the overcurrent circuit. 16 When using inductor DCR current sensing, a single NTC element is used to compensate the positive temperature coefficient of the copper winding thus sustaining the load-line accuracy. Procedures to follow in determining component values are covered in the “Component Selection and Application” section of the datasheet. In addition to the total current which is used for DROOP and OC, the individual channel average currents are also monitored and are used for balancing the load between the channels. The IBAL circuit will adjust the channel pulsewidths up or down relative to the other channels to cause the voltages presented to the ISEN pins to be equal. The ISL6260 and ISL6260B controller can be configured for three-, two- or single-channel operation. To disable channel two and/or channel three, its PWM output pin should be tied to +5V VDD and the ISEN pins should be grounded. If the ISL6208 gate driver is populated in an unused channel, its PWM input pin should be opened in order to turn off its output. In three-channel operation, the three channel PWM's are 120 degrees apart, and in two-channel operation they are 180 degrees apart. The channel PWM frequency is determined by the value of RFSET as shown in the “Component Selection and Application” section of this document. If the controller is kept in continuous conduction mode (CCM), the switching frequency may not be constant but it can maintain the switching ripple within spec. However, it will be very close to the set value at high input voltage and heavy load conditions. Selected by setting DPRSLPVR high, DPRSTP# low, together with PSI# signal (see Table 2), discontinuous conduction mode (DCM) is allowed. In DCM, the ISL6260, ISL6260B commands the ISL6208 to turn off the lower FET after its channel current decays to zero. As load is further reduced, channel switching frequency will drop, providing optimized efficiency even at light loading. Dynamic Operation Refer to Figure 39. The ISL6260 and ISL6260B respond to changes in VID command voltage by slewing to new voltages with a dV/dt set by the SOFT capacitor and by the state of DPRSLPVR. With CSOFT = 20nF and DPRSLPVR HIGH, the output voltage will move at ±2mV/µs for large changes in voltage. For DPRSLPVR LOW, the large signal dV/dt will be ±10mV/µs. As the output approaches the VID command voltage, the dV/dt rate moderates to prevent overshoot. During Geyserville III transitions where there is one LSB VID step each 5µs, the controller will follow the VID command with its dV/dt rate of ±2.5mV/µs. FN9162.1 January 3, 2006 ISL6260, ISL6260B Vout 10mV/us -2mV/us DPRSTP# PSI# When PSI# is de-asserted low, ISEN2 pin is connected to the ISEN pins of the operational phases internally to keep proper current balance and less current overshoot and undershoot when the disabled phase is enabled again. 2mV/us TABLE 2. ISL6260 ISL6260B MODE OF OPERATIONS DPRS DPR LPVR STP# PSI# IMVP-6 Logic DPRSLPVR MSB of VID FIGURE 39. DEEPER SLEEP TRANSITION SHOWING DPRSLPVR’s EFFECT ON EXIT SLEW RATE Keeping DPRSLPVR HIGH during VID transitions will result in reduced dV/dt output voltage changes with resulting minimized audio noise. For fastest recovery from Deeper Sleep to Active mode, DPRSLPVR LOW achieves maximum dV/dt. Therefore, the ISL6260 and ISL6260B are IMVP-6 compliant for DPRSTP# and DPRSLPVR logic. Intersil's R3 intrinsically has voltage-feed-forward. Highspeed input voltage steps result in insignificant output voltage perturbations. Refer to Figure 15 in the “Typical Operating Performance” section of this document for Input Transient Performance. In response to load current step increases, the ISL6260 and ISL6260B will transiently raise the switching frequency so that response time is decreased and current is shared by all the channels. Modes of Operation Programmed by Logic Signals The operational modes of ISL6260 and ISL6260B are related to the control signals of DPRSLPVR, DPRSTP#, and PSI#. ISL6260B responds PSI# signal by adding or dropping PWM2 and adjusting the overcurrent protection level accordingly. ISL6260 does not drop phases while in operation. For example, if the ISL6260B is initially used as three phase, the PSI# signal will add or drop PWM2 and leave PWM1 and PWM3 always in operation. Meanwhile, after PWM2 is dropped, the phase shift between the PWM1 and PWM3 is adjusted from 120 degree to 180 degree and the overcurrent and the way-overcurrent protection level will be adjusted to 2/3 of the initial value. If the ISL6260B is initially used as two phase operation, it is suggested that PWM1 and PWM2 pair, not PWM1 and PWM3 pair, should be used such that PSI# signal will enable or disable PWM2 with PWM1 in operation always. The overcurrent and wayovercurrent protection level in two-to-one phase mode operation will be adjusted as two to one as well. For ISL6260B, the DCM mode is independent of PSI#, it just responds to the DPRSLPVR and DPRSTP#. The following table shows the operation modes of ISL6260 and ISL6260B with combinations of control logic. 17 Other Logic 0 1 1 ISL6260 ISL6260B N phase CCM N phase CCM CPU MODE Active 0 1 0 N phase CCM N-1 phase CCM Active 1 0 1 N phase CCM N phase DCM 1 0 0 N phase DCM N-1 phase DCM Deeper sleep 0 0 1 N phase CCM N phase CCM 0 0 0 N phase CCM N-1 phase CCM 1 1 1 N phase CCM N phase CCM 1 1 0 N phase CCM N-1phase CCM Deeper sleep Protection The ISL6260 and ISL6260B provide overcurrent, overvoltage, and undervoltage protection. Overcurrent protection is tied to the voltage droop which is determined by the resistors selected as described in the “Component Selection and Application” section. After the load-line is set, the OCSET resistor can be selected to detect overcurrent at any level of droop voltage. For overcurrent less that twice the OCSET level, the overload condition must exist for 120µs in order to trip the OC fault latch. This is shown in Figure 28. For overload exceeding twice the set level, the PWM outputs will immediately shut off and PGOOD will go low to maximize protection due to hard shorts. This protection was referred to as way-overcurrent. In addition, excessive phase unbalance due to gate driver failure will be detected and will shut down the controller after 1ms. The phase unbalance is detected by the voltage on the ISEN pins if the difference is greater than 9mV. Undervoltage protection is independent of the overcurrent limit. If the output voltage is less than the VID set value by 300mV or more, a fault will latch after 1ms in that condition. The PWM outputs will turn off and PGOOD will go low. This is shown in Figure 27. Note that most practical core regulators will have the overcurrent set to trip before the 300mV undervoltage limit. There are two levels of overvoltage protection and response. For output voltage exceeding the set value by +200mV for 1ms, a fault is declared. All of the above faults have the same action taken: PGOOD is latched low and the upper and lower power FETs are turned off so that inductor current will decay through the FET body diodes. This condition can be reset by bringing VR_ON low or by bringing VDD below POR threshold. When these inputs are returned to their high operating levels, a soft-start will occur. FN9162.1 January 3, 2006 ISL6260, ISL6260B TABLE 3. SUMMARY OF THE FAULT PROTECTION AND RESET OPERATIONS OF ISL6260, ISL6260B FAULT DURATION PRIOR TO PROTECTION PROTECTION ACTIONS FAULT RESET Overcurrent 120µs PWMs tri-state, Pgood latched low VR_ON toggle or VDD toggle Way-Overcurrent <2µs PWMs tri-state, Pgood latched low VR_ON toggle or VDD toggle Overvoltage 1.7V Immediately Low side MOSFET on until Vcore <0.85V, then PWM tri-state, Pgood latched low. VDD toggle Overvoltage +200mV 1ms PWMs tri-state, Pgood latched low VR_ON toggle or VDD toggle Undervoltage -300mV 1ms PWMs tri-state, Pgood latched low VR_ON toggle or VDD toggle Phase Current Unbalance 1ms PWMs tri-state, Pgood latched low VR_ON toggle or VDD toggle Over Temperature Immediately VR_TT# goes low N/A Refer to Figure 26. The second level of overvoltage protection behaves differently. If the output exceeds 1.7V, an OV fault is immediately declared, PGOOD is latched low and the low-side FETs are turned on. The low-side FETs will remain on until the output voltage is pulled down below 0.85V at which time all FETs are turned off. If the output again rises above 1.7V, the process is repeated. This affords the maximum amount of protection against a shorted highside FET while preventing output ringing below ground. The 1.7V OV is not reset with VR_ON, but requires that VDD be lowered to reset. The 1.7V OC detector is active at all times that the controller is enabled including after one of the other faults occurs so that the processor is protected against highside FET leakage while the FETs are commanded off. The ISL6260 and ISL6260B have a thermal throttling feature. If the voltage on the NTC pin goes below the 1.18V OT threshold, the VR_TT# pin is pulled low indicating the need for thermal throttling to the system oversight processor. No other action is taken within the ISL6260 and ISL6260B in response to NTC pin voltage. Fault protection is summarized in Table 3. Component Selection and Application Soft-Start and Mode Change Slew Rates The ISL6260 and ISL6260B use 2 slew rates for various modes of operation. The first is a slow slew rate, used to reduce inrush current on start-up. It is also used to reduce audible noise when entering or exiting Deeper Sleep Mode. A faster slew rate is used to exit out of Deeper Sleep and to increase system performance by achieving active mode regulation more quickly. Note that the SOFT cap current is bidirectional and is flowing into the SOFT capacitor when the output voltage is commanded to rise, and out of the SOFT capacitor when the output voltage is commanded to fall. The two slew rates are determined by commanding 1 of 2 currents onto the SOFT pin. As can be seen in Figure 40, the SOFT pin has a capacitance to ground. Also, the SOFT pin is the input to the error amplifier and is, therefore, the 18 commanded system voltage. Depending on the state of the system, i.e. Start-Up or Active mode, and the state of the DPRSLPVR pin, one of the two currents shown in Figure 40 will be used to charge or discharge this capacitor, thereby controlling the slew rate of the commanded voltage. These currents can be found under the Soft Current section of the Electrical Specification Table. ISL6260(B) ISS I2 ERROR AMPLIFIER + SOFT CSOFT + VREF FIGURE 40. SOFT PIN CURRENT SOURCES FOR FAST AND SLOW SLEW RATES The first current, labeled ISS, is given in the Specification Table as 41μA. This current is used during Soft-Start. The second current, I2 sums with ISS to get the larger of the two currents, labeled IGV in the Electrical Specification Table. This total current is typically 200μA with a minimum of 175μA. The IMVP-6™ specification reveals the critical timing associated with regulating the output voltage. The symbol, Slewrate, as given in the IMVP-6™ specification, will determine the choice of the SOFT capacitor, CSOFT, by the following equation: I GV C SOFT = -----------------------------------SLEWRATE FN9162.1 January 3, 2006 ISL6260, ISL6260B Using a SLEWRATE of 10mV/μs, and the typical IGV value, given in the Electrical Specification Table of 200μA, CSOFT is 200μA C SOFT = ------------------ = 0.020μF 10mV ---------------1μs (EQ. 2) regulation, as indicated by PGD_IN going high, CLK_EN# goes low, triggering an internal timer for the IMVP6_PWRGD signal. This timer allows IMVP6_PWRGD to go high approximately 7ms after CLK_EN# goes low. Static Mode of Operation - Processor Die Sensing A choice of 0.015μF would guarantee a SLEWRATE of 10mV/μs is met for minimum IGV value, given in the Electrical Specification Table. Now this choice of CSOFT will then control the start-up slewrate as well. One should expect the output voltage to slew to the Boot value of 1.2V at a rate given by the following equation: I SS 41μA dV mV - = ----------------------- = 2.73 --------------- = -----------------0.015μF C SOFT dt μs (EQ. 3) Generally, when output voltage is approaching its steady state, its dv/dt will slow down to prevent overshoot. In order to compensate the slow-down effect, faster initial dv/dt slew rates can be used with small soft capacitors such as 10nF to achieve the desired overall dv/dt in the allocated time interval. Selecting RBIAS To properly bias the ISL6260 and ISL6260B, a reference current is established by placing a 147kΩ, 1% tolerance resistor from the RBIAS pin to ground. This will provide a highly accurate, 10μA current source from which OCSET reference current can be derived. Care should be taken in layout that the resistor is placed very close to the RBIAS pin and that a good quality signal ground is connected to the opposite side of the RBIAS resistor. Do not connect any other components to this pin as this would negatively impact performance. Capacitance on this pin would create instabilities and is to be avoided. Start-up Operation - CLK_EN# and PGOOD The ISL6260 and ISL6260B provide a 3.3V logic output pin for CLK_EN#. The 3V3 pin allows for a system 3.3V source to be connected to separated circuitry inside the ISL6260 and ISL6260B, solely devoted to the CLK_EN# function. The output is a 3.3V CMOS signal with 4mA of source and sinking capability. This implementation removes the need for an external pull-up resistor on this pin, and due to the normal level of this signal being a low, removes the leakage path from the 3.3V supply to ground through the pull-up resistor. This reduces 3.3V supply current, that would occur under normal operation with a pull-up resistor, and prolongs battery life. The 3.3V supply should be decoupled to digital ground, not to analog ground for noise immunity. As mentioned in the “Theory of Operation” section of this datasheet, CLK_EN# is logic level high at start-up until 20μs after the system Vccp and Vcc_mch supplies are within regulation, and the Vcc-core is in regulation at the Boot level. Approximately 20μs after these voltages are within 19 Die sensing is the ability of the controller to regulate the Core output voltage at a remotely sensed point. This allows the Voltage Regulator to compensate for various resistive drops in the power path and insure that the voltage seen at the CPU die is the correct level independent of load current. The VSEN and RTN pins of the ISL6260 and ISL6260B are connected to Kelvin sense leads at the die of the processor through the processor socket. These signal names are Vcc_sense and Vss_sense respectively. This allows the Voltage Regulator to tightly control the processor voltage at the die, independent of layout inconsistencies and drops. This Kelvin sense technique provides for extremely tight load line regulation. These traces should be laid out as noise sensitive traces. For optimum load line regulation performance, the traces connecting these two pins to the Kelvin sense leads of the processor must be laid out away from rapidly rising voltage nodes (switching nodes) and other noisy traces. To achieve optimum performance, place common mode and differential mode RC filters to analog ground on VSEN and RTN as shown in Figure 42. The filter resistors should be in order of 10Ω so that they do not interact with the 50kΩ input resistance of the differential amplifier. Due to the fact that the voltage feedback to the switching regulator is sensed at the processor die, there exists the potential of an overvoltage due to an open circuited feedback signal, should the regulator be operated without the processor installed. Due to this fact, we recommend the use of the Ropn1 and Ropn2 connected to Vout and ground as shown in Figure 42. These resistors will provide voltage feedback in the event that the system is powered up without a processor installed. These resistors are typically 100Ω. Setting the Switching Frequency - FSET The R3 modulator scheme is not a fixed frequency PWM architecture. The switching frequency can increase during the application of a load to improve transient performance. However, it also varies slightly due changes in input and output voltage and output current, but this variation is normally less than 10% in continuous conduction mode. Refer to Figure 35. A resistor connected between the VW and COMP pins of the ISL6260 and ISL6260B adjusts the switching window, and therefore adjusts the switching frequency. The Rfset resistor that sets up the switching frequency of the converter operating in CCM can be FN9162.1 January 3, 2006 ISL6260, ISL6260B determined using the following relationship, where Rfset is in kΩ and the switching period is in μs. (EQ. 4) Rfset ( kΩ ) = ( Period ( μs ) – 0.29 ) × 2.33 In discontinuous conduction mode, (DCM), the ISL6260, ISL6260B runs in period stretching mode. It should be noted that the switching frequency in the Electric Table is tested with the error amplifier output or Comp pin voltage at 2V. When Comp pin voltage is lower, the switching frequency will not be at the tested value. Voltage Regulator Thermal Throttling lntel® IMVP-6® technology supports thermal throttling of the processor to prevent catastrophic thermal damage to the voltage regulator. The ISL6260 and ISL6260B feature a thermal monitor which senses the voltage change across an externally placed negative temperature coefficient (NTC) thermistor, see Figure 41. Proper selection and placement of the NTC thermistor allows for detection of a designated temperature rise by the system. Figure 41 shows the thermal throttling feature with hysteresis. At low temperature, SW1 is on and SW2 connects to the 1.18V side. The total current going from NTC pin is 60µA. The voltage on NTC pin is higher than threshold voltage of 1.18V and the comparator output is low. VR_TT# is pulling up high by the external resistor. 54µA 6µA + 1.20V Therefore, proper NTC thermistor has to be chosen such that 2.56K resistor change will be corresponding to required temperature hysteresis. Regular external resistor may need to be in series with NTC resistors to meet the threshold voltage values. The following is an example. For Panasonic NTC with B = 4700, its resistance will drop to 0.03322 of its nominal at 105°C, and drop to 0.03956 of its nominal at 100 C°. If the requirement for the temperature hysteresis is (105-100) C°, the required resistance of NTC will be: 2 . 56 K Ω ( 0 . 03956 − 0 . 03322 ) = 404 K Ω Therefore a larger value thermistor, such as 470 K NTC should be used. At 105°C, 470K NTC resistance becomes (0.03322*470K) = 15.6K. With 60µA on NTC pin, the voltage is only (15.6K*60µA) = 0.937V. This value is much lower than the threshold voltage of 1.18V. Therefore, a resistor is needed to be in series with the NTC. The required resistance can be calculated by: 4.02K is a standard resistor value. Therefore, the NTC branch should have a 470K NTC and 4.02K resistor in series. The part number for the NTC thermistor is ERTJ0EV474J. It is a 0402 package. NTC thermistor will be placed in the hot spot of the board. A thermistor in an 0402 package costs less than in an 0603 package. - Rs = 2 . 56 K VR_TT# SW1 RNTC 1 .2 V 1 . 18 V − 54 μ A 60 μ A 1 . 18 V − 15 . 6 K Ω = 4 . 06 K Ω 60 uA NTC + VNTC - eventually go up. If NTC voltage increases to 1.20V, the comparator will then be able to flip back. The external resistance difference in these two conditions is: SW2 1.18V INTERNAL TO ISL6260 FIGURE 41. CIRCUITRY ASSOCIATED WITH THE THERMAL THROTTLING FEATURE OF THE ISL6260 When temperature increases, the NTC resistor on NTC pin decreases. The voltage on NTC pin decreases to a level lower than 1.18V. The comparator changes polarity and turns SW1 off and throws SW2 to 1.20V. This pulls VR_TT# low and sends the signal to start thermal throttle. There is a 6µA current reduction on NTC pin and 20mV voltage increase on threshold voltage of the comparator in this state. The VR_TT# signal will be used to change the CPU operation and decrease the power consumption. When the temperature goes down, the NTC thermistor voltage will 20 FN9162.1 January 3, 2006 ISL6260, ISL6260B IS E N 1 IS E N 2 10uA OCSET RTN VSEN VO' V D IF F RS2 VSUM Rpar + - RO1 VO' L2 R L2 DCR + V d c r2 RO2 C L2 VO' Ip h a s e 3 Rntc 1 IS E N 1 IS E N 2 Cn + DROOP + - Rdrp2 Σ 1 DFB Rdrp1 In te r n a l to IS L 6 2 6 0 C L1 R L1 Ip h a s e 2 VSUM Rseries + DROOP - L3 + V d cr3 RS3 R L3 0 .0 1 u F IS E N 3 10 0 .2 2 u F Ropn1 VC C _S EN S E VSS _S EN S E Ropn2 Vout - DCR VSUM VO' - DCR VSUM VO' VSUM + V d c r1 + RS1 R OCSET + OC L1 IS E N 3 IS E N 2 IS E N 1 Ip h a s e 1 IS E N 3 C L3 RO3 C b u lk VO' to V o u t ESR T o P ro ce ss o r S o c k e t K e lv in C o n n e c tio n s FIGURE 42. EQUIVALENT MODEL FOR DROOP AND DIE SENSING USING DCR SENSING Static Mode of Operation - Static Droop using DCR Sensing As previously mentioned, the ISL6260 and ISL6260B have an internal differential amplifier which provides for extremely accurate voltage regulation at the die of the processor. The load line regulation is also very accurate, and the process of selecting the components for the appropriate load line droop is explained here. Figure 43 shows the simplified model of the droop circuitry. Essentially one resistor can replace the RO resistors of each phase and one RS resistor can replace the RS resistors of each phase. The total DCR drop due to load current can be replaced by a DC source, the value of which is given by Equation 5. Iout × DCR Vdcr EQV = ------------------------------N (EQ. 5) For DCR sensing, the process of compensation for DCR resistance variation to achieve the desired load line droop has several steps and is somewhat iterative. Refer to Figure 42. where N is the number of channels designed for Active operation. N = 3 for this example. Another simplification can be done by reducing the NTC network comprised of Rntc, Rseries and Rparallel, given in Figure 43, to a single resistor given as Rn. In Figure 42 we show a 3 phase solution using DCR sensing. There are two resistors around the inductor of each phase. These are labeled RS and RO. These resistors are used to arrive at the DC voltage drop across each inductor. Each inductor will have a certain level of DC current flowing through it, this current when multiplied by the DCR of the inductor creates a small DC level of voltage. When this voltage is summed with the other channels DC voltages, the total DC load current can be derived. The first step in droop load line compensation is to adjust Rn, ROEQV and RSEQV such that sufficient droop voltage exists even at light loads between the VSUM and VO’ nodes. We recognize that these components form a voltage divider. As a rule of thumb we start with the voltage drop across the Rn network, VN, to be 0.57 x Vdcr. This ratio provides for a fairly reasonable amount of light load signal from which to arrive at droop. RO is typically 5 to 10Ω. This resistor is used to tie the outputs of all channels together and thus create a summed average of the local CORE voltage output. RS is determined through an understanding of both the DC and transient load currents. This value will be covered in the next section. First we calculate the equivalent NTC network resistance, Rn. Typical values that provide good performance are, Rseries = 3.57K_1%, Rpar = 4.53K_1% and Rntc = 10kΩ NTC, ERT-J1VR103J from Panasonic. Rn is then given by Equation 6. However, it is important to keep in mind that the output of each of these RS resistors are tied together to create the VSUM voltage node. With both the outputs of RO and RS tied together, the simplified model for the droop circuit can be derived. This is presented in Figure 43. 21 ( Rseries + Rntc ) × Rpar Rn = -------------------------------------------------------------------- = 3.4kΩ Rseries + Rntc + Rpar (EQ. 6) In our second step we calculate the series resistance from each phase to the Vsum node, labeled RS1, RS2 and RS3 in Figure 42. FN9162.1 January 3, 2006 ISL6260, ISL6260B 10uA OCSET + OC RS EQV = VSUM + DROOP - Vdcr EQV = Iout × DROOP RTN VDIFF VSEN VO' Cn + 1 - + + Rdrp1 Σ + 1 - + DFB Rdrp2 Internal to ISL6260 VSUM RS N VN Rn = DCR N (Rntc + Rseries ) × Rpar (Rntc + Rseries ) + Rpar VO' RO EQV = RO N FIGURE 43. EQUIVALENT MODEL FOR DROOP AND DIE SENSING USING DCR SENSING We do this using the assumption that we desire approximately a 0.57 gain from the DCR voltage, Vdcr, to the Rn network. We call this gain, G1. (EQ. 7) G1 = 0.57 After simplification, then RSEQV is given by the following equation: 1 RS EQV = ⎛ -------- – 1⎞ Rn = 2.56kΩ ⎝ G1 ⎠ (EQ. 8) The individual resistors from each phase to the VSUM node, labeled RS1, RS2 and RS3 in Figure 42, are then given by Equation 9, where N is 3, for the number of channels in active operation. RS = N × RS EQV = 7.69kΩ (EQ. 9) Choosing RS = 7.68k_1% is a good choice. Once we know the attenuation of the RS and RN network, we can then determine the Droop amplifier Gain required to achieve the load line. Setting Rdrp1 = 1k_1%, then Rdrp2 is can be found using Equation 10. N × Rdroop Rdrp2 = ⎛ -------------------------------- – 1⎞ × Rdrp1 ⎝ DCR × G1 ⎠ (EQ. 10) Setting N = 3 for 3 channel operation, Droop Impedance (Rdroop) = 0.0021 (V/A) as per the Intel IMVP-6 specification, DCR = 0.0012Ω typical, Rdrp1 = 1kΩ and the attenuation gain (G1) = 0.57, Rdrp2 is then 3 × 0.0021 Rdrp2 = ⎛ ------------------------------------ – 1⎞ × 1K = 8.21kΩ ⎝ 0.0012 × 0.57 ⎠ 22 (EQ. 11) Rdrp2 is selected to be a 8.25k_1% resistor. Note, we choose to ignore the RO resistors because they do not add significant error. These values are extremely sensitive to layout and coupling factor of the NTC to the inductor. As only one NTC is required in this application, this NTC should be placed as close to the Channel 1 inductor as possible and PCB traces sensing the inductor voltage should be go directly to the inductor pads. Once the board has been laid out, some adjustments may be required to adjust the full load droop voltage. This is fairly easy and can be accomplished by allowing the system to achieve thermal equilibrium at full load, and then adjusting Rdrp2 to obtain the appropriate load line slope. To see whether the NTC has compensated the temperature change of the DCR, the user can apply full load current and wait for the thermal steady state and see how much the output voltage will deviate from the initial voltage reading. A good compensation can limit the drift to 2mV. If the output voltage is decreasing with temperature increase, that ratio between the NTC thermistor value and the rest of the resistor divider network has to be increased. The user should follow the component values and layout of NTC on evaluation board as much as possible to minimize engineering time. The 2.1mV/A load line should be adjusted by Rdrp2 based on maximum current steps, not based on small current steps like 10A, as the droop gain might vary slightly between each 10A steps. Basically, if the max current is 40A, the required droop voltage is 84mV. The user should have 40A load current on the converter and look for 84mV droop. If the droop voltage is FN9162.1 January 3, 2006 ISL6260, ISL6260B less than 84mV, for example, 80mV. The new value will be calculated by: Rdrp 2 _ new = 84 mV ( Rdrp 1 + Rdrp 2 ) − Rdrp 1 80 mV For the best accuracy, the effective resistance on the DFB and VSUM pins should be identical so that the bias current of the droop amplifier does not cause an offset voltage. In the example above, the resistance on the DFB pin is Rdrp1 in parallel with Rdrop2, that is, 1K in parallel with 8.21K or 890Ω. The resistance on the VSUM pin is Rn in parallel with RSeqv or 3.4K in parallel with 2.56K or 1460Ω. The mismatch in the effective resistances is 1460-890 = 570Ω. Do not let the mismatch get larger than 600Ω. To reduce the mismatch, multiply both Rdrp1 and Rdrp2 by the appropriate factor. The appropriate factor in the example is 1460/890 = 1.64. Dynamic Mode of Operation - Dynamic Droop using DCR Sensing Droop is very important for load transient performance. If the system is not compensated correctly, the output voltage could sag excessively upon load application and potentially create a system failure. The output voltage could also take a long period of time to settle to its final value. This could be problematic if a load dump were to occur during this time. This situation would cause the output voltage to rise above the no load setpoint of the converter and could potentially damage the CPU. The L/DCR time constant of the inductor must be matched to the Rn*Cn time constant as shown in the following equation: ⎛ Rn × RS EQV⎞ L -⎟ × Cn ------------- = ⎜ ---------------------------------DCR ⎝ Rn + RS EQV⎠ (EQ. 12) Solving for Cn we now have the following equation: L ------------DCR Cn = ----------------------------------------⎛ Rn × RS EQV⎞ ⎜ -----------------------------------⎟ ⎝ Rn + RS EQV⎠ (EQ. 13) Note, RO was neglected. As long as the inductor time constant matches the Cn, Rn and Rs time constants as given above, the transient performance will be optimum. As in the Static Droop Case, this process may require a slight adjustment to correct for layout inconsistencies. For the example of L = 0.5μH, Cn is calculated below. 0.5μH -----------------0.0012 Cn = ------------------------------------------------- = 28.5nF 3.4kΩ × 2.56kΩ⎞ ⎛ -----------------------------------------⎝ 3.4kΩ + 2.56kΩ⎠ (EQ. 14) The value of this capacitor is selected to be 27nF. As the inductors tend to have 20% to 30% tolerances, this cap generally will be tuned on the board by examining the transient voltage. If the output voltage transient has an initial dip, lower than the voltage required by the load line, and slowly increases back to the steady state, the cap is too 23 small and vice versa. It is better to have the cap value a little bigger to cover the tolerance of the inductor to prevent the output voltage from going lower than the spec. This cap needs to be a high grade cap like X7R with low tolerance. There is another consideration in order to achieve better time constant match mentioned above. The NPO/COG (class-I) capacitors have only 5% tolerance and a very good thermal characteristics. But those caps are only available in small capacitance values. In order to use such capacitors, the resistors and thermistors surrounding the droop voltage sensing and droop amplifier has to be resized up to 10X to reduce the capacitance by 10X. But attention has to be paid in balancing the impedance of droop amplifier in this case. Dynamic Mode of Operation - Compensation Parameters Considering the voltage regulator as a black box with a voltage source controlled by VID and a series impedance, in order to achieve the 2.1mV/A load line, the impedance needs to be 2.1mΩ. The compensation design has to ensure the output impedance of the converter be lower than 2.1mΩ. There is a mathematical calculation file available to the user. The power stage parameters such as L and Cs are needed as the input to calculate the compensation component values. Attention has be paid to the input resistor to the FB pin. Too high of a resistor will cause an error to the output voltage regulation because of bias current flowing in the FB pin. It is better to keep this resistor below 3K when using this file. Static Mode of Operation - Current Balance using DCR or Discrete Resistor Current Sensing Current Balance is achieved in the ISL6260 and ISL6260B through the matching of the voltages present on the ISEN pins. The ISL6260 and ISL6260B adjust the duty cycles of each phase to maintain equal potentials on the ISEN pins. RL and CL around each inductor, or around each discrete current resistor, are used to create a rather large time constant such that the ISEN voltages have minimal ripple voltage and represent the DC current flowing through each channel’s inductor. For optimum performance, RL is chosen to be 10kΩ and CL is selected to be 0.22µF. When discrete resistor sensing is used, a capacitor of 10nF should be placed in parallel with RL to properly compensate the current balance circuit. ISL6260 and ISL6260B uses RC filter to sense the average voltage on phase node and forces the average voltage on the phase node to be equal for current balance. Even though the ISL6260, ISL6260B forces the ISEN voltages to be almost equal, the inductor currents will not be exactly equal. Take DCR current sensing as example, two errors have to be added to find the total current imbalance. 1) Mismatch of DCR: If the DCR has a 5% tolerance then the resistors could mismatch by 10% worst case. If each phase is carrying 20A then the phase currents mismatch by 20A*10% = 2A. 2) Mismatch of phase voltages/offset voltage of ISEN pins. The FN9162.1 January 3, 2006 ISL6260, ISL6260B phase voltages are within 2mV of each other by current balance circuit. The error current that results is given by 2mV/DCR. If DCR = 1mΩ then the error is 2A. In the above example, the two errors add to 4A. For a two phase DC/DC, the currents would be 22A in one phase and 18A in the other phase. In the above analysis, the current balance can be calculated with 2A/20A = 10%. This is the worst case calculation, for example, the actual tolerance of two 10% DCRs is 10%*sqrt(2) = 7%. current sensing resistors, and therefore is populated with a 47pF capacitor. The Rs values in the previous section, Rs = 7.68k_1% are sufficient for this approach. Now, the input to the Droop amplifier is the Vrsense voltage. This voltage is given by the following equation: Rsense Vrsense = ---------------------- × Iout N (EQ. 15) There are provisions to correct the current imbalance due to layout or to purposely divert current to certain phase for better thermal management. Customer can put a resistor in parallel with the current sensing capacitor on the phase of interest in order to purposely increase the current in that phase. But it is highly recommended for symmetrical layout. The gain of the Droop amplifier, G2, must be adjusted for the ratio of the Rsense to Droop impedance, Rdroop. We use the following equation: In the case the PC board trace resistance from the inductor to the microprocessor are not the same on all three phases, the current will not be balanced. On the phases that have too much trace resistance a resistor can be added in parallel with the ISEN capacitor that will correct for the poor layout. But it is highly recommended for symmetrical layout. Assuming N = 3, Rdroop = 0.0021(V/A) as per the Intel IMVP-6 specification, Rsense = 0.001Ω, we obtain G2 = 6.3. An estimate of the value of the resistor is: Rdroop G2 = ---------------------- × N Rsense (EQ. 16) The values of Rdrp1 and Rdrp2 are selected to satisfy two requirements. First, the ratio of Rdrp2 and Rdrp1 determine the gain G2 = (Rdrp2/Rdrp1)+1. Second, the parallel combination of Rdrp1 and Rdrp2 should equal the parallel combination of the Rs resistors. Combining these requirements gives: Rtweak = Risen* [2*Rdcr - (Rtrace - Rmin)]/[2(Rtrace - Rmin)] where Risen is the resistance from the phase node to the ISEN pin; usually 10kΩ. Rdcr is the DCR resistance of the inductor. Rtrace is the trace resistance from the inductor to the microprocessor on the phase that needs to be tweaked. It should be measured with a good microΩ meter. Rmin is the trace resistance from the inductor to the microproccessor on the phase with the least resistance. For example, if the PC board trace on one phase is 0.5mΩ and on another trace is 0.3mΩ; and if the DCR is 1.2mΩ; then the tweaking resistor is Rtweak = 10kΩ * [1.2*2 (0.5-0.3)]/[2*(0.5-0.3)] = 55kΩ. Droop using Discrete Resistor Sensing - Static/ Dynamic Mode of Operation When choosing current sense resistor, not only the tolerance of the resistance is important, but also the TCR. And its combined tolerance at a wide temperature range should be calculated. Figure 44 shows the equivalent circuit of a discrete current sense approach. Figure 36 shows a more detailed schematic of this approach. Droop is solved the same way as the DCR sensing approach with a few slight modifications. First, there is no NTC required for thermal compensation, therefore, the Rn resistor network in the previous section is not required. Secondly, there is no time constant matching required, therefore, the Cn component is not matched to the L/DCR time constant, but this component does indeed provide noise immunity, especially due to the ESL of the 24 Rdrp1 = G2/(G2-1) * Rs/N Rdrp2 = (G2-1) * Rdrp1 In the example above, Rs = 7.68K, N = 3, and G2 = 6.3 so Rdrp 3K and Rdrp2 is 15.8kΩ. These values are extremely sensitive to layout. Once the board has been laid out, some tweaking may be required to adjust the full load Droop. This is fairly easy and can be accomplished by allowing the system to achieve thermal equilibrium at full load, and then adjusting Rdrp2 to obtain the desired Droop value. Fault Protection - Overcurrent Fault Setting As previously described, the overcurrent protection of the ISL6260, ISL6260B is related to the Droop voltage. Previously we have calculated that the Droop Voltage = ILoad * Rdroop, where Rdroop is the load line slope specified as 0.0021 (V/A) in the Intel IMVP-6 specification. Knowing this relationship, the overcurrent protection threshold can be set up as a voltage Droop level. Knowing this voltage droop level, one can program in the appropriate drop across the Roc resistor. This voltage drop will be referred to as Voc. Once the droop voltage is greater than Voc, the PWM drives will turn off and PGOOD will go low. The selection of Roc is given below in Equation 17. Assuming we desire an overcurrent trip level, Ioc, of 55A, and knowing from the Intel Specification that the load line FN9162.1 January 3, 2006 + ISL6260, ISL6260B 10uA Roc + VSUM + DROOP - + VDIFF + 1 - DROOP + 1 - RTN + VSEN VO' Rdrp1 Σ + RS N DFB Rdrp2 Internal to ISL6260 RS EQV = VSUM Cn OC - Voc OCSET Vrsense EQV = Iout × Rsense N VN VO' ROEQV = RO N FIGURE 44. EQUIVALENT MODEL FOR DROOP AND DIE SENSING USING DISCRETE RESISTOR SENSING slope, Rdroop is 0.0021 (V/A), we can then calculate for Roc as shown in Equation 17. Ioc × Rdroop 55 × 0.0021 Roc = ------------------------------------- = ------------------------------- = 11.5KΩ 10μA 10x10 – 6 (EQ. 17) Note, if the droop load line slope is not -0.0021 (V/A) in the application, the overcurrent setpoint will differ from predicted. A capacitor may be added in parallel with Roc to improve noise rejection but the Roc*capacitor time constant cannot exceed 20µs. Do not remove Roc if overcurrent protection is not desired. The maximum Roc is 30K. 25 FN9162.1 January 3, 2006 ISL6260, ISL6260B Quad Flat No-Lead Plastic Package (QFN) Micro Lead Frame Plastic Package (MLFP) 2X 9 MILLIMETERS D/2 D1 D1/2 2X N 6 INDEX AREA 40 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE (COMPLIANT TO JEDEC MO-220VJJD-2 ISSUE C) 0.15 C A D A L40.6x6 0.15 C B SYMBOL MIN NOMINAL MAX NOTES A 0.80 0.90 1.00 - A1 - - 0.05 - A2 - - 1.00 A3 1 2 3 E1/2 E/2 E1 b D2 0.15 C B 0.15 C A 4X B TOP VIEW 0 A C 0.08 C SEATING PLANE A1 A3 SIDE VIEW 9 5 NX b 0.10 M C A B 4X P D2 (DATUM B) 8 7 NX k D2 2 N 4X P - 4.10 9 4.25 6.00 BSC - 5.75 BSC 9 3.95 4.10 4.25 (Ne-1)Xe REF. E2 - k 0.25 - - - L 0.30 0.40 0.50 8 L1 - - 0.15 10 N 40 2 Nd 10 3 Ne 10 3 P - - 0.60 9 θ - - 12 9 NOTES: 1. Dimensioning and tolerancing conform to ASME Y14.5-1994. 7 E2/2 NX L N e 8 2. N is the number of terminals. 8 3. Nd and Ne refer to the number of terminals on each D and E. 4. All dimensions are in millimeters. Angles are in degrees. 9 CORNER OPTION 4X (Nd-1)Xe REF. 7, 8 0.50 BSC Rev. 1 10/02 2 3 6 INDEX AREA 7, 8 E 1 (DATUM A) 5, 8 5.75 BSC 3.95 e / / 0.10 C 0.30 E1 E2 A2 0.23 9 6.00 BSC D1 9 2X 2X 0.18 D E 9 0.20 REF BOTTOM VIEW 5. Dimension b applies to the metallized terminal and is measured between 0.15mm and 0.30mm from the terminal tip. 6. The configuration of the pin #1 identifier is optional, but must be located within the zone indicated. The pin #1 identifier may be either a mold or mark feature. A1 NX b 5 7. Dimensions D2 and E2 are for the exposed pads which provide improved electrical and thermal performance. SECTION "C-C" C L L1 10 8. Nominal dimensions are provided to assist with PCB Land Pattern Design efforts, see Intersil Technical Brief TB389. C L L L1 e 10 L e C C TERMINAL TIP FOR ODD TERMINAL/SIDE 9. Features and dimensions A2, A3, D1, E1, P & θ are present when Anvil singulation method is used and not present for saw singulation. 10. Depending on the method of lead termination at the edge of the package, a maximum 0.15mm pull back (L1) maybe present. L minus L1 to be equal to or greater than 0.3mm. FOR EVEN TERMINAL/SIDE All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems. Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries. For information regarding Intersil Corporation and its products, see www.intersil.com 26 FN9162.1 January 3, 2006