MICRO-LINEAR ML4866CS

July 2000
ML4866*
3.3V Output DC–DC Step-Down Converter
GENERAL DESCRIPTION
FEATURES
The ML4866 is a high efficiency pulse width modulated
(PWM) buck regulator designed for use in 5V systems or
portable equipment that need a compact, efficienct 3.3V
supply. It has a switching frequency of 120kHz and uses
synchronous rectification to optimize power conversion
efficiency. Unlike other solutions, the ML4866 requires no
external diodes or FETs.
■
High power conversion efficiency over 2 decades of
load current
■
No external FETs or diodes; minimum external
components
■
3.5V to 6.5V input voltage range
The ML4866 can provide up to 500mA of output current,
and operates over an input voltage range of 3.5V to 6.5V
(3 to 4 cells or a 5 VDC supply). A complete switched
mode power converter can be quickly and easily
implemented with few external components. Thanks to a
built-in autoburst mode, power conversion efficiency of
this DC–DC converter can exceed 90% over more than 2
decades of output load current.
■
Significantly extends battery life over linear regulator
based solutions
■
Micropower operation
■
Low shutdown mode quiescent current
Stability and fast loop response are provided by current
programming and a current sense circuit. The ML4866
also has a SHDN pin for use in systems which have power
management control. Undervoltage lockout and soft start
are also built in.
(* Indicates Part is End Of Life as of July 1, 2000)
BLOCK DIAGRAM
5
7
VIN
1
VOUT
VL
CURRENT
SENSE
BUCK
CONTROL
UVLO/
SHUTDOWN
OSC
ERROR
AMPLIFIER
SLOPE
COMPENSATION
REFERENCE
–
–
BURST
SHDN
6
3
VREF
+
+
BURST
4
VREF
COMP
2
GND
8
1
ML4866
PIN CONFIGURATION
ML4866
8-Pin SOIC (S08)
VOUT
1
8
GND
COMP
2
7
VL
VREF
3
6
SHDN
BURST
4
5
VIN
TOP VIEW
PIN DESCRIPTION
PIN
NAME
FUNCTION
1
VOUT
Regulated 3.3V output
2
COMP
3
4
2
PIN
NAME
FUNCTION
5
V IN
Input voltage
Connection point for an external
compensation network
6
SHDN
Pulling this pin low shuts down the
regulator
V REF
1.25V reference output
7
VL
Buck inductor connection
BURST
This pin controls when the control
circuit switches between PWM and
PFM modes of operation
8
GND
Ground
ML4866
ABSOLUTE MAXIMUM RATINGS
OPERATING CONDITIONS
Absolute maximum ratings are those values beyond which
the device could be permanently damaged. Absolute
maximum ratings are stress ratings only and functional
device operation is not implied.
Temperature Range
ML4866CS ................................................. 0ºC to 70ºC
ML4866ES .............................................. -20ºC to 70ºC
ML4866IS ............................................... -40ºC to 85ºC
VIN Operating Range ................................... 3.5V to 6.5V
VIN ................................................................................................... 7V
Voltage on any other pin ......... GND - 0.3V to VIN + 0.3V
Peak Switch Current (IPEAK) ......................................... 2A
Average Switch Current (IAVG) ..................................... 1A
Junction Temperature .............................................. 150ºC
Storage Temperature Range ....................... -65ºC to 150ºC
Lead Temperature (Soldering 10 Sec.) ..................... 260ºC
Thermal Resistance (qJA) .................................... 160ºC/W
ELECTRICAL CHARACTERISTICS
Unless otherwise specified, VIN = 5V, L = 50µH, COUT = 100µF, RCOMP = 390kW, CCOMP = 15nF,
TA = Operating Temperature Range (Note 1)
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
0 < I(VREF) < –5µA, IOUT = 0mA
1.22
1.25
1.27
V
Oscillator Initial Accuracy
IOUT = 200mA, TA = 25°C
100
115
165
kHz
Oscillator Total Variation
Line and Temp
90
130
185
kHz
3
5
ms
REFERENCE
VREF
Output Voltage
PWM REGULATOR
fOSC
Soft Start VIN to VOUT Delay
BURST Burst Mode Threshold
250
BURST PWM Mode Threshold
400
500
BURST Bias Current
Output Voltage
mV
850
mV
35
µA
IOUT = 200mA
3.2
3.3
3.4
V
IOUT = 20mA, BURST = 0V
3.28
3.38
3.48
V
±2
%
Line Regulation
VIN = 4V to 6.5V, TA = 25°C
Load Regulation
IOUT = 100mA to 500mA,
TA = 25°C
±2.5
%
IOUT = 5mA to 100mA,
BURST = 0V, TA = 25°C
±2.5
%
Temperature Stability
TA = -40°C to 85°C
±1
%
Total Variation
Line, Load, Temp
±5
%
SHUTDOWN
UVLO Startup Threshold
3.2
3.5
V
UVLO Shutdown Threshold
2.9
3.1
V
SHDN Threshold
SHDN Bias Current
2
V
–5
µA
3
ML4866
ELECTRICAL CHARACTERISTICS
SYMBOL
(Continued)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
IOUT = 0mA, BURST = 5V
400
500
µA
IOUT = 0mA, BURST = 0V
120
220
µA
SHDN = 0V
20
35
µA
SUPPLY
IIN
VIN Current
Note 1: Limits are guaranteed by 100% testing, sampling, or correlation with worst case test conditions.
FUNCTIONAL DESCRIPTION
The ML4866 is a current-mode, step-down (buck)
converter designed to keep the buck inductor current in
the continuous conduction mode (CCM). Current-mode
operation provides faster output response to input voltage
and output current changes along with cycle-by-cycle
current limiting. CCM inductor current is preferred when
the highest conversion efficiencies are required.
ramping the inductor current down to 0mA. This action is
repeated until the output voltage returns to its nominal
setting and begins again when the output drops below its
nominal setting. The rate or frequency at which this
“bursting” occurs is directly proportional to the output
current. When the average output current rises above
130mA, the ML4866 returns to PWM operation.
For high efficiencies at low output current, the ML4866
contains an autoburst function which automatically
switches from pulse width modulation (PWM) to pulsed
frequency modulation (PFM) operation when the output
current drops below 100mA. Selection of either mode is
possible by applying the correct logic level signal to the
BURST pin. When operating in PWM mode, loop
compensation of the ML4866 is simplified due to its
transconductance type error amplifier.
For applications having a load current range of less than
100mA and greater than 130mA, the BURST pin should be
left open and bypassed to ground with a 15nF or larger
capacitor. It is possible to tailor an application for the
highest possible efficiency by externally forcing the
ML4866 into either control mode. Applying a logic low
level to BURST forces the IC into PFM mode. Conversely,
a logic high places it in PWM mode. Care should be
taken to avoid reducing the efficiency by placing the
controller in the least efficient mode for a given output
current.
An under voltage lockout (UVLO) circuit within the
ML4866 enables the converter when the input voltage
is greater than 3.25V and disables it when the input
voltage is below 3.10V. The IC can also be disabled
externally by applying a logic low signal to the SHDN
pin. When disabled, the ML4866 draws less than 20µA of
current.
The internal 1.25V bandgap reference is made available
via the VREF pin, and may be used for general
applications requiring less than 10µA of current. For
proper operation, this pin must always be bypassed to
GND with a 100nF capacitor.
BURST MODE
Burst (PFM) mode is a method of regulating the output
voltage by applying a variable frequency modulation
technique to the buck inductor. This method maintains
higher efficiencies at light loads than if PWM were used.
If BURST is left open, the ML4866 switches from PWM
mode to PFM mode when the output current falls below
100mA. When the output voltage falls out of regulation
while in PFM mode, the internal buck switch turns on and
ramps the inductor current up to 300mA. The buck switch
then turns off and the synchronous switch turns on,
4
VIN – VOUT
L
–VOUT
L
∆I
Figure 1. Inductor Current
ML4866
DESIGN CONSIDERATIONS
100
98
Figure 1 shows the inductor current in a step-down
converter operating in CCM. Note that the inductor
current does not reach zero during each switching cycle.
This is unlike discontinuous conduction mode (DCM)
where the inductor current is allowed to reach zero. CCM
operation generally results in lower peak to peak output
ripple voltage and higher circuit efficiencies because of
lower peak and RMS currents in the switching FETs and
buck inductor. The minimum value of inductance required
for CCM operation with a 6.5V input and a load range of
100mA to 500mA is:
96
L>
L>
VOUT ™ ( VIN ( MAX) - VOUT )
2 ™ VIN ( MAX) ™ IOUT
IL(P –P) =
( MAX)
™ ( VIN ( MIN) - VOUT
88
86
( MAX) )
(2)
2 ™ 3.465V ™ ( 4.0V - 3.465V)
= 103mA
4.0V ™ 90kHz ™ 100mH
( MAX )
+
VOUT
IL(PEAK) = IOUT ( MAX) +
( MAX )
™ ( VIN ( MIN) - VOUT ( MAX) )
VIN ( MIN) ™ fSW ™ L
(3)
For the highest efficiency, inductor core and copper losses
must be minimized. Good high frequency core material
such as Kool-Mu, ferrite or Molyperm are popular choices
for this converter. Disregarding physical size
requirements, the lowest loss inductor will generally be
the one with the highest peak current rating.
Figure 2 displays the efficiency of the ML4866 under
various input voltage and output current conditions. These
results were obtained using a Coiltronics CTX100-4
inductor having the following specifications:
Peak Current Rating - 950mA
DC Resistance - 175mW
3.5
4.0
4.5
5.0
5.5
6.0
6.5
A partial listing of inductor manufacturers with standard
parts which meet the criteria for use with the ML4866 is
given below.
Coiltronics
Dale
Coilcraft
XFMRS, Inc
Sumida
(561) 241-7876
(605) 665-9301
(847) 639-6400
(317) 834-1066
(847) 956-0666
CAPACITOR SELECTION
3.465V ™ ( 4.0V - 3.465V)
= 550mA
4.0V ™ 120kHz ™ 100mH
Nominal Inductance - 100µH
IOUT = 500mA
Figure 2. Efficiency vs. Input Voltage
VIN ( MIN) ™ fSW ( MIN) ™ L
IL(PEAK) = IOUT
IOUT = 10mA
92
(1)
To guarantee reliable operation, the peak inductor current
must be between 80% and 85% of its maximum rated
value. This value is the sum of the inductor peak to peak
current and the maximum output current:
2 ™ VOUT
94
90
3.3V ™ (6.5V - 33
. V)
> 68mH
2 ™ 6.5V ™ 100mA ™ 120kHz
IL(P –P) =
IOUT = 100mA
INPUT VOLTAGE (V)
™ fSW
( MIN)
EFFICIENCY (%)
INDUCTOR SELECTION
A typical digital system requires a peak to peak output
ripple voltage of no greater than 1% to 3% of the nominal
output voltage. In a step-down converter, the largest
contributor to ripple voltage is almost always the product
of the inductor peak-to-peak current times the output
capacitor’s equivalent series resistance. To select the
correct capacitor, first calculate the minimum
capacitance value required:
C OUT >
C OUT >
VOUT ™ ( VIN ( MAX) - VOUT )
(4)
8 ™ VP -P ( MAX) ™ VIN ( MAX) ™ L ™ fSW 2
. V)
3.3 ™ (6.5V - 33
8 ™ 33mV ™ 6.5V ™ 100mH ™ 120kHz 2
> 4.27mF
Next, calculate the maximum permissible ESR of the
output capacitor:
ESR <
(0.033)
< 0.33W
(0.1)
(5)
When limited space is available, tantalum capacitors are
the best choice. Electrolytic capacitors can be used and
will be less expensive, but the ESR for low capacitance
values as needed here will be much higher than for the
same value tantalum. Table 2 lists the ESR values for a
number of general purpose tantalum capacitors which are
widely available from a number of sources. A 47µF
capacitor was chosen for the design example.
5
ML4866
DESIGN CONSIDERATIONS
(Continued)
FREQUENCY COMPENSATION
VARYING LOAD CURRENT
Frequency compensation of the ML4866 is required when
the converter is operating in PWM mode. Two simple
methods are provided to ensure the converter is frequency
stable. Both these methods will work only if the inductor
current is selected to be in CCM at the maximum load
current (see Inductor Selection). The first, called dominant
pole compensation, is used when non-varying loads are
expected. This method requires a single capacitor
connected from the error amplifier output (COMP Pin) to
ground.
To minimize output voltage variations due to rapidly
changing load currents, use the series RC zero
compensation method to find the compensation network
component values that will improve the output voltage
response to load transients.
For loads which change suddenly, the transient response
(or bandwidth) of the circuit must be increased to prevent
the output voltage from going outside of the regulation
band. The method used to accomplish this is called
zero/pole compensation and requires a series resistor
capacitor network from COMP to ground.
To determine which method works best for a given
application, apply the components found from the
zero/pole compensation method to an actual circuit
and examine the output voltage variation. If the voltage
variation is acceptable, connect the simpler, single
capacitor and re-check the output voltage for acceptable
load transient response.
The unity gain bandwidth of the converter is extended to
15kHz using an RC network determined by:
R COMP >
f
G
, where G = O
gm
fCOMP
(7)
C COMP =
1
50p ™ R COMP
(8)
Where f0 = 15kHz, fCOMP = 640Hz, RCOMP > 375kW (use
390kW, 5%), and CCOMP = 16nF (use 15nF).
Either method of compensation for CCM mode with result
in continued stability as the ML4866 changes to DCM
mode at lighter load currents. Figure 3 shows a typical
application circuit for the ML4866.
NON-VARYING LOAD CURRENT
For the best possible response to load transients using only
a single capacitor, dominant pole compensation is
implemented with a single capacitor value of:
C COMP =
gm
2 ™ fCOMP
(6)
Where fCOMP is the unity gain crossover point (640Hz),
gm = 62.5µmho, and CCOMP > 15.5nF (choose a standard
18nF or 22nF capacitor). The value of CCOMP can be
increased but at the risk of increased output voltage
variations with transient loads.
VOUT
3.3V
33µF
CAPACITANCE
VOLTAGE
RATING
SIZE
ESR @
100kHz
4.7µF
16V
3216
0.490W
10µF
6.3V
3216
0.368W
22µF
16V
7343
0.149W
33µF
6.3V
6032
0.291W
47µF
10V
7343
0.144W
100µF
6.3V
7343
0.088W
Table 2. ESR Values for Low Cost Tantalum Capacitors
6
100µH
ML4866
VOUT
390kΩ
COMP
VREF
BURST
15nF
100nF
15nF
1
8
2
7
3
6
4
5
GND
VL
SHDN
VIN
100µF
VIN
3.5V to 6.5V
100nF
Figure 3. Typical Application Circuit
ML4866
LAYOUT
For proper performance, all components should be placed
as close to the ML4866 as possible. Particular attention
should be paid to minimize the length of the connections
between the COMP and VREF pins to GND. Also avoid
bringing these traces and the associated components
close to VL.
It is always recommended that a 10µF or greater
capacitor be connected to VIN of the ML4866. A 33µF
tantalum capacitor and 100nF film or ceramic capacitor
is recommended when powering the ML4866 from
Lithium or Alkaline cells.
Ground and power planes must be large enough to carry
the current the converter has been designed to supply.
A sample PC board layout is shown in Figure 4.
Figure 4. Sample PC Board Layout
7
ML4866
PHYSICAL DIMENSIONS
inches (millimeters)
Package: S08
8-Pin SOIC
0.189 - 0.199
(4.80 - 5.06)
8
PIN 1 ID
0.148 - 0.158 0.228 - 0.244
(3.76 - 4.01) (5.79 - 6.20)
1
0.050 BSC
(1.27 BSC)
0.017 - 0.027
(0.43 - 0.69)
(4 PLACES)
0.059 - 0.069
(1.49 - 1.75)
0º - 8º
0.055 - 0.061
(1.40 - 1.55)
0.012 - 0.020
(0.30 - 0.51)
0.004 - 0.010
(0.10 - 0.26)
0.015 - 0.035
(0.38 - 0.89)
0.006 - 0.010
(0.15 - 0.26)
SEATING PLANE
ORDERING INFORMATION
PART NUMBER
TEMPERATURE RANGE
PACKAGE
ML4866CS (End Of Life)
ML4866ES (EOL)
ML4866IS (Obsolete)
0ºC to 70ºC
-20ºC to 70ºC
-40ºC to 85ºC
8-Pin SOIC (S08)
8-Pin SOIC (S08)
8-Pin SOIC (S08)
DS4866-01
© Micro Linear 1997.
is a registered trademark of Micro Linear Corporation. All other trademarks are the property of their respective owners.
Products described herein may be covered by one or more of the following U.S. patents: 4,897,611; 4,964,026; 5,027,116; 5,281,862; 5,283,483; 5,418,502; 5,508,570;
5,510,727; 5,523,940; 5,546,017; 5,559,470; 5,565,761; 5,592,128; 5,594,376; 5,652,479; 5,661,427; 5,663,874. Japan: 2,598,946; 2,619,299. Other patents are pending.
Micro Linear reserves the right to make changes to any product herein to improve reliability, function or design. Micro Linear does not assume any liability arising out of
the application or use of any product described herein, neither does it convey any license under its patent right nor the rights of others. The circuits contained in this data
sheet are offered as possible applications only. Micro Linear makes no warranties or representations as to whether the illustrated circuits infringe any intellectual property
rights of others, and will accept no responsibility or liability for use of any application herein. The customer is urged to consult with appropriate legal counsel before
deciding on a particular application.
8
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9/8/97 Printed in U.S.A.