TI TPS62125DSGT

TPS62125
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SLVSAQ5A – MARCH 2012 – REVISED APRIL 2012
3V-17V, 300mA Step Down Converter With Adjustable Enable Threshold And Hysteresis
Check for Samples: TPS62125
FEATURES
DESCRIPTION
•
•
The TPS62125 is a high efficiency synchronous step
down converter optimized for low and ultra low power
applications providing up to 300mA output current.
The wide input voltage range of 3V to 17V supports
four cell alkaline and 1 to 4 cell Li-Ion batteries in
series configuration as well as 9V to 15V powered
applications. The device includes a precise low power
enable comparator which can be used as an input
supply voltage supervisor (SVS) to address system
specific power up and down requirements. The
enable comparator consumes only 6µA quiescent
current and features an accurate threshold of 1.2V
typical as well as an adjustable hysteresis. With this
feature, the converter can generate a power supply
rail by extracting energy from a storage capacitor fed
from high impedance sources such as solar panels or
current loops. With its DCS - Control TM scheme the
converter provides power save mode operation to
maintain highest efficiency over the entire load
current range. At light loads the converter operates in
PFM mode (pulse frequency modulation) and
transitions seamlessly and automatically in PWM
(pulse width modulation) mode at higher load
currents. The DCS - ControlTM scheme is optimized
for low output ripple voltage in PFM mode in order to
reduce output noise to a minimum and features
excellent AC load regulation. An open drain power
good output indicates once the output voltage is in
regulation.
1
•
•
•
•
•
•
•
•
•
•
•
•
•
Wide Input Voltage Range 3V to 17V
Input SVS (Supply Voltage Supervisor) with
Adjustable Threshold / Hysteresis Consuming
typ. 6µA Quiescent Current
Wide Output Voltage Range 1.2V to 10V
Typ. 13 µA Quiescent Current
350nA typ. Shutdown Current
Seamless Power Save Mode Transition
DCS-ControlTM Scheme
Low Output Ripple Voltage
Up to 1MHz Switching Frequency
Highest Efficiency over Wide VIN and VOUT
Range
Pin to Pin Compatible with TPS62160/70
100% Duty Cycle Mode
Power Good Open Drain Output
Output Discharge Function
Small 2x2mm2 SON 8 pin Package
APPLICATIONS
•
•
•
•
•
Embedded processing
4 cell alkaline, 1-4 cell Li-Ion battery powered
applications
9V - 15V standby power supply
Energy harvesting
Inverter (negative VOUT)
95
10mA
100mA
90
TPS62125
VIN
SW
R1
1.8M
VOS
CIN
10µF
EN
VOUT = 3.3V
up to 300mA
L 15µH
FB
R2
576k
EN_hys
250mA
1.0mA
80
COUT
10µF
75
70
0.25mA
65
Rpullup
GND
85
Efficiency (%)
VIN = 4V to 17V
PG
VOUT = 3.3V
L = 15mH VLF302515
COUT = 10mF
60
0.1mA
55
PWR GOOD
50
4
5
6
7
8
9
10 11
12 13 14 15 16 17
Input Voltage VIN(V)
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2012, Texas Instruments Incorporated
TPS62125
SLVSAQ5A – MARCH 2012 – REVISED APRIL 2012
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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
ORDERING INFORMATION
TA
–40°C to 85°C
(1)
(2)
PART NUMBER (1)
VOUT
PACKAGE MARKING
TPS62125
adjustable
SAQ
TPS62126 (2)
1.8V
TBD
For the most current package and ordering information see the Package Option Addendum at the end of this document, or see the TI
Web site at www.ti.com
samples available, contact TI for further information
ABSOLUTE MAXIMUM RATINGS (1)
over operating free-air temperature range (unless otherwise noted)
VALUE
Pin voltage range
(2)
UNIT
MIN
MAX
VIN
- 0.3
20
V
SW
- 0.3
VIN +0.3V
V
EN
- 0.3
VIN +0.3V
V
FB
– 0.3
3.6
V
VOS, PG
- 0.3
12
V
EN_hys
– 0.3
7
V
mA
Power Good sink
current
IPG
10
EN_hys sink current
IEN_hys
3
HBM Human body model
2
CDM Charge device model
1
ESD rating (3)
Machine model
kV
100
V
Maximum operating junction temperature, TJ
–40
125
°C
Storage temperature range, Tstg
–65
150
°C
(1)
(2)
(3)
Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only and functional operation of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute–maximum–rated conditions for extended periods may affect device reliability.
All voltage values are with respect to network ground terminal GND.
The human body model is a 100-pF capacitor discharged through a 1.5-kΩ resistor into each pin. The machine model is a 200-pF
capacitor discharged directly into each pin.
THERMAL INFORMATION
TPS62125
THERMAL METRIC (1)
DSG
UNITS
8 PINS
θJA
Junction-to-ambient thermal resistance
65.2
θJC(top)
Junction-to-case(top) thermal resistance
93.3
θJB
Junction-to-board thermal resistance
30.1
ψJT
Junction-to-top characterization parameter
0.5
ψJB
Junction-to-board characterization parameter
47.4
θJC(bottom)
Junction-to-case(bottom) thermal resistance
7.2
(1)
2
°C/W
For more information about traditional and new thermal metrics, see the IC PackageThermal Metrics application report, SPRA953
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RECOMMENDED OPERATING CONDITIONS
MIN
Supply voltage VIN
Output current capability
MAX
17
3V ≤ VIN < 6V
200
6V ≤ VIN ≤ 17V
300
Operating ambient temperature TA
(1)
, (Unless Otherwise Noted)
Operating junction temperature range, TJ
(1)
NOM
3
UNIT
V
mA
–40
85
°C
–40
125
°C
In applications where high power dissipation and/or poor package thermal resistance is present, the maximum ambient temperature may
have to be derated. Maximum ambient temperature (TA(max)) is dependent on the maximum operating junction temperature (TJ(max)) and
the maximum power dissipation of the device in the application (PD(max)). see the IC Package Thermal Metrics application report,
SPRA953.
ELECTRICAL CHARACTERISTICS
TA = –40°C to 85°C, typical values are at TA = 25°C (unless otherwise noted), VIN = 12V
PARAMETER
TEST CONDITIONS
MIN
TYP MAX
UNIT
3.0
17
V
1.2
10
V
23
µA
SUPPLY
VIN
Input voltage range (1)
VOUT
Output voltage range
IQ
Quiescent current
IOUT = 0mA, Device not switching, EN = VIN,
regulator sleeps
13
IOUT = 0mA, Device switching, VIN = 7.2V,
VOUT = 1.2V, L = 22µH
14
µA
VIN = 5V, EN = 1.1V, enable comparator active,
device DC/DC converter off
6
11
IActive
Active mode current consumption
VIN = 5 V = VOUT, TA = 25°C, high-side MOSFET
switch fully turned on (100% Mode)
230
275
µA
ISD
Shutdown current (2)
Enable comparator off, EN < 0.4V,
VOUT = SW = 0 V, VIN = 5V
0.35
2.4
µA
VUVLO
Undervoltage lockout threshold
Falling VIN
2.8
2.85
V
Rising VIN
2.9
2.95
V
1.16
1.20
1.24
V
1.12
1.15
1.19
V
ENABLE COMPARATOR THRESHOLD AND HYSTERESIS (EN, EN_hys)
VTH EN ON
EN pin threshold rising edge
VTH EN OFF
EN pin threshold falling edge
VTH EN Hys
EN pin hysteresis
IIN
Input bias current into EN pin
EN = 1.3V
VEN_hyst
EN_hys pin output low
IEN_hyst = 1mA, EN = 1.1V
IIN
Input bias current into EN_hyst pin
EN_hyst = 1.3V
EN
EN_hyst
3.0 V ≤ VIN ≤ 17V
50
0
mV
50
nA
0.4
V
0
50
nA
2.4
4
VIN = 12V, I = 100mA
1.5
2.6
VIN = 3V, I = 100mA
0.75
1.3
VIN = 12V, I = 100mA
0.6
1
750
900
POWER SWITCH
high-side MOSFET on-resistance
RDS(ON)
low-side MOSFET on-resistance
VIN = 3 V, I = 100mA
Ω
Switch current limit high-side
MOSFET
VIN = 12V
Thermal shutdown
Increasing junction temperature
150
°C
Thermal shutdown hysteresis
Decreasing junction temperature
20
°C
tONmin
Minimum ON time
VIN = 5V, VOUT = 2.5 V
500
ns
tOFFmin
Minimum OFF time
VIN = 5 V
60
ns
ILIMF
TSD
600
mA
OUTPUT
(1)
(2)
The part is functional down to the falling UVLO (Under Voltage Lockout) threshold
Current into VIN pin
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ELECTRICAL CHARACTERISTICS (continued)
TA = –40°C to 85°C, typical values are at TA = 25°C (unless otherwise noted), VIN = 12V
PARAMETER
VREF_FB
TEST CONDITIONS
MIN
Internal reference voltage of error
amplifier
TYP MAX
0.808
Referred to internal reference (VREF_FB)
Feedback voltage line regulation
IOUT = 100mA, 5V ≤ VIN ≤ 17V, VOUT = 3.3V
Feedback voltage load regulation
VOUT = 3.3V; IOUT = 1mA to 300mA, VIN = 12V
Input bias current into FB pin
VFB = 0.8 V
tStart
Regulator start-up time
Time from EN high to device starts switching,
VIN = 5V
tRamp
Output voltage ramp time
Time to ramp up VOUT = 1.8V, no load
200
ILK_SW
Leakage current into SW pin (4)
1.8
2.85
µA
IIN_VOS
Bias current into VOS pin
VOS = VIN = VSW = 1.8 V, EN = GND, device in
shutdown mode.
0
50
nA
IIN_FB
(3)
0
V
Feedback voltage accuracy
VFB
–2.5
UNIT
2.5
-0.05
(3)
-0.004
0
%
%/V
%/mA
50
50
nA
µs
POWER GOOD OUTPUT (PG)
Rising VFB feedback voltage
93
95
97
Falling VFB feedback voltage
87
90
93
VTH_PG
Power Good threshold voltage
VOL
PG pin Output low voltage
Current into PG pin IPG= 0.4mA
0.3
V
VOH
PG pin Output high voltage
Open drain output, external pull up resistor
10
V
IIN_PG
Bias current into PG pin
V(PG) = 3V, EN = 1.3V, FB = 0.85 V
50
nA
(3)
(4)
4
0
%
VOUT = 3.3V, L = 15µH, COUT = 10µF
An internal resistor divider network with typ. 1MΩ total resistance is connected between SW pin and GND.
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DEVICE INFORMATION
PIN ASSIGNMENTS
DSG PACKAGE
(TOP VIEW)
8
1
2
3
4
TH E
ER XP
M OS
A ED
L
PA
D
GND
VIN
EN
EN_hys
7
6
5
PG
SW
VOS
FB
PIN FUNCTIONS
PIN
NAME
NO.
I/O
DESCRIPTION
GND
1
PWR
GND supply pin.
VIN
2
PWR
VIN power supply pin.
EN
3
IN
Input pin for the enable comparator. Pulling this pin to GND turns the device into shutdown mode.
The DC/DC converter is enabled once the rising voltage on this pin trips the enable comparator
threshold, VTH EN ON of typ. 1.2V. The DC/DC converter is turned off once a falling voltage on this
pin trips the threshold, VTH EN OFF of typ. 1.15V. The comparator threshold can be increased by
connecting an external resistor to pin EN_hys. See also application section. This pin must be
terminated.
EN_hys
4
OUT
Enable Hysteresis Open-Drain Output. This pin is pulled to GND when the voltage on the EN pin is
below the comparator threshold VTH EN ON of typ. 1.2V and the comparator has not yet tripped. The
pin is high impedance once the enable comparator has tripped and the voltage at the pin EN is
above the threshold VTH EN ON. The pin is pulled to GND once the falling voltage on the EN pin
trips the threshold VTH EN OFF (1.15V typ.). This pin can be used to increase the hysteresis of the
enable comparator. If not used, tie this pin to GND, or leave it open.
FB
5
IN
This is the feedback pin for the regulator. An external resistor divider network connected to this pin
sets the output voltage. In case of fixed output voltage option, the resistor divider is integrated and
the pin need to be connected directly to the output voltage.
VOS
6
IN
This is the output voltage sense pin for the DCS - ControlTM circuitry. This pin must be connected
to the output voltage of the DC/DC converter.
SW
7
OUT
This is the switch pin and is connected to the internal MOSFET switches. Connect the inductor to
this pin. Do not tie this pin to VIN, VOUT or GND.
PG
8
OUT
Open drain power good output. This pin is internally pulled to GND when the device is disabled or
the output voltage is below the PG threshold. The pin is floating when the output voltage is in
regulation and above the PG threshold. For Power Good indication, the pin van be connected via
a pull up resistor to a voltage rail up to 10V. The pin can sink a current up to 0.4mA and maintain
the specified high / low voltage levels. It can be used to discharge the output capacitor with up to
10mA. In this case the current into the pin must be limited with an appropriate pull up resistor.
More details can be found in the application section. If not used, leave the pin open, or connect to
GND.
Exposed
Thermal PAD
Exposed Thermal Pad. This pad must be connected to GND.
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FUNCTIONAL BLOCK DIAGRAM
ON/
SD
VREF
EN
1.2V
VREF_FB
0.808V
Softstart
Softstart
PG Comparator
EN
VREF
PG
VTH_PG
UVLO
Comparator
FB
VIN
EN
Comparator
VUVLO
EN_hys
DC-DC
ON/OFF
EN_comp
Peak Current
Limit Comparator
GND
Limit
High Side
Timer
DCS
Control
VOS
VIN
Control
Logic
Min. On
VOS
VIN
PMOS
Min. OFF
Direct Control
& Compensation
Gate Driver
Anti
Shoot-Through
SW
VREF_FB
FB
Error
amplifier
NMOS
Comparator
GND
fixed
VOUT
Thermal
Shutdown
Zero Current
Comparator
PARAMETER MEASUREMENT INFORMATION
TPS62125
VIN
VIN
L
VOUT
SW
R1
CIN
10µF
EN
COUT
10µF (VOUT=
< 3.3V)
2x 10µF (3.3V< VOUT<
= 6.7V)
10µF+ 22µF ( VOUT> 6.7V)
FB
R2
EN_hys
VOS
Rpullup
GND
PG
PWR GOOD
L: LPS3314 10mH, 15mH, LPS4018 22mH, VLF302515 15mH
CIN / COUT: 10mF 0805/25V GRM21BR61E106
22mF GRM31CR61 16V X5R, 10mF GRM21B 16V X5R
6
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TYPICAL CHARACTERISTICS
Table 1. Table Of Graphs
PARAMETER
Efficiency
Output Voltage VOUT
peak to peak Output Ripple
Voltage VOUTpp
Switching frequency fSW
Shutdown current
Quiescent Current
EN Comparator Thresholds
RDSON
Typical Operation
Line transient response
Hotplug
Figure 1, Figure 3, Figure 5, Figure 7,
Figure 9, Figure 11
vs. Input Voltage, VOUT = 1.8V, vs. Input voltage, VOUT = 3.3V, VOUT = 5V, VOUT = 6.8V,
VOUT = 8V, VOUT = 10V
Figure 2, Figure 4, Figure 6, Figure 8,
Figure 10, Figure 12
vs. Output Current, VOUT = 3.3V
Figure 13
vs. Input Voltage, VOUT = 3.3V
Figure 14
vs. Output current, VOUT = 5.0V
Figure 15
vs. Input Voltage, VOUT = 5V
Figure 16
vs. Output current, VOUT = 6.7V
Figure 17
vs. Input Voltage, VOUT = 6.7V
Figure 18
vs. Output current, VOUT = 8V
Figure 19
vs. Input Voltage, VOUT = 8V
Figure 20
vs. Output Current, VOUT = 3.3V
Figure 21
vs. Output Current, VOUT = 3.3V
Figure 22
vs. Output Current, VOUT 5.0V
Figure 23
vs. Output Current, VOUT = 8V
Figure 24
vs. Input Voltage
Figure 25
vs. Input Voltage
Figure 26
vs. EN Voltage, rising VEN
Figure 27
vs. EN Voltage, falling VEN
Figure 28
vs. Input Voltage
Figure 29
High Side Switch
Figure 30
Low Side Switch
Figure 31
Power Save Mode VOUT=3.3V, IOUT = 1mA
Figure 32
PWM Mode VOUT= 3.3V, IOUT = 100mA
Figure 33
Load Transient 5mA to 200mA, VOUT = 3.3V
Figure 34
AC Load Regulation 5mA to 200mA 10kHz, VOUT = 3.3V
Figure 35
Load Transient 1mA to 50mA, VOUT = 5V
Figure 36
Load Transient 10mA to 200mA, VOUT = 5V
Figure 37
AC Load Regulation VOUT = 5V
Figure 38
VIN = 9V to 12V, IOUT = 100mA
Figure 39
VIN overshoot exceeding Abs Max ratings
Figure 40
VIN overshoot reduction with additional tantalum polymer capacitor
Figure 41
Short circuit and overcurrent
protection
Input SVS Operation
FIGURE
vs. Output Current VOUT = 1.8V, VOUT = 3.3V, VOUT = 5V, VOUT = 6.8V, VOUT = 8V ,
VOUT = 10V
Figure 42
VOUT = 5.0V
Figure 43
No Input SVS Operation
VOUT = 5.0V, VIN tracks VOUT
Figure 44
Operation from a 0.5mA
current source
VOUT 3.3V, 20mA pulse load
Figure 45
1.8V VOUT
Figure 46
3.3V VOUT
Figure 47
5V VOUT
Figure 48
8V VOUT
Figure 49
Startup
EN On/Off
Output Discharge
Figure 50
using PG pin, triggered by EN Comparator
Figure 51
VOUT ramp down with falling VIN
Figure 52
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95
95
90
90
85
85
80
80
Efficiency (%)
Efficiency (%)
10mA
VIN = 3V
VIN = 5V
VIN = 7.5V
VIN = 9V
VIN = 12V
VIN = 15V
75
70
65
60
50
0.01
0.1
1
10
Output Current IOUT (mA)
250mA
1.0mA
75
VOUT = 1.8V
L = 15mH LPS3314
COUT = 10mF
70
0.25mA
65
60
VOUT = 1.8V
L = 15mH LPS3314
COUT = 10mF
55
100mA
100
0.1mA
55
50
1000
Figure 1. Efficiency vs. Output Current VOUT = 1.8V
3
4
5
6
7
8 9 10 11 12 13 14 15 16 17
Input Voltage VIN (V)
Figure 2. Efficiency vs. Input Voltage, VOUT = 1.8V
100
95
10mA
95
90
85
250mA
1.0mA
80
Efficiency (%)
Efficiency (%)
85
80
VIN = 4.0V
VIN = 5V
VIN = 7.5V
VIN = 9V
VIN = 12V
VIN = 15V
75
70
65
55
50
0.01
0.1
1
10
Output Current (mA)
100
75
70
0.25mA
65
VOUT = 3.3V
L = 15mH VLF302515
COUT = 10mF
60
1000
VOUT = 3.3V
L = 15mH VLF302515
COUT = 10mF
60
0.1mA
55
50
4
5
6
7
8
9
10 11
12 13 14 15 16 17
Input Voltage VIN(V)
Figure 3. Efficiency vs. Output current, VOUT = 3.3V
8
100mA
90
Figure 4. Efficiency vs. Input voltage, VOUT = 3.3V
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100
100
95
95
90
90
10mA
250mA
85
85
VIN = 6.0V
VIN = 7.5V
VIN = 9.0V
VIN = 12V
VIN = 15V
80
75
Efficiency (%)
Efficiency (%)
100mA
70
80
70
65
60
60
55
55
50
0.01
0.1
1
10
100
Output Current IOUT (mA)
0.25mA
75
VOUT = 5V
L = 10mH LPS3314
COUT = 2x10mF
65
1.0mA
50
1000
VOUT = 5V
L = 10mH LPS3314
COUT = 2x10mF
6
Figure 5. Efficiency vs. Output Current, VOUT = 5V
7
8
9
10 11 12 13 14
Input Voltage VIN (V)
100
95
95
90
90
10mA
75
70
VOUT = 6.7V
L = 10mH LPS3314
COUT = 2x10mF
65
0.25mA
VOUT = 6.7V
L = 10mH LPS3314
COUT = 2x10mF
65
55
100
1000
Figure 7. Efficiency vs. Output current, VOUT = 6.8V
250mA
70
55
1
10
Output Currernt IOUT (mA)
17
1.0mA
75
60
0.1
16
80
60
50
0.01
100mA
85
Efficiency (%)
Efficiency (%)
85
VIN = 7.5V
VIN = 9V
VIN = 12V
VIN = 15V
15
Figure 6. Efficiency vs. Input Voltage, VOUT = 5V
100
80
0.1mA
50
7
8
9
10
11
12
13
14
Input Voltage VIN (V)
0.1mA
15
16
17
Figure 8. Efficiency vs. Input Voltage, VOUT = 6.8V
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100
100
95
95
90
90
10mA
250mA
1.0mA
85
85
VIN = 9.0V
VIN = 12V
VIN = 15V
80
Efficiency (%)
Efficiency (%)
100mA
75
VOUT = 8V
L = 10mH LPS3314
COUT = 10mF+22mF
70
65
80
0.25mA
75
70
60
60
55
55
50
0.01
0.1
1
10
100
Output Current IOUT (mA)
VOUT = 8V
L = 10mH LPS3314
COUT = 10mF + 22mF
65
50
1000
Figure 9. Efficiency vs. Output Current, VOUT = 8V
9
10
11
12
13
14
Input Voltage VIN (V)
0.1mA
15
16
17
Figure 10. Efficiency vs. Input Voltage, VOUT = 8V
100
100
95
95
90
90
85
85
10mA
100mA
250mA
VIN = 12V
VIN = 15V
80
Efficiency (%)
Efficiency (%)
1.0mA
75
VOUT = 10V
L = 10mH LPS3314
COUT = 10mF + 22mF
70
70
65
60
60
55
55
0.1
1
10
100
Output Currernt IOUT (mA)
1000
Figure 11. Efficiency vs. Output Current, VOUT = 10V
0.25mA
75
65
50
0.01
10
80
50
VOUT = 10V
L = 10mH LPS3314
COUT = 10mF+22mF
11
12
13
14
15
Input Voltage VIN (V)
0.1mA
16
17
Figure 12. Efficiency vs. Input Voltage, VOUT = 10V
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3.4
3.432
VOUT 3.3V
L = 15mH,
COUT = 10mF
VIN = 5V, TA = -40°C
VIN = 7.5V, TA = -40°C
VIN = 12V, TA = -40°C
3.35
Output Voltage (V)
3.366
3.333
VIN = 5V, TA= 25°C
VIN = 7.5V, TA = 25°C
VIN = 12V, TA = 25°C
VIN = 5V, TA = 85°C
VIN = 7.5V, TA = 85°
VIN = 12V, TA = 85°
3.300
3.267
3.234
0.01
0.1
1
10
Output Current [mA]
3.3
IOUT = 1mA
IOUT = 25mA
IOUT = 100mA
IOUT = 150mA
3.25
100
3.2
1000
5
6
7
8
9
10 11 12 13
Input Voltage (V)
14
15
16
17
Figure 13. Output Voltage vs. Output Current, VOUT = 3.3V
Figure 14. Output Voltage vs. Input Voltage, VOUT = 3.3V
5.20
5.15
5.15
Output Voltage [V]
VOUT 3.3V
L = 15mH,
COUT = 10mF
VOUT 5.0V
L = 10mH,
COUT = 2x10mF
5.10
VIN = 7.5V, TA = 25°C
VIN = 12V, TA = 25°C
VIN = 7.5V, TA = -40°C
VIN = 12V, TA = -40°C
VIN = 7.5V, TA = 85°
VIN = 12V, TA = 85°
5.05
5.05
5
5.00
4.95
4.95
4.9
4.90
0.01
0.1
1
10
Output Current [mA]
100
VOUT 5.0V
L = 10mH,
COUT = 2x10mF
5.1
Output Voltage (V)
Output Voltage [V]
3.399
1000
4.85
IOUT = 1mA
IOUT = 25mA
IOUT = 100mA
IOUT = 250mA
7
8
9
10
11
12
13
14
15
16
17
Input Voltage (V)
Figure 15. Output Voltage vs. Output current, VOUT = 5.0V
Figure 16. Output Voltage vs. Input Voltage, VOUT = 5V
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6.968
6.9
VOUT 6.7V
L = 10mH,
COUT = 2x10mF
6.85
6.901
VIN = 12V, TA = 25°C
VIN = 9V, TA = -40°C
6.834
6.8
VIN = 9V, TA = 25°C
Output Voltage [V]
Output Voltage [V]
VIN = 12V, TA = -40°C
6.767
VIN = 9V, TA = 85°
6.700
VIN = 12V, TA = 85°
VOUT 6.7V
L = 10mH,
COUT = 2x10mF
0.1
6.7
6.65
IOUT = 1mA
IOUT = 25mA
IOUT = 100mA
IOUT = 250mA
6.6
6.633
6.566
0.01
6.75
1
10
Output Current [mA]
100
6.55
6.5
1000
Figure 17. Output Voltage vs. Output Current, VOUT = 6.7V
8
11
12
13
14
Input Voltage [V]
15
16
17
8.24
VOUT 8.0V
L = 10mH,
COUT = 10mF + 22mF
8.24
VOUT 8.0V
L = 10mH,
COUT = 10mF + 22mF
8.16
8.16
VIN = 15V, TA = 25°C
Output Voltage [V]
VIN = 15V, TA = -40°C
Output Voltage [V]
10
Figure 18. Output voltage vs. Input voltage, VOUT = 6.7V
8.32
VIN = 12V, TA = -40°C
8.08
8.08
8
7.92
8.00
VIN = 12V, TA = 25°C
7.84
0.01
0.1
IOUT = 1mA
IOUT = 25mA
IOUT = 100mA
IOUT = 250mA
VIN = 12V, TA = 85°
VIN = 15V, TA = 85°
7.92
1
10
Output Current [mA]
7.84
100
1000
7.76
9
10
11
12
13
14
15
16
17
Input Voltage [V]
Figure 19. Output Voltage vs. Output Current, VOUT = 8V
12
9
Figure 20. Output Voltage vs. Input Voltage, VOUT = 8V
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1000
40
VIN = 12V
900
VOUT = 3.3V
L = 15mH
COUT = 10mF
Switch Frequency fSW( kHz)
Output Ripple Voltage (mVpp)
50
30
20
VIN = 15V
800
700
VIN = 7.5V
600
500
VIN = 5V
400
300
VIN = 5.0V
VIN = 7.5V
VIN = 12V
VIN = 15V
10
0
0
50
100
150
200
Output Current (mA)
250
100
0
300
Figure 21. Output Ripple Voltage vs. Output Current, VOUT
= 3.3V
1100
50
100
150
200
Output Current (mA)
250
300
Figure 22. Switch Frequency vs. Output Current, VOUT =
3.3V
1100
VIN = 15V
VIN = 12V
VIN = 7.5V
900
VIN = 15V
VIN = 12V
1000
Switch Frequency FSW ( kHz)
1000
Frequency ( kHz)
0
1200
1200
800
700
600
500
900
800
VIN = 10V
700
600
500
400
400
VOUT = 5.0V
L = 10mH
COUT = 2x10mF
300
200
300
VOUT = 8.0V
L = 10mH
COUT = 10mF + 22mF
200
100
100
0
VOUT = 3.3V
L = 15mH
COUT = 10mF
200
0
50
100
150
200
Output Current (mA)
250
300
Figure 23. Switch Frequency vs. Output Current, VOUT 5.0V
0
0
50
100
150
200
Output Current (mA)
250
300
Figure 24. Switch Frequency vs. Output Current, VOUT =
8V
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25
1.5
1.4
1.3
TA = 85°C
1.2
20
Quiescent Current IQ (mA)
Shutdown Current - ISD [mA]
1.1
TA = 60°C
1.0
0.9
TA = 25°C
0.8
TA = 0°C
0.7
0.6
TA = -40°C
0.5
IQ no switching: TA= 85°C
IQ no switching: TA= 60°C
15
IQ no switching: TA= 25°C
10
IQ no switching: TA= -40°C
0.4
0.3
IQ device switching: TA= 25°C,
VOUT = 1.8V, IOUT = 0mA no load,
EN = VIN
5
0.2
0.1
0
3
4
5
6
7
0
8 9 10 11 12 13 14 15 16 17
Input Voltage - VIN [V]
3
Figure 25. Shutdown Current vs. Input Voltage
4
5
6
7
Figure 26. Quiescent Current vs. Input Voltage
1000
25
VIN = 6V TA = 85°C
20
Quiescent Current (mA)
100
Quiescent Current (mA)
8 9 10 11 12 13 14 15 16 17
Input Voltage (V)
VIN = 6V TA = 25°C
VIN = 6V TA = 85°C
VIN = 6V TA = -40°C
VIN = 12V TA = 25°C
VIN = 12V TA = 85°C
VIN = 12V TA = −40°C
10
VIN = 12V TA = 85°C
VIN = 6V TA = 25°C
15
VIN = 12V TA = 25°C
VIN = 6V TA = -40°C
VIN = 12V TA = −40°C
10
1
5
0.1
0
200
400
600
800
1000
Voltage VEN (mV)
1200
1400
Figure 27. Quiescent Current vs. EN Voltage, rising VEN
14
0
0
200
400
600
800
1000
Voltage VEN (mV)
1200
1400
Figure 28. Quiescent Current vs. VEN Voltage, falling VEN
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1.25
1.245
1.24
1.235
1.23
1.225
1.22
1.215
1.21
1.205
1.2
1.195
1.19
1.185
1.18
1.175
1.17
1.165
1.16
1.155
1.15
SLVSAQ5A – MARCH 2012 – REVISED APRIL 2012
3
2.8
2.6
2.4
VTH EN ON TA = 25°C
VTH EN ON TA = -40°C
2.2
VTH EN ON TA = 85°C
TA = 85°C
2
RDSON (W)
EN Comparator Threshold - VTH EN (V)
www.ti.com
TA = 60°C
1.8
1.6
1.4
1.2
1
VTH EN OFF TA = 25°C
VTH EN OFF TA = -40°C
VTH EN OFF TA = 85°C
EN Comparator Threshold - VTH EN (V)
TA = 25°C
0.6
0.4
0.2
3
4
5
6
7
8 9 10 11 12 13 14 15 16 17
Input Voltage VIN (V)
0
3
4
5
6
7
8
9
10 11 12 13 14 15 16 17
Input Voltage VIN [V]
Figure 29. EN Comparator Thresholds vs. Input Voltage
1.25
1.245
1.24
1.235
1.23
1.225
1.22
1.215
1.21
1.205
1.2
1.195
1.19
1.185
1.18
1.175
1.17
1.165
1.16
1.155
1.15
TA = 0°C
TA = -40°C
0.8
Figure 30. RDSON High Side Switch
VOUT = 3.3 V
3.3V offset, 50mV/Div
ILoad = 1mA
VIN = 12V
L = 15 mH
COUT = 10 mF
VTH EN ON TA = 25°C
VTH EN ON TA = -40°C
VTH EN ON TA = 85°C
SW pin 10V/Div
VTH EN OFF TA = 25°C
VTH EN OFF TA = -40°C
3
4
5
6
7
Inductor current 200mA/Div
VTH EN OFF TA = 85°C
8 9 10 11 12 13 14 15 16 17
Input Voltage VIN (V)
Figure 31. RDSON Low Side Switch (Rectifier)
Figure 32. Power Save Mode VOUT=3.3V, IOUT = 1mA
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TPS62125
SLVSAQ5A – MARCH 2012 – REVISED APRIL 2012
VOUT = 3.3 V
3.3V offset, 50mV/Div
ILoad = 100mA
www.ti.com
VIN = 12V
L = 15 mH
COUT = 10 mF
VIN = 12V
L = 15 mH
COUT = 10 mF
VOUT = 3.3 V
3.3V offset, 50mV/Div
SW pin 10V/Div
SW pin 10V/Div
IOUT
5mA to 200mA
200mA/Div
Inductor current
200mA/Div
Inductor current 200mA/Div
Figure 33. PWM Mode VOUT= 3.3V, IOUT = 100mA
VOUT = 3.3 V
3.3V offset, 50mV/Div
VIN = 12V
L = 15 mH
COUT = 10 mF
Figure 34. Load Transient 5mA to 200mA, VOUT = 3.3V
VIN = 12V
L = 10 mH
COUT = 2x 10 mF
VOUT = 5.0 V
5V offset, 50mV/Div
SW pin 10V/Div
Inductor current
200mA/Div
IOUT
AC 10kHz
5mA to 200mA
200mA/Div
Inductor current 200mA/Div
ILoad = 1mA to 50mA
50mA / Div
Figure 35. AC Load Regulation, VOUT = 3.3V
VOUT = 5.0 V
5V offset, 50mV/Div
VIN = 12V
L = 10 mH
COUT = 2x 10 mF
VIN = 12V
L = 10 mH
COUT = 2x 10 mF
VOUT = 5.0 V
5V offset, 50mV/Div
Inductor current 200mA/Div
Inductor current 200mA/Div
ILoad = 10mA to 200mA
200mA / Div
ILoad = AC 5kHz 1mA to 250mA
200mA / Div
Figure 37. Load Transient 10mA to 200mA, VOUT = 5V
16
Figure 36. Load Transient 1mA to 50mA, VOUT = 5V
Figure 38. AC Load Regulation VOUT = 5V
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SLVSAQ5A – MARCH 2012 – REVISED APRIL 2012
VIN overshoot 25V
VIN = 9V to 12V
IOUT = 100mA
L = 15 mH
COUT = 10 mF
VIN = 12V, Hotplug
CIN = 10mF ceramic capacitor
VOUT
= 3.3 V, 50mV/Div
Current into
input capacitor 20A/div
IIN
Figure 39. Line transient response VIN = 9V to 12V
VIN overshoot reduction
to 15V
Figure 40. VIN Hotplug overshoot
VOUT = 5.0 V
2V/Div
VIN = 12V, Hotplug
CIN = 10mF
additional 22mF
tantalum-polymer input capacitor
type Poscap 20TQC22MYFB
VIN = 12V
CIN = 10uF ceramic
+ 22uF Poscap
VOUT
Startup
L = 10 mH
COUT = 2x 10 mF
IIN
200mA/Div
IIN
Current into CIN
IL
1A/Div
Current into
input capacitors 20A/div
IOUT
1A/Div
Figure 41. VIN Hotplug overshoot reduction with Poscap
Short
IOUT = 10mA
Short
Figure 42. Short circuit and overcurrent protection
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TPS62125
SLVSAQ5A – MARCH 2012 – REVISED APRIL 2012
VIN : ramped up/down
0V to 12V, 175mV/ms
2.5V/Div
REN1 = 820kW
REN2 = 110kW
REN2 = 82kW
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VOUT = 5.0 V
L = 10 mH
COUT = 2x10 mF
Load = 100W
VIN_Start = 10V
VIN : ramped up/down
0V to 12V, 175mV/ms
2.5V/Div
EN = VIN
VIN_Stop = 6V
UVLO
VOUT
2.5V/Div
VOUT
2.5V/Div
PG
5V/Div
PG
5V/Div
IIN
200mA/Div
IIN
200mA/Div
Figure 43. Input supply voltage supervisor (SVS), VOUT =
5.0V
VIN: 0.5mA current source
2.5V/Div
REN1 = 680kW
REN2 = 110kW
REN2 = 120kW
VIN_startup = 6.82V
VIN_stop = 4.55V
VOUT = 5.0 V
L = 10 mH
COUT = 2x10 mF
Load = 100W
VIN tracks VOUT
UVLO
Figure 44. Operation with EN = VIN, VIN tracks VOUT
CIN = 10mF ceramic + 22mF Poscap
VIN = 12V
VOUT = 1.8 V
L = 15 mH
COUT = 10 mF
Load = 180W
VOUT = 3.3 V
L = 15 mH, COUT = 10 mF
7ms/20mA Pulse Load
VOUT
2.0V/Div
Startup in 20mA Load
IOUT
20mA/Div
0.5mA Source Current
IIN: 0.5mA/Div
Figure 45. 0.5mA current source, 20mA pulse load
18
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Figure 46. Startup 1.8V VOUT
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VIN = 12V
VOUT = 3.3 V
L = 15 mH
COUT = 10 mF
Load = 330W
VIN = 12V
VOUT = 5.0 V
L = 10 mH
COUT = 2x10 mF
Load = 500W
Figure 47. Startup 3.3V VOUT
Figure 48. Startup 5.0V VOUT
VIN = 12V
VOUT = 3.3 V
COUT = 10uF
L = 15mH
RLoad = 100 W
RPullup PG = 100kW to VOUT
VIN = 12V
VOUT = 8.0 V
L = 10 mH
COUT = 2x10 mF
Load = 800W
1V/Div
2V/Div
1V/Div
Figure 49. Startup 8V VOUT
VIN
1V/Div
VIN_stop = 4.55V
REN1 = 680kW
REN2 = 110kW
REN_hys = 120kW
Figure 50. VOUT Ramp up/down with EN on/off
VIN = 12V to 0V
VOUT = 3.3 V
COUT = 10uF
L = 15mH
IOUT = 0mA
RPullup PG = 0W to VOUT
VOUT
1V/Div
VIN
2V/Div
VIN_stop = 4.55V
REN1 = 680kW
REN2 = 110kW
REN_hys = 120kW
VIN = 12V to 0V
VOUT = 3.3 V
COUT = 10uF
L = 15mH
IOUT = 0mA
RPullup PG = 100kW to VOUT
VOUT
1V/Div
PG
1V/Div
Figure 51. Output discharge using PG pin, triggered by EN
Comparator
Figure 52. VOUT ramp down with falling VIN, schematic
Figure Figure 60
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DETAILED DESCRIPTION
DCS - ControlTM
The TPS62125 high efficiency synchronous switch mode buck converter includes TI's DCS - Control™ (Direct
Control with Seamless Transition into Power Save Mode), an advanced regulation topology, which combines the
advantages of hysteretic and voltage mode control. Characteristics of DCS - ControlTM are excellent AC load
regulation and transient response, low output ripple voltage and a seamless transition between PFM and PWM
mode operation.
DCS - ControlTM includes an AC loop which senses the output voltage (VOS pin) and directly feeds the
information to a fast comparator stage. This comparator sets the switching frequency, which is constant for
steady state operating conditions, and provides immediate response to dynamic load changes. In order to
achieve accurate DC load regulation, a voltage feedback loop is used. The internally compensated regulation
network achieves fast and stable operation with small external components and low ESR capacitors. The DCS ControlTM topology supports PWM (Pulse Width Modulation) mode for medium and high load conditions and a
Power Save Mode at light loads. During PWM mode, it operates in continuous conduction. The switch frequency
is up to 1MHz with a controlled frequency variation depending on the input voltage. If the load current decreases,
the converter seamless enters Power Save Mode to maintain high efficiency down to very light loads. In Power
Save Mode the switching frequency varies linearly with the load current. Since DCS - ControlTM supports both
operation modes within one single building block, the transition from PWM to Power Save Mode is seamless
without effects on the output voltage. The TPS62125 offers both excellent DC voltage and superior load transient
regulation, combined with very low output voltage ripple, minimizing interference with RF circuits.
At high load currents the converter operates in quasi fixed frequency PWM mode operation and at light loads in
PFM (Pulse Frequency Modulation) mode to maintain highest efficiency over the full load current range. In PFM
Mode, the device generates a single switching pulse to ramp up the inductor current and recharge the output
capacitor, followed by a sleep period where most of the internal circuits are shutdown to achieve a quiescent
current of typically 13µA. During this time, the load current is supported by the output capacitor. The duration of
the sleep period depends on the load current and the inductor peak current.
Pulse Width Modulation (PWM) Operation
The TPS62125 operates with pulse width modulation in continuous conduction mode (CCM) with a nominal
switching frequency of about 1MHz. The frequency variation in PWM mode is controlled and depends on VIN,
VOUT and the inductance. The device operates in PWM mode as long the output current is higher than half the
inductor's ripple current. To maintain high efficiency at light loads, the device enters Power Save Mode at the
boundary to discontinuous conduction mode (DCM). This happens if the output current becomes smaller than
half the inductor's ripple current.
Power Save Mode
With decreasing load current, the TPS62125 transitions seamlessly from PWM Mode to Power Save Mode once
the inductor current becomes discontinuous. This ensures a high efficiency at light loads. In Power Save Mode
the converter operates in Pulse Frequency Modulation (PFM Mode) and the switching frequency decreases
linearly with the load current. DCS - ControlTM features a small and predictable output voltage ripple in Power
Save Mode. The transition between PWM Mode and Power Save Mode occurs seamlessly in both directions.
The minimum On Time TONmin for a single pulse can be estimated by:
TON =
VOUT
´ 1ms
VIN
(1)
Therefore the peak inductor current in PFM mode is approximately:
(VIN - VOUT ) ´ T
ILPFMpeak =
ON
L
(2)
The transition from PFM mode to PWM mode operation and back occurs at a load current of approximately ½
ILPFMpeak.
With:
20
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TON: high-side MOSFET switch on time [µs]
VIN: Input voltage [V]
VOUT: Output voltage [V]
L : Inductance [µH]
ILPFMpeak : PFM inductor peak current [mA]
The maximum switching frequency can be estimated by:
f SW max »
1
= 1MHz
1ms
(3)
100% DUTY CYCLE LOW DROPOUT OPERATION
The device increases the On Time of the high-side MOSFET switch as the input voltage comes close to the
output voltage in order to keep the output voltage in regulation. This reduces the switching frequency.
With further decreasing input voltage VIN, the high-side MOSFET switch is turned on completely. In this case,
the converter provides a low input-to-output voltage difference. This is particularly useful in applications with a
widely variable supply voltage to achieve longest operation time by taking full advantage of the whole supply
voltage span.
The minimum input voltage to maintain output voltage regulation depends on the load current and output voltage,
and can be calculated as:
VIm in = VOUT min + I OUT ´ ( RDSON max + RL )
(4)
With:
IOUT = output current
RDS(ON)max = maximum high side switch RDS(ON).
RL = DC resistance of the inductor
VOUTmin = minimum output voltage the load can accept
UNDER-VOLTAGE LOCKOUT
In addition to the EN Comparator, the device includes an under-voltage lockout circuit which prevents the device
from misoperation at low input voltages. Both circuits are fed to an AND gate and prevents the converter from
turning on the high-side MOSFET switch or low-side MOSFET under undefined conditions. The UVLO threshold
is set to 2.9V typical for rising VIN and 2.8V typical for falling VIN. The hysteresis between rising and falling UVLO
threshold ensures proper start up. Fully functional operation is permitted for an input voltage down to the falling
UVLO threshold level. The converter starts operation again once the input voltage trips the rising UVLO threshold
level and the voltage at the EN pin trips VTH_EN_ON.
SOFT START
The TPS62125 has an internal soft-start circuit which controls the ramp up of the output voltage and limits the
inrush current during start-up. This limits input voltage drop.
The soft-start system generates a monotonic ramp up of the output voltage and reaches an output voltage of
1.8V typ. within 240µs after the EN pin was pulled high. For higher output voltages, the ramp up time of the
output voltage can be estimated with a ramp up slew rate of about 12mV/us. TPS62125 is able to start into a pre
biased output capacitor. The converter starts with the applied bias voltage and ramps the output voltage to its
nominal value. In case the output voltage is higher than the nominal value, the device starts switching once the
output has been discharged by an external load or leakage current to its nominal output voltage value.
During start up the device can provide an output current of half of the high-side MOSFET switch current limit
ILIMF. Large output capacitors and high load currents may exceed the current capability of the device during start
up. In this case the start up ramp of the output voltage will be slower.
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ENABLE COMPARATOR (EN / EN_hys)
The EN pin is connected to an On/Shutdown detector (ON/SD) and an input of the Enable Comparator. With a
voltage level of 0.4V or less at the EN pin, the ON/SD detector turns the device into Shutdown mode and the
quiescent current is reduced to typically 350nA. In this mode the EN comparator as well the entire internal-control
circuitry are switched off. A voltage level of typ. 900mV (rising) at the EN pin triggers the ON/Shutdown detector
and activates the internal reference VREF (Typ.1.2V), the EN comparator and the UVLO comparator. In
applications with slow rising voltage levels at the EN pin, the quiescent current profile before this trip point needs
to be considered, see Figure 27. Once the ON/SD detector has tripped, the quiescent current consumption of the
device is typ. 6µA. The TPS62125 starts regulation once the voltage at the EN pin trips the threshold VEN_TH ON
(typ. 1.2V) and the input voltage is above the UVLO threshold. It enters softstart and ramps up the output
voltage. For proper operation, the EN pin must be terminated and must not be left floating. The quiescent current
consumption of the TPS62125 is typ. 13µA under no load condition (not switching). See Figure 25. The DC/DC
regulator stops operation once the voltage on the EN pin falls below the threshold VEN_TH OFF (typ. 1.15V) or the
input voltage falls below UVLO threshold. The enable comparator features a built in hysteresis of typ. 50mV. This
hysteresis can be increased with an external resistor connected to pin EN_hys. See more details in application
information section.
POWER GOOD OUTPUT / OUTPUT DISCHARGE (PG)
The Power Good Output (PG pin) is an open drain output. The circuit is active once the device is enabled. It is
driven by an internal comparator connected to the FB pin voltage and an internal reference. The PG output
provides a high level (open drain high impedance) once the feedback voltage exceeds typical 95% of its nominal
value. The PG output is driven to low level once the FB pin voltage falls below typ. 90% of its nominal value
VREF_FB. The PG output goes high (high impedance) with a delay of typically 2µs. A pull up resistor is needed to
generate a high level. The PG pin can be connected via a pull up resistors to a voltage up to 10V. This pin can
also be used to discharge the output capacitor. See section Application Information for more details.
The PG output is pulled low if the voltage on the EN pin falls below the threshold VEN_TH
is below the undervoltage lockout threshold UVLO.
OFF
or the input voltage
SHORT-CIRCUIT PROTECTION
The TPS62125 integrates a high-side MOSFET switch current limit, ILIMF, to protect the device against a short
circuit. The current in the high-side MOSFET switch is monitored by a current limit comparator and once the
current reaches the limit of ILIMF , the high-side MOSFET switch is turned off and the low-side MOSFET switch is
turned on to ramp down the inductor current. The high-side MOSFET switch is turned on again once the zero
current comparator trips and the inductor current has become zero. In this case, the output current is limited to
half of the high-side MOSFET switch current limit, ½ ILIMF, typ. 300mA.
THERMAL SHUTDOWN
As soon as the junction temperature, TJ, exceeds 150°C (typical) the device goes into thermal shutdown. In this
mode, the high-side and low-side MOSFETs are turned-off. The device continues its operation when the junction
temperature falls below the thermal shutdown hysteresis.
22
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APPLICATION INFORMATION
OUTPUT VOLTAGE SETTING
The output voltage can be calculated by:
æ
R ö
VOUT = VREF _ FB ´ çç1 + 1 ÷÷
è R2 ø
æ
R
VOUT = 0.8V ´ çç1 + 1
è R2
ö
æV
R1 = R2 ´ ç OUT - 1÷
ø
è 0.8V
ö
÷÷
ø
(5)
The internal reference voltage for the error amplifier,VREF_FB, is nominal 0.808V. However for the feedback
resistor divider selection, it is recommended to use the value 0.800V as the reference. Using this value, the
output voltage sets 1% higher and provides more headroom for load transients as well for line and load
regulation. The current through the feedback resistors R1 and R2 should be higher than 1µA. In applications
operating over full temperature range or in noisy environments, this current may be increased for robust
operation. However, higher currents through the feedback resistors impact the light load efficiency of the
converter.
Table 2 shows a selection of suggested values for the feedback divider network for most common output
voltages.
Table 2. Suggested Values for Feedback Divider Network
Output Voltage
1.2V
1.8V
3.3V
5V
6.7V
8V
R1 [kΩ]
180
300
1800
1100
1475
1800
R2 [kΩ]
360
240
576
210
200
200
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ENABLE THRESHOLD AND HYSTERESIS SETTING
ON/SD
VIN
VIN
VREF
1.2V
REN1
EN
EN
Comparator
VTH_EN
REN2
EN_hys
REN HYS
GND
Figure 53. Using the Enable Comparator Threshold and Hysteresis for an input SVS (supply voltage
supervisor)
The enable comparator can be used as an adjustable input supply voltage supervisor (SVS) to start and stop the
DC/DC converter depending on the input voltage level. The input voltage level, VIN_startup, at which the device
starts up is set by the resistors REN1 and REN2 and can be calculated by :
æ
R
V IN _ startup = V EN _ TH _ ON ´ çç1 + EN 1
REN 2
è
ö
æ
R
÷÷ = 1.2V ´ çç1 + EN 1
REN 2
ø
è
ö
÷÷
ø
(6)
The resistor values REN1 and REN2 can be calculated by:
æ V
ö
æV
ö
R E N 1 = R E N 2 ´ ç IN _ startup - 1 ÷ = R E N 2 ´ çç IN _ startup - 1 ÷÷
çV
÷
è 1 . 2V
ø
è E N _ TH _ O N
ø
R EN 2 =
R EN 1
ö
æ V IN _ startup
ç
- 1÷
÷
çV
ø
è EN _ TH _ O N
=
(7)
R EN 1
ö
æ V IN _ startup
çç
- 1 ÷÷
ø
è 1 . 2V
(8)
The input voltage level VIN_stop at which the device will stop operation is set by REN1, REN2 and REN
be calculated by:
æ
R EN 1
V IN _ stop = V EN _ TH _ OFF ´ ç 1 +
ç
R EN 2 + R EN _ hys
è
ö
æ
R EN 1
÷ = 1 . 15V ´ ç 1 +
÷
ç
R EN 2 + R EN _ hys
ø
è
ö
÷
÷
ø
HYS
and can
(9)
The resistor value REN_hyscan be calculated according to:
R EN _ hys =
R EN 1
æ V IN _ stop
ö
ç
- 1÷
çV
÷
è EN _ TH _ OFF
ø
- R EN 2 =
R EN 1
ö
æ V IN _ stop
çç
- 1 ÷÷
ø
è 1 .15V
- R EN 2
(10)
The current through the resistors REN1, REN2 and REN HYS should be higher than 1µA. In applications operating
over the full temperature range and in noisy environments, the resistor values can be reduced to smaller values.
24
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VIN
DC/DC start
VIN_startup
Hysteresis
VIN_stop
DC/DC stop
VOUT
Proper VOUT ramp up
Figure 54. Using the EN comparator as input SVS for proper VOUT ramp up
POWER GOOD (PG) PULL UP / OUTPUT DISCHARGE RESISTOR
The Power Good open collector output needs an external pull up resistor to indicate a high level. The pull up
resistor can be connected to a voltage level up to 10V. The output can sink current up to 0.4mA with specifed
output low level of less than 0.3V. The lowest value for the pull up resistor can be calculated by:
R Pullup min =
VOUT - 0.3V
0.0004 A
(11)
VOUT
TPS62125
COUT
RPullup
RIPG
PG Comparator
PG
VPG Power Good
max 10V
VTH_PG
FB
Figure 55. PG open collector output
The PG pin can be used to discharge the output capacitor. The PG output has an internal resistance RIPGof
typical 600Ω and minimum 400Ω. The maximum sink current into the PG pin is 10mA. In order to limit the
discharge current to the maximum allowable sink current into the PG pin, the external pull up resistor RPull up can
be calculated to:
RPullup min =
VOUT
I PG _ max
- RIPG _ min =
VOUT
- 400W
0.01A
(12)
In case a negative value is calculated, the external pull up resistor can be removed and the PG pin can be
directly connected to the output.
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OUTPUT FILTER DESIGN (INDUCTOR AND OUTPUT CAPACITOR)
The external components have to fulfill the needs of the application, but also the stability criteria of the devices
control loop. The TPS62125 is optimized to work within a range of L and C combinations. The LC output filter
inductance and capacitance have to be considered together, creating a double pole, responsible for the corner
frequency of the converter. Table 3 can be used to simplify the output filter component selection.
Table 3. Recommended LC Output Filter Combinations
Output Capacitor Value [µF] (2)
Inductor Value
[µH] (1)
10µF
15
√
22
√ (3)
15
√ (3)
22
√ (3)
2 x 10µF
22µF
47µF
√
√
√
√
√
√
√
√
√
√
√
√
VOUT 1.2V - 1.8V
VOUT 1.8V - 3.3V
VOUT 3.3V - 5V
10
√
√
√
15
√ (3)
√ (3)
√
22
VOUT 5V - 10V
(1)
(2)
(3)
10
√ (3)
√ (3)
√
15
√
√
√
22
√
√
√
Inductor tolerance and current de-rating is anticipated. The effective inductance can vary by 20% and -30%.
Capacitance tolerance and bias voltage de-rating is anticipated. The effective capacitance can vary by 20% and -50%.
This LC combination is the standard value and recommended for most applications.
More detailed information on further LC combinations can be found in application note SLVA515.
INDUCTOR SELECTION
The inductor value affects its peak-to-peak ripple current, the PWM-to-PFM transition point, the output voltage
ripple and the efficiency. The selected inductor has to be rated for its DC resistance and saturation current. The
inductor ripple current (ΔIL) decreases with higher inductance and increases with higher VIN or VOUT and can be
estimated according to Equation 13.
Equation 14 calculates the maximum inductor current under static load conditions. The saturation current of the
inductor should be rated higher than the maximum inductor current as calculated with Equation 14. This is
recommended because during heavy load transient the inductor current will rise above the calculated value. A
more conservative way is to select the inductor saturation current according to the high-side MOSFET switch
current limit ILIMF.
(VIN - VOUT )
DIL =
´ TON
L
(13)
ΔIL
ILmax = Ioutmax +
2
(14)
With:
TON = see equation (3)
L = Inductance
ΔIL = Peak to Peak inductor ripple current
ILmax = Maximum Inductor current
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In DC/DC converter applications, the efficiency is essentially affected by the inductor AC resistance (i.e. quality
factor) and by the inductor DCR value. To achieve high efficiency operation, care should be taken in selecting
inductors featuring a quality factor above 25 at the switching frequency. Increasing the inductor value produces
lower RMS currents, but degrades transient response. For a given physical inductor size, increased inductance
usually results in an inductor with lower saturation current.
The total losses of the coil consist of both the losses in the DC resistance (RDC) and the following frequencydependent components:
• The losses in the core material (magnetic hysteresis loss, especially at high switching frequencies)
• Additional losses in the conductor from the skin effect (current displacement at high frequencies)
• Magnetic field losses of the neighboring windings (proximity effect)
• Radiation losses
The following inductor series from different suppliers have been used with the TPS62125.
Table 4. List of Inductors
INDUCTANCE
[µH]
DCR [Ω]
DIMENSIONS
[mm3]
INDUCTOR
TYPE
SUPPLIER
10 / 15
0.33 max / 0.44 max
3.3 x 3.3 x 1.4
LPS3314
Coilcraft
22
0.36 max
3.9 x 3.9 x 1.8
LPS4018
Coilcraft
15
0.33 max
3.0 x 2.5 x 1.5
VLF302515
TDK
10/15
0.44 max / 0.7 max
3.0 x 3.0 x 1.5
LPS3015
Coilcraft
10
0.38 typ.
3.2 × 2.5 × 1.7
LQH32PN
Murata
OUTPUT CAPACITOR SELECTION
Ceramic capacitors with low ESR values provide the lowest output voltage ripple and are recommended. The
output capacitor requires either an X7R or X5R dielectric. Y5V and Z5U dielectric capacitors, aside from their
wide variation in capacitance over temperature, become resistive at high frequencies.
At light load currents the converter operates in Power Save Mode and the output voltage ripple is dependent on
the output capacitor value and the PFM peak inductor current. Higher output capacitor values minimize the
voltage ripple in PFM Mode and tighten DC output accuracy in PFM Mode. In order to achieve specified
regulation performance and low output voltage ripple, the DC-bias characteristic of ceramic capacitors must be
considered. The effective capacitance of ceramic capacitors drops with increasing DC - bias Voltage. Due to this
effect, it is recommended for output voltages above 3.3V to use at least 1 x 22µF or 2 x 10µF ceramic capacitors
on the output.
INPUT CAPACITOR SELECTION
Because of the nature of the buck converter having a pulsating input current, a low ESR input capacitor is
required for best input voltage filtering and minimizing the interference with other circuits caused by high input
voltage spikes. For most applications, a 10µF ceramic capacitor is recommended. The voltage rating and DC
bias characteristic of ceramic capacitors need to be considered. The input capacitor can be increased without
any limit for better input voltage filtering.
For applications powered from high impedance sources, a tantalum polymer capacitor should be used to buffer
the input voltage for the TPS62125. Tantalum polymer capacitors provide a constant capacitance vs. DC bias
characteristic compared to ceramic capacitors. In this case, a 10µF ceramic capacitor should be used in parallel
to the tantalum polymer capacitor to provide low ESR.
Take care when using only small ceramic input capacitors. When a ceramic capacitor is used at the input and the
power is being supplied through long wires, such as from a wall adapter, a load step at the output or VIN step on
the input can induce large ringing at the VIN pin. This ringing can couple to the output and be mistaken as loop
instability or could even damage the part by exceeding the maximum ratings. In case the power is supplied via a
connector e.g. from a wall adapter, a hot-plug event can cause voltage overshoots on the VIN pin exceeding the
absolute maximum ratings and can damage the device, too. In this case a tantalum polymer capacitor or
overvoltage protection circuit reduces the voltage overshoot, see Figure 41.
Table 5 shows a list of input/output capacitors.
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Table 5. List of Capacitor
CAPACITANCE
[µF]
SIZE
CAPACITOR TYPE
USAGE
SUPPLIER
10
0805
GRM21B 25V X5R
CIN /COUT
Murata
10
0805
GRM21B 16V X5R
COUT
Murata
22
1206
GRM31CR61 16V X5R
COUT
Murata
22
B2 (3.5x2.8x1.9)
20TQC22MYFB
CIN / input
protection
Sanyo
LAYOUT CONSIDERATIONS
As for all switching power supplies, the layout is an important step in the design. Proper function of the device
demands careful attention to PCB layout. Care must be taken in board layout to get the specified performance. If
the layout is not carefully done, the regulator could show frequency variations, poor line and/or load regulation,
stability issues as well as EMI problems. It is critical to provide a low inductance, low impedance ground path.
Therefore, use wide and short traces for the paths conducting AC current of the DC/DC converter. The area of
the AC current loop (input capacitor - TPS62125 - inductor - output capacitor) should be routed as small as
possible to avoid magnetic field radiation. Therefore the input capacitor should be placed as close as possible to
the IC pins as well as the inductor and output capacitor. Use a common Power GND node and a different node
for the signal GND to minimize the effects of ground noise. Keep the common path to the GND pin, which returns
both the small signal components and the high current of the output capacitors as short as possible to avoid
ground noise. A well proven practice is to merge small signal GND and power GND path at the exposed thermal
pad. The FB divider network and the FB line should be routed away from the inductor and the SW node to avoid
noise coupling. The VOS line should be connected as short as possible to the output, ideally to the VOUT
terminal of the inductor. Keep the area of the loop VOS node - inductor - SW node small. The Exposed Thermal
Pad must be soldered to the circuit board for mechanical reliability and to achieve appropriate power dissipation.
COUT
L
PG
GND
Approximate circuit area
2
2
= 51mm (0.079in )
VOUT
CIN
VIN
REN1
REN2
U1
GND
R1
R2
REN_hys
Figure 56. EVM board Layout
28
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TYPACAL APPLICATIONS
TPS62125
VIN = 4V to 17V
VIN
SW
R1
1.8M
VOS
EN
CIN
10µF
VOUT = 3.3V
up to 300mA
L 15µH
FB
COUT
10µF
R2
576k
EN_hys
Rpullup
GND
PG
PWR GOOD
Figure 57. TPS62125 3.3V Output Voltage Configuration
TPS62125
VIN = 6V to 17V
VIN
SW
R1
1.1M
VOS
EN
CIN
10µF
VOUT = 5V
up to 300mA
L 10µH
FB
R2
210k
EN_hys
COUT
2 x 10µF
or 1 x 22µF
Rpullup
GND
PG
PWR GOOD
Figure 58. TPS62125 5.0V Output Voltage Configuration
VIN_Start = 10V
VIN_Stop = 6V
TPS62125
VIN
REN1
820k
CIN
10µF
REN2
110k
SW
R1
1.1M
VOS
R2
EN
VOUT = 5V
up to 300mA
L 10µH
FB
R2
210k
COUT
2 x 10µF
or 1 x 22µF
EN_hys
REN_hys
82k
Rpullup
GND
PG
PWR GOOD
Figure 59. TPS62125 5V VOUT, Start up voltage VIN_Start = 10V, Stop voltage VIN_Stop = 6V, see Figure 43
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VIN_Start = 6.82V
VIN_Stop = 4.55V
TPS62125
SW
VIN
Current
Source
0.5mA
10V max
REN1
610k
CIN
22µF
Poscap
CIN
10µF
R1
1.8M
VOS
FB
EN
R2
REN2
110k
VOUT = 3.3V
up to 300mA
L 15µH
COUT
10µF
R2
576k
EN_hys
REN_hys
120k
Rpullup
100k
GND
PG
PWR Good
Figure 60. TPS62125 operation from a storage capacitor charged from a 0.5mA current source, VOUT =
3.3V, see Figure 45
CIN
10µF
TPS62125
VIN = 5V
VOS
EN
GND
R1
1.1M
FB
Cbypass
10µF
R2
210k
EN_hys
GND
L 10µH
SW
VIN
PG
COUT
2 x 10µF
or 1 x 22µF
VOUT = - 5V
up to 150mA
Figure 61. 5V to -5V inverter configuration, see SLVA514
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PACKAGE OPTION ADDENDUM
www.ti.com
9-May-2012
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package
Drawing
Pins
Package Qty
Eco Plan
(2)
Lead/
Ball Finish
MSL Peak Temp
(3)
TPS62125DSGR
ACTIVE
WSON
DSG
8
3000
Green (RoHS
& no Sb/Br)
CU NIPDAU Level-2-260C-1 YEAR
TPS62125DSGT
ACTIVE
WSON
DSG
8
250
Green (RoHS
& no Sb/Br)
CU NIPDAU Level-2-260C-1 YEAR
Samples
(Requires Login)
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
9-May-2012
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
B0
(mm)
K0
(mm)
P1
(mm)
W
Pin1
(mm) Quadrant
TPS62125DSGR
WSON
DSG
8
3000
180.0
8.4
2.3
2.3
1.15
4.0
8.0
Q2
TPS62125DSGT
WSON
DSG
8
250
180.0
8.4
2.3
2.3
1.15
4.0
8.0
Q2
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
9-May-2012
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
TPS62125DSGR
WSON
DSG
8
3000
210.0
185.0
35.0
TPS62125DSGT
WSON
DSG
8
250
210.0
185.0
35.0
Pack Materials-Page 2
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microcontroller.ti.com
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