AN708
Vishay Siliconix
Low-Power Universal-Input Power Supply
Achieves High Efficiency
Expanding global markets have created a demand for what
have become known as universal-input power supplies –– that
is, power supplies that allow devices to be plugged into wall
outlets anywhere in the world. These power supplies must be
able to operate directly from 100-, 110-, and 220-V ac power
lines without the use of selector switches or jumpers. A power
supply with the ability to operate under such conditions while
remaining cost-effective is now becoming a necessity.
In the under 30-W power range, meeting the above
requirements while maintaining high efficiency has been a
challenge. Add to this the need to meet various international
safety standards, and the circuit designer has his hands full.
The demands of low-power universal-input power supplies are
met by the Si9120 pulse width modulation (PWM) controller
from Vishay Siliconix. Using the Si9120, the flyback circuit
presented in this application note demonstrates that designing
universal-input supplies can be a simple task.
For the low power levels that are of interest here (under 30 W),
the discontinuous-mode (DCM) flyback converter is the
preferred topology. The biggest advantage of this topology is
simplicity. The parts count in the power path cannot get any
lower.
The peak-to-average primary current ratio in a DCM flyback is
high relative to other topologies; however, at low power levels,
this is not a serious drawback. On-state losses are minimal.
Magnetics are small. Also, the transformer reset voltage is set
by the minimum input voltage and remains fairly constant as
the line voltage changes. As a result, a 600-V MOSFET proves
adequate, even with ac inputs up to 300 V RMS.
The DCM flyback converter, when operated under
current-mode control, provides a natural input volt-second
limit, which helps keep the drain voltage from getting out of
control during line or load transient conditions. Also, today’s
power MOSFETs are able to withstand avalanche current
many times greater than a low power circuit can typically
deliver (see appendix A). As such, the MOSFET will serve as
a clamp for the occasional spike which may result from a short
circuit or extreme load transient.
Cross regulation is fairly good, especially if leakage
inductance between windings can be kept low.[1] In a
universal-input application, meeting VDE input-to-output
isolation requirements is essential. Depending on the end
product, this can be as high as 3750-V RMS, primary to
Document Number: 70581
secondary –– a figure that is totally inconsistent with the desire
to achieve low leakage inductance. As a result, cross
regulation between primary and secondary- referenced
windings will be poor. This complicates the regulation of the
primary-side
bootstrap
winding
used
to
avoid
secondary-to-primary feedback across the isolation boundary.
The addition of a simple spike-blanking circuit solves the
problem (see AN707, “Designing Low-Power Off-Line Flyback
Converters Using the Si9120 Switchmode Controller IC”).
When using the Si9120 for universal-input applications, it is
recommended that a bootstrap winding be employed. While
not strictly necessary, the power dissipation and chip
temperature are higher if bootstrapping is not utilized. As an
example, at VIN = 400 V dc and ICC = 1.5 mA, the power
dissipation on the chip without a bootstrap is 600 mW. If a 10 V
bootstrap supply is used, the dissipation is only 15 mW. This
becomes more of a concern as the gate charge requirements
of the power MOSFET increase, since the value of ICC for the
controller is largely dependent on gate drive demands.
Another advantage of the DCM flyback converter is its
single-pole loop response. This makes compensating the
feedback loop comparatively simple. In addition, transient
response can be quite good in DCM flyback converters. It is
possible (though not practical in a closed-loop system) to slew
the power stage from no load to full load in only one switching
cycle.
The circuit shown in Figure 1 is an 11.1-W, 3-output off-line
supply. The input voltage is specified from 90- to 260-V ac.
Outputs are +5 V at 1.5 A, +12 V at 150 mA, and –12 V at
150 mA. The design features full VDE isolation, primary side
regulation, and true foldback current limiting. Operating
frequency is 100 kHz.
DCM flyback operating principles are generally well
understood and will not be presented here. Refer to
Vishay Siliconix Application Note AN707 for a detailed design
example. References 2 and 3 are also recommended.
Sizing the input capacitor and rectifiers for universal input
requires more thought than for comparable single-input
converters. Keep in mind that while the maximum input voltage
occurs at high-input line, the maximum current stresses will
occur at low line. This implies that the input capacitor value
must be sized at low line while the voltage rating is dictated by
the high-line condition. The bridge rectifier should be rated at
600-V dc minimum. The RMS current rating is calculated
below.
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AN708
Vishay Siliconix
L1
ac In
90 – 260 V ac
50/60 Hz
D3
DB105
C17
0.1 F
X
C18
0.1 F
X
8 mH
C20
0.0047 F
Y
C19
0.0047 F
Y
+ C8
0.0047 F
450 V
C7
33 F
450 V
T1
+12 V, 150 mA
D7
MUR110
+ C21
2200 F
16 V
C22
0.47 F
50 V
RTN
R16
10 k
C6
1000 pF
D1
R2
175 k
RTN
R14
390 k
6
10
C10
0.1 F
50 V
+ C4
2200 F
16 V
U1
V CC
7
D8
5
C5
0.47 F
50 V
OUT
R4
10 4
SENSE
14
Q1
SMP4N60
–12 V, 150 mA
R3
1 k
FB
COMP
OSC
IN
11
DIS
V IN
13
Si9120
V–
SD
RESET
BIAS
12
9
V REF
16
8
OSC
OUT
MUR110
1
15
R9
2 k
C15
N/U
C11
4700 pF
R5
1.3 R1
10 C12
1000 pF
100 V
D5
R6
20 /W
1N4148
D4
D2
1N4148
1N5822
D6
R13
75 k
R12
8.2 k
R10
130 k 1N4148
C9
220 pF
C6
1000 pF
100 V
L2
6 H
+ C1
2200 F
6.3 V
C2
1000 F
6.3 V
+5 V, 1.5 A
+
C3
0.1 F
50 V
RTN
R7
1.2 k
C14
1 F
50 V
Q3
2N4403
C13
4700 pF Q2
2N7000
R15
40.2 k
R11
68 k
R8
680 FIGURE 1. Schematic for Universal-Input Power Supply
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Document Number: 70581
AN708
Vishay Siliconix
For the present example:
Output power = 5.0 V x 1.5 A + 12 V x 0.15 A x 2 = 11.1 W
choosing an operating frequency and calculating the peak
primary current, a value for primary inductance, LP , can be
determined as follows:
If efficiency is assumed to be 70%,
For 100 kHz, period = 10 s.
Input power = 11.1 W/0.7 = 15.86 W
At 50% duty factor, ton(max) = 5 s.
For a little cushion, assume a low-input line voltage of 85 V ac.
Ipk = Iin x 4
= (0.1586 A) (4)
Thus,
V dc 85 2 120V dc.
= 0.634 A pk.
For VIN (dc) = 100 V
Assuming a 20-V pk-pk input-capacitor ripple voltage the
minimum voltage is
Vmin = 120 V – 20 V = 100 V.
Iin = Pin/VDC
= 15.86 W/ 100 V = 0.1586 A
C I dt
dv
0.1586 A x 0.01 s
20 V
L V dt
I pk
(100 V) (5E–6 s)
0.634 A
= 788 H.
The actual inductance used was 735 H. [For high-volume
production applications, the design engineer should consider
the worst case tolerances for clock frequency and inductor
value.]
= 79 F
68 F is a standard value. With 68 F, the ripple voltage is
V pp I dt
C
0.16 A 0.01 s
68E–6F
= 23.5 V, an acceptable value.
The capacitor voltage rating is calculated:
V max 260 V ac x 2 368 V dc.
A 400-V capacitor is acceptable. A rating of 450-V dc is
preferable if high reliability is required or significant line
transients are expected.
See AN707 for transformer design equations and a fully
worked example.
The biggest considerations for universal input are related to
the additional insulation required to comply with VDE isolation
specifications. The physical space occupied by the insulation
typically reduces the useable fill factor to 25%. Furthermore,
the increase in leakage inductance caused by large physical
separation of the windings has the undesirable effects of
creating large voltage spikes on the power MOSFET drain,
contributing to power losses, and degrading load regulation.
Barrier tape at window ends will take up a lot of useable space,
so a core geometry with a long, low window should be selected
to minimize wasted area. This has the added benefit of
reducing leakage inductance. (See equation 6.4 of
reference 4.)
Assuming a power factor of 0.65, the RMS input current is
I ac Po
hac (PF)
11.1 W
(0.7) (85 V) (0.65)
= 0.287 A
A 1-A bridge rectifier is more than adequate.
The primary inductance value is chosen by analyzing the
lowest input voltage case. For a given load, the value of the
peak transformer primary current will remain constant
regardless of the input voltage. Since the primary inductance
is fixed, the time to ramp to a given value of current is inversely
proportional to input voltage (V = Ldi/dt). Therefore, low line is
where the most time is needed to ramp to the desired primary
current. The duty factor limit dictates an on-time limit. After
Document Number: 70581
Wind the primary first. Apply the required insulation, and then
wind the secondaries. All secondaries should be wound
together with no intervening insulation, if voltage levels allow.
Optimal cross regulation is achieved in this way.
Further reductions in leakage inductance can be realized by
using interleaved windings. First wind one half of the primary,
followed by the secondaries and remaining primary turns. The
multiple primaries are usually connected in parallel. The spike
blanking circuit described in AN707 virtually eliminates the
primary-to-secondary leakage inductance problems, at least
from the standpoint of the regulation effects.
In selecting a power MOSFET, the main concerns will be the
rDS(on) and the drain voltage ratings. The transformer primary
voltage during the off time is VP = (Vo + VD) NP/NS. Using the
5-V winding,
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AN708
Vishay Siliconix
VP = (5.0 V + 0.4 V)(45 T/3 T) = 81
Therefore,
VDS(off) = VIN(max) + VP = 368 V + 81 V = 449 V.
A 600 V MOSFET allows for a 150 V spike due to leakage
inductance at high line. The RC snubber was sized empirically
to keep the peak drain voltage below 600 V.
The SMP4N60 is the smallest 600-V device available. At 25_C
the rDS(on) is 2.0 . At 100_C, rDS(on) = 1.75 x 2 = 3.5 . The
peak drain current was previously calculated at 0.634 A. The
maximum RMS drain current is given by
I RMS + Ipk
ǒǓ
D
3
1
2
+ 0.634A
ǒǸ Ǔ
0.5 + 0.26 A
3
On-state losses are given by
Pon = IRMS2 x rDS(on)
= (0.26 A)2 x 3.5 = 237 mW.
Switching losses are estimated at 350 mW. Since the thermal
resistance is specified at 80_C/W, a total temperature rise of
47_C is expected. This permits operation up to approximately
50_C ambient temperature, while holding the maximum
junction temperature to 100_C.
Something of more concern for universal-input than for a
single-input voltage supply is the range of duty factor to be
expected. Since the on time varies inversely with input voltage,
the high-line on-time can become quite small in a
high-frequency converter. For this kind of application, try to
keep the minimum on time to not much less than 1 s. This will
help minimize noise problems with the current sense.
Also, be sure to use a non-inductive resistor for the current
sense (carbon composition or film type). Use of a wire-wound
resistor will produce large spikes which have to be filtered out.
The dual-delay current-limit comparators of the Si9120 will
frequently eliminate the need for a current-sense filter
altogether. The magnitude of the noise on the current sense
voltage will be affected by transformer parasitic capacitances
and PCB layout. As such, every design will exhibit slightly
different characteristics. Careful attention to detail in the
magnetics design and construction as well as the board layout
is a must.
For designs using current-sense resistors in the power
MOSFET’s source leg, note that the gate drive current is “seen”
by the sense resistor. In very low-power designs, this can
easily exceed the full load sense voltage causing severe noise
problems. Adding a fairly large-value gate resistor will help in
this case. Also, an RC current-sense filter becomes much
more important.
FOLDBACK CIRCUIT
Foldback current limiting is provided by Q3 and its associated
components. Under normal operating conditions, diode D6
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keeps C13 charged to VCC. Hence, Q3 is biased off. In the
event of a short circuit on any output, all winding voltages are
clamped low. This causes the voltage on C13 to drop to a level
set by divider R10 and R11. VCC is held at 8.6 V by the Si9120’s
start-up regulator. The current set by the value of R12 flows
through Q3 and R3, and causes the voltage on pin 4 to rise.
Since a peak threshold of 1.2 V is internally set on pin 4, the
voltage required across R5 to terminate a pulse is reduced by
an amount equal to the drop on R3.
ID = {1.2 - (IQ3)(R3)}/R5.
Thus as IQ3 increases, ID decreases.
See Figure 2a for foldback operating waveforms.
The foldback circuit will not perform correctly without the spike
blanking circuit. The leakage spike will peak charge C13 even
with a shorted load. However, the foldback function is
completely optional and all associated components can be
eliminated if a lower cost supply is desired.
TEST RESULTS
Data compiled on the test circuit appear in Table 1. Combined
line and load regulation measures "2.7%, well within a "5%
specification. Measured efficiency is 73.4% with no effort at
optimization. A detailed loss assessment could, no doubt, offer
some improvements. Pulse load tests show reasonable
transient response, and phase margin is measured at
60 degrees. For details on how to close the feedback loop,
refer to Vishay Siliconix application notes AN713 and AN707.
All data taken with dc input source to ensure stable readings.
TABLE 1. UNIVERSALĆINPUT SUPPLY
TEST DATA
Full Load:
VIN (dc)
Iin (mA)
+5 V
+12 V
–12 V
100 V
143.9
4.974
12.64
12.50
200 V
72.3
5.014
12.76
12.61
300 V
48.9
5.027
12.79
12.65
385 V
39.4
5.049
12.81
12.67
100 V
78.0
5.153
12.99
12.83
200 V
40.3
5.205
13.10
12.96
300 V
27.9
5.235
13.12
12.97
385 V
23.0
5.254
13.14
13.01
Half Load:
PKĆPK OUTPUT RIPPLE VOLTAGES
(SPIKES NOT INCLUDED)
5V
+12 V
–12 V
60 mV
45 mV
40 mV
Note: Worst case over full line-voltage range.
Document Number: 70581
AN708
Vishay Siliconix
a)
Short Circuit On 5 V Output
Q1 Drain Voltage (100 V/Div)
Voltage On U1, Pin 4 (Current Sense)
(0.5 V/Div)
NOTE: 0.8 V dc pedestal caused by
the foldback circuit.
b)
Q1 Drain Voltage (100 V/Div)
Q1 Gate Voltage (100 V/Div)
c)
Q1 Gate Drive (5 V/Div)
Voltage On U1, Pin 4 (Current Sense)
(0.5 V/Div)
FIGURE 2. Operating Waveforms (all photos full load, VIN = 150 V dc)
Measured efficiency at VIN = 300 VDC was 73.4%.
During testing, an input capacitor value of as little as 33 F
proved adequate versus the design value of 68 F. The
low-value capacitor produces an input ripple voltage of 30 V
pk-pk. Since the primary inductance is slightly lower than the
design maximum value, the circuit is still able to maintain
regulation with the higher input ripple voltage value. This is a
Document Number: 70581
good example of where trade-offs can be made during
development programs. By using the larger input capacitance
and primary inductance, the peak input current could be
reduced slightly, and a slight improvement in efficiency should
result. However, a larger input capacitance will decrease the
conduction angle of the input rectifiers, and consequently will
reduce the input power factor. The priorities of a particular
application will determine the optimal approach.
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AN708
Vishay Siliconix
CONCLUSION
100 k 200 k 300 k 13.0
12.0
11.0
10.0
APPENDIX A
The SMP4N60 was tested for ability to withstand repetitive
avalanche currents and for non-repetitive capability.
Inductance values of 12 H and 94 H were used. Repetitive
tests were run at 3 A, with 94 H at 25_C. Failure current was
measured at 25_C and 100_C. Results were as follows:
L = 94 H
L = 12 H
25_C
100_C
25_C
100_C
4.25 A
2.40 A
7.28 A
6.40 A
For the 11.1-W flyback supply presented here, the leakage
inductance is specified at 60 H maximum. The maximum
drain current is set to approximately 1.0 A. Therefore, based
on the above data, adequate margin is present to prevent
avalanche failure.
APPENDIX B
A number of performance parameters of the Si9120 can be
altered by setting Ibias to a value other that 15 A. At lower Ibias,
higher efficiency can be obtained. At higher Ibias, lower
propagation delays and a wider error amplifier bandwidth are
possible. Also, if a VCC supply other than 10 V is used, Rbias
should be something other than 390 k. Figure 3 below relates
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14.0
V CC (V)
The simple universal-input power supply design that has been
presented combines economy and performance which should
prove more than adequate for the majority of applications. The
overall cost of the supply should rival linear regulators of similar
power level if heatsink cost is considered. Good regulation has
been achieved while maintaining the 3750-V ac
input-to-output isolation mandated by VDE. The Si9120
eliminates the need for any external start-up circuitry. Also,
foldback current-limiting is demonstrated which requires no
feedback across the isolation boundary.
R=1k
750 k 500 k 400 k VCC to Rbias and typical Ibias. The equation given can be used
for points not on the graph.
9.0
8.0
5
15
25
35
45
55
65
75
85
mA
R bias V CC – 2.3 V – 484
Ibias
I bias
FIGURE 3. Relationship of VCC to Rbias an Typical Ibias
REFERENCES
1) Liu, K.H., “Effects of Leakage Inductance on the Cross
Regulation in a Discontinuous-Mode Flyback Converter,”
Proceedings, 1989 High Frequency Power Conference,
Naples, Florida.
2) Chryssis, G., “High Frequency Switching Power Supplies,”
McGraw Hill 1984.
3) Billings, K., “Switchmode Power Supply Handbook,”
McGraw Hill 1989.
4) McLyman, Col. W.T., “Transformer and Inductor Design
Handbook,” McGraw Hill 1988.
Document Number: 70581
AN708
Vishay Siliconix
C1 . . . . . . . . . . . . . 2200 mF, 6.3 V Al. Electrolytic – United Chemicon SXC
C2 . . . . . . . . . . . . . 1000 mF, 6.3 V Al. Electrolytic – United Chemicon SXC
C3, C10 . . . . . . . . . 0.1 mF, 50 V Ceramic, Vishay Vitramon VJ1206Y104KXAAT
C4, C21 . . . . . . . . . 2200 mF, 16 V Al. Electrolytic – United Chemicon SXC
C5, C22 . . . . . . . . . 0.47 mF, 50 V Ceramic, Vishay Vitramon VJ1210Y474KXAAT
C6, C12 . . . . . . . . . 1000 pF, 100 V Ceramic. Vishay Vitramon VJ1206Y102KXBAT
C7 . . . . . . . . . . . . . 33 mF, 450 V Al Electrolytic (400 V ok)
C9 . . . . . . . . . . . . . 220 pF, 100 V Ceramic, Vishay Vitramon VJ1206A221KXBAT
C11, C13 . . . . . . . . 4700 pF, 100 V Ceramic, Vishay Vitramon VJ1206Y472KXBAT
C14 . . . . . . . . . . . . 1 mF, 50 V Ceramic, Vishay Vitramon VJ1812Y105KXAAT
C16 . . . . . . . . . . . . 75 pF, 500 V Ceramic or Mica, Vishay Vitramon VJ1206A750KXEAT
C17, C18 . . . . . . . . 0.1 mF, 250 V,ac VDE Class X2 Wima MKS 4-R, Vishay Roederstein F17724102000
C8, C19, C20 . . . . 0.0047 mF, 250 V,ac VDE Class Y Wima MP3-Y, Vishay Roederstein F17724102000
D1, D7 . . . . . . . . . . MUR 110 Motorola 1 A 100 V
D2 . . . . . . . . . . . . . 1N5822 3 A, 40 V Schottky
D3 . . . . . . . . . . . . . Bridge 1 A, 600 V DB105
D4, D5, D6 . . . . . . 1N4148
L1 . . . . . . . . . . . . . . Common mode choke Renco 1361-2
L2 . . . . . . . . . . . . . . Inductor 6 mH, 1.5 A, Vishay Dale ILS-1206-6 mH 10%
Q1 . . . . . . . . . . . . . FET N-channel SMP4N60 Vishay Siliconix
Q2 . . . . . . . . . . . . . FET N-channel 2N7000 Vishay Siliconix
Q3 . . . . . . . . . . . . . 2N4403 PNP (or 2N2907)
R1, R4 . . . . . . . . . . 10 W, 1/8 Carbon Film or Metal Film, Vishay Dale TNPW120610R0FT2
R2 . . . . . . . . . . . . . 175 kW, 1/8 W Carbon Film or Metal Film, Vishay Dale TNPW12061753FT2
R3 . . . . . . . . . . . . . 1 kW, 1/8 W Carbon Film or Metal Film, Vishay Dale TNPW12061001FT2
R5 . . . . . . . . . . . . . 1.3 W, 1/4 W Metal Film
R6 . . . . . . . . . . . . . 20 W, 1/2 W Metal Film
R7 . . . . . . . . . . . . . 1.2 kW, 1/8 W Metal Film, Vishay Dale TNPW12061201FT2
R8 . . . . . . . . . . . . . 680 W, 1/8 W Metal Film, Vishay Dale TNPW12066800FT2
R9 . . . . . . . . . . . . . 2 kW, 1/8 W Metal Film, Vishay Dale TNPW12062001FT2
R10 . . . . . . . . . . . . 130 kW, 1/8 W Metal Film, Vishay Dale TNPW12061303FT2
R11 . . . . . . . . . . . . . 68 kW, 1/8 W Metal Film, Vishay Dale TNPW12066802FT2
R12 . . . . . . . . . . . . 8.2 kW, 1/8 W Metal Film, Vishay Dale TNPW12068201FT2
R13 . . . . . . . . . . . . 75 kW, 1/8 W 1% Metal Film, Vishay Dale TNPW12067502FT2
R14 . . . . . . . . . . . . 390 kW, 1/8 W Metal Film, Vishay Dale TNPW12063903FT2
R15 . . . . . . . . . . . . 40.2 kW, 1/8 W 1% Metal Film, Vishay Dale TNPW12064022FT2
R16 . . . . . . . . . . . . 330 W, 1/2 W 5% Carbon Composition
T1 . . . . . . . . . . . . . . Schott Corp. #67122700
Document Number: 70581
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