TI TPS65135

TPS65135
SLVS704A – NOVEMBER 2011 – REVISED NOVEMBER 2011
www.ti.com
Single-Inductor, Multiple-Output (SIMO) Regulator
Check for Samples: TPS65135
FEATURES
1
•
•
•
•
•
•
•
2
•
•
•
•
•
•
•
SIMO Regulator Technology
2.5 V to 5.5 V Input Voltage Range
750 mW Output Power at VIN = 2.9 V
Positive Output Voltage up to 6 V
Negative Output Voltage Down to –7 V
1% Output Voltage Accuracy
Output current missmatch of pos and neg rail
up to 50%
Excellent Line Regulation
Advanced Power-Save Mode for Light-Load
Efficiency
Low-Noise Operation
Out-of-Audio Mode
Short-Circuit Protection
Thermal Shutdown
3-mm × 3-mm Thin QFN Package
APPLICATIONS
•
•
•
Active-Matrix OLED Power Supply
LCD Power Supply
General dual power supply applications
DESCRIPTION
The TPS65135 is a high efficient single inductor dual output converter. Due to its single-inductor multiple-output
(SIMO) technology the converter uses a minimum of external components. The device operates with a
buck-boost topology and generates a positive and a negative output voltage above or below the input voltage
rail. The SIMO technology enables excellent line and load regulation which is for instance required to avoid
disturbance of a mobile phone display as a result of input voltage variations that occur during transmit periods in
mobile communication systems. The device can also be used as a standard +/- supply as long as the output
current missmatch between the rails is smaller than 50%.
TYPICAL APPLICATION
L1
2.2 mH
TPS65135
16
15
VIN
2.5 V to 5.5 V
1
C1
10 mF
8
4
11
C4
100 nF
12
5
L1
L2
L1
L2
VIN
OUTP
EN
OUTP
VAUX
FB
PGND
FBG
PGND
OUTN
GND
OUTN
14
13
10
9
R1
365 kW
7
6
VPOS
5 V / 80 mA
C2
4.7 mF
R2
120 kW
3
2
R3
487 kW
C3
4.7 mF
VNEG
–5 V / 80 mA
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2011, Texas Instruments Incorporated
TPS65135
SLVS704A – NOVEMBER 2011 – REVISED NOVEMBER 2011
www.ti.com
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
ORDERING INFORMATION (1)
(1)
TA
ORDERING P/N
PACKAGE
PACKAGE MARKING
–40°C to 85°C
TPS65135RTE
RTE
CCR
The RTE package is available in tape and reel. Add R suffix (TPS65135RTER) to order quantities of 3000 parts per reel. For the most
current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI Web site at
www.ti.com.
16-PIN TQFN PACKAGE
OUTN
3
VAUX
4
L1
L2
L2
13
12 PGND
11 PGND
Exposed
Thermal Die
10 OUTP
9
5
6
7
8
EN
2
14
FB
OUTN
15
FBG
1
16
GND
VIN
L1
RTE Package
(Top View)
OUTP
P0081-01
PIN FUNCTIONS
PIN
NAME
NO.
I/O
DESCRIPTION
EN
8
I
Input pin to enable the device. Pulling this pin high enables the device. This pin has an
internal 500 kΩ pulldown resistor.
FB
7
I
Feedback regulation input for the positive output voltage rail
FBG
6
I
Feedback regulation input for GND reference (regulation of the negative output voltage rail)
Analog ground
5
–
L1
15, 16
I/O
Inductor terminal
L2
13, 14
I/O
Inductor terminal
2, 3
O
Negative output
OUTP
9, 10
O
Positive output
PGND
11, 12
–
Power GND
VAUX
4
I/O
VIN
1
GND
OUTN
Exposed thermal die
Reference voltage output. This pin requires a 100-nF capacitor for stability.
I
Input supply
–
Connect this pad to analog GND.
2
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FUNCTIONAL BLOCK DIAGRAM
L1
L1
15
L2
16
L2
13
14
OVP
VIN
OUTP
VAUX
M4
1
Vpos
9
M1
OUTP
VAUX
10
M2
FB
Gate
Drive
EN
8
7
Bias (1.2V)
UVLO
Thermal
Shutdown
FBG
6
OUTN
PGND
GND
2
5
M3
Vneg
3
VAUX
4
VAUX
Regulator
Current Limit
OUTP
+
VoltageControlled
Oscillator
VCO
OUTN
Current
Sense/
Soft Start
Ipeak
–
–
+
PWM/PFM
Control
Vref
–
+
Gate Drive
Out-of-Audio
Mode
20 kHz
OUTP
OUTP
OUTN
Short-Circuit
Protection
OUTN
11
PGND
12
PGND
3
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ABSOLUTE MAXIMUM RATINGS (1) (2)
over operating free-air temperature range (unless otherwise noted)
VALUE
VIN, EN, VAUX, FB, OUTP, L2
Voltage range
MAX
–0.3
7
V
–8
7
V
–0.3
0.3
V
2
kV
L1, OUTN
FBG
Human Body Model
ESD rating
Machine Model
Charged Device Model
Continuous total power dissipation
UNIT
MIN
200
V
1
kV
See Dissipation Ratings Table
Operating junction temperature range, TJ
–40
150
°C
Operating ambient temperature range, TA
–40
85
°C
Storage temperature range, Tstg
–65
150
°C
(1)
(2)
Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute–maximum–rated conditions for extended periods may affect device reliability.
All voltage values are with respect to ground.
THERMAL INFORMATION
THERMAL METRIC
(1)
VALUE
θJA
Junction-to-ambient thermal resistance
44.8
θJCtop
Junction-to-case (top) thermal resistance
42.0
θJCbot
Junction-to-case (bottom) thermal resistance
4.3
θJB
Junction-to-board thermal resistance
16.9
ψJT
Junction-to-top characterization parameter
0.4
ψJB
Junction-to-board characterization parameter
16.8
(1)
UNIT
°C/W
For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.
RECOMMENDED OPERATING CONDITIONS
MIN
VIN
Input voltage range
TYP
2.5
(1)
L
Inductor
1
2.2
CIN
Input Capacitor (1)
4.7
10
COUTP,
COUTN
Output Capacitors (1)
4.7
10
TA
Operating ambient temperature
TJ
Operating junction temperature
(1)
MAX
UNIT
5.5
V
4.7
µH
µF
20
µF
–40
85
°C
–40
125
°C
Please refer to DETAILED DESCRIPTION for further information
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ELECTRICAL CHARACTERISTICS
VIN = 3.7 V, EN = VIN, OUTP = 5 V, OUTN = –5 V, TA = –40°C to 85°C; typical values are at TA = 25°C
(unless otherwise noted).
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
SUPPLY CURRENT
VIN
Input voltage range
IQ
Operating quiescent current into VIN
ISD
Shutdown current into VIN
VUVLO
Undervoltage lockout threshold
2.5
5.5
7
EN = low
0.1
2
VIN falling
1.8
2.1
VIN rising
2
2.3
Thermal shutdown
Thermal shutdown hysteresis
V
mA
µA
V
140
°C
5
°C
ENABLE
VH
Logic high-level voltage
VIN = 2.5 V to 5.5 V
VL
Logic low-level voltage
VIN = 2.5 V to 5.5 V
REN
Enable pulldown resistor
1.2
200
V
500
0.4
V
900
kΩ
6
V
OUTPUT
VOUTP
Positive output voltage range
OVPP
Positive overvoltage protection
VOUTN
Negative output voltage range
OVPN
Negative overvoltage protection
Imis
Output current mismatch Ipos to Ineg (1)
VOUTP
Positive output voltage regulation
VFBG
Feedback ground regulation
rDS(on)
3
IOUT = 10 mA
6.1
7
–2.5
IOUT = 10 mA
-7.1
V
–7
–7.6
–50%
V
V
50%
-1 %
1.24
+1 %
–10
0
10
M1 MOSFET on-resistance
IL1 = 100 mA
250
M2 MOSFET on-resistance
IL2 = 100 mA
200
M3 MOSFET on-resistance
IL1 = 100 mA
500
M4 MOSFET on-resistance
IL2 = 100 mA
V
mV
mΩ
300
VIN = 3.7 V
0.9
1.2
1.6
VIN = 2.5 V
1
1.5
1.9
ISW
Switch current limit (M2)
POUT
Output power
Vpos – Vneg ≤ 10 V; VIN = 2.9 V
fs
Switching frequency
IOUT neg = IOUT pos = 30 mA
1
MHz
Line regulation positive output OUTP
VIN = 2.5V to 5.5V, IOUTN = IOUTP = 5
mA
0
%/V
Line regulation negative output OUTN
VIN = 2.5V to 5.5V, IOUTN = IOUTP = 5
mA
0
%/V
Load regulation positive output OUTP
IOUTN = IOUTP = 0 mA to 80 mA
0
%/A
Load regulation negative output OUTN
IOUTN = IOUTP = 0 mA to 80 mA
0
%/A
(1)
750
A
mW
See TYPICAL CHARACTERISTICS and DETAILED DESCRIPTION for more detail
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Application for Typical Characteristics Measurements
L1
TPS65135
16
15
1
VIN
C1
8
4
11
C4
12
5
L1
L2
L1
L2
VIN
EN
OUTP
OUTP
VAUX
FB
PGND
FBG
PGND
OUTN
GND
OUTN
14
13
10
VPOS
9
R1
C2
7
6
R2
3
2
C3
R3
VNEG
Figure 1. Application Circuit for Typical Characteristics Measurements
Table 1. Component List for Typical Characteristics Circuit
Reference
Description
C1, C2, C3
10 μF, 6.3 V, 0603, X5R, ceramic
Murata, GRM188R60J106ME84D
Manufacturer and Part Number
C4
100 nF, 10 V, 0603, X7R, ceramic
Murata, GRM188R71H104KA93D
L1
2.2 μH, 2.2 A, 90 mΩ, 2.5 mm * 2.0 mm * 1.2 mm
Toko, DFE252012C
R1
Depending on the output voltage, 1%, (all measurements with +/-5 V output voltage uses 365 kΩ)
R2
Depending on the output voltage, 1%, (all measurements with +/-5 V output voltage uses 120 kΩ)
R3
Depending on the output voltage, 1%, (all measurements with +/-5 V output voltage uses 487 kΩ)
U1
TPS65135RTE
Texas Instruments
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TYPICAL CHARACTERISTICS
TABLE OF GRAPHS
VPOS= 5 V, VNEG= -5 V, unless otherwise noted
Figure
Efficiency
vs. load current with TOKO DFE252012C 2.2 µH
inductor
Figure 2
Efficiency
vs. load current with TOKO DFE252012C 4.7 µH
inductor
Figure 3
Typical Output Current Window (Mismatch)
IOUTP vs. IOUTN
Figure 4
Operation at light load current (10mA)
DCM operation
Figure 5
Operation at high load current (80mA)
CCM operation
Figure 6
Line transient response
IOUT = 10 mA
Figure 7
Line transient response
IOUT = 50 mA
Figure 8
Start-up
VIN rising, no load
Figure 9
Start-up
EN = high, no load
Figure 10
Shut-down
VIN falling, no load
Figure 11
Shut-down
EN = low, no load
Figure 12
Transient Response
IOUT = 10mA to 50mA (load between VNEG and VPOS)
Figure 13
Transient Response
IOUT = 20mA to 80mA (load between VNEG and VPOS)
Figure 14
Switching frequency
vs. load current
Figure 15
Quiescent current
vs. input voltage
Figure 16
EFFICIENCY
vs.
LOAD CURRENT (TOKO DFE252012C, 4.7 µH)
80
80
70
70
60
60
Efficiency (%)
Efficiency (%)
EFFICIENCY
vs
LOAD CURRENT (TOKO DFE252012C, 2.2 µH)
50
40
30
20
L = 2.2 µH
VPOS = 5 V
VNEG = −5 V
10
0
VIN = 2.5 V
VIN = 3.0 V
VIN = 3.7 V
VIN = 4.5 V
1
10
Current (mA)
50
40
30
20
10
100
0
VIN = 2.5 V
VIN = 3.0 V
VIN = 3.7 V
VIN = 4.5 V
L = 4.7 µH
VPOS = 5 V
VNEG = −5 V
1
G000
Figure 2.
10
Current (mA)
100
G000
Figure 3.
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OUTPUT CURRENT MISMATCH
Positive Output Current
vs.
Negative Output Current
Not allowed
Area
90
L1
VIN = 3.7 V
Vpos = 5 V
Vneg = −5 V
L = TOKO DFE252012C, 2.2µH
80
Vneg current (mA)
OPERATION AT LIGHT LOAD CURRENT, DCM
OPERATION
70
60
VPOS
50
VNEG
40
30
Not allowed
Area
10
0
VIN = 3.7V
VPOS = 5V
VNEG = -5V
IOUT = 10mA
IInductor
20
0
10
20
30
40
50
60
70
Vpos current (mA)
80
90
100
Figure 4.
Figure 5.
OPERATION AT HIGH LOAD CURRENT, CCM
OPERATION
LINE TRANSIENT RESPONSE
IOUT = 10 mA
VIN
L1
VPOS
VPOS
VIN = 2.9V → 3.4V
VPOS = 5V
VNEG = -5V
IOUT = 10mA
VNEG
VNEG
VIN = 3.7V
VPOS = 5V
VNEG = -5V
IOUT = 80mA
IInductor
Figure 6.
Figure 7.
LINE TRANSIENT RESPONSE
IOUT = 50 mA
START UP VIN RISING
VIN
VPOS
VIN = 2.9V → 3.4V
VPOS = 5V
VNEG = -5V
IOUT = 50mA
VIN
VNEG
VPOS
VIN = 3.7V
VPOS = 5V
VNEG = -5V
IOUT = 0mA
VNEG
IIN
Figure 8.
Figure 9.
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START UP EN = HIGH
SHUTDOWN VIN FALLING
EN
VIN
VNEG
VPOS
VPOS
VNEG
VIN = 3.7V
VPOS = 5V
VNEG = -5V
IOUT = 0mA
IIN
VIN = 3.7V
VPOS = 5V
VNEG = -5V
IOUT = 0mA
IIN
Figure 10.
Figure 11.
SHUTDOWN EN = LOW
TRANSIENT RESPONSE IOUT = 10mA to 50mA
EN
VNEG
VPOS
VPOS
VNEG
VIN = 3.7V
VPOS = 5V
VNEG = -5V
IOUT = 0mA
IIN
VIN = 3.7V
VPOS = 5V
VNEG = -5V
IOUT
IOUT_POS to NEG = 10mA → 50mA
Figure 12.
Figure 13.
TRANSIENT RESPONSE IOUT = 20mA to 80mA
SWITCHING FREQUENCY
vs
LOAD CURRENT
VNEG
VPOS
IOUT
IOUT_POS to NEG
VIN = 3.7V
VPOS = 5V
VNEG = -5V
= 20mA → 80mA
Switching Frequency (MHz)
2
L = 2.2 µH
VPOS = 5 V
VNEG = −5 V
1.5
1
VIN = 2.5 V
VIN = 3.1 V
VIN = 3.7 V
VIN = 4.6 V
0.5
0
0
Figure 14.
10
20
30
40
50
60
70
Output Current (mA)
80
90
100
G000
Figure 15.
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QUIESCENT CURRENT
vs
INPUT VOLTAGE
30
IOUT = 0mA
IOUT = 1mA
IOUT = 2mA
IOUT = 3mA
IOUT = 4mA
IOUT = 5mA
Input Current (mA)
25
20
15
10
5
0
2.5
L = 2.2 µH
VPOS = 5 V
VNEG = −5 V
3
3.5
4
4.5
Input Voltage (V)
5
5.5
G000
Figure 16.
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DETAILED DESCRIPTION
The TPS65135 operates with a four-switch buck-boost converter topology, generating a negative and a positive
output voltage with a single inductor. The device uses the SIMO regulator technology featuring best-in-class
line-transient regulation, buck-boost mode for the positive and negative outputs, and highest efficiency over the
entire load-current range. High efficiency over the entire load-current range is implemented by reducing the
converter switching frequency. Out-of-audio mode avoids the switching frequency going below 20 kHz.
As illustrated in the FUNCTIONAL BLOCK DIAGRAM, the converter operates with two control loops. One error
amplifier sets the output voltage for the positive output OUTP. The ground error amplifier regulates FBG to
typically 0 V. Using the external feedback divider allows setting both output voltages, OUTP and OUTN. In
principle the converter topology operates just like any other buck-boost converter topology with the difference
that the output voltage across the inductor is the sum of the positive and negative output voltage. With this
consideration all calculations of the buck-boost converter apply for this topology as well. During the first switch
cycle M1 and M2 are closed, connecting the inductor from VIN to GND. During the second switch cycle the
inductor discharges to the positive and negative outputs by closing switches M4 and M3. Because the inductor is
discharged to both of the outputs simultaneously, the output voltages can be higher or lower than the input
voltage. Because of this the converter operates best when the current out of OUTP is equal to the current flowing
into OUTN. This is for example the case when driving an AMOLED panel. Asymmetries in load current can be
canceled out by the used topology. However this is only possible for current asymmetries of up to 50%. During
light load the converter operates in discontinuous conduction mode. The converter operates in
peak-current-mode control with the switching cycle given by an internal voltage-controlled oscillator (VCO). As
the load current increases the converter operates in continuous-conduction mode. In this mode, the converter
moves to peak-current control with the switch cycle given by the fixed off-time. The SIMO regulator topology has
excellent line transient regulation when operating in discontinuous conduction mode. As the load current
increases, entering continuous conduction mode, the line transient performance is linearly decreased.
Advanced Power-Save Mode for Light-Load Efficiency
In order to maintain high efficiency over the entire load-current range, the converter reduces its switching
frequency as the load current decreases. The advanced power-save mode controls the switching frequency
using a voltage-controlled oscillator (VCO). The VCO frequency is proportional to the inductor peak current, with
a lower frequency limit of 20 kHz in typical applications the frequency does not go below 100 kHz. This avoids
disturbance of the audio band and minimizes audible noise coming from the ceramic input and output capacitors.
By maintaining a controlled switching frequency, possible EMI is minimized. This is especially important when
using the device in mobile phones. See Figure 15 for typical switching frequency versus load current. For zero
load an internal shunt regulator ensures stable output voltage regulation.
Buck-Boost Mode Operation
Buck-boost mode operation allows the input voltage to be higher or lower than the output voltage. This mode
allows the use of batteries and supply voltages that are above the output voltage of OUTP.
Inherent Excellent Line-Transient Regulation
The SIMO regulator achieves inherent superior line-transient regulation when operating in discontinuous
conduction mode, shown in Figure 7 and Figure 8. In discontinuous conduction mode the current delivered to the
output is given by the inductor peak current and falling slope of the inductor current. This is shown in Figure 17,
where the output current, given by the area A, is the same for different input voltages. Because the converter
uses peak-current-mode control, the peak current is fixed as long as the load current is fixed. The falling slope of
the inductor current is given by the sum of the output voltage and inductor value. This is also a fixed value and
independent of the input voltage. Because of this, any change in input voltage changes the converter duty cycle
but does not change the inductor peak current or the falling slope of the inductor current. Therefore the output
current, given by the area A (Figure 17), remains constant over any input voltage variation. Because the area A
is constant, the converter has an inherently perfect line regulation when operating in discontinuous conduction
mode. Entering continuous conduction mode (CCM) linearly decreases the line-transient performance. However
the line-transient response in CCM is still as good as for any standard current-mode-controlled switching
converter. The following formulas detail the relations of the TPS65135 converter topology operating in CCM.
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Vpos
Vin
L
Ip
Vneg
L
A
A
tclock
M0116-01
Figure 17. Inherently Perfect Line-Transient Regulation
The converter always sees the sum of the negative and positive output voltage, which is calculated as:
VO = VOUTP + VOUTN
(1)
The converter duty cycle is calculated using the efficiency estimation from the data sheet curves or from real
application measurements. A 70% efficiency value is a good value to go through the calculations.
VO
D=
h × VIN + VO
(2)
The output current for entering continuous conduction mode can be calculated. The switching frequency can be
obtained from the data sheet graphs. A frequency of 1.5 MHz is usually sufficient for these types of calculations.
2
IC =
VO × (1 - D )
fS × 2 × L
(3)
The inductor ripple current when operating in CCM can also be calculated.
V ×D
DIL = IN
L × fS
(4)
Last but not least, the converter switch peak current is calculated as follows.
I
1
IL _ peak = OUT + × DIL
1- D 2
(5)
Overvoltage Protection
The device monitors the positive and negative output voltage. The regulators monitor the outputs and reduce the
current limit when the output voltages exceed the overvoltage protection limit. They are clamped using a zener
diode, the positive output to typically 7V and the negative to –7.6 V.
Short-Circuit Protection
Both outputs are protected against short circuits either to GND or against the other output. The device switching
frequency and the current limit are reduced in case of a short circuit.
Soft-Start Operation
The device increases the current limit during soft-start operation to avoid high inrush currents during start-up.
The current limit typically ramps up to its full-current limit within 100 µs.
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Output-Current Mismatch
The device operates best when the current of the positive output is similar to the current of the negative output.
However the device is able to regulate an output-current mismatch between the outputs of up to 50% (See
Figure 4 for typically allowed currents, only 50% missmatch is guaranteed). If the output-current mismatch
becomes much larger one of the outputs goes out of regulation and finally the IC shuts down. In case of zero
load of one output the other output can support up to 5mA. The IC automatically recovers when the mismatch is
reduced. The below formula is used to calculate the maximum supported current missmatch.
Smaller IOUT
1£ 50%
Bigger IOUT
(6)
Input Capacitor Selection
The device typically requires a 10 µF ceramic input capacitor. Larger values can be used to lower the input
voltage ripple. Table 2 lists capacitors suitable for use on the TPS65135 input.
Table 2. Input Capacitor Selection
CAPACITOR
COMPONENT SUPPLIER
SIZE
10 µF / 6.3V
Murata GRM188R60J106ME84D
0603
10 µF / 6.3 V
Taiyo Yuden JMK107BJ106
0603
Inductor Selection/Efficiency/Line-Transient Response
The device is internally compensated and operates best with a 2.2 µH inductor. For this type of converter the
inductor selection is a key element in the design process because it has a big impact on several application
parameters. The inductor selection influences the converter efficiency a lot, also the line and load transient
response as well as the maximum output current. Because the inductor ripple current is fairly large in this type of
application, the inductor has a major impact on the overall converter efficiency. Having large inductor ripple
current causes the inductor core and magnetizing losses to become dominant. Due to this, an inductor with a
larger dc winding resistance can achieve higher converter efficiencies when having lower core and magnetizing
losses. The used inductance influences the line transient regulation, it influences the current range entering
continuous conduction mode (CCM). As discussed, the line transient performance decreases when entering
CCM. The larger the inductor value, the lower the load current when entering CCM. The formula to calculate the
current entering CCM is shown in Equation 3. The inductors listed in Table 3 achieve a good overall converter
efficiency while having a low device profile. The first two TOKO inductors achieve the highest efficiency (almoust
identical) followed by the LPS3008. The best compromize between efficiency and inductor size is given by the
XFL2006 inductor . The inductor saturation current should be 1A or higher, depending on the maximum output
current of the application it can also be lower. See Equation 5, where the converter switch current limit is
calculated. The converter switch current is equal to the peak inductor current.
Table 3. Inductor Selection
INDUCTOR VALUE
COMPONENT SUPPLIER
DIMENSIONS in mm
Isat / RDC
2.2 µH
2.2 µH
TOKO DFE252010C
2.5 x 2 x 1
1.9 A / 130 mΩ
TOKO DFE252012C
2.5 x 2 x 1.2
2.2 µH
2.2 A / 90 mΩ
Coilcraft XFL2006-222
2 × 1.9 × 0.6
0.8 A / 278 mΩ
2.2 µH
Coilcraft LPS3008-222
3 × 3 × 0.8
1.1 A / 175 mΩ
2.2 µH
Samsung CIG2MW2R2NNE
2 × 1.6 × 1
1.2 A / 110 mΩ
2.2 µH
TOKO FDSE0312-2R2
3.3 × 3.3 × 1.2
1.2 A / 160 mΩ
2.2 µH
ABCO LPF3010T-2R2
2.8 × 2.8 × 1
1.0 A / 100 mΩ
2.2 µH
Maruwa CXFU0208-2R2
2.65 × 2.65 × 0.8
0.85 A / 185 mΩ
13
Copyright © 2011, Texas Instruments Incorporated
Product Folder Link(s): TPS65135
TPS65135
SLVS704A – NOVEMBER 2011 – REVISED NOVEMBER 2011
www.ti.com
Output Capacitor Selection
A 4.7-µF output capacitor is generally sufficient for most applications, but larger values can be used as well for
improved load- and line-transient response at higher load currents. The capacitors of Table 4 is recommended
for use with the TPS65135.
Table 4. Output Capacitor Selection
CAPACITOR
COMPONENT SUPPLIER
SIZE
10 µF / 6.3V
Murata GRM188R60J106ME84D
0603
4.7 µF / 10V
Taiyo Yuden LMK107BJ475
0603
10 µF / 6.3 V
Taiyo Yuden JMK107BJ106
0603
SPACER
Setting the Output Voltages OUTP and OUTN
The feedback divider R1, R2, R3 sets the positive and negative output voltage. The device regulates the feeback
voltage FB to typically 1.24 V and the feedback FBG to typically 0V. R2 is selected to have at least 10 µA
through the feedback divider.
1.24V
R2 =
» 120kW
10mA
(7)
The positive output voltage and R1 are calculated as:
VPOS = 1.24V ×
R1 + R2
R2
(8)
æ V
ö
R1 = R2 × ç POS - 1÷
è 1.24V
ø
(9)
The negative output voltage is calculated as:
VNEG = - (VFB + VFBG ) ×
R3
R2
(10)
Since VFBG is typically regulated to 0 V, the formula can be simplified and R3 is then calculated as:
R3 =
VNEG
1.24V
× R2
(11)
PCB Layout Guidelines
PCB layout is an important task in the power supply design. Good PCB layout minimizes EMI and allows very
good output voltage regulation. For the TPS65135 the following PCB layout guidelines are recommended.
Place the power components first. The inductor and the input and output capacitors must be as close as possible
to the IC pins. Place the bypass capacitor for the reference output voltage VAUX as close as possible to pin 4.
Use bold and wide traces for power traces connecting the inductor and input and output capacitors. Use a
common ground plane or a start ground connection.
See the TPS65135EVM-063 user's guide (SLVU244) and evaluation module for a PCB layout example.
14
Copyright © 2011, Texas Instruments Incorporated
Product Folder Link(s): TPS65135
TPS65135
SLVS704A – NOVEMBER 2011 – REVISED NOVEMBER 2011
www.ti.com
TYPICAL APPLICATION
L1
2.2 mH
TPS65135
16
15
VIN
2.5 V to 5.5 V
1
C1
10 mF
8
4
11
C4
100 nF
12
5
L1
L2
L1
L2
VIN
OUTP
EN
OUTP
VAUX
FB
PGND
FBG
PGND
OUTN
GND
OUTN
14
13
10
9
R1
365 kW
7
6
VPOS
5 V / 80 mA
C2
4.7 mF
R2
120 kW
3
2
R3
487 kW
C3
4.7 mF
VNEG
–5 V / 80 mA
Figure 18. Standard Application +/- 5 V Supply
SPACER
REVISION HISTORY
Changes from Original (November 2011) to Revision A
•
Page
Changed the UVLO threshould max value for VIN falling From: 2 V To 2.1 V ...................................................................... 5
15
Copyright © 2011, Texas Instruments Incorporated
Product Folder Link(s): TPS65135
PACKAGE MATERIALS INFORMATION
www.ti.com
14-Jul-2012
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
TPS65135RTER
Package Package Pins
Type Drawing
WQFN
RTE
16
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
3000
330.0
12.4
Pack Materials-Page 1
3.3
B0
(mm)
K0
(mm)
P1
(mm)
3.3
1.1
8.0
W
Pin1
(mm) Quadrant
12.0
Q2
PACKAGE MATERIALS INFORMATION
www.ti.com
14-Jul-2012
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
TPS65135RTER
WQFN
RTE
16
3000
367.0
367.0
35.0
Pack Materials-Page 2
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