NSC LM2735YMF

LM2735
520kHz/1.6MHz – Space-Efficient Boost and SEPIC DC-DC
Regulator
General Description
Features
The LM2735 is an easy-to-use, space-efficient 2.1A low-side
switch regulator ideal for Boost and SEPIC DC-DC regulation.
It provides all the active functions to provide local DC/DC
conversion with fast-transient response and accurate regulation in the smallest PCB area. Switching frequency is internally set to either 520kHz or 1.6MHz, allowing the use of
extremely small surface mount inductor and chip capacitors
while providing efficiencies up to 90%. Current-mode control
and internal compensation provide ease-of-use, minimal
component count, and high-performance regulation over a
wide range of operating conditions. External shutdown features an ultra-low standby current of 80 nA ideal for portable
applications. Tiny SOT23-5, LLP-6, and eMSOP-8 packages
provide space-savings. Additional features include internal
soft-start, circuitry to reduce inrush current, pulse-by-pulse
current limit, and thermal shutdown.
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Input voltage range 2.7V to 5.5V
Output voltage range 3V to 24V
2.1A switch current over full temperature range
Current-Mode control
Logic high enable pin
Ultra low standby current of 80 nA in shutdown
170 mΩ NMOS switch
±2% feedback voltage accuracy
Ease-of-use, small total solution size
Internal soft-start
Internal compensation
Two switching frequencies
520 kHz (LM2735-Y)
1.6 MHz (LM2735-X)
Uses small surface mount inductors and chip capacitors
Tiny SOT23-5, LLP-6, and eMSOP-8 packages
Applications
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LCD Display Backlighting For Portable Applications
OLED Panel Power Supply
USB Powered Devices
Digital Still and Video Cameras
White LED Current Source
Automotive
Typical Boost Application Circuit
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Efficiency vs Load Current VO = 12V
© 2008 National Semiconductor Corporation
202158
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LM2735 520kHz/1.6MHz – Space-Efficient Boost and SEPIC DC-DC Regulator
September 10, 2008
LM2735
Connection Diagrams
Top View
Top View
Top View
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5-Pin SOT23
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6-Pin LLP
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8-Pin eMSOP
Ordering Information
Order Number
Description
LM2735YMF
LM2735YMFX
LM2735YSD
LM2735YSDX
520kHz
LM2735YMY
LM2735YMYX
Package Type
Package
Drawing
SOT23-5
MF05A
LLP-6
SDE06A
eMSOP-8
MUY08A
LM2735XMF
Supplied As
1000 units tape & reel
3000 units tape & reel
1000 units tape & reel
4500 units tape & reel
1000 units tape & reel
3500 units tape & reel
1000 units tape & reel
LM2735XMFX
3000 units tape & reel
SOT23-5
LM2735XQMF
MF05A
LM2735XQMFX
1000 units tape & reel
3000 units tape & reel
1.6MHz
LM2735XSD
LM2735XSDX
LM2735XMY
LM2735XMYX
Features
LLP-6
SDE06A
eMSOP-8
MUY08A
AEC-Q100 Grade 1
qualified. Automotive
Grade Production Flow*
1000 units tape & reel
4500 units tape & reel
1000 units tape & reel
3500 units tape & reel
*Automotive Grade (Q) product incorporates enhanced manufacturing and support processes for the automotive market, including defect detection methodologies.
Reliability qualification is compliant with the requirements and temperature grades defined in the AEC-Q100 standard. Automotive grade products are identified
with the letter Q. For more information go to http://www.national.com/automotive.
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2
LM2735
Pin Descriptions - 5-Pin SOT23
Pin
Name
1
SW
Function
2
GND
3
FB
Feedback pin. Connect FB to external resistor divider to set output voltage.
4
EN
Shutdown control input. Logic high enables operation. Do not allow this pin to float or be greater than VIN +
0.3V.
5
VIN
Supply voltage for power stage, and input supply voltage.
Output switch. Connect to the inductor, output diode.
Signal and power ground pin. Place the bottom resistor of the feedback network as close as possible to this
pin.
Pin Descriptions - 6-Pin LLP
Pin
Name
Function
1
PGND
Power ground pin. Place PGND and output capacitor GND close together.
2
VIN
Supply voltage for power stage, and input supply voltage.
3
EN
Shutdown control input. Logic high enables operation. Do not allow this pin to float or be greater than VIN +
0.3V.
4
FB
Feedback pin. Connect FB to external resistor divider to set output voltage.
5
AGND
6
SW
DAP
GND
Signal ground pin. Place the bottom resistor of the feedback network as close as possible to this pin & pin
4.
Output switch. Connect to the inductor, output diode.
Signal & Power ground. Connect to pin 1 & pin 5 on top layer. Place 4-6 vias from DAP to bottom layer GND
plane.
Pin Descriptions - 8-Pin eMSOP
Pin
Name
1
Function
No Connect
2
PGND
3
VIN
Supply voltage for power stage, and input supply voltage.
4
EN
Shutdown control input. Logic high enables operation. Do not allow this pin to float or be greater than VIN +
0.3V.
5
FB
Feedback pin. Connect FB to external resistor divider to set output voltage.
6
AGND
7
SW
8
DAP
Power ground pin. Place PGND and output capacitor GND close together.
Signal ground pin. Place the bottom resistor of the feedback network as close as possible to this pin & pin 5
Output switch. Connect to the inductor, output diode.
No Connect
GND
Signal & Power ground. Connect to pin 2 & pin 6 on top layer. Place 4-6 vias from DAP to bottom layer GND
plane.
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LM2735
Soldering Information
Infrared/Convection Reflow (15sec)
Absolute Maximum Ratings (Note 1)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
VIN
SW Voltage
FB Voltage
EN Voltage
ESD Susceptibility (Note 4)
Junction Temperature (Note 2)
Storage Temp. Range
Operating Ratings
(Note 1)
VIN
VSW
VEN (Note 5)
Junction Temperature Range
Power Dissipation
(Internal) SOT23-5
-0.5V to 7V
-0.5V to 26.5V
-0.5V to 3.0V
-0.5V to 7.0V
2kV
150°C
-65°C to 150°C
220°C
2.7V to 5.5V
3V to 24V
0V to VIN
−40°C to +125°C
400 mW
Electrical Characteristics
Limits in standard type are for TJ = 25°C only; limits in boldface type apply over the
junction temperature range of (TJ = -40°C to 125°C). Minimum and Maximum limits are guaranteed through test, design, or statistical
correlation. Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only.
VIN = 5V unless otherwise indicated under the Conditions column.
Symbol
VFB
ΔVFB/VIN
Parameter
Feedback Voltage
Feedback Voltage Line Regulation
IFB
Feedback Input Bias Current
FSW
Switching Frequency
DMAX
Maximum Duty Cycle
DMIN
Minimum Duty Cycle
RDS(ON)
Switch On Resistance
ICL
Switch Current Limit
SS
Soft Start
IQ
UVLO
Quiescent Current (switching)
Conditions
Min
Typ
Max
−40°C ≤ to TJ ≤ +125°C (SOT23-5)
1.230 1.255
1.280
0°C ≤ to TJ ≤ +125°C (SOT23-5)
1.236 1.255
1.274
−40°C ≤ to TJ ≤ +125°C (LLP-6)
1.225 1.255
1.285
−0°C ≤ to TJ ≤ +125°C (LLP-6)
1.229 1.255
1.281
−40°C ≤ to TJ ≤ +125°C (eMSOP-8)
1.220 1.255
1.290
0°C ≤ to TJ ≤ +125°C (eMSOP-8)
1.230 1.255
1.280
VIN = 2.7V to 5.5V
0.06
%/V
1
µA
kHz
1200
1600
2000
LM2735-Y
360
520
680
LM2735-X
88
96
LM2735-Y
91
99
LM2735-X
5
LM2735-Y
2
%
%
SOT23-5 and eMSOP-8
170
330
LLP-6
190
350
3
4
ms
7.0
11
LM2735-Y
3.4
7
Quiescent Current (shutdown)
All Options VEN = 0V
80
Undervoltage Lockout
VIN Rising
2.3
1.7
mΩ
A
LM2735-X
VIN Falling
V
0.1
LM2735-X
2.1
Units
mA
nA
2.65
V
1.9
Shutdown Threshold Voltage
(Note 5)
Enable Threshold Voltage
(Note 5)
I-SW
Switch Leakage
VSW = 24V
1.0
µA
I-EN
Enable Pin Current
Sink/Source
100
nA
VEN_TH
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4
0.4
1.8
V
Parameter
Conditions
Min
Typ
LLP-6 and eMSOP-8 Package
80
SOT23-5 Package
118
LLP-6 and eMSOP-8 Package
18
SOT23-5 Package
60
θJA
Junction to Ambient
0 LFPM Air Flow (Note 3)
θJC
Junction to Case (Note 3)
TSD
Thermal Shutdown Temperature (Note 2)
160
Thermal Shutdown Hysteresis
10
Max
Units
°C/W
°C/W
°C
Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is
intended to be functional, but specific performance is not guaranteed. For guaranteed specifications and the test conditions, see Electrical Characteristics.
Note 2: Thermal shutdown will occur if the junction temperature exceeds the maximum junction temperature of the device
Note 3: Applies for packages soldered directly onto a 3” x 3” PC board with 2oz. copper on 4 layers in still air.
Note 4: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin.
Note 5: Do not allow this pin to float or be greater than VIN +0.3V.
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LM2735
Symbol
LM2735
Typical Performance Characteristics
Current Limit vs Temperature
FB Pin Voltage vs Temperature
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Oscillator Frequency vs Temperature - "X"
Oscillator Frequency vs Temperature - "Y"
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20215809
Typical Maximum Output Current vs VIN
RDSON vs Temperature
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LM2735Y Efficiency vs Load Current, Vo = 20V
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LM2735X Efficiency vs Load Current, Vo = 12V
LM2735Y Efficiency vs Load Current, Vo = 12V
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Output Voltage Load Regulation
Output Voltage Line Regulation
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LM2735
LM2735X Efficiency vs Load Current, Vo = 20V
LM2735
Simplified Internal Block Diagram
20215818
FIGURE 1. Simplified Block Diagram
at the falling edge of the reset pulse generated by the internal
oscillator. When this pulse goes low, the output control logic
turns on the internal NMOS control switch. During this ontime, the SW pin voltage (VSW) decreases to approximately
GND, and the inductor current (IL) increases with a linear
slope. IL is measured by the current sense amplifier, which
generates an output proportional to the switch current. The
sensed signal is summed with the regulator’s corrective ramp
and compared to the error amplifier’s output, which is proportional to the difference between the feedback voltage and
VREF. When the PWM comparator output goes high, the output switch turns off until the next switching cycle begins.
During the switch off-time, inductor current discharges
through diode D1, which forces the SW pin to swing to the
output voltage plus the forward voltage (VD) of the diode. The
regulator loop adjusts the duty cycle (D) to maintain a constant output voltage .
Application Information
THEORY OF OPERATION
The LM2735 is a constant frequency PWM boost regulator IC
that delivers a minimum of 2.1A peak switch current. The regulator has a preset switching frequency of either 520 kHz or
1.60 MHz. This high frequency allows the LM2735 to operate
with small surface mount capacitors and inductors, resulting
in a DC/DC converter that requires a minimum amount of
board space. The LM2735 is internally compensated, so it is
simple to use, and requires few external components. The
LM2735 uses current-mode control to regulate the output
voltage. The following operating description of the LM2735
will refer to the Simplified Block Diagram (Figure 1) the simplified schematic (Figure 2), and its associated waveforms
(Figure 3). The LM2735 supplies a regulated output voltage
by switching the internal NMOS control switch at constant
frequency and variable duty cycle. A switching cycle begins
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LM2735
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FIGURE 2. Simplified Schematic
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FIGURE 3. Typical Waveforms
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LM2735
CURRENT LIMIT
The LM2735 uses cycle-by-cycle current limiting to protect
the internal NMOS switch. It is important to note that this current limit will not protect the output from excessive current
during an output short circuit. The input supply is connected
to the output by the series connection of an inductor and a
diode. If a short circuit is placed on the output, excessive current can damage both the inductor and diode.
20215824
Design Guide
FIGURE 4. Inductor Current
ENABLE PIN / SHUTDOWN MODE
The LM2735 has a shutdown mode that is controlled by the
Enable pin (EN). When a logic low voltage is applied to EN,
the part is in shutdown mode and its quiescent current drops
to typically 80 nA. Switch leakage adds up to another 1 µA
from the input supply. The voltage at this pin should never
exceed VIN + 0.3V.
THERMAL SHUTDOWN
Thermal shutdown limits total power dissipation by turning off
the output switch when the IC junction temperature exceeds
160°C. After thermal shutdown occurs, the output switch
doesn’t turn on until the junction temperature drops to approximately 150°C.
A good design practice is to design the inductor to produce
10% to 30% ripple of maximum load. From the previous equations, the inductor value is then obtained.
SOFT-START
This function forces VOUT to increase at a controlled rate during start up. During soft-start, the error amplifier’s reference
voltage ramps to its nominal value of 1.255V in approximately
4.0ms. This forces the regulator output to ramp up in a more
linear and controlled fashion, which helps reduce inrush current.
Where: 1/TS = FSW = switching frequency
One must also ensure that the minimum current limit (2.1A)
is not exceeded, so the peak current in the inductor must be
calculated. The peak current (ILPK ) in the inductor is calculated by:
INDUCTOR SELECTION
The Duty Cycle (D) can be approximated quickly using the
ratio of output voltage (VO) to input voltage (VIN):
or
ILpk = IIN + ΔIL
ILpk = IOUT / D' + ΔIL
When selecting an inductor, make sure that it is capable of
supporting the peak input current without saturating. Inductor
saturation will result in a sudden reduction in inductance and
prevent the regulator from operating correctly. Because of the
speed of the internal current limit, the peak current of the inductor need only be specified for the required maximum input
current. For example, if the designed maximum input current
is 1.5A and the peak current is 1.75A, then the inductor should
be specified with a saturation current limit of >1.75A. There is
no need to specify the saturation or peak current of the inductor at the 3A typical switch current limit.
Because of the operating frequency of the LM2735, ferrite
based inductors are preferred to minimize core losses. This
presents little restriction since the variety of ferrite-based inductors is huge. Lastly, inductors with lower series resistance
(DCR) will provide better operating efficiency. For recommended inductors see Example Circuits.
Therefore:
Power losses due to the diode (D1) forward voltage drop, the
voltage drop across the internal NMOS switch, the voltage
drop across the inductor resistance (RDCR) and switching
losses must be included to calculate a more accurate duty
cycle (See Calculating Efficiency and Junction Temperature
for a detailed explanation). A more accurate formula for calculating the conversion ratio is:
INPUT CAPACITOR
An input capacitor is necessary to ensure that VIN does not
drop excessively during switching transients. The primary
specifications of the input capacitor are capacitance, voltage,
RMS current rating, and ESL (Equivalent Series Inductance).
The recommended input capacitance is 10 µF to 44 µF depending on the application. The capacitor manufacturer
specifically states the input voltage rating. Make sure to check
any recommended deratings and also verify if there is any
Where η equals the efficiency of the LM2735 application.
The inductor value determines the input ripple current. Lower
inductor values decrease the size of the inductor, but increase
the input ripple current. An increase in the inductor value will
decrease the input ripple current.
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LM2735
significant change in capacitance at the operating input voltage and the operating temperature. The ESL of an input
capacitor is usually determined by the effective cross sectional area of the current path. At the operating frequencies
of the LM2735, certain capacitors may have an ESL so large
that the resulting impedance (2πfL) will be higher than that
required to provide stable operation. As a result, surface
mount capacitors are strongly recommended. Multilayer ceramic capacitors (MLCC) are good choices for both input and
output capacitors and have very low ESL. For MLCCs it is
recommended to use X7R or X5R dielectrics. Consult capacitor manufacturer datasheet to see how rated capacitance
varies over operating conditions.
OUTPUT CAPACITOR
The LM2735 operates at frequencies allowing the use of ceramic output capacitors without compromising transient response. Ceramic capacitors allow higher inductor ripple
without significantly increasing output ripple. The output capacitor is selected based upon the desired output ripple and
transient response. The initial current of a load transient is
provided mainly by the output capacitor. The output
impedance will therefore determine the maximum voltage
perturbation. The output ripple of the converter is a function
of the capacitor’s reactance and its equivalent series resistance (ESR):
20215829
FIGURE 5. Setting Vout
A good value for R1 is 10kΩ.
COMPENSATION
The LM2735 uses constant frequency peak current mode
control. This mode of control allows for a simple external
compensation scheme that can be optimized for each application. A complicated mathematical analysis can be completed to fully explain the LM2735’s internal & external compensation, but for simplicity, a graphical approach with simple
equations will be used. Below is a Gain & Phase plot of a
LM2735 that produces a 12V output from a 5V input voltage.
The Bode plot shows the total loop Gain & Phase without external compensation.
When using MLCCs, the ESR is typically so low that the capacitive ripple may dominate. When this occurs, the output
ripple will be approximately sinusoidal and 90° phase shifted
from the switching action .
Given the availability and quality of MLCCs and the expected
output voltage of designs using the LM2735, there is really no
need to review any other capacitor technologies. Another
benefit of ceramic capacitors is their ability to bypass high
frequency noise. A certain amount of switching edge noise
will couple through parasitic capacitances in the inductor to
the output. A ceramic capacitor will bypass this noise while a
tantalum will not. Since the output capacitor is one of the two
external components that control the stability of the regulator
control loop, most applications will require a minimum at 4.7
µF of output capacitance. Like the input capacitor, recommended multilayer ceramic capacitors are X7R or X5R.
Again, verify actual capacitance at the desired operating voltage and temperature.
SETTING THE OUTPUT VOLTAGE
The output voltage is set using the following equation where
R1 is connected between the FB pin and GND, and R2 is
connected between VOUT and the FB pin.
20215831
FIGURE 6. LM2735 Without External Compensation
One can see that the Crossover frequency is fine, but the
phase margin at 0dB is very low (22°). A zero can be placed
just above the crossover frequency so that the phase margin
will be bumped up to a minimum of 45°. Below is the same
application with a zero added at 8 kHz.
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LM2735
20215829
FIGURE 8. Setting External Pole-Zero
20215832
FIGURE 7. LM2735 With External Compensation
The simplest method to determine the compensation component value is as follows.
Set the output voltage with the following equation.
There is an associated pole with the zero that was created in
the above equation.
It is always higher in frequency than the zero.
A right-half plane zero (RHPZ) is inherent to all boost converters. One must remember that the gain associated with a
right-half plane zero increases at 20dB per decade, but the
phase decreases by 45° per decade. For most applications
there is little concern with the RHPZ due to the fact that the
frequency at which it shows up is well beyond crossover, and
has little to no effect on loop stability. One must be concerned
with this condition for large inductor values and high output
currents.
Where R1 is the bottom resistor and R2 is the resistor tied to
the output voltage. The next step is to calculate the value of
C3. The internal compensation has been designed so that
when a zero is added between 5 kHz & 10 kHz the converter
will have good transient response with plenty of phase margin
for all input & output voltage combinations.
Lower output voltages will have the zero set closer to 10 kHz,
and higher output voltages will usually have the zero set closer to 5 kHz. It is always recommended to obtain a Gain/Phase
plot for your actual application. One could refer to the Typical
applications section to obtain examples of working applications and the associated component values.
Pole @ origin due to internal gm amplifier:
There are miscellaneous poles and zeros associated with
parasitics internal to the LM2735, external components, and
the PCB. They are located well over the crossover frequency,
and for simplicity are not discussed.
FP-ORIGIN
PCB Layout Considerations
Pole due to output load and capacitor:
When planning layout there are a few things to consider when
trying to achieve a clean, regulated output. The most important consideration when completing a Boost Converter layout
is the close coupling of the GND connections of the COUT capacitor and the LM2735 PGND pin. The GND ends should be
close to one another and be connected to the GND plane with
at least two through-holes. There should be a continuous
ground plane on the bottom layer of a two-layer board except
under the switching node island. The FB pin is a high
impedance node and care should be taken to make the FB
trace short to avoid noise pickup and inaccurate regulation.
The feedback resistors should be placed as close as possible
to the IC, with the AGND of R1 placed as close as possible to
the GND (pin 5 for the LLP) of the IC. The VOUT trace to R2
should be routed away from the inductor and any other traces
that are switching. High AC currents flow through the VIN, SW
This equation only determines the frequency of the pole for
perfect current mode control (CMC). I.e, it doesn’t take into
account the additional internal artificial ramp that is added to
the current signal for stability reasons. By adding artificial
ramp, you begin to move away from CMC to voltage mode
control (VMC). The artifact is that the pole due to the output
load and output capacitor will actually be slightly higher in frequency than calculated. In this example it is calculated at 650
Hz, but in reality it is around 1 kHz.
The zero created with capacitor C3 & resistor R2:
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that are obtainable via the National Semiconductor website.
The demonstration board consists of a two layer PCB with a
common input and output voltage application. Most of the
routing is on the top layer, with the bottom layer consisting of
a large ground plane. The placement of the external components satisfies the electrical considerations, and the thermal
performance has been improved by adding thermal vias and
a top layer “Dog-Bone”.
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LM2735
and VOUT traces, so they should be as short and wide as possible. However, making the traces wide increases radiated
noise, so the designer must make this trade-off. Radiated
noise can be decreased by choosing a shielded inductor. The
remaining components should also be placed as close as
possible to the IC. Please see Application Note AN-1229 for
further considerations and the LM2735 demo board as an example of a four-layer layout.
Below is an example of a good thermal & electrical PCB design. This is very similar to our LM2735 demonstration boards
LM2735
Example of Proper PCB Layout
guidelines). The electrical design considerations outweigh the
thermal considerations. Other factors that influence thermal
performance are thermal vias, copper weight, and number of
board layers.
Definitions
Heat energy is transferred from regions of high temperature
to regions of low temperature via three basic mechanisms:
radiation, conduction and convection.
Radiation: Electromagnetic transfer of heat between masses
at different temperatures.
Conduction: Transfer of heat through a solid medium.
Convection: Transfer of heat through the medium of a fluid;
typically air.
Conduction & Convection will be the dominant heat transfer
mechanism in most applications.
RθJA: Thermal impedance from silicon junction to ambient air
temperature.
RθJC: Thermal impedance from silicon junction to device case
temperature.
CθJC: Thermal Delay from silicon junction to device case temperature.
CθCA: Thermal Delay from device case to ambient air temperature.
RθJA & RθJC: These two symbols represent thermal
impedances, and most data sheets contain associated values
for these two symbols. The units of measurement are °C/
Watt.
RθJA is the sum of smaller thermal impedances (see simplified
thermal model below). The capacitors represent delays that
are present from the time that power and its associated heat
is increased or decreased from steady state in one medium
until the time that the heat increase or decrease reaches
steady state on the another medium.
20215840
FIGURE 9. Boost PCB Layout Guidelines
Thermal Design
When designing for thermal performance, one must consider
many variables:
Ambient Temperature: The surrounding maximum air temperature is fairly explanatory. As the temperature increases,
the junction temperature will increase. This may not be linear
though. As the surrounding air temperature increases, resistances of semiconductors, wires and traces increase. This will
decrease the efficiency of the application, and more power
will be converted into heat, and will increase the silicon junction temperatures further.
Forced Airflow: Forced air can drastically reduce the device
junction temperature. Air flow reduces the hot spots within a
design. Warm airflow is often much better than a lower ambient temperature with no airflow.
External Components: Choose components that are efficient, and you can reduce the mutual heating between devices.
PCB design with thermal performance in mind:
The PCB design is a very important step in the thermal design
procedure. The LM2735 is available in three package options
(5 pin SOT23, 8 pin eMSOP & 6 pin LLP). The options are
electrically the same, but difference between the packages is
size and thermal performance. The LLP and eMSOP have
thermal Die Attach Pads (DAP) attached to the bottom of the
packages, and are therefore capable of dissipating more heat
than the SOT23 package. It is important that the customer
choose the correct package for the application. A detailed
thermal design procedure has been included in this data
sheet. This procedure will help determine which package is
correct, and common applications will be analyzed.
There is one significant thermal PCB layout design consideration that contradicts a proper electrical PCB layout design
consideration. This contradiction is the placement of external
components that dissipate heat. The greatest external heat
contributor is the external Schottky diode. It would be nice if
you were able to separate by distance the LM2735 from the
Schottky diode, and thereby reducing the mutual heating effect. This will however create electrical performance issues.
It is important to keep the LM2735, the output capacitor, and
Schottky diode physically close to each other (see PCB layout
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20215841
FIGURE 10. Simplified Thermal Impedance Model
14
We will talk more about calculating the variables of this equation later, and how to eventually calculate a proper junction
temperature with relative certainty. For now we need to define
the process of calculating the junction temperature and clarify
some common misconceptions.
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LM2735
RθJA [Variables]:
• Input Voltage, Output Voltage, Output Current, RDSon.
• Ambient temperature & air flow.
• Internal & External components power dissipation.
• Package thermal limitations.
• PCB variables (copper weight, thermal via’s, layers
component placement).
It would be wrong to assume that the top case temperature is
the proper temperature when calculating
value. The
value represents the thermal impedance of all six sides
of a package, not just the top side. This document will refer to
a thermal impedance called
.
represents a thermal
impedance associated with just the top case temperature.
This will allow one to calculate the junction temperature with
a thermal sensor connected to the top case.
The datasheet values for these symbols are given so that one
might compare the thermal performance of one package
against another. In order to achieve a comparison between
packages, all other variables must be held constant in the
comparison (PCB size, copper weight, thermal vias, power
dissipation, VIN, VOUT, Load Current etc). This does shed light
on the package performance, but it would be a mistake to use
these values to calculate the actual junction temperature in
your application.
LM2735
and loads. All loss elements will mutually increase the heat
on the PCB, and therefore increase each other’s temperatures.
LM2735 Thermal Models
Heat is dissipated from the LM2735 and other devices. The
external loss elements include the Schottky diode, inductor,
20215843
FIGURE 11. Thermal Schematic
20215844
FIGURE 12. Associated Thermal Model
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16
LM2735
Calculating Efficiency, and Junction
Temperature
The complete LM2735 DC/DC converter efficiency (η) can be
calculated in the following manner.
The diode, NMOS switch, and inductor DCR losses are included in this calculation. Setting any loss element to zero will
simplify the equation.
VD is the forward voltage drop across the Schottky diode. It
can be obtained from the manufacturer’s Electrical Characteristics section of the data sheet.
The conduction losses in the diode are calculated as follows:
PDIODE = VD x IO
Power loss (PLOSS) is the sum of two types of losses in the
converter, switching and conduction. Conduction losses usually dominate at higher output loads, where as switching
losses remain relatively fixed and dominate at lower output
loads.
Losses in the LM2735 Device: PLOSS = PCOND + PSW + PQ
Conversion ratio of the Boost Converter with conduction loss
elements inserted:
Depending on the duty cycle, this can be the single most significant power loss in the circuit. Care should be taken to
choose a diode that has a low forward voltage drop. Another
concern with diode selection is reverse leakage current. Depending on the ambient temperature and the reverse voltage
across the diode, the current being drawn from the output to
the NMOS switch during time D could be significant, this may
increase losses internal to the LM2735 and reduce the overall
efficiency of the application. Refer to Schottky diode
manufacturer’s data sheets for reverse leakage specifications, and typical applications within this data sheet for diode
selections.
Another significant external power loss is the conduction loss
in the input inductor. The power loss within the inductor can
be simplified to:
One can see that if the loss elements are reduced to zero, the
conversion ratio simplifies to:
PIND = IIN2RDCR
The LM2735 conduction loss is mainly associated with the
internal NFET:
And we know:
PCOND-NFET = I2SW-rms x RDSON x D
Therefore:
20215852
FIGURE 13. LM2735 Switch Current
Calculations for determining the most significant power losses are discussed below. Other losses totaling less than 2%
are not discussed.
A simple efficiency calculation that takes into account the
conduction losses is shown below:
(small ripple approximation)
PCOND-NFET = IIN2 x RDSON x D
17
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LM2735
PCONDUCTION = IIN2 x D x RDSON x 305 mW
Diode Losses
VD = 0.45V
PDIODE = VD x IIN(1-D) = 236 mW
The value for should be equal to the resistance at the junction
temperature you wish to analyze. As an example, at 125°C
and VIN = 5V, RDSON = 250 mΩ (See typical graphs for value).
Switching losses are also associated with the internal NMOS
switch. They occur during the switch on and off transition periods, where voltages and currents overlap resulting in power
loss.
The simplest means to determine this loss is to empirically
measuring the rise and fall times (10% to 90%) of the switch
at the switch node:
Inductor Power Losses
RDCR = 75 mΩ
PIND = IIN2 x RDCR = 145 mW
Total Power Losses are:
TABLE 2. Power Loss Tabulation
5V
VIN
PSWR = 1/2(VOUT x IIN x FSW x TRISE)
VOUT
12V
IOUT
500mA
POUT
6W
PDIODE
236mW
PSWF = 1/2(VOUT x IIN x FSW x TFALL)
VD
0.4V
PSW = PSWR + PSWF
FSW
1.6MHz
TRISE
6nS
PSWR
80mW
TFALL
5nS
PSWF
70mW
Typical Switch-Node Rise and Fall Times
VIN
VOUT
TRISE
TFALL
3V
5V
6nS
4nS
IQ
4mA
PQ
20mW
250mΩ
PCOND
305mW
PIND
145mW
PLOSS
856mW
5V
12V
6nS
5nS
RDSon
3V
12V
7nS
5nS
RDCR
75mΩ
5V
18V
7nS
5nS
D
0.623
η
86%
Quiescent Power Losses
IQ is the quiescent operating current, and is typically around
4mA.
PINTERNAL = PCOND + PSW = 475 mW
PQ = IQ x VIN
Calculating
Example Efficiency Calculation:
and
TABLE 1. Operating Conditions
5V
VIN
VOUT
12V
IOUT
500mA
VD
0.4V
FSW
1.60MHz
IQ
4mA
TRISE
6nS
TFALL
5nS
RDSon
250mΩ
RDCR
50mΩ
D
0.64
IIN
1.4A
We now know the internal power dissipation, and we are trying to keep the junction temperature at or below 125°C. The
and/or
. This is
next step is to calculate the value for
actually very simple to accomplish, and necessary if you think
you may be marginal with regards to thermals or determining
what package option is correct.
The LM2735 has a thermal shutdown comparator. When the
silicon reaches a temperature of 160°C, the device shuts
down until the temperature reduces to 150°C. Knowing this,
one can calculate the
or the
of a specific application.
Because the junction to top case thermal impedance is much
lower than the thermal impedance of junction to ambient air,
the error in calculating
is lower than for
. However,
you will need to attach a small thermocouple onto the top case
value.
of the LM2735 to obtain the
Knowing the temperature of the silicon when the device shuts
down allows us to know three of the four variables. Once we
calculate the thermal impedance, we then can work backwards with the junction temperature set to 125°C to see what
ΣPCOND + PSW + PDIODE + PIND + PQ = PLOSS
Quiescent Power Losses
PQ = IQ x VIN = 20 mW
Switching Power Losses
PSWR = 1/2(VOUT x IIN x FSW x TRISE) ≊ 6 ns ≊ 80 mW
PSWF = 1/2(VOUT x IIN x FSW x TFALL) ≊ 5 ns ≊ 70 mW
PSW = PSWR + PSWF = 150 mW
Internal NFET Power Losses
RDSON = 250 mΩ
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18
LM2735
maximum ambient air temperature keeps the silicon below
the 125°C temperature.
Procedure:
Place your application into a thermal chamber. You will need
to dissipate enough power in the device so you can obtain a
good thermal impedance value.
Raise the ambient air temperature until the device goes into
thermal shutdown. Record the temperatures of the ambient
air and/or the top case temperature of the LM2735. Calculate
the thermal impedances.
Example from previous calculations:
Pdiss = 475 mW
Ta @ Shutdown = 139°C
Tc @ Shutdown = 155°C
20215856
FIGURE 14. RθJA vs Internal Dissipation for the LLP-6
and eMSOP-8 Package
LLP = 55°C/W
LLP = 21°C/W
LLP & eMSOP typical applications will produce
numbers
in the range of 50°C/W to 65°C/W, and
will vary between
18°C/W and 28°C/W. These values are for PCB’s with two
and four layer boards with 0.5 oz copper, and four to six thermal vias to bottom side ground plane under the DAP.
For 5-pin SOT23 package typical applications, RθJA numbers
will range from 80°C/W to 110°C/W, and
will vary between
50°C/W and 65°C/W. These values are for PCB’s with two &
four layer boards with 0.5 oz copper, with two to four thermal
vias from GND pin to bottom layer.
Here is a good rule of thumb for typical thermal impedances,
and an ambient temperature maximum of 75°C: If your design
requires that you dissipate more than 400mW internal to the
LM2735, or there is 750mW of total power loss in the application, it is recommended that you use the 6 pin LLP or the 8
pin eMSOP package.
Note: To use these procedures it is important to dissipate an
amount of power within the device that will indicate a true
thermal impedance value. If one uses a very small internal
dissipated value, one can see that the thermal impedance
calculated is abnormally high, and subject to error. The graph
below shows the nonlinear relationship of internal power dissipation vs .
.
SEPIC Converter
The LM2735 can easily be converted into a SEPIC converter.
A SEPIC converter has the ability to regulate an output voltage that is either larger or smaller in magnitude than the input
voltage. Other converters have this ability as well (CUK and
Buck-Boost), but usually create an output voltage that is opposite in polarity to the input voltage. This topology is a perfect
fit for Lithium Ion battery applications where the input voltage
for a single cell Li-Ion battery will vary between 3V & 4.5V and
the output voltage is somewhere in between. Most of the
analysis of the LM2735 Boost Converter is applicable to the
LM2735 SEPIC Converter.
SEPIC Design Guide:
SEPIC Conversion ratio without loss elements:
Therefore:
Small ripple approximation:
In a well-designed SEPIC converter, the output voltage, and
input voltage ripple, the inductor ripple and is small in comparison to the DC magnitude. Therefore it is a safe approximation to assume a DC value for these components. The
main objective of the Steady State Analysis is to determine
the steady state duty-cycle, voltage and current stresses on
all components, and proper values for all components.
In a steady-state converter, the net volt-seconds across an
inductor after one cycle will equal zero. Also, the charge into
a capacitor will equal the charge out of a capacitor in one cycle.
Therefore:
19
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LM2735
Applying Charge balance on C1:
Since there are no DC voltages across either inductor, and
capacitor C6 is connected to Vin through L1 at one end, or to
ground through L2 on the other end, we can say that
Substituting IL1 into IL2
VC1 = VIN
Therefore:
The average inductor current of L2 is the average output load.
This verifies the original conversion ratio equation.
It is important to remember that the internal switch current is
equal to IL1 and IL2. During the D interval. Design the converter
so that the minimum guaranteed peak switch current limit
(2.1A) is not exceeded.
20215863
FIGURE 15. Inductor Volt-Sec Balance Waveform
20215880
FIGURE 16. SEPIC CONVERTER Schematic
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20
LM2735
Steady State Analysis with Loss
Elements
20215866
Using inductor volt-second balance & capacitor charge balance, the following equations are derived:
20215890
Efficiencies for Typical SEPIC Application
SEPIC Converter PCB Layout
The layout guidelines described for the LM2735 Boost-Converter are applicable to the SEPIC Converter. Below is a
proper PCB layout for a SEPIC Converter.
Therefore:
One can see that all variables are known except for the duty
cycle (D). A quadratic equation is needed to solve for D. A
less accurate method of determining the duty cycle is to assume efficiency, and calculate the duty cycle.
20215872
FIGURE 17. SEPIC PCB Layout
21
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LM2735
LLP Package
The LM2735 packaged in the 6–pin LLP:
20215873
FIGURE 18. Internal LLP Connection
For certain high power applications, the PCB land may be
modified to a "dog bone" shape (see Figure 19). Increasing
the size of ground plane, and adding thermal vias can reduce
the RθJA for the application.
20215874
FIGURE 19. PCB Dog Bone Layout
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22
LM2735
LM2735X SOT23-5 Design Example 1
20215875
LM2735X (1.6MHz): Vin = 5V, Vout = 12V @ 350mA
Part ID
Part Value
Manufacturer
Part Number
U1
2.1A Boost Regulator
NSC
LM2735XMF
C1, Input Cap
22µF, 6.3V, X5R
TDK
C2012X5R0J226M
C2 Output Cap
10µF, 25V, X5R
TDK
C3216X5R1E106M
C3 Comp Cap
330pF
TDK
C1608X5R1H331K
D1, Catch Diode
0.4Vf Schottky 1A, 20VR
ST
STPS120M
L1
15µH 1.5A
Coilcraft
MSS5131-153ML
R1
10.2kΩ, 1%
Vishay
CRCW06031022F
R2
86.6kΩ, 1%
Vishay
CRCW06038662F
R3
100kΩ, 1%
Vishay
CRCW06031003F
23
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LM2735
LM2735Y SOT23-5 Design Example 2
20215875
LM2735Y (520kHz): Vin = 5V, Vout = 12V @ 350mA
Part ID
Part Value
Manufacturer
Part Number
U1
2.1A Boost Regulator
NSC
LM2735YMF
C1, Input Cap
22µF, 6.3V, X5R
TDK
C2012X5R0J226M
C2 Output Cap
10µF, 25V, X5R
TDK
C3216X5R1E106M
C3 Comp Cap
330pF
TDK
C1608X5R1H331K
D1, Catch Diode
0.4Vf Schottky 1A, 20VR
ST
STPS120M
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L1
33µH 1.5A
Coilcraft
DS3316P-333ML
R1
10.2kΩ, 1%
Vishay
CRCW06031022F
R2
86.6kΩ, 1%
Vishay
CRCW06038662F
R3
100kΩ, 1%
Vishay
CRCW06031003F
24
LM2735
LM2735X LLP-6 Design Example 3
20215876
LM2735X (1.6MHz): Vin = 3.3V, Vout = 12V @ 350mA
Part ID
Part Value
Manufacturer
Part Number
U1
2.1A Boost Regulator
NSC
LM2735XSD
C1 Input Cap
22µF, 6.3V, X5R
TDK
C2012X5R0J226M
C2 Input Cap
No Load
C3 Output Cap
10µF, 25V, X5R
TDK
C3216X5R1E106M
C4 Output Cap
No Load
C5 Comp Cap
330pF
TDK
C1608X5R1H331K
D1, Catch Diode
0.4Vf Schottky 1A, 20VR
ST
STPS120M
L1
6.8µH 2A
Coilcraft
DO1813H-682ML
R1
10.2kΩ, 1%
Vishay
CRCW06031022F
R2
86.6kΩ, 1%
Vishay
CRCW06038662F
R3
100kΩ, 1%
Vishay
CRCW06031003F
25
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LM2735
LM2735Y LLP-6 Design Example 4
20215876
LM2735Y (520kHz): Vin = 3.3V, Vout = 12V @ 350mA
Part ID
Part Value
Manufacturer
Part Number
U1
2.1A Boost Regulator
NSC
LM2735YSD
C1 Input Cap
22µF, 6.3V, X5R
TDK
C2012X5R0J226M
C2 Input Cap
No Load
C3 Output Cap
10µF, 25V, X5R
TDK
C3216X5R1E106M
C4 Output Cap
No Load
C5 Comp Cap
330pF
TDK
C1608X5R1H331K
D1, Catch Diode
0.4Vf Schottky 1A, 20VR
ST
STPS120M
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L1
15µH 2A
Coilcraft
MSS5131-153ML
R1
10.2kΩ, 1%
Vishay
CRCW06031022F
R2
86.6kΩ, 1%
Vishay
CRCW06038662F
R3
100kΩ, 1%
Vishay
CRCW06031003F
26
LM2735
LM2735Y eMSOP-8 Design Example 5
20215877
LM2735Y (520kHz): Vin = 3.3V, Vout = 12V @ 350mA
Part ID
Part Value
Manufacturer
U1
2.1A Boost Regulator
NSC
LM2735YMY
C1 Input Cap
22µF, 6.3V, X5R
TDK
C2012X5R0J226M
TDK
C3216X5R1E106M
C2 Input Cap
No Load
C3 Output Cap
10µF, 25V, X5R
C4 Output Cap
No Load
Part Number
C5 Comp Cap
330pF
TDK
C1608X5R1H331K
D1, Catch Diode
0.4Vf Schottky 1A, 20VR
ST
STPS120M
L1
15µH 1.5A
Coilcraft
MSS5131-153ML
R1
10.2kΩ, 1%
Vishay
CRCW06031022F
R2
86.6kΩ, 1%
Vishay
CRCW06038662F
R3
100kΩ, 1%
Vishay
CRCW06031003F
27
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LM2735
LM2735X SOT23-5 Design Example 6
20215878
LM2735X (1.6MHz): Vin = 3V, Vout = 5V @ 500mA
Part ID
Part Value
Manufacturer
Part Number
U1
2.1A Boost Regulator
NSC
LM2735XMF
C1, Input Cap
10µF, 6.3V, X5R
TDK
C2012X5R0J106K
C2, Output Cap
10µF, 6.3V, X5R
TDK
C2012X5R0J106K
C3 Comp Cap
1000pF
TDK
C1608X5R1H102K
D1, Catch Diode
0.4Vf Schottky 1A, 20VR
ST
STPS120M
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L1
10µH 1.2A
Coilcraft
DO1608C-103ML
R1
10.0kΩ, 1%
Vishay
CRCW08051002F
R2
30.1kΩ, 1%
Vishay
CRCW08053012F
R3
100kΩ, 1%
Vishay
CRCW06031003F
28
LM2735
LM2735Y SOT23-5 Design Example 7
20215878
LM2735Y (520kHz): Vin = 3V, Vout = 5V @ 750mA
Part ID
Part Value
Manufacturer
Part Number
U1
2.1A Boost Regulator
NSC
LM2735YMF
C1 Input Cap
22µF, 6.3V, X5R
TDK
C2012X5R0J226M
C2 Output Cap
22µF, 6.3V, X5R
TDK
C2012X5R0J226M
C3 Comp Cap
1000pF
TDK
C1608X5R1H102K
D1, Catch Diode
0.4Vf Schottky 1A, 20VR
ST
STPS120M
L1
22µH 1.2A
Coilcraft
MSS5131-223ML
R1
10.0kΩ, 1%
Vishay
CRCW08051002F
R2
30.1kΩ, 1%
Vishay
CRCW08053012F
R3
100kΩ, 1%
Vishay
CRCW06031003F
29
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LM2735
LM2735X SOT23-5 Design Example 8
20215879
LM2735X (1.6MHz): Vin = 3.3V, Vout = 20V @ 100mA
Part ID
Part Value
Manufacturer
Part Number
U1
2.1A Boost Regulator
NSC
LM2735XMF
C1, Input Cap
22µF, 6.3V, X5R
TDK
C2012X5R0J226M
C2, Output Cap
4.7µF, 25V, X5R
TDK
C3216X5R1E475K
C3 Comp Cap
470pF
TDK
C1608X5R1H471K
D1, Catch Diode
0.4Vf Schottky 500mA, 30VR
Vishay
MBR0530
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L1
10µH 1.2A
Coilcraft
DO1608C-103ML
R1
10.0kΩ, 1%
Vishay
CRCW06031002F
R2
150kΩ, 1%
Vishay
CRCW06031503F
R3
100kΩ, 1%
Vishay
CRCW06031003F
30
LM2735
LM2735Y SOT23-5 Design Example 9
20215879
LM2735Y (520kHz): Vin = 3.3V, Vout = 20V @ 100mA
Part ID
Part Value
Manufacturer
Part Number
U1
2.1A Boost Regulator
NSC
LM2735YMF
C1 Input Cap
22µF, 6.3V, X5R
TDK
C2012X5R0J226M
C2 Output Cap
10µF, 25V, X5R
TDK
C3216X5R1E106M
C3 Comp Cap
470pF
TDK
C1608X5R1H471K
D1, Catch Diode
0.4Vf Schottky 500mA, 30VR
Vishay
MBR0530
L1
33µH 1.5A
Coilcraft
DS3316P-333ML
R1
10.0kΩ, 1%
Vishay
CRCW06031002F
R2
150.0kΩ, 1%
Vishay
CRCW06031503F
R3
100kΩ, 1%
Vishay
CRCW06031003F
31
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LM2735
LM2735X LLP-6 Design Example 10
20215876
LM2735X (1.6MHz): Vin = 3.3V, Vout = 20V @ 150mA
Part ID
Part Value
Manufacturer
Part Number
U1
2.1A Boost Regulator
NSC
LM2735XSD
C1 Input Cap
22µF, 6.3V, X5R
TDK
C2012X5R0J226M
C2 Input Cap
22µF, 6.3V, X5R
TDK
C2012X5R0J226M
C3 Output Cap
10µF, 25V, X5R
TDK
C3216X5R1E106M
C4 Output Cap
No Load
C5 Comp Cap
470pF
TDK
C1608X5R1H471K
D1, Catch Diode
0.4Vf Schottky 500mA, 30VR
Vishay
MBR0530
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L1
8.2µH 2A
Coilcraft
DO1813H-822ML
R1
10.0kΩ, 1%
Vishay
CRCW06031002F
R2
150kΩ, 1%
Vishay
CRCW06031503F
R3
100kΩ, 1%
Vishay
CRCW06031003F
32
LM2735
LM2735Y LLP-6 Design Example 11
20215876
LM2735Y (520kHz): Vin = 3.3V, Vout = 20V @ 150mA
Part ID
Part Value
Manufacturer
Part Number
U1
2.1A Boost Regulator
NSC
LM2735YSD
C1 Input Cap
10µF, 6.3V, X5R
TDK
C2012X5R0J106K
C2 Input Cap
10µF, 6.3V, X5R
TDK
C2012X5R0J106K
C3 Output Cap
10µF, 25V, X5R
TDK
C3216X5R1E106M
C4 Output Cap
No Load
C5 Comp Cap
470pF
TDK
C1608X5R1H471K
D1, Catch Diode
0.4Vf Schottky 500mA, 30VR
Vishay
MBR0530
L1
22µH 1.5A
Coilcraft
DS3316P-223ML
R1
10.0kΩ, 1%
Vishay
CRCW06031002F
R2
150kΩ, 1%
Vishay
CRCW06031503F
R3
100kΩ, 1%
Vishay
CRCW06031003F
33
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LM2735
LM2735X LLP-6 SEPIC Design Example 12
20215880
LM2735X (1.6MHz): Vin = 2.7V - 5V, Vout = 3.3V @ 500mA
Part ID
Part Value
Manufacturer
U1
2.1A Boost Regulator
NSC
LM2735XSD
C1 Input Cap
22µF, 6.3V, X5R
TDK
C2012X5R0J226M
TDK
C3216X5R1E106M
C2 Input Cap
No Load
C3 Output Cap
10µF, 25V, X5R
C4 Output Cap
No Load
Part Number
C5 Comp Cap
2200pF
TDK
C1608X5R1H222K
C6
2.2µF 16V
TDK
C2012X5R1C225K
D1, Catch Diode
0.4Vf Schottky 1A, 20VR
ST
STPS120M
L1
6.8µH
Coilcraft
DO1608C-682ML
L2
6.8µH
Coilcraft
DO1608C-682ML
R1
10.2kΩ, 1%
Vishay
CRCW06031002F
R2
16.5kΩ, 1%
Vishay
CRCW06031652F
R3
100kΩ, 1%
Vishay
CRCW06031003F
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34
LM2735
LM2735Y eMSOP-8 SEPIC Design Example 13
20215881
LM2735Y (520kHz): Vin = 2.7V - 5V, Vout = 3.3V @ 500mA
Part ID
Part Value
Manufacturer
U1
2.1A Boost Regulator
NSC
Part Number
LM2735YMY
C1 Input Cap
22µF, 6.3V, X5R
TDK
C2012X5R0J226M
C2 Input Cap
No Load
C3 Output Cap
10µF, 25V, X5R
TDK
C3216X5R1E106M
C4 Output Cap
No Load
C5 Comp Cap
2200pF
TDK
C1608X5R1H222K
C2012X5R1C225K
C6
2.2µF 16V
TDK
D1, Catch Diode
0.4Vf Schottky 1A, 20VR
ST
STPS120M
L1
15µH 1.5A
Coilcraft
MSS5131-153ML
L2
15µH 1.5A
Coilcraft
MSS5131-153ML
R1
10.2kΩ, 1%
Vishay
CRCW06031002F
R2
16.5kΩ, 1%
Vishay
CRCW06031652F
R3
100kΩ, 1%
Vishay
CRCW06031003F
35
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LM2735
LM2735X SOT23-5 LED Design Example 14
20215882
LM2735X (1.6MHz): Vin = 2.7V - 5V, Vout = 20V @ 50mA
Part ID
Part Value
Manufacturer
U1
2.1A Boost Regulator
NSC
LM2735XMF
C1 Input Cap
22µF, 6.3V, X5R
TDK
C2012X5R0J226M
C2 Output Cap
4.7µF, 25V, X5R
TDK
C3216JB1E475K
D1, Catch Diode
0.4Vf Schottky 500mA, 30VR
Vishay
MBR0530
L1
15µH 1.5A
Coilcraft
MSS5131-153ML
R1
25.5Ω, 1%
Vishay
CRCW080525R5F
R2
100Ω, 1%
Vishay
CRCW08051000F
R3
100kΩ, 1%
Vishay
CRCW06031003F
www.national.com
36
Part Number
LM2735
LM2735Y LLP-6 FlyBack Design Example 15
20215883
LM2735Y (520kHz): Vin = 5V, Vout = ±12V 150mA
Part ID
Part Value
Manufacturer
Part Number
U1
2.1A Boost Regulator
NSC
LM2735YSD
C1 Input Cap
22µF, 6.3V, X5R
TDK
C2012X5R0J226M
C2 Output Cap
10µF, 25V, X5R
TDK
C3216X5R1E106M
C3 Output Cap
10µF, 25V, X5R
TDK
C3216X5R1E106M
Cf Comp Cap
330pF
TDK
C1608X5R1H331K
D1, D2 Catch Diode
0.4Vf Schottky 500mA, 30VR
Vishay
MBR0530
R1
10.0kΩ, 1%
Vishay
CRCW06031002F
R2
86.6kΩ, 1%
Vishay
CRCW06038662F
R3
100kΩ, 1%
Vishay
CRCW06031003F
T1
37
www.national.com
LM2735
LM2735X SOT23-5 LED Design Example 16
VRAIL > 5.5V Application
202158a3
LM2735X (1.6MHz): VPWR = 9V, Vout = 12V @ 500mA
Part ID
Part Value
Manufacturer
Part Number
U1
2.1A Boost Regulator
NSC
LM2735XMF
C1, Input Cap
10µF, 6.3V, X5R
TDK
C2012X5R0J106K
C2, Output Cap
10µF, 25V, X5R
TDK
C3216X5R1E106M
C3 VIN Cap
0.1µF, 6.3V, X5R
TDK
C2012X5R0J104K
C4 Comp Cap
1000pF
TDK
C1608X5R1H102K
D1, Catch Diode
0.4Vf Schottky 1A, 20VR
ST
STPS120M
D2
3.3V Zener, SOT23
Diodes Inc
BZX84C3V3
L1
6.8µH 2A
Coilcraft
DO1813H-682ML
R1
10.0kΩ, 1%
Vishay
CRCW08051002F
R2
86.6kΩ, 1%
Vishay
CRCW08058662F
R3
100kΩ, 1%
Vishay
CRCW06031003F
R4
499Ω, 1%
Vishay
CRCW06034991F
www.national.com
38
LM2735
LM2735X SOT23-5 LED Design Example 17
Two Input Voltage Rail Application
202158a4
LM2735X (1.6MHz): VPWR = 9V in = 2.7V - 5.5V, Vout = 12V @ 500mA
Part ID
Part Value
Manufacturer
Part Number
U1
2.1A Boost Regulator
NSC
LM2735XMF
C1, Input Cap
10µF, 6.3V, X5R
TDK
C2012X5R0J106K
C2, Output Cap
10µF, 25V, X5R
TDK
C3216X5R1E106M
C3 VIN Cap
0.1µF, 6.3V, X5R
TDK
C2012X5R0J104K
C1608X5R1H102K
C4 Comp Cap
1000pF
TDK
D1, Catch Diode
0.4Vf Schottky 1A, 20VR
ST
STPS120M
L1
6.8µH 2A
Coilcraft
DO1813H-682ML
R1
10.0kΩ, 1%
Vishay
CRCW08051002F
R2
86.6kΩ, 1%
Vishay
CRCW08058662F
R3
100kΩ, 1%
Vishay
CRCW06031003F
39
www.national.com
LM2735
Physical Dimensions inches (millimeters) unless otherwise noted
6-Lead LLP Package
NS Package Number SDE06A
5-Lead SOT23-5 Package
NS Package Number MF05A
www.national.com
40
LM2735
8-Lead eMSOP Package
NS Package Number MUY08A
41
www.national.com
LM2735 520kHz/1.6MHz – Space-Efficient Boost and SEPIC DC-DC Regulator
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