Multichannel ISM Band FSK/GFSK/OOK/GOOK/ASK Transmitter ADF7012 FEATURES GENERAL DESCRIPTION Single-chip, low power UHF transmitter 75 MHz to 1 GHz frequency operation Multichannel operation using Frac-N PLL 2.3 V to 3.6 V operation On-board regulator—stable performance Programmable output power: −16 dBm to +14 dBm, 0.4 dB steps Data rates: dc to 179.2 kbps Low current consumption: 868 MHz, 10 dBm, 21 mA 433 MHz, 10 dBm, 17 mA 315 MHz, 0 dBm, 10 mA Programmable low battery voltage indicator 24-lead TSSOP The ADF7012 is a low power FSK/GFSK/OOK/GOOK/ASK UHF transmitter designed for short range devices (SRDs). The output power, output channels, deviation frequency, and modulation type are programmable by using four, 32-bit registers. The fractional-N PLL and VCO with external inductor enable the user to select any frequency in the 75 MHz to 1 GHz band. The fast lock times of the fractional-N PLL make the ADF7012 suitable in fast frequency hopping systems. The fine frequency deviations available and PLL phase noise performance facilitates narrow-band operation. There are five selectable modulation schemes: binary frequency shift keying (FSK), Gaussian frequency shift keying (GFSK), binary on-off keying (OOK), Gaussian on-off keying (GOOK), and amplitude shift keying (ASK). In the compensation register, the output can be moved in <1 ppm steps so that indirect compensation for frequency error in the crystal reference can be made. APPLICATIONS Low cost wireless data transfer Security systems RF remote controls Wireless metering Secure keyless entry A simple 3-wire interface controls the registers. In power-down, the part has a typical quiescent current of <0.1 μA. FUNCTIONAL BLOCK DIAGRAM CREG PRINTED INDUCTOR OSC1 OSC2 CLKOUT CPVDD CPGND L1 L2 CVCO VDD OOK\ASK ÷CLK VCO PA ÷R PFD/ CHARGE PUMP DVDD RFOUT RF GND DGND CREG OOK\ASK +FRACTIONAL N FSK\GFSK Σ-∆ LDO REGULATOR TxCLK PLL LOCK DETECT LE DATA SERIAL INTERFACE CLK CE FREQUENCY COMPENSATION MUXOUT MUXOUT BATTERY MONITOR CENTER FREQUENCY AGND RSET 04617-0-001 TxDATA Figure 1. Rev. 0 Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.326.8703 © 2004 Analog Devices, Inc. All rights reserved. ADF7012 TABLE OF CONTENTS Specifications..................................................................................... 3 GOOK Modulation .................................................................... 15 Timing Characteristics..................................................................... 5 Output Divider ........................................................................... 16 Absolute Maximum Ratings............................................................ 6 MUXOUT Modes....................................................................... 16 Transistor Count........................................................................... 6 Theory of Operation ...................................................................... 17 ESD Caution.................................................................................. 6 Choosing the External Inductor Value.................................... 17 Pin Configuration and Function Descriptions............................. 7 Choosing the Crystal/PFD Value............................................. 17 Typical Performance Characteristics ............................................. 8 Tips on Designing the Loop Filter ........................................... 17 315 MHz ........................................................................................ 8 PA Matching................................................................................ 18 433 MHz ........................................................................................ 9 Transmit Protocol and Coding Considerations ..................... 18 868 MHz ...................................................................................... 10 Application Examples .................................................................... 19 915 MHz ...................................................................................... 11 315 MHz Operation ................................................................... 20 Circuit Description......................................................................... 12 433 MHz Operation ................................................................... 21 PLL Operation ............................................................................ 12 868 MHz Operation ................................................................... 22 Crystal Oscillator........................................................................ 12 915 MHz Operation ................................................................... 23 Crystal Compensation Register................................................ 12 Register Descriptions ..................................................................... 24 Clock Out Circuit ....................................................................... 12 R Register..................................................................................... 24 Loop Filter ................................................................................... 13 N-Counter Latch ........................................................................ 25 Voltage-Controlled Oscillator (VCO) ..................................... 13 Modulation Register .................................................................. 26 Voltage Regulators...................................................................... 13 Function Register ....................................................................... 27 FSK Modulation.......................................................................... 13 Outline Dimensions ....................................................................... 28 GFSK Modulation ...................................................................... 14 Ordering Guide .......................................................................... 28 Power Amplifier.......................................................................... 14 REVISION HISTORY 10/04—Revision 0: Initial Version Rev. 0 | Page 2 of 28 ADF7012 SPECIFICATIONS DVDD = 2.3 V – 3.6 V; AGND = DGND = 0 V; TA = TMIN to TMAX, unless otherwise noted. Operating temperature range is −40°C to +85°C. Table 1. Parameter RF OUTPUT CHARACTERISTICS Operating Frequency Phase Frequency Detector MODULATION PARAMETERS Data Rate FSK/GFSK Data Rate ASK/OOK Deviation FSK/GFSK GFSK BT ASK Modulation Depth OOK Feedthrough (PA Off) POWER AMPLIFIER PARAMETERS Max Power Setting, DVDD = 3.6 V Max Power Setting, DVDD = 3.0 V Max Power Setting, DVDD = 2.3 V Max Power Setting, DVDD = 3.6 V Max Power Setting, DVDD = 3.0 V Max Power Setting, DVDD = 2.3 V PA Programmability POWER SUPPLIES DVDD Current Comsumption 315 MHz, 0 dBm/5 dBm 433 MHz, 0 dBm/10 dBm 868 MHz, 0 dBm/10 dBm/14 dBm 915 MHz, 0 dBm/10 dBm/14 dBm VCO Current Consumption Crystal Oscillator Current Consumption Regulator Current Consumption Power-Down Current REFERENCE INPUT Crystal Reference Frequency Single-Ended Reference Frequency Crystal Power-On Time 3.4 MHz/26 MHz Single-Ended Input Level B Version Unit Conditions/Comments 75/1000 MHz min/max VCO range adjustable using external inductor; divide-by-2, -4, -8 options may be required FRF/128 Hz min 179.2 64 kbps Kbps PFD/214 511 × PFD/214 0.5 25 −40 −80 Hz min Hz max typ dB max dBm typ dBm typ FRF = Fvco FRF = Fvco/2 14 13.5 12.5 14.5 14 13 0.4 dBm dBm dBm dBm dBm dBm dB typ FRF = 915 MHz, PA is matched into 50 Ω FRF = 915 MHz, PA is matched into 50 Ω FRF = 915 MHz, PA is matched into 50 Ω FRF = 433 MHz, PA is matched into 50 Ω FRF = 433 MHz, PA is matched into 50 Ω FRF = 433 MHz, PA is matched into 50 Ω PA output = −20 dBm to +13 dBm 2.3/3.6 V min/V max 8/14 10/18 14/21/32 16/24/35 1/8 190 280 0.1/1 mA typ mA typ mA typ mA typ mA min/max µA typ µA typ µA typ/max 3.4/26 MHz min/max 3.4/26 1.8/2.2 CMOS Levels MHz min/max ms typ Using 1 MHz loop bandwidth Based on US FCC 15.247 specfications for ACP; higher data rates are achievable depending on local regulations For example, 10 MHz PFD – deviation min = ± 610 Hz For example, 10 MHz PFD – deviation max = ± 311.7 kHz DVDD = 3.0 V, PA is matched into 50 Ω, IVCO = min VCO current consumption is programmable CE to Clock Enable Valid Refer to the LOGIC INPUTS parameter. Applied to OSC 2 – oscillator circuit disabled. Rev. 0 | Page 3 of 28 ADF7012 Parameter PHASE-LOCKED LOOP PARAMETERS VCO Gain 315MHz 433MHz 868MHz B Version Unit Conditions/Comments 22 24 80 MHz/V typ MHz/V typ MHz/V typ VCO divide-by-2 active VCO divide-by-2 active 88 0.3/2.0 −65/−70 MHz/V typ V min/max dBc IVCO is programmable 0.3 0.9 1.5 2.1 mA typ mA typ mA typ mA typ Refering to DB[7:6] in Function Register Refering to DB[7:6] in Function Register Refering to DB[7:6] in Function Register Refering to DB[7:6] in Function Register −85 −83 −80 dBc/Hz typ dBc/Hz typ dBc/Hz typ PFD = 10 MHz, 5 kHz offset, IVCO = 2 mA PFD = 10 MHz, 5 kHz offset, IVCO = 2 mA PFD = 10 MHz, 5 kHz offset, IVCO = 3 mA 915MHz Phase Noise (Out of Band)1 315MHz 433MHz −80 dBc/Hz typ PFD = 10 MHz, 5 kHz offset, IVCO = 3 mA −103 −104 dBc/Hz typ dBc/Hz typ PFD = 10 MHz, 1 MHz offset, IVCO = 2 mA PFD = 10 MHz, 1 MHz offset, IVCO = 2 mA 868MHz 915MHz Harmonic Content (Second)2 Harmonic Content (Third)2 −115 −114 −20 −30 dBc/Hz typ dBc/Hz typ dBc typ dBc typ PFD = 10 MHz, 1 MHz offset, IVCO = 3 mA PFD = 10 MHz, 1 MHz offset, IVCO = 3 mA FRF = FVCO Harmonic Content (Others)2 Harmonic Content (Second)2 Harmonic Content (Third)2 −27 −24 −14 dBc typ dBc typ dBc typ Harmonic Content (Others)2 −19 dBc typ LOGIC INPUTS Input High Voltage,VINH 0.7 × DVDD V min Input Low Voltage, VINL Input Current, IINH/IINL Input Capacitance, CIN 0.2 × DVDD ±1 4.0 V max µA max pF max 915MHz VCO Tuning Range Spurious (IVCO Min/Max) Charge Pump Current Setting [00] Setting [01] Setting [10] Setting [11] Phase Noise (In band)1 315MHz 433MHz 868MHz LOGIC OUTPUTS Output High Voltage, VOH Output High Current, IOH, Output Low Voltage, VOL 1 2 FRF = FVCO/N (where N = 2, 4, 8) DVDD − 0.4 V min CMOS output chosen 500 0.4 µA max V max IOL = 500 µA Measurements made with NFRAC = 2048. Measurements made without harmonic filter. Rev. 0 | Page 4 of 28 ADF7012 TIMING CHARACTERISTICS DVDD = 3 V ± 10%; AGND = DGND = 0 V; TA = TMIN to TMAX, unless otherwise noted. Table 2. Parameter t1 t2 t3 t4 t5 t6 t7 Limit at TMIN to TMAX (B Version) 20 10 10 25 25 10 20 Unit ns min ns min ns min ns min ns min ns min ns min Test Conditions/Comments LE setup time Data-to-clock setup time Data-to-clock hold time Clock high duration Clock low duration Clock –to-LE setup time LE pulse width t4 t5 CLOCK t2 DATA DB23 (MSB) t3 DB22 DB2 DB1 (CONTROL BIT C2) DB0 (LSB) (CONTROL BIT C1) t7 LE t1 04617-0-002 t6 LE Figure 2. Timing Diagram Rev. 0 | Page 5 of 28 ADF7012 ABSOLUTE MAXIMUM RATINGS TA = 25°C, unless otherwise noted. Table 3. Parameter DVDD to GND (GND = AGND = DGND = 0 V) Digital I/O Voltage to GND Analog I/O Voltage to GND Operating Temperature Range Maximum Junction Temperature TSSOP θJA Thermal Impedance Lead Temperature, Soldering Vapor Phase (60 sec) Infrared (15 sec) Rating −0.3 V to +3.9 V −0.3 V to DVDD + 0.3 V −0.3 V to DVDD + 0.3 V 150°C 150.4°C/W Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those listed in the operational sections of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. This device is a high performance RF integrated circuit with an ESD rating of 1 kV and it is ESD sensitive. Proper precautions should be taken for handling and assembly. 215°C 220°C TRANSISTOR COUNT 35819 (CMOS) ESD CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although this product features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. Rev. 0 | Page 6 of 28 ADF7012 PIN CONFIGURATION AND FUNCTION DESCRIPTIONS DVDD 1 24 CREG2 CREG1 2 23 RSET CPOUT 3 22 AGND 21 DVDD 20 RFOUT TxDATA 4 TxCLK 5 TSSOP ADF7012 19 RFGND TOP VIEW DGND 7 (Not to Scale) 18 VCOIN OSC1 8 17 CVCO OSC2 9 16 L2 CLKOUT 10 15 L1 CLK 11 14 CE DATA 12 13 LE 04617-0-003 MUXOUT 6 Figure 3. Table 4. Pin Functional Descriptions Pin No. 1 Mnemonic DVDD 2 CREG1 3 CPOUT 4 5 TxDATA TxCLK 6 MUXOUT 7 8 9 DGND OSC1 OSC2 10 CLKOUT 11 CLK 12 DATA 13 LE 14 CE 15 L1 16 17 L2 CVCO 18 VCOIN 19 20 RFGND RFOUT 21 DVDD 22 23 24 AGND RSET CREG2 Description Positive Supply for the Digital Circuitry. This must be between 2.3 V and 3.6 V. Decoupling capacitors to the analog ground plane should be placed as close as possible to this pin. A 2.2 µF capacitor should be added at CREG to reduce regulator noise and improve stability. A reduced capacitor improves regulator power-on time, but may cause higher spurious noise. Charge Pump Output. This output generates current pulses that are integrated in the loop filter. The integrated current changes the control voltage on the input to the VCO. Digital Data to Be Transmitted is inputted on this pin. GFSK and GOOK only. This clock output is used to synchronize microcontroller data to the TxDATA pin of the ADF7012. The clock is provided at the same frequency as the data rate. The microcontroller updates TxDATA on the falling edge of TxCLK. The rising edge of TxCLK is used to sample TxDATA at the midpoint of each bit. Provides the Lock_Detect Signal. This determines if the PLL is locked to the correct frequency and also monitors battery voltage. Other signals include Regulator_Ready, which indicates the status of the serial interface regulator. Ground for Digital Section. The reference crystal should be connected between this pin and OSC2. The reference crystal should be connected between this pin and OSC1. A TCXO reference may be used, by driving this pin with CMOS levels, and powering down the crystal oscillator bit in software. A divided-down version of the crystal reference with output driver. The digital clock output may be used to drive several other CMOS inputs, such as a microcontroller clock. The output has a 50:50 mark-space ratio. Serial Clock Input. This serial clock is used to clock in the serial data to the registers. The data is latched into the 32-bit shift register on the CLK rising edge. This input is a high impedance CMOS input. Serial Data Input. The serial data is loaded MSB first with the two LSBs being the control bits. This is a high impedance CMOS input. Load Enable, CMOS Input. When LE goes high, the data stored in the shift registers is loaded into one of the four latches, the latch being selected using the control bits. Chip Enable. Bringing CE low puts the ADF7012 into complete power-down, drawing < 1uA. Register values are lost when CE is low and the part must be reprogrammed once CE is brought high. Connected to external printed or discrete inductor. See Choosing the External Inductor Value for advice on the value of the inductor to be connected between L1 and L2. Connected to external printed or discrete inductor. A 220 nF capacitor should be tied between the CVCO and CREG2 pins. This line should run underneath the ADF7012. This capacitor is necessary to ensure stable VCO operation. The tuning voltage on this pin determines the output frequency of the voltage controlled oscillator (VCO). The higher the tuning voltage, the higher the output frequency. Ground for Output Stage of Transmitter. The modulated signal is available at this pin. Output power levels are from –16 dBm to +12 dBm. The output should be impedance matched using suitable components to the desired load. See the PA Matching section. Voltage supply for VCO and PA section. This should have the same supply as DVDD Pin 1, and should be between 2.3 V and 3.6 V. Place decoupling capacitors to the analog ground plane as close as possible to this pin. Ground Pin for the RF Analog Circuitry. External Resistor to set charge pump current and some internal bias currents. Use 3.6 kV as default. Add a 470 nF capacitor at CREG to reduce regulator noise and improve stability. A reduced capacitor improves regulator power-on time and phase noise, but may have stability issues over the supply and temperature. Rev. 0 | Page 7 of 28 ADF7012 TYPICAL PERFORMANCE CHARACTERISTICS 315 MHZ –60 –70 5 0 –80 A 3 –20 –30 –100 –110 –50 –120 –60 –70 10.0k 100.0k PHASE NOISE (Hz) 1.0M 1 [T1] 2 [T1] –90 –95 10.0M CENTER 3.5MHz Figure 4. Phase Noise Response—DVDD = 3.0 V, ICP = 0.86 mA IVCO = 2.0 mA, FOUT = 315 MHz, PFD = 3.6864 MHZ, PA Bias = 5.5 mA REF LVL 5dBm 5 0 1 2 5 0 –10 –10 –20 –20 1MA –40 –50 –50 –60 –60 –70 –70 1 A SGL 1MA 3 04617-0-005 CENTER 315MHz 50kHz/ 4 –80 2 [T1] –90 –95 SPAN 500kHz 5 0 REF LVL 5dBm 4 [T1] –42.93dBm 939.87975952MHz –55.48dBm 1.26252505GHz SPAN 7GHz 700MHz/ RBW 20.33dBm VBW 26.55310621kHz SWT 5 0 A –10 5kHz RF ATT 30dB 5kHz dBm 500ms UNIT A 1 –10 –20 –20 1MA D1 –41.5dBm –40 D2 –49dBm –50 –60 –60 –70 –70 1 [T1] 04617-0-006 –80 0.31dBm 315.40080160MHz CENTER 315MHz 40MHz/ 2 3 –30 –80 –90 –95 SPAN 400MHz 1MA 1 [T1] 3 [T1] 2 [T1] –3.49dBm 315.00012525MHz –20.33dB 26.55310621kHz –20.85dB –27.55511022kHz CENTER 315MHz Figure 6. Spurious Components—Meets FCC Specs 50kHz/ 04617-0-009 –30 –90 –95 3 [T1] Figure 8. Harmonic Response, Fifth-Order Butterworth Filter RBW 500kHz RF ATT 30dB 0.31dBm VBW 500kHz 315.40080160MHz SWT dBm 5ms UNIT 1 0.18dBm 308.61723447MHz –50.53dBm 631.26252505MHz CENTER 3.5MHz Figure 5. FSK Modulation, Power = 0 dBm, Data Rate = 1 kbps, FDEVIATION = ±50 kHz REF LVL 5dBm D1 –41.5dBm 2 1 [T1] –90 –95 SPAN 7GHz –30 –40 –80 700MHz/ –11.48dBm 939.87975952MHz –34.11dBm 1.26252505GHz RBW 1MHz RF ATT 30dB 0.18dBm VBW 1MHz 308.61723447MHz SWT 17.5ms UNIT dBm REF LVL 5dBm A –30 4 [T1] Figure 7. Harmonic Response, RFOUT Matched to 50 Ω, No Filter 5kHz RF ATT 30dB 5kHz dBm 500ms UNIT RBW 0.45dBm VBW 315.05060120MHz SWT 3 [T1] 0.27dBm 308.61723447MHz –35.43dBm 631.26252505MHz –80 04617-0-007 –140 1.0k D1 –41.5dBm 04617-0-008 –130 –50 1MA 4 2 –40 04617-0-004 dBc (Hz) 1 –10 –90 –40 RBW 1MHz RF ATT 30dB 0.27dBm VBW 1MHz 308.61723447MHz SWT 17.5ms UNIT dBm REF LVL 5dBm = NORMAL FREQUENCY = 9.08 kHz LEVEL = –84.47dBc/Hz SPAN 500kHz Figure 9. OOK Modulation, Power = 0 dBm, Data Rate = 10 kbps Rev. 0 | Page 8 of 28 ADF7012 433 MHZ 1 2.00V/ 2 1.00V/ 1.50ms 500µs TRIG'D 1 720mv RBW 10.01dBm VBW 433.91158317MHz SWT REF LVL 15dBm 15 10 30kHz RF ATT 40dB 30kHz dBm 90ms UNIT 1 A 0 –10 –20 2 CLKOUT 1MA –30 D1 –36dBm –40 –50 –60 04617-0-010 –70 04617-0-013 1 CE –80 –85 CENTER 433.9500601MHz 3.2MHz/ Figure 10. Crystal Power-On Time, 4 MHz, Time = 1.6 ms Figure 13. Spurious Components—Meets ETSI Specs –40 15 10 –80 1 A 0 3 –10 –100 2 4 –20 1MA –120 –30 –140 –40 –160 –50 10.0k 100.0k PHASE NOISE (Hz) 1.0M 1 [T1] 2 [T1] –80 –85 10.0M CENTER 3.5GHz Figure 11. Phase Noise Response—ICP = 2.0 mA, IVCO = 2.0 mA, RFOUT = 433.92 MHz, PFD = 4 MHz, PA Bias = 5.5 mA REF LVL 15dBm 15 10 RBW 10kHz RF ATT 40dB 5.60dBm VBW 300kHz 433.91158317MHz SWT dBm 44ms UNIT 15 10 0 0 –10 –20 –30 –30 D1 –36dBm 1 A SGL 1MA 2 –50 –60 –60 –70 –70 04617-0-012 1 [T1] 174kHz/ 2 [T1] –80 –85 STOP 434.79kHz D1 –30dBm 3 –40 –50 START 433.05MHz SPAN 7GHz –20 1MA –80 –85 700MHz/ –5.12dBm 1.30460922GHz –17.57dBm 1.73947896GHz RBW 1MHz RF ATT 40dB 9.51dBm VBW 1MHz 434.86973948MHz SWT 17.5ms UNIT dBm REF LVL 15dBm A 1 3 [T1] 4 [T1] Figure 14. Harmonic Response, RFOUT Matched to 50 Ω, No Filter –10 –40 10.10dBm 434.86973948MHz –15.25dBm 869.73947896MHz –70 04617-0-014 –200 1.0k D1 –36dBm 9.51dBm 434.86973948MHz –33.75dBm 869.73947896MHz CENTER 3.5GHz Figure 12. FSK Modulation, Power = 10 dBm, Data Rate = 38.4 kbps, FDEVIATION = ±19.28 kHz Rev. 0 | Page 9 of 28 D1 –36dBm 4 3 [T1] 4 [T1] 700MHz/ –43.60dBm 1.30460922GHz –43.44dBm 1.73947896GHz 04617-0-015 –180 D1 –30dBm –60 04617-0-011 dBc (Hz) RBW 1MHz RF ATT 40dB 10.10dBm VBW 1MHz 434.86973948MHz SWT 17.5ms UNIT dBm REF LVL 15dBm = NORMAL FREQUENCY = 393.38 kHz LEVEL = –102.34dBc/Hz –60 SPAN 32MHz SPAN 7GHz Figure 15. Harmonic Response, Fifth-Order Butterworth Filter ADF7012 868 MHZ 0 15 10 –40 dBc (Hz) –80 3 4 1MAX 1MA –30 –100 –40 –120 –50 –160 1.0k 10.0k 100.0k PHASE NOISE (Hz) 1.0M 1 [T1] –70 10.0M 15 10 12.27dBm 869.33867735MHz –4.00dBm 1.72865731GHz 2 [T1] –80 –85 CENTER 3.8GHz Figure 16. Phase Noise Response–ICP = 2.5 mA, IVCO = 1.44 mA, RFOUT = 868.95 MHz, PFD = 4.9152 MHz, Power = 12.5 dBm, PA Bias = Max REF LVL 15dBm D1 –30dBm –60 04617-0-016 –140 30dB RBW 10kHz RF ATT –40.44dBm VBW 10kHz MIXER –20dBm 869.20000000MHz SWT 15ms UNIT dBm 2 15 10 A LN 640MHz/ 1 1 [T1] 3 [T1] 2 [T1] –10 –20 1MAX 1MA –20 –30 –30 D2 –36dBm –40 1 –16.88dBm 2.59699399GHz –15.06dBm 3.46913828GHz SPAN 6.4GHz 30dB RBW 1kHz RF ATT 10.39dBm VBW 1kHz MIXER –20dBm 869.33867735MHz SWT 10ms UNIT dBm 0 –10 3 [T1] 4 [T1] Figure 19. Harmonic Response, RFOUT Matched to 50 Ω, No Filter REF LVL 15dBm 0 10.39dBm 869.33867735MHz –50.92dBm 1.72000000GHz –50.40dBm 2.59600000GHz 1MAX A LN 1MA D2 –30dBm –40 –50 –50 –60 2 3 –60 2 [T1] –80 –85 CENTER 868.944489MHz –70 60kHz/ –80 –85 SPAN 600kHz START 3.8GHz Figure 17. FSK Modulation, Power = 12.5 dBm, Data Rate = 38.4 kbps, FDEVIATION = ±19.2 kHz REF LVL 15dBm RBW 12.55dBm VBW 869.025050100MHz SWT 15 10 1 1 [T1] 0 2 [T1] –10 3 [T1] 30dB 2kHz RF ATT 2kHz MIXER –20dBm dBm 16s UNIT 12.55dBm 869.02505010MHz –57.89dBm 859.16695500MHz –81.97dBm 862.00000000MHz –20 1MAX A LN 1MA –30 D2 –36dBm –40 2 D1 –54dBm –60 04617-0-018 –70 3 START 856.5MHz 2.5MHz/ 04617-0-020 –70 40.44dBm 869.20000000MHz 8.02dBm 868.96673347MHz 04617-0-017 1 [T1] –80 –85 A 2 0 –20 –50 1 –10 –60 1MHz RF ATT 40dB 1MHz dBm 16ms UNIT 04617-0-019 –20 RBW 12.27dBm VBW 869.33867735MHz SWT REF LVL 15dBm = NORMAL FREQUENCY = 251.3 kHz LEVEL = –99.39dBc/Hz STOP 881.5MHz Figure 18. Spurious Components—Meets ETSI Specs Rev. 0 | Page 10 of 28 640MHz/ SPAN 6.4GHz Figure 20. Harmonic Response, Fifth-Order Chebyshev Filter ADF7012 915 MHZ RBW 50MHz RF ATT 40dB 10.25dBm VBW 50MHz 907.81563126MHz SWT dBm 6.4s UNIT –40 REF LVL 15dBm = NORMAL FREQUENCY = 992.38 kHz LEVEL = –102.34dBc/Hz –60 15 10 –80 0 2 –10 –100 –20 3 1MAX 4 1MA –120 –30 –140 –40 –160 D1 –41.5dBm 04617-0-021 –50 –200 1.0k 10.0k 100.0k PHASE NOISE (Hz) 1.0M –60 1 [T1] –70 10.0M 2 [T1] –80 –85 10.25dBm 907.81563126MHz –10.06dBm 1.83126253GHz CENTER 3.8GHz Figure 21. Phase Noise Response–ICP = 1.44 mA, IVCO = 3.0 mA, RFOUT = 915.2 MHz, PFD =10 MHz, Power = 10 dBm, PA Bias = 5.5 mA REF LVL 15dBm 15 10 RBW 10kHz RF ATT 40dB 3.88dBm VBW 300kHz 915.19098196MHz SWT 15ms UNIT dBm 15 10 0 1MA –20 –30 –30 –40 –40 –50 –50 RBW 50MHz RF ATT 40dB 9.06dBm VBW 50MHz 907.81563126MHz SWT dBm 6.4s UNIT A 1 –60 –60 –70 –70 1MAX 1MA 2 04617-0-036 1 [T1] –80 –85 CENTER 915.190982MHz 50kHz/ 2 [T1] –80 –85 REF LVL 15dBm * A SGL 0 –10 1MA –20 –30 D1 –41.5dBm D1 –49.5dBm –60 04617-0-037 –70 –80 –85 CENTER 915.2MHz 40MHz/ D1 –41.5dBm 4 3 [T1] 4 [T1] 640MHz/ –46.22dBm 2.74188377GHz –46.96dBm 3.65250501GHz SPAN 6.4GHz Figure 25. Harmonic Response, Fifth-Order Chebyshev Filter RBW 10kHz RF ATT 40dB 9.94dBm VBW 300kHz 915.23167977MHz SWT 100ms UNIT dBm 1 3 9.06dBm 907.81563126MHz –48.40dBm 1.83126253GHz CENTER 3.8GHz SPAN 500kHz Figure 22. FSK Modulation, Power = 10 dBm, Data Rate = 38.4 kbps, Fdeviation = ±19.2 kHz –50 SPAN 6.4GHz –10 –20 1MAX –40 640MHz/ 0 –10 15 10 –20.29dBm 2.74188377GHz –17.50dBm 3.65250501GHz Figure 24. Harmonic Response, RFOUT Matched to 50 Ω, No Filter REF LVL 15dBm A 1 3 [T1] 4 [T1] 04617-0-038 –180 04617-0-039 dBc (Hz) A 1 SPAN 400MHz Figure 23. Spurious Components—Meets FCC Specs Rev. 0 | Page 11 of 28 ADF7012 CIRCUIT DESCRIPTION PLL OPERATION CRYSTAL/R LOOP FILTER R PFD VCO FVCO CP2 Figure 27. Two parallel resonant capacitors are required for oscillation at the correct frequency—the value of these depend on the crystal specification. They should be chosen so that the series value of capacitance added to the PCB track capacitance adds to give the load capacitance of the crystal, usually 20 pF. Track capacitance values vary between 2 pF to 5 pF, depending on board layout. Where possible, to ensure stable frequency operation over all conditions, capacitors should be chosen so that they have a very low temperature coefficient and/or opposite temperature coefficients 04617-0-022 FADJUST = FSTEP × FEC Figure 26. FCRYSTAL × N = FPFD × N R CP1 The ADF7012 features a 15-bit fixed modulus, which allows the output frequency to be adjusted in steps of FPFD/15. This fine resolution can be used to easily compensate for initial error and temperature drift in the reference crystal. VCO/N FOUT = OSC2 CRYSTAL COMPENSATION REGISTER CP N OSC1 04617-0-023 A fractional-N PLL allows multiple output frequencies to be generated from a single-reference oscillator (usually a crystal) simply by changing the programmable N value found in the N register. At the phase frequency detector (PFD), the reference is compared to a divided-down version of the output frequency (VCO/N). If VCO/N is too low a frequency, typically the output frequency is lower than desired, and the PFD and charge-pump combination sends additional current pulses to the loop filter. This increases the voltage applied to the input of the VCO. Because the VCO of the ADF7012 has a positive frequency vs. voltage characteristic, any increase in the Vtune voltage applied to the VCO input increases the output frequency at a rate of kV, the tuning sensitivity of the VCO (MHz/V). At each interval of 1/PFD seconds, a comparison is made at the PFD until the PFD and charge pump eventually force a state of equilibrium in the PLL where PFD frequency = VCO/N. At this point, the PLL can be described as locked. (1) (3) where FSTEP = FPFD/215 and FEC = Bits F1 to F11 in the R Register. Note that the notation is twos compliment, so F11 represents the sign of the FEC number. where NFRAC can be bits M1 to M12 in the fractional N register. Example FPFD = 10 MHz FADJUST = −11 kHz FSTEP = 10 MHz/215 = 305.176 Hz FEC = −11 kHz/305.17 Hz = −36 = −(00000100100) = 11111011100 = 0x7DC CRYSTAL OSCILLATOR CLOCK OUT CIRCUIT The on-board crystal oscillator circuitry (Figure 27) allows an inexpensive quartz crystal to be used as the PLL reference. The oscillator circuit is enabled by setting XOEB low. It is enabled by default on power-up and is disabled by bringing CE low. Errors in the crystal can be corrected using the error correction register within the R register. The clock out circuit takes the reference clock signal from the oscillator section above and supplies a divided-down 50:50 mark-space signal to the CLKOUT pin. An even divide from 2 to 30 is available. This divide is set by the DB[19:22] in the R register. On power-up, the CLKOUT defaults to divide by 16. N ⎛ ⎞ FOUT = FPFD × ⎜ N INT + FRAC 12 ⎟ 2 ⎠ ⎝ (2) DVDD CLKOUT ENABLE BIT A single-ended reference may be used instead of a crystal, by applying a square wave to the OSC2 pin, with XOEB set high. OSC1 DIVIDER 1 TO 15 ÷2 Figure 28. Rev. 0 | Page 12 of 28 CLKOUT 04617-0-024 For a Fractional N PLL ADF7012 The output buffer to CLKOUT is enabled by setting Bit DB4 in the function register high. On power-up, this bit is set high. The output buffer can drive up to a 20 pF load with a 10% rise time at 4.8 MHz. Faster edges can result in some spurious feedthrough to the output. A small series resistor (50 Ω) can be used to slow the clock edges to reduce these spurs at FCLK. LOOP FILTER The loop filter integrates the current pulses from the charge pump to form a voltage that tunes the output of the VCO to the desired frequency. It also attenuates spurious levels generated by the PLL. A typical loop filter design is shown in Figure 29. CHARGE PUMP OUT VCO 04617-0-025 In FSK, the loop should be designed so that the loop bandwidth (LBW) is a minimum of two to three times the data rate. Widening the LBW excessively reduces the time spent jumping between frequencies, but results in reduced spurious attenuation. See the section Tips on Designing the Loop Filter. For OOK/ASK systems, a wider loop bandwidth than for FSK systems is desirable. The sudden large transition between two power levels results in VCO pulling (VCO temporarily goes to incorrect frequency) and can cause a wider output spectrum. By widening the loop bandwidth a minimum of 10 × data rate, VCO pulling is minimized because the loop settles quickly back to the correct frequency. The free design tool ADIsimPLL™ can be used to design loop filters for the ADI family of transmitters. VOLTAGE-CONTROLLED OSCILLATOR (VCO) The ADF7012 features an on-chip VCO with an external tank inductor, which is used to set the frequency range. The center frequency of oscillation is governed by the internal varactor capacitance and that of the external inductor combined with the bond-wire inductance. An approximation for this is given in the Equation 4. For a more accurate selection of the inductor, see the section Choosing the External Inductor Value. 1 2π ( LINT + LEXT ) × (CVAR + CFIXED ) VCO Bias Current VCO bias current may be adjusted using bits VB1 to VB4 in the function register. Additional bias current will reduce spurious levels, but increase overall current consumption in the part. A bias value of 0x5 should ensure oscillation at most frequencies and supplies. Settings 0x0, 0xE ,and 0xF are not recommended. Setting 0x3 and Setting 0x4 are recommended under most conditions. Improved phase noise can be achieved for lower bias currents. VOLTAGE REGULATORS Figure 29. FVCO = Figure 32 shows the VCO gain over temperature and frequency. VCO gain is important in determining the loop filter design— predictable changes in VCO gain resulting in a change in the loop filter bandwidth can be offset by changing the chargepump current in software. (4) The varactor capacitance can be adjusted in software to increase the effective VCO range by writing to bits VA1 and VA2 in the R register. Under typical conditions, setting VA1 and VA2 high increases the center frequency by reducing the varactor capacitance by approximately 1.3 pF. There are two band gap voltage regulators on the ADF7012 providing a stable 2.25 V internal supply: a 2.2 µF capacitor (X5R, NP0) to ground at CREG1 and a 470 nF capacitor at CREG2 should be used to ensure stability. The internal reference ensures consistent performance over all supplies and reduces the current consumption of each of the blocks. The combination of regulators, band gap reference, and biasing typically consume 1.045 mA at 3.0 V and can be powered down by bringing the CE line low. The serial interface is supplied by Regulator 1, so powering down the CE line causes the contents of the registers to be lost. The CE line must be high and the regulators must be fully powered on to write to the serial interface. Regulator power-on time is typically 100 µs and should be taken into account when writing to the ADF7012 after power-up. Alternatively, regulator status may be monitored at the MUXOUT pin once CE has been asserted, because MUXOUT defaults to the regulator ready signal. Once Regulator_ready is high, the regulator is powered up and the serial interface is active. FSK MODULATION FSK modulation is performed internally in the PLL loop by switching the value of the N register based on the status of the TxDATA line. The TxDATA line is sampled at each cycle of the PFD block (every 1/FPFD seconds). When TxDATA makes a lowto-high transition, an N value representing the deviation frequency is added to the N value representing the center frequency. Immediately the loop begins to lock to the new frequency of FCENTER + FDEVIATION. Conversely, when TxDATA makes a high-to-low transition, the N value representing the deviation is subtracted from the PLL N value representing the center frequency and the loop transitions to FCENTER − FDEVIATION. Rev. 0 | Page 13 of 28 ADF7012 PFD/ CHARGE PUMP PA STAGE VCO FSK DEVIATION FREQUENCY –FDEV THIRD-ORDER Σ-∆ MODULATOR +FDEV TxDATA FRACTIONAL-N INTEGER-N 04617-0-026 ÷N Figure 30. The deviation from the center frequency is set using bits D1 to D9 in the modulation register. The frequency deviation may be set in steps of FSTEP ( Hz) = FPFD 214 FPFD × ModulationNumber 214 TxDATA INT TxCLK ADF7012 FETCH FETCH FETCH SAMPLE SAMPLE FETCH SAMPLE Figure 31. TxCLK/TxDATA Synchronization. (5) The number of steps between symbol ‘0’ and symbol ‘1’ is determined by the setting for the index counter. The deviation frequency is therefore FDEVIATION ( Hz) = I/O µC 04617-0-040 4R For GFSK and GOOK, the incoming bit stream to be transmitted needs to be synchronized with an on-chip sampling clock which provides one sample per bit to the Gaussian FIR filter. To facilitate this, the sampling clock is routed to the TxCLK pin where data is fetched from the host microcontroller or microprocessor on the falling edge of TxCLK, and the data is sampled at the midpoint of each bit on TxCLK’s rising edge. Inserting external RC LPFs on TxDATA and TxCLK lines creates smoother edge transitions and improves spurious performance. As an example, suitable components would be a 1 kV resistor and 10 nF capacitor for a data rate of 5 kbps. (6) The GFSK deviation is set up as GFSK DEVIATION ( Hz) = where ModulationNumber is set by bits D1 to D9. FPFD × 2 m 212 (7) The maximum data rate is a function of the PLL lock time (and the requirement on FSK spectrum). Because the PLL lock time is reduced by increasing the loop-filter bandwidth, highest data rates can be achieved for the wider loop filter bandwidths. The absolute maximum limit on loop filter bandwidth to ensure stability for a fractional-N PLL is FPFD/7. For a 20 MHz PFD frequency, the loop bandwidth could be as high as 2.85 MHz. FSK modulation is selected by setting bits S1 and S2 in the modulation register low. where DividerFactor can be bits D1 to D7, and IndexCounter can be bits IC1 and IC2 in the modulation register. GFSK MODULATION POWER AMPLIFIER Gaussian Frequency Shift Keying, or GFSK, represents a filtered form of frequency shift keying. The data to be modulated to RF is prefiltered digitally using an finite impulse response filter (FIR). The filtered data is then used to modulate the sigmadelta fractional-N to generate spectrally-efficient FSK. The output stage is based on a Class E amplifier design, with an open drain output switched by the VCO signal. The output control consists of six current mirrors operating as a programmable current source. FSK consists of a series of sharp transitions in frequency as the data is switched from one level to an other. The sharp switching generates higher frequency components at the output, resulting in a wider output spectrum. With GFSK, the sharp transitions are replaced with up to 128 smaller steps. The result is a gradual change in frequency. As a result, the higher frequency components are reduced and the spectrum occupied is reduced significantly. GFSK does require some additional design work as the data is only sampled once per bit, and so the choice of crystal is important to ensure the correct sampling clock is generated. where m is the mod control (Bits MC1 to MC3 in the modulation register). The GFSK sampling clock samples data at the data rate: DataRate (bps) = FPFD DividerFactor × IndexCounter (8) To achieve maximum voltage swing, the RFOUT pin needs to be biased at DVDD. A single pull-up inductor to DVDD ensures a current supply to the output stage, PA biased to DVDD volts, and with the correct choice of value transforms the impedance. The output power can be adjusted by changing the value of bits P1 to P6. Typically, this is P1 to P6 output −20dBm at 0x0, and 13 dBm at 0x7E at 868MHz, with the optimum matching network. Rev. 0 | Page 14 of 28 ADF7012 The nonlinear characteristic of the output stage results in an output spectrum containing harmonics of the fundamental, especially the third and fifth. To meet local regulations, a lowpass filter usually is required to filter these harmonics. As is the case with GFSK, GOOK requires the bit stream applied at TxDATA to be synchronized with the sampling clock, TxCLK (see the GFSK Modulation section). 10 The output stage can be powered down by setting Bit PD2 in the function register low. 0 –10 OOK GOOK MODULATION –60 –70 –80 909.43 910.43 FREQUENCY (MHz) 10 0 –10 PRE-FILTER DATA (0 TO 1 TRANSITION) –20 –30 OOK –40 –50 –60 –70 –80 –90 885.43 GOOK 910.43 FREQUENCY (MHz) 04617-0-041 Figure 34. GOOK vs. OOK Frequency Spectra (Wideband Measurement) Figure 32. Varying PA Output for GOOK (Index Counter = 16). Rev. 0 | Page 15 of 28 04617-0-044 Figure 32 shows the step response of the Gaussian FIR filter. An index counter of 16 is demonstrated for simplicity. While the pre-filter data would switch the PA directly from off to on with a low-to-high data transition, the filtered data gradually increases the PA output in discrete steps. This has the effect of making the output spectrum more compact. DISCRETIZED FILTER OUTPUT 910.93 20 Bits D1 to D6 represent the output power for the system for a positive data bit. Divider Factor = 0x3F represents the maximum possible deviation from PA at minimum to PA at maximum output. An index counter setting of 128 is recommended. PA SETTING 16 (MAX) 15 14 13 12 11 10 9 8 7 6 5 4 3 2 1 (PA OFF) GOOK Figure 33. GOOK vs. OOK Frequency Spectra (Narrow-Band Measurement) (9) POWER (dBm) FPFD DividerFactor × IndexCounter –40 –50 The GOOK sampling clock samples data at the data rate: DataRate (bps) = –30 04617-0-043 Gaussian on-off keying (GOOK) represents a prefiltered form of OOK modulation. The usually sharp symbol transitions are replaced with smooth Gaussian-filtered transitions with the result being a reduction in frequency pulling of the VCO. Frequency pulling of the VCO in OOK mode can lead to a wider than desired bandwidth, especially if it is not possible to increase the loop filter bandwidth to > 300kHz. POWER (dBm) –20 935.93 ADF7012 OUTPUT DIVIDER Battery Voltage Read back An output divider is a programmable divider following the VCO in the PLL loop. It is useful when using the ADF7012 to generate frequencies of < 500 MHz. By setting MUXOUT to 1010 to 1101, the battery voltage can be estimated. The battery measuring circuit features a voltage divider and a comparator where the divided-down supply voltage is compared to the regulator voltage. PFD CP LOOP FILTER VCO OUTPUT DIVIDER PA ÷1/2/4/8 ÷N Table 6. 04617-0-042 REFERENCE DIVIDER Figure 35. Output Divider Location in PLL. The output divider may be used to reduce feedthrough of the VCO by amplifying only the VCO/2 component, restricting the VCO feedthrough to leakage. Because the divider is in loop, the N register values should be set up according to the usual formula. However, the VCO gain (KV) should be scaled according to the divider setting, as shown in the following example. Fout = 433 MHz, Fvco = 866 MHz, KV @ 868 MHz = 60 MHz/V The divider value is set in the R register. Table 5. OD2 0 1 0 1 MUXOUT High DVDD > 3.25 V DVDD > 3.0 V DVDD > 2.75 V DVDD > 2.35 V MUXOUT Low DVDD < 3.25 V DVDD < 3.0 V DVDD < 2.75 V DVDD < 2.35 V The accuracy of the measurement is limited by the accuracy of the regulator voltage and also the internal resistor tolerances. Regulator Ready The regulator has a power-up time, dependant on process and the external capacitor. The regulator ready signal indicates that the regulator is fully powered, and that the serial interface is active. This is the default setting on power-up at MUXOUT. Digital Lock Detect Therefore, KV for loop filter design = 30 MHz/V. OD1 0 0 1 1 MUXOUT 1010 1011 1100 1101 Divider Status Divider off Divide by 2 Divide by 4 Divide by 8 MUXOUT MODES The MUXOUT pin allows the user access to various internal signals in the transmitter, and provides information on the PLL lock status, the regulator, and the battery voltage. The MUXOUT is accessed by programming Bits M1 to M4 in the function register and observing the signal at the MUXOUT pin. Digital lock detect indicates that the status of the PLL loop. The PLL loop takes time to settle on power-up and when the frequency of the loop is changed by changing the N value. When lock detect is high, the PFD has counted a number of consecutive cycles where the phase error is < 15 ns. The lock detect precision bit in the function register determines whether this is 3 cycles (LDP = 0), or 5 cycles (LDP=1). It is recommended that LDP be set to 1. The lock detect is not completely accurate and goes high before the output has settled to exactly the correct frequency. In general, add 50% to the indicated lock time to obtain lock time to within 1 kHz. The lock detect signal can be used to decide when the power amplifier (PA) should be enabled. R Divider MUXOUT provides the output of the R divider. This is a narrow pulsed digital signal at frequency FPFD. This signal may be used to check the operation of the crystal circuit and the R divider. R divider/2 is a buffered version of this signal at FPFD/2. Rev. 0 | Page 16 of 28 ADF7012 THEORY OF OPERATION CHOOSING THE EXTERNAL INDUCTOR VALUE Standard Crystal Values The ADF7012 allows operation at many different frequencies by choosing the external VCO inductor to give the correct output frequency. Figure 36 shows both the minimum and maximum frequency vs. the inductor value. These are measurements based on 0603 CS type inductors from Coilcraft, and are intended as guidelines in choosing the inductor because board layout and inductor type varies between applications. Standard crystal values are 3.6864 MHz, 4 MHz, 4.096 MHz, 4.9152 MHz, 7.3728 MHz, 9.8304 MHz, 10 MHz, 11.0592 MHz, 12 MHz, and 14.4792 MHz. Crystals with these values are usually available in stock and cost less than crystals with nonstandard values. The inductor value should be chosen so it is between the minimum and maximum value. 1200 MIN (meas) 1100 MAX (meas) MIN (eqn) MAX (eqn) Reference spurious levels (spurs) occur at multiples of the PFD frequency. The reference spur closest to the carrier is usually highest with the spur further out being attenuated by the loop filter. The level of reference spur is lower for lower PFD frequencies. In designs with high output power where spurious levels are the main concern, a lower PFD frequency (<5 MHz) may be desirable. Beat Note Spurs 900 500 These are spurs occurring for very small or very large values in the fractional register. These are quickly attenuated by the loop filter. Selection of the PFD therefore determines their location, and ensures that they have negligible effect on the transmitter spectrum. 400 Phase Noise 800 700 600 04617-0-031 FREQUENCY (MHz) 1000 Reference Spurious Levels 300 0 5 10 15 20 INDUCTANCE (nH) 25 30 The phase noise of a frequency synthesizer improves by 3dB for every doubling of the PFD frequency. Because ACP is related to the phase noise, the PFD may be increased to reduce the ACP in the system. PFD frequencies of < 5MHz typically deliver sufficient phase noise performance for most systems. 35 Figure 36. Output Frequency vs. External Inductor Value Ibias = 2.0 mA. For frequencies between 270 MHz and 550 MHz, it is recommended to operate the VCO at twice the desired output frequency and use the divide-by-2 option. This ensures reliable operation over temperature and supply. For frequencies between 130 MHz and 270 MHz, it is recommended to operate the VCO at four times the desired output frequency and use the divide-by-4 option. Deviation Frequency The deviation frequency is adjustable in steps of FSTEP ( Hz) = FPFD 214 (10) To get the exact deviation frequency required, ensure FSTEP is a factor of the desired deviation. For frequencies below 130 MHz, it is best to use the divide-by-8 option. It is not necessary to use the VCO divider for frequencies above 550 MHz. ADIsimPLL is a PLL design tool which can perform the frequency calculations for the ADF7012, and is available at www.analog.com/pll. CHOOSING THE CRYSTAL/PFD VALUE The choice of crystal value is an important one. The PFD frequency must be the same as the crystal value or an integer division of it. The PFD determines the phase noise, spurious levels and location, deviation frequency, and the data rate in the case of GFSK. The following sections describe some factors that should be considered when choosing the crystal value. TIPS ON DESIGNING THE LOOP FILTER The loop filter design is crucial in ensuring stable operation of the transmitter, meeting Adjacent Channel Power (ACP) specifications, and meeting spurious requirements for the relevant regulations. ADIsimPLL is a free tool available to aid the design of loop filters. The user enters the desired frequency range, the reference crystal and PFD values, and the desired loop bandwidth. ADIsimPLL gives a good starting point for the filter, and the filter can be further optimized based on the criteria below. Rev. 0 | Page 17 of 28 ADF7012 Setting Tuning Sensitivity Value PA MATCHING The tuning sensitivity or kV is usually denoted in MHz/V and is required for the loop filter design. It refers to the amount that a change of a volt in the voltage applied to VCOIN pin, changes the output frequency. Typical data for the ADF7012 over a frequency range is shown. The ADF7012 exhibits optimum performance in terms of transmit power and current consumption only if the RF output port is properly matched to the antenna impedance. 120 100 KV (MHz/V) 80 ZOPT_PA depends primarily on the required output power, and the frequency range. Selecting the optimum ZOPT_PA helps to minimize the current consumption. This data sheet contains a number of matching networks for common frequency bands. Under certain conditions it is recommended to obtain a suitable ZOPT_PA value by means of a load-pull measurement. 60 DVDD 40 400 500 600 700 800 FREQUENCY (MHz) 900 1000 LPF ZOPT_PA 1100 Figure 37. kV vs. VCO Frequency Figure 38. ADF7012 with Harmonic Filter Charge-Pump Current The charge-pump current allows the loop filter bandwidth to be changed using the registers. The loop bandwidth reduces as the charge pump current is reduced and vice versa. Selecting Loop Filter Bandwidth Data Rate The loop filter bandwidth should usually be at two to three times the data rate. This ensures that the PLL has ample time to jump between the mark and space frequencies. The impedance matching values provided in the next section are for 50 Ω environments. An additional matching network may be required after the harmonic filter to match to the antenna impedance. This can be incorporated into the filter design itself in order to reduce external components. TRANSMIT PROTOCOL AND CODING CONSIDERATIONS PREAMBLE ACP In the case where the ACP specifications are difficult to meet, the loop filter bandwidth can be reduced further to reduce the phase noise at the adjacent channel. The filter rolls off at 20 dB per decade. Spurious Levels In the case where the output power is quite high, a reduced loop filter bandwidth reduces the spurious levels even further, and provides additional margin on the specification. SYNC WORD ID FIELD DATA FIELD CRC 04617-0-034 300 PA ANTENNA 04716-0-033 0 200 RFOUT 004617-0-032 20 Figure 39. Typical Format of a Transmit Protocol A dc-free preamble pattern such as 10101010… is recommended for FSK/ASK/OOK demodulation. Preamble patterns with longer run-length constraints such as 11001100…. can also be used. However, this can result in a longer synchronization time of the received bit stream in the chosen receiver. The following sections provide examples of loop filter designs for typical applications in specific frequencies. Rev. 0 | Page 18 of 28 ADF7012 APPLICATION EXAMPLES R2 C1 C2 R1 VDD C5+ 10µF C5 2.2µF VDD U1 ADF7012 C5 100pF 1 C6 2.2µF CREG1 2 3 1kΩ TxDATA TxDATA TxCLK 1kΩ TxCLK 1kΩ MUXOUT 6 MUXOUT Y1 C8 27pF OSC1 C9 1kΩ CLKOUT 9 PIN D-TYPE PLUG J3–3 J3–5 J3–7 R3 1kΩ CLK R4 1kΩ DATA R5 1kΩ LE CLK DATA 8 OSC2 9 CLKOUT 10 CREG2 CREG1 RSET CPOUT AGND DVDD TxDATA TxCLK RFOUT MUXOUT RFGND DGND VCOIN OSC1 CVCO OSC2 L2 CLKOUT L1 CLK CE DATA LE C10 470nF C11 0.22µF 23 R9 3.6kΩ 22 C12 100pF C13 2.2µF VDD 21 L1 20 C14 L2 C15 19 LF1 CF1 18 17 16 L3 11 12 15 14 VDD 13 CE R6 1kΩ R7 1kΩ R8 1kΩ TxDATA J5–1 J5–2 TxCLK MUXOUT J5–3 J5–4 CLKOUT CLK J5–5 J5–5 DATA LE J5–7 J5–8 CE VDD J5–9 J5–10 10 PIN HEADER (5X2) Figure 40. Applications Diagrams with Harmonic Filter Rev. 0 | Page 19 of 28 J4 LF2 CF2 LE J3–6 J3–8 5 7 C7 27pF J2 4 DVDD CREG2 24 04617-0-035 J1 C3 CF2 ADF7012 315 MHZ OPERATION The recommendations here are guidelines only. The design should be subject to internal testing prior to FCC site testing. Matching components need to be adjusted for board layout. The FCC standard 15.231 regulates operation in the band from 260MHz to 470MHz in the US. This is used generally in the transmission of RF control signals, such as in a satellitedecoder remote control, or remote keyless entry system. The band cannot be used to send any continuous signal. The maximum output power allowed is governed by the duty cycle of the system. A typical design example for a remote control is shown next. Design Criteria 315 MHz center frequency FSK/OOK modulation 1 mW output power House range Meets FCC 15.231 Bias Current Because low current is desired, a 2.0 mA VCO bias can be used. Additional bias current reduces any spur, but increases current consumption. The PA bias can be set to 5.5 mA and achieve 0 dBm. Loop Filter Bandwidth The loop filter is designed with ADIsimPLL Version 2.5. The loop bandwidth design is straightforward because the 20 dB bandwidth is generally of the order of >400 kHz (0.25% of center frequency). A loop bandwidth of close to 100 kHz strikes a good balance between lock time and spurious suppression. If it is found that pulling of the VCO is more than desired in OOK mode, the bandwidth could be increased. Design of Harmonic Filter The main requirements in the design of this remote are a long battery life and sufficient range. It is possible to adjust the output power of the ADF7012 to increase the range depending on the antenna performance. The center frequency is 315 MHz. Because the ADF7012 VCO is not recommended for operation in fundamental mode for frequencies below 400 MHz, the VCO needs to operate at 630 MHz. Figure 36 (Output Frequency vs. External Inductor Value) implies an inductor value of 7.6 nH or close to this. The chip inductor chosen = 7.5 nH (0402CS-7N5 from Coilcraft). Coil inductors are recommended to provide sufficient Q for oscillation. Crystal and PFD Phase noise requirements are not excessive as the adjacent channel power requirement is −20 dB. The PFD is chosen so as to minimize spurious levels (beat note and reference), and to ensure a quick crystal power-up time. PFD = 3.6864 MHz − Power-Up Time 1.6ms. Figure 10 shows a typical power-on time for a 4 MHz crystal. N-Divider The N Divider is determined as being: Nint = 85 Nfrac = (1850)/4096 VCO divide-by-2 is enabled The main requirement of the harmonic filter should ensure that the third harmonic level is < −41.5 dBm. A fifth-order Chebyshev filter is recommended to achieve this, and a suggested starting point is given next. The Pi format is chosen to minimize the more expensive inductors. Component Values—Crystal: 3.6864MHz Loop Filter ICP 0.866 mA LBW 100 kHz C1 680 pF C2 12 nF C3 220 pF R1 1.1 kV R2 3 kV Matching L1 56 nH L2 1 nF C14 Short C15 Open Harmonic Filter L4 22 nH L5 22 nH CF1 3.3 pF CF2 8.2 pF CF3 3.3 pF Deviation The deviation is set to ± 50 kHz so as to accommodate a simple receiver architecture. The modulation steps available are in 3.6864 MHz/214 : Modulation steps = 225 Hz Modulation number = 50 kHz/225 Hz = 222 Rev. 0 | Page 20 of 28 ADF7012 433 MHZ OPERATION The recommendations here are guidelines only. The design should be subject to internal testing prior to ETSI site testing. Matching components need to be adjusted for board layout. The ETSI standard EN 300-220 governs operation in the 433.050 MHz to 434.790 MHz band. For many systems, 10% duty is sufficient for the transmitter to output 10 dBm. Design Criteria Loop Filter Bandwidth The loop filter is designed with ADIsimPLL Version 2.5. The loop bandwidth design requires that the channel power be < −36 dBm at ±870 kHz from the center. A loop bandwidth of close to 160 kHz strikes a good balance between lock time for data rates, including 32 kbps and spurious suppression. If it is found that pulling of the VCO is more than desired in OOK mode, the bandwidth could be increased. Design of Harmonic Filter 433.92 MHz center frequency FSK modulation 10 mW output power 200 m range Meets ETSI 300-220 The main requirement in the design of this remote is a long battery life and sufficient range. It is possible to adjust the output power of the ADF7012 to increase the range depending on the antenna performance. The center frequency is 433.92 MHz. It is possible to operate the VCO at this frequency. Figure 36 shows the inductor value vs. center frequency. The inductor chosen is 22 nH. Coilcraft inductors such as 0603-CS-22NXJBU are recommended. Crystal and PFD The phase noise requirement is such to ensure the power at the edge of the band is < −36 dBm. The PFD is chosen so as to minimize spurious levels (beat note and reference), and to ensure a quick crystal power-up time. PFD = 4.9152 MHz − Power-Up Time 1.6 ms. Figure 10 shows a typical power-up time for a 4 MHz crystal. N-Divider The N Divider is determined as being: Nint = 88 Nfrac = (1152)/4096 VCO divide-by-2 is not enabled The main requirement of the harmonic filter should ensure that the third harmonic level is < −30 dBm. A fifth-order Chebyshev filter is recommended to achieve this, and a suggested starting point is given next. The Pi format is chosen to minimize the more expensive inductors. Component Values—Crystal: 4.9152 MHz Loop Filter Icp 2.0 mA LBW 100 kHz C1 680 pF C2 12 nF C3 270 pF R1 910 V R2 3.3 kV Matching L1 22 nH L2 10 pF C14 Short C15 Open Harmonic Filter L4 22 nH L5 22 nH CF1 3.3 pF CF2 8.2 pF CF3 3.3 pF Deviation The deviation is set to ± 50 kHz so as to accommodate a simple receiver architecture. The modulation steps available are in 4.9152 MHz/214 : Modulation steps = 300 Hz Modulation number = 50 kHz/300Hz = 167 Bias Current Because low current is desired, a 2.0 mA VCO bias can be used. Additional bias current reduces any spurious, but increases current consumption. The PA bias can be set to 5.5 mA and achieve 10 dBm. Rev. 0 | Page 21 of 28 ADF7012 868 MHZ OPERATION The recommendations here are guidelines only. The design should be subject to internal testing prior to ETSI site testing. Matching components need to be adjusted for board layout. The modulation steps available are in 4.9152 MHz/214 : Modulation steps = 300 Hz Modulation number = 19.2 kHz/300 Hz = 64. The ETSI standard EN 300-220 governs operation in the 868 MHz to 870MHz band. The band is broken down into several subbands each having a different duty cycle and output power requirement. Narrowband operation is possible in the 50kHz channels, but both the output power and data rate are limited by the −36 dBm adjacent channel power specification. There are many different applications in this band, including remote controls for security, sensor interrogation, metering and home control. Bias Current Design Criteria 868.95 MHz center frequency (band 868.7MHz − 869.2 MHz) FSK modulation 12 dBm output power 300 m range Meets ETSI 300-220 38.4 kbps data rate The design challenge is to enable the part to operate in this particular subband and meet the ACP requirement 250 kHz away from the center. The center frequency is 868.95 MHz. It is possible to operate the VCO at this frequency. Figure 31 shows the inductor value vs. center frequency. The inductor chosen is 1.9 nH. Coilcraft inductors such as 0402-CS-1N9XJBU are recommended. Crystal and PFD The phase noise requirement is such to ensure the power at the edge of the band is < −36 dBm. This requires close to −100 dBc/Hz phase noise at the edge of the band. The PFD is chosen so as to minimize spurious levels (beat note and reference), and to ensure a quick crystal power-up time. A PFD of < 6 MHz places the largest PFD spur at a frequency of greater than 862 MHz, and so reduces the requirement on the spur level to −36 dBm instead of −54 dBm. PFD = 4.9152 MHz − Power Up-Time 1.6 ms. Figure 10 shows a typical power-on time for a 4MHz crystal. Because low current is desired, a 2.5 mA VCO bias can be used. Additional bias current reduces any spurious, but increases current consumption. A 2.5 mA bias current gives the best spurious vs. phase noise trade-off. The PA bias should be set to 7.5 mA to achieve 12 dBm. Loop Filter Bandwidth The loop filter is designed with ADIsimPLL Version 2.5. The loop bandwidth design requires that the channel power be < −36 dBm at ±250 kHz from the center. A loop bandwidth of close to <60 kHz is required to bring the phase noise at the edge of the band sufficiently low to meet the ACP specification. This represents a compromise between the data rate requirement and the phase noise requirement. Design of Harmonic Filter The main requirement of the harmonic filter should ensure that the second and third harmonic levels are < −30 dBm. A fifthorder Chebyshev filter is recommended to achieve this, and a suggested starting point is given next. The Pi format is chosen to minimize the more expensive inductors. Component Values—Crystal: 4.9152 MHz Loop Filter Icp 1.44 mA LBW 60 kHz C1 1.5 nF C2 22 nF C3 560 pF R1 390 V R2 910 V Matching L1 27 nH L2 6.2 nH C14 470 pF C15 Open Harmonic Filter L4 8.2 nH L5 8.2 nH CF1 4.7 pF CF2 6.8 pF CF3 4.7 pF N-Divider The N divider is determined as being: Nint = 176 Nfrac = (3229)/4096 VCO divide-by-2 is not enabled. Deviation The deviation is set to ±19.2 kHz so as to accommodate a simple receiver architecture and also ensure that the modulation spectrum is narrow enough to meet the adjacent channel power (ACP) requirements. Rev. 0 | Page 22 of 28 ADF7012 915 MHZ OPERATION The recommendations here are guidelines only. The design should be subject to internal testing prior to FCC site testing. Matching components need to be adjusted for board layout. FCC 15.247 and FCC 15.249 are the main regulations governing operation in the 902 MHz to 928 MHz Band. FCC 15.247 requires some form of spectral spreading. Typically, the ADF7012 would be used in conjunction with the frequency hopping spread spectrum (FHSS) or it may be used in conjunction with the digital modulation standard which requires large deviation frequencies. Output power of < 1 W is tolerated on certain spreading conditions. Compliance with FCC 15.249 limits the output power to −1.5 dBm, but does not require spreading. There are many different applications in this band, including remote controls for security, sensor interrogation, metering, and home control. Deviation The deviation is set to ±19.2 kHz so as to accommodate a simple receiver architecture, and also to ensure the available spectrum is used efficiently. The modulation steps available are in 10 MHz/214 : Modulation steps = 610 Hz Modulation number = 19.2 kHz/610 Hz = 31. Bias Current Because low current is desired, a 3 mA VCO bias can be used and still ensure oscillation at 928 MHz. Additional bias current reduces any spurious noise, but increases current consumption. A 3 mA bias current gives the best spurious vs. phase noise trade-off. The PA bias should be set to 5.5 mA to achieve 10 dBm power. Design Criteria Loop Filter Bandwidth 915.2MHz center frequency FSK modulation 10 dBm output power 200 m range Meets FCC 15.247 38.4 kbps data rate The loop filter is designed with ADIsimPLL Version 2.5. A data rate of 170 kHz is chosen, which allows for data rates of > 38.4 kbps. It also attenuates the beat note spurs quickly to ensure they have no effect on system performance. Design of Harmonic Filter The center frequency is 915.2 MHz. It is possible to operate the VCO at this frequency. Figure 36 shows the inductor value vs. center frequency. The inductor chosen is 1.6 nH. Coilcraft inductors such as 0603-CS-1N6XJBU are recommended. Additional hopping frequencies can easily be generated by changing the N value. Crystal and PFD The phase noise requirement is such to ensure that the 20 dB bandwidth requirements are met. These are dependant on the channel spacing chosen. A typical channel spacing would be 400 kHz, which would allow 50 channels in 20 MHz and enable the design to avoid the edges of the band. The PFD is chosen so as to minimize spurious levels. There are beat note spurious levels at 910 MHz and 920 MHz, but the level is usually significantly less than the modulation power. They are also attenuated quickly by the loop filter to ensure a quick crystal power-up time. PFD = 10 MHz − Power-Up Time 1.8 ms (approximately). Figure 10 shows a typical power-on time for a 4 MHz crystal. N-Divider The N divider is determined as being: Nint = 91 Nfrac = (2130)/4096 VCO divide-by-2 is not enabled The main requirement of the harmonic filter should ensure that the third harmonic level is < −41.5 dBm. A fifth-order Chebyshev filter is recommended to achieve this, and a suggested starting point is given next. The Pi format is chosen to minimize the number of inductors in the system. Component Values—Crystal: 10 MHz Loop Filter Icp 1.44 mA LBW 170 kHz C1 470 pF C2 12 nF C3 120 pF R1 470 V R2 1.8 kV Matching L1 27 nH L2 6.2 nH C14 470 pF C15 Open Harmonic Filter L4 8.2 nH L5 8.2 nH CF1 4.7 pF CF2 6.8 pF CF3 4.7 pF Rev. 0 | Page 23 of 28 ADF7012 REGISTER DESCRIPTIONS OD1 OUTPUT DIVIDER 0 0 1 1 0 1 0 1 DISABLED DIVIDE BY 2 DIVIDE BY 4 DIVIDE BY 8 VA2 VA1 VCO ADJUST 0 0 1 1 0 1 0 1 NO VCO ADJUSTMENT (FCENTER – (1 × f)) (FCENTER – (2 × f)) (FCENTER – (3 × f)) ADDRESS BITS ....... F3 F2 F1 F-COUNTER OFFSET 0 0 0 0 0 ....... ....... ....... ....... ....... 1 1 . 0 0 0 0 . 0 0 0 1 . 1 0 +1023 +1022 . +1 +0 1 1 . 1 1 ....... ....... ....... ....... ....... 1 1 . 0 0 1 1 . 0 0 1 0 . 1 0 –1 –0 –1 –1023 –1024 C1 (0) DB0 DB2 F1 C2 (0) DB1 DB3 F2 DB4 DB8 F7 F3 DB9 F8 DB5 DB10 F9 DB6 DB11 F4 DB12 F11 F10 F5 DB13 R1 X1 XOEB 0 XTAL OSCILLATOR ON (DEFAULT) 1 XTAL OSCILLATOR OFF F11 e.g., F-COUNTER OFFSET = 1, FRACTIONAL OFFSET = 1/215 CL4 0 0 0 0 . . . 1 1 CL3 0 0 0 1 . . . 1 1 CL2 0 1 1 0 . . . 1 1 CL1 1 0 1 0 . . . 0 1 CLKOUT DIVIDE RATIO 2 4 6 8 . 16 (DEFAULT) . 28 30 Figure 41. Rev. 0 | Page 24 of 28 RL4 0 0 0 0 . . . 1 1 1 1 RL3 0 0 0 1 . . . 1 1 1 1 RL2 0 1 1 0 . . . 0 0 1 1 RL1 1 0 1 0 . . . 0 1 0 1 RF R COUNTER DIVIDE RATIO 1 2 3 4 . . . 12 13 14 15 04617-0-027 OD2 DB7 DB14 D1 CRYSTAL DOUBLER 0 CRYSTAL DOUBLER OFF 1 CRYSTAL DOUBLER ON F6 DB15 R3 DB19 CL1 R2 DB20 CL2 DB16 DB21 CL3 R4 XOEB DB22 CL4 DB17 DB23 VA1 DB18 DB24 VA2 11-BIT FREQUENCY ERROR CORRECTION X1 DB25 OD1 4-BIT R DIVIDER D1 DB26 DB27 CLOCK OUT DIVIDER OD2 DB28 DB29 DB30 DB31 OUTPUT VCO DIVIDER ADJUST CRYSTAL DOUBLER R REGISTER ADF7012 ADDRESS BITS C1 (1) DB0 C2 (0) DB1 DB2 0 1 2 . . . 4092 4093 4094 4095 ....... ....... ....... ....... ....... ....... ....... ....... ....... ....... DB3 0 1 0 . . . 0 1 0 1 0 0 0 . . . 1 1 1 1 M1 0 0 1 . . . 0 0 1 1 0 0 0 . . . 1 1 1 1 M2 DB8 0 0 0 . . . 1 1 1 1 0 0 0 . . . 1 1 1 1 DB4 DB9 M7 DB11 M8 DB12 M11 M10 DB10 DB13 M9 DB14 N1 M12 DB15 N2 MODULUS DIVIDE RATIO M3 DB16 N3 M1 DB5 DB17 N4 M2 DB6 DB18 N5 M3 M11 M4 DB19 N6 M10 ....... M12 M5 DB20 N7 12-BIT FRACTIONAL-N DB7 DB21 N8 8-BIT INTEGER-N M6 DB22 PRESCALER DB23 P1 N-COUNTER LATCH e.g., SETTING F = 0 IN FSK MODE TURNS ON THE Σ-∆ WHILE THE PLL IS AN INTEGER VALUE N8 0 0 0 . . . 1 1 N7 0 0 0 . . . 1 1 N6 0 0 0 . . . 1 1 N5 0 0 0 . . . 1 1 N4 0 0 0 . . . 1 1 N3 0 0 0 . . . 1 1 N2 0 0 1 . . . 1 1 N1 0 1 0 . . . 0 1 N-COUNTER DIVIDE RATIO 1 2 3 . . . 254 255 P1 PRESCALER 0 4/5 1 8/9 THE N-VALUE CHOSEN IS A MINIMUM OF P2 + 3P + 3. FOR PRESCALER 8/9 THIS MEANS A MINIMUM N-DIVIDE OF 91. Figure 42. Rev. 0 | Page 25 of 28 04617-0-028 e.g., MODULUS DIVIDE RATIO = 2048 - > 1/2 ADF7012 DB11 DB10 DB9 DB8 D1 P6 P5 P4 MOD CONTROL GOOK ADDRESS BITS C1 (0) DB0 DB12 D2 C2 (1) DB1 DB13 D3 DB2 DB14 D4 DB3 DB15 D5 S1 DB16 D6 S2 DB17 D7 DB4 DB18 D8 G1 DB19 D9 DB5 DB20 MC1 DB6 DB21 MC2 P1 DB22 MC3 P2 DB23 MUST BE LOW DB7 DB24 POWER AMPLIFIER IC1 MODULATION DEVIATION P3 GFSK MOD CONTROL IC2 G1 GAUSSIAN OOK 0 ON 1 OFF IF AMPLITUDE SHIFT KEYING SELECTED, TxDATA = 0 D6 0 . 0 0 . . 1 . . . . . . . 1 . . . . . . . 1 D2 X 0 0 1 . . 1 D1 X 0 1 0 . . 1 –16.0dBm 1/31 * 14dBm 2/31 * 14dBm . . 14dBm IF FREQUENCY SHIFT KEYING SELECTED D9 0 0 0 0 . 1 ....... ....... ....... ....... ....... ....... ....... D3 0 0 0 0 . 1 D2 0 0 1 1 . 1 D1 0 1 0 1 . . S1 MODULATION SCHEME 0 0 1 1 0 1 0 1 FSK GFSK ASK OOK POWER AMPLIFIER OUTPUT LEVEL P6 0 0 0 0 . . 1 PA OFF S2 . . . . . . . 1 . . . . . . . 1 P2 X 0 1 1 . . 1 P1 X 1 0 1 . . 1 PA OFF –16.0dBm 1/31 * 14dBm 2/31 * 14dBm . . 13dBm FSTEP = FPFD /214 F DEVIATION PLL MODE 1 × –FSTEP 2 × –FSTEP 3 × –FSTEP ....... 511 × –FSTEP IF GAUSSIAN FREQUENCY SHIFT KEYING SELECTED IC2 0 0 1 1 IC1 0 1 0 1 INDEX COUNTER 16 32 64 128 MC3 0 0 . 1 MC2 0 0 . 1 MC1 0 1 . 1 GFSK MOD CONTROL 0 1 . 7 D7 0 0 0 0 . 1... ....... ....... ....... ....... ....... ....... ....... Figure 43. Rev. 0 | Page 26 of 28 D3 0 0 0 0 . 1 D2 0 0 1 1 . 1 D1 0 1 0 1 . 1 DIVIDER FACTOR 0 1 2 3 ....... 127 04617-0-029 DB25 DB26 DB27 DB28 DB29 DB30 DB31 TEST BITS INDEX COUNTER MODULATION REGISTER ADF7012 VB4 0 0 . 1 PA2 0 0 1 . . 1 VB3 0 0 . 1 PA1 0 1 0 . . 1 VB2 0 1 . 1 PA BIAS 5µA 6µA 7µA . . 12µA VB1 1 0 . 1 VCO BIAS CURRENT 0.5mA M3 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 M2 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 ADDRESS BITS C1 (1) DB0 DB2 PD1 BLEED DOWN PD1 PLL ENABLE 0 1 BLEED OFF BLEED ON 0 1 PLL OFF PLL ON CP3 BLEED UP PD2 PA ENABLE 0 1 BLEED OFF BLEED ON 0 1 PA OFF PA ON CP2 CP1 0 0 1 1 0 1 0 1 CHARGE PUMP CURRENT 0.3mA 0.9mA 1.5mA 2.1mA VD1 VCO DISABLE PD3 CLKOUT 0 1 VCO ON VCO OFF 0 1 CLKOUT OFF CLKOUT ON M1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 C2 (1) DB1 DB3 DB4 PD3 PD2 DB5 I1 DATA INVERT CLKOUT ENABLE PA ENABLE PLL ENABLE DB6 CP1 VCO DISABLE CP4 1mA . 8mA M4 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 DATA INVERT DATA DATA MUXOUT LOGIC LOW LOGIC HIGH INVALID MODE – DO NOT USE REGULATOR READY (DEFAULT) DIGITAL LOCK DETECT ANALOG LOCK DETECT R DIVIDER/2 OUTPUT N DIVIDER/2 OUTPUT RF R DIVIDER OUTPUT DATA RATE BATTERY MEASURE IS > 3.25V BATTERY MEASURE IS > 3V BATTERY MEASURE IS > 2.75V BATTERY MEASURE IS > 2.35V NORMAL TEST MODES Σ-∆ TEST MODES Figure 44. Rev. 0 | Page 27 of 28 04617-0-030 PA3 0 0 0 . . 1 DB7 DB8 DB12 M2 I1 0 1 CP2 DB13 M3 CP3 DB14 M4 DB9 DB15 LD1 CP4 DB16 VB1 DB11 DB17 VB2 DB10 DB18 VB3 M1 DB19 BLEED CHARGE CURRENT PUMP VD1 DB20 PA1 DB24 PT2 VB4 DB25 PT3 DB21 DB26 PT4 PA2 DB27 PT5 DB22 DB28 ST1 MUXOUT DB23 DB29 ST2 VCO BIAS PT1 DB30 PA BIAS PA3 DB31 ST3 PLL TEST MODES ST4 SD TEST MODES LD PRECISION FUNCTION REGISTER ADF7012 OUTLINE DIMENSIONS 7.90 7.80 7.70 24 13 4.50 4.40 4.30 6.40 BSC 1 12 PIN 1 0.65 BSC 0.15 0.05 0.30 0.19 1.20 MAX SEATING PLANE 0.20 0.09 0.75 0.60 0.45 8° 0° 0.10 COPLANARITY COMPLIANT TO JEDEC STANDARDS MO-153AD Figure 45. 24-Lead Thin Shrink Small Outline Package [TSSOP] (RU-24) ORDERING GUIDE Model Temperature Range Package Description Package Option Frequency Range ADF7012BRU ADF7012BRU-REEL ADF7012BRU-REEL7 EVAL-ADF7012EB1 EVAL-ADF7012EB2 EVAL-ADF7012EB3 EVAL-ADF7012EB4 EVAL-ADF7012EB5 −40°C to +85°C −40°C to +85°C −40°C to +85°C TSSOP TSSOP, 13”REEL TSSOP, 7” REEL Evaluation Board Evaluation Board Evaluation Board Evaluation Board Evaluation Board RU-24 RU-24 RU-24 50 MHz to 1 GHz 50 MHz to 1 GHz 50 MHz to 1 GHz 902 MHz to 928 MHz 860 MHz to 880 MHz 418 MHz to 435 MHz 310 MHz to 330 MHz 50 MHz to 1 GHz © 2004 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D04617–0–10/04(0) Rev. 0 | Page 28 of 28