弑 求唧 窀 T抄 d咖 o矽 Ofy尺 ε货urch&盗 pI9Jjc砑 莒 粥 l 电信与数据通 信 电源高压 PWM控 制器 MA× 5003应 用介绍 国营 第 SO72厂 刘鲁 香 南京 邮 电 学 院 李静 摘要 :高 压 PWM控 制器 MA× sO03除 用于电信和 IsDN电 源外 ,还 可用于+42V汽 车系统、土业 电源 和高压电源模块 。文 中介 绍了 MA× sO00的 特 `点 ,重 ・ 点介绍 了其应用。 关键词 :电 信/数 据通信 ;电 源控制器 ;MA× 5003 1.序 言 式电源 。 目前 的电信和数据通信系统 的电源总线 电压 是 缌VDc。 在 电话网络中,该 电压 电平当时远距离 ” 2。 主要 特 点 和 优 点 环路末端设备供电时,能 充分降低线路电流 ,并 确 MA× sO∞ 单片 IC集 成了 10~sO只 晶体管 ,内 (MA× 1M)公 司生产的高压开 证安全。美国美信 部电路主要由振荡器 、斜坡产生器、电流传感比较 PWM控 关电源 制器 MA× 5003单 片 IC,主 要瞄准 器 、PWM比 较器 、误差放大器和输出驱动器等组 电信 电源和综合业 务数据 网电源而 设计 。除此之 外 ,该 PWNl控 制器还可用于+42V汽 车系统 、高压 成 。其主要特点和优点如附表所列 。 电源模块和工业 电源 。 3.应 MA× 5003是 一种 电压模式控制器 。该类型的 用介绍 : 一 MA× 5o03是 处应用灵活的电压型 PWM控 控制器除具有电流模式控制器 IC优 异的控制环路 带宽、对输入电压变化有相同的周期响应和逐周脉 制器 ,它 可以在回扫式或正向变换器拓扑中在连续 或断续电流模式下工作 。当 MA× sO03应 用于反激 式拓扑结构时,输 出功率限制在 20W以 下 。当其 在正向变换器 中作为 PWM控 制器使用时,输 出功 冲 电流限制等优点外 ,同 时还具有 内在的抗噪扰能 力 ,并 使用铰简单的补偿方案。MA× ~sOO3为 回扫 式和正 向电压型控制变 换器提供全部所需要 的功 艹 能 ,可 用作设计宽范 围输入 电压的隔离式和非隔离 附表 MA× sO∞ 特,点 及优 `点 序号 1 , ・ 优 点 点 输 入 电压 范 围 :Ⅱ △ 10V 直接 利 用 高压 操 作 电压模 式控 制 ,电 流 限制 易于补偿 ,优 良的输 入 瞬 态响应和抗 噪扰性 能 3 内置 高压启 动 电路 4 可编 程 开 关 电 流 限制 5 直达 300KIIZ的 可调频 率 ,外 部 频 率 同步化 6 软启 动 时 间和 欠 电压 锁 定 门限可调 7 输入 电压 前馈 8 ±2.5%的 精 密 内部基 准 电压 9 囤 特 率范围被限制在 ⒛-sOW。 新式 16引 脚 QsoP或 《中国咆塬博览》2003年 第 7期 在轻载 下保 持 高效 率 选择低 成本 MOSmT,设 计灵 活性 强 提 高设 计灵 活性 ,减 小 EMI,选 择 较 小的 磁 性元 件 ,提 高 系统功 率密度 提 高了系统可靠性和设计灵 活性 ,简 化 总 线分布设计 快速 线路 瞬 态响应 so封 装 提 高电压 精度和稳定性 面积 比 14引 脚 s0封 装 小 绍 % ^ ' 弑 求饵 窀鹦 图 图2 1 输 出 5V/1A的 反 激 式非 隔 离变换 器 电路 一鲳 Ⅴ 输 入 和 5W1A输 出的 隔 离式 变换 器 电路 3.1输 出 5W1A的 非 隔离型反激式变换器 MA× 50∞ 由 MA× 5003组 成 的一 种非隔离式反激型变换 器 电路 如 图 1所 示 。 该 变 换 器 DC输 入 电压 VIN〓 36~72Ⅴ (含 48Ⅴ 在 内 ),DC输 出电压 V。 叨=5V,・ 输 出电流 I。uT〓 1A。 输 入 电压从 r的 1脚 (V+脚 )加 入之后 ,内 部 的耗尽型 ∏Ⅱ 晶体管导通 ,内 部低压差线性稳压 器 LDo1和 LDo2依 次启动 ,为 IC产 生 Vcc电 源 电压 (7.4~12Ⅴ )。 一 旦 超 过 输 入 电压 欠 电压 锁 定 (UVLo)预 置 门限 ,内 部高压 ∏r晶 体管截止 , 高压被切 断 ,变 压器辅助绕组组成 的电源 电路连接 到 IC的 VDD脚 ,为 IC供 电 。 当 VDD〓 12Ⅴ 时 ,电 的 2脚 连接外部 的电阻分压器 ,其 功能是欠 电压传感和前馈输入 。IC的 3脚 为预调整 器 输 出 ,与 地 (AGND)之 间连接 一 只 0.1uF的 电 容 。 当 VN(36V时 ,该 脚应连接到 V十 脚 。IC脚 动 电容 (470nF),6脚 为 3Ⅴ 的参考 电压输 出 ,7脚 为 PWM比 较器控制输 入 ,8脚 为误差放大器输 出 (在 外部连接 RF和 CF补 偿 网络 ),9脚 为反馈输入 。 IC脚 10外 部 电阻 (51KΩ )用 作设定 PWWI增 益 并限制最大 PWbl占 空 比 (%%)。 IC脚 14驱 动 的 , MOs∏ Ⅱ 源极 电阻 Rcs,为 电流检测元件 ,其 感测 信号从 IC脚 12输 入 。 变压器是关键元件 ,其 设计步骤如下 源 电流仅为 zmA。 4 外部 电阻用作设置 PWM频 率 ,5脚 外部连接软启 : 妒 2003年 第 7期 《中国 爸你博 览》 田 斌 求料 窀 (1)确 定变压器匝数 比,检 验最大 占空因数 〓 559b× (36;冫 亻 2y) 影响变压器初 、次级绕组匝数 比的主要 因数是 开关击 穿 电压和 占空因数 。较小 的匝数 比 ,有 利于 减 小次 级整 流 电压 和 在 回扫 期 间通 过 开关 的最 大 (RMs) 电流 ,从 而影 响系统效率 。对于 鲳 V到 5V的 系统 为保持 回扫 电压在控制之下 ,匝 数 比 (N=NP/Ns) 电压 ,但 会减小 占空 比,增 加初级有效值 可选择 8∶ 1。 为避免在连续传 导模式 作 ,PWNl最 大 占空 因数 由公式 , (CCM)操 (D确 定 -” : DCMAx〓 (T::甾 +1 品丁 由于最低输入 电压 VⅧ Ⅱ 36V,N〓 8,次 级 电压 Vs〓 V。 trr+Ⅴ D=5V+0.4V〓 5,4V(VD为 肖特基整流 二 (D可 以得到最大 占空 极管正 向压 降 ),根 据公式 因数 (DCMAx)为 55%。 (2)确 定变压器初级 电感值 假设系统效率 n〓 gO%,欲 产生 5W(5V× 1A) 的输 出所需要 的输入功率 PN=6.笏 W。 设实际操作 占 空 因 数 比 最 大 占 空 因 数 低 12%,即 DC=ss%-12%〓 43%,开 关频率 /sw司 o0KIIz,初 级 电感值可由公式 (2)计 算 : (2) LP〓 (0.43× 36y)2 2× 6.75× 300KJfz 〓65/Jff (3)确 定初 、次级最大 电流和最 小 占空 因数 峰值初级 电流为 : 65/Jff× 峰值次级 电流 比 , 即 300KHz =0.8A `° Is(PK)〓 NIP(PK)〓 0・ 8A× 8〓 6.4A 当 VN(MAx)〓 ” V时 ,满 载 下最 小 占空 因数 为 严 囤 (似 ;冫 ) 亻 彳 :蕊 幻 《中@咆 你博 览》⒛ 03年 第 7期 V(含 -鲳 V),输 出是 5W1A。 从输 出到初级 侧 的反馈环 路 由电阻分压 器 (两 支 zzI。 9Κ Ω的电阻 )、 作 为误差放大器 使用 的并联稳压器 TI/31和 光耦 合器 MDC217组 成 。MA× 5003的 输入脚 V+连 接 oV的 线路 ,接 地脚 AGND则 连接 -鲳 v的 母线 。 3.350W光 耦 隔离单端 正激式变换器 由 MA× 5003组 成 的光耦 隔离的单端正 激式变 换器 电路如 图 3所 示 。该拓扑结构与反激式拓扑 比 较 ,具 有较高 的输 出功率 电平 ,并 有较小趵 纹波和 较小 的峰值与平 均值 电流之 比。这 种正激式变换器 输入 电压范 围仍为 36~” V,输 出电压 输 出电流 I。 t/r〓 10A,在 V。 uT〓 5V, vIN〓 48V和 P。 tJr〓 25W下 的 效率 η=ss%,开 关频率 冉w设 置在 犭ClKHz,输 入 与输出之间的隔离电压为 1500V(@← 1s)。 变压器 T1的 ①与②端之间的绕组 (14T)是 初 级绕组 。T1③ 与④端之间的复位绕组 (12T)与 二 极管 D5组 成复位电路 ,其 作用是防止磁芯剩磁累 加导致脉冲变压器饱和而损坏功率开关 Q1。 当系 统启动之后 ,T1⑤ 与⑥端之间的辅助绕组 (4T), 以及二极管 D3、 电容 C8、 齐纳二极管 Z1和 晶体 管 Q2组 成的电源电路 ,为 U1供 电。T1⑤ 、⑥端 之间的偏置绕组工作在反激模式 ,故 可省略一只滤 波电感 。 变 换器输出电压的稳定控制 ,是 通过 RII与 ・ R12组 成 的 电 阻 分 压 器 、 高 精 度 稳 压 器 U3 (TL431AID)和 光耦合器 U2(MOC⒛ 7)组 成的 反馈环路实现的。当输出电流从零到 10A变 化时 , K)为 峰值初级 电流乘 以匝数 洲 =DcM舡 脚 =27.5% 根据 以上确定的参数 ,来 选择变压器磁 芯和 绝 ∷ 缘铜导线规 格 。 3.2输 出 5W1A的 隔 离型反激式变换 器 与 图 1所 示 的电路相 对应 的隔离型反 激式变换 器 电路如 图 2所 示 。该应用 电路输入 电压从-%V到 : 输出电压调整精度在 ±0.3%之 内。 高频变压器采用 EFD⒛ 型铁氧体磁芯制作 ① 、②端之间初级绕组 (N1乇 )电 感值是 犭0uH。 , 次级双 肖特基整流二极管 D4的 额定电流为 20A, 反向击穿电压是 40V。 滤波电感器 L1电 感量为 4.7 u耳 ,能 通过 10A的 电流 。滤波电感容 C7、 c13和 C14选 用 笳0uF/b。 3V的 铝电解电容器 ,C15选 用 卫 日 穸 ‘ ” 1刀饣 oJo￡ vR‘ |se‘饣 Fch 球钾 窀 c殳 Az,l9Jj‘ ˇ z矽氵 o托 弥补 了砖 块密 封 式模 块 电源 制造 过程 中散 热 降温 要求高 、周期长 、输 出电压不灵活和价格较高等不 MA× ⒛ 19和 MA× 50⒛ 等器件 。其 中 MA× sO14和 MA× sO19可 分别用作组成隔离式和 非 隔离式仅激式变换器 ,最 大 占空 因数 为 85%; MA× sO15和 MA× ∞ 20可 分别用作设计光耦 隔离 足 ,具 有较高 的性价 比。 式 和 非 隔 离 式 正 激 式 拓 扑 结 构 ,最 大 占空 比 是 MA× 1M公 司生产 的能信 电源 PWNl控 制器 ICs,除 了 MA× sO03之 外 ,还 有 MA× 5014、 MA 50%。 0.1uF的 陶瓷 电容 。 图 3所 示 的 50W电 信/数 据通信服务器 电源 ×5015、 , , MA× sO14~MA× 5020均 采用 8引 脚 So封 装 ,单 价 “ 美元 (采 购批量 10000只 时售价 0。 )。 I GGGGG 图3 sOW藕 隔 离正 激 式 变换 器 电路 C7I.C13I亠 C1 19-1555; Rev 2; 4/02 KIT ATION EVALU E L B AVAILA High-Voltage PWM Power-Supply Controller General Description The MAX5003 offers some distinctive advantages: softstart, undervoltage lockout, external frequency synchronization, and fast input voltage feed-forward. The device is designed to operate at up to 300kHz switching frequency. This allows use of miniature magnetic components and low-profile capacitors. Undervoltage lockout, soft-start, switching frequency, maximum duty cycle, and overcurrent protection limit are all adjustable using a minimum number of external components. In systems with multiple controllers, the MAX5003 can be externally synchronized to operate from a common system clock. Warning: The MAX5003 is designed to operate with high voltages. Exercise caution. The MAX5003 is available in 16-pin SO and QSOP packages. An evaluation kit (MAX5003EVKIT) is also available. Applications Features ♦ Wide Input Range: 11V to 110V MAX5003 The MAX5003 high-voltage switching power-supply controller has all the features and building blocks needed for a cost-effective flyback and forward voltagemode control converter. This device can be used to design both isolated and nonisolated power supplies with multiple output voltages that operate from a wide range of voltage sources. It includes a high-voltage internal start-up circuit that operates from a wide 11V to 110V input range. The MAX5003 drives an external Nchannel power MOSFET and has a current-sense pin that detects overcurrent conditions and turns off the power switch when the current-limit threshold is exceeded. The choice of external power MOSFET and other external components determines output voltage and power. ♦ Internal High-Voltage Startup Circuit ♦ Externally Adjustable Settings Output Switch Current Limit Oscillator Frequency Soft-Start Undervoltage Lockout Maximum Duty Cycle ♦ Low External Component Count ♦ External Frequency Synchronization ♦ Primary or Secondary Regulation ♦ Input Feed-Forward for Fast Line-Transient Response ♦ Precision ±2.5% Reference over Rated Temperature Range ♦ Thermal Shutdown Ordering Information PART TEMP. RANGE PIN-PACKAGE MAX5003CEE 0°C to +70°C 16 QSOP MAX5003CSE 0°C to +70°C 16 Narrow SO MAX5003C/D MAX5003EEE MAX5003ESE (Note A) -40°C to +85°C -40°C to +85°C Dice 16 QSOP 16 Narrow SO Note: Dice are designed to operate over a -40°C to +140°C junction temperature (Tj) range, but are tested and guaranteed at TA = +25°C. Pin Configuration Telecommunication Power Supplies ISDN Power Supplies +42V Automobile Systems High-Voltage Power-Supply Modules Industrial Power Supplies TOP VIEW V+ 1 16 VDD INDIV 2 15 VCC ES 3 FREQ 4 SS 5 14 NDRV MAX5003 13 PGND 12 CS REF 6 11 AGND CON 7 10 MAXTON COMP 8 9 FB QSOP/Narrow SO ________________________________________________________________ Maxim Integrated Products For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at 1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com. 1 MAX5003 High-Voltage PWM Power-Supply Controller ABSOLUTE MAXIMUM RATINGS V+ to GND ............................................................-0.3V to +120V ES to GND ..............................................................-0.3V to +40V VDD to GND ............................................................-0.3V to +19V VCC to GND .........................................................-0.3V to +12.5V MAXTON, COMP, CS, FB, CON to GND..................-0.3V to +8V NDRV, SS, FREQ to GND ...........................-0.3V to (VCC + 0.3V) INDIV, REF to GND.................................................-0.3V to +4.5V VCC, VDD, V+, ES Current ................................................±20mA NDRV Current, Continuous...............................................±25mA NDRV Current, ≤ 1µs .............................................................±1A CON and REF Current ......................................................±20mA All Other Pins ....................................................................±20mA Continuous Power Dissipation (TA = +70°C) 16-Pin SO (derate 9.5mW/°C above +70°C)...............762mW 16-Pin QSOP (derate 8.3mW/°C above +70°C)..........667mW Maximum Junction Temperature (TJ) ..............................+150°C Operating Temperature Ranges MAX5003C_E ....................................................0°C to +70°C MAX5003E_E ..................................................-40°C to +85°C Operating Junction Temperature (TJ) .............................+125°C 16-Pin SO θJA .................................................................105°C/W 16-Pin QSOP θJA............................................................120°C/W Storage Temperature Range .............................-65°C to +150°C Lead Temperature (soldering, 10s) .................................+300°C Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS (V+ = VES = VDD = +12V, VINDIV = 2V, VCON = 0, RFREQ = RMAXTON = 200kΩ, TA = TMIN to TMAX, unless otherwise noted. Typical values are at TA = +25°C.) PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS 35 75 µA 1.2 mA SUPPLY CURRENT Shutdown Current I+ VINDIV = 0, V+ = 110V, VES = VDD = unconnected Supply Current IDD V+ = VES , VDD = 18.75 V+ ES = VDD = unconnected PREREGULATOR/STARTUP V+ Input Voltage (Note 1) INDRV = 2mA 25 V INDRV = 5mA 10.8 ES Input Voltage (Note 1) VESI VDD = unconnected, V+ = VES, INDRV = 7.5mA ES Output Voltage VESO V+ = 110V, VDD = unconnected VDD Output Voltage VDD V+ = 36V, IDD = 0 to 7.5mA, ES = unconnected VDD Input Voltage Range VDD V+ = VES = 36V, INDRV = 7.5mA 10.75 VDD Regulator Turn-Off Voltage VTO V+ = 36V, IV+ < 75µA, ES = unconnected 10.75 VCC V+ = 36V, ES = unconnected, VDD = 18.75V 9 9.75 110 V 36 V 36 V 10.5 V 18.75 V V CHIP SUPPLY (VCC) VCC Output Voltage VCC Undervoltage Lockout Voltage 7.4 12 V VCC falling 6.3 V Peak Source Current VNDRV = 0, VCC supported by VCC capacitor 570 mA Peak Sink Current NDRV Resistance High NDRV Resistance Low REFERENCE REF Output Voltage REF Voltage Regulation ROH ROL VNDRV = VCC INDRV = 50 mA INDRV = 50 mA 1000 4 1 12 mA Ω Ω VREF ∆VREF No load IREF = 0 to 1mA 3.000 5 3.098 20 V mV VCCLO OUTPUT DRIVER 2 2.905 _______________________________________________________________________________________ High-Voltage PWM Power-Supply Controller (V+ = VES = VDD = +12V, VINDIV = 2V, VCON = 0, RFREQ = RMAXTON = 200kΩ, TA = TMIN to TMAX, unless otherwise noted. Typical values are at TA = +25°C.) PARAMETERS SYMBOL CONDITIONS MIN TYP MAX UNITS 100 120 mV +1 µA CURRENT LIMIT CS Threshold Voltage VCS VCON = 1.25V 80 CS Input Bias Current ICS 0 < VCS < 0.1V -1 Overcurrent Delay tD From end of blanking time 25mV overdrive CS Blanking Time tB 240 ns 70 ns ERROR AMPLIFIER Voltage Gain AV ICOMP = 5µA; VCOMP = 0.5V, 2.5V 80 dB Unity-Gain Bandwidth BW RLOAD = 200kΩ, CLOAD = 100pF 1.2 MHz AVOL = 1V/V, CLOAD = 100pF 65 degrees φ Phase Margin 60 Output Clamp Low VCOMPL At COMP 0.25 V Output Clamp High VCOMPH At COMP 3.00 V FEEDBACK INPUT AND SET POINT FB Regulation Voltage VSET FB Bias Current IFB FB VSET Tempco TCFB FB = COMP, VCON = 1.5V 1.448 -1 VFB = 1.5V 1.485 1.522 0.1 +1 100 V µA ppm/°C UNDERVOLTAGE LOCKOUT INDIV Undervoltage Lockout INDIV Hysteresis VINDIVLO V+ = VES = VDD = 10.8V and 18.75V VINDIV falling 1.15 1.20 1.25 VINDIV rising 1.23 1.32 1.45 125 VHYST INDIV Bias Current VINDIV = 1.28V -1 0.01 V mV +1 µA 0.8 V 1 µA MAIN OSCILLATOR—EXTERNAL MODE FREQ Input Low VIL VCON = 3.0V FREQ Input High VIH VCON = 3.0V FREQ Output Low IOL VFREQ = 5V, VCON = 3.0V External Oscillator Maximum Low Time tEXT (Note 2) FREQ Range fFREQ Frequency Range fS fS = 1/4 fFREQ 2.7 8 V 13 µs 200 1200 kHz 50 300 kHz 150 FREQ HI/LO Pulse Width ns MAIN OSCILLATOR—INTERNAL MODE FREQ Resistor Range 50 RFREQ 80 Oscillator Frequency FREQ Output Current High IOH VFREQ = 0 FREQ Output Current Low IOL VFREQ = 1.5V 100 500 kΩ 120 kHz 300 µA 1 µA MAXIMUM DUTY CYCLE (MAXTON) Maximum Programmable Duty Cycle VINDIV = 1.25V 75 % _______________________________________________________________________________________ 3 MAX5003 ELECTRICAL CHARACTERISTICS (continued) ELECTRICAL CHARACTERISTICS (continued) (V+ = VES = VDD = +12V, VINDIV = 2V, VCON = 0, RFREQ = RMAXTON = 200kΩ, TA = TMIN to TMAX, unless otherwise noted. Typical values are at TA = +25°C.) PARAMETERS SYMBOL CONDITIONS MIN TYP MAX UNITS 500 kΩ PWM OSCILLATOR MAXTON Resistor Range RMAXTON Maximum On-Time Range tON 50 7.5 RMAXTON = 200kΩ, VINDIV = 1.25V µs Input Voltage Feed Forward Ratio VINDIV stepped from 1.5V to 1.875V, VCON = 3.0V (Note 3) 0.72 0.8 0.88 RAMP Voltage Low VINDIV = 1.875V 0.48 0.5 0.53 V RAMP Voltage High 2.5 V Minimum On-Time 200 ns SOFT-START SS Source Current VSS = 0.5V, VDD = unconnected, VCON = 1.5V SS Sink Current VSS = 0.4V (Note 4) 3.4 5.5 SS Time 9 µA 10 mA 0.45 s/µF PWM COMPARATOR CON Bias Current ICON -1 VCON = 0.5V and 2.5V 0.01 1 µA THERMAL SHUTDOWN Thermal Shutdown Temperature 150 °C Thermal Hysteresis 20 °C Note 1: Note 2: Note 3: Note 4: See the Typical Operating Characteristics for preregulator current-to-voltage characteristics. Maximum time FREQ can be held below VIL and still remain in external mode. Feed-forward Ratio = Duty cycle at (VINDIV = 1.5V)/Duty cycle at (VINDIV = 1.875V) Occurs at start-up and until VREF is valid. Typical Operating Characteristics (VDD = +12V, RFREQ = 200kΩ, RMAXTON = 200kΩ, TA = +25°C, unless otherwise noted.) FB SET-POINT VOLTAGE CHANGE vs. SUPPLY VOLTAGE 0.2 0 -0.2 -0.4 -0.6 0.050 0.025 0 -0.025 -0.050 -0.075 -20 0 20 40 60 TEMPERATURE (°C) 80 100 0.20 0 -0.20 -0.40 -0.60 -0.80 -1.00 -0.100 0.8 4 0.075 FREQUENCY CHANGE (%) 0.4 0.40 MAX5003-02 0.6 0.100 FB SET-POINT VOLTAGE CHANGE (%) MAX5003-01 0.8 SWITCHING FREQUENCY CHANGE vs. TEMPERATURE MAX5003-03 FB SET-POINT VOLTAGE CHANGE vs. TEMPERATURE FB SET-POINT VOLTAGE CHANGE (%) MAX5003 High-Voltage PWM Power-Supply Controller -1.20 11 12 13 14 15 VDD (V) 16 17 18 -40 -20 0 20 40 60 TEMPERATURE (°C) _______________________________________________________________________________________ 80 100 High-Voltage PWM Power-Supply Controller V+ INPUT CURRENT vs. TEMPERATURE 2.5 MAXIMUM DUTY CYCLE vs. VINDIV MAX5003-05 MAX5003-04 2.50 80 400kΩ 70 300kΩ 2.00 MAX5003-06 V+ INPUT CURRENT vs. VOLTAGE 3.0 60 1.50 1.00 VCON = VCOMP = VFB SWITCHING 60 80 100 -20 0 20 40 60 80 100 V+ (V) TEMPERATURE (°C) ERROR AMP FREQUENCY RESPONSE SWITCHING FREQUENCY AND PERIOD vs. RFREQ 60 0 400 -20 350 50 -40 40 -60 PHASE -80 20 -100 40 V+ = 110V VINDIV = 0 VDD = UNCONNECTED 38 30 20 150 -10 -160 50 -20 -180 0 35 34 32 31 0 0 10M 36 33 10 100 100 200 400 300 30 500 -40 -20 RFREQ (kΩ) FREQUENCY (Hz) V+ = 110V VINDIV = 1.5V 20 40 60 80 100 VCC LOAD REGULATION 10 MAX5003-10 28.0 0 TEMPERATURE (°C) V+ CURRENT IN BOOTSTRAPPED OPERATION vs. TEMPERATURE V+ = 50V TO 110V 9 8 27.5 7 VCC (V) IV+ (µA) 40 PERIOD -140 1M 3.0 V+ SHUTDOWN CURRENT vs. TEMPERATURE 39 200 0 100k 2.5 2.0 37 -120 10k MAX5003-08 250 10 1k 1.5 VINDIV (V) 300 30 0.1k 1.0 FREQUENCY FREQUENCY (kHz) GAIN PHASE (degrees) MAX5003-07 70 GAIN (dB) 0 -40 120 IV+ (µA) 40 PARAMETER IS RMAXTON VCON CLAMPED HIGH PERIOD (µs) 20 100kΩ 10 0 0 30 20 VCON = VCOMP = VFB SWITCHING V+ = 110V 0.50 0.5 40 MAX5003-09 1.0 200kΩ 50 MAX5003-11 1.5 DUTY CYCLE (%) IV+ (mA) IV+ (mA) 2.0 27.0 V+ = 12V 6 V+ = 13V 5 V+ = 14V 4 3 26.5 V+ = 15V 2 ES = UNCONNECTED VDD = UNCONNECTED 1 0 26.0 -50 0 50 TEMPERATURE (°C) 100 0 5 10 15 20 ICC (mA) _______________________________________________________________________________________ 5 MAX5003 Typical Operating Characteristics (continued) (VDD = +12V, RFREQ = 200kΩ, RMAXTON = 200kΩ, TA = +25°C, unless otherwise noted.) Typical Operating Characteristics (continued) (VDD = +12V, RFREQ = 200kΩ, RMAXTON = 200kΩ, TA = +25°C, unless otherwise noted.) 350 9 MAX SWITCHING FREQUENCY (kHz) MAX5003-12 10 8 7 6 5 4 3 2 V+ = VES = 12V TO 36V VDD = UNCONNECTED 1 0 300 30nC MAX5003-13 MAXIMUM FREQUENCY vs. INPUT VOLTAGE AND FET TOTAL GATE-SWITCHING CHARGE VCC LOAD REGULATION VCC (V) MAX5003 High-Voltage PWM Power-Supply Controller 25nC 250 20nC 200 10nC 150 15nC 100 50 0 0 5 10 15 20 12 13 ICC (mA) 15 14 16 V+ (V) Pin Description PIN NAME FUNCTION 1 V+ Preregulator Input. Connect to the power line for use with 25V to 110V line voltages. Bypass V+ to ground with a 0.1µF capacitor, close to the IC. Connects internally to the drain of a depletion FET preregulator. 2 INDIV Undervoltage Sensing and Feed-Forward Input. Connect to the center point of an external resistive divider connected between the main power line and AGND. Undervoltage lockout takes over and shuts down the controller when VINDIV < 1.2V. INDIV bias is typically 0.01µA. 3 ES 4 FREQ 5 SS 6 REF Reference Voltage Output (3.0V). Bypass to AGND with a 0.1µF capacitor. 7 CON Control Input of the PWM Comparator 8 COMP 9 FB 10 6 MAXTON Preregulator Output. When V+ ranges above 36V, bypass ES to AGND with a 0.1µF capacitor close to the IC. When V+ is always below 36V, connect ES to V+. Oscillator Frequency Adjust and Synchronization Input. In internal free-running mode, the voltage on this pin is internally regulated to 1.25V. Connect a resistor between this pin and AGND to set the PWM frequency. Drive between VIL and VIH at four times the desired frequency for external synchronization. Soft-Start Capacitor Connection. Ramp time to full current limit is approximately 0.5ms/nF. Limits duty cycle when VSS < VCON. Compensation Connection. Output of the error amplifier, available for compensation. Feedback Input. Regulates to VFB = VREF / 2 = 1.5V. Maximum On-Time Programming. A resistor from MAXTON to AGND sets the PWM gain and limits the maximum duty cycle. The voltage on MAXTON tracks the voltage on the INDIV pin. Maximum on-time is proportional to the value of the programming resistor. The maximum duty cycle is limited to 75%, regardless of the programming resistor. _______________________________________________________________________________________ High-Voltage PWM Power-Supply Controller PIN NAME FUNCTION 11 AGND 12 CS 13 PGND Power Ground. Connect to AGND. 14 NDRV Gate Drive for External N-Channel Power FET 15 VCC Output Driver Power-Rail Decoupling Point. Connect a capacitor to PGND with half the value used for VDD bypass very close to the pin. If synchronizing several controllers, power the fan-out buffer driving the FREQ pins from this pin. 16 VDD 9.75V Internal Linear-Regulator Output. Drive VDD to a voltage higher than 10.75V to bootstrap the chip supply. VDD is also the supply voltage rail for the chip. Bypass to AGND with a 5µF to 10µF capacitor. Analog Ground. Connect to PGND close to the IC. Current Sense with Blanking. Turns power switch off if VCS rises above 100mV (referenced to PGND). Connect a 100Ω resistor between CS and the current-sense resistor (Figure 2). Connect CS to PGND if not used. Detailed Description The MAX5003 is a PWM controller designed for use as the control and regulation core of voltage-mode control flyback converters or forward-voltage power converters. It provides the power-supply designer with maximum flexibility and ease of use. The device is specified up to 110V and will operate from as low as 11V. Its maximum operating frequency of 300kHz permits the use of miniature magnetic components to minimize board space. The range, polarity, and range of output voltages and power are limited only by design and by the external components used. This device works in isolated and nonisolated configurations, and in applications with single or multiple output voltages. All the building blocks of a PWM voltage-mode controller are present in the MAX5003 and its settings are adjustable. The functional diagram is shown on Figure 1. Modern Voltage-Mode Controllers The MAX5003 offers a voltage-mode control topology and adds features such as fast input voltage feed forward, programmable maximum duty cycle, and high operating frequencies. It has all the advantages of current-mode control—good control loop bandwidth, same-cycle response to input voltage changes, and pulse-by-pulse current limiting. It eliminates disadvantages such as the need for ramp compensation, noise sensitivity, and the analytical and design difficulties of dealing with two nested feedback loops. In summary, voltage-mode control has inherent superior noise immunity and uses simpler compensation schemes. Internal Power Regulators The MAX5003’s power stages operate over a wide range of supply voltages while maintaining low power consumption. For the high end of the range (+36V to +110V), power is fed to the V+ pin into a depletion junction FET preregulator. This input must be decoupled with a 0.1µF capacitor to the power ground pin (PGND). To decouple the power line, other large-value capacitors must be placed next to the power transformer connection. The preregulator drops the input voltage to a level low enough to feed a first low-dropout regulator (LDO) (Figure 1). The input to the LDO is brought out at the ES pin. ES must also be decoupled with a 0.1µF capacitor. In applications where the maximum input voltage is below 36V, connect ES and V+ together and decouple with a 0.1µF capacitor. The first LDO generates the power for the VDD line. The VDD line is available at the VDD pin for decoupling. The bypass to AGND must be a 5µF to 10µF capacitor. When the maximum input voltage is always below 18.75V, power may also be supplied at VDD; in this case, connect V+, ES, and VDD together. Forcing voltages at VDD above 10.75V (see Electrical Characteristics) disables the first LDO, typically reducing current consumption below 50µA (see Typical Operating Characteristics). Following the VDD LDO is another regulator that drives VCC: the power bus for the internal logic, analog circuitry, and external power MOSFET driver. This regulator is needed because the VDD voltage level would be too high for the external N-channel MOSFET gate. The _______________________________________________________________________________________ 7 MAX5003 Pin Description (continued) MAX5003 High-Voltage PWM Power-Supply Controller VFETBIAS V+ MAX5003 HIGH-VOLTAGE EPIFET 1 LINEAR VES REGULATOR VDD 16 VDD 15 VCC 14 NDRV 13 PGND 12 CS 11 AGND 10 MAXTON AGND VDD CC AGND INDIV 2 1.2V ES LINEAR VDD REGULATOR VCC VINOK VCCOK AGND VINOK UV LOCKOUT VCC REF REFOK BANDGAP REFERENCE SDN AGND 3 “1” VCC FREQ 4 D “D”FF CLK Q CLK FREQ DRIVER NDRV R REFOK PGND SDN INDIV SS 5 VCC RAMP CURRENT SENSE MAXTON VCC AGND C LIMIT REF 0.1V PGND 6 CS BLANK VCC 100ns STRETCHING VCC RAMP PGND SS CON 7 VCON PWM COMP AGND VCC VREF SDN R COMP 8 9 R ERROR AMP AGND Figure 1. Functional Diagram 8 _______________________________________________________________________________________ FB High-Voltage PWM Power-Supply Controller Undervoltage Lockout, Feed Forward, and Shutdown The undervoltage lockout feature disables the controller when the voltage at INDIV is below 1.2V (120mV hysteresis). When INDIV rises higher than 1.2V plus the hysteresis (typically 1.32V), it allows the controller to start. An external resistive divider connected between the power line and AGND generates the INDIV signal. INDIV is also used as the signal for the fast input voltage feed-forward circuit. Always connect INDIV to a voltage divider. It is not a “don’t care” condition; the signal is used to set the fast feed-forward circuit (see the Oscillator and Ramp Generator section). Choose R2 (Figure 2) between 25kΩ to 500kΩ and calculate R1 to satisfy the following equation: V SUL R1 = R2 - 1 VINDIVLO where V SUL = system undervoltage lockout and VINDIVLO = INDIV undervoltage lockout. The undervoltage lockout function allows the use of the INDIV pin as a shutdown pin with an external switch to ground. The shutdown circuit must not affect the resistive divider during normal operation. Current-Sense Comparator The current-sense (CS) comparator and its associated logic limit the current through the power switch. Current is sensed at CS as a voltage across a sense resistor between the external MOSFET source and PGND. Connect CS to the external MOSFET source through a 100Ω resistor or RC lowpass filter (Figures 2 and 3). See CS Resistor in the Component Selection section. A blanking circuit shunts CS to ground when the power MOSFET switch is turned off, and keeps it there for 70ns after turn-on. This avoids false trips caused by the switching transients. The blanking circuit also resets the RC filter, if used. When VCS > 100mV, the power MOSFET is switched off. The propagation delay from the time the switch current reaches the trip level to the driver turn-off time is 240ns. If the current limit is not used, the CS pin must be connected to PGND. Error Amplifier The internal error amplifier is one of the building blocks that gives the MAX5003 its flexibility. Its noninverting input is biased at 1.5V, derived from the internal 3V reference. The inverting input is brought outside (FB pin) and is the regulation feedback connection point. If the error amplifier is not used, connect this pin to ground. The output is available for the frequency compensation network and for connection to the input of the PWM comparator (CON). Unity-gain frequency is 1.2MHz, open-circuit gain is 80dB, and the amplifier is unitygain stable. To eliminate long overload recovery times, there are clamps limiting the output excursions close to the range limits of the PWM ramp. The voltage at the noninverting input of the error amplifier is the regulator set point, but is not accessible. Set-point voltage can be measured, if needed, by connecting COMP and FB and measuring that node with respect to ground. The error amplifier is powered from the VCC rail. PWM Comparator The pulse-width modulator (PWM) comparator stage transforms the error signal into a duty cycle by comparing the error signal with a linear ramp. The ramp levels are 0.5V min and 2.5V max. The comparator has a typical hysteresis of 5.6mV and a propagation delay of 100ns. The output of the comparator controls the external FET. Soft-Start The soft-start feature allows converters built using the MAX5003 to apply power to the load in a controllable soft ramp, thus reducing start-up surges and stresses. _______________________________________________________________________________________ 9 MAX5003 VCC regulator has a lockout line that shorts the N-channel MOSFET driver output to ground if the VCC LDO is not regulating. VCC feeds all circuits except the VCC lockout logic, the undervoltage lockout, and the power regulators. The preferred method for powering the MAX5003 is to start with the high-voltage power source (at V+ or ES, depending on the application), then use a bootstrap source from the same converter with an output voltage higher than the VDD regulator turn-off voltage (10.75V) to power VDD. This will disable the power consumption of the V DD LDO. It is also possible to power the MAX5003 with no bootstrap source from ES or V+, but do not exceed the maximum allowable power dissipation. The current consumption of the part is mostly a function of the operating frequency and the type of external power switch used—in particular, the total charge to be supplied to the gate. A reference output of 3V nominal is externally available at the REF pin, with a current sourcing capability of 1mA. A lockout circuit shuts off the oscillator and the output driver if REF falls 200mV below its set value. Minimize loading at REF, since the REF voltage is the source for the FB voltage, which is the regulator set point when the error amplifier is used. Any changes in VREF will be proportionally reflected in the regulated output voltage of the converter. MAX5003 High-Voltage PWM Power-Supply Controller It also determines power-up sequencing when several converters are used. Upon power turn-on, the SS pin acts as a current sink to reset any capacitance attached to it. Once REF has exceeded its lockout value, SS sources a current to the external capacitor, allowing the converter output voltage to ramp up. Full output voltage is reached in approximately 0.45s/µF. The SS pin is an overriding extra input to the PWM comparator. As long as its voltage is lower than VCON, it overrides VCON and SS determines the level at which the duty cycle is decided by the PWM comparator. After exceeding VCON, SS no longer controls the duty cycle. Its voltage will keep rising up to VCC. Oscillator and Ramp Generator The MAX5003 oscillator generates the ramp used by the comparator, which in turn generates the PWM digital signal. It also controls the maximum on-time feature of the controller. The oscillator can operate in two modes: free running and synchronized (sync). A single pin, FREQ, doubles as the attachment point for the frequency programming resistor and as the synchronization input. The mode recognition is automatic, based on the voltage level at the FREQ pin. In free-running mode, a 1.25V source is internally applied to the pin; the oscillator frequency is proportional to the current out of the pin through the programming resistor, with a proportionality constant of 16kHz/µA. In sync mode, the signal from the external master generator must be a digital rectangular waveform running at four times the desired converter switching frequency. Minimum acceptable signal pulse width is 150ns, positive or negative, and the maximum frequency is 1.2MHz. When the voltage at FREQ is forced above 2.7V, the oscillator goes into sync mode. If left at or below 1.5V for more than 8µs to 20µs, it enters free-running mode. The master clock generator cannot be allowed to stop at logic zero. If the system design forces such a situation, an inverter must be used at the FREQ pin. In sync mode, the oscillator signal is divided by four and decoded. The output driver is blocked during the last phase of the division cycle, giving a hardwired maximum on-time of 75%. In free-running mode, the oscillator duty cycle is 75% on, and the off portion also blocks the output driver. The maximum on-time is then absolutely limited to 75% in either mode. Maximum on-time can be controlled to values lower than 75% by a programming resistor at the MAXTON pin. 10 The PWM ramp generated goes from 0.5V min to 2.5V max, and the maximum time on is the time it takes from low to high. MAXTON is internally driven to VINDIV and a resistor must be connected from MAXTON to AGND, to program the maximum on-time. The ramp slope is directly proportional to VINDIV and inversely proportional to RMAXTON. Since the ramp voltage limits are fixed, controlling the ramp slope sets the maximum time on. Changing the ramp slope while VCON remains constant also changes the duty cycle and the energy transferred to the load per cycle of the converter. The INDIV signal is a fraction of the input voltage, so the fast input voltage feedforward works by modifying the duty cycle in the same clock period, in response to an input voltage change. Calculate the maximum duty cycle as: DMAX = MAXTON T × 100 where: DMAX = Maximum duty cycle (%) MAXTON = Maximum on-time T = Switching period Then: R 1.25V ƒ SW DMAX = 0.75 × 100 MAXTON 200kΩ VINDIV 100kHz where: RMAXTON = Resistor from the MAXTON pin to ground VINDIV = Voltage at the INDIV pin ƒSW = Output switching frequency MAXTON can then be calculated as: MAXTON = 0.75 × RMAXTON × 1.25V 200kΩ × VINDIV × 100kHz N-Channel MOSFET Output Switch Driver The MAX5003 output drives an N-channel MOSFET transistor. The output sources and sinks relatively large currents, supplying the gate with the charge the transistor needs to switch. These are current spikes only, since after the switching transient is completed the load is a high-value resistance. The current is supplied from the V CC rail and must be sourced by a large-value ______________________________________________________________________________________ High-Voltage PWM Power-Supply Controller The driver can source up to 560mA and sink up to 1A transient current with a typical on source resistance of 4Ω. The no-load output levels are VCC and PGND. Applications Information Compensation and Loop Design Considerations The circuit shown in Figure 2 is essentially an energy pump. It stores energy in the magnetic core and the air gap of the transformer while the power switch is on, and delivers it to the load during the off phase. It can operate in two modes: continuous and discontinuous. In discontinuous mode, all the energy is given to the load before the next cycle begins; in continuous mode, some energy is continuously stored in the core. The system has four operating parameters: input voltage, output voltage, load current, and duty cycle. The PWM controller senses the output voltage and the input voltage, and keeps the output voltage regulated by controlling the duty cycle. The output filter in this circuit consists of the load resistance and the capacitance on the output. To study the stability of the feedback system and design the compensation necessary for system stability under all operating conditions, first determine the transfer function. In discontinuous mode, since there is no energy stored in the inductor at the end of the cycle, the inductor and capacitor do not show the characteristic double pole, and there is only a dominant pole defined by the filter capacitor and the load resistance. There is a zero at a higher frequency, defined by the ESR of the output filter capacitor. Such a response is easy to stabilize for a wide range of operating conditions while retaining a reasonably fast loop response. In continuous mode, the situation is different. The inductor-capacitor combination creates a double pole, since energy is stored in the inductor at all times. In addition to the double pole, a right-half-plane zero appears in the frequency response curves. This response is not easy to compensate. It can result in conditional stability, a complicated compensation network, or very slow transient response. To avoid the analytical and design problems of the continuous-conduction mode flyback topology and maintain good loop response, choose a design incorporating a discontinuous-conduction mode power stage To keep the converter in discontinuous mode at all times, the value of the power transformer’s primary inductance must be calculated at minimum line voltage and maximum load, and the maximum duty cycle must be limited. The MAX5003 has a programmable duty-cycle limit function intended for this purpose. Design Methodology Following is a general procedure for developing a system: 1) Determine the requirements. 2) In free-running mode, choose the FREQ pin programming resistor. In synchronized mode, determine the clock frequency (fCLK). 3) Determine the transformer turns ratio, and check the maximum duty cycle. 4) Determine the transformer primary inductance. 5) Complete the transformer specifications by listing the primary maximum current, the secondary maximum current, and the minimum duty cycle at full power. 6) Choose the MAXTON pin programming resistor. 7) Choose a filter capacitor. 8) Determine the compensation network. Design Example 1) 36V < V IN < 72V, V OUT = 5V, I OUT = 1A, ripple < 50mV, settling time ≈ 0.5ms. 2) Generally, the higher the frequency, the smaller the transformer. A higher frequency also gives higher system bandwidth and faster settling time. The trade-off is lower efficiency. In this example, 300kHz switching frequency is the choice to favor for a small transformer. If the converter will be free running (not externally synchronized), use the following formula to calculate the RFREQ programming resistor: 100kHz RFREQ = 200k = 66.7kΩ ƒ SW where: RFREQ = Resistor between FREQ and ground ƒSW = Switching frequency (300kHz) If the converter is synchronized to an external clock, the input frequency will be 1.2MHz. The external clock runs at four times the desired switching frequency. ______________________________________________________________________________________ 11 MAX5003 capacitor (5µF to 10µF) at the VCC pin, since the rail will not support such a load. It is this current, equivalent to the product of the total gate switching charge (from the N-channel MOSFET data sheet), times the operating frequency, that determines the bulk of the MAX5003 power dissipation. MAX5003 High-Voltage PWM Power-Supply Controller +48V (36V TO 72V) XFACOILTRCTX03 VIN 8 2 CMSD4448 LP 65µH 1M 1 2 3 4 5 0.1µF 6 7 8 39k V+ VDD INDIV ES FREQ 7 15 IRFD620S VCC 14 NDRV CS REF AGND CON MAXTON FB 11, 12 +5V 1A MBRS130L 10µF PGND COMP 5 16 MAX5003 SS 4.7µF 0.1µF 33µF 13 12 22µF 11 22µF 100Ω 10 RA 41.2k 9, 10 9 62k 51k 0.1µF 470nF 0.1µF CF 390pF RCS 0.1Ω RF 200k RB 17.4k 0V Figure 2. Application Example 1: Nonisolated +48V to +5V Converter 3) The main factors influencing the choice of the turns ratio are the switch breakdown voltage and the duty cycle. With a smaller turns ratio, the secondary reflected voltage and the maximum voltage seen by the switch during flyback are reduced, which is favorable. On the other hand, a smaller turns ratio will shorten the duty cycle and increase the primary RMS current, which can impact efficiency. A good starting figure is the ratio of the input voltage to the output voltage, rounding to the nearest integer. To keep the flyback voltage under control, choose an 8to-1 ratio for the 48V to 5V system. The maximum duty cycle allowed without putting the device in continuous-conduction mode can be found using the following formula: 1 DCMAX = V MIN +1 VSEC × N where: N = NP/NS = Turns ratio VSEC = Secondary voltage VMIN = Minimum power-line voltage For a 48V to 5V system with an 8-to-1 turns ratio, the maximum duty cycle before putting the device in discontinuous mode is 55%. Assume that VIN min is 36V (minimum input voltage, neglecting drops in the power switch and in the resistance of the primary coil) and VSEC is 5.4V (5V plus a Schottky diode drop). The MAX5003 maximum duty cycle is internally limited to 75%. Generally this parameter must fall between 45% to 65% to obtain a balance between efficiency and flyback voltage while staying out of continuous conduction. If the value exceeds these bounds, adjust the turns ratio. 4) Assuming 80% efficiency, a 6.25W input is needed to produce a 5W output. Set an operating duty cycle around 12% below the maximum duty cycle to allow for component variation: 55% - 12% = 43%. Use the following formula to calculate the primary inductance: LPRI = (DC × VMIN ) 2 2 × PWRIN × ƒ SW = (0.43 × 36V) 2 × 6.25W × 300kHz DCMAX = Maximum duty cycle 12 2 ______________________________________________________________________________________ ≅ 65µH High-Voltage PWM Power-Supply Controller MAX5003 0V XFACOILTRCTX03 CMSD4448 2 LP 65µH 8 IRFD620S R1 1M 4.7µF 7 1 2 3 0.1µF 4 5 6 7 8 R2 39k VDD V+ 5 15 VCC 14 NDRV INDIV ES FREQ 16 PGND SS CS REF AGND CON MAXTON COMP FB +5V 1A 11, 12 10µF MAX5003 MBRS130L 33µF 0.1µF 13 12 22µF 11 100Ω 51Ω 22µF 9, 10 - 10 9 680Ω 62k 6 240k 7 0.01µF 1 1.3k 5 MDC217 470nF 0.1µF 2 0.1µF 3900pF 24.9k 51k VIN RCS 0.1Ω TL431 24.9k -48V -36V TO -72V Figure 3. Application Example 2: Isolated -48V to +5V Converter where: DC = Duty cycle. Set to calculated minimum duty cycle at VMIN. PWRIN = Input power, at maximum output power This gives an inductance value (LPRI) of approximately 65µH. 5) The other parameter that defines the transformer is peak current. This is given by: IPRI = 2 × PWRIN LPRI × ƒ SW = × 6.25W 65µH × 300kHz 2 = 0.8A The peak secondary current is the peak primary current multiplied by the turns ratio, or 0.8A · 8 = 6.4A. Calculating the minimum duty cycle: DC(MIN) = DC(MAX) × VIN(MIN) VIN(MAX) = 43% × 36V 72V = With these numbers, the transformer manufacturer can choose a core. 6) For this application, the MAX5003 must be programmed for a maximum duty cycle of 55% at 36V. The MAX5003 will automatically scale the limit with the reciprocal of the input voltage as it changes. The duty-cycle limit for an input voltage of 72V will be 27% (half of 55%). The duty cycle needed to stay out of continuous conduction at 72V is 37%, so there is a 10% margin. The maximum duty time scales with the voltage at the undervoltage lockout pin, VINDIV. The voltage at INDIV is set by selecting the power line undervoltage lockout trip point. The trip point for this system, running from 36V to 72V, is 32V. Then INDIV must be connected to the center point of a divider with a ratio of 32/1.25, connected between the power line and ground. Then RMAXTON is: V 100kHz DCMAX (VMIN ) RMAXTON = MIN 200kΩ 75% VUVL ƒ SW 36V 100kHz 55% = 200kΩ = 55kΩ 32V 300kHz 75% ______________________________________________________________________________________ 13 High-Voltage PWM Power-Supply Controller MAX5003 where: RMAXTON = Resistor between the MAXTON pin and ground VMIN = Minimum power-line voltage VUVL = Power-line trip voltage DCMAX(VMIN) = Maximum duty cycle at minimum power-line voltage For this application circuit, a 10% margin is reasonable, so the value used is 50kΩ. This gives a maximum duty cycle of 50%. The maximum duty cycle can now be expressed as: V - 0.5V DC(VCON,VIN ) = CON 2.0V V - 0.5V CON ≈ 2.0V V ƒ SW MIN × DCMAX(VMIN) V ƒ IN NOM 36V ƒ SW 50% VIN ƒ NOM where: VCON = Voltage at the CON pin, input of the PWM comparator DC(VCON, VIN) = Duty cycle, function of VCON and VIN 0.5V and 2.5V are the values at the beginning and end of the PWM ramp. The term ƒSW / ƒNOM varies from 0.8 to 1.2 to allow for clock frequency variation. If the clock is running at 300kHz and the input voltage is fixed, then the duty cycle is a scaled portion of the maximum duty cycle, determined by VCON. V - 0.5V DC(VCON,VMIN ) = CON 50% 2.0V V - 0.5V DC(VCON,VMAX ) = CON 25% 2.0V DC(2.5V,VMIN ) = 50% DC(2.5V,VMAX ) = 25% DC(0.5V,VMIN ) = 0 DC(0.5V,VMAX ) = 0 7) Low-ESR/ESL ceramic capacitors were used in this application. The output filter is made by two 22µF ceramic capacitors in parallel. Normally, the ESR of a capacitor is a dominant factor determining the ripple, but in this case it is the capacitor value. Calculating IOUT ƒ SW × C 14 = 1A 300kHz × 44 µF = 76mV the ripple will be a fraction of this depending on the duty cycle. For a 50% duty cycle, the ripple due to the capacitance is approximately 45mV. 8)The PWM gain can be calculated from: APWM = dVOUT dVCON = = V MIN DC MAX(VMI 2 × LPRI × ƒ SW 2.0V RL 36V 50% ≅ 3 2 × LPRI × ƒ SW 2.0V RL Note that while the above formula incorporates the product of the maximum duty cycle and VIN, it is independent of VIN. For 1A output (RL = 5Ω), the PWM gain is +3.0V/V. For a 10% load (RL = 50Ω), the gain is multiplied by the square root of 10 and becomes +10V/V. The pole of the system due to the output filter is 1 / 2πRC, where R is the load resistance and C the filter capacitor. Choosing a capacitor and calculating the pole frequency by: 1 1 ƒP = = 2π × 5Ω × 44 µF 2π × R L × C L it is 723Hz at full load. At 10% load it will be 72Hz, since the load resistor is then 50Ω instead of 5Ω. The total loop gain is equal to the PWM gain times the gain in the combination of the voltage divider and the error amplifier. The worst case for phase margin is at full load. For a phase margin of 60 degrees, this midband gain (G) must be set to be less than: G < ƒ UErrorAmp tan(PM) × APWM × ƒ P = 1 MHz 1.7 × 3 × 723Hz where: ƒU = Unity-gain frequency of error amplifier PM = Phase margin angle The DC accuracy of the regulator is a function of the DC gain. For 1% accuracy, a DC gain of 20 is required. Since the maximum midband gain for a stable response is 16, an integrator with a flat midband gain given by a zero is used. The midband gain is less than 16, to preserve stability, and the DC gain is much larger than 20, to achieve high DC accuracy. Optimization on the bench showed that a midband gain of 5 gave fast transient response and settling with no ringing. The zero was pushed as high in frequency as possible without losing stability. The zero must be a factor of two or so below the system unity-gain frequency (crossover frequency) at minimum load. With the ______________________________________________________________________________________ High-Voltage PWM Power-Supply Controller desired value, the center-point voltage will be 1.5V. The Thevenin equivalent of the resistors must be low enough so the error amplifier bias current will not introduce a division error. The two resistors must have similar temperature coefficients (tempcos), so the dividing ratio will be constant with temperature. RB / (RA + RB) = VSET / VOUT CS Resistor The CS resistor is connected in series with the source of the N-channel MOSFET and ground, sensing the switch current. Its value can be calculated from the following equation: Since VSET = 1.5V and VOUT = 5V, RA is set to 41.2kΩ and RB to 17.4kΩ. The midband gain is the ratio of RF/RA. RB does not affect the gain because it is connected to a virtual ground. For a midband gain of 5, the feedback resistor equals 200kΩ. To set the zero at 2kHz, the capacitor value is: CF = 1 / (2π x RF x fz ) = 400pF Layout Recommendations All connections carrying pulsed currents must be very short, be as wide as possible, and have a ground plane behind them whenever possible. The inductance of these connections must be kept to an absolute minimum due to the high di/dt of the currents in highfrequency switching power converters. In the development or prototyping process, multipurpose boards, wire wrap, and similar constructive practices are not suitable for these type of circuits; attempts to use them will fail. Instead, use milled PC boards with a ground plane, or equivalent techniques Current loops must be analyzed in any layout proposed, and the internal area kept to a minimum to reduce radiated EMI. The use of automatic routers is discouraged for PC board layout generation in the board area where the high-frequency switching converters are located. Designers should carefully review the layout. In particular, pay attention to the ground connections. Ground planes must be kept as intact as possible. The ground for the power-line filter capacitor and the ground return of the power switch or currentsensing resistor must be close. All ground connections must resemble a star system as much as practical. “Short” and “close” are dimensions on the order of 0.25in to 0.5in (0.5cm to about 1cm). Setting the Output Voltage The output voltage of the converter, if using the internal error amplifier, can easily be set by the value of the FB pin set voltage. This value is 1.5V. A resistive divider must be calculated from the output line to ground, with a dividing ratio such that when the output is at the Component Selection RCS = 100mV ILIM (PRI) = 100mV 2 PWROUT(MAX) × K TOL LPRI × ƒ SW × η where η = efficiency and 0.5 < KTOL < 0.75. KTOL includes the tolerance of the sensing resistor, the dispersion of the MAX5003 CS trip point, and the uncertainties in the calculation of the primary maximum current. The sensing resistor must be of the adequate power dissipation and low tempco. It must also be noninductive and physically short. Use standard surface-mount CS resistors. A 100Ω resistor is recommended between the CS resistor and the CS pin. If the current surge at the beginning of the conduction period is large and disrupts the MAX5003’s operation, add a capacitor between the CS pin and PGND, to form an RC filter. Power Switch The MAX5003 will typically drive an N-channel MOSFET power switch. The maximum drain voltage, maximum RDS(ON), and total gate switching charge are the parameters involved in choosing the FET. The maximum gate switching charge is the most important factor defining the MAX5003 internal power consumption, since the product of the switching frequency and the total gate charge is the IC current consumption. RDS(ON) is the parameter that determines the total conduction power losses in the switch, and the choice depends on the expected efficiency and the cooling and mounting method. The maximum drain voltage requirements can be different depending on the topology used. In the flyback configuration, the maximum voltage is the maximum supply voltage plus the reflected secondary voltage, any ringing at the end of the conduction period, and the spike caused by the leakage inductance. In the case of the forward converter, the reset time of the core will set the maximum voltage ______________________________________________________________________________________ 15 MAX5003 zero at 2kHz, the crossover frequency is 4kHz and the phase margin is 50°. Given the above considerations, RA, RB , RF, and CF can be chosen (Figure 2). The sum of RA and RB is chosen for low current drain. In the example, RA plus RB is 58kΩ and draws 80µA. The following ratio sets the output voltage: MAX5003 High-Voltage PWM Power-Supply Controller stress on the switch. A FET with the lowest total charge and the lowest RDS(ON) for the maximum drain voltage expected (plus some safety factor) is the best choice. The choice of package is a function of the application, the total power, and the cooling methods available. Transformer Transformer parameters, once calculated in the design process, can be used to find standard parts whenever possible. The most important factors are the saturation current, primary inductance, leakage inductance turns ratio, and losses. Packaging and EMI generation and susceptibility are closely connected, and must be considered. In general, parts with exposed air gaps (not contained inside the magnetic structure) will generate the most radiated EMI, and might need external shielding. If the design is in high-voltage power supplies, the insulation specifications are also important. Pay close attention if the circuitry is galvanically connected to the mains at any point, since serious safety and regulatory issues might exist. Capacitors As in any high-frequency power circuit, the capacitors used for filtering must meet very low ESR and ESL requirements. At the 300kHz frequency (of which the MAX5003 is capable), the most favorable technologies are ceramic capacitors and organic semiconductor (OS CON) capacitors. The temperature dependence of the capacitance value and the ESR specification is important, particularly if the ESR is used as part of the compensation network for the feedback loop. If using through-hole- mounted parts, keep lead length as short as practical. Components with specifications for switching power converters are preferred. Decoupling capacitors must be mounted close to the IC. Diodes The choice of rectifier diodes depends on the output voltage range of the particular application. For low-voltage converters, the diode drop is a significant portion of the total loss, and must be kept to a minimum. In those cases, Schottky diodes are the preferred component for the design. At higher voltages, ultra-fast recovery diodes must be used, since Schottky components will not satisfy the reverse voltage specification. For all cases, the specifications to be determined before choosing a diode are the peak current, the average current, the maximum reverse voltage, and the maximum acceptable rectification losses. Once a type is identified, a thermal analysis of the diode losses vs. total thermal resistance (from junction to ambient) must be carried out if the total power involved is significant. Industrial-frequency (60Hz) rectifiers are not recommended for any function in these converters, due to their high capacitance and recovery losses. If using overdimensioned rectifiers, the junction capacitance influence must be reviewed. ___________________Chip Information TRANSISTOR COUNT: 1050 SUBSTRATE CONNECTED TO GND Table 1. Component Manufacturers DEVICE TYPE MANUFACTURER PHONE FAX International Rectifier 310-322-3333 310-322-3332 Fairchild 408-822-2000 408-822-2102 Dale-Vishay 402-564-3131 402-563-6418 Motorola 303-675-2140 303-675-2150 Central Semiconductor 516-435-1110 516-435-1824 Central Semiconductor 516-435-1110 516-435-1824 Sanyo 619-661-6835 619-661-1055 Taiyo Yuden 408-573-4150 408-573-4159 AVX 803-946-0690 803-626-3123 Coiltronics 561-241-7876 561-241-9339 Power FETs Current-Sense Resistors Diodes Transistors Capacitors Coils Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. 16 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 © 2002 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products. 19-1914; Rev 1; 3/02 MAX5003-50W Evaluation Kit The MAX5003 50W forward converter evaluation kit (EV kit) provides a regulated +5V output voltage at currents up to 10A, when operated from a +36V to +72V input voltage range. This EV kit is fully assembled and tested. The output voltage is preset to +5V. A single-transistor forwardconverter topology with a reset winding is used for high output power and high efficiency. The use of an optocoupler in the feedback circuit provides full 1500V primary to secondary galvanic isolation. A bottom-mounted heatsink plate safely dissipates the heat generated by the power MOSFET and the output diode. The power supply is designed to fit into a small footprint. WARNING: Dangerous voltages are present on this EV kit and on equipment connected to it. Users who power-up this EV kit or power the sources connected to it must be careful to follow safety procedures appropriate to working with high-voltage electrical equipment. Under severe fault or failure conditions, this EV kit may dissipate large amounts of power, which could result in the mechanical ejection of a component or of component debris at high velocity. Operate this EV kit with care to avoid possible personal injury. Features ♦ +5V at 10A Output ♦ ±36V to ±72V Input Voltage Range ♦ 250kHz Switching Frequency ♦ Fully Isolated Design with 1500V Isolation Built into the Transformer ♦ Fully Assembled and Tested Board with Minimum PC Board Footprint ♦ 0.3% typical Line and Load Regulation ♦ 85% typical Efficiency at 25W Ordering Information PART TEMP RANGE IC PACKAGE MAX5003EVKIT50W 0°C to +50°C* 16 SO *With air flow. Component List DESIGNATOR C1, C3, C10, C15 QTY 4 DESCRIPTION 0.1µF ceramic caps (0805) DESIGNATOR R1 QTY 1 C2 1 470pF ceramic cap (0805) R2 1 39.2kΩ ±1% resistor (0805) C4, C5, C6 3 0.47µF, 100V ceramic caps (2220) R3 1 80.6kΩ ±1% resistor (0805) R4 1 1.24kΩ ±1% resistor (0805) 560µF, 6.3V electrolytic capacitors Nichicon UPW0J561MPH 47nF ceramic capacitors (0805) R5 1 56kΩ ±1% resistor (0805) R6 1 0.02Ω resistor Dale-Vishay WSL1206 0.02Ω ±1.0% R86 C7, C13, C14 3 DESCRIPTION 1MΩ ±1% resistor (0805) C8, C9 2 C11 1 22nF ceramic capacitor (0805) R8 1 100Ω ±5% resistor (0805) C12 1 1nF, 100V ceramic capacitor (0805) R9 1 470Ω ±5% resistor (0805) C16 1 4.7nF, 1500V ceramic capacitor R11, R12 2 10kΩ ±1% resistors (0805) 1 20Ω ±5% resistor (1206) D3 1 200mA, 100V diode Panasonic MA111CT R13 R14 1 10kΩ ±5% resistor (0805) D4 1 20A, 40V low forward voltage Schottky diode General Semi SBL2040CT 1 200mA, 200V, diode Panasonic MA115CT Q1 1 200V MOSFET, Rds = 0.18Ω International Rectifier IRF640N Q2 1 NPN transistor, FMMT3904 D5 R15 1 240kΩ ±5% resistor (0805) R16 1 1Ω ±5% resistor (0805) L1 1 4.7µH inductor Coiltronics HC2-4R7 T1 1 Transformer (12-pin gull wing) Coiltronics CTX03-14856 ________________________________________________________________ Maxim Integrated Products For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at 1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com. 1 Evaluates: MAX5003 General Description Evaluates: MAX5003 MAX5003-50W Evaluation Kit Component List (continued) DESIGNATOR QTY U2 1 DESCRIPTION Optocoupler QT Optoelectronics MOC217 U3 1 Shunt regulator TL431AID U1 1 MAX5003ESE, 16-pin narrow SO 1 15V Zener diode Panasonic MA8150 Z1 3) 4) 5) 6) Component Suppliers PHONE FAX Coiltronics SUPPLIER 561-241-7876 561-241-9339 Dale-Vishay 402-564-3131 402-563-6418 General Semiconductor 631-847-3000 631-847-3236 International Rectifier 310-322-3331 310-322-3332 Nichicon 847-843-7500 847-843-2798 Panasonic 201-392-7522 201-392-4441 QT Optoelectronics 408-720-1440 408-720-0848 Quick Start The MAX5003 50W EV kit is fully assembled and tested. The power supply has full isolation between the primary and secondary circuit. A heatsink is included at the noncomponent side for heatsinking the power MOSFET and the output dual diode D4. During normal operation at full output current, this heatsink becomes hot. A small fan with direct airflow towards this heatsink is recommended to keep the temperature rise to acceptable levels. This power supply is not fused at the input. For added protection, a 3A to 5A fuse should be used at the input. Appropriately sized heavy-gauge wires should be used to connect the power supply to the EV kit and load. Follow these steps to verify board operation. Do not turn on the power supply until all connections are made. 1) Connect a 220µF bulk storage capacitor at the input terminals of the EV kit. This capacitor should be rated for 100V and be able to handle 1.5A of ripple current. 2) Connect a +36V to +72V power supply to the pads labeled VIN. The positive power-supply terminal should connect to +VIN and the negative powersupply terminal should connect to -VIN. The power 2 supply must be rated to at least 3A. The input voltage to the MAX5003 EV kit should not exceed 80V at any time. Connect a variable load capable of sinking at least 10A at 5V and a voltmeter to the pads labeled +VO and -VO. Set the load current to approximately 5A. Turn on the input power and verify that the output voltage is +5V. To evaluate the load regulation of the EV kit, vary the load from 0 to 10A and record the output voltage variation as needed. For best measurement accuracy, the voltmeter must be connected right to the output pads of the EV kit. 7) To evaluate the line regulation of the EV kit, vary the input voltage from +36V to +72V and record the output voltage. Note: The MAX5003 EV kit undervoltage lockout circuitry has been designed to shut down when the input supply voltage is under 32V. Power Supply Typical Specifications Table 1 summarizes the typical performance of the 50W power supply. Table 1. Typical Specifications Output Power 50W Input Voltage (VIN) ±36V to ±72V Output Voltage (VOUT) +5V Output Current (IOUT) 10A Initial Output Accuracy ±3%* Output Voltage Regulation 0.3%, over line and load Efficiency 85% at 48V and 25W Input Output Isolation 1500V for 1s Switching Topology Feedforward Compensated Forward Converter Dimensions 4.05in x 1.3in *Initial setpoint accuracy can be improved by using tighter tolerance resistor divider (R11 and R12). _______________________________________________________________________________________ MAX5003-50W Evaluation Kit 70 MAX5003EV fig03 MAX5003EV fig01 80 0.20 0.15 60 0.10 VOLTS EFFICIENCY (%) 0.25 Evaluates: MAX5003 90 50 40 0.05 0 30 -0.05 20 -0.10 10 -0.15 0 0 10 20 30 OUTPUT POWER (W) 40 50 Figure 1. Efficiency vs. Output Power 10µs/div Figure 3. Output Transient Response (IOUT: 10A to 0.8A) MAX5003 fig04 MAX5003EV fig02 5.5 5.4 5.3 5.1 VOUT (V) VOUT (V) 5.2 5.0 1V/div 4.9 4.8 4.7 4.6 4.5 0 2 4 6 8 10 2ms/div IOUT (A) Figure 2. Output Voltage Regulation vs. Output Current Power-Supply Performance Key performance characteristics of the power supply include efficiency and output voltage regulation. Figure 1 shows the efficiency vs. output power. The efficiency reaches 85% at about 25W of output power and stays relatively flat up to 50W. Even though the efficiency is very high, heatsinking is required for the power MOSFET and output diode. The diode will dissipate about 6W with a 10A output current and the MOSFET can be expected to dissipate about 3W to 4W at full 50W load. Sufficient airflow over the power supply is recommended to cool down the power transformer and output inductor. Figure 2 shows the output voltage regulation of the power supply from 0 to 10A of output current. Voltage measurement was done across the output voltage sense points +VO and -VO. Figure 4. Output Voltage Transient At Power-Up (VIN = 48V, IOUT = 5A) Another interesting performance waveform for power supplies is the output voltage transient response to a step change in output current. Figure 3 shows load transient response when the load is stepped from 10A to 0.8A. As can be seen from Figure 3, the initial transient response time is less than 30µs. This is a side benefit of using an optocoupler in conjunction with a TL431 shunt regulator for isolation. Figure 4 shows the well-behaved startup characteristics of this power supply, which are characterized by the monotonic rise of the output voltage as well as the absence of any overshoots at the end of the rise period. _______________________________________________________________________________________ 3 MAX5003 fig05 VDS(V) Evaluates: MAX5003 MAX5003-50W Evaluation Kit 50V/div 5V/div 400ns/div Figure 5. Drain-Source Voltage Waveform The Power Circuit Topology Among the several power topologies available, the single-transistor forward topology offers a simple and lowcost solution and provides very good efficiency throughout the operating power range. However, this topology requires a transformer reset winding connected to pins T1–3 and T1–4 (Figure 7). The forward converter was chosen because it offers higher power density and higher efficiency than a flyback converter at these power levels. Transformer T1 provides 1500V isolation between primary and secondary. Efficiency is further improved by powering the control circuit from a primary bias winding (T1–5, T1–6, Figure 7) after initial startup. A 250kHz switching frequency was selected to allow small form-factor transformer, inductor, and output capacitors. Key Operating Waveforms Key operating waveforms are always useful in understanding the operation of switching power supplies. A 10× oscilloscope probe is necessary for effective probing. A digital scope is very useful in capturing startup sequences. However, extreme caution should be exercised when probing live power supplies. For example, shorting the drain-source terminals of Q1 while power is applied is sure to produce a big spark and may damage the EV kit. Figure 5 shows the drain-to-source waveform of Q1. Notice the leading-edge voltage spike. This is a result of the energy stored in transformer T1’s leakage inductance. Figure 6 shows the voltage at the output of the secondary rectifier (cathode of D4). 4 MAX5003 fig06 200ns/div Figure 6. Waveform at Cathode of D4 PC Board Layout and Component Placement As with any other switching power supply, component placement is very important. Because of the primary-tosecondary isolation, the primary and secondary grounds are separated. Figure 10 clearly shows the separation on both sides of the PC board. The layout of the board can be changed to accommodate different footprints. Also, the power MOSFET and output rectifier should be mounted on a heatsink for best thermal management. In this implementation, both of these components are on the noncomponent side of the board, with their tabs mounted to the heatsink plate. The critical layout considerations are as follows: • Distance from the secondary transformer leads to diode D4 should be kept to a minimum. This will improve EMI as well as the effective available power transfer. • Bypass capacitors C4, C5, and C6 should be as close as possible to T1–1. • The PC board trace connecting T1–2 to the drain of Q1 should be as short as possible. • The current-sense resistor R6 should be as close as possible to the source of Q1 and should return with a very short trace either to the ground plane or to the negative lead of bypass capacitors C4, C5, and C6. • The gate-drive loop, consisting of pin 14 of MAX5003, R16, Q1, R6, and pin 13 of the MAX5003, must be kept as short as possible and preferably routed over a ground plane. • Relevant trace spacing (relating to trace creepage) must be observed according to applicable safety agency guidelines. _______________________________________________________________________________________ VDD GND GND GND GND GND GND -VIN +VIN GND GND C3 0.1µF R3 80.6kΩ 1% R2 39.2kΩ 1% R1 1MΩ 1% R15 240kΩ R4 1.24kΩ 1% C2 470pF C1 0.1µF Q2 GND 5 6 U2 8 7 6 5 4 3 2 1 GND R14 10kΩ 7 COMP CON REF SS FREQ ES INDIV V+ U1 C4 0.47µF 100V VCC VDD 4 3 2 1 2 1 NC A A C U3 C10 0.1µF 10 11 12 13 FB 9 MAXTON AGND CS PGND 14 15 NC A A R C9 47nF VDD C6 0.47µF 100V 16 GND NDRV SGND MAX5003 U1 GND C5 0.47µF 100V GND C8 47nF 5 6 7 8 SGND R5 56kΩ 1% R8 100Ω R16 1Ω C11 22nF D3 MA111CT GND GND GND GND GND GND SGND R12 10kΩ 1% GND D5 MA115 5T 12T R11 10kΩ 1% R9 470Ω T1 R6 0.02Ω 1% Q1 GND 2 14T 1 5 4 6 3 5T GND 8 9 12 11 C16 4.7nF 1500V C12 1nF 100V R13 20Ω D4 C7 560µF 6.3V L1 4.7µH + C14 560µF 6.3V + SGND C15 0.1µF SGND: DENOTES SECONDARY GROUND + C13 560µF 6.3V SGND +VO -VO Evaluates: MAX5003 Z1 MAX5003-50W Evaluation Kit Figure 7. MAX5003 50W EV Kit Schematic _______________________________________________________________________________________ 5 Evaluates: MAX5003 MAX5003-50W Evaluation Kit 1.0" Figure 8. MAX5003-50W EV Kit PC Board Layout—Component Side 1.0" Figure 9. MAX5003-50W EV Kit Component Placement Guide—Component Side. Note: Q1 and D4 are placed on the bottom side where their metal tabs are exposed to heatsink plate. 1.0" Figure 10. MAX5003-50W EV Kit PC Board Layout—Solder Side Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. 6 _____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 © 2002 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products. 19-1914; Rev 1; 1/01 MAX5003 Evaluation Kit Features ♦ 5V at 1A Output The MAX5003 EV kit is a fully assembled and tested surface-mount printed circuit (PC) board. It comes with the output voltage set to 5V. This EV kit is configured as a flyback converter, and can easily be configured for either isolated or nonisolated operation by selecting the state of a mechanical switch. Additionally, for systems in which the input and output ground references are not at the same potential and for which isolation is not desired, the user has the option to install a level-shifter (not supplied) in the controller feedback loop. WARNING: Dangerous voltages are present on this EV kit and on equipment connected to it. Users who power up this EV kit or power the sources connected to it must be careful to follow safety procedures appropriate to working with high-voltage electrical equipment. ♦ 300kHz Switching Frequency Under severe fault or failure conditions, this EV kit may dissipate large amounts of power, which could result in the mechanical ejection of a component or of component debris at high velocity. Operate this kit with care to avoid possible personal injury. *With air flow. ♦ +36V to +72V Input Voltage Range ♦ Can be Configured for -48V Input and +5V Output ♦ Selectable Isolated or Nonisolated Operation ♦ Proven PC Board Layout ♦ Fully Assembled and Tested Surface-Mount Board Ordering Information PART TEMP. RANGE MAX5003EVKIT 0°C to +70°C* IC PACKAGE 16 QSOP Component List DESIGNATION QTY DESCRIPTION DESIGNATION QTY DESCRIPTION 1 33µF, 100V electrolytic capacitor Sanyo 100MV33CZ Q2 1 C2, C3 2 22µF, 10V ceramic capacitors Taiyo Yuden LMK432BJ226MM 2N3904-type NPN transistor Central Semiconductor CMPT3904 or equivalent R1 1 41.2kΩ ±1% resistor C5 C6 C7 1 1 1 2200pF ±10% ceramic capacitor 3900pF ±10% ceramic capacitor 0.01µF ceramic capacitor R2 1 17.4kΩ ±1% resistor R3 1 68kΩ ±5% resistor R4, R22, R23 3 1MΩ ±5% resistors C8 1 10µF, 16V ceramic capacitor Taiyo Yuden EMK325BJ106MN R5 1 39kΩ ±5% resistor R6 1 51kΩ ±5% resistor R7 R8, R15 1 2 200kΩ ±5% resistor 43Ω ±5% resistors DESIGNATION C1 C9 1 100pF ±10% ceramic capacitor C10 1 0.47µF ceramic capacitor C11, C13, C17 3 0.1µF ceramic capacitors C12 1 390pF ±10% ceramic capacitor R9 1 0.11Ω ±1%, 1/4W resistor Dale WSL-1206/0.11Ω/1% C16 1 4.7µF, 25V tantalum capacitor AVX TAJB475M025 D1 1 30V, 1A Schottky diode Fairchild MBRS130L 1 1 1 1 100Ω ±5% resistor 100kΩ ±5% resistor 20kΩ ±5% resistor 1.3kΩ ±5% resistor D2 1 Small-signal switching diode Central Semiconductor CMSD4448 R10 R11 R12 R13 N1 1 200V, 5.2A N-channel MOSFET International Rectifier IRF620S Q1 0 Not installed R14 1 240kΩ ±5% resistor R16, R17 2 24.9kΩ ±1% resistors R18 0 Not installed ________________________________________________________________ Maxim Integrated Products For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at 1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com. 1 Evaluates: MAX5003 General Description The MAX5003 evaluation kit (EV kit) provides a regulated 5V output voltage up to 1A while operating from a +36V to +72V input voltage. Evaluates: MAX5003 MAX5003 Evaluation Kit Component List (continued) DESIGNATION QTY DESCRIPTION R19 0 Not installed R20 1 680Ω ±5% resistor R21 1 15Ω ±5% resistor SW1 SW2 1 1 DPDT switch SPDT switch T1 1 65µH, 8:1:2.5 transformer Coiltronics CTX03-14502 U1 1 MAX5003EEE (16-pin QSOP) U2 1 2.5V voltage reference Motorola TL431BCD U3 1 Low-current optocoupler QT Opto MOC217 None None None 1 1 1 MAX5003 PC board MAX5003 data sheet MAX5003 EV kit data sheet Table 1. Switch/Jumper Functions FUNCTION Nonisolated, Non-level-shifted Feedback Mode (e.g., +48V input and +5V output) Nonisolated, Level-Shifted Feedback Mode (e.g., -48V input and +5V output) Isolated Feedback Mode (input and output supplies isolated from one another) SWITCH/ JUMPER POSITION SW1 NON SW2 DIR JU1 Closed* (default trace) SW1 NON SW2 LVL JU1 Open (Cut) SW1 ISO SW2 LVL JU1 Open (Cut) *Default setting Component Suppliers PHONE FAX AVX 803-946-0690 803-626-3123 3) Verify that switch SW1 is set to the NON position and SW2 is set to the DIR position. See Table 1 for a description of the switch settings. Central Semiconductor 516-435-1110 516-435-1824 4) Turn on the power and verify that the output voltage is +5V. Coiltronics 561-241-7876 561-241-9339 Dale-Vishay 402-564-3131 402-563-6418 Fairchild 408-822-2000 408-822-2102 International Rectifier 310-322-3331 310-322-3332 Motorola 303-675-2140 303-675-2150 QT Optoelectronics 408-720-1440 408-720-0848 SUPPLIER Sanyo 619-661-6835 619-661-1055 Taiyo Yuden 408-573-4150 408-573-4159 Note: Please indicate that you are using the MAX5003 when contacting the above component suppliers. Quick Start The MAX5003 EV kit is fully assembled and tested. Follow these steps to verify board operation in nonisolated mode. Do not turn on the power supply until all connections are completed. 1) Connect a +36V to +72V power supply to the pad labeled VIN. Do not exceed 100V input voltage. The ground connects to the GND pad (-48V). 2) Connect a voltmeter and load (if any) to the +5V pad. 2 5) Refer to the Isolated Feedback section to modify the board for isolated operation. Refer to the Nonisolated Level-Shifted Feedback section to modify the board for operation with the input and output negative supplies at different potentials. Detailed Description Feedback Mode Selection Switch SW1 selects the feedback configuration (isolated or nonisolated). If SW1 is set to the NON position, switch SW2 selects either direct feedback (for the case in which the input and output share the same ground) or level-shifted feedback. Switch SW2 is only effective when nonisolated feedback is selected. Jumper JU1 determines whether the input and output ground references are connected. Table 1 summarizes switch and jumper functions. Do not operate switches SW1 and SW2 when power is applied to the EV kit because the controller can be damaged. Isolated Feedback To configure the MAX5003 EV kit for isolated operation, turn off the power supply and cut the JU1 PC board trace. Set the SW1 switch to the ISO position, and set the SW2 switch to the LVL position (setting SW2 to the _______________________________________________________________________________________ MAX5003 Evaluation Kit Nonisolated Level-Shifted Feedback To configure the MAX5003 EV kit for operation in a system in which the negative terminal of the input power supply is at a more negative potential than the negative terminal of the output power supply (for example, in a -48V input to +5V output application), first turn off the power supply and cut the JU1 PC board trace. Set the SW1 switch to the NON position, and set the SW2 switch to the LVL position. Locate parts R18, R19, and Q1 (directly above jumper JU1 on the PC board). Solder the following parts into the R18, R19, and Q1 locations: R18 = 36.5kΩ ±1% resistor (1206), R19 = 12.4kΩ ±1% resistor (1206), and Q1 = 60V 2N2907type PNP transistor (SOT23). Note that the initial DC output voltage accuracy and the temperature variation will be degraded in this configuration. Do not operate switches SW1 and SW2 when power is applied to the EV kit because the controller can be damaged. high signal to turn on transistor Q2. For normal operation, the SHDN pad can be connected to ground or left unconnected. Note that the logic-high signal used to drive the SHDN pad is referenced to the negative side of the input supply. For more details regarding undervoltage lockout, refer to the MAX5003 data sheet. Current Limiting The MAX5003 EV kit has a current-limiting feature implemented by current-sense resistor R9. The MAX5003 turns off switching FET N1 when the voltage at the CS pin reaches 100mV. Since R9 is a 0.11Ω resistor, this limits the current in the transformer primary to 0.91A peak, which corresponds to a typical output short-circuit current of 4.5A. R10, a 100Ω resistor, is connected between the current-sense resistor and the CS pin to enable current-sense blanking after N1 is turned on, as described in the MAX5003 data sheet. Layout Considerations The MAX5003 EV kit layout is optimized for fast switching and high currents. The input and output power and ground traces must both be as short and wide as possible to minimize unwanted parasitic inductance. This board was not designed per UL spacing specifications. Undervoltage Lockout and Shutdown The MAX5003 EV kit is configured to go into undervoltage lockout when VIN drops below 32V. The MAX5003 does not have a shutdown pin, but the undervoltage lockout state is equivalent to a shutdown state. The MAX5003 EV kit contains a shutdown function consisting of an NPN switching transistor (Q2) that can pull the VINDIV pin to ground. To place the MAX5003 in undervoltage lockout, drive the SHDN pad with a +5V logic- _______________________________________________________________________________________ 3 Evaluates: MAX5003 LVL position disconnects the R1-R2 resistor-divider from the MAX5003’s FB pin, as required for isolated operation). Turn the power supply back on and verify that the output voltage is still +5V. Note that for the isolated configuration, the output ground and the input ground may differ by as much as 500V. Do not operate switches SW1 and SW2 when power is applied to the EV kit because the controller can be damaged. 4 SHDN REF R11 100kΩ R23 1MΩ VIN GND (-48V) SW1-B 4 REF R12 20kΩ 1 Q2 2 3 ISO 6 NON 2 COMP R3 68kΩ C13 0.1µF 5 4 3 R5 39kΩ U1 C12 390pF FB 9 AGND CS PGND NDRV VCC VDD MAXTON MAX5003 COMP CON REF R7 200kΩ 8 7 6 SS FREQ ES V+ VINDIV 2 C10 0.47µF C11 0.1µF C9 100pF 1 R4 1MΩ 10 11 12 13 14 15 16 C1 33µF 100V R6 51kΩ 1 C8 10µF 16V R10 100Ω C17 0.1µF VDD 3 N1 2 VDD R22 1MΩ REF R9 0.11Ω 1/4W 1% D2 2 1 REF R18 OPEN C5 2200pF T1 R21 15Ω R13 1.3kΩ NOT INSTALLED R19 OPEN Q1 R8 43Ω 5 2 8 7 3 5 6 R14 240kΩ 11 12 9 10 1 U3 MOC217 7 2 2 1 2, 3 U2 TL431 1 C2 22µF 10V R15 43Ω VDD SW2 LVL DIR 3 D1 C16 4.7µF 25V 6, 7 8 C7 0.01µF R20 680Ω C6 3900pF JU1 CUT HERE R2 17.4kΩ 1% C3 22µF 10V R17 24.9kΩ 1% R16 24.9kΩ 1% R1 41.2kΩ 1% NON 1 3 5 GND ISO SW1-A +5V Evaluates: MAX5003 MAX5003 Evaluation Kit Figure 1. MAX5003 EV Kit Schematic _______________________________________________________________________________________ MAX5003 Evaluation Kit Figure 2. MAX5003 EV Kit Component Placement Guide— Component Side 1.0" 1.0" Figure 3. MAX5003 EV Kit PC Board Layout—Component Side Figure 4. MAX5003 EV Kit PC Board Layout—Solder Side Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 _____________________ 5 © 2001 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products. Evaluates: MAX5003 1.0"