LTC3566 High Efficiency USB Power Manager Plus 1A Buck-Boost Converter DESCRIPTION FEATURES The LTC®3566 is a highly integrated power management and battery charger IC for Li-Ion/Polymer battery applications. It includes a high efficiency current limited switching PowerPath manager with automatic load prioritization, a battery charger, an ideal diode, and a high efficiency synchronous buck-boost switching regulator. Designed specifically for USB applications, the LTC3566’s switching power manager automatically limits input current to a maximum of either 100mA or 500mA for USB applications or 1A for adapter-powered applications. POWER MANAGER ■ High Efficiency Switching PowerPathTM Controller with Bat-TrackTM Adaptive Output Control ■ Programmable USB or Wall Input Current Limit (100mA/500mA/1A) ■ Full Featured Li-Ion/Polymer Battery Charger ■ “Instant-On” Operation with Discharged Battery ■ 1.5A Maximum Charge Current ■ Internal 180mΩ Ideal Diode Plus External Ideal Diode Controller Powers Load in Battery Mode ■ Low No-Load I when Powered from BAT (<30μA) Q The LTC3566’s switching input stage transmits nearly all of the 2.5W available from the USB port to the system load with minimal power wasted as heat. This feature allows the LTC3566 to provide more power to the application and eases the constraint of thermal budgeting in small spaces. 1A BUCK-BOOST DC/DC ■ High Efficiency (1A I OUT) ■ 2.25MHz Constant Frequency Operation ■ Low No-Load Quiescent Current (~13μA) ■ Zero Shutdown Current ■ Pin Control of All Functions The synchronous buck-boost DC/DC can provide up to 1A. The LTC3566 is available in a low profile 24-lead 4mm × 4mm QFN surface mount package. APPLICATIONS ■ ■ L, LT, LTC and LTM are registered trademarks of Linear Technology Corporation. PowerPath and Bat-Track are trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents including 6522118, 6404251. HDD Based MP3 Players, PDA, GPS, PMP Products Other USB Based Handheld Products TYPICAL APPLICATION LTC3566 USB Power Manager with 3.3V/1A Buck-Boost FROM AC ADAPTER Switching Regulator Efficiency to System Load (POUT/PBUS) 3.3μH 4.7μF SW 10μF CLPROG GATE 3.01k 100k 0.1μF BAT NTC T 100k CHRG LTC3566 Li-Ion 1μF SWCD1 VOUT1 ILIM1 3.3V/25mA ALWAYS ON LDO 1μF 2.2μH ILIM0 DIGITAL CONTROL + LDO3V3 VIN1 SWAB1 CHRGEN 3.3V/1A HDD 324k MODE 10μF FB1 1.3nF EN1 GND EXPOSED PAD VC1 3566 TA01 90 80 OPTIONAL PROG 2k 100 VOUT = BAT + 300mV TO OTHER DC/DCs VOUT 105k EFFICIENCY (%) VBUS FROM USB 70 BAT = 4.2V 60 BAT = 3.3V 50 40 30 20 10 0 0.01 VBUS = 5V IBAT = 0mA 10x MODE 0.1 IOUT (A) 1 3566 TA01b 3566fa 1 LTC3566 ABSOLUTE MAXIMUM RATINGS PIN CONFIGURATION (Note 1) BAT VOUT VBUS SW CHRGEN EN1 TOP VIEW 24 23 22 21 20 19 LDO3V3 1 18 GATE CLPROG 2 17 GND NTC 3 16 CHRG 25 FB1 4 15 PROG VC1 5 14 ILIM1 GND 6 GND VIN1 SWCD1 9 10 11 12 VOUT1 8 MODE 13 ILIM0 7 SWAB1 VBUS (Transient) t < 1ms, Duty Cycle < 1% ...................................... –0.3V to 7V VBUS (Static), VIN1, BAT, NTC, CHRG, MODE, ILIM0, ILIM1, EN1, CHRGEN ................................ –0.3V to 6V FB1, VC1 .............. –0.3V to Lesser of 6V or (VIN1 + 0.3V) ICLPROG ....................................................................3mA ICHRG ......................................................................50mA IPROG ........................................................................2mA ILDO3V3 ...................................................................30mA ISW, IBAT, IVOUT ............................................................2A IVOUT1, ISWAB1, ISWCD1 .............................................2.5A Operating Temperature Range (Note 2).... –40°C to 85°C Junction Temperature (Note 3) ............................. 125°C Storage Temperature Range................... –65°C to 125°C UF PACKAGE 24-LEAD (4mm × 4mm) PLASTIC QFN TJMAX = 125°C, θJA = 37°C/W EXPOSED PAD (PIN 25) IS GND, MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING PACKAGE DESCRIPTION TEMPERATURE RANGE LTC3566EUF#PBF LTC3566EUF#TRPBF 3566 24-Lead (4mm × 4mm) Plastic QFN –40°C to 85°C Consult LTC Marketing for parts specified with wider operating temperature ranges. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VBUS = 5V, VBAT = 3.8V, DVCC = 3.3V, RCLPROG = 3.01k, RPROG = 1k, VIN1 = VOUT1 = 3.8V unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS Power Path Switching Regulator VBUS Input Supply Voltage 4.35 ● ● ● ● 87 436 800 0.31 5.5 95 460 860 0.38 100 500 1000 0.50 V IBUSLIM Total Input Current 1x Mode, VOUT = BAT 5x Mode, VOUT = BAT 10x Mode, VOUT = BAT Suspend Mode, VOUT = BAT mA mA mA mA IBUSQ VBUS Quiescent Current 1x Mode, IOUT = 0mA 5x Mode, IOUT = 0mA 10x Mode, IOUT = 0mA Suspend Mode, IOUT = 0mA 7 15 15 0.044 mA mA mA mA hCLPROG (Note 4) Ratio of Measured VBUS Current to CLPROG Program Current 1x Mode 5x Mode 10x Mode Suspend Mode 224 1133 2140 11.3 mA/mA mA/mA mA/mA mA/mA 3566fa 2 LTC3566 ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VBUS = 5V, VBAT = 3.8V, DVCC = 3.3V, RCLPROG = 3.01k, RPROG = 1k, VIN1 = VOUT1 = 3.8V unless otherwise noted. SYMBOL PARAMETER CONDITIONS IOUT (PowerPath) VOUT Current Available Before Loading BAT 1x Mode, BAT = 3.3V 5x Mode, BAT = 3.3V 10x Mode, BAT = 3.3V Suspend Mode 135 672 1251 0.32 mA mA mA mA VCLPROG CLPROG Servo Voltage in Current Limit 1x, 5x, 10x Modes Suspend Mode 1.188 100 V mV VUVLO_VBUS VBUS Undervoltage Lockout Rising Threshold Falling Threshold VUVLO_VBUS -VBAT VBUS to BAT Differential Undervoltage Lockout Rising Threshold Falling Threshold VOUT VOUT Voltage 1x, 5x, 10x Modes, 0V < BAT < 4.2V, IOUT = 0mA, Battery Charger Off MIN 3.95 ● Switching Frequency 4.30 4.00 MAX 4.35 200 50 USB Suspend Mode, IOUT = 250μA fOSC TYP UNITS V V mV mV 3.4 BAT + 0.3 4.7 V 4.5 4.6 4.7 V 2.25 2.7 MHz 1.8 RPMOS_PowerPath PMOS On-Resistance 0.18 Ω RNMOS_PowerPath NMOS On-Resistance 0.30 Ω 2 3 A A IPEAK_PowerPath Peak Switch Current Limit 1x, 5x Modes 10x Mode Battery Charger VFLOAT BAT Regulated Output Voltage ICHG Constant Current Mode Charge Current IBAT ● Battery Drain Current RPROG = 5k VBUS > VUVLO, Battery Charger Off, IOUT = 0μA VBUS = 0V, IOUT = 0μA (Ideal Diode Mode) 4.179 4.165 4.200 4.200 4.221 4.235 V V 980 185 1022 204 1065 223 mA mA 2 3.5 5 μA 27 38 μA VPROG PROG Pin Servo Voltage VPROG_TRIKL PROG Pin Servo Voltage in Trickle Charge VC/10 C/10 Threshold Voltage at PROG 100 mV hPROG Ratio of IBAT to PROG Pin Current 1022 mA/mA ITRKL Trickle Charge Current BAT < VTRKL 100 mA VTRIKL Trickle Charge Threshold Voltage BAT Rising ΔVTRKL Trickle Charge Hysteresis Voltage VRECHRG Recharge Battery Threshold Voltage Threshold Voltage Relative to VFLOAT –75 –100 –125 mV tTERM Safety Timer Termination Timer Starts When BAT = VFLOAT 3.3 4 5 Hour tBADBAT Bad Battery Termination Time BAT < VTRKL 0.42 0.5 0.63 Hour hC/10 End of Charge Indication Current Ratio (Note 5) 0.088 0.1 0.112 mA/mA VCHRG CHRG Pin Output Low Voltage ICHRG = 5mA 65 100 mV ICHRG CHRG Pin Leakage Current VCHRG = 5V 1 μA VBAT < VTRIKL 2.7 1.000 V 0.100 V 2.85 3.0 135 V mV 3566fa 3 LTC3566 ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VBUS = 5V, VBAT = 3.8V, DVCC = 3.3V, RCLPROG = 3.01k, RPROG = 1k, VIN1 = VOUT1 = 3.8V unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS RON_CHG Battery Charger Power FET On Resistance (Between VOUT and BAT) 0.18 Ω TLIM Junction Temperature in Constant Temperature Mode 110 °C NTC VCOLD Cold Temperature Fault Threshold Voltage Rising Threshold Hysteresis 75.0 76.5 1.5 78.0 %VBUS %VBUS VHOT Hot Temperature Fault Threshold Voltage Falling Threshold Hysteresis 33.4 34.9 1.5 36.4 %VBUS %VBUS VDIS NTC Disable Threshold Voltage Falling Threshold Hysteresis 0.7 1.7 50 2.7 %VBUS mV INTC NTC Leakage Current VNTC = VBUS = 5V –50 50 nA VFWD Forward Voltage VBUS = 0V, IOUT = 10mA IOUT = 10mA RDROPOUT Internal Diode On-Resistance, Dropout VBUS = 0V IMAX_DIODE Internal Diode Current Limit Ideal Diode 2 15 mV mV 0.18 Ω 1.6 A Always On 3.3V Supply VLDO3V3 Regulated Output Voltage 0mA < ILDO3V3 < 25mA 3.1 3.3 3.5 V RCL_LDO3V3 Closed-Loop Output Resistance 4 Ω ROL_LDO3V Dropout Output Resistance 23 Ω Logic (ILIM0, ILIM1, EN1, CHRGEN, MODE) VIL Logic Low Input Voltage VIH Logic High Input Voltage 0.4 IPD1 ILIM0, ILIM1, EN1, MODE Pull-Down Currents 1.6 IPD1_CHRGEN CHRGEN Pull-Down Current 1.6 1.2 V V μA 10 μA 5.5 V 2.6 2.8 2.9 V V 2.25 2.7 MHz 220 13 0 400 20 1 μA μA μA Buck-Boost Regulator VIN1 Input Supply Voltage 2.7 VOUTUVLO VOUT UVLO -VOUT Falling VOUT UVLO - VOUT Rising VIN1 Connected to VOUT Through Low Impedance. Switching Regulator Disabled in UVLO fOSC Oscillator Frequency PWM Mode IVIN1 Input Current PWM Mode, IOUT1 = 0μA Burst Mode® Operation, IOUT1 = 0μA Shutdown 2.5 ● 1.8 Burst Mode is a registered trademark of Linear Technology Corporation. 3566fa 4 LTC3566 ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VBUS = 5V, VBAT = 3.8V, DVCC = 3.3V, RCLPROG = 3.01k, RPROG = 1k, VIN1 = VOUT1 = 3.8V unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN VOUT1(LOW) Minimum Regulated Output Voltage For Burst Mode Operation or Synchronous PWM Operation VOUT1(HIGH) Maximum Regulated Output Voltage 5.50 TYP MAX UNITS 2.65 2.75 V 5.60 V ILIMF1 Forward Current Limit (Switch A) ● 2 2.5 3 IPEAK1(BURST) Forward Burst Current Limit (Switch Burst Mode Operation A) ● 200 275 350 mA IZERO1(BURST) Reverse Burst Current Limit (Switch D) ● –30 0 30 mA IMAX1(BURST) Maximum Deliverable Output Current 2.7V ≤ VIN1 ≤ 5.5V, 2.75V ≤ VOUT ≤ 5.5V in Burst Mode Operation (Note 6) VFB1 Feedback Servo Voltage IFB1 FB1 Input Current VFB1 = 0.8V RDS(ON)P PMOS RDS(ON) Switches A, D 0.22 Ω RDS(ON)N NMOS RDS(ON) Switches B, C 0.17 Ω ILEAK(P) PMOS Switch Leakage Switches A, D –1 1 μA ILEAK(N) NMOS Switch Leakage Switches B, C –1 1 μA PWM Mode Burst Mode Operation 50 ● RVOUT1 VOUT1 Pull-Down in Shutdown DBUCK(MAX) Maximum Buck Duty Cycle PWM Mode DBOOST(MAX) Maximum Boost Duty Cycle PWM Mode tSS1 Soft-Start Time A 0.780 mA 0.800 –50 0.820 V 50 nA 10 Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LTC3566E is guaranteed to meet performance specifications from 0°C to 85°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: The LTC3566 includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction ● kΩ 100 % 75 % 0.5 ms temperatures will exceed 125°C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. Note 4: Total input current is the sum of quiescent current, IVBUSQ, and measured current given by: VCLPROG/RCLPROG • (hCLPROG + 1) Note 5: hC/10 is expressed as a fraction of measured full charge current with indicated PROG resistor. Note 6: Guaranteed by design. 3566fa 5 LTC3566 TYPICAL PERFORMANCE CHARACTERISTICS Ideal Diode Resistance vs Battery Voltage Ideal Diode V-I Characteristics 1.0 0.20 INTERNAL IDEAL DIODE ONLY 0.4 0.2 INTERNAL IDEAL DIODE 0.15 0.10 INTERNAL IDEAL DIODE WITH SUPPLEMENTAL EXTERNAL VISHAY Si2333 PMOS 0.05 VBUS = 0V VBUS = 5V 0.12 0.16 0.08 FORWARD VOLTAGE (V) 0 2.7 0.20 3.0 3.6 3.9 3.3 BATTERY VOLTAGE (V) 3566 G01 600 125 VBUS = 5V RPROG = 1k RCLPROG = 3.01k CHARGE CURRENT (mA) 400 300 200 25 4.2 VBUS = 5V RPROG = 1k RCLPROG = 3.01k 75 50 90 1x USB SETTING, BATTERY CHARGER SET FOR 1A 3.0 100 EFFICIENCY (%) 70 60 RCLPROG = 3.01k RPROG = 1k IVOUT = 0mA BAT = 3.8V IVOUT = 0mA 5x CHARGING EFFICIENCY 1x CHARGING EFFICIENCY 80 60 2.7 3.0 4.2 50 70 3566 G07 3.6 3.9 3.3 BATTERY VOLTAGE (V) VBUS Quiescent Current vs VBUS Voltage (Suspend) 50 1 3.0 3566 G06 Battery Charging Efficiency vs Battery Voltage with No External Load (PBAT/PBUS) 80 VBUS = 5V (SUSPEND MODE) 3566 G05 5x, 10x MODE 0.1 OUTPUT CURRENT (A) 10 0 2.7 4.2 3.3 3.6 3.9 BATTERY VOLTAGE (V) 90 40 0.01 15 5 2.7 PowerPath Switching Regulator Efficiency vs Output Current 1x MODE IVOUT = 0μA VBUS = 0V 100 0 1000 20 3566 G04 BAT = 3.8V 600 800 400 OUTPUT CURRENT (mA) Battery Drain Current vs Battery Voltage 25 100 5x USB SETTING, BATTERY CHARGER SET FOR 1A 0 3.0 3.3 3.6 2.7 3.9 BATTERY VOLTAGE (V) 200 0 3566 G03 QUIESCENT CURRENT (μA) CHARGE CURRENT (mA) 150 EFFICIENCY (%) 3.25 4.2 USB Limited Battery Charge Current vs Battery Voltage 700 100 BAT = 3.4V 3.75 3566 G02 USB Limited Battery Charge Current vs Battery Voltage 500 4.00 3.50 BATTERY CURRENT (μA) 0.04 0 VBUS = 5V 5x MODE 4.25 OUTPUT VOLTAGE (V) RESISTANCE (Ω) CURRENT (A) 4.50 BAT = 4V 0.6 0 Output Voltage vs Output Current (Battery Charger Disabled) 0.25 INTERNAL IDEAL DIODE WITH SUPPLEMENTAL EXTERNAL VISHAY Si2333 PMOS 0.8 TA = 25°C unless otherwise noted. 3.6 3.9 3.3 BATTERY VOLTAGE (V) 4.2 3566 G08 40 30 20 10 0 0 1 3 2 VBUS VOLTAGE (V) 4 5 3566 G09 3566fa 6 LTC3566 TYPICAL PERFORMANCE CHARACTERISTICS Output Voltage vs Load Current in Suspend VBUS Current vs Load Current in Suspend 5.0 0.5 3.5 2.5 0.1 BAT = 3.9V, 4.2V 0.3 0.2 0.3 0.4 0.2 LOAD CURRENT (mA) 0 0.5 0.3 0.4 0.2 LOAD CURRENT (mA) 3566 G10 600 0.5 200 15 20 10 LOAD CURRENT (mA) 25 Low-Battery (Instant-On) Output Voltage vs Temperature 4.21 3.68 4.20 3.66 BAT = 2.7V IVOUT = 100mA 5x MODE OUTPUT VOLTAGE (V) FLOAT VOLTAGE (V) THERMAL REGULATION 5 0 3566 G12 500 CHARGE CURRENT (mA) BAT = 3V BAT = 3.1V BAT = 3.2V BAT = 3.3V 2.8 Battery Charger Float Voltage vs Temperature 300 BAT = 3.6V 3566 G11 Battery Charge Current vs Temperature 400 BAT = 3.5V 3.0 2.6 0.1 0 BAT = 3.4V 3.2 0.1 VBUS = 5V BAT = 3.3V RCLPROG = 3k 0 3.4 OUTPUT VOLTAGE (V) 4.0 3.3V LDO Output Voltage vs Load Current, VBUS = 0V VBUS = 5V BAT = 3.3V RCLPROG = 3.01k 0.4 VBUS CURRENT (mA) OUTPUT VOLTAGE (V) 4.5 3.0 TA = 25°C unless otherwise noted. 4.19 4.18 3.64 3.62 100 RPROG = 2k 10x MODE 0 –40 –20 0 20 40 60 80 TEMPERATURE (°C) 4.17 –40 100 120 –15 35 10 TEMPERATURE (°C) 60 3566 G13 2.2 BAT = 3V VBUS = 0V 2.0 60 70 VBUS = 5V IVOUT = 0μA 5x MODE 12 85 VBUS Quiescent Current in Suspend vs Temperature QUIESCENT CURRENT (μA) FREQUENCY (MHz) QUIESCENT CURRENT (mA) 15 BAT = 3.6V VBUS = 0V 35 10 TEMPERATURE (°C) 3566 G15 VBUS Quiescent Current vs Temperature 2.6 VBUS = 5V –15 3566 G14 Oscillator Frequency vs Temperature 2.4 3.60 –40 85 9 1x MODE 6 IVOUT = 0μA 60 50 40 BAT = 2.7V VBUS = 0V 1.8 –40 –15 35 10 TEMPERATURE (°C) 60 85 3566 G16 3 –40 –15 35 10 TEMPERATURE (°C) 60 85 3566 G17 30 –40 –15 35 10 TEMPERATURE (°C) 60 85 3566 G18 3566fa 7 LTC3566 TYPICAL PERFORMANCE CHARACTERISTICS CHRG Pin Current vs Voltage (Pull-Down State) 3.3V LDO Step Response (5mA to 15mA) 50 VBUS = 5V BAT = 3.8V 80 ILDO3V3 5mA/DIV 60 0mA 40 VLDO3V3 20mV/DIV AC COUPLED 20 0 Battery Drain Current vs Temperature 40 BATTERY CURRENT (μA) CHRG PIN CURRENT (mA) 100 TA = 25°C unless otherwise noted. BAT = 3.8V 0 1 3 4 2 CHRG PIN VOLTAGE (V) 30 20 10 3566 G2 20μs/DIV BAT = 3.8V VBUS = 0V BUCK REGULATORS OFF 0 –40 5 –15 35 10 TEMPERATURE (°C) 60 3566 G19 3566 G21 Buck-Boost Regulator Current Limit vs Temperature 0.30 0.40 2600 PMOS VIN1 = 3V PMOS VIN1 = 3.6V 0.25 PMOS VIN1 = 4.5V 0.35 2550 0.20 0.30 14.0 VIN1 = 3V VOUT1 = 3.3V 13.5 VIN1 = 4.5V 0.10 0.20 0.05 0.15 0.10 5 25 45 65 85 105 125 TEMPERATURE (°C) 13.0 VIN1 = 3V VIN1 = 4.5V IQ (μA) 0.25 2500 ILIMF (mA) NMOS VIN1 = 3V NMOS VIN1 = 3.6V NMOS VIN1 = 4.5V 0 –55 –35 –15 Buck-Boost Regulator Burst Mode Operation Quiescent Current VIN1 = 3.6V NMOS RDS(ON) (Ω) 2450 12.5 2400 12.0 2350 11.5 2300 –55 –35 –15 3566 G22 VIN1 = 3.6V 11.0 –55 –35 –15 5 25 45 65 85 105 125 TEMPERATURE (°C) 5 25 45 65 85 105 125 TEMPERATURE (°C) 3566 G23 Buck-Boost Regulator PWM Mode Efficiency 3566 G24 Buck-Boost Regulator PWM Efficiency vs VIN1 Buck-Boost Regulator vs ILOAD 100 100 90 90 Burst Mode OPERATION 90 CURVES 80 80 70 60 50 Burst Mode OPERATION CURVES VIN1 = 3V VIN1 = 3.6V VIN1 = 4.5V PWM MODE CURVES EFFICIENCY (%) EFFICIENCY (%) 80 VIN1 = 3V VIN1 = 3.6V VIN1 = 4.5V 40 30 20 VOUT1 = 3.3V TA = 27°C TYPE 3 COMPENSATION 10 0 0.1 1 10 ILOAD (mA) 100 1000 3566 G25 100 70 60 50 ILOAD = 50mA ILOAD = 200mA ILOAD = 1000mA 40 EFFICIENCY (%) PMOS RDS(ON) (Ω) RDS(ON) for Buck-Boost Regulator Power Switches vs Temperature 0.15 85 70 40 30 20 20 VIN1 = 3V VIN1 = 3.6V VIN1 = 4.5V VOUT1 = 5V TA = 27°C TYPE 3 COMPENSATION 10 4.3 4.7 3566 G26 VIN1 = 3V VIN1 = 3.6V VIN1 = 4.5V 50 30 VOUT1 = 3.3V 10 TA = 27°C TYPE 3 COMPENSATION 0 3.1 3.9 2.7 3.5 VIN1 (V) PWM MODE CURVES 60 0 0.1 1 10 ILOAD (mA) 100 1000 3566 G27 3566fa 8 LTC3566 TYPICAL PERFORMANCE CHARACTERISTICS Buck-Boost Regulator Load Regulation Reduction in Current Deliverability at Low VIN1 VIN1 = 3V VIN1 = 3.6V VIN1 = 4.5V 3.322 VOUT1 (V) 3.311 3.300 3.289 3.278 VOUT1 = 3.3V TA = 27°C TYPE 3 COMPENSATION 3.267 1 10 300 REDUCTION BELOW 1A (mA) 3.333 TA = 25°C unless otherwise noted. Buck-Boost Regulator Load Step, 0mA to 300mA STEADY STATE ILOAD START-UP WITH A RESISTIVE LOAD START-UP WITH A CURRENT SOURCE LOAD 250 200 150 CH2 ILOAD DC 200mA/DIV 100 VOUT1 = 3.3V TA = 27°C TYPE 3 COMPENSATION 50 0 100 1A CH1 VOUT1 AC 100mV/DIV 2.7 3.1 3.5 ILOAD (mA) 3566 G28 3.9 VIN1 (V) 4.3 4.7 VIN1 = 4.2V VOUT1 = 3.3V L = 2.2μH COUT = 47μF 100μs/DIV 3566 G30 3566 G29 PIN FUNCTIONS LDO3V3 (Pin 1): 3.3V LDO Output Pin. This pin provides a regulated, always-on, 3.3V supply voltage. LDO3V3 gets its power from VOUT. It may be used for light loads such as a watchdog microprocessor or real time clock. A 1μF capacitor is required from LDO3V3 to ground. If the LDO3V3 output is not used it should be disabled by connecting it to VOUT. CLPROG (Pin 2): USB Current Limit Program and Monitor Pin. A resistor from CLPROG to ground determines the upper limit of the current drawn from the VBUS pin. A fraction of the VBUS current is sent to the CLPROG pin when the synchronous switch of the PowerPath switching regulator is on. The switching regulator delivers power until the CLPROG pin reaches 1.188V. Several VBUS current limit settings are available via user input which will typically correspond to the 500mA and the 100mA USB specifications. A multilayer ceramic averaging capacitor or R-C network is required at CLPROG for filtering. NTC (Pin 3): Input to the Thermistor Monitoring Circuits. The NTC pin connects to a battery’s thermistor to determine if the battery is too hot or too cold to charge. If the battery’s temperature is out of range, charging is paused until it re-enters the valid range. A low drift bias resistor is required from VBUS to NTC and a thermistor is required from NTC to ground. If the NTC function is not desired, the NTC pin should be grounded. FB1 (Pin 4): Feedback Input for the (Buck-Boost) Switching Regulator. When the regulator’s control loop is complete, this pin servos to a fixed voltage of 0.8V. VC1 (Pin 5): Output of the Error Amplifier and Voltage Compensation Node for the (Buck-Boost) Switching Regulator. External Type I or Type III compensation (to FB1) connects to this pin. See Applications Information section for selecting buck-boost loop compensation components. GND (Pins 6, 12): Power GND pins for the buck-boost. SWAB1 (Pin 7): Switch Node for the (Buck-Boost) Switching Regulator. Connected to internal power switches A and B. External inductor connects between this node and SWCD1. MODE (Pin 8): Logic Input. Mode enables Burst Mode functionality for the buck-boost switching regulator when pin is set high. Has a 1.6μA internal pull-down current source. VIN1 (Pin 9): Power Input for the (Buck-Boost) Switching Regulator. This pin will generally be connected to VOUT (Pin 20). A 1μF(min) MLCC capacitor is recommended on this pin. VOUT1 (Pin 10): Regulated Output Voltage for the (BuckBoost) Switching Regulator. 3566fa 9 LTC3566 PIN FUNCTIONS SWCD1 (Pin 11): Switch Node for the (Buck-Boost) Switching Regulator. Connected to internal power switches C and D. External inductor connects between this node and SWAB1. BAT (Pin 19): Single-Cell Li-Ion Battery Pin. Depending on available VBUS power, a Li-Ion battery on BAT will either deliver power to VOUT through the ideal diode or be charged from VOUT via the battery charger. ILIM0 (Pin 13): Logic Input. Control pin for ILIM0 bit of the current limit of the PowerPath switching regulator. See Table 2. Active high. Has a 1.6μA internal pull-down current source. VOUT (Pin 20): Output Voltage of the Switching PowerPath Controller and Input Voltage of the Battery Charger. The majority of the portable product should be powered from VOUT. The LTC3566 will partition the available power between the external load on VOUT and the internal battery charger. Priority is given to the external load and any extra power is used to charge the battery. An ideal diode from BAT to VOUT ensures that VOUT is powered even if the load exceeds the allotted power from VBUS or if the VBUS power source is removed. VOUT should be bypassed with a low impedance ceramic capacitor. ILIM1 (Pin 14): Logic Input. Control pin for ILIM1 bit of the current limit of the PowerPath switching regulator. See Table 2. Active high. Has a 1.6μA internal pull-down current source. PROG (Pin 15): Charge Current Program and Charge Current Monitor Pin. Connecting a resistor from PROG to ground programs the charge current. If sufficient input power is available in constant-current mode, this pin servos to 1V. The voltage on this pin always represents the actual charge current. CHRG (Pin 16): Open-Drain Charge Status Output. The CHRG pin indicates the status of the battery charger. Four possible states are represented by CHRG: charging, not charging, unresponsive battery and battery temperature out of range. CHRG is modulated at 35kHz and switches between a low and high duty cycle for easy recognition by either humans or microprocessors. See Table 1. CHRG requires a pull-up resistor and/or LED to provide indication. GND (Pin 17): GND pin for USB Power Manager. GATE (Pin 18): Analog Output. This pin controls the gate of an optional external P-channel MOSFET transistor used to supplement the ideal diode between VOUT and BAT. The external ideal diode operates in parallel with the internal ideal diode. The source of the P-channel MOSFET should be connected to VOUT and the drain should be connected to BAT. If the external ideal diode FET is not used, GATE should be left floating. VBUS (Pin 21): Primary Input Power Pin. This pin delivers power to VOUT via the SW pin by drawing controlled current from a DC source such as a USB port or wall adapter. SW (Pin 22): Power Transmission Pin for the USB PowerPath. The SW pin delivers power from VBUS to VOUT via the step-down switching regulator. A 3.3μH inductor should be connected from SW to VOUT. CHRGEN (Pin 23): Logic Input. This logic input pin independently enables the battery charger. Active low. Has a 1.6μA internal pull-down current source. EN1 (Pin 24): Logic Input. This logic input pin independently enables the buck-boost switching regulator. Active high. Has a 1.6μA internal pull-down current source. Exposed Pad (Pin 25): Ground. Buck-boost logic and USB Power Manager ground connections. The Exposed Pad should be connected to a continuous ground plane on the printed circuit board directly under the LTC3566. 3566fa 10 LTC3566 BLOCK DIAGRAM 21 VBUS SW 2.25MHz PowerPath BUCK REGULATOR 22 LDO3V3 3.3V LDO VOUT SUSPEND LDO 500μA/2.5mA BATTERY TEMPERATURE MONITOR + + CHARGE STATUS 3.6V 18 – CC/CV CHARGER CHRG 1.2V 20 GATE IDEAL +– 0.3V + – 16 NTC + + 3 – CLPROG – 2 1 15mV BAT 19 PROG 15 CHRGEN VIN1 9 ENABLE SWAB1 MODE 7 ILIM DECODE LOGIC 23 24 13 14 8 VOUT1 CHRGEN 1A, 2.25MHz BUCK-BOOST REGULATOR EN1 10 SWCD1 11 ILIM0 ILIM1 FB1 MODE VC1 4 5 GND 6, 12, 17, 25 3566 BD 3566fa 11 LTC3566 OPERATION Introduction The LTC3566 is a highly integrated power management IC which includes a high efficiency switch mode PowerPath controller, a battery charger, an ideal diode, an always-on LDO, and a 1A buck-boost switching regulator. The entire chip is controlled via direct digital inputs. Designed specifically for USB applications, the PowerPath controller incorporates a precision average input current step-down switching regulator to make maximum use of the allowable USB power. Because power is conserved, the LTC3566 allows the load current on VOUT to exceed the current drawn by the USB port without exceeding the USB load specifications. The PowerPath switching regulator and battery charger communicate to ensure that the input current never violates the USB specifications. The ideal diode from BAT to VOUT guarantees that ample power is always available to VOUT even if there is insufficient or absent power at VBUS. An “always-on” LDO provides a regulated 3.3V from available power at VOUT. Drawing very little quiescent current, this LDO will be on at all times and can be used to supply up to 25mA. The LTC3566 also has a general purpose buck-boost switching regulator, which can be independently enabled via direct digital control. Along with constant frequency PWM mode, the buck-boost regulator has a low power burst-only mode setting for significantly reduced quiescent current under light load conditions. High Efficiency Switching PowerPath Controller Whenever VBUS is available and the PowerPath switching regulator is enabled, power is delivered from VBUS to VOUT via SW. VOUT drives both the external load (including the buck-boost regulator) and the battery charger. If the combined load does not exceed the PowerPath switching regulator’s programmed input current limit, VOUT will track 0.3V above the battery (Bat-Track). By keeping the voltage across the battery charger low, efficiency is optimized because power lost to the linear battery charger is minimized. Power available to the external load is therefore optimized. If the combined load at VOUT is large enough to cause the switching power supply to reach the programmed input current limit, the battery charger will reduce its charge current by the amount necessary to enable the external load to be satisfied. Even if the battery charge current is set to exceed the allowable USB current, the USB specification will not be violated. The switching regulator will limit the average input current so that the USB specification is never violated. Furthermore, load current at VOUT will always be prioritized and only remaining available power will be used to charge the battery. If the voltage at BAT is below 3.3V, or the battery is not present and the load requirement does not cause the switching regulator to exceed the USB specification, VOUT will regulate at 3.6V, thereby providing “Instant-On” operation. If the load exceeds the available power, VOUT will drop to a voltage between 3.6V and the battery voltage. If there is no battery present when the load exceeds the available USB power, VOUT can drop toward ground. The power delivered from VBUS to VOUT is controlled by a 2.25MHz constant-frequency step-down switching regulator. To meet the USB maximum load specification, the switching regulator includes a control loop which ensures that the average input current is below the level programmed at CLPROG. The current at CLPROG is a fraction (hCLPROG–1) of the VBUS current. When a programming resistor and an averaging capacitor are connected from CLPROG to GND, the voltage 3566fa 12 LTC3566 OPERATION The input current is programmed by the ILIM0 and ILIM1 pins. It can be configured to limit average input current to one of several possible settings as well as be deactivated (USB Suspend). The input current limit will be set by the VCLPROG servo voltage and the resistor on CLPROG according to the following expression: I VBUS =IBUSQ + 1800 1600 LTC3566 IDEAL DIODE 1400 1200 1000 800 600 ON SEMICONDUCTOR MBRM120LT3 400 200 0 0 Figure 2. Ideal Diode Operation 4.2 3.9 NO LOAD 3.6 300mV 3.3 3.0 2.7 2.7 3.0 3.6 3.3 BAT (V) 3.9 60 120 180 240 300 360 420 480 FORWARD VOLTAGE (mV) (BAT – VOUT) 3566 F02 4.5 VOUT (V) VISHAY Si2333 OPTIONAL EXTERNAL IDEAL DIODE 2000 VCLPROG •(hCLPROG + 1) RCLPROG Figure 1 shows the range of possible voltages at VOUT as a function of battery voltage. 2.4 2.4 2200 CURRENT (mA) on CLPROG represents the average input current of the switching regulator. When the input current approaches the programmed limit, CLPROG reaches VCLPROG, 1.188V and power out is held constant. 4.2 3566 F01 Figure 1. VOUT vs BAT Ideal Diode from BAT to VOUT The LTC3566 has an internal ideal diode as well as a controller for an optional external ideal diode. The ideal diode controller is always on and will respond quickly whenever VOUT drops below BAT. If the load current increases beyond the power allowed from the switching regulator, additional power will be pulled from the battery via the ideal diode. Furthermore, if power to VBUS (USB or wall power) is removed, then all of the application power will be provided by the battery via the ideal diode. The transition from input power to battery power at VOUT will be quick enough to allow only a 10μF capacitor to keep VOUT from drooping. The ideal diode consists of a precision amplifier that enables a large onchip P-channel MOSFET transistor whenever the voltage at VOUT is approximately 15mV (VFWD) below the voltage at BAT. The resistance of the internal ideal diode is approximately 180mΩ. If this is sufficient for the application, then no external components are necessary. However, if more conductance is needed, an external P-channel MOSFET transistor can be added from BAT to VOUT. When an external P-channel MOSFET transistor is present, the GATE pin of the LTC3566 drives its gate for automatic ideal diode control. The source of the external P-channel MOSFET should be connected to VOUT and the drain should be connected to BAT. Capable of driving a 1nF load, the GATE pin can control an external P-channel MOSFET transistor having an on-resistance of 40mΩ or lower. Suspend LDO If the LTC3566 is configured for USB suspend mode, the switching regulator is disabled and the suspend LDO provides power to the VOUT pin (presuming there is power available to VBUS). This LDO will prevent the battery from running down when the portable product has access to a suspended USB port. Regulating at 4.6V, this LDO only becomes active when the switching converter is disabled (suspended). To remain compliant with the USB specification, the input to the LDO is current limited so that it will not exceed the 500μA low power suspend specification. If the load on VOUT exceeds the suspend current limit, the additional current will come from the battery via the ideal diode. 3566fa 13 LTC3566 OPERATION TO USB OR WALL ADAPTER 21 VBUS SW ISWITCH/N VOUT PWM AND GATE DRIVE CONSTANT CURRENT CONSTANT VOLTAGE BATTERY CHARGER IDEAL DIODE OV 15mV CLPROG 1.188V – + AVERAGE INPUT CURRENT LIMIT CONTROLLER + + – 2 – + + – GATE SYSTEM LOAD 3.5V TO (BAT + 0.3V) 22 20 OPTIONAL EXTERNAL IDEAL DIODE PMOS 18 0.3V 3.6V BAT +– 19 AVERAGE OUTPUT VOLTAGE LIMIT CONTROLLER + SINGLE CELL Li-Ion 3566 F03 Figure 3. PowerPath Block Diagram 3.3V Always-On Supply The LTC3566 includes a low quiescent current low dropout regulator that is always powered. This LDO can be used to provide power to a system pushbutton controller, standby microcontroller or real time clock. Designed to deliver up to 25mA, the always-on LDO requires at least a 1μF low impedance ceramic bypass capacitor for compensation. The LDO is powered from VOUT, and therefore will enter dropout at loads less than 25mA as VOUT falls near 3.3V. If the LDO3V3 output is not used, it should be disabled by connecting it to VOUT. VBUS Undervoltage Lockout (UVLO) An internal undervoltage lockout circuit monitors VBUS and keeps the PowerPath switching regulator off until VBUS rises above 4.30V and is about 200mV above the battery voltage. Hysteresis on the UVLO turns off the regulator if VBUS drops below 4.00V or to within 50mV of BAT. When this happens, system power at VOUT will be drawn from the battery via the ideal diode. Battery Charger The LTC3566 includes a constant-current/constant-voltage battery charger with automatic recharge, automatic termination by safety timer, low voltage trickle charging, bad cell detection and thermistor sensor input for out-oftemperature charge pausing. Battery Preconditioning When a battery charge cycle begins, the battery charger first determines if the battery is deeply discharged. If the battery voltage is below VTRKL, typically 2.85V, an automatic trickle charge feature sets the battery charge current to 10% of the programmed value. If the low voltage persists for more than 1/2 hour, the battery charger automatically terminates and indicates via the CHRG pin that the battery was unresponsive. Once the battery voltage is above 2.85V, the battery charger begins charging in full power constant-current mode. The current delivered to the battery will try to reach 1022V/ RPROG. Depending on available input power and external load conditions, the battery charger may or may not be able to charge at the full programmed rate. The external load will always be prioritized over the battery charge current. The USB current limit programming will always be observed and only additional power will be available to charge the battery. When system loads are light, battery charge current will be maximized. 3566fa 14 LTC3566 OPERATION Charge Termination The battery charger has a built-in safety timer. When the voltage on the battery reaches the pre-programmed float voltage of 4.200V, the battery charger will regulate the battery voltage and the charge current will decrease naturally. Once the battery charger detects that the battery has reached 4.200V, the four hour safety timer is started. After the safety timer expires, charging of the battery will discontinue and no more current will be delivered. Automatic Recharge After the battery charger terminates, it will remain off drawing only microamperes of current from the battery. If the portable product remains in this state long enough, the battery will eventually self discharge. To ensure that the battery is always topped off, a charge cycle will automatically begin when the battery voltage falls below 4.1V. In the event that the safety timer is running when the battery voltage falls below 4.1V, it will reset back to zero. To prevent brief excursions below 4.1V from resetting the safety timer, the battery voltage must be below 4.1V for more than 1.3ms. The charge cycle and safety timer will also restart if the VBUS UVLO cycles low and then high (e.g. VBUS, is removed and then replaced) or if the battery charger is cycled on and off by the CHRGEN digital I/O pin. Charge Current The charge current is programmed using a single resistor from PROG to ground. 1/1022th of the battery charge current is sent to PROG which will attempt to servo to 1.000V. Thus, the battery charge current will try to reach 1022 times the current in the PROG pin. The program resistor and the charge current are calculated using the following equations: RPROG = 1022V 1022V ,ICHG = ICHG RPROG In either the constant-current or constant-voltage charging modes, the voltage at the PROG pin will be proportional to the actual charge current delivered to the battery. Therefore, the actual charge current can be determined at any time by monitoring the PROG pin voltage and using the following equation: IBAT = VPROG • 1022 RPROG In many cases, the actual battery charge current, IBAT, will be lower than ICHG due to limited input power available and prioritization with the system load drawn from VOUT. Charge Status Indication The CHRG pin indicates the status of the battery charger. Four possible states are represented by CHRG which include charging, not charging, unresponsive battery and battery temperature out of range. The signal at the CHRG pin can be easily recognized as one of the above four states by either a human or a microprocessor. An open drain output, the CHRG pin can drive an indicator LED through a current limiting resistor for human interfacing or simply a pull-up resistor for microprocessor interfacing. To make the CHRG pin easily recognized by both humans and microprocessors, the pin is either low for charging, high for not charging, or it is switched at high frequency (35kHz) to indicate the two possible faults, unresponsive battery and battery temperature out of range. When charging begins, CHRG is pulled low and remains low for the duration of a normal charge cycle. When charging is complete, i.e., the BAT pin reaches 4.200V and the charge current has dropped to one tenth of the programmed value, the CHRG pin is released (Hi-Z). If a fault occurs, the pin is switched at 35kHz. While switching, its duty cycle is modulated between a high and low value at a very low frequency. The low and high duty cycles are disparate 3566fa 15 LTC3566 OPERATION enough to make an LED appear to be on or off thus giving the appearance of “blinking”. Each of the two faults has its own unique “blink” rate for human recognition as well as two unique duty cycles for machine recognition. charge threshold voltage within the bad battery timeout period. In this case, the battery charger will falsely indicate a bad battery. System software may then reduce the load and reset the battery charger to try again. The CHRG pin does not respond to the C/10 threshold if the LTC3566 is in VBUS current limit. This prevents false end of charge indications due to insufficient power available to the battery charger. Although very improbable, it is possible that a duty cycle reading could be taken at the bright-dim transition (low duty cycle to high duty cycle). When this happens the duty cycle reading will be precisely 50%. If the duty cycle reading is 50%, system software should disqualify it and take a new duty cycle reading. Table 1 illustrates the four possible states of the CHRG pin when the battery charger is active. NTC Thermistor Table 1. CHRG Output Pin STATUS MODULATION (BLINK) FREQUENCY FREQUENCY DUTY CYCLE Charging 0Hz 0Hz (Lo-Z) 100% Not Charging 0Hz 0Hz (Hi-Z) 0% NTC Fault 35kHz 1.5Hz at 50% 6.25%, 93.75% Bad Battery 35kHz 6.1Hz at 50% 12.5%, 87.5% An NTC fault is represented by a 35kHz pulse train whose duty cycle alternates between 6.25% and 93.75% at a 1.5Hz rate. A human will easily recognize the 1.5Hz rate as a “slow” blinking which indicates the out-of-range battery temperature while a microprocessor will be able to decode either the 6.25% or 93.75% duty cycles as an NTC fault. If a battery is found to be unresponsive to charging (i.e., its voltage remains below 2.85V, for 1/2 hour), the CHRG pin gives the battery fault indication. For this fault, a human would easily recognize the frantic 6.1Hz “fast” blink of the LED while a microprocessor would be able to decode either the 12.5% or 87.5% duty cycles as a bad battery fault. Note that the LTC3566 is a 3-terminal PowerPath product where system load is always prioritized over battery charging. Due to excessive system load, there may not be sufficient power to charge the battery beyond the trickle The battery temperature is measured by placing a negative temperature coefficient (NTC) thermistor close to the battery pack. To use this feature connect the NTC thermistor, RNTC, between the NTC pin and ground and a resistor, RNOM, from VBUS to the NTC pin. RNOM should be a 1% resistor with a value equal to the value of the chosen NTC thermistor at 25°C (R25). A 100k thermistor is recommended since thermistor current is not measured by the LTC3566 and will have to be budgeted for USB compliance. The LTC3566 will pause charging when the resistance of the NTC thermistor drops to 0.54 times the value of R25 or approximately 54k. For Vishay “Curve 1” thermistor, this corresponds to approximately 40°C. If the battery charger is in constant-voltage (float) mode, the safety timer also pauses until the thermistor indicates a return to a valid temperature. As the temperature drops, the resistance of the NTC thermistor rises. The LTC3566 is also designed to pause charging when the value of the NTC thermistor increases to 3.25 times the value of R25. For Vishay “Curve 1” this resistance, 325k, corresponds to approximately 0°C. The hot and cold comparators each have approximately 3°C of hysteresis to prevent oscillation about the trip point. Grounding the NTC pin disables the NTC charge pausing function. 3566fa 16 LTC3566 OPERATION Thermal Regulation Input Current Limit To optimize charging time, an internal thermal feedback loop may automatically decrease the programmed charge current. This will occur if the die temperature rises to approximately 110°C. Thermal regulation protects the LTC3566 from excessive temperature due to high power operation or high ambient thermal conditions and allows the user to push the limits of the power handling capability with a given circuit board design without risk of damaging the LTC3566 or external components. The benefit of the LTC3566 thermal regulation loop is that charge current can be set according to actual conditions rather than worst-case conditions with the assurance that the battery charger will automatically reduce the current in worst-case conditions. The input current limit comparator will shut the input PMOS switch off once current exceeds 2.5A (typical). The 2.5A input current limit also protects against a grounded VOUT1 node. Buck-Boost DC/DC Switching Regulator The LTC3566 contains a 2.25MHz constant-frequency voltage mode buck-boost switching regulator. The regulator provides up to 1A of output load current. The buck-boost can be programmed to a minimum output voltage of 2.75V and can be used to power a microcontroller core, microcontroller I/O, memory, disk drive, or other logic circuitry. To suit a variety of applications, a selectable mode function allows the user to trade off noise for efficiency. Two modes are available to control the operation of the LTC3566’s buck-boost regulator. At moderate to heavy loads, the constant frequency PWM mode provides the least noise switching solution. At lighter loads Burst Mode operation may be selected. The output voltage is programmed by a user supplied resistive divider returned to the FB1 pin. An error amplifier compares the divided output voltage with a reference and adjusts the compensation voltage accordingly until the FB1 has stabilized at 0.8V. The buckboost regulator also includes a soft-start to limit inrush current and voltage overshoot when powering on, short circuit current protection, and switch node slew limiting circuitry for reduced radiated EMI. Output Overvoltage Protection If the FB1 node were inadvertently shorted to ground, then the output would increase indefinitely with the maximum current that could be sourced from VIN1. The LTC3566 protects against this by shutting off the input PMOS if the output voltage exceeds a 5.6V (typical). Low Output Voltage Operation When the output voltage is below 2.65V (typical) during start-up, Burst Mode operation is disabled and switch D is turned off (allowing forward current through the well diode and limiting reverse current to 0mA). Buck-Boost Regulator PWM Operating Mode In PWM mode the voltage seen at FB1 is compared to a 0.8V reference. From the FB1 voltage an error amplifier generates an error signal seen at VC1. This error signal commands PWM waveforms that modulate switches A, B, C and D. Switches A and B operate synchronously as do switches C and D. If VIN1 is significantly greater than the programmed VOUT1, then the converter will operate in buck mode. In this mode switches A and B will be modulated, with switch D always on (and switch C always off), to step-down the input voltage to the programmed output. If VIN1 is significantly less than the programmed VOUT1, then the converter will operate in boost mode. In this mode switches C and D are modulated, with switch A always on (and switch B always off), to step-up the input voltage to the programmed output. If VIN1 is close to the programmed VOUT1, then the converter will operate in 4-switch mode. In this mode the switches sequence through the pattern of AD, AC, BD to either step the input voltage up or down to the programmed output. 3566fa 17 LTC3566 OPERATION Buck-Boost Regulator Burst Mode Operation Buck-Boost Regulator Soft-Start Operation In Burst Mode operation, the buck-boost regulator uses a hysteretic FB1 voltage algorithm to control the output voltage. By limiting FET switching and using a hysteretic control loop, switching losses are greatly reduced. In this mode output current is limited to 50mA typical. While operating in Burst Mode operation, the output capacitor is charged to a voltage slightly higher than the regulation point. The buck-boost converter then goes into a sleep state, during which the output capacitor provides the load current. The output capacitor is charged by charging the inductor until the input current reaches 275mA typical and then discharging the inductor until the reverse current reaches 0mA typical. This process is repeated until the feedback voltage has charged to 6mV above the regulation point. In the sleep state, most of the regulator’s circuitry is powered down, helping to conserve battery power. When the feedback voltage drops 6mV below the regulation point, the switching regulator circuitry is powered on and another burst cycle begins. The duration for which the regulator sleeps depends on the load current and output capacitor value. The sleep time decreases as the load current increases. The maximum load current in Burst Mode operation is 50mA. The buck-boost regulator will not go to sleep if the current is greater than 50mA and if the load current increases beyond this point while in Burst Mode operation the output will lose regulation. Burst Mode operation provides a significant improvement in efficiency at light loads at the expense of higher output ripple when compared to PWM mode. For many noise-sensitive systems, Burst Mode operation might be undesirable at certain times (i.e. during a transmit or receive cycle of a wireless device), but highly desirable at others (i.e. when the device is in low power standby mode). The MODE pin is used to enable or disable Burst Mode operation at any time, offering both low noise and low power operation when they are needed. Soft-start is accomplished by gradually increasing the reference voltage input to the error amplifier over a 0.5ms (typical) period. This limits transient inrush currents during start-up because the output voltage is always “in regulation”. Ramping the reference voltage input also limits the rate of increase in the VC1 voltage which helps minimize output overshoot during start-up. A soft-start cycle occurs whenever the buck-boost is enabled, or after a fault condition has occurred (thermal shutdown or UVLO). A soft-start cycle is not triggered by changing operating modes. This allows seamless operation when transitioning between Burst Mode operation and PWM mode. Low Supply Operation The LTC3566 incorporates an undervoltage lockout circuit on VOUT (connected to VIN1) which shuts down the buckboost regulator when VOUT drops below 2.6V. This UVLO prevents unstable operation. Table 2. USB Current Limit Settings ILIM1 ILIM0 USB SETTING 0 0 1x Mode (USB 100mA Limit) 0 1 10x Mode (Wall 1A Limit) 1 0 Suspend 1 1 5x Mode (USB 500mA Limit) Table 3. Switching Regulator Modes MODE SWITCHING REGULATOR MODE 0 PWM Mode 1 Burst Mode Operation 3566fa 18 LTC3566 APPLICATIONS INFORMATION CLPROG Resistor and Capacitor Choosing the PowerPath Inductor As described in the High Efficiency Switching PowerPath Controller section, the resistor on the CLPROG pin determines the average input current limit when the switching regulator is set to either the 1x mode (USB 100mA), the 5x mode (USB 500mA) or the 10x mode. The input current will be comprised of two components, the current that is used to drive VOUT and the quiescent current of the switching regulator. To ensure that the USB specification is strictly met, both components of input current should be considered. The Electrical Characteristics table gives values for quiescent currents in either setting as well as current limit programming accuracy. To get as close to the 500mA or 100mA specifications as possible, a 1% resistor should be used. Recall that IVBUS = IVBUSQ + VCLPROG/RCLPROG • (hCLPROG + 1). Because the input voltage range and output voltage range of the PowerPath switching regulator are both fairly narrow, the LTC3566 was designed for a specific inductance value of 3.3μH. Some inductors which may be suitable for this application are listed in Table 4. An averaging capacitor or an R-C combination is required in parallel with the CLPROG resistor so that the switching regulator can determine the average input current. This network also provides the dominant pole for the feedback loop when current limit is reached. To ensure stability, the capacitor on CLPROG should be 0.1μF or larger. Table 4. Recommended Inductors for PowerPath Controller INDUCTOR TYPE L (μH) MAX IDC (A) MAX DCR (Ω) SIZE IN mm (L × W × H) MANUFACTURER LPS4018 3.3 2.2 0.08 3.9 × 3.9 × 1.7 CoilCraft www.coilcraft. com D53LC DB318C 3.3 3.3 2.26 1.55 0.034 0.070 5.0 × 5.0 × 3.0 Toko 3.8 × 3.8 × 1.8 www.toko.com WE-TPC Type M1 3.3 1.95 0.065 4.8 × 4.8 × 1.8 Würth Elektronik www.we-online. com CDRH6D12 CDRH6D38 3.3 3.3 2.2 3.5 0.0625 6.7 × 6.7 × 1.5 Sumida 0.020 7.0 × 7.0 × 4.0 www.sumida.com 3566fa 19 LTC3566 APPLICATIONS INFORMATION Buck-Boost Regulator Inductor Selection Many different sizes and shapes of inductors are available from numerous manufacturers. Choosing the right inductor from such a large selection of devices can be overwhelming, but following a few basic guidelines will make the selection process much simpler. The buck-boost converter is designed to work with inductors in the range of 1μH to 5μH. For most applications a 2.2μH inductor will suffice. Larger value inductors reduce ripple current which improves output ripple voltage. Lower value inductors result in higher ripple current and improved transient response time. To maximize efficiency, choose an inductor with a low DC resistance. For a 3.3V output, efficiency is reduced about 3% for a 100mΩ series resistance at 1A load current, and about 2% for 300mΩ series resistance at 200mA load current. Choose an inductor with a DC current rating at least 2 times larger than the maximum load current to ensure that the inductor does not saturate during normal operation. If output short circuit is a possible condition, the inductor should be rated to handle the 2.5A maximum peak current specified for the buck-boost converter. Different core materials and shapes will change the size/current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or Permalloy materials are small and do not radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. Inductors that are very thin or have a very small volume typically have much higher core and DCR losses, and will not give the best efficiency. The choice of which style inductor to use often depends more on the price vs size, performance and any radiated EMI requirements than on what the LTC3566 requires to operate. The inductor value also has an effect on Burst Mode operation. Lower inductor values will cause the Burst Mode operation switching frequencies to increase. Table 5 shows several inductors that work well with the LTC3566’s buck-boost regulator. These inductors offer a good compromise in current rating, DCR and physical size. Consult each manufacturer for detailed information on their entire selection of inductors. Table 5. Recommended Inductors for Buck-Boost Regulator L (μH) MAX IDC (A) MAX DCR (Ω) SIZE IN mm (L × W × H) MANUFACTURER LPS4018 3.3 2.2 2.2 2.5 0.08 0.07 3.9 × 3.9 × 1.7 3.9 × 3.9 × 1.7 Coilcraft www.coilcraft.com D53LC 2.0 3.25 0.02 5.0 × 5.0 × 3.0 Toko www.toko.com 7440430022 2.2 2.5 0.028 4.8 × 4.8 × 2.8 Würth Elektronik www.we-online.com CDRH4D22/HP 2.2 2.4 0.044 4.7 × 4.7 × 2.4 Sumida www.sumida.com SD14 2.0 2.56 0.045 5.2 × 5.2 × 1.45 Cooper www.cooperet.com INDUCTOR TYPE 3566fa 20 LTC3566 APPLICATIONS INFORMATION VBUS and VOUT Bypass Capacitors The style and value of capacitors used with the LTC3566 determine several important parameters such as regulator control-loop stability and input voltage ripple. Because the LTC3566 uses a step-down switching power supply from VBUS to VOUT, its input current waveform contains high frequency components. It is strongly recommended that a low equivalent series resistance (ESR) multilayer ceramic capacitor be used to bypass VBUS. Tantalum and aluminum capacitors are not recommended because of their high ESR. The value of the capacitor on VBUS directly controls the amount of input voltage ripple for a given load current. Increasing the size of this capacitor will reduce the input voltage ripple. To prevent large VOUT voltage steps during transient load conditions, it is also recommended that a ceramic capacitor be used to bypass VOUT. The output capacitor is used in the compensation of the switching regulator. At least 4μF of actual capacitance with low ESR are required on VOUT. Additional capacitance will improve load transient performance and stability. Multilayer ceramic chip capacitors typically have exceptional ESR performance. MLCCs combined with a tight board layout and an unbroken ground plane will yield very good performance and low EMI emissions. There are several types of ceramic capacitors available, each having considerably different characteristics. For example, X7R ceramic capacitors have the best voltage and temperature stability. X5R ceramic capacitors have apparently higher packing density but poorer performance over their rated voltage and temperature ranges. Y5V ceramic capacitors have the highest packing density, but must be used with caution, because of their extreme nonlinear characteristic of capacitance vs voltage. The actual in-circuit capacitance of a ceramic capacitor should be measured with a small AC signal (ideally less than 200mV) as is expected in-circuit. Many vendors specify the capacitance vs voltage with a 1VRMS AC test signal and as a result overstate the capacitance that the capacitor will present in the application. Using similar operating conditions as the application, the user must measure or request from the vendor the actual capacitance to determine if the selected capacitor meets the minimum capacitance that the application requires. Buck-Boost Regulator Input/Output Capacitor Selection Low ESR MLCC capacitors should be used at both the buck-boost regulator output (VOUT1) and the buck-boost regulator input supply (VIN1). Only X5R or X7R ceramic capacitors should be used because they retain their capacitance over wider voltage and temperature ranges than other ceramic types. A 22μF output capacitor is sufficient for most applications. The buck-boost regulator input supply should be bypassed with a 2.2μF capacitor. Consult with capacitor manufacturers for detailed information on their selection and specifications of ceramic capacitors. Many manufacturers now offer very thin (<1mm tall) ceramic capacitors ideal for use in height restricted designs. Table 6 shows a list of several ceramic capacitor manufacturers. Table 6. Recommended Ceramic Capacitor Manufacturers MANUFACTURER WEBSITE AVX www.avxcorp.com Murata www.murata.com Taiyo Yuden www.t-yuden.com Vishay Siliconix www.vishay.com TDK www.tdk.com Over-Programming the Battery Charger The USB high power specification allows for up to 2.5W to be drawn from the USB port (5V x 500mA). The PowerPath switching regulator transforms the voltage at VBUS to just above the voltage at BAT with high efficiency, while limiting power to less than the amount programmed at CLPROG. In some cases the battery charger may be programmed (with the PROG pin) to deliver the maximum safe charging current without regard to the USB specifications. If there is insufficient current available to charge the battery at the programmed rate, the PowerPath regulator will reduce charge current until the system load on VOUT is satisfied 3566fa 21 LTC3566 APPLICATIONS INFORMATION and the VBUS current limit is satisfied. Programming the battery charger for more current than is available will not cause the average input current limit to be violated. It will merely allow the battery charger to make use of all available power to charge the battery as quickly as possible, and with minimal power dissipation within the battery charger. Alternate NTC Thermistors and Biasing The LTC3566 provides temperature qualified charging if a grounded thermistor and a bias resistor are connected to NTC. By using a bias resistor whose value is equal to the room temperature resistance of the thermistor (R25) the upper and lower temperatures are pre-programmed to approximately 40°C and 0°C, respectively (assuming a Vishay “Curve 1” thermistor). The upper and lower temperature thresholds can be adjusted by either a modification of the bias resistor value or by adding a second adjustment resistor to the circuit. If only the bias resistor is adjusted, then either the upper or the lower threshold can be modified but not both. The other trip point will be determined by the characteristics of the thermistor. Using the bias resistor in addition to an adjustment resistor, both the upper and the lower temperature trip points can be independently programmed with the constraint that the difference between the upper and lower temperature thresholds cannot decrease. Examples of each technique follow. NTC thermistors have temperature characteristics which are indicated on resistance-temperature conversion tables. The Vishay-Dale thermistor NTHS0603N011-N1003F, used in the following examples, has a nominal value of 100k and follows the Vishay “Curve 1” resistance-temperature characteristic. In the explanation below, the following notation is used. R25 = Value of the thermistor at 25°C RNTC|COLD = Value of thermistor at the cold trip point RNTC|HOT = Value of thermistor at the hot trip point rCOLD = Ratio of RNTC|COLD to R25 RNOM = Primary thermistor bias resistor (see Figure 4a) R1 = Optional temperature range adjustment resistor (see Figure 4b) The trip points for the LTC3566’s temperature qualification are internally programmed at 0.349 • VBUS for the hot threshold and 0.765 • VBUS for the cold threshold. Therefore, the hot trip point is set when: RNTC|HOT RNOM +RNTC|HOT • VBUS = 0.349 • VBUS and the cold trip point is set when: RNTC|COLD RNOM +RNTC|COLD • VBUS = 0.765 • VBUS Solving these equations for RNTC|COLD and RNTC|HOT results in the following: RNTC|HOT = 0.536 • RNOM and RNTC|COLD = 3.25 • RNOM By setting RNOM equal to R25, the above equations result in rHOT = 0.536 and rCOLD = 3.25. Referencing these ratios to the Vishay Resistance-Temperature Curve 1 chart gives a hot trip point of about 40°C and a cold trip point of about 0°C. The difference between the hot and cold trip points is approximately 40°C. By using a bias resistor, RNOM, different in value from R25, the hot and cold trip points can be moved in either direction. The temperature span will change somewhat due to the nonlinear behavior of the thermistor. The following equations can be used to easily calculate a new value for the bias resistor: rHOT •R25 0.536 r RNOM = COLD •R25 3.25 RNOM = rHOT= Ratio of RNTC|HOT to R25 3566fa 22 LTC3566 APPLICATIONS INFORMATION where rHOT and rCOLD are the resistance ratios at the desired hot and cold trip points. Note that these equations are linked. Therefore, only one of the two trip points can be chosen, the other is determined by the default ratios designed in the IC. Consider an example where a 60°C hot trip point is desired. From the Vishay Curve 1 R-T characteristics, rHOT is 0.2488 at 60°C. Using the above equation, RNOM should be set to 46.4k. With this value of RNOM, the cold trip point is about 16°C. Notice that the span is now 44°C rather than the previous 40°C. This is due to the decrease in “temperature gain” of the thermistor as absolute temperature increases. The upper and lower temperature trip points can be independently programmed by using an additional bias resistor as shown in Figure 4b. The following formulas can be used to compute the values of RNOM and R1: RNOM = rCOLD −rHOT •R25 2.714 R1 = 0.536 • RNOM – rHOT • R25 For example, to set the trip points to 0°C and 45°C with a Vishay Curve 1 thermistor choose: LTC3566 NTC BLOCK VBUS VBUS RNOM 100k NTC 0.765 • VBUS RNTC 100k 3.266 − 0.4368 • 100k = 104.2k 2.714 The nearest 1% value is 105k R1 = 0.536 • 105k – 0.4368 • 100k = 12.6k The nearest 1% value is 12.7k. The final solution is shown in Figure 4b and results in an upper trip point of 45°C and a lower trip point of 0°C. USB Inrush Limiting When a USB cable is plugged into a portable product, the inductance of the cable and the high-Q ceramic input capacitor form an L-C resonant circuit. If the cable does not have adequate mutual coupling or if there is not much impedance in the cable, it is possible for the voltage at the input of the product to reach as high as twice the USB voltage (~10V) before it settles out. To prevent excessive voltage from damaging the LTC3566 during a hot insertion, it is best to have a low voltage coefficient capacitor at the VBUS pin to the LTC3566. This is achievable by selecting an MLCC capacitor that has a higher voltage rating than that required for the application. For example, a 16V, X5R, 10μF capacitor in a 1206 case would be a more conservative choice than a 6.3V, X5R, 10μF capacitor in a smaller 0805 VBUS – TOO_COLD 3 RNOM = VBUS RNOM 105k NTC 0.765 • VBUS LTC3566 NTC BLOCK – TOO_COLD + 3 + – R1 12.7k – TOO_HOT 0.349 • VBUS TOO_HOT 0.349 • VBUS + RNTC 100k + + + NTC_ENABLE 0.017 • VBUS – NTC_ENABLE 0.017 • VBUS – 3566 F04a 3566 F04b (b) (a) Figure 4. NTC Circuits 3566fa 23 LTC3566 APPLICATIONS INFORMATION case. The size of the input overshoot will be determined by the “Q” of the resonant tank circuit formed by CIN and the input lead inductance. It is recommended to measure the input ringing with the selected components to verify compliance with the Absolute Maximum specifications. Alternatively, the following soft connect circuit (Figure 5) can be employed. In this circuit, capacitor C1 holds MP1 off when the cable is first connected. Eventually C1 begins to charge up to the USB input voltage applying increasing gate support to MP1. The long time constant of R1 and C1 prevent the current from building up in the cable too fast thus dampening out any resonant overshoot. Where COUT is the output filter capacitor. The output filter zero is given by: f FILTER _ ZERO = ⎛ R1 ⎞ VOUT1 = VFB1 ⎜ +1 ⎝ RFB ⎟⎠ Closing the Feedback Loop The LTC3566 incorporates voltage mode PWM control. The control to output gain varies with operation region (buck, boost, buck-boost), but is usually no greater than 20. The output filter exhibits a double pole response given by: 1 Hz 2 • π • L • COUT A troublesome feature in boost mode is the right-half plane zero (RHP), and is given by: f RHPZ = VBUS 5V USB INPUT USB CABLE C2 10μF LTC3566 R1 40k GND 3566 F05 Figure 5. USB Soft Connect Circuit VIN12 Hz 2 • π •IOUT •L • VOUT1 The loop gain is typically rolled off before the RHP zero frequency. A simple Type I compensation network (as shown in Figure 6), can be incorporated to stabilize the loop but at the cost of reduced bandwidth and slower transient response. To ensure proper phase margin, the loop must cross unity-gain a decade before the LC double pole. f UG = 1 Hz 2 • π • R1• CP1 Most applications demand an improved transient response to allow a smaller output filter capacitor. To achieve a higher bandwidth, Type III compensation is required. Two zeros are required to compensate for the double-pole response. Type III compensation also reduces any VOUT1 overshoot seen at start-up. The compensation network depicted in Figure 7 yields the transfer function: MP1 Si2333 C1 100nF Hz The unity-gain frequency of the error amplifier with the Type I compensation is given by: where VFB1 is fixed at 0.8V (see Figure 6). f FILTER _ POLE = 2 • π • RESR • COUT where RESR is the capacitor equivalent series resistance. Buck-Boost Regulator Output Voltage Programming The buck-boost regulator can be programmed for output voltages greater than 2.75V and less than 5.5V. The output voltage is programmed using a resistor divider from the VOUT1 pin connected to the FB1 pin such that: 1 VC1 1 = • VOUT1 R1• (C1+ C2) (1+ sR2C2) • (1+ s(R1+R3)C3) ⎛ sR2C1C2 ⎞ • (1+ sR3C3) s • ⎜ 1+ ⎝ C1+ C2 ⎟⎠ 3566fa 24 LTC3566 APPLICATIONS INFORMATION A Type III compensation network attempts to introduce a phase bump at a higher frequency than the LC double pole. This allows the system to cross unity gain after the LC double pole, and achieve a higher bandwidth. While attempting to cross over after the LC double pole, the system must still cross over before the boost right-half plane zero. If unity gain is not reached sufficiently before the right-half plane zero, then the –180° of phase lag from the LC double pole combined with the –90° of phase lag from the right-half plane zero will result in negating the phase bump of the compensator. The compensator zeros should be placed either before or only slightly after the LC double pole such that their positive phase contributions offset the –180° that occurs at the filter double pole. If they are placed at too low of a frequency, they will introduce too much gain to the system and the crossover frequency will be too high. The two high frequency poles should be placed such that the system crosses unity gain during the phase bump introduced by the zeros and before the boost right-half plane zero and such that the compensator bandwidth is less than the bandwidth of the error amp (typically 900 kHz). If the gain of the compensation network is ever greater than the gain of the error amplifier, then the error amplifier no longer acts as an ideal op-amp, another pole will be introduced and at the same point. Recommended Type III compensation components for a 3.3V output: R1: 324kΩ RFB: 105kΩ C1: 10pF R2: 15kΩ C2: 330pF R3: 121kΩ C3: 33pF COUT: 22μF LOUT: 2.2μH Printed Circuit Board Layout Considerations In order to be able to deliver maximum current under all conditions, it is critical that the Exposed Pad on the backside of the LTC3566 package be soldered to the PC board ground. Failure to make thermal contact between the Exposed Pad on the backside of the package and the copper board will result in higher thermal resistances. Furthermore, due to its high frequency switching circuitry, it is imperative that the input capacitors, inductors, and output capacitors be as close to the LTC3566 as possible VOUT1 + VOUT1 + ERROR AMP ERROR AMP 0.8V R1 FB1 CP1 RFB R1 FB1 R3 C3 – VC1 – VC1 0.8V R2 C1 C2 RFB 3566 F07 3566 F06 Figure 6. Error Amplifier with Type I Compensation Figure 7. Error Amplifier with Type III Compensation 3566fa 25 LTC3566 APPLICATIONS INFORMATION 1. Are the capacitors at VBUS, VIN1, and VOUT1 as close as possible to the LTC3566? These capacitors provide the AC current to the internal power MOSFETs and their drivers. Minimizing inductance from these capacitors to the LTC3566 is a top priority. 2. Are COUT and L1 closely connected? The (-) plate of COUT returns current to the GND plane, and then back to CIN. 3566 F08 3. Keep sensitive components away from the SW pins. Battery Charger Stability Considerations Figure 8. Higher Frequency Ground Currents Follow Their Incident Path. Slices in the Ground Plane Cause High Voltage and Increased Emissions. and that there be an unbroken ground plane under the LTC3566 and all of its external high frequency components. High frequency currents, such as the VBUS, VIN1, and VOUT1 currents on the LTC3566, tend to find their way along the ground plane in a myriad of paths ranging from directly back to a mirror path beneath the incident path on the top of the board. If there are slits or cuts in the ground plane due to other traces on that layer, the current will be forced to go around the slits. If high frequency currents are not allowed to flow back through their natural least-area path, excessive voltage will build up and radiated emissions will occur. There should be a group of vias under the grounded backside of the package leading directly down to an internal ground plane. To minimize parasitic inductance, the ground plane should be on the second layer of the PC board. The GATE pin for the external ideal diode controller has extremely limited drive current. Care must be taken to minimize leakage to adjacent PC board traces. 100nA of leakage from this pin will introduce an offset to the 15mV ideal diode of approximately 10mV. To minimize leakage, the trace can be guarded on the PC board by surrounding it with VOUT connected metal, which should generally be less than one volt higher than GATE. When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3566. The LTC3566’s battery charger contains both a constantvoltage and a constant-current control loop. The constantvoltage loop is stable without any compensation when a battery is connected with low impedance leads. Excessive lead length, however, may add enough series inductance to require a bypass capacitor of at least 1μF from BAT to GND. Furthermore, when the battery is disconnected, a 4.7μF capacitor in series with a 0.2Ω to 1Ω resistor from BAT to GND is required to keep ripple voltage low. High value, low ESR multilayer ceramic chip capacitors reduce the constant-voltage loop phase margin, possibly resulting in instability. Ceramic capacitors up to 22μF may be used in parallel with a battery, but larger ceramics should be decoupled with 0.2Ω to 1Ω of series resistance. In constant-current mode, the PROG pin is in the feedback loop rather than the battery voltage. Because of the additional pole created by any PROG pin capacitance, capacitance on this pin must be kept to a minimum. With no additional capacitance on the PROG pin, the battery charger is stable with program resistor values as high as 25k. However, additional capacitance on this node reduces the maximum allowed program resistor. The pole frequency at the PROG pin should be kept above 100kHz. Therefore, if the PROG pin has a parasitic capacitance, CPROG, the following equation should be used to calculate the maximum resistance value for RPROG: RPROG ≤ 1 2π • 100kHz • CPROG 3566fa 26 LTC3566 TYPICAL APPLICATIONS Direct Pin Controlled LTC3566 USB Power Manager with 3.3V/1A Buck-Boost L1 3.3μH USB 4.5V TO 5.5V VBUS C1 10μF 100k C2 22μF VOUT LTC3566 NTC GATE OPTIONAL BAT 100k T TO OTHER LOADS SW + PROG 1k Li-Ion GND CLPROG 2k 0.1μF CHRG 3.01k VIN1 2.2μF SWAB1 PARTS LIST C1: MURATA GRM21BR61A/06KE19 C2,C3: TAIYO-YUDEN JMK212BJ226MG L1: COILCRAFT LPS4018-332MLC L2: COILCRAFT LPS4018-222MLC L2 2.2μH LDO3V3 SWCD1 1μF 121k VOUT1 3.3V/1A DISK DRIVE 33pF C3 22μF 324k TO DIGITAL CONTROLLER CHRGEN FB1 MODE VC1 EN1 GND ILIM 330pF 15k 10pF 105k 2 3566 TA02 PACKAGE DESCRIPTION UF Package 24-Lead Plastic QFN (4mm × 4mm) (Reference LTC DWG # 05-08-1697 Rev B) BOTTOM VIEW—EXPOSED PAD 4.00 p 0.10 (4 SIDES) 0.70 p0.05 R = 0.115 TYP 0.75 p 0.05 PIN 1 TOP MARK (NOTE 6) PIN 1 NOTCH R = 0.20 TYP OR 0.35 s 45o CHAMFER 23 24 0.40 p 0.10 1 2 4.50 p 0.05 2.45 p 0.05 3.10 p 0.05 (4 SIDES) 2.45 p 0.10 (4-SIDES) PACKAGE OUTLINE (UF24) QFN 0105 0.25 p0.05 0.50 BSC 0.200 REF 0.00 – 0.05 0.25 p 0.05 0.50 BSC RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS NOTE: 1. DRAWING PROPOSED TO BE MADE A JEDEC PACKAGE OUTLINE MO-220 VARIATION (WGGD-X)—TO BE APPROVED 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE, IF PRESENT 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 3566fa Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 27 LTC3566 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC3440 600mA (IOUT), 2MHz Synchronous BuckBoost DC/DC Converter VIN: 2.5V to 5.5V, VOUT: 2.5V to 5.5V IQ = 25μA, ISD < 1μA, MS, DFN Package LTC3441/ LTC3442 1.2A (IOUT), Synchronous Buck-Boost DC/DC Converters, LTC3441 (1MHz), LTC3443 (600kHz) VIN: 2.5V to 5.5V, VOUT: 2.4V to 5.25V IQ = 25μA, ISD < 1μA, MS, DFN Package LTC3442 1.2A (IOUT), 2MHz Synchronous Buck-Boost DC/DC Converter VIN: 2.4V to 5.5V, VOUT: 2.4V to 5.25V IQ = 28μA, ISD < 1μA, MS Package LTC3455 Dual DC/DC Converter with USB Power Management and Li-Ion Battery Charger Efficiency >96%, Accurate USB Current Limiting (500mA/100mA), 4mm × 4mm QFN-24 Package LTC3538 800mA, 2MHz Synchronous Buck-Boost DC/DC Converter VIN: 2.4V to 5.5V, VOUT: 1.8V to 5.25V IQ = 35μA, 2mm × 3mm DFN-8 Package LTC3550 Dual Input USB/AC Adapter Li-Ion Battery Charger with adjustable output 600mA Buck Converter Synchronous Buck Converter, Efficiency: 93%, Adjustable Output at 600mA; Charge Current: 950mA Programmable, USB Compatible, Automatic Input Power Detection and Selection, 3mm × 5mm DFN-16 Package LTC3550-1 Dual Input USB/AC Adapter Li-Ion Battery Charger with 600mA Buck Converter Synchronous Buck Converter, Efficiency: 93%, Output: 1.875V at 600mA; Charge Current: 950mA Programmable, USB Compatible, Automatic Input Power Detection and Selection, 3mm × 5mm DFN-16 Package LTC3552 Standalone Linear Li-Ion Battery Charger with Adjustable Output Dual Synchronous Buck Converter Synchronous Buck Converter, Efficiency: >90%, Adjustable Outputs at 800mA and 400mA; Charge Current Programmable Up to 950mA, USB Compatible, 3mm × 5mm DFN-16 Package LTC3552-1 Standalone Linear Li-Ion Battery Charger with Dual Synchronous Buck Converter Synchronous Buck Converter, Efficiency: >90%, Output: 1.8V at 800mA, 1.575V at 400mA; Charge Current Programmable Up to 950mA, USB Compatible, 3mm × 5mm DFN-16 Package LTC3555 Switching USB Power Manager with Li-Ion/ Polymer Charger, Triple Synchronous Buck Converter Plus LDO Complete Multi-Function PMIC: Switchmode Power Manager and Three Buck Regulators Plus LDO; Charge Current Programmable Up to 1.5A from Wall Adapter Input, Thermal Regulation, Synchronous Buck Converters Efficiency: >95%, ADJ Outputs: 0.8V to 3.6V at 400mA/400mA/1A Bat-Track Adaptive Output Control, 200mΩ Ideal Diode, 4mm × 5mm QFN-28 Package LTC3556 Switching USB Power Manager with Li-Ion/ Polymer Charger, 1A Buck-Boost + Dual Sync Buck Converter + LDO Complete Multi-Function PMIC: Switching Power Manager, 1A Buck-Boost + 2 Buck Regulators + LDO, ADJ Out Down to 0.8V at 400mA/400mA/1A, Synchronous Buck/ Buck-Boost Converter Efficiency: >95%; Charge Current Programmable up to 1.5A from Wall Adapter Input, Thermal Regulation, Bat-Track Adaptive Output Control, 180mΩ Ideal Diode, 4mm × 5mm QFN-28 Package LTC3557/ LTC3557-1 USB Power Manager with Li-Ion/Polymer Charger, Triple Synchronous Buck Converter Plus LDO Complete Multi-Function PMIC: Linear Power Manager and Three Buck Regulators, Charge Current Programmable Up to 1.5A from Wall Adapter Input, Thermal Regulation, Synchronous Buck Converters Efficiency: >95%, ADJ Output: 0.8V to 3.6V at 400mA/ 400mA/600mA, Bat-Track Adaptive Output Control, 200mΩ Ideal Diode, 4mm × 4mm QFN-28 Package LTC3559 Linear USB Li-Ion/Polymer Battery Charger with Dual Synchronous Buck Converters Adjustable Synchronous Buck Converters, Efficiency: >90%, Outputs: Down to 0.8V at 400mA for Each, Charge Current Programmable Up to 950mA, USB Compatible, 3mm × 3mm QFN-16 Package LTC4055 USB Power Controller and Battery Charger Charges Single-Cell Li-Ion Batteries Directly From USB Port, Thermal Regulation, 4mm × 4mm QFN-16 Package LTC4067 Linear USB Power Manager with OVP, Ideal Diode Controller and Li-Ion Charger 13V Overvoltage Transient Protection, Thermal Regulation 200mΩ Ideal Diode with <50mΩ Option, 3mm × 4mm QFN-14 Package LTC4085 Linear USB Power Manager with Ideal Diode Controller and Li-Ion Charger Charges Single Cell Li-Ion Batteries Directly from a USB Port, Thermal Regulation, 200mΩ Ideal Diode with <50mΩ Option, 3mm × 4mm QFN-14 Package LTC4088/ LTC4088-1/ LTC4088-2 High Efficiency USB Power Manager and Battery Charger Maximizes Available Power from USB Port, Bat-Track, “Instant-On” Operation, 1.5A Maximum Charge Current, 180mΩ Ideal Diode with <50mΩ Option, 3.3V/25mA AlwaysOn LDO, 3mm × 4mm DFN-14 Package LTC4090 High Voltage USB Power Manager with Ideal Diode Controller and High Efficiency Li-Ion Battery Charger High Efficiency 1.2A Charger from 6V to 38V (60V Maximum) Input Charges Single Cell Li-Ion Batteries Directly from a USB Port, Thermal Regulation; 200mΩ Ideal Diode with <50mΩ option, 3mm × 6mm DFN-22 Package Bat-Track Adaptive Output Control 3566fa 28 Linear Technology Corporation LT 0508 REV A • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2008