19-2883; Rev 0; 7/03 KIT ATION EVALU E L B A AVAIL 5-Channel Slim DSC Power Supplies Features ♦ Step-Up DC-DC Converter, 95% Efficient The MAX1584/MAX1585 include 5 high-efficiency DCDC conversion channels: ♦ Up to 1MHz Operating Frequency • Step-up DC-DC converter with on-chip FETs • Step-down DC-DC converter with on-chip FETs • Three PWM DC-DC controllers for CCD, LCD, LED, or other functions The step-down DC-DC converter can operate directly from the battery or from the step-up output, providing boost-buck capability with a compound efficiency of up to 90%. Both devices include three PWM DC-DC controllers: the MAX1584 includes two step-up controllers and one step-down controller, while the MAX1585 includes one step-up controller, one inverting controller, and one step-down controller. All DC-DC channels operate at one fixed frequency—settable from 100kHz to 1MHz—to optimize size, cost, and efficiency. Other features include soft-start, power-OK outputs, and overload protection. The MAX1584/MAX1585 are available in space-saving, 32-pin thin QFN packages. An evaluation kit is available to expedite designs. ♦ Step-Down DC-DC Converter Operate from Battery for 95% Efficient Step-Down 90% Efficient Boost-Buck with Step-Up ♦ Three Auxiliary PWM DC-DC Controllers ♦ No Transformers (MAX1585) ♦ 1mA Shutdown Mode ♦ Internal Soft-Start Control ♦ Overload Protection ♦ Compact 32-Pin Thin QFN Package (5mm x 5mm) Ordering Information PINPACKAGE AUX FUNCTIONS MAX1584ETJ -40°C to +85°C 32 Thin QFN 5mm x 5mm 2 step-up 1 step-down MAX1585ETJ -40°C to +85°C 32 Thin QFN 5mm x 5mm 1 step-up 1 step-down 1 inverting PART TEMP RANGE Pin Configuration Applications Typical Operating Circuit MAX1585 GND DL1 DL3 DL2 PV INDL2 28 27 26 25 CC1 1 24 CC2 FB1 2 23 FB2 PGSD 3 22 PVSU 21 LXSU 20 PGSU MAX1584 MAX1585 LXSD 4 PVSD 5 ONSD 6 19 OSC CCSD 7 18 SCF FBSD 8 17 SDOK CCD -7.5V AUX3 LOGIC +3.3V 10 11 12 13 14 15 16 AUX1OK AUX2 9 FBSU LCD, CCD, LED +15V CCSU CORE +1.8V REF AUX1 29 ON1 ONSU ONSD ON1 ON2 ON3 30 ON3 STEP-DOWN SYSTEM +5V 31 ONSU STEP-UP 32 ON2 INPUT 0.7V TO 5.5V CC3 PDAs FB3 Digital Cameras THIN QFN 5mm x 5mm ________________________________________________________________ Maxim Integrated Products For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at 1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com. 1 MAX1584/MAX1585 General Description The MAX1584/MAX1585 provide a complete powersupply solution for slim digital cameras. They improve performance, component count, and size compared to conventional multichannel controllers in 2-cell AA, 1-cell Li+, and dual-battery designs. On-chip MOSFETs provide up to 95% efficiency for critical power supplies, while additional channels operate with external FETs for optimum design flexibility. This optimizes overall efficiency and cost, while also reducing board space. MAX1584/MAX1585 5-Channel Slim DSC Power Supplies ABSOLUTE MAXIMUM RATINGS PV, PVSU, PVSD, SDOK, AUX1OK, SCF, ON_, FB_ to GND..........................................................................-0.3V to +6V PGND to GND....................................................…-0.3V to +0.3V INDL2, DL1, DL3 to GND.........................-0.3V to (PVSU + 0.3V) DL2 to GND ............................................-0.3V to (INDL2 + 0.3V) PV to PVSU ...........................................................-0.3V to + 0.3V LXSU Current (Note 1) ..........................................................3.6A LXSD Current (Note 1) ........................................................2.25A REF, OSC, CC_ to GND...........................-0.3V to (PVSU + 0.3V) Continuous Power Dissipation (TA = +70°C) 32-Pin Thin QFN (derate 22mW/°C above +70°C) ....1700mW Operating Temperature Range ...........................-40°C to +85°C Junction Temperature ......................................................+150°C Storage Temperature Range .............................-65°C to +150°C Lead Temperature (soldering, 10s) .................................+300°C Note 1: LXSU has internal clamp diodes to PVSU and PGND, and LXSD has internal clamp diodes to PVSD and PGND. Applications that forward bias these diodes should take care not to exceed the device’s power dissipation limits. Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS (VPVSU = VPV = VPVSD = VINDL2 = 3.6V, TA = 0°C to +85°C, unless otherwise noted.) PARAMETER CONDITIONS MIN TYP MAX UNITS 5.5 V GENERAL Input Voltage Range (Note 2) 0.7 Step-Up Minimum Startup Voltage ILOAD < 1mA, TA = +25°C, startup voltage tempco is -2300ppm/°C (typ) (Note 3) 0.9 1.1 V Shutdown Supply Current into PV PV = 3.6V 0.1 5 µA Supply Current into PV with Step-Up Enabled ONSU = 3.6V, FBSU = 1.5V (does not include switching losses) 300 450 µA Supply Current into PV with Step-Up and Step-Down Enabled ONSU = ONSD = 3.6V, FBSU = 1.5V, FBSD = 1.5V (does not include switching losses) 450 700 µA Total Supply Current from PV and PVSU with Step-Up and One AUX Enabled ONSU = ON1 = 3.6V, FBSU = 1.5V, FB2 = 1.5V (does not include switching losses) 400 650 µA REFERENCE Reference Output Voltage IREF = 20µA 1.25 1.27 V Reference Load Regulation 10µA < IREF < 200µA 1.23 4.5 10 mV Reference Line Regulation 2.7 < PVSU < 5.5V 1.3 5 mV 1.25 1.275 V 52 80 OSCILLATOR OSC Discharge Trip Level Rising edge OSC Discharge Resistance OSC = 1.5V, IOSC = 3mA 1.225 OSC Discharge Pulse Width OSC Frequency ROSC = 47kΩ, COSC =100pF Ω 150 ns 500 kHz STEP-UP DC-DC CONVERTER Step-Up Startup-to-Normal Operating Threshold Step-Up Startup-to-Normal Operating Threshold Hysteresis 2 Rising edge or falling edge (Note 4) 2.30 2.5 80 _______________________________________________________________________________________ 2.65 V mV 5-Channel Slim DSC Power Supplies (VPVSU = VPV = VPVSD = VINDL2 = 3.6V, TA = 0°C to +85°C, unless otherwise noted.) PARAMETER CONDITIONS Step-Up Voltage Adjust Range MIN TYP 3.0 Start Delay of ONSD, ON1, ON2, ON3 after SU in Regulation MAX 5.5 V OSC cycles 1024 FBSU Regulation Voltage UNITS 1.231 1.25 1.269 V FBSU = CCSU 80 135 185 µS FBSU Input Leakage Current FBSU = 1.25V -100 +1 +100 Idle ModeTM Trip Level (Note 6) FBSU to CCSU Transconductance Current-Sense Amplifier Transresistance mA 0.275 V/A Step-Up Maximum Duty Cycle FBSU = 1V 85 90 % PVSU Leakage Current VLX = 0V, PVSU = 5.5V 0.1 5 µA LXSU Leakage Current VLXSU = VOUT = 5.5V 0.1 5 µA N channel 95 150 P channel 150 250 2.8 3.2 Switch On-Resistance N-Channel Current Limit 80 nA 150 2.4 P-Channel Turn-Off Current mΩ A 20 mA Startup Current Limit PVSU = 1.8V (Note 5) 450 mA Startup tOFF PVSU = 1.8V 700 ns Startup Frequency PVSU = 1.8V 200 kHz STEP-DOWN DC-DC CONVERTER Step-Down Output Voltage Adjust Range PVSD must be greater than output (Note 7) FBSD Regulation Voltage 1.25 5.00 V 1.231 1.25 1.269 V FBSD to CCSD Transconductance FBSD = CCSD 80 135 185 µS FBSD Input Leakage Current FBSD = 1.25V -100 +0.1 +100 nA Idle Mode Trip Level (Note 6) Current-Sense Amplifier Transresistance LXSD Leakage Current Switch On-Resistance 100 mA 0.5 V/A VLXSD = 0 to 3.6V, PVSU = 3.6V 0.1 5 N channel 95 150 P channel 150 250 0.8 0.95 P-Channel Current Limit 0.65 N-Channel Turn-Off Current Soft-Start Interval µA mΩ A 20 mA 2048 OSC cycles SDOK Output Low Voltage 0.1mA into SDOK 0.01 0.1 V SDOK Leakage Current ONSU = GND 0.01 1 µA Idle Mode is a trademark of Maxim Integrated Products, Inc. _______________________________________________________________________________________ 3 MAX1584/MAX1585 ELECTRICAL CHARACTERISTICS (continued) MAX1584/MAX1585 5-Channel Slim DSC Power Supplies ELECTRICAL CHARACTERISTICS (continued) (VPVSU = VPV = VPVSD = VINDL2 = 3.6V, TA = 0°C to +85°C, unless otherwise noted.) PARAMETER CONDITIONS MIN TYP MAX UNITS AUX1, 2, 3 DC-DC CONTROLLERS Maximum Duty Cycle FB_ = 1V 80 85 90 % FB1 and FB3 Regulation Voltage FB_ = CC_ 1.231 1.25 1.269 V FB2 (MAX1584) Regulation Voltage FB_ = CC_ 1.231 1.25 1.269 V FB2 (MAX1585) (Inverter) Regulation Voltage FB_ = CC_ -0.01 0 +0.01 V FB_ to CC_ Transconductance FB_ = CC_ 80 135 185 µS FB_ Input Leakage Current FB_ = 1.25V -100 +1 +100 nA DL_ Driver Resistance Output high or low 2.5 10 Ω DL_ Drive Current Sourcing or sinking 0.5 A 4096 OSC cycles Soft-Start Interval AUX1OK Output Low Voltage 0.1mA into AUX1OK 0.01 0.1 V AUX1OK Leakage Current ONSU = GND 0.01 1 µA OVERLOAD AND THERMAL PROTECTION Overload-Protection Fault Delay OSC cycles 100,000 SCF Leakage Current ONSU = PVSU, FBSU = 1.5V SCF Output Low Voltage 0.1mA into SCF 0.1 1 0.01 0.1 µA V Thermal Shutdown +160 °C Thermal Hysteresis 20 °C LOGIC INPUTS ON_ Input Low Level ON_ Input High Level 1.1V < PVSU < 1.8V (ONSU only) 0.2 1.8V < PVSU< 5.5V 0.4 1.1V < PVSU < 1.8V (ONSU only) 1.8V < PVSU < 5.5V ON_ Impedance to GND 4 ON_ = 3.35V VPVSU 0.2 V V 1.6 330 _______________________________________________________________________________________ kΩ 5-Channel Slim DSC Power Supplies (VPVSU = VPV = VPVSD = VINDL2 = 3.6V, TA = -40°C to +85°C, unless otherwise noted.) (Note 8) PARAMETER CONDITIONS MIN MAX UNITS 0.7 5.5 V 5 µA GENERAL Input Voltage Range (Note 2) Shutdown Supply Current into PVSU PVSU = 3.6V Supply Current into PV with Step-Up Enabled ONSU = 3.6V, FBSU = 1.5V (does not include switching losses) 450 µA Supply Current into PV with Step-Up and Step-Down Enabled ONSU = ONSD = 3.6V, FBSU = 1.5V, FBSD = 1.5V (does not include switching losses) 700 µA Total Supply Current from PV and PVSU with Step-Up and One AUX Enabled ONSU = ON1 = 3.6V, FBSU = 1.5V, FB2 = 1.5V (does not include switching losses) 650 µA REFERENCE Reference Output Voltage IREF = 20µA 1.275 V Reference Load Regulation 10µA < IREF < 200µA 1.225 10 mV Reference Line Regulation 2.7V < PVSU < 5.5V 5 mV OSCILLATOR OSC Discharge Trip Level Rising edge OSC Discharge Resistance OSC = 1.5V, IOSC = 3mA 1.225 1.275 V 80 Ω 2.30 2.65 V 3.0 5.5 V 1.225 1.275 V STEP-UP DC-DC CONVERTER Step-Up Startup-to-Normal Operating Threshold Rising edge or falling edge (Note 4) Step-Up Voltage Adjust Range FBSU Regulation Voltage FBSU to CCSU Transconductance FBSU = CCSU 80 185 µS FBSU Input Leakage Current FBSU = 1.25V -100 +100 nA Step-Up Maximum Duty Cycle FBSU = 1V 80 90 % PVSU Leakage Current VLX = 0V, PVSU = 5.5V 5 µA LXSU Leakage Current VLXSU = VOUT = 5.5V 5 µA Switch On-Resistance N channel 150 P channel 250 N-Channel Current Limit mΩ 2.4 3.2 A 1.25 5.00 V STEP-DOWN DC-DC CONVERTER Step-Down Output Voltage Adjust Range PVSD must be greater than output (Note 7) _______________________________________________________________________________________ 5 MAX1584/MAX1585 ELECTRICAL CHARACTERISTICS MAX1584/MAX1585 5-Channel Slim DSC Power Supplies ELECTRICAL CHARACTERISTICS (continued) (VPVSU = VPV = VPVSD = VINDL2 = 3.6V, TA = -40°C to +85°C, unless otherwise noted.) (Note 8) PARAMETER CONDITIONS FBSD Regulation Voltage MIN MAX UNITS 1.225 1.275 V FBSD to CCSD Transconductance FBSD = CCSD 80 185 µS FBSD Input Leakage Current FBSD = 1.25V -100 +100 nA LXSD Leakage Current VLXSD = 0 to 3.6V, PVSU = 3.6V 5 µA Switch On-Resistance N channel 150 P channel 250 P-Channel Current Limit 0.65 SDOK Output Low Voltage 0.1mA into SDOK SDOK Leakage Current ONSU = GND 0.95 mΩ A 0.1 V 1 µA AUX1, 2, 3 DC-DC CONTROLLERS Maximum Duty Cycle FB_ = 1V 80 90 % FB1 and FB3 Regulation Voltage FB_ = CC_ 1.225 1.275 V FB2 (MAX1584) Regulation Voltage FB_ = CC_ 1.225 1.275 V FB2 (MAX1585) (Inverter) Regulation Voltage FB_ = CC_ -0.01 +0.01 V FB_ to CC_ Transconductance FB_ = CC_ 80 185 µS FB_ Input Leakage Current FB_ = 1.25V -100 +100 nA DL_ Driver Resistance Output high or low 10 Ω AUX1OK Output Low Voltage 0.1mA into AUX1OK 0.1 V AUX1OK Leakage Current ONSU = GND 1 µA OVERLOAD AND THERMAL PROTECTION SCF Leakage Current ONSU = PVSU, FBSU = 1.5V 1 µA SCF Output Low Voltage 0.1mA into SCF 0.1 V 1.1V < PVSU < 1.8V (ONSU only) 0.2 1.8V < PVSU < 5.5V 0.4 LOGIC INPUTS ON_ Input Low Level ON_ Input High Level 1.1V < PVSU < 1.8V (ONSU only) 1.8V < PVSU < 5.5V VPVSU - 0.2 1.6 V V Note 2: The MAX1584/MAX1585 are powered from the step-up output (PVSU). An internal low-voltage startup oscillator drives the step-up starting at about 0.9V until PVSU reaches approximately 2.5V. When PVSU reaches 2.5V, the main control circuitry takes over. Once the step-up is up and running, it can maintain operation with very low input voltages; however, output current is limited. Note 3: Since the device is powered from PVSU, a Schottky rectifier, connected from the input battery to PVSU, is required for lowvoltage startup, or if PVSD is connected to VIN instead of PVSU. Note 4: The step-up regulator is in startup mode until this voltage is reached. Do not apply full load current during startup. A powerOK output can be used with an external PFET to gate the load until the step-up is in regulation. See the Applications Information section. 6 _______________________________________________________________________________________ 5-Channel Slim DSC Power Supplies (VPVSU = VPV = VPVSD = VINDL2 = 3.6V, TA = -40°C to +85°C, unless otherwise noted.) (Note 8) Note 5: The step-up current limit in startup refers to the LXSU switch current limit, not an output current limit. Note 6: The idle mode current threshold is the transition point between fixed-frequency PWM operation and idle mode operation (where switching rate varies with load). The specification is given in terms of inductor current. In terms of output current, the idle mode transition varies with input-output voltage ratio and inductor value. For the step-up, the transition output current is approximately 1/3 the inductor current when stepping from 2V to 3.3V. For the step-down, the transition current in terms of output current is approximately 3/4 the inductor current when stepping down from 3.3V to 1.8V. Note 7: Operation in dropout (100% duty cycle) can only be maintained for 100,000 OSC cycles before the output is considered faulted, triggering global shutdown. Note 8: Specifications to -40°C are guaranteed by design, not production tested. Typical Operating Characteristics (Circuit of Figure 1, TA = +25°C, unless otherwise noted.) STEP-DOWN EFFICIENCY vs. LOAD CURRENT VIN = 4.5V VIN = 4.2V VIN = 3.8V VIN = 3.0V 40 60 40 30 30 20 20 10 10 VOUT = 5V 0 1 10 100 VIN = 3.0V VIN = 3.8V VIN = 4.2V VIN = 4.5V 50 80 PVSD CONNECTED TO BATTERY VOUT = 1.5V DOES NOT INCLUDE CURRENT USED BY THE STEP-UP TO POWER THE IC VOUT3 = 3.3V VOUTSU = 5.0V 20 10 1 10 100 1000 1 10 100 LOAD CURRENT (mA) EFFICIENCY vs. INPUT VOLTAGE AUX1 EFFICIENCY vs. LOAD CURRENT MAX1585 AUX2 EFFICIENCY vs. LOAD CURRENT MAX1584/85 toc04 100 90 80 85 SU = 5V, 300mA SD = 1.5V, 250mA SU + AUX3 = 3.3V, 300mA AUX1 = 15V, 40mA AUX2 = -7.5V, 40mA EFFICIENCY (%) EFFICIENCY (%) 90 VIN = 4.5V VIN = 4.2V VIN = 3.8V VIN = 3.0V 70 60 50 VIN = 3.0V VIN = 3.8V VIN = 4.2V VIN = 4.5V 60 50 VOUT1 = 15V 30 3.5 70 40 40 70 4.0 INPUT VOLTAGE (V) 4.5 1000 90 80 3.0 40 LOAD CURRENT (mA) 95 75 50 LOAD CURRENT (mA) 100 80 VIN = 4.5V VIN = 4.2V VIN = 3.8V VIN = 3.0V 60 30 0 1000 70 MAX5184/85 toc06 50 70 90 MAX5184/85 toc05 60 80 EFFICIENCY (%) 70 90 EFFICIENCY (%) EFFICIENCY (%) 80 100 MAX1584/85 toc02 90 EFFICIENCY (%) 100 MAX1584/85 toc01 100 COMBINED BOOST-BUCK EFFICIENCY vs. LOAD CURRENT MAX1584/85 toc03 STEP-UP EFFICIENCY vs. LOAD CURRENT VOUT2 = -7.5V 30 1 10 100 LOAD CURRENT (mA) 1000 1 10 100 1000 LOAD CURRENT (mA) _______________________________________________________________________________________ 7 MAX1584/MAX1585 ELECTRICAL CHARACTERISTICS (continued) Typical Operating Characteristics (continued) (Circuit of Figure 1, TA = +25°C, unless otherwise noted.) MINIMUM STARTUP VOLTAGE vs. LOAD CURRENT (VSU) NO-LOAD INPUT CURRENT vs. INPUT VOLTAGE (SWITCHING) BOOST-BUCK (SU + AUX3) VSU = 5.0V, OUT3 = 3.33V 6 5 4 3 2 3.0 2.5 1.5 1.0 0.5 VSU = 5.0V 1 0 0 0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 200 400 600 800 1000 INPUT VOLTAGE (V) LOAD CURRENT (mA) REFERENCE VOLTAGE vs. TEMPERATURE REFERENCE VOLTAGE vs. REFERENCE LOAD CURRENT MAX1584/85 toc09 1.254 1.251 1.248 1.246 1.250 1.249 REFERENCE VOLTAGE (V) 0 REFERENCE VOLTAGE (V) SCHOTTKY DIODE CONNECTED FROM IN TO VSU 2.0 MAX1584/85 toc10 INPUT CURRENT (mA) 7 MAX5184/85 toc08 8 3.5 MINIMUM STARTUP VOLTAGE (V) MAX1584/85 toc07 9 1.248 1.247 1.246 1.245 1.243 1.244 -25 0 25 50 75 100 0 50 100 150 200 REFERENCE LOAD CURRENT (µA) OSCILLATOR FREQUENCY vs. ROSC SWITCHING FREQUENCY vs. TEMPERATURE COSC = 470pF 900 COSC = 330pF COSC = 220pF 700 COSC = 100pF COSC = 47pF 500 300 510 509 508 507 506 505 504 503 100 ROSC= 51kΩ COSC= 100pF 502 -100 300 MAX1584/85 toc12 MAX1584/85 toc11 1100 501 1 10 100 ROSC (kΩ) 8 250 TEMPERATURE (°C) SWITCHING FREQUENCY (kHz) -50 OSCILLATOR FREQUENCY (kHz) MAX1584/MAX1585 5-Channel Slim DSC Power Supplies 1000 -50 -25 0 25 50 75 TEMPERATURE (°C) _______________________________________________________________________________________ 100 5-Channel Slim DSC Power Supplies AUX MAXIMUM DUTY CYCLE vs. FREQUENCY WHEN THIS DUTY CYCLE IS EXCEEDED FOR 100,000 CLOCK CYCLES, THE MAX1584/MAX1585 SHUT DOWN 86 MAX1584/85 toc13 MAXIMUM DUTY CYCLE (%) 87 STEP-UP STARTUP RESPONSE MAX1584/85 toc14 88 ONSU 5V/div OUTSU 5V/div IOUTSU 200mA/div 0V 85 0V 84 0A 83 82 IIN 1.0A/div 0A 81 COSC = 330pF VIN = 3.5V 80 0 100 200 300 400 500 600 700 800 900 1000 200µs/div FREQUENCY (kHz) STEP-DOWN STARTUP RESPONSE AUX1 STARTUP RESPONSE MAX1584/85 toc15 MAX1584/85 toc16 ONSD 5V/div OUTSD 5V/div 0V 0V ON1 5V/div OUT1 10V/div 0V 0V IOUTSD 200mA/div IOUT1 100mA/div 0A 0A VIN = 3.5V VIN = 3.5V 4ms/div 2ms/div STEP-UP LOAD-TRANSIENT RESPONSE STEP-DOWN LOADTRANSIENT RESPONSE MAX1584/85 toc17 VOUTSU AC-COUPLED 500mV/div 0V 0A MAX1584/85 toc18 IOUT_SU 200mA/div VOUTSU = 5.0V VIN = 3.5V 400µs/div VOUTSD AC-COUPLED 100mV/div 0V 0A VIN = 3.5V VOUT_SD = 1.5V IOUT_SD 100mA/div 400µs/div _______________________________________________________________________________________ 9 MAX1584/MAX1585 Typical Operating Characteristics (continued) (Circuit of Figure 1, TA = +25°C, unless otherwise noted.) 5-Channel Slim DSC Power Supplies MAX1584/MAX1585 Pin Description 10 PIN NAME FUNCTION 1 CC1 AUX1 Controller Compensation Node. Connect a series resistor-capacitor from CC1 to GND to compensate the converter control loop. This pin is actively driven to GND in shutdown, overload, and thermal limit. See the AUX Compensation section. 2 FB1 AUX1 Controller Feedback Input. The feedback threshold is 1.25V. This pin is high impedance in shutdown. 3 PGSD Step-Down Power Ground. Connect all PG_ pins together and to GND with short traces as close as possible to the IC. 4 LXSD Step-Down Converter Switching Node. Connect to the inductor of the step-down converter. LXSD is high impedance in shutdown. 5 PVSD Step-Down Converter Input. PVSD can connect to PVSU, effectively making OUTSD a boost-buck output from the battery. Bypass to GND with a 1µF ceramic capacitor if connected to PVSU. PVSD can also be connected to the battery but should not exceed PVSU by more than a Schottky diode forward voltage. Bypass PVSD with a 10µF ceramic capacitor when connecting to the battery input. A 10kΩ internal resistance connects PVSU and PVSD. 6 ONSD Step-Down Converter On/Off Control Input. Logic high = on; however, turn-on is locked out until the stepup has reached regulation. This pin has an internal 330kΩ pulldown resistance to GND. 7 CCSD Step-Up Converter Compensation Node. Connect a series resistor-capacitor from CCSD to GND to compensate the converter control loop. This pin is actively driven to GND in shutdown, overload, and thermal limit. See the Step-Down Compensation section. 8 FBSD Step-Down Converter Feedback Input. Connect a resistive voltage-divider from OUTSD to FBSD to GND. The FBSD feedback threshold is 1.25V. This pin is high impedance in shutdown. 9 ON1 AUX1 Controller On/Off Input. Logic high = on; however, turn-on is locked out until 1024 OSC cycles after the step-up has reached regulation. This pin has an internal 330kΩ pulldown resistance to GND. 10 ON2 AUX2 Controller On/Off Input. Logic high = on; however, turn-on is locked out until 1024 OSC cycles after the step-up has reached regulation. This pin has an internal 330kΩ pulldown resistance to GND. 11 ON3 AUX3 Controller On/Off Input. Logic high = on; however, turn-on is locked out until 1024 OSC cycles after the step-up has reached regulation. This pin has an internal 330kΩ pulldown resistance to GND. 12 ONSU 13 REF Reference Output. Bypass REF to GND with a 0.1µF or greater capacitor. The maximum allowed load on REF is 200µA. REF is actively pulled to GND when all converters are shut down. 14 FBSU Step-Up Converter Feedback Input. Connect a resistive voltage-divider from PVSU to FBSU to GND. The FBSU feedback threshold is 1.25V. This pin is high impedance in shutdown. 15 CCSU Step-Up Converter Compensation Node. Connect a series resistor-capacitor from CCSU to GND to compensate the converter control loop. This pin is actively driven to GND in shutdown, overload, and thermal limit. See the Step-Up Compensation section. Step-Up Converter On/Off Control. Logic high = on. All other ON_ pins are locked out until 1024 OSC cycles after the step-up DC-DC converter output has reached its final value. This pin has an internal 330kΩ pulldown resistance to GND. ______________________________________________________________________________________ 5-Channel Slim DSC Power Supplies PIN NAME 16 AUX1OK 17 SDOK FUNCTION Open-Drain Power-OK Signal for AUX1 Controller. AUX1OK is low when the AUX1 controller has successfully completed soft-start. This pin is high impedance in shutdown, overload, and thermal limit. Open-Drain Power-OK Signal for Step-Down Converter. SDOK is low when the step-down has successfully completed soft-start. This pin is high impedance in shutdown, overload, and thermal limit. 18 SCF Short-Circuit Flag, Active-Low, Open-Drain Output. SCF is high impedance when overload protection occurs and during startup. SCF can drive high-side PFET switches connected to one or more outputs to completely disconnect the load when the channel turns off in response to a logic command or an overload. See the Status Outputs (SDOK, AUX1OK, SCF) section. 19 OSC Oscillator Control. Connect a timing capacitor from OSC to GND and a timing resistor from OSC to PVSU (or other DC voltage) to set the oscillator frequency between 100kHz and 1MHz. See the Setting the Switching Frequency section. This pin is high impedance in shutdown. 20 PGSU Step-Up Power Ground. Connect all PG_ pins together and to GND with short traces as close to the IC as possible. 21 LXSU Step-Up Converter Switching Node. Connect to the inductor of the step-up converter. LXSU is high impedance in shutdown. 22 PVSU Power Output of the Step-Up DC-DC Converter. Connect the output filter capacitor from PVSU to PGSU. PVSU can also power other converter channels. Connect PVSU to PV at the IC. 23 24 25 26 27 AUX2 Controller Feedback Input. This pin is high impedance in shutdown. FB2 INDL2 DL2 MAX1584 (AUX2 step-up): The FB2 feedback threshold is 1.25V. Connect a resistive voltage-divider from the output voltage to FB2 to GND to set the output voltage. AUX2 Controller Compensation Node. Connect a series resistor-capacitor from CC2 to GND to compensate the control loop. CC2 is actively driven to GND in shutdown and thermal limit. See the AUX Compensation section. CC2 PV MAX1585 (AUX2 inverter): The FB2 feedback threshold is 0V. Connect a resistive voltage-divider from the output voltage to FB2 to REF to set the output voltage. Voltage Input for the AUX2 Gate Driver. The voltage at INDL2 sets the high gate-drive voltage. MAX1585 (AUX2 inverter): Connect INDL2 to the external P channel MOSFET source (typically the battery) to ensure the P channel is completely off when D2 swings high. MAX1584 (AUX2 step-up): Connect INDL2 to PVSU for optimum N-channel gate drive. IC Power Input. Connect PVSU and PV together. AUX2 Controller Gate-Drive Output. DL2 drives between INDL2 and GND. MAX1585: DL2 drives a PFET in an inverter configuration. In shutdown, overload, and thermal limit, DL2 is driven high. MAX1584: DL2 drives an N-channel FET in a boost/flyback configuration. In shutdown, overload, and thermal limit, DL2 is driven low. ______________________________________________________________________________________ 11 MAX1584/MAX1585 Pin Description (continued) 5-Channel Slim DSC Power Supplies MAX1584/MAX1585 Pin Description (continued) PIN NAME 28 DL3 AUX3 Step-Down Controller Gate-Drive Output. Connect to the gate of a P-channel MOSFET. DL3 swings from GND to PVSU and supplies up to 500mA. DL3 is driven to PVSU in shutdown and thermal limit. 29 DL1 AUX1 Step-Up Controller Gate-Drive Output. Connect to the gate of an N-channel MOSFET. DL1 swings from GND to PVSU and supplies up to 500mA. DL1 is driven to GND in shutdown and thermal limit. 30 GND Analog Ground. Connect to all PG_ pins as close to the IC as possible. 31 CC3 AUX3 Step-Down Controller Compensation Node. Connect a series resistor-capacitor from CC3 to FB3 to compensate the converter control loop. This pin is actively driven to GND in shutdown, overload, and thermal limit. See the AUX Compensation section. 32 FB3 PWM Step-Up Controller 3 Feedback Input. Connect a resistive voltage-divider from the output voltage to FB3 to GND to set the output voltage. The FB3 feedback threshold is 1.25V. This pin is high impedance in shutdown. EP Exposed Underside Metal Pad. This pad must be soldered to the PC board to achieve package thermal and mechanical ratings. There is no internal metal or bond wire physically connecting the exposed pad to the GND pin(s). Connecting the exposed pad to ground does not remove the requirement for a good ground connection to the appropriate IC pins. PAD FUNCTION Detailed Description The MAX1584/MAX1585 are complete power-conversion ICs for slim digital still cameras. They can accept input from a variety of sources, including single-cell Li+ batteries and 2-cell alkaline or NiMH batteries, as well as systems designed to accept both battery types. The MAX1584/MAX1585 include five DC-DC converter channels to generate all required voltages (Figure 2 shows a functional diagram): • Synchronous-rectified step-up DC-DC converter with on-chip MOSFETs—Typically supplies 3.3V for main system power or 5V to power other DC-DC converters for boost-buck designs. • Synchronous-rectified step-down DC-DC converter with on-chip MOSFETs—Typically supplies 1.8V for the DSP core. Powering the step-down from the step-up output provides efficient (up to 90%) boostbuck functionality that supplies a regulated output when the battery voltage is above or below the output voltage. The step-down can also be powered from the battery if there is sufficient headroom. • AUX1 step-up controller—Typically used for 15V to bias one or more of the LCD, CCD, and LED backlights. 12 • AUX2 step-up controller (MAX1584)—Typically supplies remaining bias voltages with either a multi-output flyback transformer or a boost converter with charge-pump inverter. Alternately, can power white LEDs for LCD backlighting. • AUX2 inverter controller (MAX1585)—Typically supplies negative CCD bias when high current is needed for large pixel-count CCDs. • AUX3 step-down controller—Typically steps 5V generated at PVSU down to 3.3V for system logic in boost-buck designs. Step-Up DC-DC Converter The step-up DC-DC switching converter is typically used to generate a 5V output voltage from a 1.5V to 4.5V battery input, but any voltage from VIN to 5V can be set. An internal NFET switch and a PFET synchronous rectifier allow conversion efficiencies as high as 95%. Under moderate to heavy loading, the converter operates in a low-noise PWM mode with constant frequency and modulated pulse width. Switching harmonics generated by fixed-frequency operation are consistent and easily filtered. Efficiency is enhanced under light (<75mA typ) loading, by an idle mode that switches the step-up only as needed to service the load. In this mode, the maximum inductor current is 250mA for each pulse. ______________________________________________________________________________________ 5-Channel Slim DSC Power Supplies L3 2µH MAX1585 +15V, 80mA +CCD LCD LED INDL2 AUX2 V-MODE INV PWM P1 DL2 D2 -7.5V -CDD BIAS R8 526kΩ C8 4.7µF AUX1 V-MODE STEP-UP PWM N1 AUX3 V-MODE STEP-DOWN PWM REF L4 10µH DL3 D3 FB3 OSC 3.3V 250mA LOGIC C25 47µF 1.25V REF R14 30.1kΩ R15 18.2kΩ 20kΩ TO PVSU R7 90.9kΩ TO PVSU C23 OR VIN 10µF P2 0.1µF R6 1MΩ FB1 FB2 L6 3.6µH R9 93.1kΩ C6 4.7µF D1 DL1 R22 1.2kΩ C20 560pF TO VIN D4 PV MAX1584/MAX1585 VIN 1.5V TO 4.2V C24 10µF 330pF PVSU R1 47kΩ R2 25kΩ C1 0.01µF R3 20kΩ CCSU CCSD CC1 CC2 CC3 R5 10kΩ CURRENTMODE STEP-UP R4 61.9kΩ C5 1500pF 5V 1A MAIN SYSTEM R12 274kΩ R13 90.9kΩ FBSU PVSD TO FB3 C3 1500pF L1 5µH PGSU C4 470pF C2 4700pF LXSU C11 47µF C19 10µF ONSU ONSD ON1 ON2 ON3 CURRENTMODE STEPDOWN LXSD TO VIN OR PVSU L2 22µH +1.5V 250mA CORE C9 30µF PGSD AUX1OK R10 18.2kΩ FBSD SCF SDOK R11 90.9kΩ GND Figure 1. MAX1584/MAX1585 Typical Application for 2-Cell AA or 1-Cell Li+ Battery ______________________________________________________________________________________ 13 MAX1584/MAX1585 5-Channel Slim DSC Power Supplies PVSU INTERNAL POWER-OK NORMAL MODE STARTUP OSCILLATOR 2.35V MAX1584/ MAX1585 ONSU VREF OVER TEMP REFOK SCF 1V FAULT 100,000 IN CLOCK CYCLE FAULT TIMER CLK ONSU FLTALL TO INTERNAL POWER PV OSC 1.25V REFERENCE REF 150ns ONE-SHOT REF GND CCSU PVSU FAULT LXSU FBSU CURRENT-MODE DC-DC STEP-UP STEP-UP SOFT-START DONE (SUSSD) SOFT-START RAMP GENERATOR PGSU TO VREF ONSU FLTALL CCSD PVSD FAULT FBSD LXSD CURRENT-MODE DC-DC STEP-DOWN SOFT-START RAMP GENERATOR PGSD TO VREF SDOK ONSD SUSSD FLTALL TO AUX_ CHANNELS (SEE FIGURE 3) Figure 2. MAX1584/MAX1585 Functional Diagram 14 ______________________________________________________________________________________ 5-Channel Slim DSC Power Supplies Boost-Buck Operation The step-down input can be powered from the output of the step-up. By cascading these two channels, the step-down output can maintain regulation even as the battery voltage falls below the step-down output voltage. This is especially useful when trying to generate 3.3V from 1-cell Li+ inputs, or 2.5V from 2-cell alkaline or NiMH inputs, or when designing a power supply that must operate from both Li+ and alkaline/NiMH inputs. Compound efficiencies of up to 90% can be achieved when the step-up and step-down are operated in series. Note that the step-up output supplies both the step-up load and the step-down input current when the stepdown is powered from the step-up. The step-down input current reduces the available step-up output current for other loads. Direct Battery Step-Down Operation The step-down converter can also be operated directly from the battery as long as the voltage at PVSD does not exceed PVSU by more than a Schottky diode forward voltage. When using this connection, connect an external Schottky diode from the battery input to PVSU. On the MAX1584/MAX1585, there is an internal 10kΩ resistance from PVSU to PVSD. This adds a small addi- tional current drain (of approximately (VPVSU - VPVSD) / 10kΩ) from PVSU when PVSD is not connected directly to PVSU. Step-down direct battery operation improves efficiency for the step-down output (up to 95%), but restricts the upper limit of the output voltage to 200mV less than the minimum battery voltage. In 1-cell Li+ designs (with a 2.7V min), the output can be set up to 2.5V. In 2-cell alkaline or NiMH designs, the output can be limited to 1.5V or 1.8V, depending on the minimum-allowed cell voltage. The step-down can only be briefly operated in dropout since the MAX1584/MAX1585 fault protection detects the out-of-regulation condition and activates after 100,000 OSC cycles (200ms at fOSC = 500kHz). At that point, all MAX1584/MAX1585 channels shut down. AUX1, AUX2, and AUX3 DC-DC Controllers The three auxiliary controllers operate as fixed-frequency voltage-mode PWM controllers. They do not have internal MOSFETs, so output power is determined by external components. The controllers regulate output voltage by modulating the pulse width of the DL_ drive signal to an external MOSFET switch. The MAX1584 contains two step-up/flyback controllers (AUX1 and AUX2) and one step-down controller (AUX3). The MAX1585 contains one step-up controller (AUX1), one inverting controller (AUX2), and one step-down controller (AUX3). Figure 3 shows a functional diagram of the AUX controllers. The inverting and step-down controllers differ from the step-up controllers only in the gate-drive logic and FB polarity and threshold. The sawtooth oscillator signal at OSC governs timing. At the start of each cycle, DL_ turns on the external MOSFET switch. For step-up controllers, DL_ goes high, while for inverting and step-down controllers, DL_ goes low (to turn on PFETs). The external MOSFET then turns off when the internally level-shifted sawtooth rises above CC_ or when the maximum duty cycle is exceeded. The switch remains off until the start of the next cycle. A transconductance error amplifier forms an integrator at CC_ so that high DC loop gain and accuracy can be maintained. In step-up and step-down controllers, the FB_ threshold is 1.25V, and higher FB_ voltages reduce the MOSFET duty cycle. In inverting controllers, the FB_ threshold is 0V, and lower (more negative) FB_ voltages reduce the MOSFET duty cycle. Auxiliary controllers do not start until the step-up DC-DC output is in regulation. If the step-up, step-down, or any of the auxiliary controllers remains faulted for 100,000 ______________________________________________________________________________________ 15 MAX1584/MAX1585 Step-Down DC-DC Converter The step-down DC-DC converter is optimized for generating low output voltages (down to 1.25V) at high efficiency. Output voltages lower than 1V can be set by adding an additional resistor (see the Applications Information section). The step-down runs from the voltage at PVSD. This pin can be connected directly to the battery if sufficient headroom exists to avoid dropout; otherwise, PVSD can be powered from the output of another converter. The step-down can also operate with the step-up for boost-buck operation. Under moderate to heavy loading, the converter operates in a low-noise PWM mode with constant frequency and modulated pulse width. Efficiency is enhanced under light (<75mA typ) loading by assuming an idle mode during which the step-down switches only as needed to service the load. In this mode, the maximum inductor current is 100mA for each pulse. The stepdown DC-DC is inactive until the step-up DC-DC is in regulation. The step-down also features an open-drain SDOK output that goes low when the step-down output is in regulation. SDOK can be used to drive an external MOSFET switch that gates 3.3V power to the processor after the core voltage is in regulation. This connection is shown in Figure 13. MAX1584/MAX1585 5-Channel Slim DSC Power Supplies FB_ FB2 MAX1584 AUX1 AND AUX2, MAX1585 AUX1 STEP-UP CONTROLLER MAX1585 AUX2 INVERTING CONTROLLER CC_ CC2 R LEVEL SHIFT REFI REF 0.85 REF Q S REFI SOFT-START (REFI RAMPS FROM 0V TO REF IN 1024 OSC CYCLES) R LEVEL SHIFT DL_ Q DL2 S SOFT-START (REFI RAMPS FROM REF TO 0V IN 1024 OSC CYCLES) REF 0.85 REF CLK CLK OSC OSC FAULT PROTECTION FAULT PROTECTION ENABLE ENABLE FB3 MAX1584 AND MAX1585 AUX3 STEP-DOWN CONTROLLER CC3 R LEVEL SHIFT REFI REF 0.85 REF Q DL3 S SOFT-START (REFI RAMPS FROM 0V TO REF IN 1024 OSC CYCLES) CLK OSC FAULT PROTECTION ENABLE Figure 3. AUX Controller Functional Diagrams OSC cycles, then all MAX1584/MAX1585 channels latch off. Maximum Duty Cycle The MAX1584/MAX1585 auxiliary PWM controllers have a guaranteed maximum duty cycle of 80%. In boost designs that employ continuous current, the maximum duty cycle limits the boost ratio so that: 16 1 - VIN / VOUT ≤ 80% With discontinuous inductor current, no such limit exists for the input/output ratio since the inductor has time to fully discharge before the next cycle begins. AUX1 AUX1 can be used for conventional DC-DC boost and flyback designs (Figure 5). Its output (DL1) is designed ______________________________________________________________________________________ 5-Channel Slim DSC Power Supplies AUX2 In the MAX1584, AUX2 is identical to AUX1. In the MAX1585, AUX2 is an inverting controller that generates a regulated negative output voltage, typically for CCD and LCD bias. This is handy in height-limited designs where transformers might not be desired. The AUX2 MOSFET driver (DL2) in the MAX1585 is designed to drive P-channel MOSFETs. DL2 swings from GND to PVSU. See Figure 8 for a typical inverter configuration. AUX3 DC-DC Step-Down Controller AUX3 can be used for conventional DC-DC step-down (buck) designs (Figure 1). Its output (DL3) is designed to drive a P-channel MOSFET and swings from GND to PVSU. Its feedback (FB3) threshold is 1.25V. Master/Slave Configurations The MAX1584/MAX1585 support the MAX1801 slave PWM controllers that obtain input power, a voltage reference, and an oscillator signal directly from the MAX1584/MAX1585 master. The master/slave configuration allows channels to be easily added and minimizes system cost by eliminating redundant circuitry. The slaves also control the harmonic content of noise since their operating frequency is synchronized to that of the MAX1584/MAX1585 master converter. A MAX1801 connection to the MAX1584/MAX1585 is shown in Figure 12. Status Outputs (SDOK, AUX1OK, SCF) The MAX1584/MAX1585 include three versatile status outputs that can provide information to the system. All are open-drain outputs and can directly drive MOSFET switches to facilitate sequencing, disconnect loads during overloads, or perform other hardware-based functions. SDOK pulls low when the step-down has successfully completed soft-start. SDOK goes high impedance in shutdown, overload, and thermal limit. A typical use for SDOK is to enable 3.3V power to the CPU I/O after the CPU core is powered up (Figure 13), thus providing safe sequencing in hardware without system intervention. AUX1OK pulls low when the AUX1 controller has successfully completed soft-start. AUX1OK goes high impedance in shutdown, overload, and thermal limit. A typical use for AUX1OK is to drive a P-channel MOSFET that gates 5V power to the CCD until the +15V CCD bias (generated by AUX1) is powered up (Figure 14). SCF goes high (high impedance, open drain) when overload protection occurs. Under normal operation, SCF pulls low. SCF can drive a high-side P-channel MOSFET switch that can disconnect a load during power-up or when a channel turns off in response to a logic command or an overload. Several connections are possible for SCF. One is shown in Figure 15, where SCF provides load disconnect for the step-up on fault and power-up. Soft-Start The MAX1584/MAX1585 channels feature a soft-start function that limits inrush current and prevents excessive battery loading at startup by ramping the output voltage of each channel up to the regulation voltage. This is accomplished by ramping the internal reference inputs to each channel error amplifier from 0V to the 1.25V reference voltage over a period of 4096 oscillator cycles (16ms at 500kHz) when initial power is applied or when a channel is enabled. Soft-start is not included in the step-up converter in order to avoid limiting startup capability with loading. The step-down soft-start ramp takes half the time (2048 clock cycles) of the other channel ramps. This allows the step-down and AUX3 output (when set to 3.3V) to track each other and rise at nearly the same dV/dt rate on power-up. Once the step-down output reaches its regulation point (1.5V or 1.8V typ), the AUX3 output (3.3V typ) continues to rise at the same ramp rate. Fault Protection The MAX1584/MAX1585 have robust fault and overload protection. After power-up, the device is set to detect an out-of-regulation state that could be caused by an overload or short. If any DC-DC converter channel (step-up, step-down, or any of the auxiliary controllers) remains faulted for 100,000 clock cycles (200ms at 500kHz), then all outputs latch off until the step-up DCDC converter is reinitialized by the ONSU pin or by cycling the input power. The fault-detection circuitry for any channel is disabled during its initial turn-on softstart sequence. An exception to the standard fault behavior is that there is no 100,000 clock-cycle delay in entering the fault state if the step-up output (PVSU) is dragged below its 2.5V UVLO threshold or is shorted. The step-up UVLO immediately triggers and shuts down all channels. The step-up then continues to attempt to start. If the step-up output short remains, these attempts do not succeed since PVSU remains near ground. If a soft-short or overload remains on PVSU, the startup oscillator switches the internal N-channel MOSFET, but fault is retriggered if regulation is not achieved by the ______________________________________________________________________________________ 17 MAX1584/MAX1585 to drive an N-channel MOSFET. Its feedback (FB1) threshold is 1.25V. MAX1584/MAX1585 5-Channel Slim DSC Power Supplies VSU MAX1584 MAX1585 (PARTIAL) ROSC TO VBATT AUX PWM OSC COSC VREF (1.25V) PVSU Q1 DL_ 150ns ONE-SHOT MAX1584 MAX1585 D6 +15V 50mA LCD FB_ NOTE: THIS CIRCUIT CAN OPERATE WITH AUX1 OR AUX2 ON THE MAX1584, AND WITH AUX1 ON THE MAX1585 Figure 4. Oscillator Functional Diagram Figure 5. +15V LCD Bias with Basic Boost Topology end of the soft-start interval. If PVSU is dragged below the input, the overload is supplied by the body diode of the internal synchronous rectifier or by a Schottky diode connected from the battery to PVSU. If desired, this overload current can be interrupted by a P-channel MOSFET controlled by SCF, as shown in Figure 15. activates the internal MOSFET switch to discharge the capacitor within a 150ns interval, and the cycle repeats. The oscillation frequency changes as the main output voltage ramps upward following startup. The oscillation frequency is then constant once the main output is in regulation. Reference Low-Voltage Startup Oscillator The MAX1584/MAX1585 have internal 1.250V references. Connect a 0.1µF ceramic bypass capacitor from REF to GND within 0.2in (5mm) of the REF pin. REF can source up to 200µA and is enabled when ONSU is high and PVSU is above 2.5V. The auxiliary controllers and MAX1801 slave controllers (if connected) each sink up to 30µA REF current during startup. If the application requires that REF be loaded beyond 200µA, buffer REF with a unity-gain amplifier or op amp. The MAX1584/MAX1585 internal control and referencevoltage circuitry receive power from PVSU and do not function when PVSU is less than 2.5V. To ensure lowvoltage startup, the step-up employs a low-voltage startup oscillator that activates at 0.9V if a Schottky rectifier is connected from VBATT to PVSU (1.1V with no Schottky rectifier). The startup oscillator drives the internal N-channel MOSFET at LXSU until PVSU reaches 2.5V, at which point voltage control is passed to the current-mode PWM circuitry. Oscillator All MAX1584/MAX1585 DC-DC converter channels employ fixed-frequency PWM operation. The operating frequency is set by an RC network at the OSC pin. The range of usable settings is 100kHz to 1MHz. When MAX1801 slave controllers are added, they operate at the frequency set by OSC. The oscillator uses a comparator, a 150ns one-shot, and an internal NFET switch in conjunction with an external timing resistor and capacitor (Figure 4). When the switch is open, the capacitor voltage exponentially approaches the step-up output voltage from zero with a time constant given by the product of ROSC and COSC. The comparator output switches high when the capacitor voltage reaches VREF (1.25V). In turn, the one-shot 18 Once in regulation, the MAX1584/MAX1585 operate with inputs as low as 0.7V since internal power for the IC is supplied by PVSU. At low input voltages, the stepup can have difficulty starting into heavy loads (see the Minimum Startup Voltage vs. Load Current graph in the Typical Operating Characteristics section); however, this can be remedied by connecting an external Pchannel load switch driven by SCF so the load is not connected until the PVSU is in regulation (Figure 15). ON_ Control Inputs The step-up converter activates with a high input at ONSU. The step-down and auxiliary DC-DC converters 1, 2, and 3 activate with a high input at ONSD, ON1, ON2, and ON3, respectively. The step-down and auxil- ______________________________________________________________________________________ 5-Channel Slim DSC Power Supplies Design Procedure Setting the Switching Frequency Choose a switching frequency to optimize external component size or circuit efficiency for the particular application. Typically, switching frequencies between 400kHz and 500kHz offer a good balance between component size and circuit efficiency—higher frequencies generally allow smaller components, and lower frequencies give better conversion efficiency. The switching frequency is set with an external timing resistor (ROSC) and capacitor (COSC). At the beginning of a cycle, the timing capacitor charges through the resistor until it reaches VREF. The charge time, t1, is as follows: t1 = -ROSC x COSC x ln(1 - 1.25 / VPVSU) The capacitor voltage then decays to zero over time t2 = 150ns. The oscillator frequency is as follows: fOSC = 1 / (t1 + t2) fOSC can be set from 100kHz to 1MHz. Choose COSC between 22pF and 470pF. Determine ROSC: ROSC = (150ns - 1 / fOSC) / (COSC x ln[1 - 1.25 VPVSU]) See the Typical Operating Characteristics section for fOSC vs. ROSC using different values of COSC. Setting Output Voltages The MAX1584/MAX1585 step-up and step-down converters and the AUX1 controllers have resistoradjustable output voltages. When setting the voltage for all channels except AUX2 on the MAX1585, connect a resistive voltage-divider from the output voltage to the corresponding FB_ input. The FB_ input bias current is less than 100nA, so choose the low-side (FB_-to-GND) resistor (RL) to be 100kΩ or less. Then calculate the high-side (output-to-FB_) resistor (RH): RH = RL [(VOUT / 1.25) - 1] AUX2 is an inverter on the MAX1585, so the FB2 threshold on the MAX1585 is 0V. To set the MAX1585 AUX2 negative output voltage, connect a resistive voltage-divider from the negative output to the FB2 input, and then to REF. The FB2 input bias current is less than 100nA, so choose the REF-side (FB2-to-REF) resistor (RREF) to be 100kΩ or less. Then calculate the top-side (negative output-to-FB2) resistor: RTOP = RREF (-VOUT(AUX2) / 1.25) General Filter-Capacitor Selection The input capacitor in a DC-DC converter reduces current peaks drawn from the battery or other input power source and reduces switching noise in the controller. The impedance of the input capacitor at the switching frequency should be less than that of the input source so high-frequency switching currents do not pass through the input source. The output capacitor keeps output ripple small and ensures control-loop stability. The output capacitor must also have low impedance at the switching frequency. Ceramic, polymer, and tantalum capacitors are suitable, with ceramic exhibiting the lowest ESR and high-frequency impedance. Output ripple with a ceramic output capacitor is approximately: VRIPPLE = IL(PEAK) [1 / (2π x fOSC x COUT)] If the capacitor has significant ESR, the output ripple component due to capacitor ESR is: VRIPPLE(ESR) = IL(PEAK) x ESR Output capacitor specifics are also discussed in each converter’s Compensation section. Step-Up Component Selection The external components required for the step-up are an inductor, an input and output filter capacitor, and a compensation RC. The inductor is typically selected to operate with continuous current for best efficiency. An exception might be if the step-up ratio, (VOUT / VIN), is greater than 1 / (1 DMAX), where DMAX is the maximum PWM duty factor of 80%. When using the step-up channel to boost from a low input voltage, loaded startup is aided by connecting a Schottky diode from the battery to PVSU. See the Minimum Startup Voltage vs. Load Current graph in the Typical Operating Characteristics section. Step-Up Inductor In most step-up designs, a reasonable inductor value (LIDEAL) can be derived from the following equation, which sets continuous peak-to-peak inductor current at half the DC inductor current: LIDEAL = [2VIN(MAX) x D(1 - D)] / (IOUT x fOSC) where D is the duty factor given by: D = 1 - (VIN / VOUT) Given LIDEAL, the consistent peak-to-peak inductor current is 0.5 x IOUT / (1 - D). The peak inductor current is as follows: ______________________________________________________________________________________ 19 MAX1584/MAX1585 iary converters and cannot be activated until PVSU is in regulation. For automatic startup, connect ON_ to PVSU or a logic level greater than 1.6V. MAX1584/MAX1585 5-Channel Slim DSC Power Supplies IIND(PK) = 1.25 x IOUT / (1 - D) Inductance values smaller than LIDEAL can be used to reduce inductor size; however, if much smaller values are used, inductor current rises and a larger output capacitance might be required to suppress output ripple. Step-Up Compensation The inductor and output capacitor are usually chosen first in consideration of performance, size, and cost. The compensation resistor and capacitor are then chosen to optimize control-loop stability. In some cases, it helps to readjust the inductor or output capacitor value to get optimum results. For typical designs, the component values in the circuit of Figure 1 yield good results. The step-up converter employs current-mode control, thereby simplifying the control-loop compensation. When the converter operates with continuous inductor current (typically the case), a right-half-plane zero appears in the loop-gain frequency response. To ensure stability, the control-loop gain should cross over (drop below unity gain) at a frequency (fC) much less than that of the right-half-plane zero. The relevant characteristics for step-up channel compensation are as follows: • Transconductance (from FBSU to CCSU), g MEA (135µS) • Current-sense amplifier transresistance, R CS (0.3V/A) • Feedback regulation voltage, VFB (1.25V) • Step-up output voltage, VSU, in V • Output load equivalent resistance, R LOAD , in Ω = VSUOUT / ILOAD The key steps for step-up compensation are as follows: 1) Place fC sufficiently below the right-half-plane zero (RHPZ) and calculate CC. 2) Select RC based on the allowed load-step transient. RC sets a voltage delta on the CC pin that corresponds to load-current step. 3) Calculate the output-filter capacitor (COUT) required to allow the RC and CC selected. 4) Determine if CP is required (if calculated to be >10pF). 20 For continuous conduction, the right-half-plane zero frequency (fRHPZ) is given by the following: fRHPZ = VSUOUT (1 - D)2 / (2π x L x ILOAD) where D = the duty cycle = 1 - (VIN / VOUT), L is the inductor value, and ILOAD is the maximum output current. Typically, target crossover (f C ) for 1/6 of the RHPZ. For example, if we assume fOSC = 500kHz, VIN = 2.5V, VOUT = 5V, and IOUT = 0.5A, then RLOAD = 10Ω. If we select L = 4.7µH, then: fRHPZ = 5 (2.5 / 5)2 / (2π x 4.7 x 10-6 x 0.5) = 84.65kHz Choose fC = 14kHz. Calculate CC: CC = (VFB / VOUT)(RLOAD / RCS)(gM / 2π x fC)(1 - D) = (1.25 / 5)(10 / 0.3) x (135µS / (6.28 x 14kHz) (2/5) = 6.4nF Choose 6.8nF. Now select R C so transient-droop requirements are met. As an example, if 4% transient droop is allowed, the input to the error amplifier moves 0.04 x 1.25V, or 50mV. The error-amp output drives 50mV x 135µS, or 6.75µA across RC to provide transient gain. Since the current-sense transresistance is 0.3V/A, the value of RC that allows the required load step swing is as follows: RC = 0.3 IIND(PK) / 6.75µA In a step-up DC-DC converter, if LIDEAL is used, output current relates to inductor current by: IIND(PK) = 1.25 x IOUT / (1 - D) = 1.25 x IOUT x VOUT / VIN So for a 500mA output load step with VIN = 2.5V and VOUT = 5V: RC = [1.25(0.3 x 0.5 x 5) / 2)] / 6.75µA = 69.4kΩ Note that the inductor does not limit the response in this case since it can ramp at 2.5V / 4.7µH, or 530mA/µs. The output filter capacitor is then chosen so the COUT RLOAD pole cancels the RC CC zero: COUT x RLOAD = RC x CC For the example: COUT = 68kΩ x 6.8nF / 10Ω = 46µF Choose 47µF for COUT. If the available COUT is substantially different from the calculated value, insert the available C OUT value into the above equation and recalculate RC. Higher substituted COUT values allow a higher RC, which provides higher transient gain and consequently less transient droop. If the output filter capacitor has significant ESR, a zero occurs at the following: ZESR = 1 / (2π x COUT x RESR) ______________________________________________________________________________________ 5-Channel Slim DSC Power Supplies If CP is calculated to be <10pF, it can be omitted. Step-Down Component Selection Step-Down Inductor The external components required for the step-down are an inductor, input and output filter capacitors, and a compensation RC network. The MAX1585/1585 step-down converter provides best efficiency with continuous inductor current. A reasonable inductor value (LIDEAL) can be derived from the following: LIDEAL = [2(VIN) x D(1 - D)] / IOUT x fOSC which sets the peak-to-peak inductor current at half the DC inductor current. D is the duty cycle: D = VOUT / VIN Given LIDEAL, the peak-to-peak inductor current is 0.5 x IOUT. The absolute peak inductor current is 1.25 x IOUT. Inductance values smaller than LIDEAL can be used to reduce inductor size; however, if much smaller values are used, inductor current rises and a larger output capacitance may be required to suppress output ripple. Larger values than LIDEAL can be used to obtain higher output current, but with typically larger inductor size. Step-Down Compensation The relevant characteristics for step-down compensation are as follows: • Transconductance (from FBSD to CCSD), g MEA (135µS) • Current-sense amplifier transresistance, RCS (0.6V/A) • Feedback regulation voltage, VFB (1.25V) • Step-down output voltage, VSD, in V • Output load equivalent resistance, R LOAD , in Ω = VSD / ILOAD The key steps for step-down compensation are as follows: 1) Set the compensation RC zero to cancel the RLOAD COUT pole. 2) Set the loop crossover below 1/10 the switching frequency. If we assume VIN = 3.5V, VOUT = 1.5V, and IOUT = 250mA, then RLOAD = 6Ω. If we select fOSC = 500kHz and L = 22µH, choose fC = 24kHz and calculate CC: CC = (VFB / VOUT)(RLOAD / RCS)(gM / 2π x fC) = (1.25 / 1.5)(6 / 0.6) x (135µS / (6.28 x 40kHz)) = 4.5nF Choose 4.7nF. Now select RC so transient-droop requirements are met. For example, if 4% transient droop is allowed, the input to the error amplifier moves 0.04 x 1.25V, or 50mV. The error-amp output drives 50mV x 135µS, or 6.75µA across RC to provide transient gain. Since the current-sense transresistance is 0.6V/A, the value of RC that allows the required load step swing is as follows: RC = 0.6 x IIND(PK) / 6.75µA In a step-down DC-DC converter, If LIDEAL is used, output current relates to inductor current by the following: IIND(PK) = 1.25 x IOUT So for a 250mA output load step with VIN = 3.5V and VOUT = 1.5V: RC = (1.25 x 0.6 x 0.25) / 6.75µA = 27.8kΩ Choose 27kΩ. The inductor does somewhat limit the response in this case since it ramps at (VIN - VOUT) / 22µH, or (3.5 - 1.5) / 22µH = 90mA/µs. The output filter capacitor is then chosen so the COUT RLOAD pole cancels the RC CC zero: COUT x RLOAD = RC x CC For the example: COUT = 27kΩ x 4.7nF / 6Ω = 21µF Choose 22µF or greater. If the output filter capacitor has significant ESR, a zero occurs at: ZESR = 1 / (2π x COUT x RESR) If ZESR > fC, it can be ignored, as is typically the case with ceramic output capacitors. If ZESR is less than fC, it should be cancelled with a pole set by capacitor CP connected from CCSD to GND: CP = COUT x RESR / RC If CP is calculated to be <10pF, it can be omitted. AUX Controller Component Selection External MOSFET MAX1584/MAX1585 AUX1(step-up) controllers drive external logic-level N-channel MOSFETs. AUX3 (stepdown) controllers drive P-channel MOSFETs. AUX2 (step-up) on the MAX1584 drives an N channel, while AUX2 (inverting) on the MAX1585 drives a P channel. ______________________________________________________________________________________ 21 MAX1584/MAX1585 If ZESR > fC, it can be ignored, as is typically the case with ceramic output capacitors. If ZESR is less than fC, it should be cancelled with a pole set by capacitor CP connected from CCSU to GND: CP = COUT x RESR / RC MAX1584/MAX1585 5-Channel Slim DSC Power Supplies Significant MOSFET selection parameters are as follows: • On-resistance (RDS(ON)) • Maximum drain-to-source voltage (VDS(MAX)) • Total gate charge (QG) • Reverse transfer capacitance (CRSS) DL1 and DL3 swing between PVSU and GND. DL2 swings between INDL2 and GND. Use a MOSFET with on-resistance specified at or below the DL_ drive voltage. The gate charge, QG, includes all capacitance associated with charging the gate and helps to predict MOSFET transition time between on and off states. MOSFET power dissipation is a combination of onresistance and transition losses. The on-resistance loss is as follows: PRDSON = D x IL2 x RDS(ON) where D is the duty cycle, IL is the average inductor current, and RDS(ON) is the MOSFET on-resistance. The transition loss is approximately: PTRANS = (VOUT x IL x fOSC x tT) / 3 where VOUT is the output voltage, IL is the average inductor current, fOSC is the switching frequency, and tT is the transition time. The transition time is approximately QG / IG , where QG is the total gate charge, and IG is the gate-drive current (0.5A typ). The total power dissipation in the MOSFET is as follows: PMOSFET = PRDSON + PTRANS Diode For most AUX applications, a Schottky diode rectifies the output voltage. Schottky low forward voltage and fast recovery time provide the best performance in most applications. Silicon signal diodes (such as 1N4148) are sometimes adequate in low-current (<10mA), high-voltage (>10V) output circuits where the output voltage is large compared to the diode forward voltage. AUX Compensation The auxiliary controllers employ voltage-mode control to regulate their output voltage. Optimum compensation depends on whether the design uses continuous or discontinuous inductor current. AUX Step-Up, Discontinuous Inductor Current When the inductor current falls to zero on each switching cycle, it is described as discontinuous. The inductor is not utilized as efficiently as with continuous current, but in light-load applications, this often has little negative impact since the coil losses may already be low compared to other losses. A benefit of discontinuous 22 inductor current is more flexible loop compensation, and no maximum duty-cycle restriction on boost ratio. To ensure discontinuous operation, the inductor must have a sufficiently low inductance to fully discharge on each cycle. This occurs when: L < [VIN2 (VOUT - VIN) / VOUT3] [RLOAD / (2fOSC)] A discontinuous current boost has a single pole at the following: FP = (2VOUT - VIN) / (2π x RLOAD x COUT x VOUT) Choose the integrator cap so the unity-gain crossover, fC, occurs at fOSC / 10 or lower. For many AUX circuits, such as those powering motors, LEDs, or other loads that do not require fast transient response, it is often acceptable to overcompensate by setting fC at fOSC / 20 or lower. CC is then determined by the following: CC = [2VOUT x VIN / ((2VOUT - VIN) x VRAMP)] [VOUT / (K(VOUT - VIN))]1/2 [(VFB / VOUT)(gM / (2π x fC))] where: K = 2L x fOSC / RLOAD and VRAMP is the internal voltage ramp of 1.25V. The CC RC zero is then used to cancel the fP pole, so: RC = RLOAD x COUT x VOUT / [(2VOUT - VIN) x CC] AUX Step-Up, Continuous Inductor Current Continuous inductor current can sometimes improve boost efficiency by lowering the ratio between peak inductor current and output current. It does this at the expense of a larger inductance value that requires larger size for a given current rating. With continuous inductor-current boost operation, there is a right-halfplane zero, ZRHP, at the following: ZRHP = (1 - D)2 RLOAD / (2π x L) where (1 - D) = VIN / VOUT (in a boost converter) There is a complex pole pair at the following: f0 = VOUT / [2π x VIN (L x COUT)1/2] If the zero due to the output capacitor capacitance and ESR is less than 1/10 the right-half-plane zero: ZCOUT = 1 / (2π x COUT x RESR) < ZRHP / 10 Then choose CC so the crossover frequency fC occurs at ZCOUT. The ESR zero provides a phase boost at crossover: CC = (VIN / VRAMP)(VFB / VOUT)(gM / (2π x ZCOUT)) Choose RC to place the integrator zero, 1 / (2π x RC x CC), at f0 to cancel one of the pole pairs: RC = VIN (L x COUT)1/2 / (VOUT x CC) ______________________________________________________________________________________ 5-Channel Slim DSC Power Supplies In that case: CC = (VIN / VRAMP)(VFB / VOUT)(gM / (2π x fC)) Place: 1 / (2π x RC x CC) = 1 / (2π x RLOAD x COUT), so that RC = RLOAD x COUT / CC Or, reduce the inductor value for discontinuous operation. AUX3 Step-Down Compensation It is expected that most AUX3 step-down applications employ continuous inductor current to optimize inductor size and efficiency. To ensure stability, the controlloop gain should cross over (drop below unity gain) at a frequency (fC) much less than that of the switching frequency. The relevant characteristics for voltage-mode stepdown compensation are as follows: • Transconductance (from FB3 to CC3), gMEA (135µS) • Oscillator ramp voltage, VRAMP (1.25V) • Feedback regulation voltage, VFB (1.25V) • Output voltage, VOUT3, in V • Output load equivalent resistance, RLOAD, in Ω = VOUT3 / ILOAD • Characteristic impedance of the LC output filter, RO = (L / C)1/2 The key steps for AUX3 step-down compensation are as follows: 1) Place fC sufficiently below the switching frequency (fOSC / 10). 2) Calculate COUT. 3) Calculate the complex pole pair due to the output LC filter. 4) Add two zeros to cancel the complex pole pair. 5) Add two high-frequency poles to optimize gain and phase margin. If we assume V IN = 5V, V OUT = 3.3V, and I OUT = 300mA, then RLOAD = 11Ω. If we select fOSC = 500kHz and L = 10µH, select the crossover frequency to be 1/10 the OSC frequency: fC = fOSC / 10 = 50kHz For 3.3V output, select R14 = 30.1kΩ and R15 = 18.2kΩ. See the Setting Output Voltages section. Calculate the equivalent impedance, REQ: REQ = RSOURCE + RL + ESR + RDS(ON) where RSOURCE is the output impedance of the source (this is the output impedance of the step-up converter when the AUX3 step-down is powered from the stepup), RL is the inductor DC resistance, ESR is the filtercapacitor equivalent resistance, and RDS(ON) is the on-resistance of the external MOSFET. The output impedance of the step-up converter (RSOURCE) is approximately 1Ω at f0. Since the sum of RL + ESR + RDS(ON) is small compared to 1Ω, assume REQ = 1Ω. Choose COUT so RO is less than REQ / 2: COUT > L / [(REQ / 2)2] = 10µH / 0.25 = 40µF Choose COUT = 47µF: C4 = (VIN / VRAMP)(1 / [2π x R14 x fC]) = (5 / 1.25)(1/ [2π x 30.1k x 50kHz) = 423pF Choose C4 = 470pF. Cancel one pole of the complex pole pair by placing the R4 C4 zero at 0.75 f0. The complex pole pair is at the following: f0 = 1 / [2π(L x COUT)1/2] = 1 / [2π(10µH x 47µF)1/2] = 7.345kHz Choose R4 = 1 / (2π x C4 x 0.75 x f0) = 1 / (2π x 470pF x 0.75 x 7.345kHz) z Choose R4 = 61.9kΩ (standard 1% value). Ensure that R4 > 2 / gMEA = 14.8kΩ. If it is not greater, reselect R14 and R15. Cancel the second pole of the complex pole pair by placing the R14 C20 zero at 1.25 x f0. C20 = 1 / (2π x R14 x 1.25 x f0) = 1 / (2π x 30.1k x 1.25 x 7.345kHz) = 576pF Choose C20 = 560pF. Roll off the gain below the switching frequency by placing a pole at fOSC / 2: R22 = 1 / (2π x C20 [fOSC / 2]) = 1 / (2π x 560pF x 250kHz) = 1.137kΩ Choose R22 = 1.2kΩ. If the output filter capacitor has significant ESR, a zero occurs at the following: ZESR = 1 / (2π x COUT x RESR) Use the R4 C22 pole to cancel the ESR zero: C22 = COUT x RESR / R4 If C22 is calculated to be <10pF, it can be omitted. ______________________________________________________________________________________ 23 MAX1584/MAX1585 If ZCOUT is not less than ZRHP / 10 (as is typical with ceramic output capacitors) and continuous conduction is required, then cross the loop over before ZRHP and f0: fC < f0SC / 10, and fC < ZRHP / 10 MAX1584/MAX1585 5-Channel Slim DSC Power Supplies MAX1585 AUX2 Inverter Compensation, Discontinuous Inductor Current If the load current is very low (40mA or less), discontinuous current is preferred for simple loop compensation and freedom from duty-cycle restrictions on the inverter input-output ratio. To ensure discontinuous operation, the inductor must have a sufficiently low inductance to fully discharge on each cycle. This occurs when: L < [VIN / (|VOUT| + VIN)]2 RLOAD / (2fOSC) A discontinuous current inverter has a single pole at: fP = 2 / (2π x RLOAD x COUT) Choose the integrator cap so the unity-gain crossover, fC, occurs at fOSC / 10 or lower. Note that for many AUX circuits that do not require fast transient response, it is often acceptable to overcompensate by setting fC at fOSC / 20 or lower. CC is then determined by the following: CC = [VIN / (K1/2 x VRAMP][VREF / (VOUT + VREF)] [gM / (2π x fC)] where: K = 2L x fOSC / RLOAD, and VRAMP is the internal voltage ramp of 1.25V. The CC RC zero then is used to cancel the fP pole, so: RC = (RLOAD x COUT) / (2 CC) MAX1585 AUX2 Inverter Compensation, Continuous Inductor Current Continuous inductor current may be more suitable for larger load currents (50mA or more). It improves efficiency by lowering the ratio between peak inductor current and output current. It does this at the expense of a larger inductance value that requires larger size for a given current rating. With continuous inductor-current inverter operation, there is a right-half-plane zero, ZRHP, at: ZRHP = [(1 - D)2 / D] x RLOAD / (2π x L) where D = |VOUT| / (|VOUT| + VIN) (in an inverter). There is a complex pole pair at: f0 = (1 - D) / (2π(L x C)1/2) If the zero due to the output-capacitor capacitance and ESR is less than 1/10 the right-half-plane zero: ZCOUT = 1 / (2π x COUT x RESR) < ZRHP / 10 Then choose CC so the crossover frequency, fC, occurs at ZCOUT. The ESR zero provides a phase boost at crossover. CC = (VIN / VRAMP)[VREF / (VREF + |VOUT|)][gM / (2π x ZCOUT)] 24 Choose RC to place the integrator zero, 1 / (2π x RC x CC), at f0 to cancel one of the pole pairs: RC = (L x COUT)1/2 / [(1 - D) x CC] If ZCOUT is not less than ZRHP / 10 (as is typical with ceramic output capacitors) and continuous conduction is required, then cross the loop over before ZRHP and f0: fC < f0 / 10, and fC < ZRHP / 10 In that case: CC = (VIN / VRAMP)[VREF / (VREF + |VOUT|)][gM / (2π x fC)] Place: 1 / (2π x RC x CC) = 1 / (2π x RLOAD x COUT), so that RC = RLOAD x COUT / CC Or, reduce the inductor value for discontinuous operation. Applications Information LED, LCD, and Other Boost Applications Any AUX channel can be used for a wide variety of step-up applications. These include generating 5V or some other voltage for motor or actuator drive, generating 15V or a similar voltage for LCD bias, or generating a step-up current source to efficiently drive a series array of white LEDs to display backlighting. Figures 5 and 6 show examples of these applications. Multiple-Output Flyback Circuits Some applications require multiple voltages from a single converter channel. This is often the case when generating voltages for CCD bias or LCD power. Figure 7 shows a two-output flyback configuration with AUX_. The controller drives an external MOSFET that switches the transformer primary. Two transformer secondaries generate the output voltages. Only one positive output voltage can be fed back, so the other voltages are set by the turns ratio of the transformer secondaries. The load stability of the other secondary voltages depends on transformer leakage, inductance, and winding resistance. Voltage regulation is best when the load on the secondary that is not fed back is small compared to the load on the one that is fed back. Regulation also improves if the load current range is limited. Consult the transformer manufacturer for the proper design for a given application. ______________________________________________________________________________________ 5-Channel Slim DSC Power Supplies MAX1584/MAX1585 TO VBATT 1µF TO VBATT PVSU 1µF +15V 50mA CCD+ MAX1584 MAX1585 (PARTIAL) DL_ AUX_ PWM WHITE LEDS AUX PWM PVSU Q1 D2 -7.5V 30mA CCD- DL_ FB_ 62Ω (FOR 20mA) MAX1585 (PARTIAL) NOTE: THIS CIRCUIT CAN OPERATE WITH AUX1 OR AUX2 ON THE MAX1584, AND WITH AUX1 ON THE MAX1585. FB_ NOTE: THIS CIRCUIT CAN OPERATE WITH AUX1 OR AUX2 ON THE MAX1584, AND WITH AUX1 ON THE MAX1585. Figure 6. AUX_ Channel Powering a White LED Step-Up Current Source Figure 7. +15V and -7.5V CCD Bias with Transformer Transformerless Inverter for Negative CCD Bias (AUX2, MAX1585) connection is shown in Figure 10. This circuit is somewhat unique in that a positive-output linear regulator is able to regulate a negative voltage output. It does this by controlling the charge current flowing to the flying capacitor rather than directly regulating at the output. On the MAX1585, AUX2 is set up to drive an external Pchannel MOSFET in an inverting configuration. DL2 drives low to turn on the MOSFET, and FB2 has inverted polarity and a 0V threshold. This is useful for generating negative CCD bias without a transformer, particularly with high pixel-count cameras that have a greater negative CCD load current. Figures 1 and 8 show such a configuration for the MAX1585. Boost with Charge Pump for Positive and Negative Outputs Another method of producing bipolar output voltages without a transformer is with an AUX controller and a charge-pump circuit as shown in Figure 9. When MOSFET Q1 turns off, the voltage at its drain rises to supply current to VOUT+. At the same time, C1 charges to the voltage VOUT+ through D1. When the MOSFET turns on, C1 discharges through D3, thereby charging C3 to VOUTminus the drop across D3 to create roughly the same voltage as VOUT+ at VOUT-, but with inverted polarity. If different magnitudes are required for the positive and negative voltages, a linear regulator can be used at one of the outputs to achieve the desired voltages. One such SEPIC Boost-Buck The MAX1584/MAX1585s’ internal switch step-up and step-down can be cascaded to make a high-efficiency boost-buck converter, but it is sometimes desirable to build a second boost-buck converter with an AUX_ controller. One type of step-up/step-down converter is the SEPIC, shown in Figure 11. Inductors L1 and L2 can be separate inductors or can be wound on a single core and coupled like a transformer. Typically, a coupled inductor improves efficiency since some power is transferred through the coupling so less power passes through the coupling capacitor (C2). Likewise, C2 should have low ESR to improve efficiency. The ripple-current rating must be greater than the larger of the input and output currents. The MOSFET (Q1) drain-source voltage rating and the rectifier (D1) reverse-voltage rating must exceed the sum of the input and output voltages. Other types of step-up/step-down circuits are a flyback converter and a step-up converter followed by a linear regulator. ______________________________________________________________________________________ 25 MAX1584/MAX1585 5-Channel Slim DSC Power Supplies +15V 20mA TO VBATT MAX1585 (PARTIAL) TO VBATT FB_ AUX_ PWM INDL2 DL2 PVSU AUX2 INVERTING PWM -7.5V 20mA DL_ -7.5V 100mA RTOP FB2 MAX1584/MAX1585 (PARTIAL) RREF IN REF SHDN GND OUT Figure 8. Regulated -7.5V Negative CCD Bias Provided by Conventional Inverter (MAX1585 Only) FB_ +1.25V L1 10µH D2 TO VBATT VOUT+ +15V 20mA C2 1µF 1µF R1 1MΩ FB_ AUX_ PWM PVSU Q1 DL_ C1 1µF D1 R2 90.9kΩ D3 MAX1616 NOTE: THIS CIRCUIT CAN OPERATE WITH AUX1 OR AUX2 ON THE MAX1584, AND WITH AUX1 ON THE MAX1585. Figure 10. +15V and -7.5V CCD Bias Without Transformer from AUX-Driven Boost and Charge Pump. A positive linear regulator (MAX1616) regulates the negative output of the charge pump. VOUT-15V C3 10mA 1µF INPUT 1-CELL Li+ MAX1584 MAX1585 (PARTIAL) VSU L2 L1 NOTE: THIS CIRCUIT CAN OPERATE WITH AUX1 OR AUX2 ON THE MAX1584, AND WITH AUX1 ON THE MAX1585. D1 PV DL_ Figure 9. ±15V Output from AUX-Driven Boost with ChargePump Inversion Adding a MAX1801 Slave The MAX1801 is a 6-pin SOT slave DC-DC controller that can be connected to generate additional output voltages. It does not generate its own reference or oscillator. Instead, it uses the reference and oscillator of the MAX1584/MAX1585 (Figure 12). OUTPUT 3.3V PVSU PART OF MAX1584 MAX1585 (PARTIAL) C2 Q1 R1 FB_ R2 NOTE: THIS CIRCUIT CAN OPERATE WITH AUX1 OR AUX2 ON THE MAX1584, AND WITH AUX1 ON THE MAX1585. Figure 11. SEPIC Converter for Additional Boost-Buck Channel 26 ______________________________________________________________________________________ 5-Channel Slim DSC Power Supplies TO VBATT VOUT PVSU DL IN MAX1584 MAX1585 OSC (PARTIAL) MAX1801 OSC MAX1584 MAX1585 (PARTIAL) PVSU AUX1 PWM FB COMP REF GND DL1 D6 15V 100mA REF DCON FB1 AUX1OK PV PVSU Figure 12. Adding a PWM Channel with an External MAX1801 Slave Controller TO VBATT CURRENTMODE STEP-UP PWM MAX1584/MAX1585 (PARTIAL) AUX3 V-MODE STEP-DOWN PWM LXSU GATED +5V TO CCD VSU +5V L2 TO PVSU PGSU DL3 FBSU 3.3V LOGIC Figure 14. AUX1OK drives an external PFET that switches 5V to the CCD only after the +15V CCD bias supply is in regulation. FB3 ON3 SDOK TO VBATT OR PVSU Selection section also apply to add-on MAX1801 slave controllers. For more details, refer to the MAX1801 data sheet. Figure 13. Using SDOK to Gate 3.3V Power to CPU After the Core Voltage Is in Regulation Using SDOK and AUX1OK for Power Sequencing The SDOK goes low when the step-down reaches regulation. Some microcontrollers with low-voltage cores require the high-voltage (3.3V) I/O rail not be powered up until the core has a valid supply. The circuit in Figure 13 accomplishes this by driving the gate of a PFET connected between the 3.3V output and the processor I/O supply. Figure 14 shows a similar application where AUX1OK gates 5V power to the CCD only after the +15V output is in regulation. Alternately, power sequencing can also be implemented by connecting RC networks to delay the appropriate converter ON_ inputs. The MAX1801 controller operation and design are similar to that of the MAX1584/MAX1585 AUX controllers. All comments in the AUX Controller Component Using SCF for Full-Load Startup The SCF output goes low only after the step-up reaches regulation. It can be used to drive a P-channel MOSFET switch that turns off the load of a selected supply in the PVSD CURRENTMODE STEP-DOWN PWM LXSD TO VBATT OR PVSU +1.5V PGSD ______________________________________________________________________________________ 27 MAX1584/MAX1585 TO BATT MAX1584/MAX1585 5-Channel Slim DSC Power Supplies VSU 3.3V MAX1584 MAX1585 (PARTIAL) PVSU PV PV PVSU VSU +5V MAX1584 MAX1585 (PARTIAL) PVSD 10µF TO VBATT CURRENT-MODE STEP-UP PWM LXSU L2 LXSD CURRENT-MODE STEP-DOWN 4.7µH VSD 0.8V 22µF PGSU FBSD SCF PGSD FBSD R3 100kΩ VFBSD 1.25V R1 56kΩ OK PWR-ON OR FAULT R2 100kΩ Figure 15. SCF controls a PFET load switch to disconnect all 5V loads on fault. This also allows full-load startup. Figure 16. Setting PVSD for Outputs Below 1.25V event of an overload. Or, it can remove the load until the supply reaches regulation, effectively allowing fullload startup. Figure 15 shows such a connection for the step-up output. spreadsheet and test estimated resistor values. A good starting point is with 100kΩ at R2 and R3. Setting SDOUT Below 1.25V The step-down feedback voltage is 1.25V. With a standard two-resistor feedback network, the output voltage can be set to values between 1.25V and the input voltage. If a step-down output voltage less than 1.25V is desired, it can be set by adding a third feedback resistor from FBSD to a voltage higher than 1.25V (the stepup output is a convenient voltage for this) as shown in Figure 16. The equation governing output voltage in Figure 16’s circuit is as follows: 0 = [(VSD - VFBSD) / R1] + [(0 - VFBSD) / R2] + [(VSU VFBSD) / R3] where VSD is the output voltage, VFBSD is 1.25V, and VSU is the step-up output voltage. Any available voltage that is higher than 1.25V can be used as the connection point for R3 in Figure 16, and for the VSD term in the equation. Since there are multiple solutions for R1, R2, and R3, the above equation cannot be written in terms of one resistor. The best method for determining resistor values is to enter the above equation into a 28 Designing a PC Board Good PC board layout is important to achieve optimal performance from the MAX1584/MAX1585. Poor design can cause excessive conducted and/or radiated noise. Conductors carrying discontinuous currents and any high-current path should be made as short and wide as possible. A separate low-noise ground plane containing the reference and signal grounds should connect to the power-ground plane at only one point to minimize the effects of power-ground currents. Typically, the ground planes are best joined right at the IC. Keep the voltage-feedback network very close to the IC, preferably within 0.2in (5mm) of the FB_ pin. Nodes with high dV/dt (switching nodes) should be kept as small as possible and should be routed away from high-impedance nodes such as FB_. Refer to the MAX1584/MAX1585 evaluation kit data sheet for a full PC board example. Chip Information TRANSISTOR COUNT: 8234 PROCESS: BiCMOS ______________________________________________________________________________________ 5-Channel Slim DSC Power Supplies b CL 0.10 M C A B D2/2 D/2 PIN # 1 I.D. QFN THIN.EPS D2 0.15 C A D k 0.15 C B PIN # 1 I.D. 0.35x45 E/2 E2/2 CL (NE-1) X e E E2 k L DETAIL A e (ND-1) X e CL CL L L e e 0.10 C A C 0.08 C A1 A3 PROPRIETARY INFORMATION TITLE: PACKAGE OUTLINE 16, 20, 28, 32L, QFN THIN, 5x5x0.8 mm APPROVAL COMMON DIMENSIONS DOCUMENT CONTROL NO. REV. 21-0140 C 1 2 EXPOSED PAD VARIATIONS NOTES: 1. DIMENSIONING & TOLERANCING CONFORM TO ASME Y14.5M-1994. 2. ALL DIMENSIONS ARE IN MILLIMETERS. ANGLES ARE IN DEGREES. 3. N IS THE TOTAL NUMBER OF TERMINALS. 4. THE TERMINAL #1 IDENTIFIER AND TERMINAL NUMBERING CONVENTION SHALL CONFORM TO JESD 95-1 SPP-012. DETAILS OF TERMINAL #1 IDENTIFIER ARE OPTIONAL, BUT MUST BE LOCATED WITHIN THE ZONE INDICATED. THE TERMINAL #1 IDENTIFIER MAY BE EITHER A MOLD OR MARKED FEATURE. 5. DIMENSION b APPLIES TO METALLIZED TERMINAL AND IS MEASURED BETWEEN 0.25 mm AND 0.30 mm FROM TERMINAL TIP. 6. ND AND NE REFER TO THE NUMBER OF TERMINALS ON EACH D AND E SIDE RESPECTIVELY. 7. DEPOPULATION IS POSSIBLE IN A SYMMETRICAL FASHION. 8. COPLANARITY APPLIES TO THE EXPOSED HEAT SINK SLUG AS WELL AS THE TERMINALS. 9. DRAWING CONFORMS TO JEDEC MO220. 10. WARPAGE SHALL NOT EXCEED 0.10 mm. PROPRIETARY INFORMATION TITLE: PACKAGE OUTLINE 16, 20, 28, 32L, QFN THIN, 5x5x0.8 mm APPROVAL DOCUMENT CONTROL NO. REV. 21-0140 C 2 2 Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 ____________________ 29 © 2003 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products. MAX1584/MAX1585 Package Information (The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information, go to www.maxim-ic.com/packages.)