PHILIPS NE5210D

Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
DESCRIPTION
PIN CONFIGURATION
The NE5210 is a 7kΩ transimpedance wide band, low noise
amplifier with differential outputs, particularly suitable for signal
recovery in fiber-optic receivers. The part is ideally suited for many
other RF applications as a general purpose gain block.
D Package
GND2
1
14
OUT (–)
GND2
2
13
GND2
NC
3
12
OUT (+)
IIN
4
11
GND1
NC
5
10
GND1
VCC1
6
9
GND1
VCC2
7
8
GND1
FEATURES
• Low noise: 3.5pA/√Hz
• Single 5V supply
• Large bandwidth: 280MHz
• Differential outputs
• Low input/output impedances
• High power supply rejection ratio
• High overload threshold current
• Wide dynamic range
• 7kΩ differential transresistance
TOP VIEW
SD00318
• Wideband gain block
• Medical and scientific instrumentation
• Sensor preamplifiers
• Single-ended to differential conversion
• Low noise RF amplifiers
• RF signal processing
APPLICATIONS
• Fiber-optic receivers, analog and digital
• Current-to-voltage converters
ORDERING INFORMATION
DESCRIPTION
14-Pin Plastic Small Outline (SO) Package
TEMPERATURE RANGE
ORDER CODE
DWG #
0 to +70°C
NE5210D
SOT108-1
ABSOLUTE MAXIMUM RATINGS
SYMBOL
VCC
PARAMETER
RATING
Power supply
UNIT
6
V
TA
Operating ambient temperature range
0 to +70
°C
TJ
Operating junction temperature range
-55 to +150
°C
Storage temperature range
-65 to +150
°C
TSTG
PDMAX
Power dissipation, TA=25°C (still air)1
IINMAX
Maximum input current2
1.0
W
5
mA
NOTES:
1. Maximum dissipation is determined by the operating ambient temperature and the thermal resistance: θJA=125°C/W.
2. The use of a pull-up resistor to VCC for the PIN diode, is recommended.
RECOMMENDED OPERATING CONDITIONS
SYMBOL
VCC
PARAMETER
RATING
UNIT
Supply voltage
4.5 to 5.5
V
TA
Ambient temperature range
0 to +70
°C
TJ
Junction temperature range
0 to +90
°C
1995 Apr 26
1
853-1654 15170
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
DC ELECTRICAL CHARACTERISTICS
Min and Max limits apply over operating temperature range at VCC=5V, unless otherwise specified. Typical data applies at VCC=5V and
TA=25°C.
SYMBOL
PARAMETER
TEST CONDITIONS
LIMITS
Min
Typ
Max
UNIT
VIN
Input bias voltage
0.6
0.8
0.95
V
VO±
Output bias voltage
2.8
3.3
3.7
V
VOS
Output offset voltage
0
80
mV
ICC
Supply current
21
26
32
mA
IOMAX
Output sink/source current1
3
4
mA
IIN
Input current (2% linearity)
Test Circuit 8, Procedure 2
±120
±160
µA
IINMAX
Maximum input current
overload threshold
Test Circuit 8, Procedure 4
±160
±240
µA
NOTES:
1. Test condition: output quiescent voltage variation is less than 100mV for 3mA load current.
AC ELECTRICAL CHARACTERISTICS
Typical data and Min/Max limits apply at VCC=5V and TA=25°C.
SYMBOL
PARAMETER
TEST CONDITIONS
LIMITS
Min
Typ
Max
UNIT
RT
Transresistance
(differential output)
DC tested, RL=∞
Test Circuit 8, Procedure 1
4.9
7
10
kΩ
RO
Output resistance
(differential output)
DC tested
16
30
42
Ω
RT
Transresistance
(single-ended output)
DC tested, RL=∞
2.45
3.5
5
kΩ
RO
Output resistance
(single-ended output)
DC tested
8
15
21
Ω
f3dB
Bandwidth (-3dB)
Test Circuit 1, TA=25°C
200
280
RIN
Input resistance
60
Ω
CIN
Input capacitance
7.5
pF
∆R/∆V
Transresistance power
supply sensitivity
∆R/∆T
IN
IT
MHz
VCC=5±0.5V
9.6
20
%/V
Transresistance ambient
temperature sensitivity
∆TA=TA MAX-TA MIN
0.05
0.1
%/°C
RMS noise current spectral density
(referred to input)
f=10MHz, TA=25°C
Test Circuit 2
3.5
6
pA/√Hz
Integrated RMS noise current over
the bandwidth (referred to input)
CS=01
CS=1pF
TA=25°C
Test Circuit 2
∆f=100MHz
37
∆f=200MHz
56
∆f=300MHz
71
∆f=100MHz
40
∆f=200MHz
66
∆f=300MHz
89
nA
PSRR
Power supply rejection ratio2
(VCC1=VCC2)
DC tested, ∆VCC=0.1V
Equivalent AC test circuit 3
20
36
dB
PSRR
Power supply rejection ratio2
(VCC1)
DC tested, ∆VCC=0.1V
Equivalent AC test circuit 4
20
36
dB
PSRR
Power supply rejection ratio2
(VCC2)
DC tested, ∆VCC=0.1V
Equivalent AC test circuit 5
65
dB
f=0.1MHz, Test Circuit 6
23
dB
PSRR
1995 Apr 26
Power supply rejection
configuration)
ratio2
(ECL
2
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
AC ELECTRICAL CHARACTERISTICS (Continued)
SYMBOL
PARAMETER
TEST CONDITIONS
VOMAX
Maximum output voltage swing differential
VINMAX
tR
LIMITS
Min
Typ
RL=∞
Test Circuit 8, Procedure 3
2.4
3.2
Maximum input amplitude for
output duty cycle of 50±5%3
Test Circuit 7
650
Rise time for 50 mVP-P
output signal4
Test Circuit 7
UNIT
Max
VP-P
mVP-P
0.8
1.2
ns
NOTES:
1. Package parasitic capacitance amounts to about 0.2pF
2. PSRR is output referenced and is circuit board layout dependent at higher frequencies. For best performance use RF filter in VCC line.
3. Guaranteed by linearity and overload tests.
4. tR defined as 20-80% rise time. It is guaranteed by a -3dB bandwidth test.
TEST CIRCUITS
SINGLE-ENDED
DIFFERENTIAL
NETWORK ANALYZER
RT S-PARAMETER TEST SET
PORT 1
V OUT
V IN
RO ZO
PORT 2
R 2 S21 R
11 S22
33
S22
RT V OUT
V IN
R O 2Z O
R 4 S21 R
11 S22
66
S22
5V
VCC1
0.1µF
ZO = 50
VCC2
OUT
33
0.1µF
ZO = 50
R = 1k
IN
DUT
33
0.1µF
OUT
RL = 50
50
GND1
GND2
Test Circuit 1
SPECTRUM ANALYZER
5V
VCC1
OUT
NC
IN
AV = 60DB
VCC2
33
DUT
33
0.1µF
ZO = 50
0.1µF
OUT
RL = 50
GND1
GND2
Test Circuit 2
1995 Apr 26
3
SD00319
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
TEST CIRCUITS (Continued)
NETWORK ANALYZER
5V
10µF
S-PARAMETER TEST SET
0.1µF
PORT 1
PORT 2
CURRENT PROBE
1mV/mA
10µF
0.1µF
16
VCC1
CAL
VCC2
33
0.1µF
OUT
50
100
BAL.
IN
33
TRANSFORMER
NH0300HB
TEST
UNBAL.
OUT
0.1µF
GND1
GND2
Test Circuit 3
NETWORK ANALYZER
5V
10µF
S-PARAMETER TEST SET
0.1µF
PORT 1
CURRENT PROBE
1mV/mA
10µF
0.1µF
5V
PORT 2
16
VCC2
10µF
CAL
VCC1
33
0.1µF
OUT
0.1µF
IN
50
100
BAL.
33
TRANSFORMER
NH0300HB
TEST
UNBAL.
OUT
GND1
GND2
0.1µF
Test Circuit 4
1995 Apr 26
4
SD00320
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
TEST CIRCUITS (Continued)
NETWORK ANALYZER
5V
10µF
S-PARAMETER TEST SET
0.1µF
PORT 1
CURRENT PROBE
1mV/mA
10µF
0.1µF
5V
PORT 2
16
VCC2
VCC1
10µF
CAL
33
0.1µF
OUT
0.1µF
IN
50
100
BAL.
33
TRANSFORMER
NH0300HB
TEST
UNBAL.
OUT
GND1
0.1µF
GND2
Test Circuit 5
NETWORK ANALYZER
S-PARAMETER TEST SET
GND
PORT 1
PORT 2
CURRENT PROBE
1mV/mA
10µF
0.1µF
16
GND1
CAL
GND2
33
0.1µF
OUT
50
100
BAL.
IN
33
TRANSFORMER
NH0300HB
TEST
UNBAL.
OUT
VCC1
5.2V
VCC2
0.1µF
10µF
0.1µF
Test Circuit 6
1995 Apr 26
5
SD00321
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
TEST CIRCUITS (Continued)
PULSE GEN.
VCC1
VCC2
33
0.1µF
OUT
0.1µF 1k IN
A
DUT
OUT
ZO = 50Ω
OSCILLOSCOPE
33
B
0.1µF
ZO = 50Ω
50
GND1
GND2
Measurement done using
differential wave forms
Test Circuit 7
SD00322
1995 Apr 26
6
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
TEST CIRCUITS (Continued)
Typical Differential Output Voltage
vs Current Input
5V
+
OUT +
IN
VOUT (V)
DUT
–
OUT –
IIN (µA)
GND1
GND2
2.00
DIFFERENTIAL OUTPUT VOLTAGE (V)
1.60
1.20
0.80
0.40
0.00
–0.40
–0.80
–1.20
–1.60
–2.00
–400
–320
–240
–160
–80
0
80
160
240
320
400
CURRENT INPUT (µA)
NE5210 TEST CONDITIONS
Procedure 1
RT measured at 60µA
RT = (VO1 – VO2)/(+60µA – (–60µA))
Where: VO1 Measured at IIN = +60µA
VO2 Measured at IIN = –60µA
Procedure 2
Linearity = 1 – ABS((VOA – VOB) / (VO3 – VO4))
Where: VO3 Measured at IIN = +120µA
VO4 Measured at IIN = –120µA
R T ( 120A) V
OA
OB
V
R T ( 120A) V
OB
OB
V
Procedure 3
VOMAX = VO7 – VO8
Where: VO7 Measured at IIN = +260µA
VO8 Measured at IIN = –260µA
Procedure 4
IIN Test Pass Conditions:
VO7 – VO5 > 20mV and V06 – VO5 > 20mV
Where: VO5 Measured at IIN = +160µA
VO6 Measured at IIN = –160µA
VO7 Measured at IIN = +260µA
VO8 Measured at IIN = –260µA
Test Circuit 8
1995 Apr 26
7
SD00323
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
TYPICAL PERFORMANCE CHARACTERISTICS
NE5210 Supply Current
vs Temperature
NE5210 Output Bias Voltage
vs Temperature
28
26
24
22
20
10 20 30 40 50 60 70
3.42
3.38
PIN 12
3.34
AMBIENT TEMPERATURE (°C)
10 20 30 40 50 60 70
DIFFERENTIAL OUTPUT VOLTAGE (V)
PIN 14
OUTPUT BIAS VOLTAGE (V)
5.5V
5.0V
4.5V
800
750
10 20 30 40 50 60 70
3.9
3.5
+125°C
+85°C
0
INPUT CURRENT (µA)
3.1
4.5V
2.9
10 20 30 40 50 60 70
80
2.0
5.5V
4.5V
0
4.5V
5.0V
–2.0
–300.0
DIFFERENTIAL OUTPUT SWING (V)
0
4.5V
–20
5.0V
–40
5.5V
–60
80
4.0
3.8
DC TESTED
RL = ∞
3.6
5.5V
3.4
3.2
5.0V
3.0
2.8
4.5V
2.6
2.4
2.2
–10 0
10 20 30 40 50 60 70
AMBIENT TEMPERATURE (°C)
80
5.5V
0
INPUT CURRENT (µA)
+300.0
Differential Output Voltage
vs Input Current
DIFFERENTIAL OUTPUT VOLTAGE (V)
NE5210 Differential Output Swing
vs Temperature
VOS = VOUT12 – VOUT14
+300.0
5.0V
AMBIENT TEMPERATURE (°C)
20
AMBIENT TEMPERATURE (°C)
5.0V
3.3
2.7
–10 0
80
NE5210 Output Offset Voltage
vs Temperature
10 20 30 40 50 60 70
5.5V
3.7
AMBIENT TEMPERATURE (°C)
–80
–10 0
3.0
Differential Output Voltage
vs Input Current
4.1
700
–10 0
–55°C
2.5
–300.0
80
NE5210 Output Bias Voltage
vs Temperature
900
850
+125°C
AMBIENT TEMPERATURE (°C)
NE5210 Input Bias Voltage
vs Temperature
INPUT BIAS VOLTAGE (mV)
PIN 14
3.30
–10 0
80
+25°C +85°C
VCC = 5.0V
3.46
OUTPUT VOLTAGE (V)
30
18
–10 0
OUTPUT OFFSET VOLTAGE (mV)
4.5
3.50
OUTPUT BIAS VOLTAGE (V)
TOTAL SUPPLY CURRENT (mA)
(I CC1+ I CC2)
32
Output Voltage
vs Input Current
2.0
0
–55°C
–2.0
–300.0
+25°C
+85°C
+125°C
0
INPUT CURRENT (µA)
+300.0
SD00324
1995 Apr 26
8
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
TYPICAL PERFORMANCE CHARACTERISTICS (Continued)
8
5.5V
7
5.5V
6
4.5V
4
5.0V
3
5
4.5V
4
5.0V
3
2
2
1
1
0
0
–1
–1
10
100
FREQUENCY (MHz)
1000
1
Gain vs Frequency
–55°C
6
–55°C
4
3
+125°C
25°C
2
4
3
0
–1
–1
10
100
FREQUENCY (MHz)
1000
180
PIN 12
VCC = 5V
TA = 25°C
90
4
0
3
2
6
–90
1
GAIN (dB)
5
–180
10
100
FREQUENCY (MHz)
1000
7.8
7.6
5.5V
5.0V
4.5V
7.4
–10 0
10 20 30 40 50 60 70
80
PIN 12
SINGLE-ENDED
400
5.5V
RL = Ω
350
5.0V
300
4.5V
10 20 30 40 50 60 70
80
AMBIENT TEMPERATURE (°C)
NE5210 Typical
Bandwidth Distribution
(70 Parts from 4 Wafer Lots)
PIN 14
VCC = 5V
TA = 25°C
360
50
270
40
5
4
180
3
2
90
0
–1
1
8.0
200
–10 0
1000
1
0
–1
1
10
100
FREQUENCY (MHz)
8
7
8.2
250
Gain and Phase Shift
vs Frequency
PHASE ( o )
GAIN (dB)
6
25°C
+125°C
1
Gain and Phase Shift
vs Frequency
7
85°C
1
0
1
–55°C
5
2
+85°C
1
RL = ∞
450
BANDWIDTH (MHz)
5
8.4
NE5210 Bandwidth vs Temperature
PIN 14
VCC = 5V
7
GAIN (dB)
GAIN (dB)
6
8.6
AMBIENT TEMPERATURE (°C)
8
PIN 12
VCC = 5V
+125°C
1000
Gain vs Frequency
8
7
10
100
FREQUENCY (MHz)
POPULATION (%)
1
8
PIN 12
VCC = 5V
RL = 50Ω
PHASE ( o )
GAIN (dB)
6
5
8
PIN 12
VCC = 5V
RL = 50Ω
GAIN (dB)
7
NE5210 Differential Transresistance
vs Temperature
Gain vs Frequency
DIFFERENTIAL TRANSRESISTANCE (kΩ )
Gain vs Frequency
PIN 12
SINGLE-ENDED
RL = 50Ω
VCC = 5.0V
TA = 25°C
30
20
10
0
0
10
100
FREQUENCY (MHz)
1000
223
255
287
319
351
FREQUENCY (MHz)
383
SD00325
1995 Apr 26
9
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
TYPICAL PERFORMANCE CHARACTERISTICS (Continued)
NE5210 Output Resistance
vs Temperature
NE5210 Output Resistance
vs Temperature
16
16
PIN 14 ROUT
5.0V
14
5.0V
PIN 12 ROUT
13
12
–10 0 10 20 30 40 50 60 70 80
AMBIENT TEMPERATURE (°C)
POWER SUPPLY REJECTION RATIO (dB)
OUTPUT RESISTANCE (Ω )
VCC = 5.0V
TA = 25°C
50
PIN 12
40
30
20
10
PIN 14
0
0.1
1
10
FREQUENCY (MHz)
5.0V
5.5V
13
PIN 14
OUTPUT REFERRED
16
4.5V
15
5.0V
5.5V
14
13
–10 0 10 20 30 40 50 60 70 80
AMBIENT TEMPERATURE (°C)
NE5210 Power Supply Rejection Ratio
vs Temperature
80
60
4.5V
14
12
–10 0 10 20 30 40 50 60 70 80
AMBIENT TEMPERATURE (°C)
Output Resistance
vs Frequency
70
15
100 200
40
39
38
Group Delay
10
VCC1 = VCC2 = 5.0V
∆VCC = ±0.1V
DC TESTED
OUTPUT REFERRED
DELAY (ns)
15
17
PIN 12
OUTPUT REFERRED
OUTPUT RESISTANCE (Ω )
VCC = 5.0V
DC TESTED
OUTPUT RESISTANCE (Ω )
OUTPUT RESISTANCE (Ω )
17
NE5210 Output Resistance
vs Temperature
37
36
8
VCC = 5V
6
TA = 25°C
4
2
0
35
34
33
–10 0 10 20 30 40 50 60 70 80
AMBIENT TEMPERATURE (°C)
0.1 20 40
60 80 100 120 140 160 180 200
FREQUENCY (MHz)
Output Step Response
VCC = 5V
TA = 25°C
20mV/Div
0
2
4
6
8
10
(ns)
12
14
16
18
20
SD00326
1995 Apr 26
10
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
THEORY OF OPERATION
Transimpedance amplifiers have been widely used as the
preamplifier in fiber-optic receivers. The NE5210 is a wide
bandwidth (typically 280MHz) transimpedance amplifier designed
primarily for input currents requiring a large dynamic range, such as
those produced by a laser diode. The maximum input current before
output stage clipping occurs at typically 240µA. The NE5210 is a
bipolar transimpedance amplifier which is current driven at the input
and generates a differential voltage signal at the outputs. The
forward transfer function is therefore a ratio of the differential output
voltage to a given input current with the dimensions of ohms. The
main feature of this amplifier is a wideband, low-noise input stage
which is desensitized to photodiode capacitance variations. When
connected to a photodiode of a few picoFarads, the frequency
response will not be degraded significantly. Except for the input
stage, the entire signal path is differential to provide improved
power-supply rejection and ease of interface to ECL type circuitry. A
block diagram of the circuit is shown in Figure 1. The input stage
(A1) employs shunt-series feedback to stabilize the current gain of
the amplifier. The transresistance of the amplifier from the current
source to the emitter of Q3 is approximately the value of the
feedback resistor, RF=3.6kΩ. The gain from the second stage (A2)
and emitter followers (A3 and A4) is about two. Therefore, the
differential transresistance of the entire amplifier, RT is
RT OUTPUT +
A3
INPUT
A1
A2
RF
A4
OUTPUT –
SD00327
Figure 1. NE5210 – Block Diagram
BANDWIDTH CALCULATIONS
The input stage, shown in Figure 3, employs shunt-series feedback
to stabilize the current gain of the amplifier. A simplified analysis can
determine the performance of the amplifier. The equivalent input
capacitance, CIN, in
parallel with the source, IS, is approximately 7.5pF, assuming that
CS=0 where CS is the external source capacitance.
V OUT(diff)
2R F 2(3.6K) 7.2kW
I IN
Since the input is driven by a current source the input must have a
low input resistance. The input resistance, RIN, is the ratio of the
incremental input voltage, VIN, to the corresponding input current, IIN
and can be calculated as:
V
RF
3.6K 51W
R IN IN 71
I IN
1 A VOL
The single-ended transresistance of the amplifier is typically 3.6kΩ.
The simplified schematic in Figure 2 shows how an input current is
converted to a differential output voltage. The amplifier has a single
input for current which is referenced to Ground 1. An input current
from a laser diode, for example, will be converted into a voltage by
the feedback resistor RF. The transistor Q1 provides most of the
open loop gain of the circuit, AVOL≈70. The emitter follower Q2
minimizes loading on Q1. The transistor Q4, resistor R7, and VB1
provide level shifting and interface with the Q15 – Q16 differential
pair of the second stage which is biased with an internal reference,
VB2. The differential outputs are derived from emitter followers Q11 –
Q12 which are biased by constant current sources. The collectors of
Q11 – Q12 are bonded to an external pin, VCC2, in order to reduce
the feedback to the input stage. The output impedance is about 17Ω
single-ended. For ease of performance evaluation, a 33Ω resistor is
used in series with each output to match to a 50Ω test system.
More exact calculations would yield a higher value of 60Ω.
Thus CIN and RIN will form the dominant pole of the entire amplifier;
f 3dB 2p
1
R IN C IN
Assuming typical values for RF = 3.6kΩ, RIN = 60Ω, CIN = 7.5pF
f 3dB 2p
1
354MHz
7.5pF 60
VCC1
VCC2
R3
R1
Q2
INPUT
R13
Q4
Q11
+
Q3
Q1
R12
Q12
Q15
R2
R14
GND1
Q16
R7
PHOTODIODE
OUT–
R15
+
OUT+
VB2
R5
R4
GND2
SD00328
Figure 2. Transimpedance Amplifier
1995 Apr 26
11
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
For a given wavelength λ; (meters)
Energy of one Photon = hc watt sec (Joule)
l
Where h=Planck’s Constant = 6.6 × 10-34 Joule sec.
c = speed of light = 3 × 108 m/sec
c / λ = optical frequency (Hz)
No. of incident photons/sec= where P=optical incident power
VCC
IC1
R1
INPUT
Q2
IB
IIN
R3
Q3
Q1
R2
VIN
IF
P
No. of incident photons/sec = hs
l
VEQ3
where P = optical incident power
RF
P
No. of generated electrons/sec = h @ hs
l
R4
where η = quantum efficiency
SD00329
Figure 3. Shunt-Series Input Stage
no. of generated electron hole paris
no. of incident photons
+
The operating point of Q1, Figure 2, has been optimized for the
lowest current noise without introducing a second dominant pole in
the pass-band. All poles associated with subsequent stages have
been kept at sufficiently high enough frequencies to yield an overall
single pole response. Although wider bandwidths have been
achieved by using a cascode input stage configuration, the present
solution has the advantage of a very uniform, highly desensitized
frequency response because the Miller effect dominates over the
external photodiode and stray capacitances. For example, assuming
a source capacitance of 1pF, input stage voltage gain of 70, RIN =
60Ω then the total input capacitance, CIN = (1+7.5) pF which will
lead to only a 12% bandwidth reduction.
P
hs
N I + h @
@ e Amps (Coulombsń sec.)
l
where e = electron charge = 1.6 × 10-19 Coulombs
h @e
Responsivity R = hs Amp/watt
l
I + P@R
Assuming a data rate of 400 Mbaud (Bandwidth, B=200MHz), the
noise parameter Z may be calculated as:1
Z +
NOISE
where Z is the ratio of RMS noise output to the peak response to a
single hole-electron pair. Assuming 100% photodetector quantum
efficiency, half mark/half space digital transmission, 850nm
lightwave and using Gaussian approximation, the minimum required
optical power to achieve 10-9 BER is:
Most of the currently installed fiber-optic systems use non-coherent
transmission and detect incident optical power. Therefore, receiver
noise performance becomes very important. The input stage
achieves a low input referred noise current (spectral density) of
3.5pA/√Hz. The transresistance configuration assures that the
external high value bias resistors often required for photodiode
biasing will not contribute to the total noise system noise. The
equivalent input RMS noise current is strongly determined by the
quiescent current of Q1, the feedback resistor RF, and the
bandwidth; however, it is not dependent upon the internal
Miller-capacitance. The measured wideband noise was 66nARMS in
a 200MHz bandwidth.
P avMIN + 12 hc B Z + 12 2.3 @ 10 *19
l
200 @ 10 6 2063
+ 1139nW + * 29.4dBm
where h is Planck’s Constant, c is the speed of light, λ is the
wavelength. The minimum input current to the NE5210, at this input
power is:
I avMIN + qP avMIN l
hc
DYNAMIC RANGE CALCULATIONS
The electrical dynamic range can be defined as the ratio of
maximum input current to the peak noise current:
*9
@ 10 *19
+ 1139 @ 10 @ 1.6
2.3 @ 10 *19
= 792nA
Electrical dynamic range, DE, in a 200MHz bandwidth assuming
IINMAX = 240µA and a wideband noise of IEQ=66nARMS for an
external source capacitance of CS = 1pF.
D E + 20log
Choosing the maximum peak overload current of IavMAX=240µA, the
maximum mean optical power is:
(Max. input current) (PK)
(Peak noise current) (RMS) @ Ǹ 2
P avMAX +
(240 @ 10 *6)
+ 20 log
+ 68dB
(Ǹ 2 66 10 *9)
hcI avMAX
*19
+ 2.3 @ 10 *19 240 @ 10 *6
l q
1.6 @ 10
Thus the optical dynamic range, DO is:
In order to calculate the optical dynamic range the incident optical
power must be considered.
1995 Apr 26
I EQ
66 @ 10 *9
+
+ 2063
qB
(1.6 @ 10 *19)(200 @ 10 6)
DO = PavMAX - PavMIN = -4.6 -(-29.4) = 24.8dB.
12
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
quiescent values of 3.3V (for a 5V supply), then the circuit may be
oscillating. Input pin layout necessitates that the photodiode be
physically very close to the input and Ground 1. Connecting Pins 3
and 5 to Ground 1 will tend to shield the input but it will also tend to
increase the capacitance on the input and slightly reduce the
bandwidth.
This represents the maximum limit attainable with the NE5210
operating at 200MHz bandwidth, with a half mark/half space digital
transmission at 850nm wavelength.
APPLICATION INFORMATION
Package parasitics, particularly ground lead inductances and
parasitic capacitances, can significantly degrade the frequency
response. Since the NE5210 has differential outputs which can feed
back signals to the input by parasitic package or board layout
capacitances, both peaking and attenuating type frequency
response shaping is possible. Constructing the board layout so that
Ground 1 and Ground 2 have very low impedance paths has
produced the best results. This was accomplished by adding a
ground-plane stripe underneath the device connecting Ground 1,
Pins 8–11, and Ground 2, Pins 1 and 2 on opposite ends of the
SO14 package. This ground-plane stripe also provides isolation
between the output return currents flowing to either VCC2 or Ground
2 and the input photodiode currents to flowing to Ground 1. Without
this ground-plane stripe and with large lead inductances on the
board, the part may be unstable and oscillate near 800MHz. The
easiest way to realize that the part is not functioning normally is to
measure the DC voltages at the outputs. If they are not close to their
As with any high-frequency device, some precautions must be
observed in order to enjoy reliable performance. The first of these is
the use of a well-regulated power supply. The supply must be
capable of providing varying amounts of current without significantly
changing the voltage level. Proper supply bypassing requires that a
good quality 0.1µF high-frequency capacitor be inserted between
VCC1 and VCC2, preferably a chip capacitor, as close to the package
pins as possible. Also, the parallel combination of 0.1µF capacitors
with 10µF tantalum capacitors from each supply, VCC1 and VCC2, to
the ground plane should provide adequate decoupling. Some
applications may require an RF choke in series with the power
supply line. Separate analog and digital ground leads must be
maintained and printed circuit board ground plane should be
employed whenever possible.
Figure 4 depicts a 50Mb/s TTL fiber-optic receiver using the BPF31,
850nm LED, the NE5210 and the NE5214 post amplifier.
+VCC
GND
47µF
C1
C2
.01µF
D1
LED
1
LED
IN1B
20
CPKDET
3
THRESH
4
GNDA
5
FLAG
100pF
IN1A
19
L2
10µH
6
C10
C11
µ
.01µF
10 F
L3
10µH
C12
C13
.01µF
JAM
7
VCCD
8
VCCA
9
GNDD
10
TTLOUT
CAZP 18
CAZN
NE5214
2
17
GND
VCC
7
9
GND
VCC
6
10
GND
NC
5
IIN
4
8
100pF
C9
R3
47k
L1
10µH
C7
C8
11
0.1µF
GND
NE5210
R2
220
OUT1B 16
12
OUT
NC
3
IN8B
15
13
GND
GND
2
OUT1A
14
14
OUT
GND
1
IN8A
13
RHYST
12
C4
.01µF
R1
100
C5
1.0µF
C3
10µF
.01µF
C6
BPF31
OPTICAL
INPUT
RPKDET 11
10µF
R4
4k
VOUT (TTL)
NOTE:
The NE5210/NE5217 combination can operate at data rates in excess of 100Mb/s NRZ
The capacitor C7 decreases the NE5210 bandwidth to improve overall S/N ratio in the DC–50MHz band, but does create extra high frequency noise
on the NE5210 VCC pin(s).
Figure 4. A 50Mb/s Fiber Optic Receiver
1995 Apr 26
13
SD00330
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
1
14
OUT (–)
GND 2
13
2
GND 2
GND 2
12
3
OUT (+)
NC
INPUT
11
4
NC
10
GND 1
GND 1
5
GND 1
VCC1
9
6
ECN No.: 06027
1992 Mar 13
VCC 2
7
8
GND 1
SD00488
Figure 5. NE5210 Bonding Diagram
carriers, it is impossible to guarantee 100% functionality through this
Die Sales Disclaimer
process. There is no post waffle pack testing performed on
Due to the limitations in testing high frequency and other parameters
individual die.
at the die level, and the fact that die electrical characteristics may
shift after packaging, die electrical parameters are not specified and
Since Philips Semiconductors has no control of third party
die are not guaranteed to meet electrical characteristics (including
procedures in the handling or packaging of die, Philips
temperature range) as noted in this data sheet which is intended
Semiconductors assumes no liability for device functionality or
only to specify electrical characteristics for a packaged device.
performance of the die or systems on any die sales.
All die are 100% functional with various parametrics tested at the
wafer level, at room temperature only (25°C), and are guaranteed to
be 100% functional as a result of electrical testing to the point of
wafer sawing only. Although the most modern processes are
utilized for wafer sawing and die pick and place into waffle pack
1995 Apr 26
Although Philips Semiconductors typically realizes a yield of 85%
after assembling die into their respective packages, with care
customers should achieve a similar yield. However, for the reasons
stated above, Philips Semiconductors cannot guarantee this or any
other yield on any die sales.
14