AD AD6636

150 MSPS Wideband
Digital Down-Converter (DDC)
AD6636
6 programmable digital AGC loops with 96 dB range
Synchronous serial I/O operation (SPI®-, SPORT-compatible)
Supports 8-bit or 16-bit microport modes
3.3 V I/O, 1.8 V CMOS core
User-configurable built-in self-test (BIST) capability
JTAG boundary scan
FEATURES
4/6 independent wideband processing channels
Processes 6 wideband carriers (UMTS, CDMA2000)
4 single-ended or 2 LVDS parallel input ports
(16 linear bit plus 3-bit exponent) running at 150 MHz
Supports 300 MSPS input using external interface logic
3 16-bit parallel output ports operating up to 200 MHz
Real or complex input ports
Quadrature correction and dc correction for complex inputs
Supports output rate up to 34 MSPS per channel
RMS/peak power monitoring of input ports
Programmable attenuator control for external gain ranging
3 programmable coefficient FIR filters per channel
2 decimating half-band filters per channel
APPLICATIONS
Multicarrier, multimode digital receivers
GSM, EDGE, PHS, UMTS, WCDMA, CDMA2000, TD-SCDMA
Micro and pico cell systems, software radios
Broadband data applications
Instrumentation and test equipment
Wireless local loop
In-building wireless telephony
CLKB
CIC5
M = 1-32
FIR1
HB1
M = Byp, 2
FIR2
HB2
M = Byp, 2
MRCF
DRCF
M = 1-16
CRCF
M = 1-16
LHB
L = Byp, 2
NCO
CIC5
M = 1-32
FIR1
HB1
M = Byp, 2
FIR2
HB2
M = Byp, 2
MRCF
DRCF
M = 1-16
CRCF
M = 1-16
LHB
L = Byp, 2
EXPB [2:0]
CLKC
ADC C/CI
NCO
CMOS
REAL
PORTS
A, B,
C,D
EXPD [2:0]
______
RESET
SYNC [3:0]
LVDS
PORTS
AB, CD
FIR2
HB2
M = Byp, 2
MRCF
DRCF
M = 1-16
CRCF
M = 1-16
AGC
FIR2
HB2
M = Byp, 2
NCO
CIC5
M = 1-32
FIR1
HB1
M = Byp, 2
FIR2
HB2
M = Byp, 2
MRCF
DRCF
M = 1-16
CRCF
M = 1-16
LHB
L = Byp, 2
NCO
CIC5
M = 1-32
FIR1
HB1
M = Byp, 2
FIR2
HB2
M = Byp, 2
MRCF
DRCF
M = 1-16
CRCF
M = 1-16
LHB
L = Byp, 2
PLL CLOCK
MULTIPLIER
NOTE: CHANNELS RENDERED AS
16-BIT
MICROPORT INTERFACE
MRCF
DRCF
M = 1-16
SPORT/SPI INTERFACE
ARE AVAILABLE ONLY IN 6-CHANNEL PART
M = DECIMATION
CRCF
M = 1-16
PA
LHB
L = Byp, 2
FIR1
HB1
M = Byp, 2
PEAK/
RMS
MEAS.
I,Q
CORR.
FIR1
HB1
M = Byp, 2
CIC5
M = 1-32
NCO
CMOS
EXPC [2:0] COMPLEX
PORTS
(AI, AQ)
CLKD (BI, BQ)
ADC D/CQ
CIC5
M = 1-32
DATA ROUTER MATRIX
ADC B/AQ
PB
LHB
L = Byp, 2
PC
JTAG
L = INTERPOLATION
04998-0-001
EXPA [2:0]
NCO
PARALLEL PORTS
ADC A/AI
INPUT MATRIX
CLKA
DATA ROUTING
FUNCTIONAL BLOCK DIAGRAM
Figure 1.
Rev. 0
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infringements of patents or other rights of third parties that may result from its use.
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www.analog.com
Fax: 781.326.8703
© 2004 Analog Devices, Inc. All rights reserved.
AD6636
TABLE OF CONTENTS
Product Description......................................................................... 3
FIR Half-Band Block.................................................................. 29
Product Highlights ....................................................................... 4
Intermediate Data Router ......................................................... 32
Specifications..................................................................................... 5
Mono-Rate RAM Coefficient Filter (MRCF) ......................... 32
Electrical Characteristics ............................................................. 5
Decimating RAM Coefficient Filter (DRCF) ........................ 33
General Timing Characteristics ................................................. 6
Channel RAM Coefficient Filter (CRCF) ............................... 35
Microport Timing Characteristics ............................................. 7
Interpolating Half-Band Filter.................................................. 36
Serial Port Timing Characteristics ............................................. 8
Output Data Router ................................................................... 37
Explanation of Test Levels for Specifications............................ 8
Automatic Gain Control............................................................ 39
Absolute Maximum Ratings............................................................ 9
Parallel Port Output ................................................................... 43
Thermal Characteristics .............................................................. 9
User-Configurable Built-In Self-Test (BIST) .......................... 47
ESD Caution.................................................................................. 9
Chip Synchronization ................................................................ 47
Pin Configuration and Function Descriptions........................... 10
Serial Port Control ..................................................................... 48
Pin Listing for Power, Ground, Data and Address Buses ...... 12
Microport .................................................................................... 52
Timing Diagrams............................................................................ 13
JTAG Boundary Scan................................................................. 53
Theory of Operation ...................................................................... 19
Memory Map .................................................................................. 54
ADC Input Port .......................................................................... 19
Reading the Memory Map Table.............................................. 54
PLL Clock Multiplier ................................................................. 20
Global Register Map .................................................................. 56
ADC Gain Control ..................................................................... 21
Input Port Register Map ............................................................ 59
ADC Input Port Monitor Function.......................................... 22
Channel Register Map ............................................................... 62
Quadrature I/Q Correction Block............................................ 24
Output Port Register Map ......................................................... 67
Input Crossbar Matrix ............................................................... 26
Design Notes ................................................................................... 70
Numerically Controlled Oscillator (NCO) ............................. 26
Outline Dimensions ....................................................................... 72
Fifth-Order CIC Filter ............................................................... 28
Ordering Guide .......................................................................... 72
REVISION HISTORY
8/04—Revision 0: Initial Version
Rev. 0 | Page 2 of 72
AD6636
PRODUCT DESCRIPTION
The AD6636 is a digital down-converter intended for IF
sampling or oversampled baseband radios requiring widebandwidth input signals. Optimized for the demanding filtering
requirements of wideband standards, such as CDMA2000,
UMTS, and TD-SCDMA, the AD6636 is designed for radio
systems that use either an IF sampling ADC or a baseband
sampling ADC.
The AD6636 channels have the following signal processing
stages: a frequency translator, a fifth-order cascaded integrated
comb filter, two sets of cascaded fixed-coefficient FIR and halfband filters, three cascaded programmable coefficient sum-ofproduct FIR filters, an interpolating half-band filter (IHB), and
a digital automatic gain control (AGC) block. Multiple modes
are supported for clocking data into and out of the chip and
provide flexibility for interfacing to a wide variety of digitizers.
Programming and control are accomplished via serial or
microport interfaces.
Input ports can take input data at up to 150 MSPS. Up to
300 MSPS input data can be supported using two input ports
(some external interface logic is required) and two internal
channels processing in tandem. Biphase filtering in output data
router is selected to complete the combined filtering mode. The
four input ports can operate in CMOS mode, or two ports can
be combined for LVDS input mode. The maximum input data
rate for each input port is 150 MHz.
Frequency translation is accomplished with a 32-bit complex
numerically controlled oscillator (NCO). It has greater than
110 dBc SDFR. This stage translates either a real or complex
input signal from IF (intermediate frequency) to a baseband
complex digital output. Phase and amplitude dither can be
enabled on-chip to improve spurious performance of the NCO.
A 16-bit phase-offset word is available to create a known phase
relationship between multiple AD6636 chips or channels. The
NCO also can be bypassed so that baseband I and Q inputs can
be provided directly from baseband sampling ADC through
input ports.
Following frequency translation is a fifth-order CIC filter with a
programmable decimation between 1 and 32. This filter is used
to lower the sample rate efficiently, while providing sufficient
alias rejection at frequencies with higher frequency offsets from
the signal of interest.
Following the CIC5 are two sets of filters. Each set has a nondecimating FIR filter and a decimate-by-2 half-band filter. The
FIR1 filter provides about 30 dB of rejection, while the HB1
filter provides about 77 dB of rejection. They can be used
together to achieve a 107 dB stopband alias rejection, or they
can be individually bypassed to save power. The FIR2 filter
provides about 30 dB of rejection, while the HB2 filter provides
about 65 dB of rejection. The filters can be used either together
to achieve more than 95 dB stopband alias rejection, or can be
individually bypassed to save power. FIR1 and HB1 filters can
run with a maximum input rate of 150 MSPS. In contrast, FIR2
and HB2 can run with a maximum input rate of 75 MSPS
(input rate to FIR2 and HB2 filters).
The programmable filtering is divided into three cascaded RAM
coefficient filters (RCFs) for flexible and power efficient
filtering. The first filter in the cascade is the MRCF, consisting
of a programmable nondecimating FIR. It is followed by
programmable FIR filters (DRCF) with decimation from 1
to 16. They can be used either together to provide high rejection
filters, or independently to save power. The maximum input rate
to the MRCF is one-fourth of PLL clock rate.
The CRCF (Channel RCF) is the last programmable FIR filter
with programmable decimation from 1 to 16. It typically is used
to meet the spectral mask requirements for the air standard of
interest. This could be an RRC, anti-aliasing filter or any other
real data filter. Decimation in preceding blocks is used to keep
the input rate of this stage as low as possible for the best filter
performance.
The last filter stage in the chain is an interpolate-by-2 half-band
filter, which is used to up-sample the CRCF output to produce
higher output oversampling. Signal rejection requirements for
this stage are relaxed because preceding filters already have
filtered the blockers and adjacent carriers.
Each input port of the AD6636 has its own clock used for
latching onto the input data, but Input Port A clock (CLKA) is
used also as the input for an on-board PLL clock multiplier. The
output of the PLL clock is used for processing all filters and
processing blocks beyond the data router following CIC filter.
The PLL clock can be programmed to have a maximum clock
rate of 200 MHz.
A data routing block (DR) is used to distribute data from the
CICs to the various channel filters. This block allows multiple
back end filter chains to work together to process high
bandwidth signals or to make even sharper filter transitions
than a single channel can perform. It also can allow complex
filtering operations to be achieved in the programmable filters.
The digital AGC provides the user with scaled digital outputs
based on the rms level of the signal present at the output of the
digital filters. The user can set the requested level and time
constant of the AGC loop for optimum performance of the
postprocessor. This is a critical function in the base station for
CDMA applications where the power level must be well
controlled going into the RAKE receivers. It has programmable
clipping and rounding control to provide different output
resolutions.
Rev. 0 | Page 3 of 72
AD6636
The overall filter response for the AD6636 is the composite of
all the combined filter stages. Each successive filter stage is
capable of narrower transition bandwidths, but requires a
greater number of CLK cycles to calculate the output. More
decimation in the first filter stage minimizes overall power
consumption. Data from the device is interfaced to a
DSP/FPGA/baseband processor via either high speed parallel
ports (preferred) or a DSP-compatible microprocessor interface.
The AD6636 is available both in 4-channel and 6-channel
versions. The data sheet primarily discusses the 6-channel part.
The only difference between the 6-channel and 4-channel
devices is that on the 4-channel version, Channels 4 and 5 are
not available (see Figure 1). The 4-channel device still has the
same input ports, output ports, and memory map. The memory
map section for Channels 4 and 5 can be programmed and read
back, but it serves no purpose.
PRODUCT HIGHLIGHTS
•
Six independent digital filtering channels
•
101 dB SNR noise performance, 110 dB spurious
performance
•
Four input ports capable of 150 MSPS input data rates
•
RMS/peak power monitoring of input ports and 96 dB
range AGCs before the output ports
•
Three programmable RAM coefficient filters, three halfband filters, two fixed coefficient filters, and one fifth-order
CIC filter per channel
•
Complex filtering and biphase filtering (300 MSPS ADC
input) by combining filtering capability of multiple
channels
•
Three 16-bit parallel output ports operating at up to
200 MHz clock
•
Blackfin®- and TigerSHARC®-compatible 16-bit
microprocessor port
•
Synchronous serial communications port is compatible
with most serial interface standards, SPORT, SPI, and SSR
Rev. 0 | Page 4 of 72
AD6636
SPECIFICATIONS
Table 1. Recommended Operating Conditions
Parameter
VDDCORE
VDDIO
TAMBIENT
Temp
Full
Full
Full
Test Level
IV
IV
IV
Min
1.7
3.0
−40
Typ
1.8
3.3
+25
Max
1.9
3.6
+85
Unit
V
V
°C
Temp
Test Level
Min
Typ
Max
Unit
Full
Full
Full
Full
Full
25°C
IV
IV
IV
IV
IV
V
3.3
2.0
−0.3
3.6
+0.8
10
10
V CMOS
V
V
µA
µA
pF
Full
Full
Full
IV
IV
IV
3.3
2.0
25°C
25°C
25°C
25°C
25°C
V
V
V
V
V
450
50
mA
mA
400
25
mA
mA
25°C
25°C
V
V
250
15
mA
mA
25°C
25°C
V
V
175
10
mA
mA
25°C
25°C
25°C
25°C
V
V
V
V
975
800
500
350
mW
mW
mW
mW
ELECTRICAL CHARACTERISTICS
Table 2. Electrical Characteristics1
Parameter
LOGIC INPUTS (NOT 5 V TOLERANT)
Logic Compatibility
Logic 1 Voltage
Logic 0 Voltage
Logic 1 Current
Logic 0 Current
Input Capacitance
LOGIC OUTPUTS
Logic Compatibility
Logic 1 Voltage (IOH = 0.25 mA)
Logic 0 Voltage (IOL = 0.25 mA)
SUPPLY CURRENTS
WCDMA (61.44 MHz) Example1
IVDDCORE
IVDDIO
CDMA 2000 (61.44 MHz) Example1
IVDDCORE
IVDDIO
TDS-CDMA (76.8 MHz) Example1, 2
IVDDCORE
IVDDIO
GSM (65 MHz) Example1, 2
IVDDCORE
IVDDIO
TOTAL POWER DISSIPATION
WCDMA (61.44 MHz)1
CDMA 2000 (61.44 MHz)1
TDS-CDMA, (76.8 MHz)1, 2
GSM, (65 MHz)1, 2
1
2
One input port, all six channels, and the relevant signal processing blocks are active.
PLL is turned off for power savings.
Rev. 0 | Page 5 of 72
1
1
4
VDDIO − 0.2
0.2
0.4
V CMOS
V
V
AD6636
GENERAL TIMING CHARACTERISTICS
Table 3. General Timing Characteristics1, 2
Parameter
CLK TIMING REQUIREMENTS
tCLK
CLKx Period (x = A, B, C, D)
tCLKL
CLKx Width Low (x = A, B, C, D)
tCLKH
CLKx Width High (x = A, B, C, D)
tCLKSKEW
CLKA to CLKx Skew (x = B, C, D)
INPUT WIDEBAND DATA TIMING REQUIREMENTS
tSI
INx [15:0] to ↑CLKx Setup Time (x = A, B, C, D)
tHI
INx [15:0] to ↑CLKx Hold Time (x = A, B, C, D)
tSEXP
EXPx [2:0] to ↑CLKx Setup Time (x = A, B, C, D)
tHEXP
EXPx [2:0] to ↑CLKx Hold Time (x = A, B, C, D)
tDEXP
↑CLKx to EXPx[2:0] Delay (x = A, B, C, D)
PARALLEL OUTPUT PORT TIMING REQUIREMENTS (MASTER)
tDPREQ
↑PCLK to ↑Px REQ Delay (x = A, B, C)
tDPP
↑PCLK to Px [15:0] Delay (x = A, B, C)
tDPIQ
↑PCLK to Px IQ Delay (x = A, B, C)
tDPCH
↑PCLK to Px CH[2:0] Delay (x = A, B, C)
tDPGAIN
↑PCLK to Px Gain Delay (x = A, B, C)
tSPA
Px ACK to ↑PCLK Setup Time (x = A, B, C)
tHPA
Px ACK to ↑PCLK Hold Time (x = A, B, C)
PARALLEL OUTPUT PORT TIMING REQUIREMENTS (SLAVE)
tPCLK
PCLK Period
tPCLKL
PCLK Low Period
tPCLKH
PCLK High Period
tDPREQ
↑PCLK to ↑Px REQ Delay (x = A, B, C)
tDPP
↑PCLK to Px [15:0] Delay (x = A, B, C)
tDPIQ
↑PCLK to Px IQ Delay (x = A, B, C)
tDPCH
↑PCLK to Px CH[2:0] Delay (x = A, B, C)
tDPGAIN
↑PCLK to Px Gain Delay (x = A, B, C)
tSPA
Px ACK to ↓PCLK Setup Time (x = A, B, C)
tHPA
Px ACK to ↓PCLK Hold Time (x = A, B, C)
MISC PINS TIMING REQUIREMENTS
tRESET
RESET Width Low
tDIRP
CPUCLK/SCLK to IRP Delay
tSS
SYNC(0, 1, 2, 3) to ↑CLKA Setup Time
tHS
SYNC(0, 1, 2, 3) to ↑CLKA Hold Time
1
2
Temp
Test Level
Min
Typ
Full
Full
Full
Full
Full
Full
Full
Full
Full
Full
I
IV
IV
IV
IV
IV
IV
IV
IV
IV
6.66
1.71
1.70
tCLK − 1.3
0.5 × tCLK
0.5 × tCLK
Full
Full
Full
Full
Full
Full
Full
IV
IV
IV
IV
IV
IV
IV
1.77
2.07
0.48
0.38
0.23
4.59
0.90
Full
Full
Full
Full
Full
Full
Full
Full
Full
Full
IV
IV
IV
IV
IV
IV
IV
IV
IV
IV
5.0
1.7
0.7
4.72
4.8
4.83
4.88
5.08
6.09
1.0
Full
Full
Full
Full
IV
V
IV
IV
30
All timing specifications are valid over the VDDCORE range of 1.7 V to 1.9 V and the VDDIO range of 3.0 V to 3.6 V.
CLOAD = 40 pF on all outputs, unless otherwise noted.
Rev. 0 | Page 6 of 72
0.75
1.13
3.37
1.11
5.98
Unit
ns
ns
ns
ns
10.74
3.86
5.29
5.49
5.35
4.95
0.5 × tPCLK
0.5 × tPCLK
8.87
8.48
10.94
10.09
11.49
7.5
0.87
0.67
Max
ns
ns
ns
ns
ns
ns
ns
ns
ns
ns
ns
ns
ns
ns
ns
ns
ns
ns
ns
ns
ns
ns
ns
ns
ns
ns
AD6636
MICROPORT TIMING CHARACTERISTICS
Table 4. Microport Timing Characteristics1, 2
Parameter
MICROPORT CLOCK TIMING REQUIREMENTS
tCPUCLK
CPUCLK Period
tCPUCLKL
CPUCLK Low Time
tCPUCLKH CPUCLK High Time
INM MODE WRITE TIMING (MODE = 0)
tSC
Control3 to ↑CPUCLK Setup Time
tHC
Control3 to ↑CPUCLK Hold Time
tSAM
Address/Data to ↑CPUCLK Setup Time
tHAM
Address/Data to ↑CPUCLK Hold Time
tDRDY
↑CPUCLK to RDY (DTACK) Delay
tACC
Write Access Time
INM MODE READ TIMING (MODE = 0)
tSC
Control3 to ↑CPUCLK Setup Time
tHC
Control3 to ↑CPUCLK Hold Time
tSAM
Address to ↑CPUCLK Setup Time
tHAM
Address to ↑CPUCLK Hold Time
tDD
↑CPUCLK to Data Delay
tDRDY
↑CPUCLK to RDY (DTACK) Delay
tACC
Read Access Time
MNM MODE WRITE TIMING (MODE = 1)
tSC
Control3 to ↑CPUCLK Setup Time
tHC
Control3 to ↑CPUCLK Hold Time
tSAM
Address/Data to ↑CPUCLK Setup Time
tHAM
Address/Data to ↑CPUCLK Hold Time
tDDTACK
↑CPUCLK to DTACK (RDY) Delay
tACC
Write Access Time
MNM MODE READ TIMING (MODE = 1)
tSC
Control3 to ↑CPUCLK Setup Time
tHC
Control3 to ↑CPUCLK Hold Time
tSAM
Address to ↑CPUCLK Setup Time
tHAM
Address to ↑CPUCLK Hold Time
tDD
CPUCLK to Data Delay
tDDTACK
↑CPUCLK to DTACK (RDY) Delay
tACC
Read Access Time
1
2
3
Temp
Test Level
Min
Typ
Full
Full
Full
IV
IV
IV
10.0
1.53
1.70
0.5 × tCPUCLK
0.5 × tCPUCLK
Full
Full
Full
Full
Full
IV
IV
IV
IV
IV
0.80
0.09
0.76
0.20
3.51
6.72
ns
ns
ns
ns
ns
Full
IV
3 × tCPUCLK
9 × tCPUCLK
ns
Full
Full
Full
Full
Full
Full
IV
IV
IV
IV
V
IV
1.00
0.03
0.80
0.20
4.50
6.72
ns
ns
ns
ns
ns
ns
Full
IV
3 × tCPUCLK
9 × tCPUCLK
ns
Full
Full
Full
Full
Full
IV
IV
IV
IV
IV
1.00
0.00
0.00
0.57
4.10
5.72
ns
ns
ns
ns
ns
Full
IV
3 × tCPUCLK
9 × tCPUCLK
ns
Full
Full
Full
Full
Full
Full
IV
IV
IV
IV
V
IV
1.00
0.00
0.00
0.57
4.20
6.03
ns
ns
ns
ns
ns
ns
Full
IV
3 × tCPUCLK
9 × tCPUCLK
ns
Unit
ns
ns
ns
5.0
5.0
All timing specifications are valid over the VDDCORE range of 1.7 V to 1.9 V and the VDDIO range of 3.0 V to 3.6 V.
CLOAD = 40 pF on all outputs, unless otherwise noted.
Specification pertains to control signals: R/W (WR), DS (RD), and CS.
Rev. 0 | Page 7 of 72
Max
AD6636
SERIAL PORT TIMING CHARACTERISTICS
Table 5. Serial Port Timing Characteristics1, 2
Parameter
SERIAL PORT CLOCK TIMING REQUIREMENTS
tSCLK
SCLK Period
tSCLKL
SCLK Low Time
tSCLKH
SCLK High Time
SPI PORT CONTROL TIMING REQUIREMENTS (MODE = 0)
tSSI
SDI to ↓SCLK Setup Time
tHSI
SDI to ↓SCLK Hold Time
tSSCS
SCS to ↑SCLK Setup Time
tHSCS
SCS to ↑SCLK Hold Time
↑SCLK to SDO Delay Time
SPORT MODE CONTROL TIMING REQUIREMENTS (MODE = 1)
tSSI
SDI to ↓SCLK Setup Time
tHSI
SDI to ↓SCLK Hold Time
tSSRFS
SRFS to ↓SCLK Setup Time
tHSRFS
SRFS to ↓SCLK Hold Time
tSSTFS
STFS to ↑SCLK Setup Time
tHSTFS
STFS to ↑SCLK Hold Time
tSSCS
SCS to ↑SCLK Setup Time
tDSDO
1
2
Temp
Test Level
Min
Typ
Full
Full
Full
IV
IV
IV
10.0
1.60
1.60
0.5 × tSCLK
0.5 × tSCLK
Full
Full
Full
IV
IV
IV
1.30
0.40
4.12
Unit
ns
ns
ns
ns
ns
ns
Full
IV
−2.78
Full
IV
4.28
Full
Full
Full
Full
Full
Full
Full
IV
IV
IV
IV
IV
IV
IV
0.80
0.40
1.60
−0.13
1.60
−0.30
4.12
ns
ns
ns
ns
ns
ns
ns
ns
tHSCS
SCS to ↑SCLK Hold Time
Full
IV
−2.76
tDSDO
↑SCLK to SDO Delay Time
Full
IV
4.29
All timing specifications are valid over the VDDCORE range of 1.7 V to 1.9 V and the VDDIO range of 3.0 V to 3.6 V.
CLOAD = 40 pF on all outputs, unless otherwise noted.
EXPLANATION OF TEST LEVELS FOR SPECIFICATIONS
I
II
III
IV
V
VI
Max
100% production tested.
100% production tested at 25°C, and sample tested at specified temperatures.
Sample tested only.
Parameter guaranteed by design and analysis.
Parameter is typical value only.
100% production tested at 25°C, and sampled tested at temperature extremes.
Rev. 0 | Page 8 of 72
ns
7.96
7.95
ns
ns
AD6636
ABSOLUTE MAXIMUM RATINGS
Stresses above those listed under the Absolute Maximum
Ratings may cause permanent damage to the device. This is a
stress rating only; functional operation of the device at these or
any other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
Table 6.
Parameter
ELECTRICAL
VDDCORE Supply Voltage
(Core Supply)
VDDIO Supply Voltage
(Ring or IO Supply)
Input Voltage
Output Voltage
Load Capacitance
ENVIRONMENTAL
Operating Temperature
Range (Ambient)
Maximum Junction
Temperature under Bias
Storage Temperature Range
(Ambient)
Rating
2.2 V
4.0 V
−0.3 to +3.6 V (not 5 V tolerant)
−0.3 to VDDIO + 0.3 V
200 pF
THERMAL CHARACTERISTICS
256-ball CSP_BGA package:
θJA = 25.4°C /W, no airflow
−40°C to +85°C
θJA = 23.3°C /W, 0.5 m/s airflow
125°C
θJA = 22.6°C /W, 1.0 m/s airflow
−65°C to +150°C
θJA = 21.9°C /W, 2.0 m/s airflow
Thermal measurements made in the horizontal position on a
4-layer board with vias.
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on
the human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
Rev. 0 | Page 9 of 72
AD6636
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
A
GND
INC3
IND4
IND7
CLKD
CLKC
IND11
GND
VDDCORE
IND14
IND15
SYNC1
TDO
PBGAIN
PB11
GND
A
B
IND0
VDDIO
INC2
IND5
IND6
IND8
IND10
IND12
IND13
INC14
SYNC3
SYNC0
TRST
PBCH2
VDDIO
PB12
B
C
EXPA1
EXPD1
INC0
INC1
IND3
INC5
IND9
INC10
INC13
SYNC2
TMS
TCLK
PBCH0
PB8
PB15
PB10
C
D
EXPB0
EXPC2
EXPC1
EXPD0
IND2
INC4
INC7
INC9
INC12
TDI
PBCH1
PBIQ
PB14
PB9
PB13
PACH1
D
E
INA14
INA15
EXPA0
LVDS_RSET
GND
IND1
INC6
INC8
INC11
INC15
PBREQ
PBACK
PB4
PB5
PB1
PCLK
E
F
INA12
INA13
EXPB1
EXPC0
EXPD2
GND
VDDIO
VDDIO
VDDIO
VDDIO
GND
PB6
PB0
PB7
PAREQ
PA0
F
G
INA11
INB13
INB15
EXPB2
EXPA2
VDDCORE
GND
GND
GND
GND
VDDCORE
PB3
PAGAIN
PB2
PACH0
PA2
G
H VDDCORE
INA10
INB12
INB11
INB14
VDDCORE
GND
GND
GND
GND
VDDCORE
PACH2
PAIQ
PAACK
PA1
GND
H
J
GND
INA9
INB10
INB8
INB9
VDDCORE
GND
GND
GND
GND
VDDCORE
PA3
PA7
PA5
PA4
VDDCORE
J
K
CLKA
INA8
INA7
INB6
INB7
VDDCORE
GND
GND
GND
GND
VDDCORE
PA12
PA15
PA9
PA8
PA6
K
L
CLKB
INA6
INB4
INB1
INB3
GND
VDDIO
VDDIO
VDDIO
VDDIO
GND
PC3
PCACK
PCCH1
PA13
PA10
L
M
INA5
INB5
INB2
INB0
GND
DTACK
(RDY, SDO)
D13
D15
D5
A5
PC12
PC7
PC2
PC0
PCCH0
PA11
M
N
INA4
INA3
INA0
R/W (WR,
STFS)
CS (SCS)
CHIPID2
D12
D2
D1
A4
A0 (SDI)
PC15
PC5
PC1
PCCH2
PA14
N
P
INA2
INA1
RESET
DS (RD,
SRFS)
SMODE
CHIPID3
GND
D9
D4
A6
A2
PC11
PC10
PC4
PCIQ
PCGAIN
P
R
CPUCLK
(SCLK)
VDDIO
MSB_
FIRST
EXT_
FILTER
CHIPID1
D14
D10
D11
D6
D0
A3
A1
PC9
PC6
VDDIO
PCREQ
R
T
GND
IRP
MODE
CHIPID0
D7
D8
D3
VDDCORE
GND
GND
A7
PC14
PC13
PC8
GND
GND
T
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
= VDDCORE
= VDDIO
= GROUND
04998-0-002
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
Figure 2. CSP_BGA Pin Configuration
Table 7. Pin Names and Functions
Name
Type
POWER SUPPLY
VDDCORE
Power
VDDIO
Power
GND
Ground
INPUT (ADC) PORTS (CMOS/LVDS)
CLKA
Input
Pin No.
Function
See Table 8
See Table 8
See Table 8
1.8 V Digital Core Supply.
3.3 V Digital I/O Supply.
Digital Core and I/O Ground.
K1
CLKB
CLKC
CLKD
INA[0:15]
INB[0:15]
INC[0:15]
IND[0:15]
EXPA[0:2]
EXPB[0:2]
EXPC[0:2]
EXPD[0:2]
CLKA, CLKB
L1
A6
A5
See Table 8
See Table 8
See Table 8
See Table 8
E3, C1, G5
D1, F3, G4
F4, D3, D2
D4, C2, F5
K1, L1
Clock for Input Port A. Used to clock INA[15:0] and EXPA[2:0] data. Additionally,
this clock is used to drive internal circuitry and PLL clock multiplier.
Clock for Input Port B. Used to clock INB[15:0] and EXPB[2:0] data.
Clock for Input Port C. Used to clock INC[15:0] and EXPC[2:0] data.
Clock for Input Port D. Used to clock IND[15:0] and EXPD[2:0] data.
Input Port A (Parallel).
Input Port B (Parallel).
Input Port C (Parallel).
Input Port D (Parallel).
Exponent Bus Input Port A. Gain control output.
Exponent Bus Input Port B. Gain control output.
Exponent Bus Input Port C. Gain control output.
Exponent Bus Input Port D. Gain control output.
LVDS Differential Clock for LVDS_A Input Port (LVDS_CLKA+, LVDS_CLKA−).
Input
Input
Input
Input
Input
Input
Input
Bidirectional
Bidirectional
Bidirectional
Bidirectional
Input
Rev. 0 | Page 10 of 72
AD6636
Name
CLKC, CLKD
INA[0:15], INB[0:15]
Type
Input
LVDS Input
Pin No.
A6, A5
See Table 8
Function
LVDS Differential Clock for LVDS_C Input Port (LVDS_CLKC+, LVDS_CLKC−).
In LVDS input mode, INA[0 :15] and INB[0 :15] form a differential pair
LVDS_A+[0:15] (positive node) and LVDS_A–[0:15] (negative node), respectively.
In LVDS input mode, INC[0 :15] and IND[0 :15] form a differential pair
LVDS_C+[0:15] (positive node) and LVDS_C–[0:15] (negative node), respectively.
INC[0:15], IND[0:15]
LVDS Input
See Table 8
OUTPUT PORTS
PCLK
PA[0:15]
PACH[0:2]
PAIQ
PAGAIN
Bidirectional
Output
Output
Output
Output
E16
See Table 8
G15, D16, H12
H13
G13
PAACK
PAREQ
PB[0:15]
PBCH[0:2]
PBIQ
PBGAIN
Input
Output
Output
Output
Output
Output
H14
F15
See Table 8
C13, D11, B14
D12
A14
PBACK
PBREQ
PC[0:15]
PCCH[0:2]
PCIQ
PCGAIN
Input
Output
Output
Output
Output
Output
E12
E11
See Table 8
M15, L14, N15
P15
P16
PCACK
PCREQ
MISC PINS
RESET
IRP
SYNC[0:3]
Input
Output
L13
R16
Parallel Output Port Clock. Master mode output, Slave mode input.
Parallel Output Port A Data Bus.
Channel Indicator Output Port A.
Parallel Port A I/Q Data Indicator. Logic 1 indicates I data on data bus.
Parallel Port A Gain Word Output Indicator. Logic 1 indicates gain word on
data bus.
Parallel Port A Acknowledge (Active High).
Parallel Port A Request (Active High).
Parallel Output Port B Data Bus.
Channel Indicator Output Port B.
Parallel Port B I/Q Data Indicator. Logic 1 indicates I data on data bus.
Parallel Port B Gain Word Output Indicator. Logic 1 indicates gain word on
data bus.
Parallel Port B Acknowledge (Active High).
Parallel Port B Request (Active High).
Parallel Output Port C Data Bus.
Channel Indicator Output Port C.
Parallel Port C I/Q Data Indicator. Logic 1 indicates I data on data bus.
Parallel Port C Gain Word Output Indicator. Logic 1 indicates gain word on
data bus.
Parallel Port C Acknowledge (Active High).
Parallel Port C Request (Active High).
Input
Output
Input
LVDS_RSET
Input
EXT_FILTER
Input
MICROPORT CONTROL
D[0:15]
Bidirectional
A[0:7]
Input
Input
DS(RD)
P3
T2
B12, A12, C10,
B11
E4
R4
Master Reset (Active Low).
Interrupt Pin.
Synchronization Inputs. SYNC pins are independent of channels or input ports and
independent of each other.
LVDS Resistor Set Pin (Analog Pin). See Design Notes.
PLL Loop Filter (Analog Pin). See Design Notes.
See Table 8
See Table 8
P4
DTACK (RDY)1
Output
M6
R/W (WR)
Input
N4
MODE
Input
T3
CS
CPUCLK
CHIPID[0:3]
Input
Input
Input
N5
R1
T4, R5, N6, P6
Bidirectional Microport Data. This bus is three-stated when CS is high.
Microport Address Bus.
Active Low Data Strobe when MODE = 1.
Active Low Read Strobe when MODE = 0.
Active Low Data Acknowledge when MODE = 1.
Microport Status Pin when MODE = 0.
Read/Write Strobe when MODE = 1.
Active Low Write Strobe when MODE = 0.
Mode Select Pin.
When SMODE = 0: Logic 0 = Intel mode; Logic 1 = Motorola mode.
When SMODE = 1: Logic 0 = SPI mode; Logic 1 = SPORT mode.
Active Low Chip Select. Logic 1 three-states the microport data bus.
Microport CLK Input (Input Only).
Chip ID Input Pins.
Rev. 0 | Page 11 of 72
AD6636
Name
Type
SERIAL PORT CONTROL
SCLK
Input
SDO
Output
SDI2
Input
STFS
Input
SRFS
Input
Input
SCS
MSB_FIRST
Input
Pin No.
Function
R1
M6
N11
N4
P4
N5
R3
SMODE
Input
P5
Serial Clock.
Serial Port Data Output.
Serial Port Data Input.
Serial Transmit Frame Sync.
Serial Receive Frame Sync.
Serial Chip Select.
Select MSB First into SDI Pin and MSB First Out of SDO Pin. Logic 0 = MSB first;
Logic 1 = LSB first.
Serial Mode Select. Pull high when serial port is used and low when microport is
used.
JTAG
TRST1
TCLK2
TMS1
TDO
TDI1
Input
Input
Input
Output
Input
B13
C12
C11
A13
D10
1
2
Test Reset Pin. Pull low when JTAG is not used.
Test Clock.
Test Mode Select.
Test Data Output. Three-stated when JTAG is in reset.
Test Data Input.
Pin with a pull-up resistor of nominal 70 kΩ.
Pin with a pull-down resistor of nominal 70 kΩ.
PIN LISTING FOR POWER, GROUND, DATA AND ADDRESS BUSES
Table 8.
Name
VDDCORE
VDDIO
GND
INA[0:15]
INB[0:15]
INC[0:15]
IND[0:15]
PA[0:15]
PB[0:15]
PC[0:15]
D[0:15]
A[0:7]
Pin No.
A9, G6, G11, H1, H6, H11, J6, J11, J16, K6, K11, T8
B2, B15, F7, F8, F9, F10, L7, L8, L9, L10, R2, R15
A1, A8, A16, E5, F6, F11, G7, G8, G9, G10, H7, H8, H9, H10, H16, J1, J7, J8, J9, J10, K7, K8, K9, K10, L6, L11, M5,
P7, T1, T9, T10, T15, T16
N3, P2, P1, N2, N1, M1, L2, K3, K2, J2, H2, G1, F1, F2, E1, E2
M4, L4, M3, L5, L3, M2, K4, K5, J4, J5, J3, H4, H3, G2, H5, G3
C3, C4, B3, A2, D6, C6, E7, D7, E8, D8, C8, E9, D9, C9, B10, E10
B1, E6, D5, C5, A3, B4, B5, A4, B6, C7, B7, A7, B8, B9, A10, A11
F16, H15, G16, J12, J15, J14, K16, J13, K15, K14, L16, M16, K12, L15, N16, K13
F13, E15, G14, G12, E13, E14, F12, F14, C14, D14, C16, A15, B16, D15, D13, C15
M14, N14, M13, L12, P14, N13, R14, M12, T14, R13, P13, P12, M11, T13, T12, N12
R10, N9, N8, T7, P9, M9, R9, T5, T6, P8, R7, R8, N7, M7, R6, M8
N11, R12, P11, R11, N10, M10, P10, T11
Rev. 0 | Page 12 of 72
AD6636
TIMING DIAGRAMS
04998-0-003
RESET
tRESL
Figure 3. Reset Timing Requirements
tCLKH
04998-0-004
CLKx
tCLKL
Figure 4. CLK Switching Characteristics
(x = A, B, C, D for Individual Input Ports)
tCLK
tCLKL
CLKA
tCLKH
04998-0-005
tCLKSKEW
CLKx
Figure 5. CLK Skew Characteristics
(x = B, C, D for Individual Input Ports)
tCPUCLKH
04998-0-006
CPUCLK
tCPUCLKL
Figure 6. CPUCLK Switching Characteristics
04998-0-007
tSCLKH
SCLK
tSCLKL
Figure 7. SCLK Switching Characteristics
CLKA
tHSYNC
04998-0-008
tSSYNC
SYNC [3:0]
Figure 8. SYNC Timing Inputs
Rev. 0 | Page 13 of 72
AD6636
tCLK
tCLKL
CLKx
tCLKH
04998-0-009
tDEXP
EXPx[2:0]
Figure 9. Gain Control Word Output Switching Characteristics
(x = A, B, C, D for Individual Input Ports)
CLKx
tSI
tHI
tSEXP
tHEXP
04998-0-010
INx[15:0]
EXPx[15:0]
Figure 10. Input Port Timing for Data
(x = A, B, C, D for Individual Input Ports)
PCLK
tSPA
tHPA
PxACK
tDPREQ
PxREQ
Px [15:0]
PxIQ
PxCH [2:0]
I [15:0]
tDPP
tDPP
Q [15:0]
RSSI [11:0]
tDPP
I [15:0]
tDPIQ
tDPIQ
tDPCH
tDPCH
PxCH [2:0] = CHANNEL #
tDPP
Q [15:0]
RSSI [11:0]
PxCH [2:0] = CHANNEL #
tDPGAIN
PxGAIN
Figure 11. Master Mode PxACK to PCLK Switching Characteristics
(x = A, B, C, D for Individual Output Ports)
Rev. 0 | Page 14 of 72
tDPP
tDPGAIN
04998-0-011
tDPP
AD6636
PCLK
PxACK
tDPREQ
TIED LOGIC HIGH ALL THE TIME
PxREQ
tDPP
I [15:0]
tDPP
Q [15:0]
tDPP
RSSI [11:0]
I [15:0]
tDPIQ
PxIQ
Q [15:0]
tDPP
RSSI [11:0]
tDPIQ
tDPCH
PxCH [2:0]
tDPP
tDPCH
PxCH [2:0] = CHANNEL #
PxCH [2:0] = CHANNEL #
tDPGAIN
04998-0-012
Px [15:0]
tDPP
tDPGAIN
PxGAIN
Figure 12. Master Mode PxREQ to PCLK Switching Characteristics
CPUCLK
RD
tSC
tHC
tSC
tHC
WR
CS
tSAM
tHAM
A [7:0]
VALID ADDRESS
tSAM
tHAM
D [15:0]
VALID DATA
tDRDY
tACC
NOTE:
tACC ACCESS TIME DEPENDS ON THE ADDRESS ACCESSED. IT CAN VARY FROM 3 TO 9 CPUCLK CYCLES.
Figure 13. INM Microport Write Timing Requirements
Rev. 0 | Page 15 of 72
04998-0-013
RDY
AD6636
CPUCLK
tHC
tSC
RD
WR
tHC
tSC
CS
tSAM
A [7:0]
tHAM
VALID ADDRESS
tDD
D [15:0]
VALID DATA
tDRDY
tACC
NOTE:
tACC ACCESS TIME DEPENDS ON THE ADDRESS ACCESSED. IT CAN VARY FROM 3 TO 9 CPUCLK CYCLES.
04998-0-014
RDY
Figure 14. INM Microport Read Timing Requirements
CPUCLK
tSC
tHC
tSC
tHC
tSC
tHC
DS
R/W
CS
tHAM
tSAM
A [7:0]
VALID ADDRESS
tSAM
D [15:0]
tHAM
VALID DATA
tACC
NOTE:
tACC ACCESS TIME DEPENDS ON THE ADDRESS ACCESSED. IT CAN VARY FROM 3 TO 9 CPUCLK CYCLES.
Figure 15. MNM Microport Write Timing Requirements
Rev. 0 | Page 16 of 72
04998-0-015
tDDTACK
DTACK
AD6636
CPUCLK
tSC
tHC
tSC
tHC
DS
R/W
tSC
tHC
tSAM
tHAM
CS
A [7:0]
VALID ADDRESS
tDD
VALID
DATA
D [15:0]
tDDTACK
DTACK
04998-0-016
tACC
NOTE:
tACC ACCESS TIME DEPENDS ON THE ADDRESS ACCESSED. IT CAN VARY FROM 3 TO 9 CPUCLK CYCLES.
Figure 16 MNM Microport Read Timing Requirements
SCLK
tSSCS
tHSCS
SCS
SMODE
LOGIC 1
tHSI
tSSI
SDI
D0
tSSRFS
D1
D2
D3
D4
D5
D6
D7
tHSRFS
MODE
LOGIC 1
Figure 17. SPORT Mode Write Timing Characteristics
Rev. 0 | Page 17 of 72
04998-0-017
SRFS
AD6636
SCLK
tSSCS
tHSCS
SCS
SMODE
LOGIC 1
tDSDO
SDO
D0
D1
D2
D3
D4
D5
D6
D7
tHSTFS
tSSTFS
04998-0-018
STFS
MODE
LOGIC 1
Figure 18. SPORT Mode Read Timing Characteristics
SCLK
tSSCS
tHSCS
SCS
SMODE
LOGIC 1
tHSI
tSSI
D0
D1
D2
D3
D4
D5
D6
D7
04998-0-019
SDI
MODE
LOGIC 0
Figure 19. SPI Mode Write Timing Characteristics
SCLK
SCS
tSSCS
tHSCS
SMODE
LOGIC 0
tDSDO
D0
D1
D2
D3
D4
D5
LOGIC 0
MODE
Figure 20. SPI Mode Read Timing Characteristics
Rev. 0 | Page 18 of 72
D6
D7
04998-0-020
SDO
AD6636
THEORY OF OPERATION
These four input ports can operate at up to 150 MSPS. Each
input port has its own clock (CLKA, CLKB, CLKC, and CLKD)
used for registering input data into the AD6636. To allow slow
input rates while providing fast processing clock rates, the
AD6636 contains an internal PLL clock multiplier that supplies
the internal signal processing clock. CLKA is used as an input to
the PLL clock multiplier. Additional programmability allows the
input data to be clocked into the part either on the rising edge
or the falling edge of the input clock.
In addition, the front end of the AD6636 contains circuitry that
enables high speed signal-level detection, gain control, and
quadrature I/Q correction. This is accomplished with a unique
high speed level-detection circuit that offers minimal latency
and maximum flexibility to control all four input signals
(typically ADC inputs) individually. The input ports also
provide input power-monitoring functions via various modes,
and magnitude and phase I/Q correction blocks. See the
Quadrature I/Q Correction Block section for details.
Each individual processing channel can receive input data from
any of the four input ports individually. This is controlled using
3-bit crossbar mux-select bit words in ADC input control
register. Each individual channel has a similar 3-bit selection. In
addition to the four input ports, an internal test signal (PN—
pseudorandom noise sequence) can also be selected. This
internal test signal is discussed in the User-Configurable BuiltIn Self-Test (BIST) section.
Input Data Format
Each input port consists of a 16-bit mantissa and a 3-bit
exponent (16 + 3 floating-point input, or up to 16-bit fixedpoint input). When interfacing to standard fixed-point ADCs,
the exponent bit should either be connected to ground or be
programmed as outputs for gain control output. If connected to
a floating-point ADC (also called gain ranging ADC), the
exponent bits from the ADC can be connected to the input
exponent bits of the AD6636. The mantissa data format is twos
complement, and the exponent is unsigned binary.
The 3-exponent bits are shared with the gain range control bits
in the hardware. When floating-point ADCs are not used, these
three pins on each ADC input port can be used as gain range
control output bits.
Input Timing
The data from each high speed input port is latched either on
the rising edge or the falling edge of the port’s individual CLKx
(where x stands for A, B, C, or D input ports). The ADC clock
invert bit in ADC clock control register selects the edge of the
clock (rising or falling) used to register input data into the
AD6636.
CLKx
tSI
INx [15:0]
EXPx [2:0]
tHI
DATA n
DATA n + 1
04998-0-021
The AD6636 features four identical, independent high speed
ADC input ports named A, B, C, and D. These input ports have
the flexibility to allow independent inputs, diversity inputs, or
complex I/Q inputs. Any of the ADC input ports can be routed
to any of the six tuner channels; that is, any of the six AD6636
channels can receive input data from any of the input ports.
Time-multiplexed inputs on a single port are not supported in
the AD6636.
Figure 21. Input Data Timing Requirements
(Rising Edge of Clock, x = A, B, C, or D for Four Input Ports)
CLKx
tSI
INx [15:0]
EXPx [2:0]
tHI
DATA n
DATA n + 1
04998-0-022
ADC INPUT PORT
Figure 22. Input Data Timing Requirements
(Falling Edge of Clock, x = A, B, C, or D for Four Input Ports)
The clock signals (CLKA, CLKB, CLKC, and CLKD) can
operate at up to 150 MHz. In applications using high speed
ADCs, the ADC sample clock, data valid, or data ready strobe
are typically used to clock the AD6636.
Connection to Fixed-Point ADC
For fixed-point ADCs, the AD6636 exponent inputs, EXP[2:0],
are not typically used and should be tied low. Alternatively,
because these pins are shared with gain range control bits, if the
gain ranging block is used, these pins can be used as outputs of
the gain range control block. The ADC outputs are tied directly
to the AD6636 inputs, MSB-justified. Therefore, for fixed-point
ADCs, the exponents are typically static and no input scaling is
used in the AD6636. Figure 23 shows a typical interconnection.
Rev. 0 | Page 19 of 72
AD6636
D13 (MSB)
Input Ports A and B, then the mux select bits should indicate
Input Port A, and the complex input bit should be selected.
IN15
AD6645
When the input ports are paired for complex input operation,
only one set of exponent bits is driven externally with gain
control output. So when Input Ports A and B form a complex
input, then EXPA[2:0] are output and, similarly, for Input Ports
C and D, EXPC[2:0] are output.
14-BIT ADC
AD6636
IN2
IN1
IN0
GAIN RANGING CONTROL
BITS OR GROUNDED
EXPONENT BITS
EXP2
EXP1
EXP0
LVDS Input Ports
AD6636 input ports can be configured in two different modes:
CMOS or LVDS. In CMOS input mode, the four input ports can
be configured as two complex input ports. In LVDS mode, two
CMOS input ports each are combined to form one LVDS input
port.
04998-0-023
D0 (LSB)
Figure 23. Typical Interconnection of the AD6645 Fixed-Point ADC
and the AD6636
Scaling with Floating-Point ADC
Table 9. Weighting Factors for Different Exp[2:0] Values
ADC Input
Level
Largest
Smallest
AD6636
Exp[2:0]
000 (0)
001 (1)
010 (2)
011 (3)
100 (4)
101 (5)
110 (6)
111 (7)
Data
Divide-By
/1 (>> 0)
/2 (>>1)
/4 (>>2)
/8 (>>3)
/16 (>> 4)
/32 (>> 5)
/64 (>> 6)
/128(>> 7)
Signal
Attenuation (dB)
0
6
12
18
24
30
36
42
Complex (I/Q) Input Ports
The four individual ADC input ports of the AD6636 can be
configured to function as two complex input ports. Additionally,
if required, only two input ports can be made to function as a
complex port, while the remaining two input ports function as
real individual input ports.
In complex mode, Input Port A is paired with Input Port B to
receive I and Q data, respectively. Similarly, Input Port C can be
paired with Input Port D to receive I and Q data, respectively.
These two pairings are controlled individually using Bits 24 and
25 of ADC input control register.
CMOS Input Ports INA[15:0] and INB[15:0] form the positive
and negative differential nodes, LVDS_A+[15:0] and
LVDS_A−[15:0], respectively. Similarly, INC[15:0] and
IND[15:0] form the positive and negative differential nodes,
LVDS_C+[15:0] and LVDS_C− [15:0], respectively. CLKA and
CLKB form the differential pair, LVDS_CLKA+ and
LVDS_CLKA− pins. Similarly, CLKC and CLKD form the
differential pair LVDS_CLKC+ and LVDS_CLKC− pins.
By default, the AD6636 powers up in CMOS mode and can be
programmed to CMOS mode by using the CMOS mode bit (Bit
10 of the LVDS control register). Writing Logic 1 to Bit 8 of the
LVDS control register enables an autocalibrate routine that
calibrates the impedance of the LVDS pads to match the output
impedance of the LVDS signal source impedance. The LVDS
pads in the AD6636 have an internal impedance of 100 Ω across
the differential signals; therefore, an external resistor is not
required.
PLL CLOCK MULTIPLIER
In the AD6636, the input clock rate must be the same as the
input data rate. In a typical digital down-converter architecture,
the clock rate is a limitation on the number of filter taps that
can be calculated in the programmable RAM coefficient filters
(MRCF, DRCF, and CRCF). For slower ADC clock rates (or for
any clock rate), this limitation can be overcome by using a PLL
clock multiplier to provide a higher clock rate to the RCF filters.
Using this clock multiplier, the internal signal processing clock
rate can be increased up to 200 MHz. The CLKA signal is used
as an input to the PLL clock multiplier.
As explained previously, each individual channel can receive
input signals from any of the four input ports using the crossbar
mux select bits in the ADC input control register. In addition to
the three bits, a 1-bit selection is provided for choosing the
complex input port option for any individual channel. For
example, if Channel 0 needs to receive complex input from
Rev. 0 | Page 20 of 72
PLL CLOCK GENERATION
1
ADC_CLK
0
CLKA
DIVIDE BY N
(1, 2, 4 OR 8)
PLL CLOCK
MULITPLIER
(4x TO 20x)
0
PLL_CLK
1
BYPASS_PLL
2
N
5 1 FOR BYPASS
M
Figure 24. PLL Clock Generation
04998-0-024
An example of the exponent control feature combines the
AD6600 and the AD6636. The AD6600 is an 11-bit ADC with
three bits of gain ranging. In effect, the 11-bit ADC provides the
mantissa, and the three bits of the relative signal strength
indicator (RSSI) are the exponent. Only five of the eight
available steps are used by the AD6600. See the AD6600 data
sheet for details.
AD6636
The PLL clock multiplier is programmable and uses input clock
rates between 4 MHz and 150 MHz to give a system clock rate
(output) of as high as 200 MHz.
The output clock rate is given by
PLL _ CLK =
CLKA × M
N
where:
CLKA is the Input Port A clock rate.
M is a 5-bit programmable multiplication factor.
N is a predivide factor.
M is a 5-bit number between 4 and 20 (both values included). N
(predivide) can be 1, 2, 4, or 8. The multiplication factor M is
programmed using a 5-bit PLL clock multiplier word in the
ADC clock control register. A value outside the valid range of 4
to 20 bypasses the PLL clock multiplier and, therefore, the PLL
clock is the same as the input clock. The predivide factor N is
programmed using a 2-bit ADC pre-PLL clock divider word in
the ADC clock control register, as listed in Table 10.
Table 10. PLL Clock Generation Predivider Control
Predivide Word [1:0]
00
01
10
11
Divide-by Value for the Clock
Divide-by-1, bypass
Divide-by-2
Divide-by-4
Divide-by-8
For best signal processing advantage, the user should program
the clock multiplier to give a system clock output as close as
possible to, but not exceeding, 200 MHz. The internal blocks of
the AD6636 that run off of the PLL clock are rated to run at a
maximum of 200 MHz. The default power-up state for the PLL
clock multiplier is the bypass state, where CLKA is passed on as
the PLL clock.
Function
The gain-control block features a programmable upper
threshold register and a lower threshold register. The ADC
input data is compared to both these registers. If ADC input
data is larger than the upper threshold register, then the gain
control output is decremented by 1. If ADC input data is smaller
than the lower threshold register, then the gain control output is
incremented by 1. When decrementing the gain control output,
the change is immediate. But when incrementing the output, a
dwell-time register is used to delay the change. If the ADC input
is larger than the upper threshold register value, the gaincontrol output is decremented immediately to prevent overflow.
When the ADC input is lower than the lower threshold register,
a dwell timer is loaded with the value in the programmable
20-bit dwell-time register. The counter decrements once every
input clock cycle, as long as the input signal remains below the
lower threshold register value. If the counter reaches 1, the gain
control output is incremented by 1. If the signal goes above the
lower threshold register value, the gain adjustment is not made,
and the normal comparison to lower and upper threshold
registers is initiated once again. Therefore, the dwell timer
provides temporal hysteresis and prevents the gain from
switching continuously.
In a typical application, if the ADC signal goes below the lower
threshold for a time greater than the dwell time, then the gain
control output is incremented by 1. Gain control bits control the
gain ranging block, which appears before the ADC in the signal
chain. With each increment of the gain control output, gain in
the gain-ranging block is increased by 6.02 dB. This increases
the dynamic range of the input signal into the ADC by 6.02 dB.
This gain is compensated for in the AD6636 by relinearizing, as
explained in the Relinearization section. Therefore, the AD6636
can increase the dynamic range of the ADC by 42 dB, provided
that the gain-ranging block can support it.
Relinearization
Each ADC input port has individual, high speed gain-control
logic circuitry. Such gain-control circuitry is useful in applications that involve large dynamic-range inputs or in which
gain-ranging ADCs are employed. The AD6636 gain-control
logic allows programmable upper and lower thresholds and a
programmable dwell-time counter for temporal hysteresis.
The gain in the gain-ranging block (external) is compensated
for by relinearizing, using the exponent bits EXP[2:0] of the
input port. For this purpose, the gain control bits are connected
to the EXP[2:0] bits, providing an attenuation of 6.02 dB for
every increase in the gain control output. After the gain in the
external gain-ranging block and the attenuation in the AD6636
(using EXP bits), the signal gain is essentially unchanged. The
only change is the increase in the dynamic range of the ADC.
Each input port has a 3-bit output from the gain control block.
These three output pins are shared with the 3-bit exponent
input pins for each input port. The operation is controlled by
the gain control enable bit in gain control register of the
individual input ports. A Logic 1 in this bit programs the
EXP[2:0] pins as gain-control outputs, and a Logic 0 configures
the pins as input exponent pins. To avoid bus contention, these
pins are set, by default, as input exponent pins.
External gain-ranging blocks or gain-ranging ADCs have a
delay associated with changing the gain of the signal. Typically,
these delays can be up to 14 clock cycles. The gain change in the
AD6636 (via EXP[2:0]) must be synchronized with the gain
change in the gain-ranging block (external). This is allowed in
the AD6636 by providing a flexible delay, programmable 6-bit
word in the gain control register. The value in this 6-bit word
gives the delay in input clock cycles. A programmable pipeline
ADC GAIN CONTROL
Rev. 0 | Page 21 of 72
AD6636
delay given by the 6-bit value (maximum delay of 63 clock
cycles) is placed between the gain control output and the
EXP[2:0] input. Therefore, the external gain-ranging block’s
settling delays are compensated for in the AD6636.
ADC INPUT PORT MONITOR FUNCTION
Note that any gain changes that are initiated during the
relinearization period are ignored. For example, if the AD6636
detects that a gain adjustment is required during the relinearization period of a previous gain adjustment, then the new
adjustment is ignored.
The AD6636 provides a power-monitor function that can
monitor and gather statistics about the received signal in a
signal chain. Each input port is equipped with an individual
power-monitor function that can operate both in real and in
complex modes of the input port. This function block can
operate in one of three modes, which measure the following
over a programmable period of time:
•
Peak power
To set up the gain control block for individual input ports, the
individual upper threshold registers and lower threshold
registers should be written with appropriate values. The 10-bit
values written into upper and lower threshold registers are
compared to the 10 MSB bits of the absolute magnitude
calculated using the input port data. The 20-bit dwell timer
register should have the appropriate number of clock cycles to
provide temporal hysteresis.
•
Mean power
•
Number of samples crossing a threshold
A 6-bit relinearization pipeline delay word is set to synchronize
with the settling delay in the external gain ranging circuitry.
Finally, the gain control enable bit is written with Logic 1 to
activate the gain control block. On enabling, the gain control
output bits are made 000 (output on EXP[2:0] pins), which
represent the minimum gain for the external gain-ranging
circuitry and corresponding minimum attenuation during
relinearization. The normal functioning takes over, as explained
previously in this section.
The three modes of operation can function continuously over a
programmable time period. This time period is programmed as
the number of input clock cycles in a 24-bit ADC monitor
period register (AMPR). This register is separate for each input
port. An internal magnitude storage register (MSR) is used to
monitor, accumulate, or count, depending on the mode of
operation.
Setting Up the Gain Control Block
Complex Inputs
For complex inputs (formed by pairing two input ports), only
one set of EXP[2:0] pins should be used as the gain control
output. For the pair of Input Ports A and B, gain control
circuitry for Input Port A is active, and EXPA[2:0] should be
connected externally as the gain control output. The gain
control circuitry for Input Port B is not activated (shut down),
and EXPB[2:0] is forced to be equal to EXP[2:0].
LOWER
THRESHOLD
REGISTER
B
A
FROM INPUT
PORTS
COMPARE
A>B
A
INC
EXP [2:0]
INCREASE
EXTERNAL GAIN
B
FROM
MEMORY
MAP
DECREASE
EXTERNAL GAIN
COMPARE
A<B
DWELL
TIMER
EXP GEN
DEC
LOWER
THRESHOLD
REGISTER
Figure 25. AD6636 Gain Control Block Diagram
04998-0-025
FROM
MEMORY
MAP
These functions are controlled via the 2-bit power-monitor
function select bits of the power monitor control register for
each individual input port. The input ports can be set for
different modes, but only one function can be active at a time
for any given input port.
Peak Detector Mode (Control Bits 00)
The magnitude of the input port signal is monitored over a
programmable time period (given by AMPR) to give the peak
value detected. This mode is set by programming Logic 0 in the
power-monitor function select bits of the power-monitor
control register for each individual input port. The 24-bit
AMPR must be programmed before activating this mode.
After enabling this mode, the value in the AMPR is loaded into
a monitor period timer and the countdown is started. The
magnitude of the input signal is compared to the MSR, and the
greater of the two is updated back into the MSR. The initial
value of the MSR is set to the current ADC input signal
magnitude. This comparison continues until the monitor period
timer reaches a count of 1.
When the monitor period timer reaches a count of 1, the value
in the MSR is transferred to the power-monitor holding register,
which can be read through the microport or the serial port. The
monitor period timer is reloaded with the value in the AMPR,
and the countdown is started. Also, the first input sample’s
magnitude is updated in the MSR, and the comparison and
update procedure, as explained above, continues. If the interrupt
is enabled, an interrupt is generated, and the interrupt status
register is updated when the AMPR reaches a count of 1.
Rev. 0 | Page 22 of 72
AD6636
Figure 26 is a block diagram of the peak detector logic. The
MSR contains the absolute magnitude of the peak detected by
the peak detector logic.
FROM
MEMORY
MAP
POWER MONITOR
PERIOD REGISTER
TO
INTERRUPT
CONTROLLER
DOWN
COUNTER
IS COUNT = 1?
TO
INTERRUPT
CONTROLLER
IS COUNT = 1?
CLEAR
LOAD
POWER MONITOR
HOLDING
REGISTER
ACCUMULATOR
TO
MEMORY
MAP
LOAD
CLEAR
MAGNITUDE
STORAGE
REGISTER
LOAD
POWER MONITOR
HOLDING
REGISTER
TO
MEMORY
MAP
Figure 27. ADC Input Mean Power-Monitoring Block Diagram
Threshold Crossing Mode (Control Bits 10)
LOAD
COMPARE
A>B
04998-0-026
FROM
INPUT
PORTS
Figure 26. ADC Input Peak Detector Block Diagram
Mean Power Mode (Control Bits 01)
In this mode, the magnitude of the input port signal is
integrated (by adding an accumulator) over a programmable
time period (given by AMPR) to give the integrated magnitude
of the input signal. This mode is set by programming Logic 1 in
the power monitor function select bits of the power monitor
control register for each individual input port. The 24-bit
AMPR, representing the period over which integration is
performed, must be programmed before activating this mode.
After enabling this mode, the value in the AMPR is loaded into
a monitor period timer, and the countdown is started
immediately. The 15-bit magnitude of input signal is rightshifted by nine bits to give 6-bit data. This 6-bit data is added to
the contents of a 24-bit holding register, thus performing an
accumulation. The integration continues until the monitor
period timer reaches a count of 1.
When the monitor period timer reaches a count of 1, the value
in the MSR is transferred to the power-monitor holding register
(after some formatting), which can be read through the
microport or the serial port. The monitor period timer is
reloaded with the value in the AMPR, and the countdown is
started. Also, the first input sample signal magnitude is updated
in the MSR, and the accumulation continues with the
subsequent input samples. If the interrupt is enabled, an
interrupt is generated, and the interrupt status register is
updated when the AMPR reaches a count of 1. Figure 27
illustrates the mean power-monitoring logic.
The value in the MSR is a floating-point number with 4 MSBs
and 20 LSBs. If the 4 MSBs are EXP and the 20 LSBs are MAG,
the value in dBFS can be decoded using the following equation:
⎤
⎡⎛ MAG ⎞
Mean Power = 10 log ⎢⎜ 20 ⎟ 2 −( EXP −1) ⎥
2
⎠
⎦
⎣⎝
In this mode of operation, the magnitude of the input port
signal is monitored over a programmable time period (given by
AMPR) to count the number of times it crosses a certain
programmable threshold value. This mode is set by programming Logic 1x (where x is a don’t care bit) in the power-monitor
function select bits of the power monitor control register for
each individual input port. Before activating this mode, the user
needs to program the 24-bit AMPR and the 10-bit upper
threshold register for each individual input port. The same
upper threshold register is used for both power monitoring and
gain control (see the ADC Gain Control section).
After entering this mode, the value in the AMPR is loaded into
a monitor period timer, and the countdown is started. The
magnitude of the input signal is compared to upper threshold
register (programmed previously) on each input clock cycle. If
the input signal has magnitude greater than the upper threshold
register, then the MSR register is incremented by 1. The initial
value of the MSR is set to zero. This comparison and increment
of the MSR register continues until the monitor period timer
reaches a count of 1.
When the monitor period timer reaches a count of 1, the value
in the MSR is transferred to the power monitor holding register,
which can be read through the microport or the serial port. The
monitor period timer is reloaded with the value in the AMPR,
and the countdown is started. The MSR register is also cleared
to a value of zero. If interrupts are enabled, an interrupt is
generated, and the interrupt status register is updated when the
AMPR reaches a count of 1. Figure 28 illustrates the threshold
crossing logic. The value in the MSR is the number of samples
that have an amplitude greater than the threshold register.
FROM
MEMORY
MAP
POWER MONITOR
PERIOD REGISTER
DOWN
COUNTER
TO
INTERRUPT
CONTROLLER
IS COUNT = 1?
LOAD
FROM
INPUT
PORTS
FROM
MEMORY
MAP
CLEAR
A COMPARE
A>B
COMPARE
A>B
LOAD
POWER MONITOR
HOLDING
REGISTER
TO
MEMORY
MAP
B
UPPER
THRESHOLD
REGISTER
Figure 28. ADC Input Threshold Crossing Block Diagram
Rev. 0 | Page 23 of 72
04998-0-028
DOWN
COUNTER
FROM
INPUT
PORTS
04998-0-027
LOAD
FROM
MEMORY
MAP
POWER MONITOR
PERIOD REGISTER
AD6636
Additional Control Bits
For additional flexibility in the power monitoring process, two
control bits are provided in the power-monitor control register.
The two control bits are the disable monitor period timer bit
and the clear-on-read bit. These options have the same function
in all three modes of operation.
If the clear-on-read bit is Logic 0, the read operation to the
microport or serial port does not clear the MSR value after it is
transferred into the holding register. The value from the
previous monitor time period persists, and it continues to be
compared, accumulated, or incremented, based on new input
signal magnitude values.
Disable Monitor Period Timer Bit
QUADRATURE I/Q CORRECTION BLOCK
When the disable monitor period timer bit is written with
Logic 1, the timer continues to run but does not cause the
contents of the MSR to be transferred to the holding register
when the count reaches 1. This function of transferring the
MSR to the power monitor holding register and resetting the
MSR is now controlled by a read operation on the microport or
serial port.
When the I and Q paths are digitized using separate ADCs, as in
quadrature IF down-conversion, a mismatch often occurs
between I and Q due to variations in the ADCs from the
manufacturing process. The AD6636 is equipped with two
quadrature correction blocks that can be used to correct I/Q
mismatch errors in a complex baseband input stream. These I/Q
mismatches can result in spectral distortions, and removing
them is useful.
Clear-on-Read Bit
This control bit is valid only when the disable monitor period
timer bit is Logic 1. When both of these bits are set, a read
operation to either the microport or the serial port reads the
MSR value and the monitor period timer is reloaded with the
AMPR value. The MSR is cleared (written with current input
signal magnitude in peak power and mean power mode; written
with a zero in threshold crossing mode), and normal operation
continues.
When the monitor period timer is disabled and the clear-onread bit is set, a read operation to the power monitor holding
register clears the contents of the MSR and, therefore, the power
monitor loop restarts.
Two such blocks are present, one each for the I/Q signal formed
by combining the A and B inputs and the C and D inputs,
respectively. The I/Q correction block can be enabled when the
Port A (or Port C) complex data active bit is enabled in the
ADC input control register. This block is bypassed when real
input data is present on the ADC input ports, because there is
no possibility of I/Q mismatch in real data.
The I/Q or quadrature correction block consists of three
independent subblocks: dc correction, phase correction, and
amplitude correction. Three individual bits in the AB (or CD)
correction control registers can be used to enable or disable
each of these subblocks independently. Figure 29 shows the
contents and definitions of the registers related to the
quadrature correction block.
DC
ESTIMATE
I [15:0] FROM
INPUT PORT
I_OUT [15:0]
TO NEXT BLOCK
PHASE ESTIMATE
[13:0]
MAGNITUDE
MAGNITUDE
ERROR
ESTIMATE [13:0] ESTIMATION
Q [15:0] FROM
INPUT PORT
PHASE
ERROR
ESTIMATION
Q_OUT [15:0]
TO NEXT BLOCK
DC
ESTIMATE
Figure 29. Quadrature Correction Block Diagram
Rev. 0 | Page 24 of 72
PHASE
ESTIMATE
[13:0]
04998-0-029
When a microport or serial port read is performed on the
power monitor holding register, the MSR value is transferred to
the holding register. After the read operation, the timer is
reloaded with the AMPR value. If the timer reaches 1 before the
microport or serial port read, the MSR value is not transferred
to the holding register, as in normal operation. The timer still
generates an interrupt on the AD6636 interrupt pin and updates
the interrupt status register. An interrupt appears on the IRP
pin, if interrupts are enabled in the interrupt enable register.
AD6636
Phase Correction
Table 11. Correction Control Registers
Register
I/Q Correction Control
DC Offset Correction I
DC Offset Correction Q
Amplitude Offset
Correction
Phase Offset Correction
Bits
15–12
11–8
7–4
3
2
1
0
31–16
15–0
31–16
Decription
Amplitude Loop BW
Phase Loop BW
DC Loop BW
Reserved (Logic 0)
Amplitude Correction
Enable
Phase Correction Enable
DC Correction Enable
DC Offset Q
DC Offset I
Amplitude Correction
15–0
Phase Correction
DC Correction
All ADCs have a nominal dc offset related to them. If the ADCs
in the I and Q path have different dc offsets due to variations in
manufacturing process, the dc correction circuit can be used to
compensate for these dc offsets. Writing Logic 1 into the dc
correction enable bit of the AB (or CD) correction control
register enables the dc correction block. Two dc estimation
blocks are used, one each for the I and Q paths. The estimated
dc value is subtracted from the I and Q paths. Therefore, the dc
signal is removed independently from the I and Q path signals.
A cascade of two low-pass decimating filters estimates the dc
offset in the feedback loop. A decimating first-order CIC filter is
followed by an interpolating second-order CIC filter. The
decimation and interpolation values of the CIC filters are the
same and are programmable between 212 and 224 in powers of 2.
The 4-bit dc loop BW word in the I/Q correction control AB (or
CD) register is used to program this decimation (interpolation)
value. When the dc loop BW is a 0, decimation is 212, and when
the dc loop BW is 11, decimation is 224.
When the dc correction circuit is enabled, the dc correction
values are estimated. The values, which are estimated independently in the I and Q paths, are subtracted independently from
their respective datapaths. These dc correction values are also
available for output continuously through the dc correction I
and dc correction Q registers. These registers contain register
16-bit dc offset values whose MSB-justified values are
subtracted directly from MSB-justified ADC inputs for the I
and Q paths.
When the dc correction circuit is disabled, the value in the dc
correction register is used for continuously subtracting the dc
offset from I and Q datapaths. This method can be used to
manually set the dc offset instead of using the automatic dc
correction circuit.
When using complex ADC input, the I and Q datapaths
typically have phase offset, caused mainly by the local oscillator
and demodulator IC. The AD6636 phase-offset correction
circuit can be used to compensate for this phase offset.
When the phase correction enable bit is Logic 1, the phase error
between I and Q is estimated (ideally, the phase should be 90°).
The phase mismatch is estimated over a period of time
determined by the integrator loop bandwidth. This integrator is
implemented as a first-order CIC decimating filter, whose
decimation value can vary between 212 and 224 in powers of 2.
Phase loop BW (Bits [11:8]) of the I/Q correction control
register determine this decimation value. When phase loop BW
equals 0, the decimation value is 212, and when phase loop BW is
11, the decimation value is 224.
While the phase offset correction circuit is enabled, the
tan(phase_mismatch) is estimated continuously. This value is
multiplied with Q path data and added to I path data
continuously. The estimated value is also updated in the phase
offset correction register. The tan(phase_mismatch) can be
±0.125 with a 14-bit resolution. This converts to a phase
mismatch of about ±7.125°.
When the phase offset correction circuit is disabled, the value in
the phase correction register multiplied with the Q path data
and added to the I path data continuously. This method can be
used to manually set the phase offset instead of using the
automatic phase offset correction circuit.
Amplitude Correction
When using complex ADC input, the I and Q datapaths
typically have amplitude offset, caused mainly by the local
oscillator and the demodulator IC. The AD6636 amplitude
offset correction circuit can be used to compensate for this
amplitude offset.
When the amplitude correction enable bit is Logic 1, the
amplitude error between the I and Q datapaths is estimated. The
amplitude mismatch is estimated over a period of time
determined by the integrator loop bandwidth. This integrator is
implemented as a first-order CIC decimating filter, whose
decimation value can vary between 212 and 224 in powers of 2.
Phase loop BW (Bits [11:8]) of the I/Q correction control
register determines this decimation value. When the phase loop
BW equals 0, the decimation value is 212, and when phase loop
BW is 11, the decimation value is 224.
While the amplitude offset correction circuit is enabled, the
difference (MAG(Q) – MAG(I)) is estimated continuously. This
value is multiplied with the Q path data and added to the Q
path data continuously. The estimated value is also updated in
the phase offset correction register. The difference (MAG(Q) –
MAG(I)) can be between 1.125 and 0.875 with a 14-bit
resolution.
Rev. 0 | Page 25 of 72
AD6636
When amplitude offset correction circuit is disabled, the value
in the amplitude offset correction register multiplied with the
Q path data and added to Q path data continuously. This
method can be used to manually set the amplitude offset
instead of using the automatic amplitude offset correction
circuit.
INPUT CROSSBAR MATRIX
The AD6636 has four ADC input ports and six channels. Two
input ports can be paired to support complex input ports.
Crossbar mux selection allows each channel to select its input
signal from the following sources: four real input ports, two
complex input ports, and internally generated pseudorandom
sequence (referred to as a PN sequence, which can be either real
or complex). Each channel has an input crossbar matrix to
select from the above-listed input signal choices.
The amplitude of the sine and cosine are represented using 17
bits. The worst-case spurious signal from the NCO is better
than −100 dBc for all output frequencies.
Because all the filtering in the AD6636 is low-pass filtering, the
carrier of interest is tuned down to dc (frequency = 0 Hz). This
is illustrated in Figure 30. Once the signal of interest is tuned
down to dc, the unwanted adjacent carriers can be rejected
using the low-pass filtering that follows.
NCO Frequency
The NCO frequency value is given by the 32-bit twos
complement number entered in the NCO frequency register.
Frequencies between −CLK/2 and CLK/2 (CLK/2 excluded)
are represented using this frequency word:
0x8000 0000 represents a frequency given by −CLK/2.
0x0000 0000 represents dc (frequency is 0 Hz).
The selection of the input signal for a particular channel is
made using a 3-bit crossbar mux select word and a 1-bit
complex data input bit selection in the ADC input control
register. Each channel has a separate selection for individual
control. Table 12 lists the valid combinations of the crossbar
mux select word, the complex data input bit values, and the
corresponding input signal selections.
0x7FFF FFFF represents CLK/2 − CLK/232.
The NCO frequency word can be calculated using following the
equation:
NCO _ FREQ = 2 32
NUMERICALLY CONTROLLED OSCILLATOR (NCO)
Each channel consists of an independent complex NCO and a
complex mixer. This processing stage comprises a digital tuner
consisting of three multipliers and a 32-bit complex NCO. The
NCO serves as a quadrature local oscillator capable of producing an NCO frequency of between −CLK/2 and +CLK/2 with a
resolution of CLK/232 in complex mode, where CLK is the input
clock frequency.
The frequency word used for generating the NCO is a 32-bit
word. This word is used to generate a 20-bit phase word. A
16-bit phase offset word is added to this phase word. 18 bits of
this phase word are used to generate the sine and cosine of the
required NCO frequency.
mod( f ch , f clk )
f clk
where:
NCO_FREQ is the 32-bit twos complement number representing the NCO frequency register.
fch is the desired carrier frequency.
fclk is the clock rate for the channel under consideration.
mod( ) is a remainder function. For example, mod(110, 100) =
10 and, for negative numbers, mod(−32, 10) = −2.
Note that this equation applies to the aliasing of signals in the
digital domain (that is, aliasing introduced when digitizing
analog signals).
Table 12. Crossbar Mux Selection for Channel Input Signal
Complex Input Bit
0
0
0
0
0
1
Crossbar Mux Select Bit
000
001
010
011
100
000
1
001
1
010
Input Signal Selection
Input Port A magnitude and exponent pins drive the channel.
Input Port B magnitude and exponent pins drive the channel.
Input Port C magnitude and exponent pins drive the channel.
Input Port D magnitude and exponent pins drive the channel.
Internal PN sequence’s magnitude and exponent bits drive the channel.
Input Ports A and B form a pair to drive I and Q paths of the channel, respectively. Input
Port A exponent pins drive the channel exponent bits.
Input Ports C and D form a pair to drive I and Q paths of the channel, respectively. Input
Port C exponent pins drive the channel exponent bits.
Internal PN sequence’s magnitude and exponent bits drive the channel.
Rev. 0 | Page 26 of 72
AD6636
WIDEBAND INPUT SPECTRUM (–fsamp/2 TO fsamp/2)
SIGNAL OF INTEREST IMAGE
–fs/2
–7fs/8
–3fs/8
–5fs/16
SIGNAL OF INTEREST
–fs/4
–3fs/16
–fs/8
–fs/16
DC
fs/16
fs/8
3fs/16
fs/4
5fs/16
3fs/8
7fs/8
fs/2
5fs/16
3fs/8
7fs/8
fs/2
WIDEBAND INPUT SPECTRUM (30MHz FROM HIGH SPEED ADC)
NCO TUNES SIGNAL TO
–fs/2
–7fs/8
–3fs/8
–5fs/16
–fs/4
–3fs/16
–fs/8
–fs/16
DC
SIGNAL OF INTEREST IMAGE
fs/16
fs/8
3fs/16
fs/4
FREQUENCY TRANSLATION (SINGLE 1MHz CHANNEL TUNED TO BASEBAND)
04998-0-030
SIGNAL OF INTEREST
AFTER FREQUENCY TRANSLATION
Figure 30. Frequency Translation Principle Using the NCO and Mixer
For example, if the carrier frequency is 100 MHz and the clock
frequency is 80 MHz,
mod( f ch , f clk ) 20
= 0.25
=
f clk
80
This, in turn, converts to 0x4000 0000 in the 32-bit twos
complement representation for NCO_FREQ.
Clear Phase Accumulator on Hop
If the carrier frequency is 50 MHz and the clock frequency is
80 MHz,
mod( f ch , f clk )
f clk
=
useful for baseband sampling applications, in which the input
Port A (or C) is connected to the I signal path within the filter
and the Input Port B (or D) is connected to the Q signal path.
This might be desired, if the digitized signal has already been
converted to baseband in prior analog stages or by other digital
preprocessing.
10
= 0.125
80
This, in turn, converts to 0xE000 0000 in the twos complement
32-bit representation.
Mixer
When clear NCO accumulator bit of NCO control register is set
(Logic 1), the NCO phase accumulator is cleared prior to a
frequency hop. Refer to the Chip Synchronization section for
details on frequency hopping. This ensures a consistent phase of
the NCO on each hop. The NCO phase offset is unaffected by
this setting and is still in effect. If phase-continuous hopping is
needed, this bit should be cleared (NCO accumulator is not
cleared). The last phase in the NCO phase register is the
initiating point for the new frequency.
Phase Dither
The NCO is accompanied by a mixer. Its operation is similar to
an analog mixer. It does the down-conversion of input signals
(real or complex) by using the NCO frequency as a local
oscillator. For real input signals, this mixer performs a real
mixer operation (with two multipliers). For complex input
signals, the mixer performs a complex mixer operation (with
four multipliers). The mixer adjusts its operation based on the
input signal (real or complex) provided to each individual
channel.
Bypass
The NCO and the mixer can be bypassed individually in each
channel by writing Logic 1 in the NCO bypass bit in the NCO
control register of the channel under consideration. When
bypassed, down-conversion is not performed and the AD6636
channel functions simply as a real filter on complex data. This is
The AD6636 provides a phase dither option for improving the
spurious performance of the NCO. Writing Logic 1 in the phase
dither enable bit of NCO control register of individual channels
enables phase dither. When phase dither is enabled, random
phase is added to LSBs of the phase accumulator of the NCO.
When phase dither is enabled, spurs due to phase truncation in
the NCO are randomized.
The energy from these spurs is spread into the noise floor and
the spurious free dynamic range is increased at the expense of a
very slight decrease in the SNR. The choice of whether to use
phase dither in a system is ultimately decided by the system
goals. If lower spurs are desired at the expense of a slightly
raised noise floor, phase dither should be employed. If a low
noise floor is desired and the higher spurs can be tolerated or
filtered by subsequent stages, then phase dither is not needed.
Rev. 0 | Page 27 of 72
AD6636
Amplitude Dither
Amplitude dither can be used to improve spurious performance
of the NCO. Amplitude dither is enabled by writing Logic 1 in
the amplitude dither enable bit of the NCO control register of
the channel under consideration. Random amplitude is added
to the LSBs of the sine and cosine amplitudes, when this feature
is enabled. Amplitude dither improves performance by
randomizing the amplitude quantization errors within the
angular-to-Cartesian conversion of the NCO. This option
might reduce spurs at the expense of a slightly raised noise
floor. Amplitude dither and phase dither can be used together,
separately, or not at all.
register is used to set the CIC decimation factor. A binary value
of one less than the decimation factor is written into this
register. The decimation ratio of 1 can be achieved by bypassing
the CIC filter stage. The frequency response of the filter is given
by the following equations. The gain and pass-band droop of
the CIC should be calculated by these equations. Both parameters can be offset in the RCF stage.
H (z ) =
1
2 (SCIC +5)
NCO Frequency Hold-Off Register
When the NCO frequency registers are written by the
microport or serial port, data is passed to a shadow register.
Data can be moved to the main registers when the channel
comes out of sleep mode, or when a sync hop occurs. In either
event, a counter can be loaded with the NCO frequency holdoff register value. The 16-bit unsigned integer counter starts
counting down, clocked by the input port clock selected at the
crossbar mux. When the counter reaches 0, the new frequency
value in the shadow register is written to the NCO frequency
register. Writing 1 in this hold-off register updates the NCO
frequency register as soon as the start sync or hop sync occurs.
See the Chip Synchronization section for details.
Phase Offset
The phase offset register can be written with a value that is
added as an offset to the phase accumulator of the NCO. This
16-bit register is interpreted as a 16-bit unsigned integer. A
0x0000 in this register corresponds to a 0 radian offset and a
0xFFFF corresponds to an offset of 2π × (1 − 1/216) radians.
This register allows multiple NCOs (multiple channels) to be
synchronized to produce complex sinusoids with a known and
steady phase difference.
H( f ) =
FIFTH-ORDER CIC FILTER
The signal processing stage immediately after the NCO is a CIC
filter stage. This stage implements a fixed-coefficient,
decimating, cascade integrated comb filter. The input rate to this
filter is the same as the data rate at the input port; the output
rate from this stage is dependent on the decimation factor.
f CIC =
f in
M cic
1
2
(SCIC +5)
⎞
⎟
⎟
⎠
5
⎛
⎞⎞
⎛M
⎜ SIN⎜ CIC × f ⎟ ⎟
⎜ f in ⎟ ⎟
⎜
⎠⎟
⎝
×⎜
⎜
⎛
f ⎞ ⎟
⎟ ⎟
⎜ SIN ⎜⎜ π
⎟ ⎟
⎜
f
in
⎠ ⎠
⎝
⎝
5
where:
fin is the data input rate to the channel under consideration.
SCIC, the scale factor, is a programmable unsigned integer
between 0 and 20.
The attenuation of the data into the CIC stage should be
controlled in 6 dB increments. For the best dynamic range, SCIC
should be set to the smallest value possible (lowest attenuation
possible) without creating an overflow condition. This can be
accomplished safely using the following equation, where
input_level is the largest possible fraction of the full-scale value
at the input port. This value is output from the NCO stage and
pipelined into the CIC filter.
( (
))
SCIC = ceil log2 MCIC5 × input _ level - 5
OLCIC =
Hop Sync
A hop sync should be issued to the channel, when the channel’s
NCO frequency needs to be changed from one frequency to a
different frequency. This feature is discussed in detail in the
Chip Synchronization section.
⎛ 1 − Z − MCIC
× ⎜⎜
−1
⎝ 1− Z
(M CIC 5 ) × input _ level
2 SCIC
+5
Bypass
The fifth-order CIC filter can be bypassed when no decimation
is required of it. When it is bypassed, the scaling operation is
not performed. In bypass mode, the output of the CIC filter is
the same as the input of the CIC filter.
CIC Rejection
Table 13 illustrates the amount of bandwidth as a percentage of
the data rate into the CIC stage, which can be protected with
various decimation rates and alias rejection specifications. The
maximum input rate into the CIC is 150 MHz (the same as the
maximum input port data rate). The data may be scaled to any
other allowable sample rate.
The decimation ratio, MCIC, can be programmed from 2 to 32
(only integer values). The 5-bit word in the CIC decimation
Rev. 0 | Page 28 of 72
AD6636
Table 13. SSB CIC5 Alias Rejection Table (fin = 1)
−60 dB
8.078
6.367
5.022
4.107
3.463
2.989
2.627
2.342
2.113
1.924
1.765
1.631
1.516
1.416
1.328
1.25
1.181
1.119
1.064
1.013
0.967
0.925
0.887
0.852
0.819
0.789
0.761
0.734
0.71
0.687
0.666
−70 dB
6.393
5.11
4.057
3.326
2.808
2.425
2.133
1.902
1.716
1.563
1.435
1.326
1.232
1.151
1.079
1.016
0.96
0.91
0.865
0.824
0.786
0.752
0.721
0.692
0.666
0.641
0.618
0.597
0.577
0.559
0.541
−80 dB
5.066
4.107
3.271
2.687
2.27
1.962
1.726
1.54
1.39
1.266
1.162
1.074
0.998
0.932
0.874
0.823
0.778
0.737
0.701
0.667
0.637
0.61
0.584
0.561
0.54
0.52
0.501
0.484
0.468
0.453
0.439
−90 dB
4.008
3.297
2.636
2.17
1.836
1.588
1.397
1.247
1.125
1.025
0.941
0.87
0.809
0.755
0.708
0.667
0.63
0.597
0.568
0.541
0.516
0.494
0.474
0.455
0.437
0.421
0.406
0.392
0.379
0.367
0.355
−100 dB
3.183
2.642
2.121
1.748
1.48
1.281
1.128
1.007
0.909
0.828
0.76
0.703
0.653
0.61
0.572
0.539
0.509
0.483
0.459
0.437
0.417
0.399
0.383
0.367
0.353
0.34
0.328
0.317
0.306
0.297
0.287
FIR HALF-BAND BLOCK
The output of the CIC filter is pipelined into the FIR HB (halfband) block. Each channel has two sets of cascading fixedcoefficient FIR and fixed-coefficient half-band filters. The halfband filters decimate by 2. Each of these filters (FIR1, HB1,
FIR2, HB2) are described in the following sections.
3-Tap Fixed-Coefficient Filter (FIR1)
The 3-tap FIR filter is useful in certain filter configurations in
which extra alias protection is needed for the decimating HB1
filter. It is a simple sum-of-products FIR filter with three filter
taps and 2-bit fixed coefficients. Note that this filter does not
decimate. The coefficients of this symmetric filter are {1, 2, 1}.
The normalized coefficients used in the implementation are
{0.25, 0.5, 0.25}.
The user can either use or bypass this filter. Writing Logic 0 to
the FIR1 enable bit in the FIR-HB control register bypasses this
fixed-coefficient filter. The filter is useful only in certain filter
configurations and bypassing it for other applications results in
power savings.
0
0.34
0.66
–8.33
FIR1 RESPONSE
–16.67
–25.00
–33.33
–41.67
–50.00
–58.33
–66.67
–75.00
–81
–83.33
–91.67
Example Calculations
–100.00
Goal: Implement a filter with an input sample rate of 100 MHz
requiring 100 dB of alias rejection for a ± 1.4 MHz pass band.
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
FRACTION OF FIR1 INPUT SAMPLE RATE
0.9
Figure 31. FIR1 Filter Response to the Input Rate of the Filter
Solution: First determine the percentage of the sample rate that
is represented by the pass band.
BW fraction = 100 ×
1.4 MHz
100 MHz
= 1.4
Rev. 0 | Page 29 of 72
04998-0-031
MCIC5
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
17
18
19
20
21
22
23
24
25
26
27
28
29
30
31
32
In the −100 dB column in Table 13, find the value greater than
or equal to the pass-band percentage of the clock rate. Then
find the corresponding rate decimation factor (MCIC). For an
MCIC of 6, the frequency that has −100 dB of alias rejection is
1.48%, which is slightly larger than the 1.4% calculated.
Therefore, for this example, the maximum bound on CIC
decimation rate is 6. A higher MCIC means less alias rejection
than the 100 dB required.
dBc
Table 13 can be used to decide the minimum decimation
required in the CIC stage to preserve a certain bandwidth. The
CIC5 stage can protect a much wider bandwidth to any given
rejection, when a decimation ratio lower than that identified in
the table is used. The table helps to calculate an upper boundary
on decimation, MCIC, given the desired filter characteristics.
AD6636
0
This filter runs at the same sample rate as the CIC filter output
rate and is given by
f FIR1 =
0.43
0.57
10
20
30
f in
M cic
40
50
dBc
HB1 RESPONSE
where:
60
70
fin is the input rate in to the channel.
–77
80
90
MCIC is the decimation ratio in the CIC filter stage.
100
120
0
0.1
Decimate-by-2 Half-Band Filter (HB1)
Table 14. Fixed Coefficients for HB1 Filter
Coefficient
Number
C1, C11
C3, C9
C5, C7
C6
Normalized
Coefficient
0.013671875
−0.103515625
0.58984375
1
Decimal Coefficient
(10-Bit)
7
−53
302
512
Similar to the FIR1 filter, this filter can be used or bypassed.
Writing Logic 0 to the HB1 enable bit in the FIR-HB control
register bypasses this fixed-coefficient HB filter. The filter is
useful only in certain filter configurations and bypassing it for
other applications results in power savings. For example, it is
useful in narrow-band and wideband output applications in
which more filtering is required as compared to very wide
bandwidth applications in which a higher output rate might
prohibit the use of a decimating filter. The response of the filter
is shown in Figure 32.
0.9
Figure 32. HB1 Filter Response to the Input Rate of the Filter
The filter has a maximum input sample rate of 150 MHz
and, when filter is not bypassed, the maximum output rate is
75 MHz.
The filter has a ripple of 0.0012 dB and rejection of 77 dB. For
an alias rejection of 77 dB, the alias-protected bandwidth is 14%
of the filter input sample rate. The bandwidth of the filter for a
ripple of 0.00075 dB is also the same as the alias-protected
bandwidth, due to the nature of half-band filters. The 3 dB
bandwidth of this filter is 44% of the filter input sample rate.
For example, if the sample rate into the filter is 50 MHz, then
the alias-protected bandwidth of the HB1 filter is 7 MHz. If the
bandwidth of the required carrier is greater than 7 MHz, then
HB1 might not be useful.
0
0.43
0.57
–10
–20
–30
–40
–50
dBc
The next stage of the FIR-HB block is a decimate-by-2 halfband filter. The 11-tap, symmetrical, fixed-coefficient HB1 filter
has low power consumption due to its polyphase implementation. The filter has 22 bits of input and output data with 10-bit
coefficients. Table 14 lists the coefficients of the half-band filter.
The normalized coefficients used in the implementation and
the 10-bit decimal equivalent value of the coefficients are also
listed. Other coefficients are zeros.
0.2
0.3
0.4
0.5
0.6
0.7
0.8
FRACTION OF HB1 INPUT SAMPLE RATE
04998-0-032
110
The maximum input and output rates for this filter are
150 MHz.
–60
–70
FIR1 + HB1 RESPONSE
–80
–90
–100
–107
The input sample rate of this filter is the same as the CIC filter
output rate and is given by
f HB1 =
f in
M cic
–120
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
FRACTION OF HB1 INPUT SAMPLE RATE
0.9
04998-0-033
–110
Figure 33. Composite Response of FIR1 and HB1 Filters to Their Input Rate
where:
fin is the input rate in to the channel.
MCIC is the decimation ratio in the CIC filter stage.
Rev. 0 | Page 30 of 72
AD6636
6-Tap Fixed Coefficient Filter (FIR2)
Decimate-by-2 Half-Band Filter (HB2)
Following the first cascade of the FIR1 and HB1 filters is the
second cascade of the FIR2 and HB2 filters. The 6-tap, fixedcoefficient FIR2 filter is useful in providing extra alias
protection for the decimating HB2 filter in certain filter
configurations. It is a simple sum-of-products FIR filter with six
filter taps and 5-bit fixed coefficients. Note that this filter does
not decimate. The normalized coefficients used in the
implementation and the 5-bit decimal equivalent value of the
coefficients are listed in Table 15.
The second stage of the second cascade of the FIR-HB block is a
decimate-by-2 half-band filter. The 27-tap, symmetric, fixedcoefficient HB2 filter has low power consumption due to its
polyphase implementation. The filter has 20 bits of input and
output data with 12-bit coefficients. The normalized coefficients
used in the implementation and the 10-bit decimal equivalent
value of the coefficients are listed in Table 16. Other coefficients
are zeros.
Table 16. HB2 Filter Fixed Coefficients
Table 15. 6-Tap FIR1 Filter Coefficients
Coefficient
Number
C0, C5
C1, C4
C2, C3
Normalized
Coefficient
−0.125
0.1875
0.9375
Decimal Coefficient
(5-Bit)
−2
3
15
The user can either use or bypass this filter. Writing Logic 0 to
FIR2 enable bit in the FIR-HB control register bypasses this
fixed-coefficient filter. The filter is useful only in certain filter
configurations and bypassing it for other applications results in
power savings. The filter is especially useful in increasing the
stop-band attenuation of the HB2 filter that follows. Therefore,
it is optimal to use both FIR2 and HB2 in a configuration.
This filter runs at a sample rate given by one of the following
equations:
fFIR2 = fHB1, if HB1 is bypassed
fFIR2 =
f HB1
, if HB1 is not bypassed
2
Coefficient
Number
C1, C27
C3, C25
C5, C23
C7, C21
C9, C19
C11, C17
C13, C15
C14
Normalized
Coefficient
0.00097656
−0.00537109
0.015
−0.0380859
0.0825195
0.1821289
0.6259766
1
Decimal Coefficient
(12-Bit)
2
−11
32
−78
169
−373
1282
2048
Similar to the HB1 filter, the user can either use or bypass this
filter. Writing Logic 0 to the HB1 enable bit in the FIR-HB
control register bypasses this fixed-coefficient HB filter. The
filter is useful only in certain filter configurations and bypassing
it for other applications results in power savings. For example,
the filter is useful in narrow-band applications in which more
filtering is required, as compared to wide-band applications, in
which a higher output rate might prohibit the use of a decimating filter. The response of the HB2 filter is shown in Figure 35.
0.01
0.34
0.66
–9.99
where fHB1 is the input rate of the HB1 filter.
–19.99
The maximum input and output rate for this filter is 75 MHz.
The response of the FIR2 filter is shown in Figure 34.
–29.99
–39.99
–49.99
0.39
0.61
dBc
0
–8.33
–65
–70.00
–16.67
–25.00
FIR2 RESPONSE
–80.00
–30
HB2 RESPONSE
–90.00
–33.33
–100.00
–41.67
–110.00
–50.00
–120.00
–58.33
0
–66.67
–75.00
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
FRACTION OF HB2 INPUT SAMPLE RATE
0.9
Figure 35. HB2 Filter Response to the Input Rate of the Filter
–83.33
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
FRACTION OF FIR2 INPUT SAMPLE RATE
0.9
04998-0-034
–91.67
–100.00
Figure 34. FIR2 Filter Response to the Input Rate of the Filter
Rev. 0 | Page 31 of 72
04998-0-035
dBc
–60.00
AD6636
The filter input sample rate is the same as the FIR2 filter output
rate and is given by one of the following equations:
fHB2 = fFIR2 =
f HB1
, if HB1 is not bypassed
2
where:
fFIR1 is the input rate of the FIR1 filter.
fHB1 is the input rate of the HB1 filter.
The input to the filter has a maximum of 75 MHz. The
maximum output rate when not bypassed is 37.5 MHz.
The filter has a ripple of 0.00075 dB and rejection of 81 dB. For
an alias rejection of 81 dB, the alias-protected bandwidth is 33%
of the filter input sample rate. The bandwidth of the filter for a
ripple of 0.00075 dB is the same as alias-protected bandwidth,
due to the nature of half-band filters. The 3 dB bandwidth of
this filter is 47% of the filter input sample rate. For example, if
the sample rate into the filter is 25 MHz, then the aliasprotected bandwidth of the HB2 filter is 8.25 MHz (33% of
25 MHz). If the bandwidth of the required carrier is greater
than 8.25 MHz, then HB2 might not be useful.
0.01
0.34
Table 17. Data Router Select Settings
MRCF Data Select [2:0]
000
001
010
011
1x0
1x1
fHB2 = fFIR2 = fHB1, if HB1 is bypassed
0.66
Data Source
Channel 0
Channel 1
Channel 2
Channel 3
Channel 4
Channel 5
Allowing different channel back ends to select different channel
front ends is useful in the polyphase implementation of filters.
When multiple AD6636 channels are used to process a single
carrier, a single-channel front end feeds more than one channel
back end. After processing through the channel back ends (RCF
filters), the data is interleaved back from all the polyphased
channels.
MONO-RATE RAM COEFFICIENT FILTER (MRCF)
The MRCF is a programmable sum-of-products FIR filter. This
filter block comes after the first data router and before the
DRCF and CRCF programmable filters. It consists of a
maximum of eight taps with 6-bit programmable coefficients.
Note that this block does not decimate and is used as a helper
filter for the DRCF and CRCF filters that follow in the signal
chain.
–9.99
The number of filter taps that are to be calculated is programmable using the 3-bit number-of-taps word in the MRCF
control register of the channel under consideration. The 3-bit
word programmed is one less than the number of filter taps.
The coefficients themselves are programmed in eight MRCF
coefficient memory registers for individual channels. The input
and output data to the block are both 20-bit.
–19.99
–29.99
–39.99
dBc
–49.99
–60.00
FIR2 + HB2
RESPONSE
–70.00
–80.00
–90
–90.00
Symmetry
–100.00
–120.00
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
FRACTION OF HB2 INPUT SAMPLE RATE
0.9
04998-0-036
–110.00
Figure 36. Composite Response of FIR1 and HB1 filters to Their Input Rates
INTERMEDIATE DATA ROUTER
Following the FIR-HB cascade filters is the intermediate data
router. This data router consists of muxes that allow the I and Q
data from any channel front end (input port + NCO + CIC +
FIR-HB) to be processed by any channel back end (MRCF +
DRCF + CRCF). The choice of channel front end is made by
programming a 3-bit MRCF data select word in the MRCF
control register. The valid values for this word and their
corresponding settings are listed in Table 17.
Though the MRCF filter does not require symmetrical filters, if
the filter is symmetrical, then the symmetry bit in the MRCF
control register should be set. When this bit is set, only half of
the impulse response needs to be programmed into the MRCF
coefficient memory registers. For example, if the number of
filter taps is equal to five or six and the filter is symmetrical,
then only three coefficients need to be written into the
coefficient memory. For both symmetrical and asymmetrical
filters, the number of filter taps is limited to eight.
Clock Rate
The MRCF filter runs on an internal high speed PLL clock. This
clock rate can be as high as 200 MHz. If the half clock rate bit in
the MRCF control register is set, then only half the PLL clock
rate is used (maximum of 100 MHz). This results in power
savings, but can only be used if certain conditions are met.
Rev. 0 | Page 32 of 72
AD6636
Because this filter is nondecimating, the input and output rates
are both same and equal to one of the following:
fMRCF = fHB2, if HB2 is bypassed
fMRCF =
Bypass
The DRCF filter can be used in normal operation or bypassed
using the DRCF bypass bit in the DRCF control register. When
the DRCF filter is bypassed, no scaling is applied and the output
of the filter is the same as the input to the DRCF filter.
f HB2
, if HB2 is not bypassed
2
If fPLLCLK is the PLL clock and if
f MRCF × N TAPS ≥
The decimation rate is programmable using the 4-bit DRCF
decimation rate word in the DRCF control register. Again, the
value written is the decimation rate minus one.
Scaling
f PLLCLK
2
then half of the PLL clock can be used for processing (power
savings). Otherwise, the PLL clock should be used.
The output of the DRCF filter can be scaled using the 2-bit
DRCF scaling word in the DRCF control register. Table 19 lists
the valid values for the 2-bit word and their corresponding
settings.
Bypass
The MRCF filter can be used in normal operation or bypassed
using the MRCF bypass bit in the MRCF control register. When
the filter is bypassed, the output of the filter is the same as the
input of the filter. Bypassing the MRCF filter when not required
results in power savings.
Scaling
The output of the MRCF filter can be scaled by using the 2-bit
MRCF scaling word in the MRCF control register. Table 18
shows the valid values for the 2-bit word and their corresponding settings.
Table 18. MRCF Scaling Factor Settings
MRCF Scale Word [1:0]
00
01
10
11
Scaling Factor
18.06 dB attenuation
12.04 dB attenuation
6.02 dB attenuation
No scaling, 0 dB
Table 19. DRCF Scaling Factor Settings
DRCF Scale Word [1:0]
00
01
10
11
Scaling Factor
18.06 dB attenuation
12.04 dB attenuation
6.02 dB attenuation
No scaling, 0 dB
Symmetry
The DRCF filter does not require symmetrical filters. However,
if the filter is symmetrical, then the symmetry bit in the DRCF
control register should be set. When this bit is set, only half of
the impulse response needs to be programmed into the DRCF
coefficient memory registers. For example, if the number of
filter taps is equal to 15 or 16 and the filter is symmetrical, then
only eight coefficients need to be written into the coefficient
memory. Because a total of 64 taps can be written into the
memory registers, the DRCF can perform 64 asymmetrical filter
taps or 128 symmetrical filter taps.
Coefficient Offset
DECIMATING RAM COEFFICIENT FILTER (DRCF)
Following the MRCF is the programmable DRCF FIR filter.
This filter can calculate up to 64 asymmetrical filter taps or up
to 128 symmetrical filter taps. The filter is also capable of a
programmable decimation rate of from 1 to 16. A flexible
coefficient offset feature allows loading multiple filters into the
coefficient RAM and changing the filters on the fly. The
decimation phase feature allows a polyphase implementation,
where multiple AD6636 channels are used for processing a
single carrier.
The DRCF filter has 20-bit input and output data and 14-bit
coefficient data. The number of filter taps to calculate is
programmable and is set in the DRCF taps register. The value
of the number of taps minus one is written to this register.
For example, a value of 19 in the register corresponds to
20 filter taps.
More than one set of filter coefficients can be loaded into
coefficient RAM at any given time (given sufficient RAM
space). The coefficient offset can be used in this case to access
the two or more different filters. By changing the coefficient
offset, the filter coefficients being accessed can be changed on
the fly. This decimal offset value is programmed in the DRCF
coefficient offset register. When this value is changed during the
calculation of a particular output data sample, the sample
calculation is completed using the old coefficients, and the new
coefficient offset from the next data sample calculation is used.
Decimation Phase
When more than one channel of AD6636 is used to process one
carrier, polyphase implementation of corresponding channels’
DRCF or CRCF is possible using the decimation phase feature.
This feature can be used only under certain conditions. The
decimation phase is programmed using the 4-bit DRCF
decimation phase word of the DRCF control register.
Rev. 0 | Page 33 of 72
AD6636
Maximum Number of Taps Calculated
7.
The output rate of the DRCF filter is given by
f DRCF =
f MRCF
M DRCF
where:
fMRCF is the data rate out of the MRCF filter and into the DRCF
filter.
MDRCF is the decimation rate in the DRCF filter.
The DRCF filter consists of two multipliers (one each for the
I and Q paths). Each multiplier, working at the high speed clock
rate (PLL clock), can do one multiply (or one tap) per high
speed clock cycle. Therefore, the maximum number of filter
taps that can be calculated (symmetrical or asymmetrical filter)
is given by
⎛f
Maximum Number of Taps = ceil ⎜⎜ PLLCLK
⎝ f DRCF
⎞
⎟ −1
⎟
⎠
where:
fPLLCLK is the high speed internal processing clock generated by
the PLL clock multiplier.
Note that each write or read access increments the internal
RAM address. Therefore, all coefficients should be read first
before reading them back. Also, for debugging purposes, each
RAM address can be written individually by making the start
address and stop addresses the same. Therefore, to program one
RAM location, the user writes the address of the RAM location
to both the start and stop address registers, and then writes the
coefficient memory register.
Programming DRCF Registers for a Symmetric Filter
To program the DRCF registers for a symmetrical filter:
1.
Write NTAPS – 1 in the DRCF taps register, where NTAPS
is the number of filter taps. The absolute maximum value
for NTAPS is 128 in symmetric filter mode.
2.
Write ceil(64 – NTAPS/2) for the DRCF coefficient offset
register, where the ceil function takes the closest integer
greater than or equal to the argument.
3.
Write 1 for the symmetrical filter bit in the DRCF control
register.
4.
Write the start address for the coefficient RAM, typically
equal to coefficient offset register, in the DRCF start
address register.
5.
Write the stop address for the coefficient RAM, typically
equal to ceil(NTAPS/2) – 1, in the DRCF stop address
register.
6.
Write all coefficients to the DRCF coefficient memory
register, starting with the middle of the filter and working
towards the end of the filter. When coefficients are
numbered 0 to NTAPS – 1, the middle coefficient is given
by the coefficient number ceil(NTAPS/2). If in 8-bit
microport mode or serial port mode, write the lower byte
of the memory register first and then the higher byte. After
each write access to the DRCF coefficient memory register,
the internal RAM address is incremented starting with the
start address and ending with stop address.
fDRCF is the output rate of the DRCF filter calculated above.
Programming DRCF Registers for an Asymmetrical Filter
To program the DRCF registers for an asymmetrical filter:
1.
Write NTAPS – 1 in the DRCF taps register, where NTAPS
is the number of filter taps. The absolute maximum value
for NTAPS is 64 in asymmetrical filter mode.
2.
Write 0 for the DRCF coefficient offset register.
3.
Write 0 for the symmetrical filter bit in the DRCF control
register.
4.
Write the start address for the coefficient RAM, typically
equal to the coefficient offset register in the DRCF start
address register.
5.
In the DRCF stop address register, write the stop address
for the coefficient RAM, typically equal to the following:
Coefficient Offset + NTAPS − 1
6.
Write all coefficients in reverse order (start with last
coefficient) to the DRCF coefficient memory register. If in
8-bit microport mode or serial port mode, write the lower
byte of the memory register first and then the higher byte.
After each write access to the DRCF coefficient memory
register, the internal RAM address is incremented starting
with the start address and ending with the stop address.
Note that each write or read access increments the internal
RAM address. Therefore, all coefficients should be read first
before reading them back. Also, for debugging purposes, each
RAM address can be written individually by making the start
and stop addresses the same. Therefore, to program one RAM
location, the user writes the address of the RAM location to
both the start and stop address registers, and then writes the
coefficient memory register.
Rev. 0 | Page 34 of 72
AD6636
Coefficient Offset
CHANNEL RAM COEFFICIENT FILTER (CRCF)
Following the DRCF is the programmable decimating CRCF
FIR filter. The only difference between the DRCF and CRCF
filters is the coefficient bit width. The DRCF has 14-bit
coefficients, while the DRCF has 20-bit coefficients.
This filter can calculate up to 64 asymmetrical filter taps or up
to 128 symmetrical filter taps. The filter is capable of a
programmable decimation rate from 1 to 16. The flexible
coefficient offset feature allows loading multiple filters into the
coefficient RAM and changing the filters on the fly. The
decimation phase feature allows for a polyphase implementation in which multiple AD6636 channels are used to process a
single carrier.
The CRCF filter has 20-bit input and output data and 14-bit
coefficient data. The number of filter taps to calculate is
programmable and is set in the CRCF taps register. The value of
the number of taps minus one is written to this register. For
example, a value of 19 in the register corresponds to 20 filter
taps. The decimation rate is programmable using the 4-bit
CRCF decimation rate word in the CRCF control register.
Again, the value written is the decimation rate minus one.
More than one set of filter coefficients can be loaded into the
coefficient RAM at any time (given sufficient RAM space). The
coefficient offset can be used in this case to access the two or
more different filters. By changing the coefficient offset, the
filter coefficients being accessed can be changed on the fly. This
decimal offset value is programmed in the CRCF coefficient
offset register. When this value is changed during the calculation of a particular output data sample, the sample calculation is
completed using the old coefficients and the new coefficient
offset is brought into effect from the next data sample
calculation.
Decimation Phase
When more than one channel of the AD6636 is used to process
one carrier, polyphase implementation of the corresponding
channels’ DRCF or CRCF is possible using the decimation
phase feature. This feature can be used only under certain
conditions. The decimation phase is programmed using the
4-bit CRCF decimation phase word of the CRCF control
register.
Maximum Number of Taps Calculated
The output rate of the CRCF filter is given by
Bypass
The CRCF filter can be used in normal operation or bypassed
using the CRCF bypass bit in the CRCF control register. When
the CRCF filter is bypassed, no scaling is applied and the output
of the filter is the same as the input to the CRCF filter.
Scaling
The output of the CRCF filter can be scaled using the 2-bit
CRCF scaling word in the CRCF control register. Table 20
shows the valid values for the 2-bit word and the corresponding
settings. | ∑COEFF | is the sum of all coefficients (in normalized
form) used to calculate the FIR filter.
Table 20. CRCF Scaling Factor Settings
CRCF Scale Word [1:0]
00
01
10
11
Scaling Factor
18.06 dB attenuation
12.04 dB attenuation
6.02 dB attenuation
No scaling, 0 dB
f CRCF =
f DRCF
M CRCF
where:
fDRCF is the data rate out of the DRCF filter and into the CRCF
filter.
MCRCF is the decimation rate in the CRCF filter.
The CRCF filter consists of two multipliers (one each for the I
and Q paths). Each multiplier, working at the high speed clock
rate (PLL clock), can multiply (or tap once). Therefore, the
maximum number of filter taps that can be calculated
(symmetrical or asymmetrical filter) is given by
⎛f
Maximum Number of Taps = ceil ⎜⎜ PLLCLK
⎝ f CRCF
⎞
⎟ −1
⎟
⎠
where:
Symmetry
The CRCF filter does not require symmetrical filters. However,
if the filter is symmetrical, then the symmetry bit in the CRCF
control register should be set. When this bit is set, only half the
impulse response needs to be programmed into the CRCF
coefficient memory registers. For example, if the number of
filter taps is equal to 15 or 16 and the filter is symmetric, then
only eight coefficients need to be written into the coefficient
memory. Because a total of 64 taps can be written into the
memory registers, the CRCF can perform 64 asymmetrical filter
taps or 128 symmetrical filter taps.
fPLLCLK is the high speed internal processing clock generated by
the PLL clock multiplier.
fCRCF is the output rate of the CRCF filter as calculated
previously.
Rev. 0 | Page 35 of 72
AD6636
Programming CRCF Registers for an Asymmetrical Filter
5.
In the CRCF stop address register, write the stop
address for the coefficient RAM, typically equal to
ceil(NTAPS/2) – 1.
6.
Write all coefficients to the CRCF coefficient memory
register, starting with middle of the filter and working
towards the end of the filter. When coefficients are
numbered 0 to NTAPS – 1, the middle coefficient is given
by the coefficient number ceil(NTAPS/2). In 8-bit
microport mode or serial port mode, write the lower byte
of the memory register first and then the higher byte. In
16-bit microport mode, write the lower 16-bits of the
CRCF memory register first and then the high four bits.
After each write access to the CRCF coefficient memory
register, the internal RAM address is incremented starting
with the start address and ending with the stop address.
To program the CRCF registers for an asymmetrical filter:
1.
Write NTAPS – 1 in the CRCF taps register, where NTAPS
is the number of filter taps. The absolute maximum value
for NTAPS is 64 in asymmetrical filter mode.
2.
Write 0 for the CRCF coefficient offset register.
3.
Write 0 for the symmetrical filter bit in the CRCF control
register.
4.
In the CRCF start address register, write the start address
for the coefficient RAM, typically equal to the coefficient
offset register.
5.
In the CRCF stop address register, write the stop address
for the coefficient RAM, typically equal to the following:
Coefficient Offset + NTAPS – 1
6.
Write all coefficients in reverse order (start with last
coefficient) to the CRCF coefficient memory register. In
8-bit microport mode or serial port mode, write the lower
byte of the memory register first and then the higher byte.
In 16-bit microport mode, write the lower 16-bits of the
CRCF memory register first and then the high four bits.
After each write access to the CRCF coefficient memory
register, the internal RAM address is incremented starting
with the start address and ending with the stop address.
Note that each write or read access increments the internal
RAM address. Therefore, all coefficients should be read first
before reading them back. Also, for debugging purposes, each
RAM address can be written individually by making the start
and stop addresses the same. Therefore, to program one RAM
location, the user writes the address of the RAM location to
both the start and stop address registers, and then writes the
coefficient memory register.
Programming CRCF Registers for a Symmetrical Filter
To program the CRCF registers for a symmetrical filter:
1.
Write NTAPS – 1 in the CRCF taps register, where NTAPS
is the number of filter taps. The absolute maximum value
for NTAPS is 128 in symmetrical filter mode.
2.
Write ceil(64 – NTAPS/2) for the CRCF coefficient offset
register, where the ceil function takes the closest integer
greater than or equal to the argument.
3.
Write 1 for the symmetrical filter bit in the CRCF control
register.
4.
In the CRCF start address register, write the start address
for the coefficient RAM, typically equal to the coefficient
offset register.
Note that each write or read access increments the internal
RAM address. Therefore, all coefficients should be read first
before reading them back. Also, for debugging purposes, each
RAM address can be written individually by making the start
and stop addresses the same. Therefore, to program one RAM
location, the user writes the address of the RAM location to
both the start and stop address registers, and then writes the
coefficient memory register.
INTERPOLATING HALF-BAND FILTER
The AD6636 has interpolating half-band FIR filters that
immediately follow the CRCF programmable FIR filters and
precede the second data router. Each interpolating half-band
filter takes 22-bit I and 22-bit Q data from the preceding CRCF
and outputs rounded 22-bit I and 22-bit Q data to the second
data router. A 10-tap fixed-coefficient filter is implemented in
this stage.
The maximum input rate into this block is 17 MHz. Consequently, the maximum output is constrained to 34 MHz. The
normalized coefficients used in the implementation and the
10-bit decimal equivalent value of the coefficients are listed in
Table 21. Other coefficients are 0.
Table 21. Interpolating HB Filter Fixed Coefficients
Coefficient
Number
C1, C11
C3, C9
C5, C7
C6
Normalized
Coefficient
0.02734375
−0.12890625
0.603515625
1
Decimal Coefficient
(10-Bit)
14
−66
309
512
The half-band filters interpolate the incoming data by 2×. For a
channel running at 2× the chip rate, the half-band can be used
to output channel data at 4× the chip rate. The interpolation
operation creates an image of the baseband signal, which is
filtered out by the half-band filter.
Rev. 0 | Page 36 of 72
AD6636
The second subblock can perform two special functions, either
complex filter completion or biphase filtering. The combined
data is passed on to the AGCs.
The image rejection of this filter is about 55 dB, but is still
sufficient, because the image is from the desired signal, not an
interfering signal. Note that the interpolating half-band filter
can be enabled by writing a Logic 1 to Bit 9 of the MRCF
control registers.
Interleaving Data
In some cases, filtering using a single channel is insufficient.
For such setups, it is advantageous to combine the filtering
resources of more than one channel.
The frequency response of the interpolating half-band FIR is
shown in Figure 37 with respect to the chip rate. The input rate
to this filter is 2× the chip rate, and the output rate is 4× the
chip rate.
0.75
1.25
–20
INTERPOLATING
HALFB AND
FILTER RESPONSE
dBc
–40
–53
For example, two channels need to work together to produce a
filter at an output rate of 10 MHz when the input rate is
100 MHz. Each channel is decimated by a factor of 20 (total
decimation) to achieve the desired output rate of 5 MHz each.
This compares to a decimation of 10, if a single channel were
filtering.
–60
04998-0-037
–80
–100
0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
1.8
FREQUENCY AS FRACTION OF INPUT RATE
The same coefficients are programmed in both channels’ RCF
filters, and the decimation phases are set to 0 and 1. The
decimation phases can be set to 0 for one channel, and 1 for the
second channel in the pair. This causes the first channel to
produce the even outputs, and the second to produce the odd
outputs of the filter. The streams can then be recombined
(interleaved) to produce the desired 10 MHz output rate. The
benefit is that now each channel’s RCF has time to calculate
twice as many taps, because it has a lower output rate.
Figure 37. Interpolating Half-Band Frequency Response
OUTPUT DATA ROUTER
The output data router circuit precedes the six AGCs of the final
output block and immediately follows the interpolating halfband filters. This block consists of two subblocks. The first
subblock is responsible for combining (interleaving) data from
more than one channel into a single stream of data.
CH0
AGC0
STR0
AGC0
CH1
AGC1
STR1
AGC1
PARALLEL
PORT A
CH2
AGC2
STR2
AGC2
COMPLEX
FILTER
COMPLETION
STREAM
CONTROL
CH3
STR3
PARALLEL
PORT B
AGC3
AGC3
CH4
STR4
PARALLEL
PORT C
AGC4
AGC4
CH5
STR5
AGC5
AGC5
Figure 38. Output Data Router Block Diagram
Rev. 0 | Page 37 of 72
04998-0-038
0
Multiple channels can be set up to work on the ADC input port
data with the same NCO and filter setups. The decimation
phase values in one of the RCF filters are set such that the
channel filters are exactly out of phase with each other. In the
data router, these multiple channels are interleaved (combined)
to form a single stream of data. Because each individual channel
is decimated more than it would be if a single channel were
filtering, a larger number of filter taps can be calculated.
AD6636
The interleaving function is a simple time-multiplexing
function, with lower data rate on the input side and higher data
rate on the output side. The output data rate is the sum of all
input stream data rates that are combined.
The terms calculated are as follows:
•
(ICi, QCi) from first channel
•
(Icq, QCq) from the second channel
The channels that need to be combined are programmable with
sufficient flexibility. Table 22 gives the combinations that are
possible using a 4-bit word (stream control bits) in the Parallel
Port Control 2 register.
Using these terms, the complex filter is completed by applying
the following formula:
After interleaving of data (see the Output Data Router section),
the data is passed to the second subblock, in which either
complex filter completion or biphase filtering can be
performed.
The channels to be combined can be programmed using a 3-bit
complex control word in the Parallel Output Control 2 register.
The values for the 3-bit control word and the corresponding
settings are listed in Table 23.
Complex Filter Completion
These outputs go to the six available AGCs. Not all AGCs need
to be used in the different applications, so unused AGCs can be
bypassed and the output data streams ignored by the parallel
output ports. For example, if Streams 0 and 1 are combined for a
complex filter, AGC 1 can be bypassed, because Stream 1 is
already combined into Stream 0 and sent to AGC 0.
In normal operation, each individual channel’s filter performs
real coefficient, complex data filtering.
Two channels are used to perform complex coefficient data
filtering. One channel is loaded with the real part (in-phase) of
the coefficients; the other channel is loaded with the imaginary
part (quadrature) of the coefficients.
(I + jQ) (Ci + jCq) = (ICi − QCq) + j(ICq + QCi)
Table 22. Stream Control Bit Combinations
Stream Control Bits
0000
0001
0010
0011
0100
0101
0110
0111
1000
1001
Any other state
Output Streams
Ch 0/1 combined, Ch 2, Ch 3, Ch 4, Ch 5 independent
Ch 0/1/2 combined, Ch 3, Ch 4, Ch5 independent
Ch 0/1/2/3 combined; Ch 4, Ch 5 independent
Ch 0/1/2/3/4 combined; Ch 5 independent
Ch 0/1/2/3/4/5 combined
Ch 0/1/2 combined, Ch 3/4/5 combined
Ch 0/1 combined, Ch 2/3 combined, Ch 4/5 combined
Ch 0/1 combined, Ch 2/3 combined, Ch 4, Ch 5 independent
Ch 0/1/2 combined, Ch 3/4 combined, Ch 5 independent
Ch 0/1/2/3 combined, Ch 4/5 combined.
Independent channels
No. of Streams
5
4
3
2
1
2
3
3
3
2
6
Table 23. Definitions for Complex Control Register Selections
Complex Control Word
000
001
010
011
101
110
111
Data Routing
No complex filters
Stream 0/1 combined
Stream 0/1 combined, Stream 2/3
combined
Stream 0/1 combined, Stream 2/3
combined, Stream 4/5 combined
Stream 0/1 Combined
Stream 0/1 combined, Stream 2/3
combined
Stream 0/1 combined, Stream 2/3
combined, Stream 4/5 combined
Comments
Stream control register controls AGC usage.
Allows Ch 0 and Ch 1 to form a complex filter.
Allows Ch 0 and Ch 1 to form a complex filter and Ch 2 and Ch 3 to
form a complex filter.
Allows Ch 0 and Ch 1 to form a complex filter, Ch 2 and Ch 3 to form a
complex filte,r and Ch 4 and Ch 5 to form a complex filter.
Allows Ch 0 and Ch 1 to form a biphase filter.
Allows Ch 0 and Ch 1 to form a biphase filter, and Ch 2 and Ch 3 to
form a biphase filter.
Allows Ch 0 and Ch 1 to form a biphase filter, Ch 2 and Ch 3 to form a
biphase filter, and Ch 4 and Ch 5 to form a biphase filter.
Rev. 0 | Page 38 of 72
AD6636
Biphase Filtering Option
The second special function that can be performed by the
second subblock of the output data router is called the biphase
filtering option. With this option, the AD6636 can be used to
process data from ADCs that run faster than the input clock
frequency by using two channels or two streams to form a
biphase filter.
For example, a 300 MHz ADC can be used with a clock rate of
150 MHz driving the ADC. The ADC data can be decimated by
2 to produce even and odd data streams of data. The even
stream can be clocked into ADC Input Port A, and the odd
stream can be clocked into ADC Input Port B. These input ports
drive separate channels or separate groups of channels. The
filters of the RCF can be designed to place a 300 MHz sample
time difference (1/300 MHz = 3.3 ns) between the even and odd
path filters.
After the channel-filter coefficients have appropriate delay, a
complex addition of the odd and even sample channels can be
performed to create a single filter. This equivalent filter looks
like a single channel with a 300 MHz input rate, even though
the clock rate of the chip runs at only 150 MHz.
A biphase filter summation is implemented by the following
equation:
Output = (Ie × Ce + Io × Co) + j(Qe × Ce + Qo × Co)
where:
Ie × Ce, Qe × Ce are even in-phase and quadrature-phase
samples from one stream.
Io × Co and Qo × Co are odd in-phase and quadrature-phase
samples from the other stream.
Ce and Co are the even and odd coefficients, which differ by
1 high speed sample time (300 MHz in the previous example).
Users can program certain streams to be summed using the
biphase filtering option. This option can be programmed using
the same 3-bit complex control word in the Parallel Output
Control 2 register. The values for the 3-bit control word and
their corresponding settings are listed in Table 23.
AUTOMATIC GAIN CONTROL
The AD6636 is equipped with six independent automatic gain
control (AGC) loops that directly follow the second data router
and immediately precede the parallel output ports. Each AGC
circuit has 96 dB of range. It is important that the decimating
filters of the AD6636 preceding the AGC reject unwanted
signals, so that each AGC loop is operating only on the carrier
of interest, and carriers at other frequencies do not affect the
ranging of the loop.
The AGC compresses the 24-bit complex output from the
second data router into a programmable word size of 4 to 8, 10,
12, or 16 bits. Because the small signals from the lower bits are
pushed in to higher bits by adding gain, the clipping of the
lower bits does not compromise the SNR of the signal of
interest.
The AGC maintains a constant mean power on the output
despite the level of the signal of interest, allowing operation in
environments where the dynamic range of the signal exceeds
the dynamic range of the output resolution. The output width of
the AGC is set by writing a 3-bit AGC word length word in the
AGC control register of the individual channel’s memory map.
The AGC can be bypassed, if needed, and, when bypassed, the
24-bit complex input word is still truncated to a 16-bit value
that is output through the parallel port output. The six AGCs
available on the AD6636 are programmable through the six
channel memory maps. AGCs corresponding to individual
channels can be bypassed by writing Logic 1 to AGC bypass bit
in the AGC control register.
Three sources of error can be introduced by the AGC function:
underflow, overflow, and modulation. Underflow is caused by
truncation of bits below the output range. Overflow is caused by
clipping errors when the output signal exceeds the output range.
Modulation error occurs when the output gain varies while
receiving data.
The desired signal level should be set based on the probability
density function of the signal, so that the errors due to underflow and overflow are balanced. The gain and damping values
of the loop filter should be set, so that the AGC is fast enough to
track long-term amplitude variations of the signal that might
cause excessive underflow or overflow, but slow enough to avoid
excessive loss of amplitude information due to the modulation
of the signal.
AGC Loop
The AGC loop is implemented using a log-linear architecture. It
contains four basic operations: power calculation, error calculation, loop filtering, and gain multiplication.
The AGC can be configured to operate in either desired signal
level mode or desired clipping level mode. The mode is set by
the AGC clipping error bit of the AGC control register. The
AGC adjusts the gain of the incoming data according to how far
it is from a given desired signal level or desired clipping level,
depending on the selected mode of operation.
Two datapaths to the AGC loop are provided: one before the
clipping circuitry and one after the clipping circuitry, as shown
in Figure 39. For the desired signal level mode, only the I/Q
path from before the clipping is used. For the desired clipping
level mode, the difference of the I/Q signals from before and
after the clipping circuitry is used.
Rev. 0 | Page 39 of 72
AD6636
I
Q
22
BITS
I
PROGRAMMABLE
BIT WIDTH
Q
CLIP
GAIN MULTIPLIER
CLIP
USED ONLY FOR
DESIRED CLIPPING
LEVEL MODE
MEAN SQUARE (I2 + Q2)
AVERAGE 1 – 16384 SAMPLES
DECIMATE 1 – 4096 SAMPLES
SQUARE ROOT
log2(x)
2×
POWER OF 2
E ERROR
R DESIRED
04998-0-039
ERROR
THRESHOLD
K × z –1
1 – (1 + P) × z –1 + P × z –2
K1 GAIN
K2 GAIN
P POLE
Figure 39: Block Diagram of the AGC
Desired Signal Level Mode
In this mode of operation, the AGC strives to maintain the
output signal at a programmable set level. The desired signal
level mode is selected by writing Logic 0 into the AGC clipping
error enable bit of the AGC control register. The loop finds the
square (or power) of the incoming complex data signal by
squaring I and Q and adding them.
The AGC loop has an average and decimate block. This average
and decimate operation takes place on power samples and
before the square root operation. This block can be programmed to average from 1 to 16,384 power samples, and the
decimate section can be programmed to update the AGC once
every 1 to 4,096 samples. The limitation on the averaging
operation is that the number of averaged power samples should
be a multiple of the decimation value (1×, 2×, 3×, or 4×).
The averaging and decimation effectively means that the AGC
can operate over averaged power of 1 to 16,384 output samples.
Updating the AGC once every 1 to 4,096 samples and operating
on average power facilitates the implementation of the loop
filter with slow time constants, where the AGC error converges
slowly and makes infrequent gain adjustments. It is also useful
when the user wants to keep the gain scaling constant over a
frame of data or a stream of symbols.
with these operations, attenuation scaling is provided before the
CIC filter.
This scaling operation accounts for the division associated with
the averaging operation as well as the traditional bit growth in
CIC filters. Because this scaling is implemented as a bit-shift
operation, only coarse scaling is possible. Fine scaling is
implemented as an offset in the request level, as explained later
in this section. The attenuation scaling SCIC is programmable
from 0 to 14 using a 4-bit CIC scale word in the AGC average
samples register and is given by
[
(
SCIC = ceil log 2 M CIC × N avg
)]
where:
MCIC is the decimation ratio (1 to 4,096).
NAVG is the number of averaged samples programmed as a
multiple of the decimation ratio (1, 2, 3, or 4).
Due to the limitation that the number of average samples must
be a multiple of the decimation value, only the multiple
numbers 1, 2, 3, or 4 are programmed. This is set using the AGC
average samples word in the AGC average sample register.
These averaged samples are then decimated with decimation
ratios programmable from 1 to 4,096. This decimation ratio is
defined in the 12-bit AGC update decimation register.
For example, if a decimation ratio Mcic is 1,000 and Navg is 3
(decimation of 1,000 and averaging of 3,000 samples), then the
actual gain due to averaging and decimation is 3,000 or 69.54
dB (log2 (3000)). Because attenuation is implemented as a bitshift operation, only multiples of 6.02 dB attenuations are
possible. SCIC in this case is 12, corresponding to 72.24 dB. This
way, SCIC scaling always attenuates more than is sufficient to
compensate for the gain in the average and decimate sections
and, therefore, prevents overflows in the AGC loop. But it is also
evident that the SCIC scaling induces a gain error (the difference
between gain due to CIC and attenuation provided by scaling)
of up to 6.02 dB. This error should be compensated for in the
request signal level, as explained later in this section.
The average and decimate operations are tied together and
implemented using a first-order CIC filter and FIFO registers.
Gain and bit growth are associated with CIC filters and depend
on the decimation ratio. To compensate for the gain associated
A logarithm to the Base 2 is applied to the output from the
average and decimate section. These decimated power samples
are converted to rms signal samples by applying a square root
operation. This square root is implemented using a simple shift
Rev. 0 | Page 40 of 72
AD6636
operation in the logarithmic domain. The rms samples obtained
are subtracted from the request signal level R specified in the
AGC desired level register, leaving an error term to be processed
by the loop filter, G(z).
programmable threshold E, K1 or K2 is used. This allows a fast
loop when the error term is high (large convergence steps
required) and a slower loop function when error term is smaller
(almost converged).
The user sets this programmable request signal level R according to the output signal level that is desired. The request signal
level R is programmable from −0 dB to −23.99 dB in steps of
0.094 dB.
The open-loop gain used in the second-order loop G(z) is given
by one of the following equations:
K = K1, if Error < Error Threshold
K = K2, if Error > Error Threshold
The request signal level should also compensate for errors, if
any, due to the CIC scaling, as explained previously in this
section. Therefore, the request signal level is offset by the
amount of error induced in CIC, given by
The open-loop transfer function for the filter, including the gain
parameter, is
G (z ) =
Offset = 10 × log(MCIC × Navg) − SCIC × 3.01 dB
Kz −1
1 − (1 + P )z −1 + Pz −2
where Offset is in dB.
If the AGC is properly configured in terms of offset in request
level, then there are no gains in the AGC loop except for the
filter gain K. Under these circumstances, a closed-loop
expression for the AGC loop is given by
Continuing the previous example, this offset is given by
Offset = 72.24 − 69.54 = 2.7 dB
So the request signal level is given by
G closed ( z ) =
⎡ (DSL − Offset ) ⎤
R = −ceil ⎢
⎥ × 0.094 dBFS
0.094
⎣
⎦
where:
R is the request signal level.
DSL (desired signal level) is the output signal level that the user
desires.
Therefore, in the previous example, if the desired signal level is
−13.8 dB, the request level R is programmed to be −16.54 dB,
compensating for the offset.
This request signal level is programmed in the 8-bit AGC
desired level register. This register has a floating-point representation, where the 2 MSBs are exponent bits and the 6 LSBs are
mantissa bits. The exponent is in steps of 6.02 dB, and the
mantissa is in steps of 0.094 dB. For example, a value 10’100101
represents 2 × 6.02 + 37 × 0.094 = 15.518 dB.
The AGC provides a programmable second-order loop filter.
The programmable parameters gain 1 (K1), gain 2 (K2), error
threshold E, and pole P completely define the loop filter
characteristics. The error term after subtracting the request
signal level is processed by the loop filter, G(z). The open loop
poles of the second-order loop filter are 1 and P, respectively.
The loop filter parameters, pole P and gain K, allow the
adjustment of the filter time constant that determines the
window for calculating the peak-to-average ratio.
G(z )
Kz −1
=
1 + G ( z ) 1 + (K − 1 − P )z −1 + Pz −2
The gain parameters K1, K2, and pole P are programmable
through AGC loop gain 1, 2, and AGC pole location registers
from 0 to 0.996 in steps of 0.0039 using 8-bit representation. For
example, 1000 1001 represent (137/256 = 0.535156). The error
threshold value is programmable between 0 dB and 96.3 dB in
steps of 0.024 dB. This value is programmed in the 12-bit AGC
error threshold register, using floating-point representation. It
consists of four exponent bits and eight mantissa bits. Exponent
bits are in steps of 6.02 dB and mantissa bits are in steps of
0.024 dB. For example, 0111’10001001 represents 7 × 6.02 + 137
× 0.024 = 45.428 dB.
The user defines the open-loop pole P and gain K, which also
directly impact the placement of the closed-loop poles and filter
characteristics. These closed-loop poles, P1, P2, are the roots of
the denominator of the previous closed-loop transfer function
and are given by
P1 , P2 =
(1 + P − K ) + (1 +P − K ) 2 − 4 P
2
Typically, the AGC loop performance is defined in terms of its
time constant or settling time. In this case, the closed-loop poles
should be set to meet the time constants required by the AGC
loop.
Depending on the value of the error term that is obtained after
subtracting the request signal level from the actual signal level,
either gain value, K1 or K2, is used. If the error is less than the
Rev. 0 | Page 41 of 72
AD6636
The relationship between the time constant and the closed-loop
poles that can be used for this purpose is
⎡
⎤
M CIC
⎥
P1, 2 = exp ⎢
⎢⎣ Sample Rate × τ 1, 2 ⎥⎦
Desired Clipping Level Mode
where τ1, 2 are the time constants corresponding to poles P1, 2.
The time constants can also be derived from settling times as
given by
τ=
2 % settling time
4
or
signal level more slowly compared to no averaging. The same
applies to the manner in which the AGC addresses a sudden
decrease in the signal level.
5 % settling time
3
MCIC (CIC decimation is from 1 to 4,096), and either the settling
time or time constant are chosen by the user. The sample rate is
the sample rate of the stream coming into the AGC. If channels
were interleaved in the output data router, then the combined
sample rate into the AGC should be considered. This rate
should be used in the calculation of poles in the previous
equation, where the sample rate is mentioned.
The loop filter output corresponds to the signal gain that is
updated by the AGC. Because all computation in the loop filter
is done in logarithmic domain (to the Base 2) of the samples,
the signal gain is generated using the exponent (power of 2) of
the loop filter output.
The gain multiplier gives the product of the signal gain with
both the I and Q data entering the AGC section. This signal
gain is applied as a coarse 4-bit scaling and then as a fine scale
8-bit multiplier. Therefore, the applied signal gain is from 0 to
96.3 dB in steps of 0.024 dB. The initial signal gain is programmable using the AGC signal gain register. This register is again a
4 exponent + 8 mantissa bit floating-point representation
similar to the error threshold. This is taken as the initial gain
value before the AGC loop starts operating.
The products of the gain multiplier are the AGC scaled outputs
with a 19-bit representation. These are in turn used as I and Q
for calculating the power, and the AGC error and loop are
filtered to produce the signal gain for the next set of samples.
These AGC scaled outputs can be programmed to have 4-, 5-,
6-, 7-, 8-, 10-, 12-, or 16-bit widths by using the AGC output
word length word in the AGC control register. The AGC scaled
outputs are truncated to the required bit widths by using the
clipping circuitry, as shown in Figure 39.
Each AGC can be configured so that the loop locks onto a
desired clipping level or a desired signal level. Desired clipping
level mode is selected by writing Logic 1 in the AGC clipping
error mode bit in the AGC control register. For signals that tend
to exceed the bounds of the peak-to-average ratio, the desired
clipping level option provides a way to prevent truncating those
signals and still provide an AGC that attacks quickly and settles
to the desired output level. The signal path for this mode of
operation is shown with dotted lines in Figure 39; the operation
is similar to the desired signal level mode.
First, the data from the gain multiplier is truncated to a lower
resolution (4, 5, 6, 7, 8, 10, 12, or 16 bits) as set by the AGC
output word length word in the AGC control register. An error
term (for both I and Q) is generated that is the difference
between the signals before and after truncation. This term is
passed to the complex squared magnitude block, for averaging
and decimating the update samples and taking their square root
to find rms samples as in desired signal level mode. In place of
the request desired signal level, a desired clipping level is
subtracted, leaving an error term to be processed by the secondorder loop filter.
The rest of the loop operates the same way as the desired signal
level mode. This way, the truncation error is calculated and the
AGC loop operates to maintain a constant truncation error
level. The only register setting that is different from the desired
signal level mode settings is that the desired clipping level is
stored in the AGC desired level registers instead of in the
request signal level.
AGC Synchronization
When the AGC output is connected to a RAKE receiver, the
RAKE receiver can synchronize the average and update section
to update the average power for AGC error calculation and loop
filtering. This external sync signal synchronizes the AGC
changes to the RAKE receiver and makes sure that the AGC
gain word does not change over a symbol period, which,
therefore, provides a more accurate estimation. This synchronization can be accomplished by setting the appropriate bits of
the AGC control register.
Sync Select Alternatives
The AGC can receive a sync as follows:
Average Samples Setting
Though it is complicated to express the exact effect of the
number of averaging samples by using equations, intuitively it
has a smoothing effect on the way the AGC loop addresses a
sudden increase or a spike in the signal level. If averaging of
four samples is used, the AGC addresses a sudden increase in
•
Channel sync: The sync signal is used to synchronize the
NCO of the channel under consideration.
•
Pin sync: Select one of the four SYNC pins.
•
Sync now bit: Through the AGC control register.
Rev. 0 | Page 42 of 72
AD6636
When the channel sync select bit of the AGC control register is
Logic 1, the AGC receives the SYNC signal used by the NCO of
the corresponding channel for the start. When this bit is Logic 0,
the pin sync defined by the 2-bit SYNC pin select word in the
AGC control register is used to provide the sync to the AGC.
Apart from these two methods, the AGC control register also
has a sync now bit that can be used to provide a sync to the
AGC by writing to this register through the microport or serial
port.
PARALLEL PORT OUTPUT
Sync Process
Each parallel port can output data from any or all of the AGCs,
using the 1-bit enable bit for each AGC in the parallel port
control register. Even when the AGC is not required for a
certain channel, the AGC can be bypassed, but the data is still
received from the bypassed AGC. The parallel port functionality
is programmable through the two parallel port control registers.
Regardless of how a sync signal is received, the syncing process
is the same. When a sync is received, a start hold-off counter is
loaded with the 16-bit value in the AGC hold-off register, which
initiates the countdown. The countdown is based on the ADC
input clock. When the count reaches 1, a sync is initiated. When
a sync is initiated, the CIC decimation filter dumps the current
value to the square root, error estimation, and loop filter blocks.
After dumping the current value, it starts working toward the
next update value. Additionally on a sync, AGC can be
initialized if the initialize AGC on sync bit is set in the AGC
control register. During initialization, the CIC accumulator is
cleared and new values for CIC decimation, number of
averaging samples, CIC scale, signal gain, open-loop gains K1
and K2, and pole parameter P are loaded from their respective
registers. When the initialize on sync bit is cleared, these
parameters are not loaded from the registers.
This sync process is also initiated when a channel comes out of
sleep by using the start sync to the NCO. An additional feature
is the first sync only bit in the AGC control register. When this
bit is set, only the first sync initiates the process and the
remaining sync signals are ignored. This is useful when syncing
using a pin sync. A sync is required only on the first pulse on
this pin. These additional features make AGC synchronization
more flexible and applicable to varied circumstances.
The AD6636 incorporates three independent 16-bit parallel
ports for output data transfer. The three parallel output ports
share a common clock, PCLK. Each port consists of a 16-bit
data bus, REQuest signal, ACKnowledge signal, three channel
indicator pins, one I/Q indicator pin, one gain word indicator
pin, and a common shared PCLK pin. The parallel ports can be
configured to function in master mode or slave mode. By
default, the parallel ports are in slave mode on power-up.
Each parallel port can be programmed individually to operate
in either interleaved I/Q mode or parallel I/Q mode. The mode
is selected using a 1-bit data format bit in the parallel port
control register. In both modes, the AGC gain word output can
be enabled using a 1-bit append gain bit in the parallel port
control register for individual output ports. There are six enable
bits per output port, one for each AGC in the corresponding
parallel port.
Interleaved I/Q Mode
Parallel port channel mode is selected by writing a 0 to the data
format bit for the parallel port in consideration. In this mode, I
and Q words from the AGC are output on the same 16-bit data
bus on a time-multiplexed basis. The 16-bit I word is output
followed by the 16-bit Q word. The specific AGCs output by the
port are selected by setting individual bits for each of the AGCs
in the parallel port control register. Figure 40 shows the timing
diagram for the interleaved I/Q mode.
PCLKn
PxACK
tDPREQ
PxREQ
tDPP
Px [15:0]
I [15:0]
Q [15:0]
tDPIC
PxIQ
tDPCH
PxCH [2:0] = CHANNEL NO.
LOGIC LOW ‘0’
PxGAIN
Figure 40. Interleaved I/Q Mode without an AGC Gain Word
Rev. 0 | Page 43 of 72
04998-0-040
PxCH [2:0]
AD6636
When an output data sample is available for output from an
AGC, the parallel port initiates the transfer by pulling the
PxREQ signal high. In response, the processor receiving the
data needs to pull the PxACK signal high, acknowledging that it
is ready to receive the signal. In Figure 40, PxACK is already
pulled high and, therefore, the 16-bit I data is output on the data
bus on the next PCLK rising edge after PxREQ is driven logic
high. The PxIQ signal also goes high to indicate that I data is
available on the data bus. The next PCLK cycle brings the
Q data onto the data bus. In this cycle, the PxIQ signal is driven
low. When I data and Q data are output, the channel indicator
pins PxCH[2:0] indicate the data source (AGC number).
port. Therefore, a minimum of three or four PCLK cycles are
required to output one sample of output data on the parallel
port without or with the AGC gain word, respectively.
Parallel IQ Mode
In this mode, eight bits of I data and eight bits of Q data are
output on the data bus simultaneously during one PCLK cycle.
The I byte is the most significant byte of the port, while the Q
byte is the least significant byte. The PAIQ and PBIQ output
indicator pins are set high during the PCLK cycle. Note that if
data from multiple AGCs are output consecutively, the PAIQ
and PBIQ output indicator pins remain high until data from all
channels is output.
Figure 40 is the timing diagram for interleaved I/Q mode with
the AGC gain word disabled. Figure 41 is a similar timing
diagram with the AGC gain word. I and Q data are as explained
for Figure 40. In the PCLK cycle after the Q data, the AGC gain
word is output on the data bus and the PxGAIN signal is pulled
high to indicate that the gain word is available on the parallel
The PACH[2:0] and PBCH[2:0] pins provide a 3-bit binary
value indicating the source (AGC number) of the data currently
being output. Figure 42 is the timing diagram for parallel I/Q
mode.
PCLKn
PxACK
tDPREQ
PxREQ
tDPP
Px [15:0]
I[15:0]
Q[15:0]
GAIN [11:0] +
0000
tDPIQ
PxIQ
tDPCH
PxCH [2:0] = CHANNEL #
tDPGAIN
PxGAIN
Figure 41. Interleaved I/Q Mode with an AGC Gain Word
Rev. 0 | Page 44 of 72
04998-0-041
PxCH [2:0]
AD6636
PCLKn
PxACK
tDPREQ
PxREQ
tDPP
I [15:8]
Q [7:0]
Px [15:0]
tDPIQ
PxIQ
tDPCH
04998-0-042
PxCH [2:0] =
AGC NO.
PxCH [2:0]
LOGIC LOW 0
PxGAIN
Figure 42. Parallel I/Q Mode without an AGC Gain Word
When an output data sample is available for output from an
AGC, the parallel port initiates the transfer by pulling the
PxREQ signal high. In response, the processor receiving the
data needs to pull the PxACK signal high, acknowledging that it
is ready to receive the signal. In Figure 42, the PxACK is already
pulled high and, therefore, the 8-bit I data and 8-bit Q data are
simultaneously output on the data bus on the next PCLK rising
edge after PxREQ is driven logic high. The PxIQ signal also
goes high to indicate that I/Q data is available on the data bus.
When I/Q data is being output, the channel indicator pins
PxCH[2:0] indicate the data source (AGC number).
Figure 42 is the timing diagram for interleaved I/Q mode with
the AGC gain word disabled. Figure 43 is a similar timing
diagram with the AGC gain word enabled. I and Q data are as
shown in Figure 39. In the PCLK cycle after the I/Q data, the
AGC gain word is output on the data bus, and the PxGAIN
signal is pulled high to indicate that the gain word is available
on the parallel port. During this PCLK cycle, the PxIQ signal is
pulled low to indicate that I/Q data is not available on the data
bus. Therefore, in parallel I/Q mode, a minimum of two PCLK
cycles is required to output one sample of output data on the
parallel port without and with the AGC gain word, respectively.
The order of data output is dependent on when data arrives at
the port, which is a function of total decimation rate, DRCF/
CRCF decimation phase, and start hold-off values. Priority
order from highest to lowest is, AGCs 0, 1, 2, 3, 4, and 5 for both
parallel I/Q and interleaved modes of output.
Master/Slave PCLK Modes
The parallel ports can operate in either master or slave mode.
The mode is set via PCLK master mode bit in the Parallel Port
Control 2 register. The parallel ports power up in slave mode to
avoid possible contentions on the PCLK pin.
In master mode, PCLK is an output derived by dividing
PLL_CLK down by the PCLK divisor. The PCLK divisor can
have a value of 1, 2, 4, or 8, depending on the 2-bit PCLK divisor
word setting in the Parallel Port Control 2 register. The highest
PLCK rate in master mode is 200 MHz. Master mode is selected
by setting the PCLK master mode bit in the Parallel Port
Control 2 register.
PCLK rate =
PLL _ CLK rate
PCLK divisor
In slave mode, external circuitry provides the PCLK signal.
Slave-mode PCLK signals can be either synchronous or
asynchronous. The maximum slave mode PCLK frequency is
also 200 MHz.
Rev. 0 | Page 45 of 72
AD6636
PCLK
PxACK
tDPREQ
PxREQ
tDPP
I [15:8]
Q [15:8]
Px [15:0]
GAIN [11:0] +
0000
tDPIQ
PxIQ
tDPCH
PxCH [2:0]
tDPGAIN
PxGAIN
04998-0-043
PxCH [2:0] = CHANNEL #
Figure 43. Parallel I/Q Mode with an AGC Gain Word
Parallel Port Pin Functions
Table 24 describes the functions of the pins used by the parallel ports.
Table 24. Parallel Port Pin Functions
Pin Name
PCLK
I/O
I/O
PAREQ, PBREQ,
PCREQ
O
PAACK, PBACK,
PCACK
I
PAIQ, PBIQ,
PCIQ
PAGAIN,
PBGAIN,
PCGAIN
PACH[2:0],
PBCH[2:0],
PCCH[2:0]
PADATA[15:0],
PBDATA[15:0],
PCDATA[15:0]
Function
PCLK can operate as a master or as a slave. This setting is dependent on the 1-bit PCLK master mode bit in the
Parallel Port Control 2 register. As an output (master mode), the maximum frequency is CLK/N, where CLK is
AD6636 clock and N is an integer divisor of 1, 2, 4, or 8. As an input (slave mode), it can be asynchronous or
synchronous relative to the AD6636 CLK. This pin powers up as an input to avoid possible contentions. Parallel
port output pins change on the rising edge of PCLK.
Active high output. Synchronous to PCLK. A logic high on this pin indicates that data is available to be shifted out
of the port. When an acknowledge signal is received, data starts shifting out and this pin remains high until all
pending data has been shifted out.
Active high asynchronous input. Applying a logic low on this pin inhibits parallel port data shifting. Applying a
logic high to this pin when REQ is high causes the parallel port to shift out data according to the programmed
data mode.
ACK is sampled on the rising edge of PCLK. Assuming that REQ is asserted, the latency from the assertion of ACK
to data appearing at the parallel port output is no more than 1.5 PCLK cycles. ACK can be held high continuously;
in this case, when data becomes available, shifting begins 1 PCLK cycle after the assertion of REQ (see Figure 40,
Figure 41, Figure 42, and Figure 43).
High whenever I data is present on the parallel port data bus; otherwise low. In parallel I/Q mode, both I data and
Q data are available at the same time and, therefore, the PxIQ signal is pulled high.
High whenever the AGC gain word is present on the parallel port data bus; otherwise low.
These pins identify data in both of the parallel port modes. The 3-bit value identifies the source of the data (AGC
number) on the parallel port when it is being shifted out.
Parallel output port data bus. Output format is twos complement. In parallel I/Q mode, 8-bit data is present; in
interleaved I/Q mode, 16-bit data is available.
Rev. 0 | Page 46 of 72
AD6636
USER-CONFIGURABLE BUILT-IN SELF-TEST (BIST)
Start with Soft Sync
Each channel of AD6636 includes a BIST block. The BIST, along
with an internal test signal (pseudorandom test input signal),
can be used to generate a signature. This signature can be
compared with a known good device and an untested device to
see if the untested device is functional.
The AD6636 can synchronize channels or chips under microprocessor control. The start hold-off counter, in conjunction
with the soft start enable bit and the channel enable bits, enables
this synchronization.
BIST timer bits in the BIST control register can be programmed
with a timer value that determines the number of clock cycles
that the output of the channels (output of AGC) have
accumulated. When the disable signature generation bit is
written with Logic 0, the BIST timer is counted down and a
signature register is written with the accumulated output of the
AD6636 channel.
When the BIST timer expires, the signature register for I and Q
paths can be read back to compare it with the signature register
from a known good device.
To synchronize the start of multiple channels via microprocessor control:
1.
Write the channel enable register to enable one or more
channels, if the channels are inactive.
2.
Write the NCO start hold-off counter(s) to the appropriate
value (greater than 1 and less than 216).
3.
Write the soft sync channel enable bit(s) and soft start
synchronization enable bit high in the soft synchronization
configuration register. This starts the countdown by the
start hold-off counter. When the count reaches 1, the
channels are activated or resynchronized.
CHIP SYNCHRONIZATION
The AD6636 offers two types of synchronization: start sync and
hop sync. Start sync is used to bring individual channels out of
sleep after programming. It can also be used while AD6636 is
operational to resynchronize the internal clocks. Hop sync is
used to change or update the NCO frequency tuning word and
the NCO phase offset word.
Two methods can be used to initiate a start sync or hop sync:
•
•
Soft sync is provided by the memory map registers and is
applied to channels directly through the microport or serial
port interface.
Pin sync is provided using four hard-wired SYNC[3:0] pins.
Each channel is programmed to listen to one of these SYNC
pins and do a start sync or a hop sync when a signal is
received on these pins.
The pin synchronization configuration register (Address 0x04)
is used to make pin synchronization even more flexible. The
part can be programmed to be edge-sensitive or level-sensitive
for SYNC pins. In edge-sensitive mode, a rising edge on the
SYNC pins is recognized as a synchronization event.
Start
Start refers to the startup of an individual channel or chip, or of
multiple chips. If a channel is not used, it should be put into
sleep mode to reduce power dissipation. Following a hard reset
(low pulse on the RESET pin), all channels are placed into sleep
mode. Alternatively, channels can be put to sleep manually by
writing 0 to the sleep register.
Start with Pin Sync
Four sync pins (0, 1, 2, and 3) provide very accurate synchronization among channels. Each channel can be programmed to
monitor any of the four sync pins.
To start the channels with a pin sync:
1.
Write the channel register to enable one more channels, if
the channels are inactive.
2.
Write the NCO start hold-off counter(s) to the appropriate
value (greater than 1 and less than 216 − 1).
3.
Program the channel NCO control registers to monitor the
appropriate SYNC pins.
4.
Write the start synchronization enable bit and SYNC pin
enable bits high in the pin synchronization configuration
register. This starts the countdown of the start hold-off
counter. When the count reaches 1, the channels are
activated or resynchronized.
Hop
Hop is a jump from one NCO frequency and/or phase offset to
a new NCO frequency and/or phase offset. This change in
frequency and/or phase offset can be synchronized via
microprocessor control (soft sync) or via an external sync signal
(pin sync).
Rev. 0 | Page 47 of 72
AD6636
Hop with Soft Sync
The AD6636 can synchronize a change in NCO frequency
and/or phase offset of multiple channels or chips under
microprocessor control. The NCO hop hold-off counter, in
conjunction with the soft hop enable bit and the channel enable
bits, enables this synchronization.
To synchronize the hop of multiple channels via microprocessor
control:
1.
Write the NCO frequency register(s) or phase offset
register(s) to the new value.
2.
Write the NCO frequency hold-off counter(s) to the
appropriate value (greater than 1 and less than 2^16).
3.
Write the soft hop synchronization enable bit and the
corresponding soft sync channel enable bits high in the soft
synchronization configuration register. This starts the
countdown by the frequency hold-off counter. When the
count reaches 1, the new frequency and/or phase offset is
loaded into the NCO.
Hop with Pin Sync
Four sync pins (0, 1, 2 and 3) provide very accurate synchronization among channels. Each channel can be programmed to
look at any of the four sync pins.
To control the hop of channel NCO frequencies:
1.
Write the NCO frequency register(s) or phase offset
register(s) to the new value.
2.
Write the NCO frequency hold-off counter(s) to the
appropriate value (greater than 1 and less than 216).
3.
Program the channel NCO control registers to monitor the
appropriate SYNC pins.
4.
Write the hop synchronization enable bit and SYNC pin
enable bits high in the pin synchronization configuration
register. This enables the countdown of the frequency
hold-off counter. When the reaches 1, the new frequency
and/or phase offset is loaded into the NCO.
SERIAL PORT CONTROL
The AD6636 serial port allows the programming and readback
of all control registers and coefficient memory, serially, in 1-byte
words. The serial port can work in two modes, selected using
the MODE pin (SPI = 0, SPORT = 1). In both SPI and SPORT
modes, the AD6636 is compatible with Blackfin, TigerSHARC,
and other DSPs. The serial port and microport share some of
the I/O pins; therefore, only one of these ports is operational at
a time.
The selection between the serial port and microport modes is
made using the SMODE pin (serial port = 1, microport = 0).
The serial port has a chip select pin (active low signal), which
should be pulled low for any operation on the serial port. Serial
data can be shifted into the part or out of the part as either MSB
first or LSB first using the MSBFIRST pin (1 = MSB first, 0 =
LSB first).
Hardware Interface
The pins listed in Table 25 comprise the physical interface
between the user’s programming device and the AD6636 serial
port. All serial pins are inputs except for SDO, which is an opendrain output and should be pulled high by an external pull-up
resistor (typical value of 1 kΩ).
Table 25. Serial Port Pin Names and Functions
Pin Name
SCLK
MSBFIRST
STFS
SRFS
SDI
SDO
SCS
SMODE
MODE
Function
Serial clock in both SPI and SPORT modes. Serial data is clocked in on the rising edge of SCLK.
Indicates whether the first bit shifted in or out of the serial port is the MSB (1) or LSB (0) of the data word.
Serial transmit frame sync in SPORT mode; ignored in SPI mode.
Serial receive frame sync in SPORT mode; ignored in SPI mode.
Serial data input in both modes.
Serial data output in both modes.
Active low serial chip select in both modes. Setting this pin high holds the serial port in reset. It should be pulled low for
any read/write operation on serial port.
Serial mode. Partis programmed through the serial port when this pin is Logic 1.
Mode pin. Selects between SPI (0) and SPORT (1) modes.
Rev. 0 | Page 48 of 72
AD6636
SPI Mode Write Operation
If the example is for MSBFIRST = 1, then the instruction words
are 0x07 (Address 7) and 0x07 (number of addresses to write).
The data corresponds to Addresses 0x07 to 0x01, in that order.
The instruction words and data are MSB first.
In SPI mode, the SCLK runs only when data is being transferred, so no external framing is necessary. The SPI serial mode
supports slave operations only. Input data on SDI pin is
registered on the rising edge of SCLK and, therefore, the DSP or
master device should be set to change data on the falling edge of
SCLK. All input and output transfers take place in 8-bit
transactions.
SPI Mode Read Operation
Data on the SDO pin is shifted out on the positive edge of
SCLK. Therefore, the DSP or other master device should
register data on the falling edge of SCLK. All input and output
transfers take place in 8-bit transactions. The SDO pin is high
impedance when data is not being output.
For a write operation, the user must write two 8-bit instruction
words to the serial port to instruct the AD6636 internal control
logic about the data to be written. The first instruction word is
an address location. If the MSBFIRST pin is Logic 1, this
address is the ending address; if it is Logic 0, this address
corresponds to the starting address. The second instruction
word contains a 1-bit read/write indicator (MSB bit: 1 = read,
0 = write), followed by a 7-bit field to indicate the number of
address locations to write (N).
Each read cycle consists of SCS going low, eight clock cycles
generated on SCLK pin, followed by SCS pulled high. Data
corresponding to the addresses to be read is transferred out on
the SDO pin and is registered by the master device on the
falling edge. The data is MSB first or LSB first based on the
status of MSBFIRST pin.
For example, consider reading Addresses 0x01 to 0x07 of the
AD6636 register map, when operating in SPI mode and
MSBFIRST = 0. The instruction words are Addresses 0x01 and
0x87 (MSB = 1 for read). The following seven read cycles
transfer one byte at a time, sequentially out of Addresses 0x01 to
0x07, in that order. The instruction words should be written
LSB first, and data comes out on the SDO with the LSB first.
Following the instruction words are the N write operations
(each one byte long), where N is the number of address
locations to write. After each write cycle, the internal address is
incremented (MSBFIRST = 0) or decremented (MSBFIRST =
1). In this case, MSBFIRST indicates the first bit coming out of
or into the SPI port as well as which byte is written first (most
significant byte of the N-byte transfer).
If the example is for MSBFIRST = 1, then the instruction words
are 0x07 (Address 7) and 0x87 (MSB = 1 for read, followed by
the number of address locations to read). The data coming out
on SDO corresponds to Addresses 0x07 to 0x01, in that order.
The instruction words are written MSB first, and data comes
out on the SDO with MSB first.
For example, consider writing Addresses 0x01 to 0x07 of
AD6636 register map, when operating in SPI mode and
MSBFIRST = 0. The instruction words are Addresses 0x01 and
0x07 (MSB = 0 for write). The following seven write cycles
transfer one byte at a time sequentially into Addresses 0x01 to
0x07, in that order. The instruction words and data should be
written with LSB first.
SCLK
tSSCS
tHSCS
SCS
SMODE
LOGIC 1
tHSI
D0
D1
D2
D3
D4
D5
D6
D7
LOGIC 0
MODE
Figure 44. SPI Write to the AD6636 Serial Port and Transfer of 1-Byte Data to Internal Registers
Rev. 0 | Page 49 of 72
04998-0-044
tSSI
SDI
AD6636
SCLK
SCS
SMODE
SDI
INSTRUCTION BYTE 1
INSTRUCTION BYTE 2
SDO
04998-0-045
VALID OUTPUT
MODE
Figure 45. SPI Readback Timing
SPORT Mode Write Operation
In SPORT mode, the SCLK runs continuously, and external
SRFS and STFS signals are used for framing of the input and
out words. Incoming framing signals SRFS (receive/input) and
STFS (transmit/output) are valid when they are high for one
SCLK cycle. All input and output data must be transmitted or
received in 8-bit words by using the appropriate framing signals.
During a write cycle, the data is registered on the rising edge of
SCLK. Therefore, the programming device outputs data on the
falling edge of the SCLK. The SCS pin is low for both read and
write cycles.
For a write operation, the user must write two 8-bit instruction
words to the SPORT to instruct the AD6636 internal control
logic about the data to be written. The first instruction word is
an address location. If MSBFIRST is Logic 1, this address is the
ending address; if MSBFIRST is Logic 0, this address is the
starting address. The second instruction word contains a 1-bit
read/write indicator (MSB bit: 1 = read, 0 = write), followed by a
7-bit field to indicate the number of address locations to write
(N). Each write cycle takes nine clock cycles, with SRFS high on
the first clock cycle and the 8-bit instruction word on the next
eight clock cycles.
Following the instruction words is N write operations (each one
byte long), where N is the number of address locations to write.
Each write operation must include SRFS high for one clock
cycle and the 8-bit data. After each write cycle, the internal
address is incremented (MSBFIRST = 0) or decremented
(MSBFIRST = 1). In this case, MSBFIRST indicates the first bit
coming out of or into the SPORT, as well as the byte that is
written first (most significant byte of the N-byte transfer, when
MSBFIRST = 1).
For example, consider writing Addresses 0x01 to 0x07 of
AD6636 register map, when operating in SPORT mode and
MSBFIRST = 0. The instruction words are Addresses 0x01 and
0x07 (MSB = 0 for write).
The following seven write cycles transfer one byte at a time,
sequentially into Addresses 0x01 to 0x07, in that order. The
instruction words and data are written with the LSB first.
If the example is for MSBFIRST = 1, then the instruction words
are 0x07 (Address 7) and 0x07 (the number of addresses to
write). The data corresponds to Addresses 0x07 through to
0x01, in that order. The instruction words and data are
MSB first.
SPORT Mode Read Operation
Data on the SDO pin is shifted out on the positive edge of
SCLK. Therefore, the DSP or other master device registers data
on the falling edge of SCLK. All input and output transfers take
place in 8-bit transactions. The SDO pin is high impedance
when data is not being output.
A read operation is similar to a write operation in its format.
The first two instruction words are written on the SDI pin, the
only difference being that the MSB bit of the second instruction
word is that Logic 1 indicates a read operation. After the
instruction words are written, the master device initiates N read
cycles. Each read cycle consists of an STFS framing signal valid
for one clock cycle and the 8-bit data coming out on the SDO
pin. The SCS pin must be low during the read cycle. Data
corresponding to the addresses to be read is transferred out on
the SDO pin and should be registered by the master device. The
data is MSB first or LSB first, based on the status of the
MSBFIRST pin.
For example, consider reading Addresses 0x01 to 0x07 of
AD6636 register map, when operating in SPORT mode and
MSBFIRST = 0. The instruction words are 0x01 and 0x87
(MSB = 1 for read). The following seven read cycles transfer one
byte at time, sequentially out of Addresses 0x01 to 0x07, in that
order. The instruction words are written LSB first and the data
comes out on the SDO with the LSB first.
Rev. 0 | Page 50 of 72
AD6636
SCLK
SCS
SMODE
SRFS
SDI
D1
D2
D3
D4
D5
D6
D7
04998-0-046
D0
MODE
Figure 46. SPORT Serial Write
SCLK
SCS
SMODE
SRFS
SDI
INSTRUCTION BYTE 1
INSTRUCTION BYTE 2
STFS
SDO
04998-0-047
VALID OUTPUT
MODE
Figure 47. SPORT Serial Readback
BLACKFIN
(MASTER)
Connecting the AD6636 Serial Port to a Blackfin DSP
SCK
SCLK
SPISS
SRFS
AND
STFS
MOSI
MISO
SDI
SDO
SCS
PF2
PROGRAMMABLE FLAG
MODE
In SPI mode, the Blackfin DSP must act as a master to the
AD6636 by providing the SCLK. SDO is an open-drain output,
so that multiple slave devices can be connected together.
Figure 48 shows typical interconnections.
VDDIO
GND
AD6636
(SLAVE)
SMODE
Figure 48. SPI Mode Serial Port Connections to Blackfin DSP
In SPORT mode, the Blackfin provides the SCLK, SRFS, and
STFS signals, as shown in Figure 49.
Rev. 0 | Page 51 of 72
04998-0-048
If the example is for MSBFIRST = 1, then the instruction words
are 0x07 (Address 7) and 0x87 (MSB = 1 for read, followed by
the number of address locations to read). The data coming out
on the SDO corresponds to Addresses 0x07 to 0x01, in that
order. The instruction words are written MSB first and data
comes out on the SDO with the MSB first.
AD6636
BLACKFIN
SCK
SCLK
TFS
SRFS
RFS
STFS
DT
DR
Intel (INM) Mode
VDDIO
SDI
SDO
SCS
GND
AD6636
MODE
SMODE
04998-0-049
PF2
PROGRAMMABLE FLAG
Figure 49. SPORT Mode Serial Port Connections to Blackfin DSP
MICROPORT
The microport on the AD6636 can be used for programming
the part, reading register values, and reading output data (I, Q,
and RSSI words).
Note that, at any given point in time, either the microport or the
serial port can be active, but not both. Some of the balls on the
package are shared between the microport and the serial port
and have dual functionality based on the SMODE pin. The
microport is selected by pulling the SMODE pin low (ground).
Both read and write operations can be performed using the
microport. The direct addressing scheme is used and any
internal register can be accessed using an 8-bit address. The
data bus can be either 8-bit or 16-bit as set by the chip I/O
access control register. Microport operation is synchronous to
CPUCLK, which must be supplied external to the AD6636 part.
CPUCLK should be less than CLKA and 100 MHz.
The microport can operate in Intel mode (separate read and
write strobes) or in Motorola mode (single read/write strobe).
The MODE pin is used to select between Intel (INM, MODE =
0) and Motorola (MNM, MODE = 1) modes. Some AD6636
pins have dual functionality based on the MODE pin. Table 26
lists the pin functions for both modes.
The programming port performs synchronous Intel-style reads
and writes on the positive edge of the CPUCLK input when
RESET is inactive (active low signal). The CPUCLK pin is
driven by the programming device (CPUCLK of DSP or
FPGA). During a write access, the A[7:0] address bus provides
the address for access, and the D[15:0] bus (D[7:0] if the 8-bit
data bus is used) is driven by the programming device. The data
bus is driven by the AD6636 during a read operation. Intel
mode uses separate read (RD) and write (WR) active-low data
strobes to indicate both the type of access and the valid data for
that access.
The chip select (CS) is an active-low input that signals when an
access is active on its programming port pins. During an access,
the AD6636 drives RDY low to indicate that it is performing the
access. When the internal read or write access is complete, the
RDY pin pulled high. Because the RDY pin is an open-drain
output with a weak internal pull-up resistor (70 kΩ), an external
pull-up resistor is recommended (see Figure 50). Figure 13 and
Figure 14 are the timing diagrams for read and write cycles
using the microport in INM mode.
For an asynchronous write operation in Intel (INM) mode, the
CPUCLK should be running. Set up the data and address buses.
Pull the WR signal low and then pull the CS signal low. The
RDY goes low to indicate that the access is taking place
internally. When RDY goes high, the write cycle is complete and
CS can be pulled high to disable the microport.
For an asynchronous read operation on the Intel mode
microport, set up the address bus and three-state the data bus.
Pull the RD signal low and then pull the CS signal low. The
RDY goes low to indicate an internal access. When RDY goes
low, valid data is available on the data bus for read.
Table 26. Microport Programming Pins
Motorola (MNM) Mode
Pin Name
RESET
SMODE
MODE
A[7:0]
D[15:0]
R/W (WR)
DS (RD)
DTACK (RDY)
CS
The programming port performs synchronous Motorola-style
reads and writes on the positive edge of CPUCLK when RESET
is inactive (active low signal). The A[7:0] bus provides the
address to access and the D[15:0] bus (D[7:0], if the 8-bit data
bus is used) is externally driven with data during a write (driven
by the AD6636 during a read). Motorola mode uses the R/W
line to indicate the type of access (Logic 1 = read, Logic 0 =
write), and the active-low data strobe (DS) signal is used to
indicate valid data.
Intel Mode
RESET
Logic 0
Logic 0
A[7:0]
D[15:0]
WR
RD
RDY
CS
Motorola Mode
RESET
Logic 0
Logic 1
A[7:0]
D[15:0]
R/W
DS
DTACK
CS
Rev. 0 | Page 52 of 72
AD6636
The chip select (CS) is an active-low input that signals when an
access is active on its programming port pins. When the
read/write cycle is complete, the AD6636 drives DTACK low.
The DTACK signal goes high again after either the CS or DS
signal is driven high. Because the DTACK pin is an open-drain
output with a weak internal pull-up resistor (70 kΩ), an external
pull-up resistor is recommended (see Figure 50). Figure 15 and
Figure 16 are the timing diagrams for read and write cycles
using the microport in MNM mode.
For an asynchronous write operation on the Motorola mode
microport, the CPUCLK should be running. Set up the data and
address buses. Pull the R/W and DS signals low and then pull
the CS signal low. The DTACK goes low after a few clock cycles
to indicate that the write access is complete and that CS can be
pulled high to disable the microport. For an asynchronous read
operation on the Motorola mode microport, set up the address
bus and three-state the data bus. Pull the RD signal low and
then pull the CS signal low. The DTACK goes low after a few
clock cycles to indicate that valid data is on the data bus.
Accessing Multiple AD6636 Devices
JTAG BOUNDARY SCAN
The AD6636 supports a subset of the IEEE Standard 1149.1
specification. For details of the standard, see the IEEE Standard
Test Access Port and Boundary-Scan Architecture, an IEEE-1149
publication.
The AD6636 has five pins associated with the JTAG interface.
These pins, listed in Table 27, are used to access the on-chip test
access port. All input JTAG pins are pull-up except for TCLK,
which is pull-down.
Table 27. Boundary Scan Test Pins
Name
TRST
TCLK
TMS
TDI
TDO
Description
Test Access Port Reset
Test Clock
Test Access Port Mode Select
Test Data Input
Test Data Output
The AD6636 supports three op codes, listed in Table 28. These
instructions set the mode of the JTAG interface.
Table 28. Boundary Scan Op Codes
If multiple AD6636 devices are on a single board, the microport
pins for these devices can be shared. In this configuration, a
single programming device (DSP, FPGA, or microcontroller)
can program all AD6636 devices connected to it.
Instruction
BYPASS
SAMPLE/PRELOAD
EXTEST
Each AD6636 has four CHIPID pins that can be connected in
16 different ways. During a write/read access, the internal
circuitry checks to see if the CHIPID bits in the chip I/O access
control register (Address 0x02) are the same as the logic levels
of the CHIPID pins (hardwired to the part). If the CHIPID bits
and the CHIPID pins have the same value, then a write/read
access is completed; otherwise, the access is ignored.
A BSDL file for this device is available. Contact Analog Devices
Inc. for more information.
To program multiple devices using the same microport control
and data buses, the devices should have separate CHIPID pin
configurations. A write/read access can be made only on the
intended chip; all other chips would ignore the access.
Op Code
11
01
00
EXTEST (2'b00)
Places the IC into an external boundary-test mode and selects
the boundary-scan register to be connected between TDI and
TDO. During this operation, the boundary-scan register is
accessed to drive-test data off-chip via boundary outputs and
receive test data off-chip from boundary inputs.
SAMPLE/PRELOAD (2'b01)
Allows the IC to remain in normal functional mode and selects
the boundary-scan register to be connected between TDI and
TDO. The boundary-scan register can be accessed by a scan
operation to take a sample of the functional data entering and
leaving the IC. Also, test data can be preloaded into the
boundary scan register before an EXTEST instruction.
BYPASS (2'b11)
Allows the IC to remain in normal functional mode and selects
a 1-bit bypass register between TDI and TDO. During this
instruction, serial data is transferred from TDI to TDO without
affecting operation of the IC.
Rev. 0 | Page 53 of 72
AD6636
MEMORY MAP
READING THE MEMORY MAP TABLE
Open Locations
Each row in the memory map table has four address locations.
The memory map is roughly divided into four regions: global
register map (Addresses 0x00 to 0x0B), input port register map
(Addresses 0x0C to 0x67), channel register map (Addresses
0x68 to 0xBB), and output port register map (Addresses 0xBC
to 0xE7). The channel register map is shared by all six channels,
and access to individual channels is given by the channel I/O
access control register (Address 0x02).
All locations marked as open are currently not used. When
required, these locations should be written with 0s. Writing to
these locations is required only when part of an address
location is open (for example, Address 0x78). If the whole
address location is open (for example, Address 0x00), then this
address location does not need to be written. If the open
locations are readback using the microport or serial port, the
readback value is undefined (each bit can be independently 1 or
0), and these bits have no significance.
In the memory map, Table 29, the addresses are given in the
right column. The column with the heading Byte 0 has the
address given in the right column. The column Byte 1 has the
address given by 1 more than the address listed in the right
column (address offset of 1). Similarly, the address offset for the
Byte 2 column is 2, and for the Byte 3 column is 3. For example,
the second row lists 0x04 as the address in the right column.
The pin synchronization configuration register has Address
0x04, the soft synchronization configuration register has
Address 0x05, and the LVDS control register lists Addresses
0x07 and 0x06.
Bit Format
All registers are in little-endian format. For example, if a register
takes 24 bits or three address locations, then the most
significant byte is at the highest address location and the least
significant byte is at a lowest address location. In all registers,
the least significant bit is Bit 0 and the most significant bit is
Bit 7. For example, the NCO frequency <31:0> register is
32 bits wide. Bit 0 (LSB) of this register is written at Bit 0 of
Address 0x70 and Bit 32 (MSB) of this register is written at
Bit 7 of Address 0x73.
When referring to a register that takes up multiple address
locations, it is referred to by the address location of the most
significant byte of the register. For example, the text reads, “Port
A dwell timer at Address 0x2A.” Note that only the four most
significant bits of this register are at this location, and this
register also takes up Addresses 0x29 and 0x28.
If an address location has more than one register or has one
register with some open bits, then the order of these registers is
as given in the table. For example, Address 0x33 reads
Open <7:5>, Port A Signal Monitor <4:0>.
The open <7:5> is located at Bits <7:5> and the Port A signal
monitor <4:0> is located at Bits <4:0>.
Another example is Address 0x35:
Open <15:10>, Port A Upper Threshold <9:0>
Here, Bits <7:2> of Address 0x35 are open <15:10>. Bits <1:0>
of Address 0x35 and Bits <7:0> of Address 0x34 make up the
Port A upper threshold <9:0> register (Bit 1 of Address 0x35 is
the MSB of the Port A upper threshold register).
Default Values
On coming out of reset, some of the address locations (but not
all) are loaded with default values. When available, the default
values for the registers are given in the table. If the default value
is not listed, then these address locations are in an undefined
state (Logic 0 or Logic 1) on RESET.
Logic Levels
In the explanation of various registers, “bit is set” is synonymous
with “bit is set to Logic 1” or “writing Logic 1 for the bit.”
Similarly “clear a bit” is synonymous with “bit is set to Logic 0”
or “writing Logic 0 for the bit.”
Rev. 0 | Page 54 of 72
AD6636
Table 29. Memory Map
8-Bit Hex
Address
0x03
0x07
Byte 3
Byte 2
Open <7:6>, Channel
Open<7:6>, Channel I/O
Enable <5:0>
Access Control<5:0>
Open <15:11>, LVDS Control<10:0> (default 0x06FC)
Byte 1
Byte 0
Open<7:0>
Chip I/O Access Control
<7:0> (default 0x00)
Soft Synchronization
Pin Synchronization
Configuration<7:0>
Configuration<7:0>
Interrupt Status <15:0> (read only, default 0x00)
0x0B
Interrupt Mask <15:0>
ADC Input Port Register Map—Addresses 0x0C to 0x67
0x0F
ADC Input Control <31:0>
0x13
Open<15:0>
ADC CLK Control <15:0> (default 0x0000)
0x17
Port AB, IQ Correction Control<15:0> (default 0x0000)
Port CD, IQ Correction Control <15:0> (default 0x0000)
0x1B
Port AB, DC Offset Correction I<15:0>
Port AB, DC Offset Correction Q<15:0>
0x1F
Port CD, DC Offset Correction I<15:0>
Port CD, DC Offset Correction Q<15:0>
0x23
Port AB, Phase Offset Correction <15:0>
Port AB, Amplitude Offset Correction <15:0>
0x27
Port CD, Phase Offset Correction <15:0>
Port CD, Amplitude Offset Correction <15:0>
0x2B
Port A Gain Control <7:0> Open<23:20>, Port A Dwell Timer <19:0>
0x2F
Open<7:0>
Port A Power Monitor Period <23:0>
0x33
Port A Power Monitor Output <23:0>
Open<7:5>, Port A Signal
Monitor<4:0>
0x37
Open<15:10>, Port A Lower Threshold <9:0>
Open <15:10>, Port A Upper Threshold <9:0>
0x3B
Port B Gain Control <7:0> Open<23:20>, Port B Dwell Timer <19:0>
0x3F
Open<7:0>
Port B Power Monitor Period <23:0>
0x43
Port B Power Monitor Output <23:0>
Open<7:5>, Port B Signal
Monitor<4:0>
0x47
Open<15:10>, Port B Lower Threshold <9:0>
Open <15:10>, Port B Upper Threshold <9:0>
0x4B
Port C Gain Control <7:0> Open<23:20>, Port C Dwell Timer <19:0>
0x4F
Open<7:0>
Port C Power Monitor Period <23:0>
0x53
Port C Power Monitor Output <23:0>
Open<7:5>, Port C Signal
Monitor<4:0>
0x57
Open<15:10>, Port C Lower Threshold <9:0>
Open <15:10>, Port C Upper Threshold <9:0>
0x5B
Port D Gain Control <7:0> Open<23:20>, Port D Dwell Timer <19:0>
0x5F
Open<7:0>
Port D Power Monitor Period <23:0>
0x63
Open<7:5>, Port D Signal Port D Power Monitor Output <23:0>
Monitor<4:0>
0x67
Open<15:10>, Port D Lower Threshold <9:0>
Open <15:10>, Port D Upper Threshold <9:0>
Channel Register Map—Addresses 0x68 to 0xBB
0x6B
Open<15:0>
Open<15:9>, NCO Control<8:0>
0x6F
NCO Start Hold-Off Counter<15:0>
NCO Frequency Hold-Off Counter<15:0>
0x73
NCO Frequency <31:0> (default 0x0000 0000)
0x77
Open<15:0>
NCO Phase Offset<15:0> (default 0x0000)
0x7B
Open<7:1>, CIC
Open<7:5>, CIC
Open<7:5>, CIC Scale
Open<7:4>, FIR-HB
Bypass<0>
Decimation<4:0>
Factor<4:0>
Control<3:0>
0x7F
Open<15:0>
Open<15:13>, MRCF Control<12:0>
0x83
Open<7:6>, MRCF
Open<7:6>, MRCF
Open<7:6>, MRCF
Open<7:6>, MRCF
Coefficient 3 <5:0>
Coefficient 2 <5:0>
Coefficient 1 <5:0>
Coefficient 0 <5:0>
0x87
Open<7:6>, MRCF
Open<7:6>, MRCF
Open<7:6>, MRCF
Open<7:6>, MRCF
Coefficient 7 <5:0>
Coefficient 6 <5:0>
Coefficient 5 <5:0>
Coefficient 4 <5:0>
0x8B
Open<15:13>, DRCF Control Register<12:0>
Open <7:6>, DRCF
Open<7>, DRCF Taps
Coefficient Offset<5:0>
<6:0>
0x8F
Open<15:0>
Open <7:6>, DRCF Final
Open <7:6>, DRCF Start
Address<5:0>
Address<5:0>
0x93
Open<15:0>
Open<15:14>, DRCF Coefficient Memory <13:0>
Rev. 0 | Page 55 of 72
8-Bit Hex
Address
0x00
0x04
0x08
0x0C
0x10
0x14
0x18
0x1B
0x20
0x24
0x28
0x2C
0x30
0x34
0x38
0x3C
0x40
0x44
0x48
0x4C
0x50
0x54
0x58
0x5C
0x60
0x64
0x68
0x6C
0x70
0x74
0x78
0x7C
0x80
0x84
0x88
0x8C
0x90
AD6636
8-Bit Hex
Address
0x97
Byte 3
Byte 2
Open<15:13>, CRCF Control Register<12:0>
Byte 1
Byte 0
Open <7:6>, CRCF
Open<7>, CRCF Taps
Coefficient Offset<5:0>
<6:0>
0x9B
Open<15:0>
Open <7:6>, CRCF
Open <7:6>, CRCF Start
Final Address<5:0>
Address<5:0>
0x9F
Open<7:0>
Open<23:20>, CRCF Coefficient Memory <19:0>
0xA3
Open<15:11>, AGC Control Register<10:0>
AGC Hold-Off Register<15:0>
0xA7
Open<15:12>, AGC Update Decimation<11:0>
Open<15:12>, AGC Signal Gain <11:0>
0xAB
Open<15:12>, AGC Error Threshold <11:0>
Open<7:6>, AGC Average
AGC Pole Location
Samples<5:0>
<7:0>
0xAF
Open<7:0>
AGC Desired Level<7:0>
AGC Loop Gain2 <7:0>
AGC Loop Gain1 <7:0>
0xB3
Open<7:0>
BIST I Path Signature Register<23:0> (read-only, default 0xAD6636)
0xB7
Open<7:0>
BIST Q Path Signature Register<23:0> (read-only, default 0xAD6636)
0xBB
Open <15:0>
BIST Control <15:0>
Output Port Register Map—Addresses 0xBC to 0xE7
0xBF
Open<7:0>
Parallel Port Output Control <23:0>
0xC3
Open<15:0>
Open<15:10>, Output Port Control <9:0>
0xC7
AGC0, I Output<15:0> (read only)
AGC0, Q Output <15:0> (read only)
0xCB
AGC1, I Output <15:0> (read only)
AGC1, Q Output <15:0> (read only)
0xCF
AGC2, I Output <15:0> (read only)
AGC2, Q Output <15:0> (read only)
0xD3
AGC3, I Output <15:0> (read only)
AGC3, Q Output <15:0> (read only)
0xD7
AGC4, I Output <15:0> (read only)
AGC4, Q Output <15:0> (read only)
0xDB
AGC5, I Output <15:0> (read only)
AGC5, Q Output <15:0> (read only)
0xDF
Open<15:12>, AGC0 RSSI Output<11:0> (read only)
Open<15:12>, AGC1 RSSI Output<11:0> (read only)
0xE3
Open<15:12>, AGC2 RSSI Output<11:0> (read only)
Open<15:12>, AGC3 RSSI Output<11:0> (read only)
0xE7
Open<15:12>, AGC4 RSSI Output<11:0> (read only)
Open<15:12>, AGC5 RSSI Output<11:0> (read only)
8-Bit Hex
Address
0x94
0x98
0x9C
0xA0
0xA4
0xA8
0xAC
0xB0
0xB4
0xB8
0xBC
0xC0
0xC4
0xC8
0xCC
0xD0
0xD4
0xD8
0xDC
0xE0
0xE4
GLOBAL REGISTER MAP
Chip I/O Access Control Register <7:0>
<7>: Synchronous Microport Bit. When this bit is set, the
microport assumes that its controls signals (such as R/W, DS,
and CS,) are synchronous to the CPUCLK. When cleared,
asynchronous control signals are assumed, and the microport
control signals are resynchronized with CPUCLK inside the
AD6636 part. Synchronous microport (when bit is set) has the
advantage of requiring a fewer number of clock cycles for
read/write access.
<0>: Byte Mode Bit. The byte mode bit selects the bit width for
the microport operation. Table 30 shows details.
Table 30. Microport Data Bus Width Selection
Chip Access Control
Register <0>
0 (default)
1
Microport Data Bus Bit Width
8-bit mode, using D<7:0>
16-bit mode, using D<15:0>
Channel I/O Access Control Register <5:0>
<6>: This bit is open.
<5:2>: Chip ID Bits. The chip ID bits are used to compare
against the chip ID input pins, enabling or disabling I/O access
for this specific chip. When more than one AD6636 part is
sharing the microport, different CHIPID pins can be used to
differentiate among the parts. A particular part gives I/O access
only when the CHIPID pins have the same value as these chip
ID bits.
These bits enable/disable the channel I/O access capability.
<5>: Channel 5 Access Bit. When the Channel 5 access bit is set
to Logic 1, any I/O write operation (from either the microport
or the serial port) that addresses a register located within the
channel register map updates the Channel 5 registers. Similarly,
for a read operation, the contents of the desired address in the
channel register map are output when this bit is set to Logic 1.
<1>: This bit is open.
Rev. 0 | Page 56 of 72
AD6636
<4>: Edge-Sensitivity Bit. When this bit is set, the rising edge on
the SYNC pin(s) is detected as a synchronization event (edgesensitive detection). When cleared, Logic 1 on the SYNC pin(s)
is detected as a synchronization event (level-sensitive
detection).
<4>: Channel 4 Access Bit. Similar to Bit <5> for Channel 4.
<3>: Channel 3 Access Bit. Similar to Bit <5> for Channel 3.
<2>: Channel 2 Access Bit. Similar to Bit <5> for Channel 2.
<1>: Channel 1 Access Bit. Similar to Bit <5> for Channel 1.
<0>: Channel 0 Access Bit. Similar to Bit <5> for Channel 0.
Note: If the access bits are set for more than one channel, during
write access all channels with access are written with same the
data. This is especially useful when more than one channel has
similar configurations. During a read operation, if more than
one channel has access, the read access is given to the channel
with the lowest channel number. For example, if both Channel 4
and Channel 2 have access bits set, then read access is given to
Channel 2.
Channel Enable Register <5:0>
<5>: Channel 5 Enable Bit. When this bit is set, Channel 5 logic
is enabled. When this bit is cleared, Channel 5 is disabled and
the channel’s logic does not consume any power. On power-up,
this bit comes up with Logic 0 and the channel is disabled. A
start sync does not start Channel 5 unless this bit is set before
issuing the start sync.
<4>: Channel 4 Enable Bit. Similar to Bit <5> for Channel 4.
<3>: Enable Synchronization from SYNC3 Bit. When this bit is
set, the SYNC3 pin can be used for synchronization. When this
bit is cleared, the SYNC3 pin is ignored. This is a global enable
for all SYNC pins, and each individual channel selects which
pin it listens to.
<2>: Enable Synchronization from SYNC2 Bit. Similar to
Bit <3> for the SYNC[2] pin.
<1>: Enable Synchronization from SYNC1 Bit. Similar to
Bit <3> for the SYNC1 pin.
<0>: Enable Synchronization from SYNC0 bit. Similar to
Bit <3> for the SYNC0 pin.
Soft Synchronization Configuration <7:0>
<7>: Soft Hop Synchronization Enable Bit. When this bit is set,
hop synchronization is enabled for all channels selected using
Bits 5:0. When this bit is cleared, hop synchronization is not
performed for any channels selected using Bits 5:0.
<6>: Soft Start Synchronization Enable Bit. When this bit is set,
start synchronization is enabled for all channels selected using
Bits 5:0. When this bit is cleared, start synchronization is not
performed for any channels selected using Bits 5:0.
<3>: Channel 3 Enable Bit. Similar to Bit <5> for Channel 3.
<2>: Channel 2 Enable Bit. Similar to Bit <5> for Channel 2.
Bits<5:0> form the SOFT_SYNC control bits. These bits can be
written to by the controller to initiate the synchronization of a
selected channel.
<1>: Channel 1 Enable Bit. Similar to Bit <5> for Channel 1.
<0>: Channel 0 Enable Bit. Similar to Bit <5> for Channel 0.
Pin Synchronization Configuration <7:0>
<7>: Hop Synchronization Enable Bit. This bit is a global enable
for any hop synchronization involving SYNC pins. When this
bit is set, hop synchronization is enabled for all channels that
are programmed for pin synchronization. When this bit is
cleared, hop synchronization is not performed for any channel
that is programmed for pin synchronization.
<6>: Start Synchronization Enable Bit. This bit is a global enable
for any start synchronization involving SYNC pins. When this
bit is set, start synchronization is enabled for all channels that
are programmed for pin synchronization. When this bit is
cleared, start synchronization is not performed for any channel
that is programmed for pin synchronization.
<5>: First Sync Only Bit. When this bit is set, the NCO
synchronization logic recognizes only the first synchronization
event as valid. All other requests for synchronization events are
ignored as long as this bit is set. When cleared, all synchronization events are acted upon.
<5>: Soft Sync Channel 5 Enable Bit. When this bit is set, it
enables Channel 5 to receive a hop sync or start sync, as defined
by Bits 7 and 6, respectively. When cleared, Channel 5 does not
receive any soft sync.
<4>: Soft Sync Channel 4 Enable Bit. Similar to Bit <5> for
Channel 4.
<3>: Soft Sync Channel 3 Enable Bit. Similar to Bit <5> for
Channel 3.
<2>: Soft Sync Channel 2 Enable Bit. Similar to Bit <5> for
Channel 2.
<1>: Soft Sync Channel 1 Enable Bit. Similar to Bit <5> for
Channel 1.
<0>: Soft Sync Channel 0 Enable Bit. Similar to Bit <5> for
Channel 0.
Rev. 0 | Page 57 of 72
AD6636
LVDS Control Register <10:0>
<10>: CMOS Mode Bit. When this bit is set, the ADC ports
operate in CMOS mode. When this bit is cleared, the ADC ports
operate in LVDS mode. The default is Logic 1 or CMOS mode.
In LVDS mode, two CMOS ADC port pins are used to form one
differential pair of LVDS ADC ports.
<9>: Reserved. This bit should always be written Logic 1.
<8>: Autocalibrate Enable Bit. When this bit is set, the autocalibration cycle is invoked for the LVDS pads. At the end of
calibration, this calibration value is set for the LVDS pads.
When this bit is cleared, the output for the LVDS controller is
taken from manual calibration value (Bits <7:0> of this
register).
<7:4>: These bits are open.
<3:0>: Manual Calibration Value Bits. The value of these bits is
used for manual LVDS calibration. When the autocalibrate bit is
set, these bits are don’t care.
Interrupt Status Register <15:0>
This register is read-only.
<15>: AGC 5 RSSI Update Interrupt Bit. If the AGC 5 update
interrupt enable bit is set, this bit is set by the AD6636 whenever
AGC 5 updates a new RSSI word (the new word should be
different from the previous word). If the AGC 5 update
interrupt enable bit is cleared, then this bit is not set (not
updated). An interrupt is not generated in this case.
AD6636 does not set this bit on signature generation and an
interrupt is not generated.
<8>: Channel 4 Data Ready Interrupt Bit. Similar to Bit <9> for
Channel 4.
<7>: Channel 3 Data Ready Interrupt Bit. Similar to Bit <9> for
Channel 3.
<6>: Channel 2 Data Ready Interrupt Bit. Similar to Bit <9> for
Channel 2.
<5>: Channel 1 Data Ready Interrupt Bit. Similar to Bit <9> for
Channel 1.
<4>: Channel 0 Data Ready Interrupt Bit. Similar to Bit <9> for
Channel 0.
<3>: ADC Port D Power Monitoring Interrupt Bit. This bit is set
by the AD6636 whenever the ADC Port D power monitor
interrupt enable bit is set and the Port D power monitor timer
runs out (end of the Port D power monitor period). If the ADC
Port D power monitoring interrupt enable bit is cleared, the
AD6636 does not set this bit and does not generate an interrupt.
Note: In real input CMOS mode, all four input ports exist. In
complex input CMOS mode, only ADC Ports A and C function.
In real input LVDS mode, only ADC Ports A and C function.
<2>: ADC Port C Power Monitoring Interrupt Bit. Similar to
Bit <3> for ADC Port C.
<1>: ADC Port B Power Monitoring Interrupt Bit. Similar to
Bit <3> for ADC Port B.
Note: For Bits <15:10>, no interrupt is generated, if the new
RSSI word is the same as the previous RSSI word.
<14>: AGC 4 RSSI Update Interrupt Bit. Similar to Bit <15> for
the AGC 4.
<13>: AGC 3 RSSI Update Interrupt Bit. Similar to Bit <15> for
the AGC 3.
<12>: AGC 2 RSSI Update Interrupt Bit. Similar to Bit <15> for
the AGC 2.
<11>: AGC 1 RSSI Update Interrupt Bit. Similar to Bit <15> for
the AGC 1.
<10>: AGC 0 RSSI Update Interrupt Bit. Similar to Bit <15> for
the AGC 0.
<9>: Channel 5 Data Ready Interrupt Bit. This bit is set to
Logic 1 whenever the channel BIST signature registers are
loaded with data. The conditions required for setting this bit
are: the channel BIST signature registers is programmed for
BIST signature generation and the Channel 5 data ready enable
bit in the interrupt enable register is cleared. If the Channel 5
data ready enable bit in the interrupt enable register is set, the
<0>: ADC Port A Power Monitoring Interrupt Bit. Similar to
Bit <3> for ADC Port A.
Interrupt Enable Register <15:0>
<15>: AGC 5 RSSI Update Enable Bit. When this bit is set, the
AGC 5 RSSI update interrupt is enabled, allowing an interrupt
to be generated when the RSSI word is updated. When this bit is
cleared, an interrupt cannot be generated for this event. Also,
see the Interrupt Status Register <15:0> section.
<14>: AGC 4 RSSI Update Enable Bit. Similar to Bit <15> for
the AGC 4.
<13>: AGC 3 RSSI Update Enable Bit. Similar to Bit <15> for
the AGC 3.
<12>: AGC 2 RSSI Update Enable Bit. Similar to Bit <15> for
the AGC 2.
<11>: AGC 1 RSSI Update Enable Bit. Similar to Bit <15> for
the AGC 1.
<10>: AGC 0 RSSI Update Enable Bit. Similar to Bit <15> for
the AGC 0.
Rev. 0 | Page 58 of 72
AD6636
<9>: Channel 5 Data Ready Enable Bit. When this bit is set, the
Channel 5 data ready interrupt is enabled, allowing an interrupt
to be generated when Channel 5 BIST signature registers are
updated. When this bit is cleared, an interrupt cannot be
generated for this event.
<8>: Channel 4 Data Ready Enable Bit. Similar to Bit <9> for
Channel 4.
<7>: Channel 3 Data Ready Enable Bit. Similar to Bit <9> for
Channel 3.
<6>: Channel 2 Data Ready Enable Bit. Similar to Bit <9> for
Channel 2.
<5>: Channel 1 Data Ready Enable Bit. Similar to Bit <9> for
Channel 1.
<4>: Channel 0 Data Ready Enable Bit. Similar to Bit <9> for
Channel 0.
<3>: ADC Port D Power Monitoring Enable Bit. When this bit is
set to Logic 1, the ADC Port D power monitoring interrupt is
enabled allowing an interrupt to be generated when ADC Port
D power monitoring registers are updated. When set to Logic 1,
the ADC Port D power monitoring interrupt is disabled.
<2>: ADC Port C Power Monitoring Enable Bit. Similar to
Bit <3> for ADC Port C.
<1>: ADC Port B Power Monitoring Enable Bit. Similar to
Bit <3> for ADC Port B.
<0>: ADC Port A Power Monitoring Enable Bit. Similar to
Bit <3> for ADC Port A.
INPUT PORT REGISTER MAP
ADC Input Control Register <27:0>
Note that complex input mode is available only in CMOS input
mode.
<24>: Port A Complex Data Active Bit. When this bit is set, the
data input on Ports A and B is interpreted as complex input
(Port A for the in-phase signal and Port B for the quadrature
phase signal). This complex input is passed on as input from
ADC Port A. When this bit is cleared, the data on ADC Port A
and ADC Port B is interpreted as real and independent input.
Note that complex input mode is available only in CMOS input
mode.
<23>: Channel 5 Complex Data Input Bit. When this bit is set,
Channel 5 gets complex input data from the source that is
selected by the crossbar mux select bits. When this bit is cleared,
Channel 5 receives real input data. (See Table 31.)
<22:20>: Channel 5 Crossbar Mux Select Bits. These bits select
the source of input data for Channel 5. (See Table 31.)
Table 31. Channel 5 Input Configuration
Complex Data
Input Bit
0
Crossbar Mux
Select Bits
000
0
001
0
010
0
011
0
100
1
000
1
001
1
010
These bits are general control bits for the ADC input logic.
<27>: PN Active Bit. When this bit is set, the pseudorandom
number generator is active. When this bit is cleared, the PN
generator is disabled and the seed is set to its default value.
Configuration
ADC Port A drives input
(real).
ADC Port B drives input
(real).
ADC Port C drives input
(real).
ADC Port D drives input
(real).
PN sequence drives input
(real).
Ports A and B drive
complex input.
Ports C and D drive
complex input.
PN sequence drives
complex input.
<19>: Channel 4 Complex Data Input Bit. Similar to Bit <23>
for Channel 4.
<26>: EXP Lock Bit. When this bit is set along with the PN
active bit, then the EXP signal for pseudorandom input is
locked to 000 (giving full-scale input). When this bit is cleared,
EXP bits for pseudorandom input are randomly generated input
data bits.
<25>: Port C Complex Data Active Bit. When this bit is set, the
data inputs on Ports C and D are interpreted as complex inputs
(Port C for the in-phase signal and Port D for the quadrature
phase signal). This complex input is passed on as the input from
ADC Port C. When this bit is cleared, the data on ADC Port C
and ADC Port D interpreted as real and independent input.
<18:16>: Channel 4 Crossbar Mux Select Bits. Similar to Bits
<22:20> for Channel 4.
<15>: Channel 3 Complex Data Input Bit. Similar to Bit <23>
for Channel 3.
<14:12>: Channel 3 Crossbar Mux Select Bits. Similar to Bits
<22:20> for Channel 3.
<11>: Channel 2 Complex Data Input Bit. Similar to Bit <23>
for Channel 2.
<10:8>: Channel 2 Crossbar Mux Select Bits. Similar to Bits
<22:20> for Channel 2.
Rev. 0 | Page 59 of 72
AD6636
<7>: Channel 1 Complex Data Input Bit. Similar to Bit <23> for
Channel 1.
decimation of 224. Each increment of these bits increases the
decimation value by a power of 2.
<6:4>: Channel 1 Crossbar Mux Select Bits. Similar to Bits
<22:20> for Channel 1.
<7:4>: DC Loop BW. These bits set the decimation and
interpolation value used in the low-pass filters for the dc offset
estimation feedback loop. A value of 0 sets a decimation/
interpolation of 212 and a value of 11 sets decimation/
interpolation of 224. Each increment of these bits increases the
decimation/interpolation value by a power of 2.
<3>: Channel 0 Complex Data Input Bit. Similar to Bit <23> for
Channel 0.
<2:0>: Channel 0 Crossbar Mux Select Bits. Similar to Bits
<22:20> for Channel 0.
<3>: Reserved.
ADC CLK Control Register <11:0>
These bits control the ADC clocks and internal PLL clock.
<11>: ADC Port D CLK Invert Bit. When this bit is set, the
inverted ADC Port D clock is used to register ADC input port
D data into the part. When this bit is cleared, the clock is used as
is, without any inversion or phase change.
<10>: ADC Port C CLK Invert Bit. Similar to Bit <11> for ADC
Port C.
<9>: ADC Port B CLK Invert Bit. Similar to Bit <11> for ADC
Port B.
<8>: ADC Port A CLK Invert Bit. Similar to Bit <11> for ADC
Port A.
<7:6>: ADC Pre PLL Clock Divider Bits. These bits control the
PLL clock divider. The PLL clock is derived from the ADC
Port A clock.
Table 32. PLL Clock Divider Select Bits
PLL Clock Divider Bits <12:11>
00
01
10
11
Divide-by Value
Divide-by-1, bypass
Divide-by-2
Divide-by-4
Divide-by-8
<5:1>: PLL Clock Multiplier Bits. These bits control the PLL
clock multiplier. The output of the PLL clock divider is
multiplied with the binary value of these bits. The valid range
for the multiplier is from 4 to 20. A value outside this range
powers down the PLL, and the PLL clock is the same as the
ADC Port A clock.
<0>: This bit is open (write Logic 0).
Port AB, I/Q Correction Control <15:0>
<15:12>: Amplitude Loop BW. These bits set the decimation
value used in the integrator for the amplitude offset-estimation
feedback loop. A value of 0 sets a decimation of 212 and a value
of 11 sets decimation of 224. Each increment of these bits
increases the decimation value by a power of 2.
<11:8>: Phase Loop BW. These bits set the decimation value
used in the integrator for the phase offset-estimation feedback
loop. A value of 0 sets a decimation of 212 and a value of 11 sets
<2>: Port AB Amplitude Correction Enable Bit. When the
amplitude correction enable bit is set, the amplitude correction
function of the I/Q correction logic for the AB port is enabled.
When this bit cleared, the amplitude correction value is given by
the value of the AB amplitude correction register. If the Port A
complex data active bit of the ADC input control register is
cleared (real input mode), this bit is a don’t care.
<1>: Port AB Phase Correction Enable Bit. When this bit is set,
the phase correction function of the I/Q correction logic for the
AB port is enabled. When this bit is cleared, the phase correction value is given by the value of the AB phase correction
register. If the Port A complex data active bit of the ADC input
control register is cleared (real input mode), this bit is a don’t
care.
<0>: Port AB DC Correction Enable Bit. When this bit is set, the
dc offset correction function of the I/Q correction block for the
AB port is enabled. When this bit is cleared, the dc offset
correction value is given by the value of the AB offset correction
registers. If the Port A complex data active bit of the ADC input
control register is cleared (real input mode), this bit is a don’t
care.
Port CD, I/Q Correction Control <15:0>
<15:12>: Amplitude Loop BW. These bits set the decimation
value used in the integrator for the amplitude offset estimation
feedback loop. A value of 0 sets a decimation of 212 and a value
of 11 sets decimation of 224. Each increment of these bits
increases the decimation value by a power of 2.
<11:8>: Phase Loop BW. These bits set the decimation value
used in the integrator for the phase offset estimation feedback
loop. A value of 0 sets a decimation of 212 and a value of 11 sets
decimation of 224. Each increment of these bits increases the
decimation value by a power of 2.
<7:4>: DC Loop BW. These bits set the decimation and
interpolation value used in the low pass filters for the dc offset
estimation feedback loop. A value of 0 sets a decimation/
interpolation of 212 and a value of 11 sets decimation/
interpolation of 224. Each increment of these bits increases the
decimation/interpolation value by a power of 2.
<3>: Reserved.
Rev. 0 | Page 60 of 72
AD6636
<2>: Port CD Amplitude Correction Enable Bit. When this bit is
set, the amplitude correction function of the I/Q correction
logic for the AB port is enabled. When this bit is cleared, the
amplitude correction value is given by the value of the AB
amplitude correction register. If the Port A complex data active
bit of the ADC input control register is cleared (real input
mode), this bit is a don’t care.
<1>: Port CD Phase Correction Enable Bit. When this bit is set,
the phase correction function of the I/Q correction logic for the
AB port is enabled. When this bit is cleared, the phase
correction value is given by the value of the AB phase
correction register. If the Port A complex data active bit of the
ADC input control register is cleared (real input mode), this bit
is a don’t care.
<0>: Port CD DC Correction Enable Bit. When the dc
correction enable bit is set, the dc offset correction function of
the I/Q correction block for the AB port is enabled. When
cleared, the dc offset correction value is given by the value of
the AB offset correction registers. If the Port A complex data
active bit of the ADC input control register is cleared (real input
mode), this bit is a don’t care.
Port AB, DC Offset Correction I <15:0>
This register holds the in-phase signal dc offset correction value
for complex data stream when dc correction is enabled. This
value should be set manually when automatic correction is
disabled. This 16-bit value is subtracted from the 16-bit ADC
Port A data (in-phase signal). This data is a don’t care in real
input mode.
Port AB, DC Offset Correction Q <15:0>
This register holds the quadrature phase signal dc offset
correction value for complex data stream when dc correction
enabled. This value should be set manually when automatic
correction is disabled. This 16-bit value is subtracted from the
16-bit ADC Port B data (quadrature phase signal). This data is a
don’t care in real input mode.
Port CD, DC Offset Correction I <15:0>
This register holds the in-phase signal dc offset correction value
for complex data stream when dc correction is enabled. This
value should be set manually when automatic correction is
disabled. This 16-bit value is subtracted from the 16-bit ADC
Port C data (in-phase signal). This data is a don’t care in real
input mode.
Port AB, Phase Offset Correction <15:0>
This register holds the phase offset correction value for complex
data stream when the AB port phase correction is enabled. This
value is set manually when automatic correction is disabled.
This value is calculated as tan(phase_mismatch), where
phase_mismatch is the mismatch in phase between I (in-phase
signal) and Q (quadrature phase signal). This 14-bit value is
multiplied with 16-bit Q (quadrature phase signal, Input Port B)
and added to 16-bit I (in-phase signal, Input Port A). This data
is a don’t care in real input mode.
Port AB, Amplitude Offset Correction <15:0>
This register holds the amplitude offset correction value for
complex data stream when the AB port amplitude correction is
enabled. This value is set manually when automatic correction
is disabled. This value is calculated as (Mag(Q) − Mag(I)),
where I is the in-phase signal and Q is the quadrature phase
signal. This 14-bit value is multiplied with 16-bit Q (quadrature
phase signal, Input Port B) and added to 16-bit Q (quadrature
phase signal, Input Port B). This data is a don’t care in real input
mode.
Port CD, Phase Offset Correction <15:0>
This register holds the phase offset correction value for the
complex data stream when CD port phase correction is enabled.
This value should be set manually when automatic correction is
disabled. This value should be calculated as tangent
(phase_mismatch), where phase_mismatch is the mismatch in
phase between I (in-phase signal) and Q (quadrature phase
signal). This 14-bit value is multiplied with 16-bit Q
(quadrature phase signal, Input Port D) and added to 16-bit I
(in-phase signal, Input Port C). This data is a don’t care in real
input mode.
Port CD, Amplitude Offset Correction <15:0>
This register holds the amplitude offset correction value for
complex data stream when CD port amplitude correction is
enabled. This value is set manually when automatic correction
is disabled. This value is calculated as (Mag(Q) − Mag(I)),
where I is the in-phase signal and Q is the quadrature phase
signal. This 14-bit value are multiplied with 16-bit Q
(quadrature phase signal, Input Port D) and added to 16-bit Q
(quadrature phase signal, Input Port D). This data is a don’t care
in real input mode.
Port A Gain Control <7:0>
<7>: This bit is open.
Port CD, DC Offset Correction Q <15:0>
This register holds the quadrature phase signal dc offset
correction value for complex data stream when dc correction is
enabled. This value should be set manually when automatic
correction is disabled. This 16-bit value is subtracted from the
16-bit ADC Port D data (quadrature phase signal). This data is a
don’t care in real input mode.
<6:1>: This 6-bit word specifies the relinearization pipe delay to
be used in the ADC input gain control block. The decimal
representation of these bits is the number of input clock cycle
pipeline delays between the external EXP data output and the
internal application of relinearization based on EXP.
Rev. 0 | Page 61 of 72
AD6636
<0>: Gain Control Enable Bit. This bit controls the configuration of the EXP<2:0> bits for Channel A. When the gain control
enable bit is Logic 1, the EXP<2:0> bits are configured as
outputs. When this bit is cleared, the EXP<2:0> bits are inputs.
Port A Dwell Timer <19:0>
This register is used to set the dwell time for the gain control
block. When gain control block is active and detects a decrease
in the signal level below the lower threshold value (programmable), a dwell time counter is initiated to provide temporal
hysteresis. Doing so prevents the gain from being switched
continuously. Note that the dwell timer is turned on only after a
drop below the lower threshold is detected in the signal level.
Port A Power Monitor Period <23:0>
This register is used in the power monitoring logic to set the
period of time for which ADC input data is monitored. This
value represents the monitor period in number of ADC port
clock cycles.
Table 33. Monitor Function Select Bits
This register is read-only and contains the current status of the
power monitoring logic output. The output is dependent on the
power monitoring mode selected. When the power monitor
block is enabled, this register is updated at the end of each
power monitor period. This register is updated even if an
interrupt signal is not generated.
Port A Upper Threshold <9:0>
This register serves the dual purpose of specifying the upper
threshold value in the gain control block and in the power
monitoring block, depending on which block is active. Any
ADC port input data having a magnitude greater than this value
triggers a gain change in the gain control block. Any ADC port
input data having a magnitude greater than this value is
monitored in the power monitoring block (in peak detect or
threshold crossing mode). The value of the register is compared
with the absolute magnitude of the input port data. For real
input, the absolute magnitude is the same as the input data; for
positive and negative data, the absolute magnitude is the value
of the data after removing the negative sign.
Port A Lower Threshold <9:0>
This register is used in the gain control block and represents the
magnitude of the lower threshold for ADC port input data. Any
ADC input data having a magnitude below the lower threshold
initiates the dwell time counter. The value of the register is
compared with the absolute magnitude of the input port data.
For real input, the absolute magnitude is the same as the input
data; for positive and negative data, the absolute magnitude is
the value of the data after removing the negative sign.
This register controls the functions of the power monitoring
block.
<3>: Clear-on-Read Bit. When this bit is set, the power monitor
holding register is cleared every time this register is read. This
bit controls whether the power monitoring function is cleared
after a read of the power monitor period register. If this bit is
set, the monitoring function is cleared after the read. If this bit is
Logic 0, the monitoring function is not cleared. This bit is a
don’t care if the disable integration counter bit is clear.
<2:1>: Monitor Function Select Bits. Table 33 lists the functions
of these bits.
Port A Power Monitor Output <23:0>
Port A Signal Monitor <4:0>
<4>: Disable Power Monitor Period Timer Bit. When this bit is
set, the power monitor period timer no longer controls the
update of the power monitor holding register. A user read to the
power monitor holding register updates this register. When this
bit is cleared, the power monitor period register controls the
timer and, therefore, controls the update rate of the power
monitor holding register.
Monitor Function Select
00
01
10
11
Function Enabled
Peak Detect Mode
Mean Power Monitor Mode
Threshold Crossing Mode
Invalid Selection
<0>: Monitor Enable Bit. When this bit is set, the power
monitoring function is enabled and operates as selected by
Bits <2:1> of the signal monitor register. When this bit is
cleared, the power monitoring function is disabled and the
signal monitor register <2:1> bits are don’t care. This bit defaults
to 0 on power-up.
Note: Gain control, dwell timer, power monitor period, signal
monitor, power monitoring output, lower threshold and upper
threshold registers for Ports B, C, and D work similarly to the
corresponding registers definitions for Port A.
CHANNEL REGISTER MAP
Channel control registers are common to all six channels, and
access to specific channels is determined by the channel I/O
access register (Address 0x02).
NCO Control <15:0>
These bits control the NCO operation.
<8:7>: NCO Sync Start Select Bits. These bits determine which
SYNC input pin is used by this channel for a start synchronization operation. Table 34 describes the selection.
Table 34. Sync Start Select Bits
NCO Control <8:7>
00
01
10
11
Rev. 0 | Page 62 of 72
SYNC Pin Used for Start Synchronization
SYNC0
SYNC1
SYNC2
SYNC3
AD6636
<6:5>: NCO Sync Hop Select Bits. These bits determine which
SYNC input pin is used by this channel for a hop synchronization operation. Table 35 describes the selection.
Table 35. Sync Hop Select Bits
NCO Control <6:5>
00
01
10
11
SYNC Pin Used for Hop Synchronization
SYNC0
SYNC1
SYNC2
SYNC3
<4>: This bit is open.
<3>: NCO Bypass Bit. When this bit is set, the NCO is bypassed
shuts down for power savings. This bit can be used for power
savings, when NCO frequency of dc or 0 Hz is required. When
this bit is cleared, the NCO operates as programmed.
<2>: Clear NCO Accumulator Bit. When this bit is set, the clear
NCO accumulator bit synchronously clears the phase accumulator on all frequency hops in this channel. When this bit is
cleared, the accumulator is not cleared and phase continuous
hops are implemented.
<1>: Phase Dither Enable Bit. When this bit is set, phase
dithering in the NCO is enabled. When this bit is cleared, phase
dithering is disabled.
<0>: Amplitude Dither Enable Bit. When this bit is set,
amplitude dithering in the NCO is enabled. When this bit is
cleared, amplitude dithering is disabled.
NCO Phase Offset <15:0>
The value in the register is loaded into the phase accumulator of
the NCO block every time a start sync or hop sync is received
by the channel. This allows individual channels to be started
with a known nonzero phase. The NCO phase offset is not
loaded on a hop sync, if Bit <2> of the NCO control register
(clear phase accumulator on hop) is cleared. This NCO offset
register value is interpreted as a 16-bit unsigned integer. A
0x0000 in this register corresponds to a 0 radian offset, and a
0xFFFF corresponds to an offset of 2π (1 − 1/(216)) radians.
CIC Bypass <0>
When this bit is set, the entire CIC filter is bypassed. The
output of CIC filter is driven straight from the input without
any change. When this bit is cleared, the CIC filter operates in
normal mode as programmed. Writing Logic 1 to this bit
disables both the CIC decimation operation and the CIC
scaling operation.
CIC Decimation <4:0>
This 5-bit word specifies the CIC filter decimation value minus
1. A value of 0x00 is a decimation of 1 (bypass), and 0x1F is a
decimation of 32. Writing a value of 0 in this register bypasses
CIC filtering, but does not bypass the CIC scaling operation.
CIC Scale Factor <4:0>
This 5-bit word specifies the CIC filter scale factor used to
compensate for the gain provided by the CIC filter. The
recommended value is given by the following equation:
Channel Start Hold-Off Counter <15:0>
When a start synchronization (software or hardware) occurs on
the channel, the value in this register is loaded into a downcounter. When the counter has finished counting down to 0, the
channel operation is started.
NCO Frequency Hop Hold-Off Counter <15:0>
When a hop sync occurs, a counter is loaded with the NCO
frequency hold-off register value. The 16-bit counter starts
counting down. When it reaches 0, the new frequency value in
the shadow register is written to the NCO frequency register.
(See the Numerically Controlled Oscillator (NCO) section.)
CIC Scale Register = ceil(5 × log2 (MCIC)) − 5
where:
MCIC is the decimation rate of the CIC (one more than the value
in the CIC decimation register).
ceil operation gives the closest integer greater than or equal to
the argument.
The valid range for this register is decimal 0 to 20.
FIR-HB Control <3:0>
NCO Frequency <31:0>
The value in this register is used to program the NCO tuning
frequency. The value to be programmed is given by the
following equation:
NCO Frequency Register =
The value given by the equation should be loaded into the
register in binary format.
NCO _ FREQUENCY
× 232
CLK
where:
NCO_FREQUENCY is the desired NCO tuning frequency.
<3>: FIR1 Enable Bit. When this bit is set, the FIR1 fixedcoefficient filter is enabled. When cleared, FIR1 is bypassed.
<2>: HB1 Enable Bit. When this bit is set, the HB1 half-band
filter is enabled. When cleared, HB1 is bypassed.
<1>: FIR2 Enable Bit. When this bit is set, the FIR2 fixedcoefficient filter is enabled. When cleared, FIR2 is bypassed.
<0>: HB2 Enable bit. When this bit is set, the HB2 half-band
filter is enabled. When cleared, HB2 is bypassed.
CLK is the ADC clock rate.
Rev. 0 | Page 63 of 72
AD6636
MRCF Control Register <12:0>
MRCF Coefficient Memory
<12:10>: MRCF Data Select Bits. These bits are used to select
the input source for the MRCF filter. Each MRCF filter can be
driven by output from the HB2 filter of any channel independently. Table 36 shows the selections available.
The MRCF coefficient memory consists of eight coefficients,
each six bits wide. The memory extends from Address 0x80 to
Address 0x87. The coefficients should be written in twos
complement format.
DRCF Control Register <11:0>
Table 36. MRCF Data Select Bits
MRCF Data Select<2:0>
000
001
010
011
1x0
1x1
MRCF Input Source
MRCF input taken from Channel 0
MRCF input taken from Channel 1
MRCF input taken from Channel 2
MRCF input taken from Channel 3
MRCF input taken from Channel 4
MRCF input taken from Channel 5
<9>: Interpolating Half-Band Enable Bit. When this bit is set,
the interpolating half-band filter, driven by the output of the
CRCF block, is enabled. When cleared, the interpolating halfband filter is bypassed and its output is the same as its input.
The interpolating half-band filter doubles the data rate.
<8>: This bit is open.
<7>: Half-Rate Bit. When this bit is set, the MRCF filter operates
using half the PLL clock rate. This is used for power savings
when there is sufficient time to complete MRCF filtering using
only half the PLL clock rate. When this bit is cleared, the MRCF
filter operates at the full PLL clock rate. (See the Mono-Rate
RAM Coefficient Filter section.)
<6:4>: MRCF Number of Taps Bits. This 3-bit word should be
written with one less than the number of taps that are calculated
by the MRCF filter. The filter length is given by the decimal
value of this register plus 1. A value of 0 represents a 1-tap filter
and maximum value of 7 represents an 8-tap filter.
<3:2>: MRCF Scale Factor Bits. The output of the MRCF filter is
scaled according to the value of these bits. Table 37 describes
the attenuation corresponding to each setting.
Table 37. MRCF Scale Factor
MRCF Scale<1:0>
00
01
10
11
Scale Factor
18.06 dB attenuation (left-shift 3 bits)
12.04 dB attenuation (left-shift 2 bits)
6.02 dB attenuation (left-shift 1 bit)
No Scaling (0 dB)
<11>: DRCF Bypass Bit. When this bit is set, the DRCF filter is
bypassed and, therefore, its output is the same as its input. When
this bit is cleared, the DRCF has normal operation as programmed by the rest of this control register.
<10>: Symmetry Bit. When this bit is set, it indicates that the
DRCF is implementing a symmetrical filter and only half the
impulse response needs to be written into the DRCF coefficient
RAM. When this bit is cleared, the filter is asymmetrical and
complete impulse response of the filter should be written to the
coefficient RAM. When this filter is symmetrical, it can
implement up to 128 filter taps.
<9:8>: DRCF Multiply Accumulate Scale Bits.The output of the
DRCF filter is scaled according to the value of these bits.
Table 38 lists the attenuation corresponding to each setting.
Table 38. DRCF Multiply Accumulate Scale Bits
DRCF Scale<1:0>
00
01
10
11
Scale Factor
18.06 dB attenuation (left-shift 3 bits)
12.04 dB attenuation (left-shift 2 bits)
6.02 dB attenuation (left-shift 1 bit)
No Scaling (0 dB)
<7:4>: DRCF Decimation Rate. This 4-bit word should be
written with one less than the decimation rate of the DRCF
filter. A value of 0 represents a decimation rate of 1 (no rate
change), and the maximum value of 15 represents a decimation
of 16. Filtering can be implemented irrespective of the
decimation rate.
<3:0>: DRCF Decimation Phase Bits. This 4-bit word represents
the decimation phase used by the DRCF filter. The valid range is
0 up to MDRCF − 1, where MDRCF is the decimation rate of the
DRCF filter. This word is primarily used for synchronization of
multiple channels of the AD6636, when more than one channel
is used for filtering one signal (one carrier).
DRCF Coefficient Offset <7:0>
<1>: This bit is open.
<0>: MRCF Bypass Bit. When this bit is set, the MRCF filter is
bypassed and, therefore, the output of the MRCF is the same as
its input. When this bit is cleared, the MRCF has normal
operation as programmed by its control register.
This register is used to specify which section of the 64-word
coefficient memory is used for a filter. It can be used to select
between multiple filters that are loaded into memory and
referenced by this pointer. This register is shadowed, and the
filter pointer is updated every time a new filter is started. This
allows the coefficient offset to be written even while a filter is
being computed without disturbing operation. The next sample
comes out of the DRCF with the new filter.
Rev. 0 | Page 64 of 72
AD6636
DRCF Taps <6:0>
This register is written with one less than the number of taps
that are calculated by the DRCF filter. The filter length is given
by the decimal value of this register plus 1. A value of 0
represents a 1-tap filter, and a value of 0x28 (40 decimal)
represents a 41-tap filter.
<3:0>: CRCF Decimation Phase. This 4-bit word represents the
decimation phase used by the CRCF filter. The valid range is 0
to MCRCF − 1, where MCRCF is the decimation rate of the CRCF
filter. This word is primarily used for synchronization of
multiple channels of the AD6636, when more than one channel
is used for filtering one signal (one carrier).
DRCF Start Address <5:0>
CRCF Coefficient Offset <5:0>
This register is written with the starting address of the DRCF
coefficient memory to be updated.
This register is used to specify which section of the 64-word
coefficient memory is used for a filter. It can be used to select
between multiple filters that are loaded into memory and
referenced by this pointer. This register is shadowed, and the
filter pointer is updated every time a new filter is started. This
allows the coefficient offset to be written even while a filter is
being computed without disturbing operation. The next sample
comes out of the CRCF with the new filter.
DRCF Final Address <5:0>
This register is written with the ending address of the DRCF
coefficient memory to be updated.
DRCF Coefficient Memory <13:0>
DRCF Memory. This memory consists of 64 words, and each
word is 14 bits wide. The data written to this memory space is
expected to be 14-bit, twos complement format. See the
Decimating RAM Coefficient Filter section for the method to
program the coefficients into the coefficient memory.
CRCF Control Register <11:0>
<11>: CRCF Bypass Bit. When this bit is set, the DRCF filter is
bypassed and, therefore, its output is the same as its input. When
this bit is cleared, the CRCF has normal operation as programmed by its control register.
<10>: Symmetry Bit. When this bit is set, it indicates that the
CRCF is implementing a symmetrical filter and only half the
impulse response needs to be written into the CRCF coefficient
RAM. When this bit is cleared, the filter is asymmetrical and the
complete impulse response of the filter should be written into
the coefficient RAM. When this filter is symmetrical, it can
implement up to 128 filter taps.
<9:8>: CRCF Multiply Accumulate Scale Bits. The output of the
CRCF filter is scaled according to the value of these bits.
Table 39 lists the attenuation corresponding to each setting.
Table 39. CRCF Multiply Accumulate Scale Bits
CRCF Scale<1:0>
00
01
10
11
Scale Factor
18.06 dB attenuation (left-shift 3 bits)
12.04 dB attenuation (left-shift 2 bits)
6.02 dB attenuation (left-shift 1 bit)
No Scaling (0 dB)
CRCF Taps <6:0>
This register is written with one less than the number of taps
that are calculated by the CRCF filter. The filter length is given
by the decimal value of this register plus 1. A value of 0
represents a 1-tap filter, and a value of 0x28 (40 decimal)
represents a 41-tap filter.
CRCF Coefficient Memory
CRCF Memory. This memory has 64 words that have 20 bits
each. The memory contains the CRCF filter coefficients. The
data written to this memory space is 20-bit in twos complement
format. See the Channel RAM Coefficient Filter section for the
method to program the coefficients into the coefficient
memory.
AGC Control Register <10:0>
<10>: Channel Sync Select Bit. When this bit is set, the AGC
uses the sync signal from the channel for its synchronization.
When this bit is cleared, the SYNC pin used for synchronization
is defined by Bits <9:8> of this register.
<9:8>: SYNC Pin Select Bits. When Bit <10> of this register is
cleared, these bits specify the SYNC pin used by AGC for
synchronization. These bits are don’t care when Bit <10> of the
AGC control register is set to Logic 1.
Table 40. SYNC Pin Select Bits
<7:4>: CRCF Decimation Rate. This 4-bit word should be
written with one less than the decimation rate of the CRCF
filter. A value of 0 represents a decimation rate of 1 (no rate
change) and the maximum value of 15 represents a decimation
of 16. Filtering operation is done irrespective of the
decimation rate.
AGC Control Bits <9:8>
00
01
10
11
Rev. 0 | Page 65 of 72
SYNC Pin Used by AGC
SYNC0
SYNC1
SYNC2
SYNC3
AD6636
<7:5>: AGC Word Length Control Bits. These bits define the
word length of the AGC output. The output word can be 4 to 8,
10, 12, or 16 bits wide. Table 41 shows the possible selections.
Table 41. AGC Word Length Control Bits
AGC Control Bits <7:5>
000
001
010
011
100
101
110
111
Output Word Length (Bits)
16
12
10
8
7
6
5
4
<4>: AGC Mode Bit. When this bit is set, the AGC operates to
maintain a desired signal level. When this bit is cleared, it
operates to maintain a constant clipping level. See the
Automatic Gain Control section for details about these modes.
<3>: AGC Sync Now Bit. This bit is used to synchronize a
particular AGC irrespective of the channel through the
programming ports (microport or serial port). When this bit is
set, the AGC block updates a new output sample (RSSI sample)
and starts working toward a new update sample.
<2>: Initialize on Sync Bit. This bit is used to determine whether
or not the AGC should initialize on a sync. When this bit is set,
during a synchronization the CIC filter is cleared and new
values for CIC decimation, number of averaging samples, CIC
scale, signal gain Gs’ gain K, and pole parameter P are loaded.
When Bit <2> = 0, the above-mentioned parameters are not
updated, and the CIC filter is not cleared. In both cases an AGC
update sample is output from the CIC filter and the decimator
starts operating towards the next output sample whenever a
sync occurs.
initialize the AGC, as defined by the control word. The AGC
loop is updated with a new sample from the CIC filter whenever
a sync occurs. If this register is Logic 1, the AGC is updated
immediately when the sync occurs. If this register Logic 0, the
AGC cannot be synchronized.
AGC Update Decimation <11:0>
This 12-bit register sets the AGC decimation ratio from 1 to
4096. An appropriate scaling factor should be set to avoid loss of
bits. The decimation ratio is given by the decimal value of the
AGC update decimation<11:0> register contents plus 1, that is,
12’0x000 describes a decimation ratio of 1, and 12’0xFFF
describes a decimation ratio of 4096.
AGC Signal Gain <11:0>
This register is used to set the initial value for a signal gain used
in the gain multiplier. This 12-bit value sets the initial signal
gain in the range of 0 dB and 96.296 dB in steps of 0.024 dB.
Initial signal gain (SG) in dB should be converted to a register
setting using the following formula:
⎡
⎤
SG
×256⎥
Register Value = round ⎢
⎣ 20 log 10 (2)
⎦
AGC Error Threshold <11:0>
This 12-bit register is the comparison value used to determine
which loop gain value (K1 or K2) to use for optimum operation.
When the magnitude-of-error signal is less than the AGC error
threshold value, then K1 is used; otherwise, K2 is used. The word
format of the AGC error threshold register is four bits to the left
of the binary point and eight bits to the right. See the Automatic
Gain Control section for details.
⎡ Error Threshold
⎤
× 256 ⎥
Register Value = round ⎢
20
log
(
2
)
10
⎣
⎦
AGC Average Samples <5:0>
<1>: First Sync Only. This bit is used to ignore repetitive
synchronization signals. In some applications, the synchronization signal occurs periodically. If this bit is cleared, each
synchronization request resynchronizes the AGC. If this bit is
set, only the first occurrence causes the AGC to synchronize
and updates the AGC gain values periodically, depending on the
decimation factor of the AGC CIC filter.
This 6-bit register contains the scale used for the CIC filter and
the number of power samples to be averaged before being sent
to the CIC filter.
<0>: AGC Bypass Bit. When this bit is set, the AGC section is
bypassed. The N-bit representation from the interpolating halfband filters is still reduced to a lower bit width representation as
set by Bits <7:5> of the AGC control register. A truncation at the
output of the AGC accomplishes this task.
<1:0>: Number of AGC Average Samples. This defines the
number of samples to be averaged before they are sent to the
CIC decimating filter. See Table 42.
AGC Hold-Off Register <15:0>
The AGC hold-off counter is loaded with the value written to
this address when either a soft sync or pin sync comes into the
channel. The counter begins counting down. When it reaches 1,
a sync is sent to the AGC. This sync might or might not
<5:2>: CIC Scale. This 4-bit word defines the scale used for the
CIC filter. Each increment of this word increases the CIC scale
by 6.02 dB.
Table 42. Number of AGC Average Samples
AGC Average Samples <1:0>
00
01
10
11
Rev. 0 | Page 66 of 72
Number of Samples Taken
1
2
3
4
AD6636
AGC Pole Location <7:0>
This 8-bit register is used to define the open-loop filter pole
location P. Its value can be set from 0 to 0.996 in steps of 0.0039.
This value of P is updated in the AGC loop each time the AGC
is initialized. This open-loop pole location directly impacts the
closed-loop pole locations, as explained in the Automatic Gain
Control section.
AGC Desired Level <7:0>
This register contains the desired signal level or desired clipping
level, depending on operational mode. This desired request level
(R) can be set in dB from 0 to 23.99 in steps of 0.094 dB. The
request level (R) in dB should be converted to a register setting
using the following formula:
⎡
⎤
R
× 64 ⎥
Register Value = round ⎢
20
log
(
2
)
10
⎣
⎦
<14:0>: BIST Timer Bits. The <14:0> bits of this register form a
15-bit word that is loaded into the BIST timer. After loading the
BIST timer, the signature register is enabled for operation while
the timer is actively counting down. (See the User-Configurable
Built-In Self-Test (BIST) section.)
OUTPUT PORT REGISTER MAP
This part of the memory map deals with the output data and
controls for parallel output ports.
Parallel Port Output Control <31:0>
<23>: Port C Append RSSI Bit. When this bit is set, an RSSI
word is appended to every I/Q output sample, irrespective of
whether the RSSI word is updated in the AGC. When this bit is
cleared, an RSSI word is appended to an I/Q output sample only
when the RSSI word is updated. The RSSI word is not output for
subsequent I/Q samples until the next time the RSSI is updated
in the AGC.
AGC Loop Gain2 <7:0>
This 8-bit register is used to define the second possible openloop gain, K2. Its value can be set from 0 to 0.996 in steps of
0.0039. This value of K2 is updated each time the AGC is
initialized. When the magnitude-of-error signal in the loop is
greater than the AGC error threshold, then K2 is used by the
loop. K2 is updated only when the AGC is initialized.
AGC Loop Gain1 <7:0>
This 8-bit register is used to define the open-loop gain K1. Its
value can be set from 0 to 0.996 in steps of 0.0039. This value of
K is updated in the AGC loop each time the AGC is initialized.
When the magnitude-of-error signal in the loop is less than the
AGC error threshold, then K1 is used by the loop. K1 is updated
only when the AGC is initialized.
I Path Signature Register <15:0>
<22>: Port C, Data Format Bit. When this bit is set, the port is
configured for 8-bit parallel I/Q mode. When cleared, the port is
configured for 16-bit interleaved I/Q mode. See the Parallel Port
Output section for details.
<21>: Port C, AGC 5 Enable Bit. When this bit is set, AGC 5 data
(I/Q data) is output on parallel Output Port C (data bus). When
this bit is cleared, AGC 5 data does not appear on Output
Port C.
<20>: Port C, AGC 4 Enable Bit. Similar to Bit <21> for AGC 4.
<19>: Port C, AGC 3 Enable Bit. Similar to Bit <21> for AGC 3.
<18>: Port C, AGC 2 Enable Bit. Similar to Bit <21> for AGC 2.
<17>: Port C, AGC 1 Enable Bit. Similar to Bit <21> for AGC 1.
This 16-bit signature register is for the I path of the channel
logic. The signature register records data on the networks that
leave the channel logic, just before entering the second data
router.
Q Path Signature Register <15:0>
This 16-bit signature register is for the Q path of the channel
logic. The signature register records data on the networks that
leave the channel logic, just before entering the second data
router.
BIST Control <23:0>
<15>: Disable Signature Generation Bit. When this bit is active
high, the signature registers do not produce a pseudorandom
output value, but instead directly load the 24-bit input data.
When this bit is cleared, the signature register produces a
pseudorandom output for every clock cycle that it is active. See
the User-Configurable Built-In Self-Test (BIST) section for
details.
<16>: Port C, AGC 0 Enable Bit. Similar to Bit <21> for AGC 0.
<15>: Port B Append RSSI Bit. When this bit is set, an RSSI
word is appended to every I/Q output sample, irrespective of
whether or not the RSSI word is updated in the AGC. When this
bit is cleared, an RSSI word is appended to an I/Q output sample
only when the RSSI word is updated. The RSSI word is not
output for subsequent I/Q samples until the next time the RSSI
is updated in the AGC.
<14>: Port B, Data Format Bit. When this bit is set, the port is
configured for 8-bit parallel I/Q mode. When this bit is cleared,
the port is configured for 16-bit interleaved I/Q mode. See the
Parallel Port Output section.
<13>: Port B, AGC 5 Enable Bit. When this bit is set, AGC 5 data
(I/Q data) is output on parallel output Port A (data bus). When
this bit is cleared, AGC 5 data does not appear on output Port C.
Rev. 0 | Page 67 of 72
AD6636
<12>: Port B, AGC 4 Enable Bit. Similar to Bit <13> for AGC 4.
<6:4>: Complex Control Bits. These bits are described in
Table 44.
<11>: Port B, AGC 3 Enable Bit. Similar to Bit <13> for AGC 3.
Table 44. Complex Control Bits
<10>: Port B, AGC 2 Enable Bit. Similar to Bit <13> for AGC 2.
<9>: Port B, AGC 1 Enable Bit. Similar to Bit <13> for r AGC 1.
Complex Control <6:4>
000
No complex filters
<8>: Port B, AGC 0 Enable Bit. Similar to Bit <13> for AGC 0.
001
Str0/1 combined
<7>: Port A Append RSSI Bit. When this bit is set, an RSSI word
is appended to every I/Q output sample, irrespective of whether
or not the RSSI word is updated in the AGC. When this bit is
cleared, an RSSI word is appended to an I/Q output sample only
when the RSSI word is updated. The RSSI word is not output for
subsequent I/Q samples until the next time RSSI is updated
again in the AGC.
010
Str0/1 combined,
Str2/3 combined
011
Str0/1 combined,
Str2/3 combined,
Str 4/5 combined
101
Str0/1 combined
110
Str0/1 combined,
Str2/3 combined
111
Str0/1 combined,
Str2/3 combined,
Str 4/5 combined
<6>: Port A, Data Format Bit. When this bit is set, the port is
configured for 8-bit parallel I/Q mode. When this bit is cleared,
the port is configured for 16-bit interleaved I/Q mode. See the
Parallel Port Output section.
<5>: Port A, AGC 5 Enable Bit. When this bit is set, AGC 5 data
(I/Q data) is output on parallel output Port A (data bus). When
this bit is cleared, AGC 5 data does not appear on output Port C.
Comment
Stream control register controls
AGC usage.
Ch 0 and Ch 1 form a complex
filter.
Ch 0 and Ch 1 form a complex
filter; Ch 2 and Ch 3 form a
complex filter.
Ch 0 and Ch 1 form a complex
filter; Ch 2 and Ch 3 form a
complex filter; Ch 4 and Ch 5
form a complex filter.
Ch 0 and Ch 1 form a biphase
filter.
Ch 0 and Ch 1 form a biphase
filter; Ch 2 and Ch 3 to form a
biphase filter.
Ch 0 and Ch 1 to form a biphase
filter; Ch 2 and Ch 3 to form a
biphase filter; Ch 4 and Ch 5 to
form a biphase filter.
<4>: Port A, AGC 4 Enable Bit. Similar to Bit <5> for AGC 4.
<3:0>: Stream Control Bits. These bits are described in Table 45.
<3>: Port A, AGC 3 Enable Bit. Similar to Bit <5> for AGC 3.
Table 45. Stream Control Bits
<2>: Port A, AGC 2 Enable Bit. Similar to Bit <5> for AGC 2.
Stream
Control Bits
0000
<1>: Port A, AGC 1 Enable Bit. Similar to Bit <5> for AGC 1.
<0>: Port A, AGC 0 Enable Bit. Similar to Bit <5> for AGC 0.
0001
Output Port Control <9:0>
0010
<9:8>: PCLK Divisor Bits. When a parallel port is in master
mode, the PCLK is derived from the PLL_CLK. These bits
define the value of the divisor used to divide the PLL_CLK to
obtain the PCLK. These bits are don’t care in slave mode.
Table 43. PCLK Divisor Bits
PCLK Divisor <7:6>
00
01
10
11
0011
0100
0101
0110
Divisor Value
1
2
4
8
0111
1000
<7>: PCLK Master Mode Bit. When the PCLK master mode bit
is set, the PCLK pin is configured as an output and the PCLK is
driven by the PLL_CLK. Data is transferred out of the AD6636
synchronous to this output clock. When this bit is cleared, the
PCLK pin is configured as an input. The user is required to
provide a PCLK, and data is transferred out of the AD6636
synchronous to this input clock. On power-up, this bit is cleared
to avoid contention on the PCLK pin.
1001
Default
Rev. 0 | Page 68 of 72
Output Streams (str0, str1,
str2, str3, str4, str5)
Ch 0/1 combined; Ch 2, Ch 3,
Ch 4, Ch 5 independent
Ch 0/1/2 combined; Ch 3, Ch 4,
Ch 5 independent
Ch 0/1/2/3 combined; Ch 4, Ch
5 independent
Ch 0/1/2/3/4 combined; Ch 5
independent
Ch 0/1/2/3/4/5 combined
Ch 0/1/2 combined, Ch 3/4/5
combined
Ch 0/1 combined, Ch 2/3
combined, Ch 4/5 combined
Ch 0/1 combined, Ch 2/3
combined, Ch 4, Ch 5
independent
Ch 0/1/2 combined, Ch 3/4
combined, 5 independent
Ch 0/1/2/3 combined, Ch 4/5
combined.
Independent channels
Number of
Streams
5
4
3
2
1
2
3
3
3
2
6
AD6636
AGC 0, I Output <15:0>
AGC 4, Q Output <15:0>
This read-only register provides the latest in-phase output
sample from AGC 0. Note that AGC 0 might be bypassed, and
that AGC 0 here is representative of the datapath only.
This read-only register provides the latest quadrature-phase
output sample from AGC 4. Note that AGC 4 might be bypassed
and that AGC 4 here is representative of the datapath only.
AGC 0, Q Output <15:0>
AGC 5, I Output <15:0>
This read-only register provides the latest quadrature-phase
output sample from AGC 0. Note that AGC 0 might be
bypassed, and that AGC 0 here is representative of the
datapath only.
This read-only register provides the latest in-phase output
sample from AGC 5. Note that AGC 5 might be bypassed and
that AGC 5 here is representative of the datapath only.
AGC 1, I Output <15:0>
This read-only register provides the latest quadrature-phase
output sample from AGC 5. Note that AGC 5 might be bypassed
and that AGC 5 here is representative of the datapath only.
AGC 5, Q Output <15:0>
This read-only register provides the latest in-phase output
sample from AGC 1. Note that AGC 1 might be bypassed and
that AGC 1 here is representative of the datapath only.
AGC 1, Q Output <15:0>
This read-only register provides the latest quadrature-phase
output sample from AGC 1. Note that AGC 1 might be bypassed
and that AGC 1 here is representative of the datapath only.
AGC 2, I Output <15:0:
This read-only register provides the latest in-phase output
sample from AGC 2. Note that AGC 2 might be bypassed and
that AGC 2 here is representative of the datapath only.
AGC 2, Q Output <15:0>
This read-only register provides the latest quadrature-phase
output sample from AGC 2. Note that AGC 2 might be bypassed
and that AGC 2 here is representative of the datapath only.
AGC 3, I Output <15:0>
This read-only register provides the latest in-phase output
sample from AGC 3. Note that AGC 3 might be bypassed and
that AGC 3 here is representative of the datapath only.
AGC 3, Q output <15:0>
This read-only register provides the latest quadrature-phase
output sample from AGC 3. Note that AGC 3 might be bypassed
and that AGC 3 here is representative of the datapath only.
AGC 4, I Output <15:0>
This read-only register provides the latest in-phase output
sample from AGC 4. Note that AGC 4 might be bypassed and
that AGC 4 here is representative of the datapath only.
AGC 0, RSSI Output <11:0>
This read-only register provides the latest RSSI output sample
from AGC 0. This register is updated only when AGC 0 is
enabled and operating.
AGC 1, RSSI Output <11:0>
This read-only register provides the latest RSSI output sample
from AGC 1. This register is updated only when AGC 1 is
enabled and operating.
AGC 2, RSSI Output <11:0>
This read-only register provides the latest RSSI output sample
from AGC 2. This register is updated only when AGC 2 is
enabled and operating.
AGC 3, RSSI Output <11:0>
This read-only register provides the latest RSSI output sample
from AGC 3. This register is updated only when AGC 3 is
enabled and operating.
AGC 4, RSSI Output <11:0>
This read-only register provides the latest RSSI output sample
from AGC 4. This register is updated only when AGC 4 is
enabled and operating.
AGC 5, RSSI Output <11:0>
This read-only register provides the latest RSSI output sample
from AGC 5. This register is updated only when AGC 5 is
enabled and operating.
Rev. 0 | Page 69 of 72
AD6636
DESIGN NOTES
VDDCORE (1.8V)
The following guidelines describe circuit connections, layout
requirements, and programming procedures for the AD6636.
The designer should review these guidelines before starting the
system design and layout.
•
Input clocks (CLKA, CLKB, CLKC, CLKD) and input port
pins (INA[15:0] to IND[15:0], EXPA[2:0] to EXPD[2:0]) are
not 5 V tolerant. Care should be taken to drive these pins
within the limits of VDDIO (3.0 V to 3.6 V).
•
When the ADC output has less than 16 bits of resolution,
it should be connected to the MSBs of the input port (MSBjustified). The remaining LSBs should be connected to
ground.
•
The number format used in this part is twos complement.
All input ports and output ports use twos complement data
format. The formats for individual internal registers are
given in the memory map description of these registers.
•
In both microport and serial port operation, the DTACK
(RDY, SDO) pin is an open-drain output and, therefore,
should be pulled high externally using a pull-up resister.
The recommended value for the pull-up resistor is from
1 kΩ and 5 kΩ.
EXT_FILTER
04998-0-051
The AD6636 requires the following power-up sequence: The
VDDCORE (1.8 V) must settle into nominal voltage levels
before the VDDIO attains the minimum. This ensures that,
on power-up, the JTAG does not take control of the I/O
pins.
10kΩ
AD6636
Figure 51. EXT_FILTER Circuit for PLL Clock
•
By default, the PLL CLK is disabled. It can be enabled by
programming the PLL multiplier and divider bits in the
ADC CLK control register. When the PLL CLK is enabled
by programming this register, it takes about 50 to 200 µs to
settle down. While the PLL loop settles down, the voltage at
the EXT_FILTER pin increases from 0 V to VDDCORE (1.8
V) and settles there. Channel registers and output port
registers (Addresses 0x68 to 0xE7) should not be programmed before the PLL loop settles down.
•
The LVDS_RSET pin is used to calibrate the current in the
LVDS pads. The recommended circuit for this pin is shown
in Figure 52. This resistor should be placed as close as
possible to the AD6636 part. This resistor is not required, if
CMOS mode input is used.
LVDS_RSET
10kΩ
AD6636
04998-0-052
•
0.01µF
Figure 52. LVDS_RSET Circuit for LVDS Calibration
3.3V
To reset the AD6636 part, the user needs to provide a
minimum pulse of 30 ns to the RESET pin. The RESET pin
should be connected to GND (or pulled low) during powerup of the part. The RESET pin can be pulled high after the
power supplies have settled to nominal values (1.8 V and 3.3
V). At this point, a pulse (pull low and high again) should be
provided to give a RESET to the part.
•
Most AD6636 pins are driven by both JTAG circuitry and
normal function circuitry specific to each pin. TRST is the
reset pin for JTAG. When TRST is pulled low, JTAG is in
reset and all pins function in normal mode (driven by
functional circuit). If JTAG is not used in the design, the
TRST pin should be pulled low at all times.
DTACK (RDY, SDO)
AD6636
04998-0-050
1kΩ
•
Figure 50. DTACK, SDO Pull-Up Resistor Circuit
•
A simple RC circuit is used on the EXT_FILTER pin to
balance the internal RC circuit on this pin and maintain a
good PLL clock lock. The recommended circuit is shown in
Figure 51, with the RC circuit connected to VDDCORE.
This RC circuit should be placed as close as possible to the
AD6636 part. This layout ensures that the PLL clock is void
of noise and spurs and the PLL lock is maintained closely.
Rev. 0 | Page 70 of 72
AD6636
The number of clock cycles required for each channel can
be 3 (interleaved I + Q + gain word), or 2 (parallel I /Q +
gain) or 2 (interleaved I + Q) or 1 (interleaved I/Q).
Designers should make sure that sufficient time is allowed
to output these channels on one output port. Also note that
the I, Q, and gain for a particular channel all come out on a
single output port and cannot be divided among output
ports.
If JTAG is used, the designer should ensure that the TRST
pin is pulled low during power-up. After the power
supplies have settled to nominal values (1.8 V and 3.3 V),
the TRST pin can be pulled high for JTAG control. When
JTAG control is no longer required, the TRST pin should
ideally be pulled low again.
•
•
The CPUCLK (SCLK) is the clock used for programming
via the microport (serial port). This clock needs to be
provided by the designer to the part (slave clock). The
designer should ensure that this clock’s frequency is less
than or equal to the frequency of the CLKA signal.
Additionally, the frequency of the CPUCLK (SCLK) should
always be less than 100 MHz.
CLKA, CLKB, CLKC, and CLKD are used as individual
clocks to input data into Input Ports A, B, C, and D,
respectively. All these clocks are required to have same
frequency and should ideally be generated from the same
clock source. Note that CLKA is used to drive the internal
circuitry and the PLL clock multiplier. Therefore, even if
Input Port A is not used, CLKA should be driven by the
input clock.
•
The microport data bus is 16 bits wide. Both 8-bit and 16 bit
modes are available using this part. If 8-bit mode is used, the
MSB of the data bus (D[15:8]) can be left floating or
connected to GND.
•
The output parallel port has a one clock cycle overhead. If
two channels (with the same data rates) are output on one
output port in 16-bit interleaved I/Q mode along with an
AGC word, this requires three clock cycles for one sample
from each channel (one clock each for I data, Q data, and
gain data). Therefore, the total number of clock cycles
required to output the data is 3 clocks/channel × 2 channels
+ 1 (overhead) = 7 clock cycles.
•
When CRCF and DRCF filters are disabled, the coefficient
memory cannot be read back, because the clock to the
coefficient RAM is also cut off.
•
In the Intel mode microport, the beginning of a read and
write access is indicated by the RDY pin going low. The
access is complete only when the RDY pin goes high. In the
Motorola mode microport, the completion of a read and
write access is indicated by the DTACK going low. In both
modes, CS, RD (DS), and WR (R/W) should be active until
access is complete; otherwise, an incomplete access results.
•
In both Intel and Motorola modes, if CS is held low even
after microport read or write access is complete, the
microport initiates a second access. This is a problem while
writing or reading from coefficient RAM, where each access
writes to or reads from a different RAM address. This can be
fixed by writing to one coefficient RAM address at a time,
that is, the coefficient start and stop address registers have
the same value.
•
In SPI mode programming, the SCS pin needs to go high
(inactive) after writing or reading each byte (eight clock
cycles on the SCLK pin).
Rev. 0 | Page 71 of 72
AD6636
OUTLINE DIMENSIONS
A1 CORNER
INDEX AREA
17.20
17.00 SQ
16.80
BALL A1
CORNER
16
14
15
12
13
10
11
8
9
6
7
4
5
2
3
1
A
B
C
D
E
F
G
H
J
K
L
M
N
P
R
T
15.00
BSC SQ
TOP VIEW
1.00
BSC
BOTTOM VIEW
1.85*
1.71
1.40
DETAIL A
DETAIL A
1.31*
1.21
1.10
0.30 MIN*
SEATING
PLANE
0.70
COPLANARITY
0.60
0.20
0.50
BALL DIAMETER
COMPLIANT TO JEDEC STANDARDS MO-192-AAF-1
EXCEPT FOR DIMENSIONS INDICATED BY A "*" SYMBOL.
Figure 53. 256-Lead Chip Scale Ball Grid Array [CSP_BGA]
(BC-256-2)
Dimensions shown in millimeters
ORDERING GUIDE
Model
AD6636BBCZ1
AD6636CBCZ1
AD6636BC/PCB
1
Temperature Range
−40°C to +85°C
−40°C to +85°C
Package Description
6-Channel Part, 256-Lead CSP_BGA
4-Channel Part, 256-Lead CSP_BGA
Evaluation Board with AD6636 (6-Channel Part) and Software
Z = Pb-free part.
© 2004 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D04998–0–8/04(0)
Rev. 0 | Page 72 of 72
Package Option
BC-256-2
BC-256-2
PCB assembled