Technical Information for Bipolar Semiconductors

AN2012-01
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Technical Information
Infineon Technologies Bipolar GmbH & Co. KG …for energy efficiency!
Bipolar Semiconductors
Published by
Infineon Technologies Bipolar GmbH & Co. KG
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Order Number: B157-H9716-X-X-7600
Date: 04 / 2012
Attention please!
The information given in this document shall in no event
be regarded as a guarantee of conditions or characteristics
(“Beschaffenheitsgarantie”). With respect to any examples
or hints given herein, any typical values stated herein and/
or any information regarding the application of the device,
Infineon Technologies hereby disclaims any and all warranties and liabilities of any kind, including without limitation warranties of non-infringement of intellectual property
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Products and Innovations
The goal of highest reliability and efficiency in a core technology is always a moving
target; therefore we understand that continuous improvement is essential. On this basis
we have established comprehensive standards with our technologies and our products,
in the power classes ranging from around 10kW to over 30MW per component. These
include for example:
■
■
■
■
PowerBLOCK modules in press-pack technology with currents up to 1100 Ampere
Diodes and thyristors with a silicon diameter up to six inches and blocking voltages
up to 9500 Volts
Light-triggered thyristors with integrated protection functions
Freewheeling diodes for the highest requirements in fast switching applications
such as with IGBTs or IGCTs
600A/9.5 kV Thyristor Technology
for Soft Starter and Power-Supplies
The 9.5 kV thyristor disc is developed and designed for the special requirements in
medium voltage soft starter as well as for medium voltage power supply applications.
For these kinds of applications it is necessary to use several thyristors in series connection. They are optimized to achieve an excellent voltage sharing under all operating
conditions.
The device is designed for a high surge current capability. To ensure a narrow spread of
dynamic parameters which enables best cost designs with less devices in series high
technology production processes are used for this type.
Of course the thyristor is suitable for general purpose line voltage rectifier applications,
e.g. for power supplies or standard electrical drives.
Contents
1. Introduction
1.1Diode
1.2Thyristor
2. Type and polarity designation
2.1 Designation of the terminals
2.2 Constructions 2.2.1General
2.2.2 Disc cells
2.2.3PowerBLOCK-Module
2.2.4 Stud type and flat case constructions
3. Electrical properties
6
6
7
9
9
9
9
9
9
10
11
3.1 Forward direction11
3.1.1 Forward off-state current iD
3.1.2 Forward off-state voltage vD
12
12
3.1.3 Forward breakover voltage V(BO)
3.1.4 Open gate forward breakover voltage V(BO)0
3.1.5 Holding current IH
3.1.6 Latching current IL
3.1.7 On-state current iT, ITAV, ITRMS iF, IFAV, IFRMS
3.1.8 On-state voltage vT, vF
3.1.9 On-state characteristic
3.1.10 Equivalent line approximation with VT(TO), VF(TO) and rT
3.1.11 Maximum average on-state current ITAVM, IFAVM
3.1.12 Maximum RMS on-state current ITRMSM, IFRMSM
3.1.13 Overload on-state current IT(OV), IF(OV)
3.1.14 Maximum overload on-state current IT(OV)M, IF(OV)M
3.1.15 Surge on-state current ITSM, IFSM
3.1.16 Maximum rated value ∫i²dt
13
13
13
13
13
14
14
14
15
15
15
15
17
17
3.1.2.1 Repetitive peak forward off-state voltage VDRM
3.1.2.2 Non-repetitive peak forward off-state voltage VDSM
3.1.2.3 Forward direct off-state voltage VD (DC) 12
13
13
3.2 Reverse direction
18
3.2.1 Reverse current iR
3.2.2 Reverse voltage vR
18
18
3.2.2.1 Repetitive peak reverse voltage VRRM
3.2.2.2 Non-repetitive peak reverse voltage VRSM
3.2.2.3 Direct reverse voltage VR(DC)
3.3 Control properties of thyristors
3.3.1 Positive gate control
3.3.1.1 Gate current iG
3.3.1.2 Gate voltage VG
3.3.1.3 Gate trigger current IGT
18
18
18
19
19
19
19
19
1
3.3.1.4
3.3.1.5
3.3.1.6
3.3.1.7
3.3.1.8
3.3.1.9
3.3.1.10
Gate trigger voltage VGT
Gate non-trigger current IGD
Gate non-trigger voltage VGD
Control characteristic
Control circuit
Minimum duration of the trigger pulse tgmin
Maximum permissible peak trigger current
19
19
19
20
20
23
23
3.4 Carrier storage effect and switching characteristics
24
3.4.1 Turn-on 24
3.4.1.2Thyristor
25
3.4.1.1.1
3.4.1.1.2
3.4.1.2.1
3.4.1.2.2
3.4.1.2.3
3.4.1.2.4
Peak value of the forward recovery voltage VFRM
On-state recovery time tfr
Gate controlled delay time tgd
Critical rate of rise of the on-state current (di/dt)cr
Repetitive turn-on current IT(RC)M
Critical rate of rise of off-state voltage (dv/dt)cr
3.4.2Turn-off
3.4.2.1
3.4.2.2
3.4.2.3
3.4.2.4
Recovery charge Qr
Peak reverse recovery current IRM
Reverse recovery time trr
Turn-off time tq
3.5 Power dissipation (losses)
3.5.1
3.5.2
3.5.3
3.5.4
Total power dissipation Ptot
Off-state losses PD, PR
On-state losses PT, PF
Switching losses PTT, PFT+PRQ
3.5.4.1 Turn-on losses PTT, PFT
3.5.4.2 Turn-off losses PRQ
3.5.5 Gate dissipation PG
3.6 Insulation test voltage VISOL 4. Thermal properties
4.1Temperatures
4.1.1
4.1.2
4.1.3
4.1.4
4.1.5
4.1.6
Junction temperature Tvj, Tvj max
Case temperature TC Heatsink temperature TH
Cooling medium temperature TA
Junction operating temperature range Tcop
Storage temperature range Tstg
4.2 Thermal resistances
4.2.1
4.2.2
4.2.3
4.2.4
4.2.5
4.2.6
4.2.7
2
24
3.4.1.1Diode
Internal thermal resistance RthJC
Thermal transfer resistance RthCH
Heatsink thermal resistance RthCA
Total thermal resistance RthJA
Transient internal thermal resistance ZthJC
Transient heatsink thermal resistance ZthCA
Total transient thermal resistance ZthJA
25
25
26
26
27
27
27
27
29
30
30
33
33
33
33
34
34
35
35
35
36
36
36
36
37
37
37
37
37
37
37
38
38
38
38
39
4.3Cooling39
4.3.1
4.3.2
4.3.3
4.3.4
Natural air cooling Forced air cooling
Water cooling
Oil cooling
39
39
39
39
5. 5. Mechanical properties 40
5.1 Tightening torque 40
5.2 Clamping force 40
5.3 Creepage distance 40
5.4 Humidity classification 40
5.5Vibration40
5.6UL-registration40
6. Notes for applications 41
6.1
6.2
6.3
6.4
6.5
Case non-rupture current
Thermal load cycling
Parallel connection
Series connection
Pulsed Power
6.5.1
6.5.2
6.5.3
6.5.4
Applications with DC
Current rise time at turn-on Zero crossing of current and voltage during turn-on
Turn-off with a high di/dt versus a negative voltage
41
41
42
44
46
46
46
47
47
7. Protection49
7.1 Overvoltage protection 49
7.1.1
7.1.2
7.1.3
7.1.4
Individual snubbering (RC-snubber)
Input snubbering for AC-controllers
Supply snubbers for line commutated converters
Additional options for protection versus energy intensive overvoltages
49
52
53
55
7.2 Overcurrent protection 56
7.2.1 Short-term protection with superfast semiconductor fuses 56
7.2.1.1 Selection of fuses 56
7.2.2.1
7.2.2.2
7.2.2.3
7.2.2.4
59
59
59
59
7.2.2 Further protection concepts: short-term protection of high power semiconductors 59
High speed DC-circuit breakers
Crowbar (electronic short circuit)
Line side circuit breaker
Blocking of trigger pulses
7.2.3 Long-term protection 59
7.2.4 Fully rated protection 59
7.3Dynamic current limiting with inductors in the load circuit 60
7.4 Reduction of interference pulses in the gate circuit 61
3
8.Mounting
62
8.1 Disc cases
62
8.1.1
8.1.2
8.1.3
8.1.4
Mounting of disc cells
62
Positioning the heatsinks
66
Connection of busbars66
Connection of the control leads 67
8.2 Stud cases
8.2.1
8.2.2
8.2.3
8.2.4
Mounting stud cases
67
Positioning the heatsinks67
Connection of busbars68
Connection of the control leads
68
8.3 Flat base cases
8.3.1
8.3.2
8.3.3
8.3.4
Mounting flat base devices
Positioning the heatsinks
Connection of busbars
Connection of the control leads
8.4PowerBLOCK-Modules
8.4.1
8.4.2
8.4.3
8.4.4
68
68
69
69
69
69
Mounting PowerBLOCK-modules
69
69
Positioning the heatsinks Connection of busbars70
Connection of the control leads
70
9.Maintenance
10.Storage
11. Type designation
12. Circuit topologies
A1. Abbreviations
A2. List of Figures
A3. List of tables A4. Conditions of use
4
67
70
70
71
72
74
76
78
79
Preface
Power semiconductors are the central components in converters technology.
Due to constant advancement these components find further use in ever new and more
complex applications.
Based on the suggestions and questions we have been approached with we compiled this
Technical Information (TI) as a reference document.
This Technical Information describes all essential technical terms for bipolar power
semiconductors (diodes and thyristors) and thus provides assistance in working and
designing as well as a reference document for the development and projection of inverter
circuitry with bipolar components.
It is aimed at the relevant specialists in industry, research, development and training.
General information regarding converters, their circuits and specialties can be found in
the pertinent literature.
At this point we refer to the appropriate standards which always need to be regarded in
their latest version.
The current technical data of Infineon power semiconductors can be down-loaded from
www.Infineon.com.
This Technical Information is meant to assist in better understanding the terms and the
application of data sheet specifications of bipolar power semiconductors.
Definitions and abbreviations used are mainly in accordance with DIN / IEC / EN.
Please note that no guaranty can be given that circuits, appliances and processes
described here are free of patent rights.
5
1. Introduction
This TI is to give detailed definitions to specifications used in the data sheets. Further,
the user is to be assisted to transfer the data sheet specifications correctly in his
application.
The following information is generally valid for all Infineon pressure contact components
(disc cells and PowerBLOCK-Modules). Exceptions are individually marked.
Information given here is valid in accordance with the currently valid norms and
standards.
1.1 Diode
Technischer Erläuterungen - Bilder
A diode is a component with one P and one N conducting semiconductor zone.
The PN-junction is responsible for the elementary features of this semiconductor
(see Figure 1).
Diode
Anode A
+
A
P
N
-
Kathode K
Cathode K
K
Figure 1: Schematic construction of a diode
The characteristic of a diode is depicted in Figure 2. It consists of two sections:
the blocking
characteristic
and on-state characteristic.
Technischer
Erläuterungen
- Bilder
iF
Durchlaßkennlinie
High conduction characteristic
Abb./Fig.1 Schematischer Aufbau einer Diode
vR
vF
Rückwärts-Sperrkennlinie
Reverse blocking characteristic
Durchlassrichtung
Forward direction
Sperrrichtung
Reverse Direction
iR
Figure
2 Characteristics
of a diode
Abb./Fig.2
Kennlinien
einer Diode
6
When a voltage up to several kV is applied in reverse direction, reverse currents in the range of mA will flow via the
main terminals anode and cathode.
When a voltage is applied in forward direction, currents up to several kA will flow via the main terminals anode and
cathode.
Technischer Erläuterungen - Bilder
1.2 Thyristor
A thyristor is a component with a total of four alternating P and N conducting semiconductor zones. These will thus
form three PN-junctions (see Figure 3).
Thyristor
+
Anode A
P
N
P
N
Steueranschluß G
Gate G
-
A
G
Kathode K
Cathode K
K
Figure 3: Schematic construction of a thyristor
The characteristics of a conventional (reverse blocking) thyristor are depicted in Figure 4. They consist of three
sections: The blocking and the on-state characteristic in forward direction and the blocking characteristic in
reverse direction.
Technischer Erläuterungen - Bilder
Abb./Fig.3 Schematischer Aufbau eines Thyristors
iT,iD
Durchlaßkennlinie
High conduction characteristic
Vorwärts-Sperrkennlinie
Forward blocking characteristic
IH
vR
V(BO)O
vD, vT
Rückwärts-Sperrkennlinie
Reverse blocking characteristic
Schaltrichtung
Forward direction
Sperrrichtung
Reverse Direction
iR
Abb./Fig.4
Kennlinien eines Thyristors
Figure
4 Characteristics
of a thyristor
As can be seen from the characteristics, the thyristor is initially blocked in forward and reverse directions. Generally
the blocking capability is approximately the same in both directions.
When voltages up to several kV are applied in forward or reverse direction, only small blocking currents will flow via
the main terminals anode and cathode. An additional control current IG between control terminal (gate) and cathode
7
will trigger the thyristor when a forward voltage vD is present, i.e. it turns on to the
on-state characteristic. However, it may not be turned off via the control terminal. Only
when the forward current by changes in the load circuit drops below the holding current
IH, the thyristor will once again block.
Fast thyristors are available in 2 basic versions:
n Symmetrically blocking thyristors
(SCR → Silicon Controlled Rectifier)
These thyristors show approximately equal blocking capability in both directions.
Individual types are differentiated by their blocking capability, their current carrying
capability, their turn-off time and the gate-cathode structure.
n Asymmetrically blocking thyristors
(ASCR → Asymmetric Silicon Controlled Rectifier)
These thyristors provide full blocking capability in forward direction and little blocking
capability in reverse. Here the reverse blocking PN-junction is replaced by a stop layer
which allows a significant reduction of the silicon height.
The advantages compared to symmetrically blocking thyristor are a shorter turn-off time
for the same on-state voltage or a lower on-state voltage for the same turn-off time.
8
2. Type and polarity designation
2.1 Designation of the terminals
Technischer Erläuterungen - Bilder
Technischer Erläuterungen - Bilder
Diode as disc cell,
ND or DZ-PowerBLOCK-Module
Thyristor as disc cell or
TZ-PowerBLOCK-Module
Diode als
Scheibenzelle
Technischer
Technischer
Erläuterungen
Erläuterungen
- Bilder
- Bilder
Thyristor als Scheibenzelle
Thyristor
als Scheibenzelle
Thyristor
als TZ-Powerblockmodul
Thyristor als TZ-Powerblockmodul
DiodeDiode
als Scheibenzelle
als DZ-Powerblockmodul
Diode als DZ-Powerblockmodul
Anode
Anode
Kathode
cathode
Kathode
cathode
Anode
Diode als Diode
Scheibenzelle
als Scheibenzelle
Diode als Diode
DZ-Powerblockmodul
als DZ-Powerblockmodul
Anode
Anode
Kathode
cathode
Kathode
cathode
Anode
Dioden als DD-Powerblockmodul
Dioden als DD-Powerblockmodul
Diodes as DD-PowerBLOCK-Module
Kathode1
Anode1
Anode1
Kathode1
cathode1
Anode2
cathode1
Anode2
Kathode2
cathode2
Dioden alsDioden
DD-Powerblockmodul
als DD-Powerblockmodul
Anode1
Anode1
Kathode1
cathode1
Kathode1
cathode1
Anode2
Anode2
Kathode2
cathode2
Abb./Fig.5 Bezeichnungen der Anschlüsse
Abb./Fig.5 Bezeichnungen der Anschlüsse
Anode
Kathode
cathode
Kathode
cathode
Thyristor als
Thyristor
Scheibenzelle
als Scheibenzelle
Thyristor als
Thyristor
TZ-Powerblockmodul
als TZ-Powerblockmodul
Steueranschluss
Hilfskathode
gate
aux. cathode
Steueranschluss
Hilfskathode
Kathode
Kathode
gate
aux.
cathode cathode
cathode
Anode
Thyristor als TT-Powerblockmodul
Thyristor als TT-Powerblockmodul
Thyristors as
TT-PowerBLOCK-Module
Kathode2
Kathode1
Steueranschluss
Steueranschluss
Hilfskathode HilfskathodeKathode2
cathode2
Anode1
Anode1
Kathode1
cathode1
Anode2
cathode1
gate
aux.gate
cathodeKathode2
aux. cathodecathode2
cathode2
Anode2
Thyristor als
Thyristor
TT-Powerblockmodul
als TT-Powerblockmodul
Hilfskathode2
Steueranschluss 2
Hilfskathode1
Steueranschluss 1
aux. cathode2
gate 2 Kathode2
Hilfskathode2
aux.
cathode1
Steueranschluss 2Kathode2
gate 1
Hilfskathode1
Kathode1
Kathode1
Kathode2 Steueranschluss 1
cathode2 Anode1 gate 1
Anode1
aux.
cathode1 cathode1
cathode1
Anode2
aux. cathode2
gate 2 cathode2
cathode2
Anode2
Hilfskathode2
Hilfskathode2
Steueranschluss
Steueranschluss
2
2
Hilfskathode1
Hilfskathode1
Steueranschluss
Steueranschluss
1
1
aux.
cathode2
aux. cathode2
gate
2
aux.
cathode1
aux. cathode1 gate 2
gate 1
gate
1
Abb./Fig.5 Bezeichnungen
Abb./Fig.5 Bezeichnungen
der Anschlüsse
der Anschlüsse
Figure 5 Designation of the terminals
2.2 Constructions
2.2.1 General
The semiconductor element (pellet) is built into a case and thus protected from adverse influences of the external
environment.
All semiconductors described here are constructed in pressure contact technology.
The pressure contact technology is known for:
n very high load cycling capability
n very good over-load capability
2.2.2 Disc cells
When mounting disc cells the pressure for the components has to be applied from the exterior. Double sided
cooling allows the heat generated through the losses to be dissipated in the best possible way from the disc cells.
They are thus used for applications with highest power requirements.
2.2.3 PowerBLOCK-Module
The PowerBLOCK-Module is a case concept which in itself provides sufficient pressure to the semiconductor
element. In addition, defined isolation against the base plate is provided. This simplifies the application of the
modules significantly, as a complete rectifier for example may be constructed on a common heatsink. Due to the
single sided cooling and the limits of the isolation voltage, possibilities of its application in the high power area
are limited.
9
Scheibenzelle
Disc case
PowerBLOCK-Modul
PowerBLOCK-module
Schnitt durch eine Scheibenzelle
Cross-sectional view of a disc
Aufbau eines PowerBLOCK-Moduls
Assembly of a PowerBLOCK-module
Figure 6 Construction concepts of pressure contact components
2.2.4 Stud type and flat case constructions
In stud (screw) type and flat case constructions the semiconductor element is already
pressed correctly. These case types are now out-dated and mostly replaced by the more
powerful PowerBLOCK-Module.
10
3. Electrical properties
The electrical properties of diodes and thyristors are temperature dependent and therefore valid only in conjunction
with a temperature specification.
All values mentioned in the data sheets are applicable to mains frequency 40 to 60Hz if not otherwise specified.
Maximum values are those values given by the manufacture as the absolute limits which generally even for short
times may not be exceeded as this may lead to a functional deterioration or destruction of the components.
Characteristic values are ranges of data distribution at defined conditions and may form the basis of incoming
inspection.
3.1 Forward direction
For diodes
the forward direction is the direction between the main terminals in which the diode has reached conduction mode
even at a low voltage of just a few volts (see Figure 1, direction anode-cathode).
For thyristors
the forward direction is the direction between the main terminals in which the thyristor may operate in two stable
modes – the on- and the off-state - (see Figure 3, direction anode-cathode).
Addition of the words “positive” or “forward” is used to expressly distinguish currents and voltages in forward
direction from those in reverse direction.
The forward characteristic of the thyristor consists of an off-state and an on-state region (see Figure 4).
The forward off-state characteristic is that part of the forward characteristic of a thyristor which illustrates the
instantaneous values of the forward off-state current and the forward off-state voltage.
11
1
Technischer Erläuterungen - Bilder
0,9
vD
Technischer Erläuterungen - Bilder
ID,R (VDRM,RRM; Tvj) / ID,R (VDRM,RRM; Tvjmax) = 0,96 (Tvj max - Tvj)
0,8
ID,R (VDRM,RRM;Tvj) / ID,R (VDRM,RRM; Tvj max)
ID,R (VDRM,RRM;Tvj) / ID,R (VDRM,RRM; Tvj max)
Technischer Erläuterungen - Bilder
0,7
0,6
0,5
0,4
0,3
0,2
0,1
VDSM
VDRM
1
0,9
VDWM
ID,R (VDRM,RRM; Tvj) / ID,R (VDRM,RRM; Tvjmax) = 0,96 (Tvj max - Tvj)
0,8
0,7
0,6
0,5
t
0,4
0,3
0,2
VRWM
0,1
VRRM
0
0,5
0
0,5
0,55
0,55
0,6
0,6
0,65
0,65
0,7
0,75
Tvj / Tvj max
0,7
0,8
0,75
0,85
Tvj / Tvj max
0,8
0,9
0,95
0,85
0,9
Figure 7 Typical dependence of the off-state current iD,R(VDRM,RRM) referenced to ID,R(VDRM,RRM; Tvj max)
Abb./Fig.7 Typische T - Abhängigkeit des auf I , (V
;T
) normierten Sperrstroms
on the junction temperature Tvj referenced to Tvj max
DR
DRM, RRM
0,95
1
vR
Abb./Fig.8 Definition der Sperrspannungsbelastungen
Abb./Fig.7 Typische Tvj - Abhängigkeit des auf ID,R(VDRM, RRM ;Tvj max) normierten Sperrstroms
vj
VRSM
1
Figure 8 Definition of the off-state voltage occurrences
vj max
3.1.1 Forward off-state current iD
iD is the current which flows in forward direction through the main terminals in the
off-state condition of the thyristor. In the data sheet it is specified for the voltage VDRM
and the maximum junction temperature Tvj max.
This current depends on the junction temperature Tvj (see Figure 7).
3.1.2 Forward off-state voltage vD
vD is the voltage which is applied across the main terminals in forward direction during
the off-state condition of the thyristor.
3.1.2.1 Repetitive peak forward off-state voltage VDRM
VDRM is the maximum value of repetitive voltages in the forward off-state direction
including all repetitive peak voltages.
In DC applications a reduction to VD (DC) is necessary. See also section 3.1.2.3.
In view of transient voltages occurring in operation, thyristors are usually operated at
supply voltages of which the peak value is equal to the maximum rated repetitive peak
off-state voltage divided by a safety factor of between 1.5 and 2.5.
ˆ
V
line
=
V
V
DWM,RWM
bzw.
DRM
V
RRM
1,5...2,5
A low safety factor is used where the transient voltages mostly known. These are
generally self commutated converters with large energy storage. For converters supplied
from mains with unknown transient levels a safety voltage margin of 2.0 to 2.5 is
preferable.
12
If transient voltages are likely to occur in operation, which exceed the maximum permissible repetitive peak off-state
voltage, a suitable transient voltage protection network has to be provided (see 7.1).
3.1.2.2 Non-repetitive peak forward off-state voltage VDSM
VDSM is the maximum rated non-repetitive peak value of a voltage in forward direction on the thyristor which must
not be exceeded.
3.1.2.3 Forward direct off-state voltage VD (DC)
VD (DC) is the permanently allowable direct voltage in forward direction in off-state mode. For the semiconductors
described here the value is rated at approximately half repetitive peak off-state voltage. This is valid for a failure
probability of approximately 100 fit (failure in time; 1fit = 1*10-9 failures per hour, i.e. one failure in 109 operating
hours of the device). Probabilities of failure to be expected for varying DC-voltages are available on request.
3.1.3 Forward breakover voltage V(BO)
V(BO) is the value of the off-state voltage in forward direction at which for a given gate current the thyristor switches
from the off-state to the on-state.
Exception: For light triggered thyristors (LTT’s) with integrated breakover diode (BOD) V(BO) is the minimum voltage at
which protective triggering of the thyristor occurs
3.1.4 Open gate forward breakover voltage V(BO)0
V(BO)0 is the breakover voltage at zero gate current. Triggering the thyristor by exceeding the V(BO)0 may cause
destruction of the device.
Exception: Light triggered thyristors are protected by an integrated breakover diode (BOD).
3.1.5 Holding current IH
IH is the minimum value of on-state current required to maintain the thyristor in on-state. IH drops with raising
junction temperature (see Figure 9).
Light triggered thyristors show a significantly lower holding current than comparable electrically triggered thyristors.
3.1.6 Latching current IL
IL is the on-state current required to maintain the thyristor in the on-state once the gate current has decayed. It
depends on the rate of change, peak and duration of the gate current as well as on the junction temperature (see
Figure 9).
Exception: Light triggered thyristors show a significantly lower latching current than comparable electrically triggered
thyristors.
3.1.7 On-state current iT, ITAV, ITRMS iF, IFAV, IFRMS
The on-state current is the current which flows via the main terminals in the on-state of the thyristor (iT, ITAV, ITRMS) or
the diode (iF, IFAV, IFRMS). It is differentiated in:
iT, iF = instantaneous value
ITAV, IFAV = average value
ITRMS, IFRMS = RMS (route mean square)
13
Technischer Erläuterungen - Bilder
2
IH(Tvj) / IH(25°C), I L(Tvj) / IL(25°C)
1,8
IH
1,6
1,4
1,2
IL
1
IL
0,8
0,6
IH
0,4
0,2
0
-40
-20
0
20
40
60
80
100
120
140
Tvj [°C]
Figure 9 Typical dependence of the latching current IL and holding current lH normalized to Tvj=25°C of the junction
temperature Tvj
Abb./Fig.9 Typische Tvj - Abhängigkeit
des auf 25°C normierten Einraststroms I und Haltestroms I
L
H
3.1.8 On-state voltage vT, vF
vT, vF is the voltage across the main terminals at the defined on-state current. It depends
on the junction temperature. Values given in the data sheet are valid for the completely
turned on thyristor (vT) or for the diode (vF).
3.1.9 On-state characteristic
The on-state characteristic is the relation of the instantaneous values of on-state current
and on-state voltage for the diode or for the completely turned on thyristor at a defined
junction temperature.
3.1.10 Equivalent line approximation with VT(TO), VF(TO) and rT
The equivalent line is an approximation to the on-state characteristic of a thyristor (VT(TO),
rT) or of a diode (VF(TO), rT) to calculate the on-state power dissipation.
Given are:
VT(TO), VF(TO) = threshold voltage
rT = differential resistance or slope resistance
The value of VT(TO), VF(TO) results from the intersection of the equivalent line
approximation and the voltage axis, the value of rT is calculated from the rate of raise
of the equivalent line. Depending on the cooling it may be necessary to adapt the
equivalent lines shown in the data sheet to the application. In some data sheets there
may hence be an additional low level value for VT(TO), VF(TO)and rT.
For components with high blocking voltages (T…1N, T…3N, D…1N) equivalent lines are
shown in addition as an approximation to a typical on-state characteristic which
describes approx. the 50% value in the statistical distribution. In applications in which
many equal components are used the conduction losses of the entire installation can be
calculated using the typical equivalent line approximation.
14
4000
3500
3500
3000
3000
2500
2500
iT [A]
iT [A]
4000
2000
ΔiT
2000
1500
1500
1000
1000
500
500
ΔvT
ΔiT
rT = Δ
vT
0
0
0
0,5
1
1,5
2
2,5
3
3,5
0
vT [V]
0,5
vT0
1
1,5
2
2,5
3
3,5
vT [V]
Figure10 Example of an on-state characteristic and the matching equivalent line approximation
3.1.11 Maximum average on-state current ITAVM, IFAVM
ITAVM, IFAVM is the maximum permissible continuous average value of the on-state current in a single phase half-wave
resistive load circuit according to DIN VDE 0558, part 1 rated at a defined case temperature TC and a frequency of 40
to 60Hz.
A diagram is given in the data sheets of the thyristors or diodes with low blocking voltages which shows the
maximum average on-state current versus the maximum allowable case temperature TC for various current
conduction angles.
This diagram takes only the conduction losses into account. For components with high blocking voltages (>2200V)
additional turn-off losses and to some degree blocking and turn-on losses need to be considered.
For components with very high blocking voltages (>4kV) this diagram is, therefore, omitted in the data sheet.
3.1.12 Maximum RMS on-state current ITRMSM, IFRMSM
ITRMSM, IFRMSM is the maximum value of RMS on-state current permissible considering electrical and thermal stresses
of all assembly parts of the device. This current must not be exceeded for flat base and stud type cases and modules
even under the best cooling conditions of the thyristor (ITRMSM) or the diode (IFRMSM).
3.1.13 Overload on-state current IT(OV), IF(OV)
IT(OV), IF(OV) is the maximum allowable value of on-state current that the thyristor (IT(OV)) or the diode (IF(OV)) may
conduct in short time operation without losing its control property. In the diagram for overload on-state current it is
given as the peak value at 50Hz sinusoidal half-waves for different preloads versus time t.
This illustration does not take into account increased blocking or turn-off losses as they occur for devices with high
blocking voltages. For components with very high blocking voltages (>4kV) this diagram is, therefore, omitted in the
data sheet.
3.1.14 Maximum overload on-state current IT(OV)M, IF(OV)M
IT(OV)M, IF(OV)M is the value of on-state current at which the device must be turned off in order not to destroyed the
thyristor (IT(OV)M) or the diode (IF(OV)M). These values are intended for the design of the protection networks. The
thyristor may temporarily lose its forward blocking capability when the current flowing through it reaches this value
and may temporarily lose its control properties.
The maximum overload on-state current characteristic shows this value as the peak value of a 50Hz sinusoidal
half-wave versus time t. Two conditions are differentiated: no load operation preceding and operation with
maximum average on-state currents preceding.
15
The maximum overload on-state current characteristics given in the individual data
sheet applies to a reverse blocking voltage of 80% of the repetitive peak reverse voltage.
In cases where the actual reverse voltage is lower, a higher maximum overload on-state
current is allowable which is shown in Figure 11 and Figure 12 for a preceding
continuous maximum overload on-state current ITAVM. The conditions for a device
without preceding load can not be determined from this.
This illustration does not take into account increased blocking or turn-off losses as they
occur for devices with high blocking voltages. For components with very high blocking
voltages (>4kV) this diagram is, therefore, omitted in the data sheet. The protection
Technischer
Erläuterungen
- Bilder
concepts for
these devices
are described in chapter 7.2.
1
0,9
0,8
IT(OV)M / ITSM
0,7
0,6
VRM =
0,5
0-50 V
0,4
0,33 VRRM
0,3
0,67 VRRM
0,2
0,1
0
0
20
40
60
80 100 120 140 160 180 200 220 240 260 280 300
t [ms]
Figure 11 Typical dependence of the maximum overload on-state current IT(OV)M, IF(OV)M (in relation to the surge
current ITSM or IFSM for 10ms and Tvj max) on the number of half-sinewaves at 50Hz.
Parameter:
voltage VIT(OV)M
Abb./Fig.10 Typische
Abhängigkeit reverse
des auf ITSMblocking
normierten Grenzstroms
RM von t für mehrere 50Hz Halbwellen Paramter: VRM
Figure 12 Typical dependence of the maximum overload on-state current IT(OV)M, IF(OV)M (in relation to the surge
current ITSM or IFSM for 10ms and Tvj max) on the time t for a number of half-sinewaves at 50Hz. Parameter: reverse blocking voltage VRM
16
3.1.15 Surge on-state current ITSM, IFSM
ITSM, IFSM is the maximum permissible peak value of a single half sine-wave 50Hz current pulse. It is specified at 25°C
(equates to a short circuit from no load condition) or at
turn-on at maximum permissible junction temperature (equates short circuit after
permanent load with maximum permissible current). When stressing a semiconductor with the surge on-state
current, the device loses its blocking capability. Therefore, no negative voltage shall subsequently be applied. This
stress may be repeated during fault conditions in a non-periodic way provided the junction temperature has dropped to
values within the permissible operating temperature area.
When exceeding the maximum permissible value destruction of the device is risked (for details please see chapter
7.2 over current protection).
3.1.16 Maximum rated value ∫i²dt
∫i²dt is the square of the surge on-state current integrated over time.
The maximum rated ∫i²dt-value serves to determine the short-circuit protection (see 7.2).
For half-sinewaves with periods shorter than 10ms the maximum rated ∫i²dt-value is shown in Figure 13. Regarding voltage stress and repetition the same applies as for the surge on-state current. When exceeding the maximum
permissible value, destruction of the device is risked. In addition, in particular for large diameter thyristors, it has to
be observed that the permissible critical turn-on current rate of change (di/dt)cr may not be exceeded.
Technischer Erläuterungen - Bilder
1
0,9
∫i² dt ( tP ) / ∫i² dt (10ms)
0,8
a
Thyristoren / Thyristors
a: PB20, PB34, PB50
TO42, TO48, TSW, TFL
b: PB60
TO58
c: PB70
≥TO75
0,7
0,6
b
0,5
Dioden / Diodes
0,4
b: VDRM,RRM ≥ 1000V
c: VDRM,RRM < 1000V
High Power T…1N, T...3N, D…1N
c: für alle Typen, for all types
c
0,3
High Power D…1N, T…1N
c: für alle Typen, for all types
0,2
0,1
0
0
1
2
3
4
5
6
7
8
9
10
tP [ms]
Abb./Fig.12 Typische Abhängigkeit des auf i² dt (10ms) normierten Grenzlastintegrals i² dt von der Halbschwingungsdauer tP
Figure 13 Typical dependence of the òi² dt normalized to the value òi² dt (10ms) on the half-sinewave duration tP
17
3.2 Reverse direction
The reverse direction is the direction from one main terminal to the other in which
the thyristor and diode is in a stable high resistance state of operation (direction
cathode-anode).
If values (voltages and currents) and data in reverse direction are to be distinguished
from those in forward direction, then the term “reverse” or “negative” is used.
The reverse blocking characteristic of a thyristor or a diode represents the instantaneous
values of reverse current and reverse voltage.
3.2.1 Reverse current iR
iR is the current flowing in reverse direction through the main terminal of the thyristor or
diode. The reverse current depends on the reverse voltage and the junction temperature
Tvj (Figure 7)
3.2.2 Reverse voltage vR
VR is the voltage applied across the main terminals of the thyristor or diode in reverse
direction.
3.2.2.1 Repetitive peak reverse voltage VRRM
VRRM is the maximum permissible instantaneous value of repetitive voltages in reverse
direction including all repetitive peak voltages.
In DC applications a reduction to VR (DC) is necessary.
See also section 3.2.2.3.
For supply voltage see section 3.1.2.1.
3.2.2.2 Non-repetitive peak reverse voltage VRSM
VRSM is the maximum allowable non-repetitive peak value of a transient voltage in
reverse direction which must not be exceeded even for the shortest duration. The value
resulting is:
For blocking voltages < 800V:
VRSM = VRRM + 50V (at Tvj = 25°C ... Tvj max)
For blocking voltages ≥ 800V:
VRSM = VRRM + 100V (at Tvj = 25°C ... Tvj max)
3.2.2.3 Direct reverse voltage VR(DC)
VR (DC) is the permanently allowable direct voltage in reverse direction, analogous to
forward direct off-state voltage 3.1.2.3.
18
3.3 Control properties of thyristors
3.3.1 Positive gate control
3.3.1.1 Gate current iG
iG is the current flowing through the control path (terminals G – HK).
Thyristors shall only be pulse triggered during the forward off-state phase.
Positive trigger pulses during the reverse off-state phase will lead to significantly increased off-state losses due to
the transistor effects caused. These losses adversely affect the functionality and may lead to destruction.
Exception: For light triggered thyristors control pulses during the reverse off-state phase are permissible.
3.3.1.2 Gate voltage VG
VG is the positive voltage across the gate terminal (G) and the cathode (K) or auxiliary cathode (HK).
3.3.1.3 Gate trigger current IGT
IGT is the minimum value of gate current which causes the thyristor to trigger. It depends on the voltage across the
main terminals and the junction temperature. At the given value of the gate trigger current all thyristors of a given
type will trigger. The gate trigger current increases with lower junction temperature and is thus specified at 25°C.
The trigger pulse generator has to safely exceed the data sheet value IGTmax (see also 3.3.1.8).
Exception: For light triggered thyristors the minimum light power PL is specified which causes all thyristors of a given
type to trigger.
3.3.1.4 Gate trigger voltage VGT
VGT is the voltage which occurs across gate terminal and cathode when the gate trigger current IGT flows. It depends
on the voltage across the main terminals and the junction temperature. At the given value of the gate trigger voltage
all thyristors of a given type will trigger. The gate trigger voltage drops with increasing junction temperature and is
thus specified at 25°C. VGT is measured when a specified load current flows.
3.3.1.5 Gate non-trigger current IGD
IGD is the value of the gate current which does just not cause the thyristor to trigger. It depends on the voltage across
the main terminals and the junction temperature. At the given maximum value no thyristor of a given type triggers.
The gate non-trigger current decreases with increasing junction temperature and is thus specified at Tvj max.
3.3.1.6 Gate non-trigger voltage VGD
VGD is the value of the gate voltage which does just not cause the thyristor to trigger.
It depends on the voltage across the main terminals and the junction temperature. At the given maximum value no
thyristor of a given type triggers. The gate non-trigger voltage decreases with increasing junction temperature and is
thus specified at Tvj max.
19
Technischer Erläuterungen - Bilder
100
10
c
vG [V]
b
1
0,1
10
Tvj =
-40 °C
Tvj =
+25°C
Tvj =
+125°C
a
100
iG [mA]
1000
10000
Figure 14 Example for control characteristic vG = f (iG) with trigger area for VD = 12 V
3.3.1.7 Control characteristic
Abb./Fig.13 Steuercharakteristik vG = f (iG) mit Zündbereichen für VD = 12 V
It shows the limits of statistical distribution of the input characteristics of a thyristor
type. Within the distribution of the input characteristics the temperature dependent
trigger areas are detailed as well as the curves of the maximum permissible gate power
dissipation PGM (a – 20W / 10ms, b – 40W / 1ms, c – 60W / 0.5ms).
3.3.1.8 Control circuit
In a normal application the design of the control circuit should be done in accordance
with the control data which are detailed in connection with the critical rise time of the
on-state current, the gate control delay time and the latching current (see Figure 15). The
minimum control data given in 3.3.1.3 and 3.3.1.4 are valid only for applications with
low requirements with regard to critical current rise time and gate control delay time. In
reality overdriving IGT specified in the data sheet 4- to 5-fold assures safe operation even
with high requirements for current rise time and gate control delay time.
Terms used in this context are:
diG/dt = gate current slew rate
iGM = peak gate current
tG = duration of the trigger pulse
VL = open circuit voltage of the control circuit
With increasing slew rate of the on-state current diT/dt as well as repetitive turn-on
current IT(RC)M from the snubber an effect from the load circuit to the gate current iG is
notable (see 3.4.1.2 and Figure 21).
20
Technischer Erläuterungen - Bilder
A
RG 2
CG
iG
+
vC
=
–
vG
RG 1
Steuerelektronik
control circuit
G
HK
RGK
K
Abb./Fig.14
Prinzipschaltbild
eines Steuergenerators
für Thyristoren
Figure
15 Concept
of a trigger
circuit for thyristors
Initially there is only a small area around the gate area on the pellet conductive during turn-on of the thyristor which
leads to high current density and increased voltage. Due to internal coupling this voltage also appears at the control
terminals and, therefore, leads to an intermediate drop of the gate trigger current. In order to avoid the possible
destruction of the thyristor, iG should not drop below the value of the gate trigger current IGT. To prevent the gate
pulse from dropping too low, a compensation by means of a higher open circuit voltage VC of the trigger circuit may
be necessary. For parallel or series connection of thyristors high, steep rising and synchronous trigger pulses are
necessary in order to achieve equalised turn-on. See also distribution of gate control delay time values (3.4.1.2.1).
Exception: To control light triggered thyristors, laser diodes emitting light in the region of 900 to 1000nm are
required. Minimum values for light power PL are given which in conjunction with the given turn-on voltage will assure
safe triggering of the thyristors. The light power is specified at the output of the fibre optic cable. With regard to
even turn-on here too overdriving is recommended in particular for parallel and series connection with high di/dt
requirements.
Infineon recommends the application of the laser diodes SPL PL90 aligned in the appropriate fitting (see Figure 16)
and offers these together with suitable fibre optic cables as ancillary equipment.
Figure 16 LTT with fibre optic cable
21
The laser diodes SPL PL 90 comply with the following laser classes:
If the laser diode is correctly terminated with the fibre optic cable
the control system complies with laser class 1. No operational
hazard.
With open operation of the laser diode or in case of a broken fibre
optic cable, the system equates to the laser class 3b according
to IEC 60825–1. In this case hazard of operation exists due to
invisible radiation. Direct or indirect exposure to the eyes or skin
is to be avoided.
Technischer Erläuterungen - Bilder
250
200
PL [mW]
150
100
50
0
500
600
700
800
900
1000
ILaserdiode [mA]
1100
1200
1300
1400
Figure 17 Laser diode SPL PL 90 typical dependence of the light power on the control current
Abb./Fig.16 Laserdiode SPL PL 90 typ. Abhängigkeit der Lichtleistung vom Steuerstrom
To control light-triggered thyristors, we recommend a current pulse for the laser diode
SPL PL90 as in Figure 18. As the laser diode SPL PL90 is not suitable for long-term
control, we recommend controlling the laser diode with a frequency of approximately
6kHz, while using the pulse in Figure 18.
22
Figure 18 Recommended current pulse for laserdiode SPL PL 90
3.3.1.9 Minimum duration of the trigger pulse tgmin
The trigger pulse has to be applied at least until the latching current of the thyristor (3.1.6) has been exceeded, as
otherwise the thyristor will return to its off-state. The gate trigger current of the thyristor must remain at least at its
rated value until the end of the trigger pulse.
In applications with very low current rise times or low load currents often a trigger profile with multiple pulses is
used (e.g. with a frequency of repetition of 6kHz).
For light triggered thyristors make sure that when using multiple pulses the laser diode does not heat up
inadmissibly. The light power of a current controlled laser diode drops with increasing temperature.
3.3.1.10 Maximum permissible peak trigger current
In applications with a high rate of rise of current iGT may be overdriven even harder than described in 3.3.1.8. For
this the gate current should be increased for a time tGM ≈ 10-20µs to the 8- to 10-fold value of IGT and than continue
for a sufficient time tG with a reduced amplitude. The open circuit voltage of the trigger circuit should at least apply
30V in order to assure a high reactionless gate current.
23
Technischer Erläuterungen - Bilder
iG
IGM ≈ 8-10 IGT
≤
IGM
0.5-1µs
IG ≈ 2-4 IGT
tGM
t
100µs < tG< tP
Figure 19 Safe overdrive of the gate trigger current
3.4 Carrier storage effect and switching characteristics
When the state of operation changes in power semiconductors, the stationary values
of current
and voltage
do not change immediately due to the carrier storage effect.
Abb./Fig.17 Zeitlicher
Verlauf des empfohlenen
Gateimpulses
Additionally, in thyristors only small areas around the gate structure become conductive
when triggered. The switching losses resulting from this have to be dissipated as heat
from the semiconductor.
3.4.1 Turn-on
3.4.1.1 Diode
When passing from a non-conducting or blocked state to a conducting state, voltage
peaks occur at the diode due to the carrier storage effect (see Figure 20).
Technischer Erläuterungen - Bilder
VFRM
VF, iF
90%
IFM
50%
diF/dt
vF
0,1 vF
tfr
Abb./Fig.18
Schematische
Darstellung
des Einschaltvorgangs
Dioden turn-on process
Figure
20 Schematic
representation
of von
a diode
24
t
3.4.1.1.1 Peak value of the forward recovery voltage VFRM
VFRM is the highest voltage value occurring during the forward recovery time (see Figure 20). It increases with rising
junction temperature and current slew rate.
In mains operation (50 / 60Hz) with its moderate current slew rates VFRM is negligible. In self-commutated converters
with fast switches di/dt>>1000A/us (IGBT’s, GTO’s and IGCT’s), however, it may reach values up to several hundred
volts. Although the forward recovery voltage exists for just a few microseconds and thus does not contribute to the
sum of losses of the diodes in a significant way, its effect on the switching semiconductor has to be considered
when designing the converter.
In diagrams for diodes optimized for these applications data is included which details the forward recovery voltage
as a function of the current slew rate.
3.4.1.1.2 On-state recovery time tfr
According to DIN IEC 60747-2 tfr is the time the diode needs to become fully conducting and a static on-state voltage
vF appears, when suddenly switched from zero to a defined on-state (see Figure 20).
3.4.1.2 Thyristor
The turn-on process is initiated at forward off-state voltage vD by a gate current with a slew rate diG/dt and a
magnitude iGM. For light triggered thyristors this applies to an equally specified trigger pulse on the laser diode.
During the gate controlled delay time tgd the blocking voltage across the thyristor drops to 90% (see Figure 21).
As initially only a small area around the gate structure becomes conductive, the initial current density and thus the
critical rate of rise of on-state current (di/dt)cr is a gauge for the robustness of the thyristor during turn-on.
Technischer Erläuterungen - Bilder
Hauptstromkreis
iT vT
main circuit
diT /dt
100%
90%
50%
10%
vCC
C
R
i T, vT
A
K
G iG
Steuergenerator
gate trigger
generator
vT
10%
90%
L
ITM
50%
iG
iT
t
tgd
diG /dt
a
b
IGM
Steuerstromkreis
gate circuit
t
Figure 21 Schematic representation of a thyristor turn-on process
Abb./Fig.19
Schematische
des off
Einschaltvorgangs
a - gate
current Darstellung
with turned
load circuit von Thyristoren
b - gate current with steeply rising on-state current (see also 3.3.1.8)
25
tgd [µs]
1000
100
10
a
b
1
0,1
10
100
iGM=iGT
iGM=4-5* iGT
1000
i GM [mA] 10000
Figure 22 Typical dependence of the gate controlled delay time tgd and the maximum gate current iGM
a) maximum value
b) typical value
3.4.1.2.1 Gate controlled delay time tgd
tgd is the period between the gate current reaching 10% of its maximum value IGM and the
time when the anode-cathode voltage drops below 90% of the applied forward off-state
voltage vD (see Figure 21).
It reduces significantly with increasing gate current (light power for LTTs) (see Figure 22).
In high power thyristors the tgd depends also on vD.
The value given in the data sheet is defined according to DIN IEC 60747 – 6 and is valid
for Tvj = 25°C and specified trigger pulse.
3.4.1.2.2 Critical rate of rise of the on-state current (di/dt)cr
Once the voltage has collapsed due to the thyristor triggering a small area of the
cathode around the gate structure begins to conduct on-state current. This current
conducting area then spreads out depending on the current density with a speed of
typically 0.1mm/µs. The current carrying capability of the system is therefore limited
in the beginning. Damage or destruction of the thyristor is impossible, however, when
the value given in the data sheet for the critical current slew rate is not exceeded. For
S-thyristors and thyristors with large square sections the gate is distributed (finger
structure). Therefore, these types show a higher (di/dt)cr.
According to DIN IEC 60747 – 6 the critical current rise time (di/dt)cr refers to loading
with on-state current over the period of a dampened half sine-wave. It is defined as the
angle of a straight line through the 10% and 50% points of the rising on-state current
(see Figure 21, Figure 47) whilst the following conditions apply:
Junction temperature Tvj = Tvj max
Forward off-state voltage vD = 0.67 VDRM,
Peak current value iTM = 2 ITAVM
Frequency of repetition f0 = 50 Hz
26
The trigger pulse is defined in the individual data sheets (see also 3.3.1.8).
Exception: Light triggered thyristors are tested with a forward off-state voltage of vD = VDRM.
3.4.1.2.3 Repetitive turn-on current IT(RC)M
IT(RC)M is the maximum permissible peak value of the on-state current immediately after turn-on with undefined
rate of rise. In general this turn-on current is caused by the discharge of the RC-snubber network. The maximum
permissible repetitive turn-on current also applies to the following steep current rise up to the critical rate of rise
of the on-state current (di/dt)cr.
For Infineon components the following values apply
IT(RC)M = 100A
Exception: Component with the type designation T…1N or T…3N
IT(RC)M = 150A
For applications above 60Hz the values for both the critical current rise time (di/dt)cr as well as the repetitive turn-on
current IT(RC)M have to be reduced. Further details for particular conditions on request.
3.4.1.2.4 Critical rate of rise of off-state voltage (dv/dt)cr
(dv/dt)cr is the maximum value for the rate of rise of a voltage applied in forward direction running almost linearly
from 0% to 67% of VDRM at which a thyristor will not switch to the on-state.
For an exponential rate of voltage rise it is a line which crosses the exponential function starting from 0% to 63% of
the maximum value.
It applies for open trigger circuit and maximum permissible junction temperature. Exceeding (dv/dt)cr may cause
destruction.
Exception: Aside from the over-voltage protection (BOD) light triggered thyristors have an integrated dv/dt protection. This causes the thyristors to trigger safely over the entire gate structure when the dv/dt gets to high.
3.4.2 Turn-off
Turning off is usually started by application of a reverse voltage. The load current of the thyristor or the diode does
not cease at the zero crossing but continues to flow briefly in reverse direction as reverse recovery current until the
carriers have left the junction region.
The softness factor FRRS describes the relation of the rates of rise of the currents during the turn-off process.
3.4.2.1 Recovery charge Qr
Qr is the total amount of charge flowing out of the semiconductor after switching from on-state to reverse off-state.
It increases with rising junction temperature as well as magnitude and fall time of the on-state current. If not
otherwise specified, the given values are valid for vR = 0.5VRRM and vRM = 0.8VRRM and are not exceeded by 95% of
the individual types of thyristors or diodes. For this an appropriately designed RC-snubber network is specified.
For components with the type designation T…1N, T…3N and D…1N the given values in the data sheet are maximum
values which are 100% tested in production.
The recovery charge Qr is mainly dependent on the junction temperature Tvj and on the rate of fall of the decaying
current (see Figure 24 and Figure 25).
27
Technischer Erläuterungen - Bilder
i,v
tp
FRRS
-di
di r
dt
dt
tint
ITM , IFM
trr
vT , v F
-di/dt
Qr
t
0,25 IRM
dir/dt
IRM
vR
vRM
0,9 IRM
Technischer
Erläuterungen
- Bilder
Figure
23 Schematic
representation
of the thyristor and diode turn-off process
1,1
Abb./Fig.20 Schematische Darstellung des Ausschaltvorgangs von Thyristoren und Dioden
Qr (Tvj) / Qr (Tvj max)
1,0
0,9
0,8
0,7
0,6
-80
-60
-40
-20
Tvj = Tvj -Tvj max [°C]
0
20
Figure 24 Typical Tvj-dependence of the recovery charge Qr normalized to Qr(Tvj max)
Q r (di/dt) / Q r (di/dt=10A/µs)
Abb./Fig.21
1,2Typische Tvj-Abhängigkeit der auf Qr(Tvj max) normierten Sperrverzögerungsladung Qr
1
0,8
0,6
0,4
0,2
0
0
1
2
3
4
5
6
7
8
9
10
di/dt [A/µs]
Figure 25 Typical di/dt-dependence of the recovery charge Qr normalized to Qr(di/dt=10A/µs)
28
11
3.4.2.2 Peak reverse recovery current IRM
IRM is the maximum value of the reverse recovery current. The dependences and operating conditions given for Qr
also apply. If IRM is not shown in the diagrams, its value may be approximately determined as follows:
IRMtIrrRM≈
-di/dt
2
⋅-di/dt
Q r ⋅ Q⋅ Q
r r
1...1,3
I 1...1,3
RM
For components with the type designation T…1N, T…3N and D…1N the given values in the data sheet are maximum
values which are 100% tested in production.
The peak reverse recovery current IRM is mainly dependent on the junction temperature Tvj and on the rate of fall of
the decaying current (see Figure 26 and Figure 27).
Technischer Erläuterungen - Bilder
1,1
IRM (Tvj) / IRM (Tvj max)
1,0
0,9
0,8
0,7
0,6
-80
-70
-60
-50
-40
-30
-20
Tvj = Tvj - Tvj max [°C]
-10
0
10
20
Figure 26 Typical Tvj-dependence of the peak reverse recovery current IRM normalized to IRM (Tvj max)
IRM (di/dt) / IRM (di/dt=10A/µs)
Abb./Fig.22
Typische Tvj-Abhängigkeit der auf IRM (Tvj max) normierten Rückstromspitze IRM
1,2
1
0,8
0,6
0,4
0,2
0
0
1
2
3
4
5
6
7
8
9
10
11
di/dt [A/µs]
Figure 27 Typical di/dt-dependence of the peak reverse recovery current IRM normalized to IRM (di/dt=10/µs)
29
3.4.2.3 Reverse recovery time trr
trr is the time interval between the zero crossing of the current and the time at which a
straight line through the 90% and 25% points of the decaying reverse recovery current
crosses the zero line (see Figure 23 ). Should trr not be specified, its value may be
approximately calculated with the following formula:
trr ≈
2⋅Qr
IRM
3.4.2.4 Turn-off time tq
tq is the time interval between the zero crossing of the current commutated in reverse
direction and the reapplication of forward off-state voltage at which a thyristor does not
turn-on without a control pulse.
The actual pause time realised in the application before the forward off-state voltage
reoccurs is called hold-off time. This time must always be greater than the turn-off time.
The turn-off time mainly depends on the fall time of the on-state current, the rate of rise
of the forward off-state voltage and the junction temperature (see Figure 29 - Figure 31).
To determine tq the duration tP of the forward current has to be chosen long enough so
that the thyristor at the point of commutation is completely turned on (see Figure 28).
The values given in the data sheets are valid for following conditions:
Junction temperature Magnitude of on-state current
Fall rate of the on-state current Reverse voltage Rate of rise of the forward off-state voltage Forward off-state voltage Tvj = Tvj max
iTM ≥ ITAVM
-diT/dt = 10 A/µs
VRM = 100V
dvD/dt = 20V/µs
VDM = 0.67VDRM
Exception: Fast thyristors were commutated off with a current rate of fall of –di/dt=20A/
µs. The dvD/dt may vary here and is specified by the 5th letter in the type designation
(see section 2.3).
For phase controlled thyristors usually typical values for the turn-off time are given as
they are mainly employed in line commutated converters. In these applications the
hold-off time is generally much longer than the turn-off time of the thyristor.
If the hold-off time is shorter than the turn-off time, the thyristor will once again
turn-on with rising forward off-state voltage without application of a trigger pulse and
destruction may be caused (tq-limit values on request if necessary).
30
If the thyristor is operated together with an inverse diode (for example free wheeling diode), much longer turn-off
times have to be taken into consideration due to the low commutation voltage (typically 30% longer). Additionally,
in such applications the inductance of the free wheeling circuit should be minimised as otherwise the turn-off time
may increase to significantly higher values.
Technischer Erläuterungen - Bilder
iT
50%
ITM
-diT/dt
tP
t
iR
dvD/dt
vD
63%
tq
VDM
vT
VRM
vR
t
vR
Abb./Fig.23 Schematische Darstellung vom Ausschalten und Freiwerden eines Thyristors
Figure 28 Schematic representation of the turn-off behaviour of a thyristor
Technischer Erläuterungen - Bilder
1,2
1,1
tq (Tvj) / tq (Tvj max)
1
0,9
0,8
0,7
0,6
0,5
0,4
-80
-70
-60
-50
-40
-30
-20
Tvj = Tvj - Tvj max [°C]
-10
0
10
20
Figure 29 Typical dependence of the turn-off time tq normalized to Tvj max on the junction temperature Tvj
Abb./Fig.24 Typische Tvj-Abhängigkeit der auf tq(Tvj max) normierten Freiwerdezeit tq
31
Technischer Erläuterungen - Bilder
tq(-diT/dt)
/ tq(-di
T/dt=10A/
tq(-di
/ t q(-di T/dts)
T/dt)
norm)
1,3
1,2
-diT/dtnorm:
N-Thyristor: 10A/µs
S-Thyristor: 20A/µs
1,1
1,2
1,0
1,1
0,9
1,0
0,8
0,9
0,7
0
1
2
0
3
5
4
5
6
-diT/dt / -diT/dtnorm
10
-diT/dt [A/ s]
7
8
15
9
10
20
Abb./Fig.25
/dt-Abhängigkeit derofaufthe
tq(diturn-off
tq
T/dtnorm) normierten
Figure
30Typische
TypicaldiTdependence
time tFreiwerdezeit
to the -diT/dtnorm on the off-commutating rate of
q normalized
Technischer
- Bilder
fall -diErläuterungen
T/dt
1,8
tq (dvD/dt) / tq (dvD/dt=20V/µs)
1,7
1,6
1,5
1,4
1,3
1,2
1,1
1
0,9
0
100
200
dvD/dt [V/µs]
300
400
500
Figure 31 Typical dependence of the turn-off time tq normalized to the dvD/dt = 20V/µs on the rate of rise of
off-state voltage dvD/dt
Abb./Fig.26 Typische duD/dt-Abhängigkeit der auf tq(duD/dt = 20V/µs) normierten Freiwerdezeit tq
32
3.5 Power dissipation (losses)
For thyristor and diode the dissipation (or losses) are classified as off-state, on-state, turn-on and turn-off losses.
The thyristor also shows control losses. Under given cooling conditions their sum determines the current loading
capability.
For mains operation up to 60Hz with its moderate dynamic requirements the dimensioning can be exclusively done
based on the on-state losses, as the sum of the others is comparatively negligible.
For semiconductors with high blocking voltages (> 2200V) or large square sections with a pellet Ø ≥ 80mm even for
mains operation the turn-off losses should be regarded in the calculation.
3.5.1 Total power dissipation Ptot
Ptot is the average value of the sum of the individual losses.
3.5.2 Off-state losses PD, PR
PD, PR are the losses caused by off-state current and off-state voltage in forward direction (PD) and in reverse
direction (PR).
3.5.3 On-state losses PT, PF
PT, PF is the electric power converted to heat when only the conducting state in forward direction is considered. The
average value of the on-state loss PTAV or PFAV is calculated with the values of the equivalent straight line according to
the following formula:
PTAV = VT(TO) • ITAV + rT • I²TRMS= VT(TO) • ITAV + rT • I²TAV • F² (for thyristors)
PFAV = VF(TO) • IFAV + rT • I²FRMS= VF(TO) • IFAV + rT • I²FAV • F² (for diodes)
For formfactors F refer to Table 1
The diagrams in the data sheets show the relation of the average value of on-state dissipation power and on-state
current for various shapes of current.
Instead of calculating the on-state losses with vT0, vF0 and rT, alternatively the on-state voltage can be calculated with
a more precise approximation with the following relation:
v T = A + B ⋅ i T + C ⋅ Ln( i T + 1) + D ⋅
iT
The factors A, B, C and D are listed in the datasheets.
Exception: PowerBLOCK-Modules are not listed with the ABCD coefficients.
33
Stromform
Current waveform
Scheitelfaktor
peak factor
0°
î
I AV
Formfaktor
form factor
IRMS
I AV
Formfaktor²
form factor²
î
IRMS
M=
sinus 180° el
2
π = 3,14
π / 2 = 1,57
2,47
sinus 120° el
2,23
4,18
1,875
3,52
sinus 90° el
2,83
6,29
2,22
4,93
sinus 60° el
3,88
10,9
2,77
7,66
sinus 30° el
5,88
23,42
3,98
15,8
DC
1
1
1
1
rect 180° el
2 = 1,41
2
2 = 1,41
2
rect 120° el
3 = 1,73
3
3 = 1,73
3
rect 90° el
4 =2
4
4 =2
4
rect 60° el
6 = 2,45
6
6 = 2,45
6
rect 30° el
12 = 3,46
S=
0°
Mittelfaktor
average factor
F2
180°
0
0
F=
180°
12
12 = 3,46
12
Table 1 Form factors for phase angle control conditions
3.5.4 Switching losses PTT, PFT+PRQ
PTT, PFT+PRQ are the portions of electric power converted to heat when turning on (PTT for
thyristors, PFT for diodes) and turning off (PRQ). The average switching losses increase
with increasing rates of rise and fall of the on-state current at turn-on and turn-off as
well as with the frequency of repetition. Up to medium size thyristors and diodes with
blocking voltages up to 2200V and applications at mains frequencies of up to 60Hz the
switching losses are mostly negligible compared to the on-state losses.
For semiconductors with high blocking voltages > 2200V or large square sections with a
pellet Ø ≥ 80mm even for mains operation the turn-off losses should be regarded in the
calculation (on request if necessary).
The turn-off losses of diodes, however, are generally still negligible.
3.5.4.1 Turn-on losses PTT, PFT
PTT, PFT is that dissipative portion which exceeds the on-state loss PT (for thyristors) or PF
(for diodes) during turn-on. It is caused on the one hand by the carrier storage effect and
on the other hand by the delayed propagation of the current carrying area.
To be able to turn on with the greatest possible square section many thyristors are
equipped with trigger amplification. This consist of one or several amplifying gates
(= auxiliary thyristors). In thyristors with large square sections the amplifying gate is
branched (finger structure). This causes a wider area to become conductive at the time of
triggering and thus reduces the turn-on losses.
34
The sum of turn-on and on-state losses PTT, PFT + PT, PF important for the dissipation calculation may be drawn from
the progression of the on-state current and the on-state voltage during and after turning on.
PTT +PT =
PFT +PF =
1
tT
tT
1
tT
tT
∫ i (t) ⋅ v (t)dt
T
T
(for thyristors)
0
∫ iF (t) ⋅ vF (t)dt
(for diodes)
0
In practice the turn-on losses are generally neglected.
3.5.4.2 Turn-off losses PRQ
Turn-off losses occur due to the carrier storage effect. They depend on the progression of the reverse delay current
as well as on the magnitude and rate of rise of the reverse off-state voltage and may therefore be influenced by the
snubber (see Figure 23 ).
PRQ =
1
tint
tint
∫ i (t) ⋅ v (t)dt
R
R
0
For the time period tint to be determined by integration the turn-off losses are calculated as follows:
An approximation of the turn-off losses may be calculated as follows:
PRQ = ERQ * f ≈ Qr * vR * 0.4 * f for the on-state limit characteristic
PRQ = ERQ * f ≈ Qr * vR * 0.5 * f for the typical on-state characteristic
ERQ = turn-off loss energy
f = frequency
Qr = maximum recovery charge
vR = (reverse voltage) driving voltage after commutation
3.5.5 Gate dissipation PG
PG is the electrical power converted into heat due the gate current flowing between gate terminal and cathode. This
is distinguished into peak gate dissipation PGM (product of the peak values of gate current and gate voltage) and
average gate dissipation PGAV (average value of gate dissipation referenced to the cycle duration).
3.6 Insulation test voltage VISOL
The insulation test voltage VISOL is the RMS-value of a sinewave voltage between the base plate and the terminal of
thyristor or diode modules. For DC-requirements VISOL DC is equal to the peak value of the specified RMS-value (i.e.
1.41* VISOL). During the test all terminals are connected with each other and VISOL is applied versus the base plate.
35
4. Thermal properties
In order to maintain the thermal equilibrium the electric power loss converted to heat
in the semiconductors has to be dissipated. For this purpose heatsinks with defined
cooling properties are available. To describe this function thermal equivalent circuits, by
Technischer
Erläuterungen - Bilder
analogy to electrical ciruits, according to Figure 32 are used.
a
b
Ptot
b
Ptot
Tvj
Tvj
RthjC
RthjC[A]
TC
QW
TC
RthCH
TH
QW
RthCA
RthCH[A]
RthCA[A]
RthHA
TH
RthHA[A]
TA
Tvj
RthjC
RthjC[K]
TC
TC
RthCH
RthCH[K]
TH
RthHA[K]
TA
RthCA[K]
TH
RthCA
RthHA
TA
Figure 32 Thermal equivalent circuits for diodes and thyristors
Rth JC= steady state thermal resistance junction - case
Rth CH= steady state transfer thermal resistance case - heatsink
Rth HA= steady state thermal resistance heatsink
Abb./Fig.27 Thermisches Ersatzschaltbild für Dioden und Thyristoren
a - single sided cooling
b - double sided cooling
4.1 Temperatures
4.1.1 Junction temperature Tvj, Tvj max
The junction temperature is the most important reference for all fundamental electrical
properties. It represents a mean spatial temperature within the semiconductor systems
and is, therefore, known more precisely as the equivalent junction temperature or virtual
junction temperature.
To observe the maximum permissible junction temperature Tvj max is important for the
function and reliability of the device. To exceed this maximum value may change the
properties of the semiconductor irreversibly and destroy it.
4.1.2 Case temperature TC
TC is the maximum temperature at the contact area of the thyristor or diode case of a disc
cell or the base plate of a PowerBLOCK-module.
36
4.1.3 Heatsink temperature TH
TH is the temperature of the heatsink resulting from the semiconductor through the contact area of the heatsink and
its surrounding cooling media.
The heatsinks offered by Infineon have been tested and specified with components mounted. The heatsink data
given, therefore, include the thermal transfer resistance RthCH between device and heatsink. This value can,
therefore, be disregarded in the calculation.
4.1.4 Cooling medium temperature TA
TA is the temperature of the cooling medium prior to entering the heatsink. For air cooling this is defined at the inlet
side of the heatsink. For fluid cooling it is defined at the inlet of the heatsink.
4.1.5 Junction operating temperature range Tcop
Tcop is the case temperature range in which the power semiconductor may be operated.
4.1.6 Storage temperature range Tstg
Tstg is the temperature range in which the power semiconductor may be stored without the application of electricity.
Independently of the maximum permissible junction temperature unlimited in time, the maximum permissible
storage temperature for epoxy disc cells and for PowerBLOCK-modules is Tstg = 150°C with a time limit to 672h
according to DIN IEC 60747-1.
4.2 Thermal resistances
4.2.1 Internal thermal resistance RthJC
RthJC is the ratio of the difference between the junction temperature Tvj and the case temperature TC to the total power
dissipation Ptot:
R thJC =
Tvj - TC
Ptot
It depends on the internal design as well as the shape and frequency of the on-state current.
The thermal resistance for double sided cooling compared to single sided cooling is lower due to paralleling of the
individual thermal resistances (see Figure 32).
The thermal resistance depends on the type and shape of the semiconductor. It is therefore not 100% measured,
but established instead during the initial type approval qualification tests.
4.2.2 Thermal transfer resistance RthCH
RthCH is the ratio of the difference between the temperature of the contact areas of the device and the heatsink
TC – TH to the total power dissipation Ptot:
R thCH =
TC - TH
Ptot
The values given are valid only when mounted correctly (see section 8).
37
4.2.3 Heatsink thermal resistance RthCA
RthCA is the ratio of the difference between the case temperature TC and the coolant
temperature TA to the total power dissipation Ptot:
R thCA =
TC - TA
Ptot
4.2.4 Total thermal resistance RthJA
RthJA is the ratio of the difference between the equivalent junction temperature Tvj and the
coolant temperature TA to the total power dissipation Ptot:
R thJA =
Tvj - TA
Ptot
= R thJC +R thCA
4.2.5 Transient internal thermal resistance ZthJC
ZthJC describes the progression of the component’s thermal resistance over time. In the
data sheets ZthJC is given for constant DC-current and partly also for pulse currents.
Additionally, the partial thermal resistances Rthn and time constants tn are compiled in a
table as an analytical function.
nmax
Z(th)JC = ∑ R thn(1 -e
-t tn
)
n=1
4.2.6 Transient heatsink thermal resistance ZthCA
ZthCA describes the progression of the heatsink thermal resistance over time. ZthCA is
defined in individual data sheets. Additionally, the values RthCAn and tn of the analytical
function are compiled in a table. There is no generally defined transient thermal
resistance for heatsinks. On the one hand, it depends on the contact region between
power semiconductor and heatsink. On the other hand, the cooling method (natural/
forced) and the flow of the cooling medium have a strong influence.
In case of natural cooling and oil cooling, the flow of the cooling medium is caused by
the convection of the air or oil. As the power dissipation defines the convection, the
actual power dissipation is specified for natural cooling and oil cooling. The correct
direction and position of the heatsink has to be observed.
In case of forced cooling and water cooling, the flow of the cooling medium is specified.
Short-term temperature variations due to pulse currents are widely independent of these
parameters. They are equalised through the large thermal capacity of the heatsink.
38
The heatsinks offered by Infineon have been tested and specified with components mounted. These given heatsink
data include the transfer thermal resistance RthCH between device and heatsink. This value is, therefore, to be
disregarded.
4.2.7 Total transient thermal resistance ZthJA
ZthJA describes the progression of the total thermal resistance over time. The calculation of the junction temperature
for short-term loads is to be based on the total transient thermal resistance. ZthJA is the sum of:
ZthJA = ZthJC + ZthCA
4.3 Cooling
4.3.1 Natural air cooling
In natural air cooling (air convection cooling) the power losses are dissipated due to natural convection of the air.
Generally the current loading capability of power semiconductors is defined at an ambient temperature TA = 45°C.
4.3.2 Forced air cooling
In forced air cooling the cooling air is forced through the fins of the heatsink by means of a fan. Generally the current
carrying capability of power semiconductors is defined at an ambient temperature TA = 35°C.
4.3.3 Water cooling
In water cooling the power losses are dissipated by means of water. Generally, the current loading capability of
power semiconductors is defined at an inlet water temperature TA = 25°C.
4.3.4 Oil cooling
In oil cooling the power losses are dissipated by means of oil. Generally, the current loading capability of power
semiconductors is defined at an inlet oil temperature TA = 70°C.
39
5. 5. Mechanical properties
5.1 Tightening torque
When mounting PowerBLOCK modules and studs, Infineon recommends keeping the
tightening torques as given in the data sheet, as otherwise the correct function within
the specifications cannot be guaranteed (see also 8.2).
5.2 Clamping force
The clamping force given in the data sheet is necessary for perfect electrical and thermal
contact of devices with flat base or disc housing. It must be largely homogeneous across
the contact surfaces (see also 8).
The limits of the clamping force for devices in disc housings are given in the relevant
data sheets. These have to be precisely observed. Deviations may alter the data and
require special agreement. The clamping force recommended should approximately be
in the middle between the given limits.
5.3 Creepage distance
The creepage distance between anode and cathode or anode and gate is defined
according to DIN VDE 0110.
5.4 Humidity classification
The values given comply with DIN IEC 60721-3 (3K3).
5.5 Vibration
The values given follow DIN IEC 60068, part 2-6.
It is given in the data sheet as a multiple of the gravitational constant (1g = 9.81m/s²).
5.6 UL-registration
PowerBLOCK modules normally comply with the standard for electrically insulated
semiconductor components of the Underwriters Laboratories Inc.
The appropriate file number is listed in the individual data sheets in the section
Mechanical Properties.
40
6. Notes for applications
6.1 Case non-rupture current
The case non-rupture current is the peak value of a current pulse in reverse direction which causes neither a
mechanical destruction of the case nor the escape of combustive plasma.
The non-repetitive surge currents ITSM, IFSM and ∫i²dt values given in the data sheets define the limit of electrical
stress in forward direction. They are used to design the short circuit protection. By definition thyristors and diodes
will not be destructed by this stress. In any case thyristors have to be triggered by sufficient gate current.
If the short circuit current in forward direction is higher than the given maximum values, at first electrical destruction occurs. The mechanical destruction of the device housing occurs only at substantially higher stress as the total
active region of the semiconductor partakes in carrying the current.
If a thyristor or a diode becomes defective in reverse direction, a short circuit current flows in reverse direction. The
cathode region not destroyed at that stage does not partake in the current flow. A small edge around the destroyed
spot melts and an arc develops inside the case. The melted material vaporizes to hot plasma which depending on
its intensity may lead to the destruction of the case. Often a hole in the case results through which hot plasma
escapes. In high power installations with strong magnetic fields it may lead to the short circuit and destruction of
the equipment.
Destructive tests carried out on thyristors and diodes in reverse direction show great variance in the distribution of
the case non-rupture current depending on the location of the destroyed spot on the silicon pellet. Infineon always
places the destruction spot at the edge as thereby the most critical case non-rupture currents occur. The rate of rise
of the short circuit current which depends on the inductances of the short circuited section of the installation is also
of influence. Infineon specifies the case non-rupture current for a 50Hz half-sinewave.
For diodes and thyristors the case non-rupture current may be lower than the non-repetitive surge on-state current
ITSM or IFSM. In these instances the case non-rupture current is given as the peak value of a half-sinewave of 50Hz
additionally in the data sheets for disc cells. The I²t-value resulting from this can be recalculated to the peak value
of a half-sinewave of 60Hz.
Recalculations of this case non-rupture current to other current wave forms, as for example occur when a short
circuit is turned off due to a fuse failure, are not or only partly correct even when they are based on an appropriate
current-time-integral.
To avoid damage the user has to provide appropriate protection measures in particular in high power installations.
6.2 Thermal load cycling
Thermal load cycling in semiconductor systems results in mechanical stresses or sliding action due to the
different coefficients of expansion of the materials. The load cycle capability of components, therefore, depends on
the magnitude as well as the progression of the temperature shifts in the device and on the number of cycles. Rapid
temperature changes of low magnitude as they often occur in permanent operation with a frequency of repetition f0
≥ 40Hz bear no influence on the load cycle capability. Only in operation with heavy load changes or low frequency
of repetition, the magnitude of the rapid temperature changes in the device ΔTvj are to be observed with regard to
sufficient lifetime for thermal load cycling.
41
6.3 Parallel connection
When connecting thyristors or diodes in parallel, equal distribution of the load current in
the branches should be aimed for. Reasons for deviations from current sharing are:
n Different slope resistances in parallel branches. These are caused by the variance in
distribution of the on-state characteristics of the devices and through the construction
in the paralleled circuits (see Figure 34).
Dynamic influences, such as:
n variance of the gate controlled delay time
n differences in the dynamic turn-on behaviour
n additionally induced voltages caused by the mechanical construction
In addition, it should be taken into consideration that all RC-snubbers of the paralleled
branches will discharge across the thyristor which triggers first.
Equal current sharing in the paralleled branches can be achieved by the following
measures:
n Application of thyristors or diodes with approximately the same on-state voltages.
On requestErläuterungen
the supply of- such
Technischer
Bildercomponents in groups with the same vT- or vF-class is
possible.
n Identification of the vT- or vF-class respectively is provided on the ceramic disc cell,
by means of a “V” followed by a 4-digit number printed on it. “V” is an abbreviation
standing for the on-state voltage. The 4-digit number indicates the maximum on-state
voltage of the corresponding vT-/vF-class and the class width (see Figure 33)
V1435
143x10mV = 1,43V
5x10mV = 50mV
Max vT der Klasse
Max vT of class
Klassenbreite
class width
1,38V < vT <= 1,43V
Figure 33 Example of vT/vF class definition
n Equal slope resistances as far as possible. Additional series resistances in the
individual branches of the paralleled thyristors or diodes e.g. fuses will improve the
symmetry.
n Application of series inductances to equalise current sharing of the thyristors.
42
n Minimal deviation in gate controlled delay time values. To minimise this, triggering of the thyristors with
synchronous, steep and high current pulses is required.
iGM ≥ 4...10 IGT
dG/dt ≥ iGM/(0.5-1µs)
The anode-cathode voltage across the paralleled devices drops to the on-state voltage of the first thyristor which
triggers. Consequently, the voltage dependent trigger delay of the thyristors turning on later and the start of
turn-on of these thyristors is retarded accordingly.
This has to be considered in particular for light triggered thyristors as these require a higher anode-cathode
voltage to safely turn-on.
For high power thyristors (T…1N) the data sheet recommends a trigger pulse with gate controlled delay time. With
this, the deviation of the gate controlled delay times tgd may be reduced to values Δtgd < 0.5µs under the listed
conditions. In conjunction with the snubber this is generally sufficient for safe triggering of the thyristors which
makes additional selection needless.
To parallel light triggered thyristors (T…3N) Infineon recommends the use of laser diodes SPL PL90 with the
appropriate fibre optic cable and a control pulse for the laser diode of 1.3A for 2µs followed by 0.8A for 8µs (see
Figure 18).
n The gate pulses described above also assure that the differences in the dynamic on-state characteristics are
minimised.
n In particular for large thyristors and those with high blocking voltages the risk exists that some of these will return
to the forward off-state after triggering due to a too low on-state current density. Overloading of the current
carrying thyristors after renewed load current increase can be avoided by repetitive triggering.
Erläuterungen
In Technischer
general, a current
sharing- Bilder
imbalance of less than 10% is aimed for.
IF, I T
I1
I2
Vparallel
vF, vT
Figure 34 Current sharing imbalance due to different on-state voltages in parallel connection
Abb./Fig.28 Stromfehlverteilung infolge unterschiedlicher Durchlassspannungen bei Parallelschaltung
43
6.4 Series connection
When connecting thyristors or diodes in series, equal distribution of the off-state voltage
should be aimed for. Reasons for deviations from the ideal voltage sharing are:
n Different leakage currents
Without additional external components, an unfavourable voltage sharing may
occur during the steady off-state condition in both directions as the voltage across the
individual thyristors or diodes results out of the uniform reverse current in the series
circuit (see Figure 35).
n Variance of the gate controlled delay time
During turn-on the thyristors triggered last are exposed by higher off-state voltage.
n Variance of the reverse recovery charge
Differences of the reverse recovery charge Qr cause different reverse recovery times
trr and peak reverse recovery currents IRM which means that the thyristors or diodes
take up off-state voltage at different times (see Figure 36 ). The variance of the reverse
recovery charge ΔQr of two thyristors or diodes connected in series effects a voltage
deviation ΔV ≈ ΔQr/C where C is the capacitor of the parallel snubber circuits (see
section 7.1).
iD , iR
Technischer Erläuterungen - Bilder
Iseries
V1
vD, vR
Figure 35 Voltage sharing imbalance due to different leakage currents in series connection
Abb./Fig.29 Spannungsfehlverteilung infolge unterschiedlicher Sperrströme bei Reihenschaltung
44
V2
i,v
iT , i F
vT , v F
t
iR
Qr
vR
V= Qr/C
Figure 36 Voltage sharing imbalance due to different turn-off properties
hlaufteilung infolge unterschiedlichen Ausschaltverhaltens
Equal off-state voltage for thyristors and diodes connected in series may be achieved by the following meaures:
n Steady state voltage sharing during the off-state phase
For this the RC-snubber is often sufficient. In case the DC off-state voltage is applied for longer periods, an
additional voltage sharing resistor paralleled to each thyristor or diodes is necessary. It should carry about two to
five times the leakage current of the applied power semiconductor at operating temperature in order to externally
force a steady state voltage symmetry. If the operating temperature is less then the maximum allowable junction
temperature for continuous operation, the leakage current drops per 10°C to approx. 66% of the initial value.
For example for thyristors with a maximum allowable junction temperature the following applies
Tvj max = 125°C:
0.66 ID oR 0.66 IR at Tvj = 115°C
0.44 ID oR 0.44 IR at Tvj = 105°C etc.
n Dynamic voltage sharing at turn-on
To reduce the variance of the gate controlled delay times, triggering of electrically triggered thyristors is necessary
with synchronous, steep and high trigger pulses.
iGM ≥ 4...10 IGT
diG/dt ≥ iGM/(0.5-1µs)
Such strong trigger pulses reduces the spread of the gate controlled delay time to values Δtgd < 1µs. It has to
be ensured that the reverse blocking voltage of the thyristor which is last to turn on (in a series connection)
45
increases only slowly. Often the RC-snubber is sufficient for this. In case the
inductance of this circuit working jointly with the RC networks are not sufficient to
reduce the reverse voltage increase additional saturable inductances are to be
implemented.
For high power thyristors (T…1N) a trigger pulse for a gate controlled delay time is
recommended in the data sheet. With this or better pulses the variance of the gate
control delay times may be reduced to values Δtgd < 0.5µs under the given conditions.
For series connection of light triggered thyristors (T…3N) which are exposed to high
current rise times Infineon recommends the use of laser diodes SPL PL90 with the
appropriate fibre optic cable and a control pulse of 1.3A for 2µs followed by 0.8A for
8µs.
n Dynamic voltage sharing at turn-off
During turn-off it is possible to improve the imbalance of off-state voltage sharing
both by sufficient dimensioning of the paralleled snubbers as well as by a small
variance of the recovery charge ΔQr of the thyristors in series. The supply of thyristors
and diodes in groups with the same Qr-class is possible on request.
6.5 Pulsed Power
Pulsed power applications are generally applications with very low duty cycle.
To dimension semiconductors for pulsed power applications generally the following has
to be observed:
6.5.1 Applications with DC
Often the power semiconductors in pulsed power applications are exposed to high
DC-voltages. For this the limitations regarding reduced voltage stress are to be observed
(see 3.1.2.3 and 3.2.2.3).
6.5.2 Current rise time at turn-on
Due to the finite propagation in the triggered area (~ 0.1mm/µs) when the thyristor is
turned on, the load current is initially concentrated to a small area. If the current density
exceeds the critical value, destruction of the device is likely. Therefore the peak current
amplitude in short pulse durations drops significantly (see Figure 37).
46
Technischer Erläuterungen - Bilder
ITSM(tP) / ITSM(10ms)
10
ITSM-Limit
di/dt-Limit
Safe
Operation
Area
1
0,1
0,01
tp [ms]
0,1
1
10
Figure 37 Schematic representation of the Safe Operation Area (SOA) of a thyristor optimised for pulsed power
with single sine wave current pulses
Abb./Fig.31 Prinzipielle Darstellung der Safe Operating Area (SOA) eines Thyristors für einzelne Strompulse.
6.5.3 Zero crossing of current and voltage during turn-on
With positive voltage applied to a thyristor may be turned on by a trigger pulse. After the gate controlled delay time
of up to several µs the voltage collapses sharply, the load current rises and the propagation in the conduction zone
starts. If during this process a reversal of voltage and current to negative values occurs, a constriction of the
conductive zone results. The energy is concentrated to a small section and can destroy the semiconductor,
regardless of whether it is light triggered or electrically triggered.
Such operating conditions are to be avoided by the use of appropriate free-wheeling circuits.
6.5.4 Turn-off with a high di/dt versus a negative voltage
These operating conditions are to be avoided if possible as they require very extensive snubbering if controllable at
all. The voltage peaks result from the snap-off of the reverse recovery current and the inductances in the circuit and
have to be limited to values permissible for the semiconductor.
Figure 38 and Figure 39 depict circuits suitable for pulsed power applications and the stresses to which the
Technischer
Erläuterungen - Bilder
semiconductors
are exposed. With the circuit shown in Figure 39 the thyristors are stressed more (see Figure 40 –
Var.2). By using circuit shown in Figure 38 the turn-off versus voltage is avoided, however (see Figure 40–Var.1).
Thyristorschalter
thyristor switch
C
Freilaufdioden
free wheeling diodes
L
R
Kreisinduktivität
circuit inductance
Kreiswiderstand
circuit resistance
Last
load
Figure 38 Thyristor switch with free-wheeling circuit at the capacitor side
Abb./Fig.32 Thyristorschalter mit Freilaufkreis am Kondensator
47
Technischer Erläuterungen - Bilder
Thyristorschalter
thyristor switch
C
Freilaufdioden
free wheeling diodes
L
R
Kreisinduktivität
circuit inductance
Kreiswiderstand
circuit resistance
Last
load
Technischer Erläuterungen - Bilder
Figure 39 Thyristor switch with free-wheeling circuit at the load side
Abb./Fig.33 Thyristorschalter mit Freilaufkreis an der Last
iT, vT
iT, vT
Var 1: Freilaufdiode am Kondensator
iT Var 2
iT Var
2
iT Var 1iT Var 1
vTvVar
2 2
T Var
vT Var v
1 T Var 1
t
VRM >> 100V
VRM >> 100V
Figure 40 Current and voltage waveforms at the thyristor
Abb./Fig.34 Stromverlauf durch den Thyristor in der Schaltung gemäß xx xx
48
Var 1: Freilaufdiode
am wheeling
Kondensator diode at capacitor
free
free wheeling diode at capacitor
Var 2: Freilaufdiode
an Last
Var 2: Freilaufdiode
an Last
free
wheeling
free wheeling
diode
at load diode at load
t
7. Protection
Thyristors and diodes have to be reliably protected versus too high currents and voltages as well as interference
pulses in the control circuit.
7.1 Overvoltage protection
On the whole, overvoltages in an installation have the following causes:
n Internal overvoltages Due to the carrier storage effect of the power semiconductors
n External overvoltages Due to switching processes on the line and
atmospherical influences such as
- switching of transformers without load
- switching of inductive loads
- blowing of fuses
- lightening strikes
As thyristors and diodes may be destructed by overvoltages in the micro second region, their overvoltage protection
requires particular attention. When designing appropriate snubbering the blocking capability (VDRM, VRRM) as well as
the critical rate of voltage rise (dv/dt)cr has to be considered.
7.1.1 Individual snubbering (RC-snubber)
During turn-off the load current of the thyristor or the diode does not stop to flow at the zero crossing but continues
briefly in reverse direction as reverse recovery current due to the carrier storage effect (Figure 23 ). Once the peak
reverse recovery current is reached, the more or less steeply falling reverse delay current causes a voltage peak
at the inductances of the load circuit which is superimposed onto the driving voltage and may thus put the
semiconductor at risk.
This overvoltage may be effectively reduced by the individual snubbering of the semiconductor with an RC-snubber.
To dimension this snubber it is necessary to know the most important factors of influence such as the magnitude
of iTM or iFM and rate of fall of -diT/dt or –diF/dt of the on-state current, peak reverse recovery current IRM, reverse
off-state voltage vRM, repetitive reverse peak off-state voltage VRRM of the semiconductor as well as the critical rate of
voltage rise (dv/dt)cr for thyristors. In mains commutated converters RC-snubbers for thyristors and diodes can be
used under normal operating conditions according to Table 2 under the following conditions:
n Short-circuit voltage of the converter supply transformer uK ≥ 4%. When connected directly to the mains the
protection choke has to be dimensioned accordingly.
n Safety margin between the repetitive peak off-state voltage and the peak value of the supply voltage ≥ 2.2.
49
Durchlassstrom ITAV, IFAV
nominal voltage VN
Anschlusspannung VN
≤230V
≤400V
≤500V
≤690V
on-state current ITAV, IFAV
≤ 50 A ≤ 100 A ≤ 200 A ≤ 500 A ≤ 1000 A ≤ 2000 A
C [µF] 0,22
0,33
0,68
1,5
3,3
6,8
R [Ω]
47
33
22
12
6,8
6,8
P[W]
≥5
≥ 10
≥ 15
≥ 30
≥ 70
≥ 150
C [µF] 0,12
0,22
0,47
1,0
2,2
4,7
R [Ω]
82
56
33
22
15
12
P[W]
≥7
≥ 15
≥ 30
≥ 70
≥ 125
≥ 300
C [µF] 0,10
0,18
0,39
0,82
1,8
3,3
R [Ω]
120
68
39
27
18
15
P[W]
≥ 10
≥ 25
≥ 50
≥ 100
≥ 200
≥ 400
C [µF]
0,27
0,56
1,0
1,8
R [Ω]
47
33
22
22
P[W]
≥ 70
≥ 125
≥ 250
≥ 500
Table 2 RC-snubbers for individual snubbering in mains applications
Especially in cases of steep rate of fall of the on-state current or low safety margin of
the blocking capability the RC-snubbers recommended above should be checked for
suitability. In these cases often capacitors with greater capacity as well as appropriately
re-dimensioned resistors are required. The best equivalent resistance for the most
favourable non-periodic dampened overvoltage progression is calculated as follows:
C’ =
Qr
R' = (1,5...2) ⋅
VR * 2
L'
C'
Where R’ and C’ are equivalent values of the RC-series snubber and L‘ is the equivalent
value of the converter inductance.
Schaltung
R'
C'
L'
M1
R
C
LS+LG
M2
R
C
2 LS
B2
1
/2 R
C
LS
M6
1
/2 R
2C
2 LS
B6
3
/5 R
5
/3 C
2 LS
Table 3 Equivalent values for converter circuits
R, C = values of the RC-snubber
LS = stray inductance of the converter transformer (one phase)
LG = inductance of the smoothing choke
50
For thyristors it also has to be observed that the resistor of the RC-snubber has to have the value of
R'
VDWM
IT(RC)M
in order for the thyristor not to be stressed with too high a discharge current from the snubber during turn-on (see
also 3.4.1.2.3).
The dissipation power of the resistor is calculated according to the following formula
PR=k*VR2*C*f
k = 2*10-6 k = 4*10-6 k = 6*10-6 for uncontrolled rectifiers
for controlled single and two pulse circuits and in AC-controllers
for controlled three and six pulse circuits and in three-phase controllers
It should be ensured here that the values with the following units are used in the formula
PR [W]
VR [V]
C [µF]
f [Hz]
If required, the snubbers according to Figure 41 may be modified so that the reduction of the over-voltage and thus
Technischer Erläuterungen - Bilder
less stress for the thyristor during turn-on is achieved.
a
b
c
Do
R
R
C
C
Ro
Do
Re
Co
R
C
Ro
Re
Co
Figure 41 Examples for extending RC-snubbers for thyristors
Abb./Fig.35 Erweiterte TSE-Beschaltung
a – with bipolar voltage surge suppressor
b – with RCD combination to dampen the turn-on current
c – with RCD combination to dampen the dv/dt and forward off-state voltage
Note: Do = fast diode particularly regarding turn-on
51
The RC-snubber may usually be omitted for rectifier operation when transformer
Technischer Erläuterungen
- Bilder
snubbering exists (see
7.1.3), provided thyristors with a critical rate of voltage rise
(dv/dt)cr ≥ 500 V/µs are used.
7.1.2 Input snubbering for AC-controllers
In AC- and three phase controllers thyristors are used in anti-parallel configuration both
for phase control as well as full wave operation for example in soft starters. Figure 42
RB
CB
L
N
Figure 42 Snubber circuit for AC-controllers
shows the snubber circuit.
The values for RC-series snubbers recommended in Table 2 apply for the snubbering of
thyristors under normal operating conditions as well as the following circumstances:
Abb./Fig.36 Eingangsschutzbeschaltung für Wechselrichter
n Inductive phase angle between supply voltage and current ≤ 30°el (cos φ ≥ 0.866).
This assures that possible oscillation caused by the series connection of snubber
capacitors and inductances is suppressed.
n Safety margin between the repetitive peak off-state voltage of the thyristors and the
peak value of the supply voltage ≥ 2.2 (see 3.1.2.1).
n Critical rate of voltage rise of the thyristors (dv/dt)cr ≥ 500V/µs.
Note: The on-state current ITAV given in Table 2 is to be seen with sufficient accuracy as
the average value of a thyristor in one-way configuration. To determine the load current
the RMS-value ITRMS of the individual thyristor in anti-parallel configuration and the RMSvalue IRMS of the total circuit may be derived from the following formulae:
For high power semiconductors and light triggered thyristors implemented in large
52
IRMS
ITRMS
IAV
ITAV
ITRMS =
ITAV ⋅ π
IRMS =
2
ITAV ⋅ π
2
Figure 43 Calculation of the current for an AC-controller
installations, it is common to optimise the snubbering according to the circuit
parameters and the semiconductor type used. In this, the rate of voltage rise can be disregarded as the critical rate
of voltage rise of these thyristors is plainly better than the criteria mentioned above.
Generic recommendations for snubber designs therefore do not make much sense.
7.1.3 Supply snubbers for line commutated converters
Energy intensive overvoltages from the mains or caused by the switching of converter transformers or chokes are
preferably dampened by combined snubber circuits. For converters with thyristors or diodes they are placed on the
AC-side and consist of auxiliary rectifiers with diodes and protection capacitors with discharge resistors. These discharge resistors are necessary because the diode bridge prevents the discharge of the snubber capacity. Therefore,
they have to be designed in a way that this capacitance is discharged within one period (see Figure 44 and Table 4).
Additional individual snubbering of all thyristors and diodes in the converter as well as the auxiliary rectifier is
Technischer Erläuterungen - Bilder
~
~
R1
+
–
~
C1
~
~
R1
R1
R1
R2
+
C1
R2
–
Abb./Fig.37
Summenschutzbeschaltung
auf controlled
der Wechselstromseite
Figure 44 Combined
snubber
on the AC-side of a the
rectifier eines Gleichrichters
53
stack output current IDC
supply voltage VN
Satzausgangsstrom IDC
circuit B6C
Anschlussspannung VN
Schaltung B6C
= 500V
= 200A
= 1000A
= 2500A
R1 [Ω]
6,8
3,9
1,8
1
C1 [µF]
6,8
10
22
33
R2 [Ω]
15
12
4,7
3,3
P2 [W]
32
40
104
150
Satzausgangsstrom IDC
Schaltung B6C
stack current IDC
supply voltage VN
Anschlussspannung VN
circuit B6C
= 690V
= 200A
= 750A
= 1500A
= 3000A
= 4000A
R1 [Ω]
22
8,2
3,9
2,7
1,8
C1 [µF]
2,2
4,7
10
15
22
R2 [Ω]
47
22
12
6,8
4,7
P2 [W]
20
43
78
140
201
Satzausgangsstrom IDC
Schaltung B6C
stack current IDC
supply voltage VN
circuit B6C
Anschlussspannung VN
= 5000A
= 1000V
= 500A
= 1000A
= 2000A
= 3000A
R1 [Ω]
18
8,2
5,6
3,9
C1 [µF]
2,2
4,7
6,8
10
R2 [Ω]
47
22
15
12
P2 [W]
42
90
133
166
Table 4 Components for a combined snubber on the AC-side of controlled three phase bridge
generally not necessary as the combined snubber acts also as an RC-network. Exempt from this are some double converter circuits such as two three-phase anti-parallel
bridges. For the design of the combined snubber the following has to be observed:
n Series resistor R1
is there to prevent possible oscillations when the converter transformer is switched.
At the same time it limits the discharge peak originating from the protection capacitor
through the diodes of the auxiliary rectifier during turn-on and overvoltage stressing.
n Protection capacitor C1
has to absorb the energy building up when the converter transformer or a choke is
switched off, so that the voltage will not exceed the maximum permissible repetitive
peak off-state voltage of the thyristors or diodes to be protected; not included are the
arcing losses of the switch.
54
n Discharge resistor R2
is sufficiently dimensioned - according to practical experience - when the discharge time constant t of successive
overvoltage energy equals R2˙ C1 = 80ms.
n Auxiliary rectifier diodes
To be considered for their selection apart from the required blocking capability is also the permissible surge
current in dependence of the charge surge current of the protection capacitor. As overvoltages only occur for short
periods and far apart, utilisation of the diodes is infrequent and, therefore, their power dissipation is low. As a
consequence heatsinks are generally not necessary.
7.1.4 Additional options for protection versus energy intensive overvoltages
RLC – filters
consist of the stray inductivity of the converter transformer or the inductivity of the commutating chokes and the
RC-networks grounded at the star point. They are suitable to dampen transient overvoltages of short duration and
low energy because, with regard to the discharge current of the capacitors, the resistors may not be chosen too low.
Apart from that, due to the occurring losses, the size of the capacitances is limited (see Figure 45).
Spark-gap arrestor
can be used when energy intensive overvoltages are expected from the line. Due to their delayed turn-on after their
trigger voltage is reached, usually additional protection measures versus overvoltages are necessary (see Figure 45).
DC-snubbering
Overvoltages originating from the load side may be dampened with DC-snubbers (see Figure 45).
Instead of RC-networks voltage dependent resistors such as metal oxide varistors may be used. On the one hand,
it is to be kept in mind that varistors are generally not suitable to limit repetitive overvoltages as they otherwise
become thermally instable and are subject to severe ageing. On the other hand, it should be noted that protection
for energy intensive overvoltages (usually spark-gap arrestors) must not be thwarted by incorrectly dimensioned
varistors.
55
Technischer Erläuterungen - Bilder
L1
L2
L3
Funkenstrecke
spark gap
L
MO-Varistor
RLC-Filter
DC-seitiger Filter
DC side filter
Figure 45 Additional options for protection versus energy intensive overvoltages
Abb./Fig.38 Zusätzliche Schutzmöglichkeiten gegen energiereiche Überspannungen
7.2 Overcurrent protection
Thyristors and diodes can be loaded with high operating currents but can be destroyed
by overcurrents and thus require suitable protection meaures. The selection of
appropriate protection depends on the type of overcurrent. In general, it is differentiated
between short-term and long-term protection.
7.2.1 Short-term protection with superfast semiconductor fuses
The short-term protection limits the overcurrent originating from a short circuit to a value
which does not put the thyristors or diodes at risk in at time span up to a half-sinewave
and is achieved by the use of special semiconductor fuses with superfast open-circuiting
characteristic. In worst case when turning off they make use of the ∫i²dt-value given in
the data sheet for the individual type.
The semiconductors loose their off-state and blocking capability entirely or partially
when stressed with the ∫i²dt-value until the junction temperature has dropped back to
the permissible value for permanent operation. This stressing may, therefore, only be
repeated after a few seconds and should only rarely occur with a limited number of
pulses over the entire period of operation of the converter (see also 3.1.16).
7.2.1.1 Selection of fuses
The fuses may optionally be placed in the phase or in the branch (arm). The branch fuse
enables the most secure short-term protection and permits maximum current loading of
the thyristors or diodes. A construction with phase fuses reduces the complexity.
56
However, for possible feedback from a load with back e.m.f. an additional fuse on the output of the converter has
to be implemented as a short circuit current from the load fed back into the DC-bus is not registered by the phase
fuses.
For some thyristors or diodes with high current loading capability paralleling of two fuses is necessary.
When selecting a fuse the following has to be taken into account:
n Fuse voltage rating
It has to be higher than the voltage which drives the short-circuit current.
n Voltage which drives the short-circuit current:
It is generally the same as the supply voltage; only for AC-converter operation will it be up to the 1.8-fold value of
the supply voltage.
n Reoccurring voltage VRMS
This results from the voltage VKRMS driving the short-circuit current divided by the number N of the series fuses
placed in the short circuit path multiplied with the safety factor Fs = 1.3. The following formula applies:
VRMS =
VKRMS
N
* Fs
for example in B2 and B6-circuits VRMS = ½ *1.3*VKRMS = 0.65*VKRMS
n Fuse arcing voltage
During the quenching process the fuse produces an arcing voltage which depends on the construction of the fuse
and the reoccurring voltage. These voltage peaks may not exceed the surge peak voltage of the semiconductors in
order not to harm any reversed biased components in the circuit.
n Nominal fuse current rating:
This usually refers to sine wave AC-current and will be above or below the rated value for deviating current wave
forms. The nominal current of the fuse should be somewhat higher than the phase or branch current to be
expected.
n ∫i²t turn-off value
This is the sum of melting and arc integral and has, therefore, to be lower than the ∫i²dt-value of the thyristor.
57
iT,F
Schmelzintegral
melting integral
Löschintegral
arc integral
t
tP ≈ 3-5 ms
Figure 46 Turn-off characteristic of superfast fuses
Schaltung
circuit
+
π
Ud
M1
Id
I2
U1
I1
Zweigstrom
arm current
RMS
2
Id ( AV ) = 1,57 Id ( AV )
4
Id ( AV ) = 0,79 Id ( AV )
4
Id( AV ) = 0,79 Id( AV )
Strangstrom
phase current
RMS
+
Ud
Id
I1
+
I2
π
Ud
U2
U1
B2
π
U2
M2
Abb./Fig.39 Abschaltcharakteristik superflinker Sicherungen
Id
I2
U1
I1
π
2 2
Id ( AV ) = 1,11 Id ( AV )
U2
U1
I1 sqrt(2)
I2
M6
1
+
3
Id( AV ) = 0,58 Id( AV )
Ud
Id
U2
U1
Id
I2
1
Ud
U v2
B6
+
3
Id( AV ) = 0,58 Id( AV )
-
ITRMS
ITAV
1
U2
U1
I1
W1C, W3C
2
Id ( AV ) = 0,71 IPhase ( R M S )
idealisierte Betrachtung für Widerstandslast und Vollaussteuerung
ideal view for resistive load and full conduction
Table 5 Calculation of branch (arm) and phase currents
58
2
3
Id ( AV ) = 0,82 Id( AV )
During the increase of the short-circuit current the fuse-link melts first. The arc resulting hereby is then quenched by
the covering filler – usually quartz sand. These fuses open within 3 to 5ms (see Figure 46)
The RMS-value of branch or phase current may be derived the output current of the various converter circuits using
the formulas in Table 5.
These factors apply for resistive load and zero-delay output
7.2.2 Further protection concepts: short-term protection of high power semiconductors
7.2.2.1 High speed DC-circuit breakers
electro-dynamic triggering within a few milliseconds in a short circuit situation. They are rarely used due to the high
costs.
7.2.2.2 Crowbar (electronic short circuit)
are mostly used in voltage source inverters with turn-off components (IGBT, GTO, IGCT). Once the DC-bus voltage
exceeds a defined protection level, the crowbar is triggered and discharges the DC-bus capacitors. When the pulse
current reverses polarity, it is fed via a special diode or via the free wheeling diodes in the inverter circuit.
7.2.2.3 Line side circuit breaker
The semiconductors have to carry the short-circuit current until the circuit breaker disconnects the mains. In large
installations this happens after three to five half-waves.
7.2.2.4 Blocking of trigger pulses
When exceeding a defined level the trigger pulses for the thyristors are suppressed. The thyristors are then stressed
with a current half-wave followed by negative and positive off-state voltage. This requires sufficient blocking
capability of the semiconductors.
7.2.3 Long-term protection
This can be achieved by suitable thermal and magnetic overcurrent protection schemes or fuses. The turn-off
characteristics of these protection units should be lower than the overcurrent in short-term operation. The blocking
capability of the thyristors or diodes will therefore remain. The long-term protection for thyristors may therefore also
be achieved by blocking of the trigger pulses. If the maximum blocking capability is not required, the interrupt
characteristics can be set on the maximum overload on-state current characteristics according to section 3.1.14.
7.2.4 Fully rated protection
This consists of long and short-term protection and in practice is only achieved by a combination of several
protection meauseres.
59
7.3 Dynamic current limiting with inductors
in the load circuit
If the inductance in the load circuit is low, too high rates of rise for the current may occur
when a thyristor
turns-on. To avoid destruction insertion of additional inductors LZ is necessary which
causes a reduction in the rate of rise of the turn-on current (see Figure 47). This measure
also reduces the turn-on losses.
In case of linear inductances the current density in the propagating triggered silicon area
is reduced during the current rise.
In saturation chokes the high rate of rise for the current will only occur after the step time
tSt (see Figure 47) when already a larger proportion of the silicon pellet is conductive. The
step current iTSt (see Figure 47) should, at the beginning of the step time, equate approximately the repetitive turn-on current IT(RC)M (see 3.4.1.2.3).
In case the step current is lower, it can be increased by a resistor Rp in parallel to the
choke. If at the time instant 0 a voltage V0 is applied, the current iRSt resulting is as
follows:
iRSt =
V0
Rp
Technischer Erläuterungen - Bilder
ITM
iT
L
R
a
Lz
diT/dtcrit
b
vC
a: ohne zusätzlicher Induktivität
without additional inductance
c
C
iT, vT
A
K
G
Steuergenerator
gate trigger
generator
b: mit linearer Induktivität Lz
with linear inductance Lz
c: mit sättigbarer Induktivität Lz
with saturable inductance Lz
iSt ≤ iT(RC)M
tSt
t
Figure 47 Schematic progression of the turn-on current of thyristors with various series inductances
Abb./Fig.40 Schutzbeschaltung zur Begrenzung der Einschaltstromsteilheit
a: maximum permissible region
b: non-permissible operation without limiting of the rate of rise of current
c: permissible operation with linear series inductance in the load circuit
d: permissible operation with series saturation choke in the load circuit
60
7.4 Reduction of interference pulses in the gate circuit
Converters produce steep current and voltage changes in the load circuit. This bears the risk of interference pulses
appearing at the gate terminal of thyristors as a consequence of inductive or capacitive coupling onto the gate leads
and trigger electronics. The thyristors can therefore be triggered inadvertently and cause an operation fault in the
installation.
Technischer Erläuterungen - Bilder
The usual measures to reduce coupling and to avoid interference pulses are twisting and possibly shortening of the
gate leads as well as improved shielding even of trigger transformers and possibly trigger electronics. In addition
the gate circuit can be protected (see Figure 48).
A
+
vL
=
–
DX
< 300mm G
Steuerelektronik
control circuit
RX
CX
HK
K
Abb./Fig.41 Prinzipielle Schutzbeschaltung der Steuerstrecke
Figure 48 Example of gate protection of thyristors
For standard phase controlled thyristors the following is recommended:
n Cx = 10...47nF
n Rx according to tX = RxCx = 10...20µs
n Dx fast diode
The discharge resistor Rx may not be omitted as otherwise some thyristor data such as the critical rate of voltage rise
(dv/dt)cr could detiriorate. If the snubber influences the control current adversely, this has to be taken into account
when dimensioning the trigger circuit (see also 3.3.1.8).
61
8. Mounting
The proper and careful mounting of semiconductors is mandatory for reliable and
undisturbed operation as this achieves both thermal and electrical contact.
8.1 Disc cases
8.1.1 Mounting of disc cells
Infineon offers a multitude of heatsinks and stacks. These are designed in conjunction
with Infineon semiconductor components. For many of these heatsinks detailed thermal
data is available on request.
As some of the disc cell heatsinks are complex to clamp, it is recommended in these
cases to purchase both the components and the heatsinks as a complete stack from
Infineon.
When mounting the components onto the heatsinks or busbar connecting with clamping
plates both thermal and electrical contact is achieved.
For this reason the procedures listed in the following must be closely adhered to:
n The contact surfaces of disc cells and heatsinks as well as the insulation must not be
damaged and have to be free of deposits.
n The contact area of disc cells and heatsinks must not exceed the values for flatness
and surface roughness Rz 10µm for the heatsink.
n Prior to mounting the contact surfaces should be coated with approx. 50µm – 100µm
of suitable electrically conductive heat transfer compound (e.g. Klüber Wolfracoat C),
depending on condition of the heatsink contact surface. If a terminal busbar is placed
between disc cell and heatsink, than this should also be treated accordingly.
Typical mounting arrangements are outlines in Figure 49 and Figure 50.
62
Technischer
Technischer
Erläuterungen
Erläuterungen
- Bilder- Bilder
Technischer Erläuterungen - Bilder
Figure 49 Typical clamping arrangement for disc cells
Abb./Fig.42 Typische
Abb./Fig.42
Spannvorrichtungen
Typische Spannvorrichtungen
für Scheibenzellen
für Scheibenzellen
Figure 50 Typical clamping arrangement V176 for disc cells
n Sufficient stiffness of the parts to be clamped has to be assured so that the required clamping forces will not
deform the heatsink contact surfaces and that a homogeneous pressure distribution is achieved (see Figure 49
and Figure 50). The deflection must not exceed the value of 0.3µm per mm of contact surface diameter D in fully
clamped condition. (Example: contact surface diameter 80 x 0.3µm/mm = 24µm maximum delection)
n A maximum of 0.5% of the surface of any contact area may show pitting greater than the specified roughness.
However, the nickel layer must not be damaged.
n Recommendation for dimensioning (see right side of Figure 50): The height of the pressure distribution bolt is to
be dimensioned with h = 0.4D. The application of pressure force is to be performed with Ø d =0.4D.
We recommend the use of steel (e.g. X20Cr13 conforming to EN 10099).
63
n If centering is not achieved otherwise, when mounting the components it has to be
assured that centre holes and pins in the heatsink half shells are present.
Also it has to be observed that the contact surface of the semiconductor is completely
contacted. That means that the contact surface of the heatsink or busbar is at least as
wide as the contact diameter of the semiconductor.
n When selecting the centering pins, the correct diameter and in particular the correct
length has to be assured. Because the contacts of the semiconductors are made of
very soft (easily deformable) copper, too long centering pins can push through to the
pellet and damage the semiconductor.
n During assembly or disassembly the bolts must be tightened or loosened crosswise
and alternately at a small angle to avoid damaging the disc cell.
n For single sided cooling of disc cells stacking has to be carried out with suitable
clamping arrangements such as the types V50, V61 and V72. It now has to be assured
that the mounting bolts are tightened in several steps and crosswise. For the types
listed above the required clamping force has automatically been reached when the
free ends of the clamping arrangement touch the contact surface.
n When using clamping arrangements with load current conducting centre bolts such as
V50M, V61M and V72M, the maximum torques for their threads have to be observed.
Figure 51 shows the typical dependence of the thermal resistance RthJC of disc cells on
the clamping force.
1,5
1,4
RthJC nominal = RthJC(Fmin)
RthJC / RthJC nominal
1,3
1,2
1,1
1,0
0,9
0,8
0,25
0,50
0,75
1,00
F / Fmin
Figure 51 Typical dependence of RthJC on the clamping force F
64
1,25
1,50
n As depicted above too low a clamping force results in an increase of the thermal resistance which leads to a
reduction of the semiconductor’s current loading capability. Also the on-state voltage increases and the surge
current behaviour may change adversely.
A severe reduction in clamping force may also let the thermal cycling capability deteriorate.
n Too high a clamping force may lead to ageing and damage of the disc cells internal contacts (metallisation) which
once again can severely reduce the thermal load cycling capability.
If it is intended to mount the disc cells with a clamping force significantly above the upper limit given in the data
sheet, it is recommended to forward an inquiry to Infineon.
n The clamping force selected should thus be in the top third of the specified force range. This should ensure that
even in the case of minimum expansion and compression processes with the materials employed, the level of
force does not fall below the minimum requirement.
n For testing purposes disc cells have to be clamped with at least 10% of the minimum nominal clamping force or
1kN (which ever value is lower will suffice) in order to assure safe contact between element and contact surfaces
of the pellet.
n For tests with load currents at least the minimal clamping force has to be applied, as the data sheet values are
only valid for the specified clamping force window.
n Correct measurements in an unclamped condition are not possible.
n In the case of the Medium Power ceramic housings with multi use gate, we recommend the use of the flat
connector for the gate connection
65
8.1.2 Positioning the heatsinks
Mounting disc cells in heatsinks for forced air cooling (F) and water cooling (W) can be
done in any position as long as the coolant quantities are adhered to.
For natural air cooling (S) the heatsinks are to be positioned such that the fins take a
vertical position and the cooling air can pass uninhibited.
The heatsinks are to be mounted with sufficient distance from the ground and other
equipment.
If several heatsinks are mounted on top of each other, a sufficiently large gap has to be
left in particular for natural air cooling to avoid mutual heating effects. If necessary, a
higher coolant temperature has to be taken into account for the upper heatsinks.
If several devices are combined in a stack, the following points are to be noted.
n In series connection of several devices a multiple of the blocking voltage of each
individual device may be achieved. This need to be taken into consideration when
designing the insulation of the clamping arrangements.
n In parallel connection of several devices side by side, clamping of the devices bet-
ween two continuous heatsink half-shells is not permitted. The height tolerances of
the devices prevent homogenous application of the pressure force. Instead, the heatsink half-shells should be mechanically separated in order for the two clamp systems
to work mechanically independent.
n If individual busbar connections are connected in a stack, it must be prevented that
unacceptable forces are applied to the stack especially during assembly.
n If disc cells on heatsinks are heated up by other equipment or components such as
fuses or transformers, their load must be reduced accordingly.
n The heatsinks carry potential and are thus to be mounted isolated.
8.1.3 Connection of busbars
It needs to be noted that:
n no additional pull or push forces are applied to the disc cells,
n mechanical oscillations which occur will not cause a ground or short circuit,
n additional heating up of the disc cells by load current carrying components, in
particular directly connected fuses is avoided by design.
66
8.1.4 Connection of the control leads
The following has to be noted:
n Bending or pre-fracturing of the control terminals by improper assembly must be avoided.
n A safe contact of the pin connection is to be assured.
n The gate leads need to be positioned EMC-compatibly and gate protection circuits are positioned in close
proximity to the semiconductors.
n The insulation between control and load circuit needs to be coordinated correctly. This is particularly important
for the galvanic separation of the trigger circuit with trigger transformers.
8.2 Stud cases
8.2.1 Mounting stud cases
Devices with stud case have to be fastened with a torque wrench which is to be positioned radially to the case in
order not to damage the ceramic insulation body. The torque values detailed in the individual datasheets is to be
adhered to with a tolerance of +10%/-20%.
Damaged threads or insufficient hole depth may let the torque be reached without the contact areas touching. The
heat transfer occurs only via the thread in that case which may lead to a thermal overload of the device.
The following notes must definitely be taken into account:
n The contact areas of stud cases and the heatsinks as well as the insulation body must not be damaged and must
be free of deposits.
n In the contact areas of stud cases and heatsinks the deviation in flatness should be within 10µm and the surface
roughness Rz of the heatsink must not exceed 10µm for stud mounting.
n Prior to assembly the contact areas should be coated with a suitable electrically conductive heat transfer
compound (e.g. Klüber Wolfracoat C), with a thickness of approx. 50-100µm. If a terminal busbar is placed between stud and heatsink, this should also be coated.
8.2.2 Positioning the heatsinks
Mounting stud cases in heatsinks for forced air cooling (F) can be done in any position as long as the required
coolant quantities are adhered to.
For natural air cooling (S) the heatsinks are to be positioned such that the fins take a vertical position and the
cooling air can pass uninhibited.
67
The heatsinks are to be mounted with sufficient distance from the ground and other
equipment.
If several heatsinks are mounted on top of each other, a sufficiently large gap has to be
left in particular for natural air cooling to avoid mutual heating effects. If necessary, a
higher coolant temperature has to be taken into account for the upper heatsinks.
If components on heatsinks are heated up by other equipment or components such as
transformers, their load must be reduced accordingly.
The heatsinks carry potential and are thus to be mounted isolated.
8.2.3 Connection of busbars
It needs to be noted that:
nno additional pull or push forces are applied to the components,
nmechanical oscillations which occur will not cause a ground or short circuit,
nadditional heating up of the semiconductors by load current carrying components, in
particular directly connected fuses is avoided by design.
nthe minimum bending radius of the flexible leads must be observed.
8.2.4 Connection of the control leads
See section 8.1.4
8.3 Flat base cases
8.3.1 Mounting flat base devices
The required clamping force is applied by the clamping plate supplied. When heatsinks
made of copper or aluminium are used then the length of the four bolts is to be such
that it reaches into the threaded section at least 50% further than the bolt diameter. The
required clamping force has been reached when the mounting bolts have been tightened in several steps and crosswise in such a way that the clamping plate is in parallel
position to the contact surface.
The following notes should definitely be taken into account:
nThe contact areas of flat base devices and the heatsinks as well as the insulation body
must not be damaged and must be free of deposits.
nIn the contact areas of flat base devices and heatsinks the deviation in flatness
should be within 10µm and the surface roughness Rz of the heatsink must not exceed
10µm for flat base mounting.
68
nPrior to assembly the contact areas should be coated with a suitable electrically conductive heat transfer
compound (e.g. Klüber Wolfracoat C), with a thickness of approx. 50-100µm. If a terminal busbar is placed
between flat base and heatsink, this should also be coated.
8.3.2 Positioning the heatsinks
See also section 8.2.2.
8.3.3 Connection of busbars
See also section 8.2.3.
8.3.4 Connection of the control leads
See section 8.1.4
8.4 PowerBLOCK-Modules
8.4.1 Mounting PowerBLOCK-modules
The contact surfaces of the modules and the heatsink have to be free of damage and deposits. The contact surface
of the heatsink may not exceed the value of 10µm for flatness and roughness Rz. Prior to mounting the contact surfaces should be coated with a layer of approximately 50µm – 100µm of suitable heat transfer compound (e.g. DOW
CORNING DC340), depending on condition of the heatsink contact surface.
A maximum of 0.5% of the surface of any contact area may show pitting greater than the specified roughness.
However, the nickel layer of the module base plate must not be damaged.
All mounting bolts are to be tightened evenly with the specified torque.
8.4.2 Positioning the heatsinks
Mounting PowerBLOCK-modules in heatsinks for forced air cooling (F) and water cooling (W) can be done in any
position as long as the coolant quantities are adhered to.
For natural air cooling (S) the heatsinks are to be positioned such that the fins take a vertical position and the
cooling air can pass uninhibited.
The heatsinks are to be mounted with sufficient distance from the ground and other equipment.
If several heatsinks are mounted on top of each other, a sufficiently large gap has to be left in particular for natural
air cooling to avoid mutual heating effects. If necessary, a higher coolant temperature has to be taken into account
for the upper heatsinks.
If several modules are connected in series on the same heatsink, the specified isolation voltage is generally no
longer sufficient. Infineon does not recommend this form of construction.
If modules on heatsinks are heated up by other equipment or components such as transformers, their load must be
reduced accordingly.
The number of modules per heatsink is to be chosen so that cross talking between them is avoided or considered in
the calculation.
69
8.4.3 Connection of busbars
It needs to be noted that:
nno additional pull or push forces are applied to the modules,
nmechanical oscillations which occur will not cause a ground or short circuit,
nadditional heating up of the modules by load current carrying components, in
particular directly connected fuses is avoided by design.
8.4.4 Connection of the control leads
See section 8.1.4
9. Maintenance
Thyristors and diodes as solid state components are virtually maintenance free. Their
isolation paths, however, are not protected against splashing or dropping water as well
as contamination. In order not to affect the insulation capability and the heat transfer,
the components and in particular their isolation paths as well as the heatsinks are to be
cleaned regularly.
10.Storage
After receipt of the shipment, the disc cells and PowerBLOCK-modules may be stored in
their original packaging for a period of at least 2 years, subject to suitable conditions
prevailing. For this, climatic conditions should conform to IEC 60721-3-1 Class 1K2.
70
11.Type designation
Disc cell
PowerBLOCK Module
T 930N 36 T O
TT 162 N 16 K O F -K
F
TTPowerBLOCK-module with 2
T
Symmetrically blocking thyristor
D
Diode
930
Maximum average on-state current (A)
DDPowerBLOCK-module with 2 diodes
0
Medium power ceramic disc
ND, DZ, TZPowerBLOCK-module with 1
1
High power ceramic disc
3Light triggered thyristor in ceramic
thyristors
thyristor or 1 diode
TD, DTPowerBLOCK-module with 1
thyristor and 1 diode
housing (LTT → Light Triggered
Thyristor)
Application:
162Maximum average on-state current
(A)
NPhase control diode, phase control
thyristor
NHPulsed Power Diode with Soft
Recovery, LTT with high turn-on-di/dt
Application:
NPhase control diode, phase control
thyristor
KPhase control diode with cathode on
case (stud and flat base housings)
SFast diode
SHFast diodes soft recovery - GCT, IGCT
S Fast diode
Blocking voltage:
16Repetitive forward off-state and
reverse peak voltage in 100V
and IGBT free wheeling diodes
UFast diode with cathode on case
(only stud and flat base housings)
A
Avalanche diode
16 = 1600V
Design:
APower block module with
Advanced Medium Power
BAvalanche diode with cathode on
Technology
the case (only stud and flat base
housings)
Blocking voltage:
36repetitive forward off-state and
K Power block module
Turn-off time:
Ono guaranteed turn-off time (see
data sheet)
reverse peak voltage in 100V
36 = 3600V
BStud base metric thread with cable
CStud base metric thread with solder
EFlat base housing with cable
T
terminal
Disc cell
Turn-off time:
Ono guaranteed turn-off time (see data
Critical rate of rise of off-state
voltage:
Design:
C500 V/µs
F 1000 V/µs
G1500 V/µs
Connection type:
-K construction with common cathode
-A construction with common anode
Special type:
sheet)
B01…nConstruction variant
Critical rate of rise of off-state voltage:
S01…nspecial electric selection
C500 V/µs
F1000 V/µs
G1500 V/µs
H2000 V/µs
Special type:
B01…n
Construction variant
S01…n
special electric selection
71
12.Circuit topologies
ideal with inductive filtering
AC-content
of the DCvoltage WU
Frequency
of the
superimpo
sed ACvoltage
rms
voltage per
arm
rms
current per
arm
%
Hz
U2RMS
I2RMS
1,8
+
Phase curren
I1RMS
M1
2
Id
I2
Voltage
diagram
1,6
1,4
1,2
121
Ud
U1
1
Ii0
0,8
0,6
0,4
50
0,2
-
I1
0
50
100
150
200
250
300
350
360° el
2.22 *
Udi
1.57 *
Id
B2
U2
* Id
U1
1,6
1,4
1,2
Ud
1
48
0,8
0,6
U2
Iin0
0,4
100
0,2
-
0
0
50
100
150
200
250
300
350
360° el
Id
1.11 *
Udi
0.707 *
Id
+
2
1,8
1,6
1,4
I1
I2
1,2
1
48
0,8
0,6
Ii0
0,4
100
0,2
0
0
50
100
150
200
250
300
350
360° el
1.11 *
Udi
U2
U1
U2
* Id
U1
1,8
+
1.21*
M2
2
Id
I2
0
Ud
Two-pulse
bridge
connection
B2
B2C
I1
U2
Two-pulse
centre-tap
connection
M2
M2C
Effective circuit
U1
Single pulse
connection
M1
M1C
Vector diagram
of the
component side
AC-voltage
connection of
converter
transformer
according to
VDE 0558
U1
Circuit topology
according to
DIN 41761
Id
-
M3
I1 sqrt(3)
2
U1
1,8
U2
Three-pulse
star
connection
M3
M3C
0.47*
1,6
I2
1,4
1,2
18
1
0,8
Id
0,6
+
0,4
150
Ud
0,2
0
0
50
100
150
200
250
300
350
360° el
-
0.855 *
Udi
U2
U1
0.58 *
Id
M6
I2
U2
Six-pulse
star
connection
M6
M6C
U1
I1 sqrt(2)
e.g. Dyn 5
1,6
1,4
1,2
4.2
1
Id
0,8
+
0,6
0,4
300
0,2
0
0
50
100
150
200
250
300
350
Ud
360° el
0.74 *
Udi
0.577*
U2
* Id
U1
2
1,8
0.408 *
Id
e.g. Dyn 5
M3.2
U2
U1
I2
I1
1,4
1,2
0,8
+
0,6
0,4
0
0
50
100
150
200
250
300
350
360° el
Ud
Id
0.855 *
Udi
0.289 *
Id
B6
+
U1
2
1,8
1,6
I2
1,4
1,2
Uv2
4.2
1
Ud
0,8
0,6
0,4
0,2
0
0
50
100
150
200
250
300
350
360° el
300
0.427 *
Udi
0.82 *
Id
-
e.g. Yy0
W1C
W3C
1
0,8
0,6
ITRMS
ITAV
0,4
0,2
I1
0
0
U2
U1
-0,2
-0,4
-0,6
-0,8
-1
72
300
0,2
e.g. Yyn0, yn6
Six-pulse
bridge
connection
B6
B6C
Anti-parallel
connection
W1C
W3C
4.2
1
Id
50
100
150
200
250
300
350
0.408*
U2
* Id
U1
2
1,8
1,6
U2
Double
three-pulse
star
connection
M3.2
M3.2C
0.82*
U2
U1
Phase current
r
Transformer
nominal power
PTR=
M1
M2
1.21*
I1RMS
P2
P1
PTR
RMS
IpRMS
U2
* Id
U1
3.49*Pdi
2.69*Pdi
3.1*Pdi
1.57*Id
1.57*Pdi
1.11*Pdi
1.34*Pdi
0.707*Id
1.11*Pdi
1.11*Pdi
1.11*Pdi
0.707*Id
1.48*Pdi
1.21*Pdi
1.35*Pdi
0.58*Id
1.81*Pdi
1.28*Pdi
1.55*Pdi
0.408*Id
1.48*Pdi
1.05*Pdi
1.26*Pdi
0.289*Id
1.05*Pdi
1.05*Pdi
1.05*Pdi
0.58*Id
U2
* Id
U1
*
B2
M3
M6
U2
* Id
U1
0.47*
U2
* Id
U1
0.577*
U2
* Id
U1
*
M3.2
0.408*
U2
* Id
U1
*
B6
W1C
W3C
0.82*
P1 + P2
2
Branch current
U2
* Id
U1
Peak blocking
voltage
average
Ipar
Id
Current
conduction
angle
Nominal DC-voltage
(VDE 0588 /
IEC60146-1-1)
Uim
Ud
U2RMS* 2
180°el
2*U2RMS* 2
180°el
2
* U 2RMS
0.45*U2RMS
0.5*Id
2 2
* U2RMS
0 9*U2RMS
.
0.5*Id
U2eff* 2
180°el
2 2
* U2RMS
0.9*U2RMS
0.33*Id
1.73*U2RMS*
120°el
2
3 3
2*
* U2RMS
1.17*U2RMS
0.17*Id
2*U2RMS* 2
60°el
3 2
* U2RMS
1.35*U2RMS
0.17*Id
2*U2RMS* 2
120°el
3 3
2*
* U2RMS
1.17*U2RMS
0.33*Id
1.73*U2RMS*
120°el
2
3 2
* U v 2RMS
1.35*Uv2RMS
I1RMS
*0.707
I1RMS
*0.45
U1RMS* 2
180°el
73
A1. Abbreviations
C
Cnull
E
Etot
f
f0
F
G
iD
iG
IA
IGD
iGM
IGT
IH
IL
iR
IRM
IRMS
IRMS(case)
iT/iF
ITAV/IFAV
ITAVM/IFAVM
ITINT/IFINT
ITM/IFM
IT(OV)/IF(0V)
IT(OV)M/IF(OV)M
IT(RC)M
ITRM/IFRM
ITRMSM/IFRMSM
ITSM/IFSM
∫i²dt
diG/dt
diT/dt
(diT/dt)cr
L
M
P
PD
PG
PR
PRQ
PRSM
PT/PF
PTAV/PFAV
PTT/PFT
PRQ
Ptot
Qr
R
rT
RthCA
RthCH
RthJA
RthJC
t
74
Kapazität
Nullkapazität
Verlustenergie
Gesamtverlust-Energie
Frequenz
Wiederholungsfrequenz
Anpresskraft
Gewicht
Vorwärts-Sperrstrom
Steuerstrom
Ausgangsstrom
nicht zündender Steuerstrom
Spitzensteuerstrom
Zündstrom
Haltestrom
Einraststrom
Rückwärts-Sperrstrom
Rückstromspitze
Strom-Effektivwert
Gehäusegrenzstrom
Durchlassstrom Thyristor/ Diode (Augenblickswert)
Durchlassstrom Thyristor/ Diode (Mittelwert)
Dauergrenzstrom Thyristor/ Diode (Mittelwert)
Höchstzulässiger Durchlassstrom bei Aussetzbetrieb
Durchlassstrom Thyristor/ Diode (Spitzenwert)
höchstzulässiger Überstrom bei Kurzzeitbetrieb
Grenzstrom
höchstzulässiger periodischer Einschaltstrom (aus RC)
höchstzulässiger periodischer Spitzenstrom
höchstzulässiger effektiver Durchlassstrom
Stoßstrom-Grenzwert
Grenzlastintegral
Steilheit des Steuerstroms
Steilheit des Durchlassstroms
kritische Stromsteilheit
Induktivität
Anzugsdrehmoment
Verlustleistung
Sperrverlustleistung (Vorwärtsrichtung)
Steuerverlustleistung
Sperrverlustleistung (Rückwärtsrichtung)
Ausschaltverlustleistung
Stoßsperrverlustleistung
Durchlassverlustleistung Thyristor/ Diode
Durchlassverlustleistung Thyristor/ Diode (Mittelwert)
Einschaltverlustleistung Thyristor/ Diode
Ausschaltverlustleistung
Gesamtverlustleistung
Sperrverzögerungsladung
Widerstand
Ersatzwiderstand
Wärmewiderstand Gehäuse-Kühlmittel
Wärmewiderstand Gehäuse-Kühlkörper
Gesamtwärmewiderstand
innerer Wärmewiderstand
Zeit
capacitance
zero capacitance
energy
total energy
frequency
repetition frequency
clamping force
weight
forward off-state current
gate current
RMS on-state current
gate non-trigger current
peak gate current
gate trigger current
holding current
latching current
reverse current
peak reverse recovery current
RMS current
peak case non-rupture current
on-state current thyristor/diode (instantaneous value)
on-state current thyristor/diode (average value)
maximum average on-state current thyristor/diode
maximum permissible on-state current in intermittent duty
on-state current thyristor/diode (peak value)
maximum permissible on-state current in short-time duty
maximum permissible overload on-state current
maximum permissible turn-on current (from snubber)
maximum permissible repetitive peak on-state current
maximum permissible RMS on-state current
maximum rated on-state surge current
maximum rated value ∫i²dt
rate of rise of gate current
rate of rise of on-state current
critical rate of rise of on-state current
inductance
tightening torque
power losses
forward off-state power losses
gate power losses
reverse power losses
turn-off power losses
surge non-repetitive reverse power losses
on-state power losses thyristor/diode
on-state power losses thyristor/diode (average value)
turn-on power losses thyristor/diode
turn-off power losses
total power losses
recovered charge
resistance
slope resistance
thermal resistance, case to coolant
thermal resistance, case to heat sink
thermal resistance, junction to coolant
thermal resistance, junction to case
time
T
TA
TC
tG
tgd
tfr
TH
tp
tq
trr
Tstg
Tvj
tvj max
Top
tW
VA
V(B0)
V(B0)0
VD
VDM
VD (DC)
VDRM
VDSM
VG
VGD
VGT
VFRM
VISOL
VL
VR
VRM
VR (DC)
VRM
VRRM
VRSM
VT/VF
V(T0)
dvD/dt
(dv/dt)cr
VL
VW
W
Wtot
Z(th)CA
Z(th)JA
Z(th)JC
θ
Periodendauer
Kühlmitteltemperatur
Gehäusetemperatur
Steuerimpulsdauer
Zündverzug
Durchlassverzögerungszeit
Kühlkörpertemperatur
Strompulsdauer (Sinusform)
Freiwerdezeit
Sperrverzugszeit
Lagertemperatur
Sperrschichttemperatur
höchstzulässige Sperrschichttemperatur
Betriebstemperatur
Stromflusszeit (Trapezform)
Ausgangsspannung
Kippspannung
Nullkippspannung
Vorwärts-Sperrspannung (Augenblickswert)
Vorwärts-Sperrspannung (Spitzenwert)
Vorwärts-Gleichsperrspannung
höchstzulässige periodische VorwärtsSpitzensperrspannung
höchstzulässige VorwärtsStoßspitzensperrspannung
Steuerspannung
nicht zündende Steuerspannung
Zündspannung
Durchlassverzögerungsspannung
Isolations-Prüfspannung
Steuergenerator-Leerlaufspannung
Rückwärts-Sperrspannung (Augenblickswert)
Rückwärts-Sperrspannung (Spitzenwert)
Rückwärts-Gleichsperrspannung
Rückwärts-Sperrspannung (Spitzenwert)
Höchstzulässige periodische Rückwärts-Sperrspannung
höchstzulässige Rückwärts-Stoßspitzensperrspannung
Durchlassspannung Thyristor/Diode (Augenblickswert)
Schleusenspannung
Steilheit der Vorwärts-Spannung
kritische Spannungssteilheit
Luftmenge
Wassermenge
Verlust-Energie
Gesamtenergie
Transienter äußerer Wärmewiderstand
Transienter Gesamtwärmewiderstand
Transienter innerer Wärmewiderstand
Stromflusswinkel
period
coolant temperature
case temperature
trigger pulse duration
gate controlled delay time
forward recovery time
heatsink temperature
current pulse duration (sinusoidal)
circuit commutated turn-off time
reverse recovery time
storage temperature
junction temperature
maximum permissible junction temperature
operating temperature
current pulse duration (trapezoidal)
output voltage
forward breakover voltage
forward breakover voltage, gate open
forward off-state voltage (instantaneous value)
forward off-state voltage (peak value)
forward DC off-state voltage
maximum permissible repetitive peak
forward off-state voltage
maximum permissible non-repetitive peak
forward off-state voltage
gate voltage
gate non trigger voltage
gate trigger voltage
forward recovery voltage
insulation test voltage
no-load voltage of trigger pulse generator
reverse voltage (instantaneous value)
reverse voltage (peak value)
reverse DC voltage
reverse voltage (peak value)
maximum permissible repetitive peak reverse voltage
maximum permissible non-repetitive peak reverse voltage
on-state voltage thyristor/diode (instantaneous value)
threshold voltage
rate of rise of forward off-state voltage
critical rate of rise of off-state voltage
air quantity
water quantity
energy
total energy
transient thermal impedance, case to coolant
transient thermal impedance, junction to coolant
transient thermal impedance, junction to case
current conduction angle
75
A2. List of Figures
Figure 1: Schematic construction of a diode 7
Figure 2 Characteristics of a diode
8
Figure 3: Schematic construction of a thyristor
9
Figure 4 Characteristics of a thyristor
9
Figure 5 Designation of the terminals
11
Figure 6 Construction concepts of pressure contact components
12
Figure 7 Typical dependence of the off-state current iD,R(VDRM,RRM) referenced to ID,R(VDRM,RRM; Tvj max)
on the junction temperature Tvj referenced to Tvj max14
Figure 8 Definition of the off-state voltage occurrences
15
Figure 9 Typical dependence of the latching current IL and holding current lH normalized to Tvj=25°C of the junction temperature Tvj17
Figure10 Example of an on-state characteristic and the matching equivalent line approximation
18
Figure 11 Typical dependence of the maximum overload on-state current IT(OV)M, IF(OV)M
(in relation to the surge current ITSM or IFSM for 10ms and Tvj max) on the number of half-sinewaves at 50Hz.
19
Figure 12 Typical dependence of the maximum overload on-state current IT(OV)M, IF(OV)M (in relation to the surge current ITSM or IFSM
for 10ms and Tvj max) on the time t for a number of half-sinewaves at 50Hz. Parameter: reverse blocking voltage VRM20
Figure 13 Typical dependence of the òi² dt normalized to the value òi² dt (10ms) on the half-sinewave duration tP21
Figure 14 Example for control characteristic vG = f (iG) with trigger area for VD = 12 V
23
Figure 15 Concept of a trigger circuit for thyristors
24
Figure 16 LTT with fibre optic cable
25
Figure 17 Laser diode SPL PL 90 typical dependence of the light power on the control current
26
Figure 18 Recommended current pulse for laserdiode SPL PL 90
26
Figure 19 Safe overdrive of the gate trigger current
27
Figure 20 Schematic representation of a diode turn-on process
28
Figure 21 Schematic representation of a thyristor turn-on process
29
Figure 22 Typical dependence of the gate controlled delay time tgd and the maximum gate current iGM30
Figure 23 Schematic representation of the thyristor and diode turn-off process
32
Figure 24 Typical Tvj-dependence of the recovery charge Qr normalized to Qr(Tvj max)33
Figure 25 Typical di/dt-dependence of the recovery charge Qr normalized to Qr(di/dt=10A/µs)33
Figure 26 Typical Tvj-dependence of the peak reverse recovery current IRM normalized to IRM (Tvj max)34
Figure 27 Typical di/dt-dependence of the peak reverse recovery current IRM normalized to IRM (di/dt=10/µs)
34
Figure 28 Schematic representation of the turn-off behaviour of a thyristor
36
Figure 29 Typical dependence of the turn-off time tq normalized to Tvj max on the junction temperature Tvj37
Figure 30 Typical dependence of the turn-off time tq normalized to the -diT/dtnorm on the off-commutating rate of fall -diT/dt37
Figure 31 Typical dependence of the turn-off time tq normalized to the dvD/dt = 20V/µs on the rate of rise of off-state voltage dvD/dt38
76
Figure 32 Thermal equivalent circuits for diodes and thyristors
42
Figure 33 Example of vT/vF class definition
48
Figure 34 Current sharing imbalance due to different on-state voltages in parallel connection
50
Figure 35 Voltage sharing imbalance due to different leakage currents in series connection
51
Figure 36 Voltage sharing imbalance due to different turn-off properties
51
Figure 37 Schematic representation of the Safe Operation Area (SOA) of a thyristor optimised
for pulsed power with single sine wave current pulses
53
Figure 38 Thyristor switch with free-wheeling circuit at the capacitor side 54
Figure 39 Thyristor switch with free-wheeling circuit at the load side
54
Figure 40 Current and voltage waveforms at the thyristor
55
Figure 41 Examples for extending RC-snubbers for thyristors
58
Figure 42 Snubber circuit for AC-controllers
58
Figure 43 Calculation of the current for an AC-controller
59
Figure 44 Combined snubber on the AC-side of a the controlled rectifier
60
Figure 45 Additional options for protection versus energy intensive overvoltages
63
Figure 46 Turn-off characteristic of superfast fuses
65
Figure 47 Schematic progression of the turn-on current of thyristors with various series inductances
68
Figure 48 Example of gate protection of thyristors
69
Figure 49 Typical clamping arrangement for disc cells
71
Figure 50 Typical clamping arrangement V176 for disc cells
71
Figure 51 Typical dependence of RthJC on the clamping force F
73
77
A3. List of tables
Table 1 Form factors for phase angle control conditions
39
Table 2 RC-snubbers for individual snubbering in mains applications
56
Table 3 Equivalent values for converter circuits
57
Table 4 Components for a combined snubber on the AC-side of controlled three phase bridge
61
Table 5 Calculation of branch (arm) and phase currents
66
Table 6 Abbreviations83
78
A4. Conditions of use
The data contained in this Technical Information is exclusively intended for technically trained staff. You or your
technical departments will have to evaluate the suitability of the products for the intended application and the completeness of the product data provided with respect to such application.
No guarantee of any kind will be given for the product or its properties.
Should you require product information in addition to the contents of this Technical Information which concerns the
specific application and use of the products, please contact the sales office which is responsible for your area (see
www.Infineon.com, sales&contact). For those interested we may provide product datasheets and application notes.
Due to technical requirements our products may contain substances which can endanger your health. For information regarding the substances contained in the specific product please also contact the sales office responsible for
your area.
Should you intend to use the products in aviation applications or in uses where health or life is endangered or in life
support, please contact Infineon.
Please note that for any such application we strongly recommend:
n
to jointly perform a risk and quality assessment,
n
to draw up a quality assurance agreement,
nto establish joint measures for ongoing product monitoring and that delivery of product may depend on such
measures.
If, and to the extent necessary, please forward equivalent notices to your customers.
Changes to this Technical Information are reserved.
79
Notes
80
Products and Innovations
The goal of highest reliability and efficiency in a core technology is always a moving
target; therefore we understand that continuous improvement is essential. On this basis
we have established comprehensive standards with our technologies and our products,
in the power classes ranging from around 10kW to over 30MW per component. These
include for example:
■
■
■
■
PowerBLOCK modules in press-pack technology with currents up to 1100 Ampere
Diodes and thyristors with a silicon diameter up to six inches and blocking voltages
up to 9500 Volts
Light-triggered thyristors with integrated protection functions
Freewheeling diodes for the highest requirements in fast switching applications
such as with IGBTs or IGCTs
600A/9.5 kV Thyristor Technology
for Soft Starter and Power-Supplies
The 9.5 kV thyristor disc is developed and designed for the special requirements in
medium voltage soft starter as well as for medium voltage power supply applications.
For these kinds of applications it is necessary to use several thyristors in series connection. They are optimized to achieve an excellent voltage sharing under all operating
conditions.
The device is designed for a high surge current capability. To ensure a narrow spread of
dynamic parameters which enables best cost designs with less devices in series high
technology production processes are used for this type.
Of course the thyristor is suitable for general purpose line voltage rectifier applications,
e.g. for power supplies or standard electrical drives.
AN2012-01
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Technical Information
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Bipolar Semiconductors
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Order Number: B157-H9716-X-X-7600
Date: 04 / 2012
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