AD OP495GBC

a
FEATURES
Rail-to-Rail Output Swing
Single-Supply Operation: +3 V to +36 V
Low Offset Voltage: 300 mV
Gain Bandwidth Product: 75 kHz
High Open-Loop Gain: 1000 V/mV
Unity-Gain Stable
Low Supply Current/Per Amplifier: 150 mA max
Dual/Quad Rail-to-Rail
Operational Amplifiers
OP295/OP495
PIN CONNECTIONS
8-Lead Narrow-Body SO
(S Suffix)
OUT A
1
8
V+
–IN A
2
7
OUT B
+IN A
3
6
–IN B
V–
4
5
+IN B
APPLICATIONS
Battery Operated Instrumentation
Servo Amplifiers
Actuator Drives
Sensor Conditioners
Power Supply Control
14-Lead Epoxy DIP
(P Suffix)
OUT A
–IN A
GENERAL DESCRIPTION
Rail-to-rail output swing combined with dc accuracy are the key
features of the OP495 quad and OP295 dual CBCMOS operational amplifiers. By using a bipolar front end, lower noise and
higher accuracy than that of CMOS designs has been achieved.
Both input and output ranges include the negative supply, providing the user “zero-in/zero-out” capability. For users of 3.3
volt systems such as lithium batteries, the OP295/OP495 is
specified for three volt operation.
Maximum offset voltage is specified at 300 µV for +5 volt operation, and the open-loop gain is a minimum of 1000 V/mV. This
yields performance that can be used to implement high accuracy
systems, even in single supply designs.
The ability to swing rail-to-rail and supply +15 mA to the load
makes the OP295/OP495 an ideal driver for power transistors
and “H” bridges. This allows designs to achieve higher efficiencies and to transfer more power to the load than previously possible without the use of discrete components. For applications
OP295
1
14 OUT D
13 –IN D
2
+IN A
3
V+
4
+IN B
5
10 +IN C
–IN B
6
9
–IN C
OUT B
7
8
OUT C
12 +IN D
OP495
11
V–
8-Lead Epoxy DIP
(P Suffix)
OUT A
1
8
V+
–IN A
2
7
OUT B
+IN A
3
6
–IN B
V–
4
5
+IN B
OP295
16-Lead SO (300 Mil)
(S Suffix)
OUT A
1
16 OUT D
–IN A
2
15
+IN A
3
14 +IN D
V+
4
+IN B
5
13
OP495
–IN D
V–
12 +IN C
–IN B
6
11
OUT B
7
10 OUT C
NC
8
9
–IN C
NC
NC = NO CONNECT
that require driving inductive loads, such as transformers, increases in efficiency are also possible. Stability while driving
capacitive loads is another benefit of this design over CMOS
rail-to-rail amplifiers. This is useful for driving coax cable or
large FET transistors. The OP295/OP495 is stable with loads in
excess of 300 pF.
The OP295 and OP495 are specified over the extended industrial (–40°C to +125°C) temperature range. OP295s are available in 8-pin plastic and ceramic DIP plus SO-8 surface mount
packages. OP495s are available in 14-pin plastic and SO-16
surface mount packages. Contact your local sales office for
MIL-STD-883 data sheet.
REV. B
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
© Analog Devices, Inc., 1995
One Technology Way, P.O. Box 9106, Norwood. MA 02062-9106, U.S.A.
Tel: 617/329-4700
Fax: 617/326-8703
OP295/OP495–SPECIFICATIONS
ELECTRICAL CHARACTERISTICS (@ V = +5.0 V, V
S
Parameter
Symbol
INPUT CHARACTERISTICS
Offset Voltage
VOS
Input Bias Current
IB
Input Offset Current
IOS
Input Voltage Range
Common-Mode Rejection Ratio
Large Signal Voltage Gain
VCM
CMRR
AVO
Offset Voltage Drift
OUTPUT CHARACTERISTICS
Output Voltage Swing High
∆VOS/∆T
VOH
Output Voltage Swing Low
VOL
Output Current
POWER SUPPLY
Power Supply Rejection Ratio
IOUT
Supply Current Per Amplifier
DYNAMIC PERFORMANCE
Skew Rate
Gain Bandwidth Product
Phase Margin
NOISE PERFORMANCE
Voltage Noise
Voltage Noise Density
Current Noise Density
ISY
PSRR
CM
= +2.5 V, TA = +258C unless otherwise noted)
Conditions
Min
–40°C ≤ TA ≤ +125°C
Max
Units
30
300
800
20
30
±3
±5
+4.0
µV
µV
nA
nA
nA
nA
V
dB
V/mV
V/mV
µV/°C
8
–40°C ≤ TA ≤ +125°C
±1
–40°C ≤ TA ≤ +125°C
0 V ≤ VCM ≤ 4.0 V, –40°C ≤ TA ≤ +125°C
RL = 10 kΩ, 0.005 ≤ VOUT ≤ 4.0 V
RL = 10 kΩ, –40°C ≤ TA ≤ +125°C
Typ
0
90
1000
500
110
10,000
1
RL = 100 kΩ to GND
RL = 10 kΩ to GND
IOUT = 1 mA, –40°C ≤ TA ≤ +125°C
RL = 100 kΩ to GND
RL = 10 kΩ to GND
IOUT = 1 mA, –40°C ≤ TA ≤ +125°C
± 1.5 V ≤ VS ≤ ± 15 V
± 1.5 V ≤ VS ≤ ± 15 V,
–40°C ≤ TA ≤ +125°C
VOUT = 2.5 V, RL = ∞, –40°C ≤ TA ≤ +125°C
4.98
4.90
± 11
5.0
4.94
4.7
0.7
0.7
90
± 18
90
110
5
2
2
V
V
V
mV
mV
mV
mA
dB
85
150
dB
µA
SR
GBP
θO
RL = 10 kΩ
0.03
75
86
V/µs
kHz
Degrees
en p-p
en
in
0.1 Hz to 10 Hz
f = 1 kHz
f = 1 kHz
1.5
51
<0.1
µV p-p
nV/√Hz
pA/√Hz
Specifications subject to change without notice.
ELECTRICAL CHARACTERISTICS (@ V = +3.0 V, V
S
Parameter
INPUT CHARACTERISTICS
Offset Voltage
Input Bias Current
Input Offset Current
Input Voltage Range
Common-Mode Rejection Ratio
Large Voltage Gain
Offset Voltage Drift
OUTPUT CHARACTERISTICS
Output Voltage Swing High
Output Voltage Swing Low
POWER SUPPLY
Power Supply Rejection Ratio
Symbol
VOS
IB
IOS
VCM
CMRR
AVO
∆VOS/∆T
CM
= +1.5 V, TA = +258C unless otherwise noted)
Conditions
Min
0 V ≤ VCM ≤ 2.0 V, –40°C ≤ TA ≤ +125°C
RL = 10 kΩ
0
90
VOH
VOL
RL = 10 kΩ to GND
RL = 10 kΩ to GND
2.9
PSRR
± 1.5 V ≤ VS ≤ ± 15 V
± 1.5 V ≤ VS ≤ ± 15 V,
–40°C ≤ TA ≤ +125°C
VOUT = 1.5 V, RL = ∞, –40°C ≤ TA ≤ +125°C
90
ISY
Typ
Max
Units
30
8
±1
500
20
±3
+2.0
µV
nA
nA
V
dB
V/mV
µV/°C
110
750
1
0.7
2
110
V
mV
dB
85
150
dB
µA
Supply Current Per Amplifier
DYNAMIC PERFORMANCE
Slew Rate
Gain Bandwidth Product
Phase Margin
SR
GBP
θO
RL = 10 kΩ
0.03
75
85
V/µs
kHz
Degrees
NOISE PERFORMANCE
Voltage Noise
Voltage Noise Density
Current Noise Density
en p-p
en
in
0.1 Hz to 10 Hz
f = 1 kHz
f = 1 kHz
1.6
53
<0.1
µV p-p
nV/√Hz
pA/√Hz
Specifications subject to change without notice.
–2–
REV. B
OP295/OP495
ELECTRICAL CHARACTERISTICS (@ V = ±15.0 V, T = +258C unless otherwise noted)
S
Parameter
Symbol
INPUT CHARACTERISTICS
Offset Voltage
VOS
Input Bias Current
IB
Input Offset Current
IOS
Input Voltage Range
Common-Mode Rejection Ratio
Large Signal Voltage Gain
Offset Voltage Drift
VCM
CMRR
AVO
∆VOS/∆T
OUTPUT CHARACTERISTICS
Output Voltage Swing High
VOL
Output Current
IOUT
POWER SUPPLY
Power Supply Rejection Ratio
Conditions
ISY
Supply Voltage Range
VS
Typ
Max
Units
30
300
800
20
30
±3
±5
+13.5
110
4000
1
µV
µV
nA
nA
nA
nA
V
dB
V/mV
µV/°C
± 15
± 25
V
V
V
V
mA
90
85
110
7
±1
–15.0 V ≤ VCM ≤ +13.5 V, –40°C ≤ TA ≤ +125°C
RL = 10 kΩ
RL = 100 kΩ to GND
RL = 10 kΩ to GND
RL = 100 kΩ to GND
RL = 10 kΩ to GND
–15
90
1000
14.95
14.80
VS = ± 1.5 V to ± 15 V
VS = ± 1.5 V to ± 15 V, –40°C ≤ TA ≤ +125°C
VO = 0 V, RL = ∞, VS = ± 18 V,
–40°C ≤ TA ≤ +125°C
PSRR
Supply Current
Min
–40°C ≤ TA ≤ +125°C
VCM = 0 V
VCM = 0 V, –40°C ≤ TA ≤ +125°C
VCM = 0 V
VCM = 0 V, –40°C ≤ TA ≤ +125°C
VOH
Output Voltage Swing Low
A
–14.95
–14.85
dB
dB
175
+36 (± 18)
+3 (± 1.5)
µA
V
DYNAMIC PERFORMANCE
Slew Rate
Gain Bandwidth Product
Phase Margin
SR
GBP
θO
RL = 10 kΩ
0.03
85
83
V/µs
kHz
Degrees
NOISE PERFORMANCE
Voltage Noise
Voltage Noise Density
Current Noise Density
en p-p
en
in
0.1 Hz to 10 Hz
f =1 kHz
f = 1 kHz
1.25
45
<0.1
µV p-p
nV/√Hz
pA/√Hz
Specifications subject to change without notice.
WAFER TEST LIMITS (@ V = +5.0 V, V
S
CM
= 2.5 V, TA = +258C unless otherwise noted)
Parameter
Symbol
Offset Voltage
Input Bias Current
Input Offset Current
Input Voltage Range1
Common-Mode Rejection Ratio
Power Supply Rejection Ratio
Large Signal Voltage Gain
Output Voltage Swing High
Supply Current Per Amplifier
Vos
IB
IOS
VCM
CMRR
PSRR
AVO
VOH
ISY
Conditions
Limit
Units
0 V ≤ VCM ≤ 4 V
± 1.5 V ≤ VS ≤ ± 15 V
RL = 10 kΩ
RL = 10 kΩ
VOUT = 2.5 V, RL = ∞
300
20
±2
0 to +4
90
90
1000
4.9
150
µV max
nA max
nA max
V min
dB min
µV/V
V/mV min
V min
µA max
NOTES
Electrical tests and wafer probe to the limits shown. Due to variations in assembly methods and normal yield loss, yield after packaging is not guaranteed for standard
product dice. Consult factory to negotiate specifications based on dice lot qualifications through sample lot assembly and testing.
1
Guaranteed by CMRR test.
ORDERING GUIDE
Model
Temperature
Range
OP295GP –40°C to +125°C
OP295GS
–40°C to +125°C
OP295GBC +25°C
REV. B
Package
Description
Package
Option
8-Pin Plastic DIP
8-Pin SOIC
DICE
N-8
SO-8
Model
Temperature
Range
OP495GP –40°C to +125°C
OP495GS
–40°C to +125°C
OP495GBC +25°C
–3–
Package
Description
Package
Option
14-Pin Plastic DIP N-14
16-Pin SOL
R-16
DICE
OP295/OP495
ABSOLUTE MAXIMUM RATINGS 1
DICE CHARACTERISTICS
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .± 18 V
Input Voltage2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .± 18 V
Differential Input Voltage2. . . . . . . . . . . . . . . . . . . . . . . +36 V
Output Short-Circuit Duration . . . . . . . . . . . . . . . . . Indefinite
Storage Temperature Range
P, S Package . . . . . . . . . . . . . . . . . . . . . . . . –65°C to +150°C
Operating Temperature Range
OP295G, OP495G . . . . . . . . . . . . . . . . . . . –40°C to +125°C
Junction Temperature Range
P, S Package . . . . . . . . . . . . . . . . . . . . . . . . –65°C to +150°C
Lead Temperature Range (Soldering, 60 Sec) . . . . . . . +300°C
Package Type
uJA3
uJC
Unit
8-Pin Plastic DIP (P)
8-Pin SOIC (S)
14-Pin Plastic DIP (P)
16-Pin SO (S)
103
158
83
98
43
43
39
30
°C/W
°C/W
°C/W
°C/W
OP295 Die Size 0.066 × 0.080 inch, 5,280 sq. mils.
Substrate (Die Backside) Is Connected to V+.
Transistor Count, 74.
NOTES
1
Absolute maximum ratings apply to both DICE and packaged parts, unless
otherwise noted.
2
For supply voltages less than ± 18 V, the absolute maximum input voltage is equal
to the supply voltage.
3
θJA is specified for the worst case conditions, i.e., θJA is specified for device in socket
for cerdip, P-DIP, and LCC packages; θJA is specified for device soldered in circuit
board for SOIC package.
OP495 Die Size 0.113 × 0.083 inch, 9,380 sq. mils.
Substrate (Die Backside) Is Connected to V+.
Transistor Count, 196.
+ OUTPUT SWING – Volts
Typical Characteristics
120
100
VS = +36V
VS = +5V
80
VS = +3V
– OUTPUT SWING – Volts
SUPPLY CURRENT PER AMPLIFIER – µA
140
60
40
20
–50
–25
0
25
50
75
100
TEMPERATURE – °C
Supply Current Per Amplifier vs. Temperature
15.2
VS = ±15V
15.0
RL = 100k
14.8
RL = 10k
14.6
14.4
RL = 2k
14.2
–14.4
RL = 2k
–14.6
R L = 10k
–14.8
–15.0
RL = 100k
–15.2
–50
–25
0
25
50
TEMPERATURE – °C
75
100
Output Voltage Swing vs. Temperature
–4–
REV. B
Typical Characteristics–OP295/OP495
5.10
3.10
VS = +5V
OUTPUT VOLTAGE SWING – Volts
OUTPUT VOLTAGE SWING – Volts
VS = +3V
3.00
RL = 100k
2.90
RL = 10k
2.80
2.70
RL = 2k
2.60
2.50
–50
–25
0
25
50
75
5.00
RL = 100k
RL = 10k
4.90
4.80
4.70
RL = 2k
4.60
4.50
–50
100
–25
0
25
50
75
100
TEMPERATURE – °C
TEMPERATURE – °C
Output Voltage Swing vs. Temperature
Output Voltage Swing vs. Temperature
200
500
BASED ON 1200 OP AMPS
BASED ON 600 OP AMPS
VS = +5V
175
VS = +5V
TA = +25°C
450
TA = +25°C
400
150
350
300
UNITS
UNITS
125
100
250
200
75
150
50
100
25
50
0
–250 –200 –150 –100
–50
0
50
100
150
200
0
–100
250
–50
INPUT OFFSET VOLTAGE – µV
OP295 Input Offset (VOS) Distribution
300
500
BASED ON 1200 OP AMPS
BASED ON 600 OP AMPS
225
200
VS = +5V
450
–40° ≤ T A ≤ +85°C
400
175
350
150
300
UNITS
UNITS
250
OP495 Input Offset (VOS) Distribution
250
125
VS = +5V
–40° ≤ T A ≤ +85°C
250
100
200
75
150
50
100
25
50
0
0
0
0.4
0.8
1.2
1.6
2.0
2.4
2.8
3.2
0
TC – V OS – µV/°C
OP295 TC–VOS Distribution
REV. B
0
50
100
150
200
INPUT OFFSET VOLTAGE – µV
0.4
0.8
1.2
1.6
2.0
TC – V OS – µV/°C
2.4
OP495 TC–VOS Distribution
–5–
2.8
3.2
OP295/OP495–Typical Characteristics
100
20
VS = ±15V
VO = ±10V
VS = +5V
OPEN-LOOP GAIN – V/µV
INPUT BIAS CURRENT – nA
16
12
8
RL = 100k
10
RL = 10k
4
RL = 2k
0
–50
–25
0
25
50
TEMPERATURE – °C
75
1
–50
100
0
–25
25
50
75
100
TEMPERATURE – °C
Open-Loop Gain vs. Temperature
Input Bias Current vs. Temperature
12
40
VS = +5V
VO = +4V
SOURCE
35
OPEN-LOOP GAIN – V/µV
VS = ±15V
SINK
25
SOURCE
20
SINK
15
VS = +5V
10
8
RL = 100k
6
RL = 10k
4
RL = 2k
2
5
0
–50
–25
0
25
50
75
0
–50
100
–25
0
25
50
75
100
TEMPERATURE – °C
TEMPERATURE – °C
Output Current vs. Temperature
OUTPUT VOLTAGE ∆ TO RAIL
OUTPUT CURRENT – mA
10
30
Open-Loop Gain vs. Temperature
1V
100mV
SOURCE
VS = +5V
TA = +25°C
10mV
SINK
1mV
100µV
1µA
10µA
100µA
1mA
10mA
LOAD CURRENT
Output Voltage to Supply Rail vs. Load Current
–6–
REV. B
OP295/OP495
APPLICATIONS
Rail-to-Rail Applications Information
0.1µF
The OP295/OP495 has a wide common-mode input range extending from ground to within about 800 mV of the positive
supply. There is a tendency to use the OP295/OP495 in buffer
applications where the input voltage could exceed the commonmode input range. This may initially appear to work because of
the high input range and rail-to-rail output range. But above the
common-mode input range the amplifier is, of course, highly
nonlinear. For this reason it is always required that there be
some minimal amount of gain when rail-to-rail output swing is
desired. Based on the input common-mode range this gain
should be at least 1.2.
LED
3
VIN
R2
27kΩ
1/2
OP295/
OP495
7
R3
8
OP295/
OP495
3
C1
1500pF
R5
10kΩ
C2
10µF
VOUT
1
4
R4
The input noise is controlled by the MAT03 transistor pair and
the collector current level. Increasing the collector current reduces the voltage noise. This particular circuit was tested with
1.85 mA and 0.5 mA of current. Under these two cases, the input voltage noise was 3.1 nV/√Hz and 10 nV/√Hz, respectively.
The high collector currents do lead to a tradeoff in supply current, bias current, and current noise. All of these parameters will
increase with increasing collector current. For example, typically
the MAT03 has an hFE = 165. This leads to bias currents of
11 µA and 3 µA, respectively. Based on the high bias currents,
this circuit is best suited for applications with low source impedance such as magnetic pickups or low impedance strain gages.
Furthermore, a high source impedance will degrade the noise
performance. For example, a 1 kΩ resistor generates 4 nV/√Hz
of broadband noise, which is already greater than the noise of
the preamp.
VOUT = 4.5V
1 TO 10µF
Figure 1. 4.5 Volt, Low Drop-Out Reference
Low Noise, Single Supply Preamplifier
Most single supply op amps are designed to draw low supply
current, at the expense of having higher voltage noise. This
tradeoff may be necessary because the system must be powered
by a battery. However, this condition is worsened because all
circuit resistances tend to be higher; as a result, in addition to
the op amp’s voltage noise, Johnson noise (resistor thermal
noise) is also a significant contributor to the total noise of the
system.
The collector current is set by R1 in combination with the LED
and Q2. The LED is a 1.6 V “Zener” that has a temperature coefficient close to that of Q2’s base-emitter junction, which provides a constant 1.0 V drop across R1. With R1 equal to 270 Ω,
the tail current is 3.7 mA and the collector current is half that,
or 1.85 mA. The value of R1 can be altered to adjust the collector current. Whenever R1 is changed, R3 and R4 should also be
adjusted. To maintain a common-mode input range that includes ground, the collectors of the Q1 and Q2 should not go
above 0.5 V—otherwise they could saturate. Thus, R3 and R4
have to be small enough to prevent this condition. Their values
and the overall performance for two different values of R1 are
summarized in Table I. Lastly, the potentiometer, R8, is needed
to adjust the offset voltage to null it to zero. Similar performance can be obtained using an OP90 as the output amplifier
with a savings of about 185 µA of supply current. However, the
output swing will not include the positive rail, and the bandwidth will reduce to approximately 250 Hz.
The choice of monolithic op amps that combine the characteristics of low noise and single supply operation is rather limited.
Most single supply op amps have noise on the order of 30 nV/√Hz
to 60 nV/√Hz and single supply amplifiers with noise below
5 nV/√Hz do not exist.
In order to achieve both low noise and low supply voltage operation, discrete designs may provide the best solution. The circuit
on Figure 2 uses the OP295/OP495 rail-to-rail amplifier and a
matched PNP transistor pair—the MAT03—to achieve zero-in/
zero-out single supply operation with an input voltage noise of
3.1 nV/√Hz at 100 Hz. R5 and R6 set the gain of 1000, making
this circuit ideal for maximizing dynamic range when amplifying
low level signals in single supply applications. The OP295/OP495
provides rail-to-rail output swings, allowing this circuit to operate with 0 to 5 volt outputs. Only half of the OP295/OP495 is
used, leaving the other uncommitted op amp for use elsewhere.
REV. B
Q2
Figure 2. Low Noise Single Supply Preamplifier
0.001µF
6
Q1
R6
10Ω
6
R8
100Ω
+5V
4
MAT- 03
R7
510Ω
20k
REF43
5
2
16k
10Ω
2
1
The OP295/OP495 can be used to gain up a 2.5 V or other low
voltage reference to 4.5 volts for use with high resolution A/D
converters that operate from +5 volt only supplies. The circuit
in Figure 1 will supply up to 10 mA. Its no-load drop-out voltage is only 20 mV. This circuit will supply over 3.5 mA with a
+5 volt supply.
2
10µ F
Q2
2N3906
Low Drop-Out Reference
+5V
R1
–7–
OP295/OP495
Table I. Single Supply Low Noise Preamp Performance
R1
R3, R4
en @ 100 Hz
en @ 10 Hz
ISY
IB
Bandwidth
Closed-Loop Gain
IC = 1.85 mA
IC = 0.5 mA
270 Ω
200 Ω
3.15 nV/√Hz
4.2 nV/√Hz
4.0 mA
11 µA
1 kHz
1000
1.0 kΩ
910 Ω
8.6 nV/√Hz
10.2 nV/√Hz
1.3 mA
3 µA
1 kHz
1000
unless this was a low distortion application such as audio. If this
is used to drive inductive loads, be sure to add diode clamps to
protect the bridge from inductive kickback.
Direct Access Arrangement
OP295/OP495 can be used in a single supply Direct Access Arrangement (DAA) as is shown an in Figure 4. This figure shows
a portion of a typical DM capable of operating from a single
+5 volt supply and it may also work on +3 volt supplies with
minor modifications. Amplifiers A2 and A3 are configured so
that the transmit signal TXA is inverted by A2 and is not inverted by A3. This arrangement drives the transformer differentially so that the drive to the transformer is effectively doubled
over a single amplifier arrangement. This application takes advantage of the OP295/OP495’s ability to drive capacitive loads,
and to save power in single supply applications.
Driving Heavy Loads
The OP295/OP495 is well suited to drive loads by using a
power transistor, Darlington or FET to increase the current to
the load. The ability to swing to either rail can assure that the
device is turned on hard. This results in more power to the load
and an increase in efficiency over using standard op amps with
their limited output swing. Driving power FETs is also possible
with the OP295/OP495 because of its ability to drive capacitive
loads of several hundred picofarads without oscillating.
390pF
37.4kΩ
20kΩ
0.1µF
OP295/
OP495
A1
RXA
0.0047µF
Without the addition of external transistors the OP295/OP495
can drive loads in excess of ± 15 mA with ± 15 or +30 volt
supplies. This drive capability is somewhat decreased at lower
supply voltages. At ± 5 volt supplies the drive current is ± 11 mA.
3.3kΩ
475Ω
OP295/
OP495
Driving motors or actuators in two directions in a single supply
application is often accomplished using an “H” bridge. The
principle is demonstrated in Figure 3a. From a single +5 volt
supply this driver is capable of driving loads from 0.8 V to 4.2 V
in both directions. Figure 3b shows the voltages at the inverting
and noninverting outputs of the driver. There is a small crossover
glitch that is frequency dependent and would not cause problems
20kΩ
A2
22.1kΩ
0.1µF
20kΩ
750pF
TXA
0.033µF
20kΩ
1:1
20kΩ
2.5V REF
OP295/
OP495
A3
+5V
2N2222
Figure 4. Direct Access Arrangement
2N2222
A Single Supply Instrumentation Amplifier
The OP295/OP495 can be configured as a single supply instrumentation amplifier as in Figure 5. For our example, VREF is set
V+
equal to
and VO is measured with respect to VREF. The in2
put common-mode voltage range includes ground and the output swings to both rails.
10k
OUTPUTS
0 ≤ VIN ≤ 2.5V 5k
1.67V
10k
10k 2N2907
2N2907
V+
Figure 3a. “H” Bridge
1/2
OP295/
OP495
VIN
3
5
8
6
4
1/2
OP295/
OP495
7
VO
1
2
100
R1
R2
R3
R4
100k
20k
20k
100k
90
VREF
RG
(
VO = 5 +
200k
RG
)V
IN
+ VREF
10
0%
Figure 5. Single Supply Instrumentation Amplifier
2V
2V
1ms
Resistor RG sets the gain of the instrumentation amplifier. Minimum gain is 6 (with no RG). All resistors should be matched in
absolute value as well as temperature coefficient to maximize
Figure 3b. “H” Bridge Outputs
–8–
REV. B
OP295/OP495
common-mode rejection performance and minimize drift. This
instrumentation amplifier can operate from a supply voltage as
low as 3 volts.
To calibrate, immerse the thermocouple measuring junction in a
0°C ice bath, adjust the 500 Ω Zero Adjust pot to zero volts out.
Then immerse the thermocouple in a 250°C temperature bath
or oven and adjust the Scale Adjust pot for an output voltage of
2.50 V, which is equivalent to 250°C. Within this temperature
range, the K-type thermocouple is quite accurate and produces
a fairly linear transfer characteristic. Accuracy of ± 3°C is achievable without linearization.
A Single Supply RTD Thermometer Amplifier
This RTD amplifier takes advantage of the rail-to-rail swing of
the OP295/OP495 to achieve a high bridge voltage in spite of a
low 5 V supply. The OP295/OP495 amplifier servos a constant
200 µA current to the bridge. The return current drops across
the parallel resistors 6.19 kΩ and the 2.55 MΩ, developing a
voltage that is servoed to 1.235 V, which is established by the
AD589 bandgap reference. The 3-wire RTD provides an equal
line resistance drop in both 100 Ω legs of the bridge, thus improving the accuracy.
Even if the battery voltage is allowed to decay to as low as 7 volts,
the rail-to-rail swing allows temperature measurements to
700°C. However, linearization may be necessary for temperatures above 250°C where the thermocouple becomes rather
nonlinear. The circuit draws just under 500 µA supply current
from a 9 V battery.
The AMP04 amplifies the differential bridge signal and converts
it to a single-ended output. The gain is set by the series resistance of the 332 Ω resistor plus the 50 Ω potentiometer. The
gain scales the output to produce a 4.5 V full scale. The
0.22 µF capacitor to the output provides a 7 Hz low-pass filter
to keep noise at a minimum.
ZERO ADJ
200Ω
10-TURNS
50Ω
26.7k
0.5%
1
3
8 0.22µF
2
2
6.19k
1%
1/2
OP295/
OP495
1
100Ω
0.5%
3
The output voltage from the DAC is the binary weighted voltage of the reference, which is gained up by the output amplifier
such that the DAC has a 1 mV per bit transfer function.
332Ω
7
AMP04
2.55M
1%
Figure 8 shows a complete voltage output DAC with wide output voltage swing operating off a single +5 V supply. The serial
input 12-bit D/A converter is configured as a voltage output
device with the 1.235 V reference feeding the current output pin
(IOUT) of the DAC. The VREF which is normally the input now
becomes the output.
+5V
26.7k
0.5%
100Ω
RTD
A 5 V Only, 12-Bit DAC That Swings 0 V to 4.095 V
VO
6
4
+5V
+5V
4.5V = 450°C
0V = 0°C
5
R1
17.8kΩ
1.235
+5V
+1.23V
37.4k
3
IOUT
8
VDD
RFB
DAC8043
VREF
+5V
2
1
2
GND CLK SRI LD
4
7
6
1.235V
ISOTHERMAL
BLOCK
1N914
ALUMEL
1.5M
1%
COLD
JUNCTIONS
K-TYPE
THERMOCOUPLE
40.7µV/°C
475Ω
1%
SPAN ADJ
500Ω
10-TURN
2.1k
1%
ZERO
ADJUST
4
1
VIN
0 + 3V
VO
0V = 0°C
5V = 500°C
6 REF02
2
GND
100kΩ
10-TURN
5V
10kΩ
182k 1.21M
10-TURN 1%
1%
8
OP295/
OP495
3
CR
CHROMEL
NULL ADJ
1.33MΩ 20k
2
AL
Figure 9 shows a self powered 4–20 mA current loop transmitter. The entire circuit floats up from the single supply (12 V to
36 V) return. The supply current carries the signal within the 4
to 20 mA range. Thus the 4 mA establishes the baseline
4.99k
1%
24.9k
1%
R3
5kΩ
4–20 mA Current Loop Transmitter
SCALE
ADJUST
24.3k
1%
100kΩ
Figure 8. A 5 Volt 12-Bit DAC with 0 V to +4.095 Output
Swing
9V
7.15k
1%
R2
41.2k
TOTAL POWER DISSIPATION = 1.6mW
24.9k
AD589
1
R4
DIGITAL
CONTROL
The OP295/OP495’s 150 µA quiescent current per amplifier
consumption makes it useful for battery powered temperature
measuring instruments. The K-type thermocouple terminates
into an isothermal block where the terminated junctions’ ambient temperatures can be continuously monitored and corrected
by summing an equal but opposite thermal EMF to the amplifier, thereby canceling the error introduced by the cold junctions.
D
(4.096V)
4096
4
5
Figure 6. Low Power RTD Amplifier
A Cold Junction Compensated, Battery Powered
Thermocouple Amplifier
VO =
OP295/
OP495
AD589
AD589
8
3
4
100Ω
3
8
2
4
1
1/2
OP295/
OP495
2N1711
4–20mA
RL
100Ω
220pF
100k
HP
5082-2800 1%
Figure 7. Battery Powered, Cold-Junction Compensated
Thermocouple Amplifier
+12V
TO
+36V
220Ω
100Ω
1%
Figure 9. 4–20 mA Current Loop Transmitter
REV. B
–9–
OP295/OP495
current budget with which the circuit must operate. This circuit
consumes only 1.4 mA maximum quiescent current, making 2.6
mA of current available to power additional signal conditioning
circuitry or to power a bridge circuit.
current limit loop. At this point A2’s lower output resistance
dominates the drive to the power MOSFET transistor, thereby
effectively removing the A1 voltage regulation loop from the
circuit.
A 3 Volt Low-Dropout Linear Voltage Regulator
If the output current greater than 1 amp persists, the current
limit loop forces a reduction of current to the load, which causes
a corresponding drop in output voltage. As the output voltage
drops, the current limit threshold also drops fractionally, resulting in a decreasing output current as the output voltage decreases, to the limit of less than 0.2 A at 1 V output. This
“fold-back” effect reduces the power dissipation considerably
during a short circuit condition, thus making the power supply
far more forgiving in terms of the thermal design requirements.
Small heat sinking on the power MOSFET can be tolerated.
Figure 10 shows a simple 3 V voltage regulator design. The
regulator can deliver 50 mA load current while allowing a 0.2 V
dropout voltage. The OP295/OP495’s rail-to-rail output swing
handily drives the MJE350 pass transistor without requiring special drive circuitry. At no load, its output can swing less than the
pass transistor’s base-emitter voltage, turning the device nearly
off. At full load, and at low emitter-collector voltages, the transistor beta tends to decrease. The additional base current is easily handled by the OP295/OP495 output.
The OP295’s rail-to-rail swing exacts higher gate drive to
the power MOSFET, providing a fuller enhancement to the
transistor. The regulator exhibits 0.2 V dropout at 500 mA of
load current. At 1 amp output, the dropout voltage is typically
5.6 volts.
The amplifier servos the output to a constant voltage, which
feeds a portion of the signal to the error amplifier.
Higher output current, to 100 mA, is achievable at a higher
dropout voltage of 3.8 V.
IL < 50mA
MJE 350
VO
44.2k
1%
VIN
5V TO 3.2V
8
1
1/2
OP295/
OP495
4
100µF
3
6V
30.9k
1%
8
2
1/2
OP295/
OP495
100k
5%
AD589
210k
1%
205k
1%
45.3k
1%
45.3k
1%
VO
5
A2
7
1N4148
1.235V
IO (NORM) = 0.5A
IO (MAX) = 1A
+5V
G
1000pF
43k
RSENSE
0.1Ω
1/4W
IRF9531
S
D
6
0.01µF
3
A1
1
1/2
4
OP295/
OP495
Figure 10. 3 V Low Dropout Voltage Regulator
Figure 11 shows the regulator’s recovery characteristic when its
output underwent a 20 mA to 50 mA step current change.
2
REF43
4
2V
STEP
CURRENT
CONTROL
WAVEFORM
90
124k
1%
2.500V
6
Square Wave Oscillator
The circuit in Figure 13 is a square wave oscillator (note the
positive feedback). The rail-to-rail swing of the OP295/OP495
helps maintain a constant oscillation frequency even if the supply voltage varies considerably. Consider a battery powered system where the voltages are not regulated and drop over time.
The rail-to-rail swing ensures that the noninverting input sees
the full V+/2, rather than only a fraction of it.
20mA
OUTPUT
10
0%
20mV
124k
1%
Figure 12. Low Dropout, 500 mA Voltage Regulator with
Fold-Back Current Limiting
100
50mA
2
1ms
Figure 11. Output Step Load Current Recovery
Low-Dropout, 500 mA Voltage Regulator with Fold-Back
Current Limiting
Adding a second amplifier in the regulation loop as shown in
Figure 12 provides an output current monitor as well as foldback current limiting protection.
Amplifier A1 provides error amplification for the normal voltage
regulation loop. As long as the output current is less than 1 ampere, amplifier A2’s output swings to ground, reverse biasing the
diode and effectively taking itself out of the circuit. However, as
the output current exceeds 1 amp, the voltage that develops
across the 0.1 Ω sense resistor forces the amplifier A2’s output
to go high, forward-biasing the diode, which in turn closes the
The constant frequency comes from the fact that the 58.7 kΩ
feedback sets up Schmitt Trigger threshold levels that are directly proportional to the supply voltage, as are the RC charge
voltage levels. As a result, the RC charge time, and therefore the
frequency, remains constant independent of supply voltage. The
slew rate of the amplifier limits the oscillation frequency to a
maximum of about 800 Hz at a +5 V supply.
Single Supply Differential Speaker Driver
Connected as a differential speaker driver, the OP295/OP495
can deliver a minimum of 10 mA to the load. With a 600 Ω
load, the OP295/OP495 can swing close to 5 volts peak-to-peak
across the load.
–10–
REV. B
OP295/OP495
V+
100k
58.7k
3
8
2
4
1/2
OP295/
OP495
1
FREQ OUT
fOSC =
100k
1
< 350Hz @ V+ = +5V
RC
R
C
Figure 13. Square Wave Oscillator Has Stable Frequency
Regardless of Supply Changes
90.9k
10k
V+
2.2µF
VIN
10k
1/4
OP295/
OP495
20k
1/4
OP295/
OP495
100k
SPEAKER
1/4
OP295/
OP495
20k
V+
Figure 14. Single Supply Differential Speaker Driver
High Accuracy, Single-Supply, Low Power Comparator
The OP295/OP495 makes an accurate open-loop comparator.
With a single +5 V supply, the offset error is less than 300 µV. Figure 15 shows the OP295/OP495’s response time when operating
open-loop with 4 mV overdrive. It exhibits a 4 ms response time at
the rising edge and a 1.5 ms response time at the falling edge.
1V
100
90
INPUT
(5mV OVERDRIVE
@ OP295 INPUT)
OUTPUT
10
0%
2V
5ms
Figure 15. Open-Loop Comparator Response Time with
5 mV Overdrive
OP295/OP495 SPICE MODEL Macro-Model
* Node Assignments
*
Noninverting Input
*
Inverting Input
*
Positive Supply
*
Negative Supply
*
Output
*
*
.SUBCKT OP295
1
2 99
50
20
*
* INPUT STAGE
*
I1
99 4
2E-6
R1
1
6
5E3
REV. B
R2
2
5
5E3
CIN 1
2
2E-12
IOS
1
2
0.5E-9
D1
5
3
DZ
D2
6
3
DZ
EOS 7
6
POLY (1) (31,39) 30E-6 0.024
Q1
8
5 4
QP
Q2
9
7
4
QP
R3
8
50 25.8E3
R4
9
50 25.8E3
*
* GAIN STAGE
*
R7
10 98 270E6
G1
98 10 POLY (1) (9,8) –4.26712E-9 27.8E-6
EREF 98 0
(39, 0) 1
R5
99 39 100E3
R6
39 50 100E3
*
* COMMON MODE STAGE
*
ECM 30 98 POLY(2) (1,39) (2,39) 0 0.5 0.5
R12 30 31 1E6
R13 31 98 100
*
* OUTPUT STAGE
*
I2
18 50 1.59E-6
V2
99 12 DC 2.2763
Q4
10 14 50 QNA 1.0
R11 14 50 33
M3
15 10 13 13 MN L=9E-6 W=102E-6 AD=15E-10 AD=15E-10
M4
13 10 50 50 MN L=9E-6 W=50E-6 AD=75E-11 AS=75E-11
D8
10 22 DX
V3
22 50 DC 6
M2
20 10 14 14 MN L=9E-6 W=2000E-6 AD=30E-9 AS=30E-9
Q5
17 17 99 QPA 1.0
Q6
18 17 99 QPA 4.0
R8
18 99 2.2E6
Q7
18 19 99 QPA 1.0
R9
99 19 8
C2
18 99 20E-12
M6
15 12 17 99 MP L=9E-6 W=27E-6 AD=405E-12 AS=405E-12
M1
20 18 19 99 MP L=9E-6 W=2000E-6 AD=30E-9 AS=30E-9
D4
21 18 DX
V4
99 21 DC 6
R10 10 11 6E3
C3
11 20 50E-12
.MODEL QNA NPN (IS=1.19E-16 BF=253 NF=0.99 VAF=193 IKF=2.76E-3
+ ISE=2.57E-13 NE=5 BR=0.4 NR=0.988 VAR=15 IKR=1.465E-4
+ ISC=6.9E-16 NC=0.99 RB=2.0E3 IRB=7.73E-6 RBM=132.8 RE=4
RC=209
+ CJE=2.1E-13 VJE=0.573 MJE=0.364 FC=0.5 CJC=1.64E-13 VJC=0.534
MJC=0.5
+ CJS=1.37E-12 VJS=0.59 MJS=0.5 TF=0.43E-9 PTF=30)
.MODEL QPA PNP (IS=5.21E-17 BF=131 NF=0.99 VAF=62 IKF=8.35E-4
+ ISE=1.09E-14 NE=2.61 BR=0.5 NR=0.984 VAR=15 IKR=3.96E-5
+ ISC=7.58E-16 NC=0.985 RB=1.52E3 IRB=1.67E-5 RBM=368.5 RE=6.31
RC=354.4
+ CJE=1.1E-13 VJE=0.745 MJE=0.33 FC=0.5 CJC=2.37E-13 VJC=0.762
MJC=0.4
+ CJS =7.11E-13 VJS=0.45 MJS=0.412 TF=1.0E-9 PTF=30)
.MODEL MN NMOS (LEVEL=3 VTO=1.3 RS=0.3 RD=0.3
+ TOX=8.5E-8 LD=1.48E-6 NSUB=1.53E16 UO=650 DELTA=10 VMAX=2E5
+ XJ=1.75E-6 KAPPA=0.8 ETA=0.066 THETA=0.01 TPG=1 CJ=2.9E-4
PB=0.837
+ MJ=0.407 CJSW=0.5E-9 MJSW=0.33)
.MODEL MP PMOS (LEVEL=3 VTO=–1.1 RS=0.7 RD=0.7
+ TOX=9.5E-8 LD=1.4E-6 NSUB=2.4E15 UO=650 DELTA=5.6 VMAX=1E5
+ XJ=1.75E-6 KAPPA=1.7 ETA=0.71 THETA=5.9E-3 TPG=–1 CJ=1.55E-4
PB=0.56
+ MJ=0.442 CJSW=0.4E-9 MJSW=0.33)
.MODEL DX D(IS=1E-15)
.MODEL DZ D (IS=1E-15, BV=7)
.MODEL QP PNP (BF=125)
.ENDS
–11–
OP295/OP495
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm)
8-Lead Narrow-Body SO (S Suffix)
8 Lead Plastic DIP (P Suffix)
8
0.280 (7.11)
0.240 (6.10)
1
8
5
0.1574 (4.00)
0.1497 (3.80)
4
PIN 1
0.070 (1.77)
0.045 (1.15)
0.430 (10.92)
0.348 (8.84)
0.015
(0.381) TYP
0.210
(5.33)
MAX
0.022 (0.558)
0.014 (0.356)
0.100
(2.54)
BSC
SEATING
PLANE
0.1968 (5.00)
0.1890 (4.80)
0.195 (4.95)
0.115 (2.93)
0.015 (0.381)
0.008 (0.204)
0.0040 (0.10)
0.0500
(1.27)
BSC
0.0098 (0.25)
0.0075 (0.19)
8°
0°
0.0500 (1.27)
0.0160 (0.41)
16-Lead Wide-Body SO (S Suffix)
8
0.280 (7.11)
0.240 (6.10)
PIN 1
1
9
16
0.2992 (7.60)
0.2914 (7.40)
7
0.325 (8.25)
0.300 (7.62)
0.795 (20.19)
0.725 (18.42)
0.100
(2.54)
BSC
0.070 (1.77)
0.045 (1.15)
SEATING
PLANE
8
1
0.015 (0.38)
0.008 (0.20)
0.4133 (10.50)
0.3977 (10.10)
15 °
0°
0.0118 (0.30)
0.0040 (0.10)
0.0500 (1.27)
BSC
0.0192 (0.49)
0.0138 (0.35)
0.1043 (2.65)
0.0926 (2.35)
0.0125 (0.32)
0.0091 (0.23)
0.0291 (0.74)
x 45 °
0.0098 (0.25)
8°
0°
0.0500 (1.27)
0.0157 (0.40)
PRINTED IN U.S.A.
0.130
(3.30)
MIN
0.160 (4.06)
0.115 (2.92)
0.4193 (10.65)
0.3937 (10.00)
PIN 1
0.015
(0.381)
MIN
0.210
(5.33)
MAX
0.022 (0.558)
0.014 (0.36)
0.0192 (0.49)
0.0138 (0.35)
0°- 15°
14-Lead Plastic DIP (P Suffix)
14
0.0196 (0.50)
x 45 °
0.0099 (0.25)
0.0688 (1.75)
0.0532 (1.35)
0.0098 (0.25)
0.130
(3.30)
MIN
0.160 (4.06)
0.115 (2.93)
0.2440 (6.20)
0.2284 (5.80)
4
1
0.325 (8.25)
0.300 (7.62)
C1806a–10–7/95
5
–12–
REV. B