MOTOROLA MC13156DW

Order this document by MC13156/D
The MC13156 is a wideband FM IF subsystem targeted at high
performance data and analog applications. Excellent high frequency
performance is achieved at low cost using Motorola’s MOSAIC 1.5 bipolar
process. The MC13156 has an onboard grounded collector VCO transistor
that may be used with a fundamental or overtone crystal in single channel
operation or with a PLL in multichannel operation. The mixer is useful to
500 MHz and may be used in a balanced–differential, or single–ended
configuration. The IF amplifier is split to accommodate two low cost
cascaded filters. RSSI output is derived by summing the output of both IF
sections. A precision data shaper has a hold function to preset the shaper for
fast recovery of new data.
Applications for the MC13156 include CT–2, wideband data links and
other radio systems utilizing GMSK, FSK or FM modulation.
• 2.0 to 6.0 Vdc Operation
•
•
•
•
•
•
WIDEBAND FM IF
SYSTEM FOR DIGITAL AND
ANALOG APPLICATIONS
SEMICONDUCTOR
TECHNICAL DATA
DW SUFFIX
PLASTIC PACKAGE
CASE 751E
(SO–24L)
24
1
Typical Sensitivity at 200 MHz of 2.0 µV for 12 dB SINAD
RSSI Dynamic Range Typically 80 dB
High Performance Data Shaper for Enhanced CT–2 Operation
Internal 330 Ω and 1.4 kΩ Terminations for 10.7 MHz and 455 kHz Filters
FB SUFFIX
PLASTIC QFP PACKAGE
CASE 873
32
1
Split IF for Improved Filtering and Extended RSSI Range
3rd Order Intercept (Input) of –25 dBm (Input Matched)
PIN CONNECTIONS
Function
Simplified Block Diagram
LO
In
LO
Emit
24
23
VEE1
22
CAR
Det
RSSI
21
20
VEE2
19
DS
Hold
Data
Out
DS
Gnd
DS
In
18
17
16
15
Mixer
Quad
Demod Coil
14
13
Data
Slicer
Bias
5.0
pF
Bias
LIM Amp
IF Amp
1
2
3
4
5
RF
In 1
RF
In 2
Mix
Out
VCC1
IF
In
6
7
IF
IF
DEC 1 DEC 2
8
9
10
IF
Out
VCC2
LIM
In
11
SO–24L
QFP
RF Input 1
RF Input 2
Mixer Output
VCC1
1
2
3
4
31
32
1
2
IF Amp Input
IF Amp Decoupling 1
IF Amp Decoupling 2
VCC Connect (N/C Internal)
5
6
7
–
3
4
5
6
IF Amp Output
VCC2
Limiter IF Input
Limiter Decoupling 1
8
9
10
11
7
8
9
10
Limiter Decoupling 2
VCC Connect (N/C Internal)
Quad Coil
Demodulator Output
12
–
13
14
11
12, 13, 14
15
16
Data Slicer Input
VCC Connect (N/C Internal)
Data Slicer Ground
Data Slicer Output
15
–
16
17
17
18
19
20
Data Slicer Hold
VEE2
RSSI Output/Carrier Detect In
Carrier Detect Output
18
19
20
21
21
22
23
24
VEE1 and Substrate
LO Emitter
LO Base
VCC Connect (N/C Internal)
22
23
24
–
25
26
27
28, 29, 30
12
ORDERING INFORMATION
LIM
LIM
DEC 1 DEC 2
Device
NOTE: Pin Numbers shown for SOIC package only. Refer to Pin Assignments Table.
This device contains 197 active transistors.
MC13156DW
MC13156FB
Operating
Temperature Range
TA = –40 to +85°C
 Motorola, Inc. 1998
MOTOROLA RF/IF DEVICE DATA
Package
SO–24L
QFP
Rev 2.1
1
MC13156
MAXIMUM RATINGS
Rating
Pin
Symbol
Value
Unit
Power Supply Voltage
16, 19, 22
Vdc
–
VEE(max)
TJ(max)
–6.5
Junction Temperature
150
°C
Storage Temperature Range
–
Tstg
–65 to +150
°C
NOTES: 1. Devices should not be operated at or outside these values. The “Recommended Operating
Conditions” table provides for actual device operation.
2. ESD data available upon request.
RECOMMENDED OPERATING CONDITIONS
Rating
Power Supply Voltage @ TA = 25°C
–40°C ≤ TA ≤ +85°C
Input Frequency
Ambient Temperature Range
Input Signal Level
Pin
Symbol
Value
Unit
4, 9
16, 19, 22
VCC
VEE
0 (Ground)
–2.0 to –6.0
Vdc
1, 2
fin
TA
Vin
500
MHz
–40 to +85
°C
200
mVrms
–
1, 2
DC ELECTRICAL CHARACTERISTICS (TA = 25°C, VCC1 = VCC2 = 0, no input signal.)
Pin
Symbol
Total Drain Current (See Figure 2)
VEE = –2.0 Vdc
VEE = –3.0 Vdc
VEE = –5.0 Vdc
VEE = –6.0 Vdc
19, 22
ITotal
Drain Current, I22 (See Figure 3)
VEE = –2.0 Vdc
VEE = –3.0 Vdc
VEE = –5.0 Vdc
VEE = –6.0 Vdc
22
Drain Current, I19 (See Figure 3)
VEE = –2.0 Vdc
VEE = –3.0 Vdc
VEE = –5.0 Vdc
VEE = –6.0 Vdc
19
Characteristic
Min
Typ
Max
Unit
–
3.0
–
–
4.8
5.0
5.2
5.4
–
8.0
–
–
–
–
–
–
3.0
3.1
3.3
3.4
–
–
–
–
–
–
–
–
1.8
1.9
1.9
2.0
–
–
–
–
1.0
1.1
1.2
Vdc
–
1.7
–
mA
mA
I22
mA
I19
mA
DATA SLICER (Input Voltage Referenced to VEE = –3.0 Vdc, no input signal; See Figure 15.)
Input Threshold Voltage (High Vin)
15
Output Current (Low Vin)
Data Slicer Enabled (No Hold)
V15 > 1.1 Vdc
V18 = 0 Vdc
17
V15
I17
AC ELECTRICAL CHARACTERISTICS (TA = 25°C, VEE = –3.0 Vdc, fRF = 130 MHz, fLO = 140.7 MHz, Figure 1 test
circuit, unless otherwise specified.)
Characteristic
Pin
Symbol
Min
Typ
Max
Unit
1, 14
–
–
–100
–
dBm
Conversion Gain
Pin = –37 dBm (Figure 4)
1, 3
–
–
22
–
dB
Mixer Input Impedance
Single–Ended (Table 1)
1, 2
Rp
Cp
–
–
1.0
4.0
–
–
kΩ
pF
Mixer Output Impedance
3
–
–
330
–
Ω
IF RSSI Slope (Figure 6)
20
–
0.2
0.4
0.6
µA/dB
IF Gain (Figure 5)
5, 8
–
–
39
–
dB
Input Impedance
5
–
–
1.4
–
kΩ
Output Impedance
8
–
–
290
–
Ω
12 dB SINAD Sensitivity (See Figures 17, 23)
fin = 144.45 MHz; fmod = 1.0 kHz; fdev = ±75 kHz
MIXER
IF AMPLIFIER SECTION
2
MOTOROLA RF/IF DEVICE DATA
MC13156
AC ELECTRICAL CHARACTERISTICS (continued) (TA = 25°C, VEE = –3.0 Vdc, fRF = 130 MHz, fLO = 140.7 MHz, Figure 1 test
circuit, unless otherwise specified.)
Characteristic
Pin
Symbol
Min
Typ
Max
Unit
20
–
0.2
0.4
0.6
µA/dB
Limiter Gain
–
–
–
55
–
dB
Input Impedance
10
–
–
1.4
–
kΩ
Output Current – Carrier Detect (High Vin)
21
–
–
0
–
µA
Output Current – Carrier Detect (Low Vin)
21
–
–
3.0
–
mA
Input Threshold Voltage – Carrier Detect
Input Voltage Referenced to VEE = –3.0 Vdc
20
–
0.9
1.2
1.4
Vdc
LIMITING AMPLIFIER SECTION
Limiter RSSI Slope (Figure 7)
CARRIER DETECT
Figure 1. Test Circuit
MC13156
1:4
(1) TR 1
RF Input
130MHz
50
Mixer
1
Local
Oscillator
Input
140.7MHz
200m Vrms
24
200
23
2
1.0 n
Mixer
Output
330
A
3
4
IF Input
VEE
VCC
22
100 n
21
Bias
5
20
50
VEE
19
A
100 n
+
1.0 n
Data Slicer
Hold
Data Output
A
17
Bias
8
VEE
1.0 µ
18
Data
Slicer
1.0 n
9
Limiter
Input
10
SMA
RSSI
Output
A
7
IF Output
Carrier
Detect
A
IF Amp
6
1.0 n
1.0 n
330
+
1.0 n
1.0 µ
VCC
VEE
16
1.0 n
V
LIM Amp
15
100 n
1.0 n
50
1.0 n
11
1.0 n
14
12
13
100 k
100 k
5.0 p
150 p
(3)
1.0 µH
NOTES: 1. TR 1 Coilcraft 1:4 impedance transformer.
2. VCC is DC Ground.
3. 1.5 µH variable shielded inductor:
Toko Part # 292SNS–T1373 or Equivalent.
MOTOROLA RF/IF DEVICE DATA
3
MC13156
DOCUMENT CONTAINS SCANNED IMAGES WHICH
COULD NOT BE PROCESSED FOR PDF FILES. FOR
COMPLETE DOCUMENT WITH IMAGES PLEASE
ORDER FROM MFAX OR THE LITERATURE
DISTRIBUTION CENTER
4
MOTOROLA RF/IF DEVICE DATA
MC13156
Figure 2. Total Drain Current versus Supply
Voltage and Temperature
Figure 3. Drain Currents versus Supply Voltage
4.0
I 19 , I 22 DRAIN CURRENTS (mA)
55°C
5.5
25°C
5.0
–10°C
4.5
–40°C
4.0
3.5
1.0
MIXER GAIN (dB)
TA = 25°C
TA = 85°C
6.0
2.0
3.0
4.0
5.0
6.0
3.2
2.8
2.4
I19
2.0
2.0
3.0
4.0
5.0
6.0
7.0
VEE, SUPPLY VOLTAGE (–Vdc)
Figure 4. Mixer Gain versus Input Signal Level
Figure 5. IF Amplifier Gain versus Input
Signal Level and Ambient Temperature
25.0
40
22.5
38
20.0
TA = 25°C
17.5
15.0
10.0
–90
I22
VEE, SUPPLY VOLTAGE (–Vdc)
12.5
36
34
32
–80
–70
–60
–50
–40
–30
–20
VEE = –5.0 Vdc
f = 10.7 MHz
26
–65
–10
85°C
55°C
25°C
–10°C
–40°C
30
28
–60
–55
–50
–45
–40
–35
–30
Pin, IF INPUT SIGNAL LEVEL (dBm)
Figure 6. IF Amplifier RSSI Output Current versus
Input Signal Level and Ambient Temperature
Figure 7. Limiter Amplifier RSSI Output Current
versus Input Signal Level and Temperature
17.5
TA = 25° to 85°C
VEE = –5.0 Vdc
f = 10.7 MHz
–10°C
–40°C
15.0
12.5
10.0
7.5
5.0
2.5
0
–50
–40
–30
–20
–10
Pin, IF INPUT SIGNAL LEVEL (dBm)
MOTOROLA RF/IF DEVICE DATA
0
10
LIMITER AMPLIFIER RSSI OUTPUT CURRENT (µ A)
Pin, RF INPUT SIGNAL LEVEL (dBm)
20.0
IF AMPLIFIER RSSI CURRENT (µ A)
3.6
1.6
1.0
7.0
IF AMPLIFIER GAIN (dB)
TOTAL DRAIN CURRENT, I TOTAL (mA)
6.5
30
25
TA = 25° to 85°C
VEE = – 5.0 Vdc
f = 10.7 MHz
–10°C
–40°C
20
15
10
5.0
0
–70
–60
–50
–40
–30
–20
–10
0
10
Pin, INPUT SIGNAL LEVEL (dBm)
5
6
19
10
12
11
9
22
23
24
4
Linear Amplifier
1
RFin1
1.0 k
2
RFin2
1.0 k
Mixer
7 1.4 k
Quadrature Detector
14
Demod
32 k
6
IFdec2
32 k
IFdec1
Mix IFin
5
3 Output
Quad coil
5.0 p
13
330
IF Amplifier
16 k
8
400 µ
290
DS in
15
RSSI
20 Out
Data Slicer
IFout
RSSI
64 k
64 k
Carrier
Detect
Output
21
64 k
28 µ
18
16
17
Carrier Detect
DSHold
DSGnd
DS
Output
Figure 8.
VEE2
LIM in
LIMdec2
LIMdec1
V CC2
VEE1
Oemitter
LO base
VCC1
Local Oscillator
Figure 8. MC13156DW Internal Circuit Schematic
MC13156
MOTOROLA RF/IF DEVICE DATA
MC13156
CIRCUIT DESCRIPTION
General
The MC13156 is a low power single conversion wideband
FM receiver incorporating a split IF. This device is designated
for use as the backend in digital FM systems such as CT–2
and wideband data links with data rates up to 500 kbaud. It
contains a mixer, oscillator, signal strength meter drive, IF
amplifier, limiting IF, quadrature detector and a data slicer
with a hold function (refer to Figure 8, Simplified Internal
Circuit Schematic).
Current Regulation
Temperature compensating voltage independent current
regulators are used throughout.
Mixer
The mixer is a double–balanced four quadrant multiplier
and is designed to work up to 500 MHz. It can be used in
differential or in single–ended mode by connecting the other
input to the positive supply rail.
Figure 4 shows the mixer gain and saturated output
response as a function of input signal drive. The circuit used
to measure this is shown in Figure 1. The linear gain of the
mixer is approximately 22 dB. Figure 9 shows the mixer gain
versus the IF output frequency with the local oscillator of
150 MHz at 100 mVrms LO drive level. The RF frequency is
swept. The sensitivity of the IF output of the mixer is shown in
Figure 10 for an RF input drive of 10 mVrms at 140 MHz and
IF at 10 MHz.
The single–ended parallel equivalent input impedance of
the mixer is Rp ~ 1.0 kΩ and Cp ~ 4.0 pF (see Table 1 for
details). The buffered output of the mixer is internally loaded
resulting in an output impedance of 330 Ω.
Local Oscillator
The on–chip transistor operates with crystal and LC
resonant elements up to 220 MHz. Series resonant, overtone
crystals are used to achieve excellent local oscillator stability.
3rd overtone crystals are used through about 65 to 70 MHz.
Operation from 70 MHz up to 180 MHz is feasible using the
on–chip transistor with a 5th or 7th overtone crystal. To
enhance operation using an overtone crystal, the internal
transistor’s bias is increased by adding an external resistor
from Pin 23 to VEE. –10 dBm of local oscillator drive is
needed to adequately drive the mixer (Figure 10).
The oscillator configurations specified above, and two
others using an external transistor, are described in the
application section:
1) A 133 MHz oscillator multiplier using a 3rd overtone
1) crystal, and
2) A 307.8 to 309.3 MHz manually tuned, varactor controlled
2) local oscillator.
RSSI
The Received Signal Strength Indicator (RSSI) output is a
current proportional to the log of the received signal
MOTOROLA RF/IF DEVICE DATA
amplitude. The RSSI current output is derived by summing
the currents from the IF and limiting amplifier stages. An
external resistor at Pin 20 sets the voltage range or swing of
the RSSI output voltage. Linearity of the RSSI is optimized by
using external ceramic or crystal bandpass filters which have
an insertion loss of 8.0 dB. The RSSI circuit is designed to
provide 70+ dB of dynamic range with temperature
compensation (see Figures 6 and 7 which show RSSI
responses of the IF and Limiter amplifiers). Variation in the
RSSI output current with supply voltage is small (see
Figure 11).
Carrier Detect
When the meter current flowing through the meter load
resistance reaches 1.2 Vdc above ground, the comparator
flips, causing the carrier detect output to go high. Hysteresis
can be accomplished by adding a very large resistor for
positive feedback between the output and the input of the
comparator.
IF Amplifier
The first IF amplifier section is composed of three
differential stages with the second and third stages
contributing to the RSSI. This section has internal dc
feedback and external input decoupling for improved
symmetry and stability. The total gain of the IF amplifier block
is approximately 39 dB at 10.7 MHz. Figure 5 shows the gain
and saturated output response of the IF amplifier over
temperature, while Figure 12 shows the IF amplifier gain as a
function of the IF frequency.
The fixed internal input impedance is 1.4 kΩ. It is designed
for applications where a 455 kHz ceramic filter is used and no
external output matching is necessary since the filter requires
a 1.4 kΩ source and load impedance.
For 10.7 MHz ceramic filter applications, an external
430 Ω resistor must be added in parallel to provide the
equivalent load impedance of 330 Ω that is required by the
filter; however, no external matching is necessary at the input
since the mixer output matches the 330 Ω source impedance
of the filter. For 455 kHz applications, an external 1.1 kΩ
resistor must be added in series with the mixer output to
obtain the required matching impedance of 1.4 kΩ of the filter
input resistance. Overall RSSI linearity is dependent on
having total midband attenuation of 12 dB (6.0 dB insertion
loss plus 6.0 dB impedance matching loss) for the filter. The
output of the IF amplifier is buffered and the impedance is
290 Ω.
Limiter
The limiter section is similar to the IF amplifier section
except that four stages are used with the last three
contributing to the RSSI. The fixed internal input impedance
is 1.4 kΩ. The total gain of the limiting amplifier section is
approximately 55 dB. This IF limiting amplifier section
internally drives the quadrature detector section.
7
MC13156
Figure 10. Mixer IF Output Level versus
Local Oscillator Input Level
Figure 9. Mixer Gain versus IF Frequency
–5.0
MIXER IF OUTPUT LEVEL (dBm)
20
MIXER GAIN (dB)
15
VEE = –3.0 Vdc
Vin = 1.0 mVrms (–47 dBm)
RO = 330 Ω
Rin = 50 Ω
BW(3.0 dB) = 21.7 MHz
fIF = fLO – fRF
fLO = 150 MHz
VLO = 100 mVrms
10
5.0
0
–5.0
0.1
1.0
10
VEE = –3.0 Vdc
TA = 25°C
–10
–15
–20
–25
–30
fRF = 140 MHz; fLO = 150 MHz
RF Input Level = –27 dBm
(10 mVrms)
Rin = 50 Ω; RO = 330 Ω
–35
–40
–45
–50
100
–40
–30
fIF, IF FREQUENCY (MHz)
Figure 11. RSSI Output Current versus
Supply Voltage and RF Input Signal Level
15
10
TA = 25°C
50
IF AMPLIFIER GAIN (dB)
I 20 , RSSI OUTPUT CURRENT ( µ A)
Vin =
–20 dBm
20
0
Figure 12. IF Amplifier Gain versus IF Frequency
30
25
–10
60
40
35
–20
LO DRIVE (dBm)
–40 dBm
–60 dBm
–80 dBm
10
–100 dBm
2.0
30
Vin = 100 µV
Rin = 50 Ω
RO = 330 Ω
BW(3.0 dB) = 26.8 MHz
TA = 25°C
20
10
5.0
0
1.0
40
3.0
4.0
5.0
6.0
0
0.1
7.0
1.0
VEE, SUPPLY VOLTAGE (–Vdc)
10
100
f, FREQUENCY (MHz)
V 14 , RECOVERED AUDIO OUTPUT (mVrms)
Figure 13. Recovered Audio Output Voltage
versus Supply Voltage
400
300
200
fmod = 1.0 kHz
fdev = ±75 kHz
fRF = 140 MHz
RF Input Level = 1.0 mVrms
TA = 25°C
100
0
1.0
2.0
3.0
4.0
5.0
6.0
7.0
VEE, SUPPLY VOLTAGE (–Vdc)
8
MOTOROLA RF/IF DEVICE DATA
MC13156
Quadrature Detector
The quadrature detector is a doubly balanced four
quadrant multiplier with an internal 5.0 pF quadrature
capacitor to couple the IF signal to the external parallel RLC
resonant circuit that provides the 90 degree phase shift and
drives the quadrature detector. A single pin (Pin 13) provides
for the external LC parallel resonant network and the internal
connection to the quadrature detector.
The bandwidth of the detector allows for recovery of
relatively high data rate modulation. The recovered signal is
converted from differential to single ended through a
push–pull NPN/PNP output stage. Variation in recovered
audio output voltage with supply voltage is very small (see
Figure 13). The output drive capability is approximately
±9.0 µA for a frequency deviation of ±75 kHz and 1.0 kHz
modulating frequency (see Application Circuit).
Data Slicer
The data slicer input (Pin 15) is self centering around 1.1 V
with clamping occurring at 1.1 ± 0.5 Vbe Vdc. It is designed to
square up the data signal. Figure 14 shows a detailed
schematic of the data slicer.
The Voltage Regulator sets up 1.1 Vdc on the base of
Q12, the Differential Input Amplifier. There is a potential of
1.0 Vbe on the base–collector of transistor diode Q11 and
2.0 Vbe on the base–collector of Q10. This sets up a 1.5 Vbe
(~ 1.1 Vdc) on the node between the 36 kΩ resistors which is
connected to the base of Q12. The differential output of the
data slicer Q12 and Q13 is converted to a single–ended
output by the Driver Circuit. Additional circuitry, not shown in
Figure 14, tends to keep the data slicer input centered at
1.1 Vdc as input signal levels vary.
The Input Diode Clamp Circuit provides the clamping at
1.0 Vbe (0.75 Vdc) and 2.0 Vbe (1.45 Vdc). Transistor diodes
Q7 and Q8 are on, thus, providing a 2.0 Vbe potential at the
base of Q1. Also, the voltage regulator circuit provides a
potential of 2.0 Vbe on the base of Q3 and 1.0 Vbe on the
emitter of Q3 and Q2. When the data slicer input (Pin 15) is
MOTOROLA RF/IF DEVICE DATA
pulled up, Q1 turns off; Q2 turns on, thereby clamping the
input at 2.0 Vbe. On the other hand, when Pin 15 is pulled
down, Q1 turns on; Q2 turns off, thereby clamping the input at
1.0 Vbe.
The recovered data signal from the quadrature detector is
ac coupled to the data slicer via an input coupling capacitor.
The size of this capacitor and the nature of the data signal
determine how faithfully the data slicer shapes up the
recovered signal. The time constant is short for large peak to
peak voltage swings or when there is a change in dc level at
the detector output. For small signal or for continuous bits of
the same polarity which drift close to the threshold voltage,
the time constant is longer. When centered there is no input
current allowed, which is to say, that the input looks high in
impedance.
Another unique feature of the data slicer is that it responds
to various logic levels applied to the Data Slicer Hold Control
pin (Pin 18). Figure 15 illustrates how the input and output
currents under “no hold” condition relate to the input voltage.
Figure 16 shows how the input current and input voltage
relate for both the “no hold” and “hold” condition.
The hold control (Pin18) does three separate tasks:
1) With Pin 18 at 1.0 Vbe or greater, the output is shut off
(sets high). Q19 turns on which shunts the base drive
from Q20, thereby turning the output off.
2) With Pin 18 at 2.0 Vbe or greater, internal clamping diodes
are open circuited and the comparator input is shut off and
effectively open circuited. This is accomplished by turning
off the current source to emitters of the input differential
amplifier, thus, the input differential amplifier is shut off.
3) When the input is shut off, it allows the input capacitor to
hold its charge during transmit to improve recovery at the
beginning of the next receive period. When it is turned on,
it allows for very fast charging of the input capacitor for
quick recovery of new tuning or data average. The above
features are very desirable in a TDD digital FM system.
9
MC13156
Figure 14. Data Slicer Circuit
9
15
VCC
DS In
8.0 k
8.0 k
Data Out
17
Q15
Q14
Q10
Q3
36 k
Q20
Q1
Q12
Q5
Q13
36 k
Q7
Q2
Q8
16
DS Gnd
32 k
Q4
Q18
Q6
Q11
Q9
Q19
Q17
Q16
64 k
16 k
VEE
16 k
64 k
64 k
19
Input Diode
Clamp Circuit
(Q1 to Q9)
Voltage
Regulator
(Q10, Q11)
Differential
Input Amplifier
(Q12, Q13)
Figure 15. Data Slicer Input/Output Currents
versus Input Voltage
0.1
0.5
–0.1
–0.5
–0.3
–0.5
0.6
Input Current
(I15)
0.8
1.0
VEE = –3.0 Vdc
V18 = 0 Vdc
(No Hold)
1.2
1.4
V15, INPUT VOLTAGE (Vdc)
10
1.6
–1.5
VEE = –3.0 Vdc
I 15 , INPUT CURRENT ( µA)
1.5
Output Current
(I17)
150
I 17 , OUTPUT CURRENT (mA)
I 15 , INPUT CURRENT (mA)
0.3
18
DS Hold
Figure 16. Data Slicer Input Current
versus Input Voltage
2.5
0.5
Driver and
Output Circuit
(Q14, Q20)
100
50
0
–50
No Hold
Hold
–2.5
1.8
Hold
V18 ≥ 1O
No Hold
V18 = 0 Vdc
–100
–1.0
–0.5
0
0.5
1.0
1.5
2.0
2.5
3.0
V15, INPUT VOLTAGE (Vdc)
MOTOROLA RF/IF DEVICE DATA
MC13156
Figure 17. MC13156DW Application Circuit
+
1.0 µ
(6)
0.146 µ
MMBR5179
15 k
100 p
MC13156
7.5 p
144.455 MHz
RF Input
68 p
50 p
Mixer
1
(1)
0.1 µ
SMA
(5) 0.82 µ
24
5.6 k
470
43 p
133.755 MHz
Osc/Tripler
23
2
10 n
(4) 3rd O.T.
XTAL
1.0 k
10 n
3
(2) 10.7 MHz
Ceramic
Filter
4
22
VEE
VCC
21
Bias
100 k
5
10 n
47 k
IF Amp
6
VEE
430
19
10 n
Data Slicer
Hold
18
7
Data
Slicer
10 k
Bias
8
VCC
RSSI
Output
20
10 n
10 n
Carrier
Detect
17
Data
Output
(2) 10.7 MHz
Ceramic
Filter
9
10
VCC
VEE
LIM Amp
16
100 n
15
180 p
10 n
100 k
11
14
100 k
430
10 n
13
12
5.0 p
150 p
+
10 k
(3)
1.5 µ
VCC
1.0 µ
NOTES: 1. 0.1 µH Variable Shielded Inductor: Coilcraft part # M1283–A or equivalent.
2. 10.7 MHz Ceramic Filter: Toko part # SK107M5–A0–10X or Murata Erie part # SFE10.7MHY–A.
3. 1.5 µH Variable Shielded Inductor: Toko part # 292SNS–T1373.
4. 3rd Overtone, Series Resonant, 25 PPM Crystal at 44.585 MHz.
5. 0.814 µH Variable Shielded Inductor: Coilcraft part # 143–18J12S.
6. 0.146 µH Variable Inductor: Coilcraft part # 146–04J08.
MOTOROLA RF/IF DEVICE DATA
11
MC13156
APPLICATIONS INFORMATION
Component Selection
The evaluation PC board is designed to accommodate
specific components, while also being versatile enough to
use components from various manufacturers and coil types.
Figures NO TAG and NO TAG show the placement for the
components specified in the application circuit (Figure 17).
The applications circuit schematic specifies particular
components that were used to achieve the results shown in
the typical curves and tables but equivalent components
should give similar results.
Input Matching Networks/Components
The input matching circuit shown in the application circuit
schematic is passive high pass network which offers effective
image rejection when the local oscillator is below the RF input
frequency. Silver mica capacitors are used for their high Q
and tight tolerance. The PC board is not dedicated to any
particular input matching network topology; space is provided
for the designer to breadboard as desired.
Alternate matching networks using 4:1 surface mount
transformers or BALUNS provide satisfactory performance.
The 12 dB SINAD sensitivity using the above matching
networks is typically –100 dBm for fmod = 1.0 kHz and
fdev = ±75 kHz at fIN = 144.45 MHz and fOSC = 133.75 MHz
(see Figure 23).
It is desirable to use a SAW filter before the mixer to
provide additional selectivity and adjacent channel rejection
and improved sensitivity. The SAW filter should be designed
to interface with the mixer input impedance of approximately
1.0 kΩ. Table 1 displays the series equivalent single–ended
mixer input impedance.
Local Oscillators
VHF Applications – The local oscillator circuit shown in the
application schematic utilizes a third overtone crystal and an
RF transistor. Selecting a transistor having good phase noise
performance is important; a mandatory criteria is for the
device to have good linearity of beta over several decades of
collector current. In other words, if the low current beta is
suppressed, it will not offer good 1/f noise performance. A
third overtone series resonant crystal having at least 25 ppm
tolerance over the operating temperature is recommended.
The local oscillator is an impedance inversion third overtone
Colpitts network and harmonic generator. In this circuit a 560
to 1.0 kΩ resistor shunts the crystal to ensure that it operates
in its overtone mode; thus, a blocking capacitor is needed to
eliminate the dc path to ground. The resulting parallel LC
network should “free–run” near the crystal frequency if a
short to ground is placed across the crystal. To provide
sufficient output loading at the collector, a high Q variable
inductor is used that is tuned to self resonate at the 3rd
harmonic of the overtone crystal frequency.
The on–chip grounded collector transistor may be used for
HF and VHF local oscillator with higher order overtone
crystals. Figure 18 shows a 5th overtone oscillator at
93.3 MHz and Figure 19 shows a 7th overtone oscillator at
148.3 MHz. Both circuits use a Butler overtone oscillator
configuration. The amplifier is an emitter follower. The crystal
is driven from the emitter and is coupled to the high
impedance base through a capacitive tap network. Operation
at the desired overtone frequency is ensured by the parallel
resonant circuit formed by the variable inductor and the tap
capacitors and parasitic capacitances of the on–chip
transistor and PC board. The variable inductor specified in
the schematic could be replaced with a high tolerance, high Q
ceramic or air wound surface mount component if the other
components have good tolerances. A variable inductor
provides an adjustment for gain and frequency of the
resonant tank ensuring lock up and startup of the crystal
oscillator. The overtone crystal is chosen with ESR of
typically 80 Ω and 120 Ω maximum; if the resistive loss in the
crystal is too high, the performance of the oscillator may be
impacted by lower gain margins.
Table 1. Mixer Input Impedance Data
(Single–ended configuration, VCC = 3.0 Vdc, local oscillator drive = 100 mVrms)
12
Frequency
(MHz)
Series Equivalent
Complex Impedance
(R + jX)
(Ω)
Parallel
Resistance
Rp
(Ω)
Parallel
Capacitance
Cp
(pF)
90
190 – j380
950
4.7
100
160 – j360
970
4.4
110
130 – j340
1020
4.2
120
110 – j320
1040
4.2
130
97 – j300
1030
4.0
140
82 – j280
1040
4.0
150
71 – j270
1100
4.0
160
59 – j260
1200
3.9
170
52 – j240
1160
3.9
180
44 – j230
1250
3.8
190
38 – j220
1300
3.8
MOTOROLA RF/IF DEVICE DATA
MC13156
A series LC network to ground (which is VCC) is comprised
of the inductance of the base lead of the on–chip transistor
and PC board traces and tap capacitors. Parasitic
oscillations often occur in the 200 to 800 MHz range. A small
resistor is placed in series with the base (Pin 24) to cancel the
negative resistance associated with this undesired mode of
oscillation. Since the base input impedance is so large a
small resistor in the range of 27 to 68 Ω has very little effect
on the desired Butler mode of oscillation.
The crystal parallel capacitance, Co, provides a feedback
path that is low enough in reactance at frequencies of 5th
overtone or higher to cause trouble. Co has little effect near
resonance because of the low impedance of the crystal
motional arm (Rm–Lm–Cm). As the tunable inductor which
forms the resonant tank with the tap capacitors is tuned off
the crystal resonant frequency, it may be difficult to tell if the
oscillation is under crystal control. Frequency jumps may
occur as the inductor is tuned. In order to eliminate this
behavior an inductor (Lo) is placed in parallel with the crystal.
Lo is chosen to resonant with the crystal parallel capacitance
(Co) at the desired operation frequency. The inductor
provides a feedback path at frequencies well below
resonance; however, the parallel tank network of the tap
capacitors and tunable inductor prevent oscillation at these
frequencies.
UHF Application
Figure 20 shows a 318.5 to 320 MHz receiver which drives
the mixer with an external varactor controlled (307.8 to
309.3 MHz) LC oscillator using an MPS901 (RF low power
transistor in a TO–92 plastic package; also MMBR901 is
available in a SOT–23 surface mount package). With the
50 kΩ 10 turn potentiometer this oscillator is tunable over a
range of approximately 1.5 MHz. The MMBV909L is a low
voltage varactor suitable for UHF applications; it is a dual
back–to–back varactor in a SOT–23 package. The input
matching networ k us es a 1:4 impedanc e matching
transformer (Recommended sources are Mini–Circuits and
Coilcraft).
Using the same IF ceramic filters and quadrature detector
circuit as specified in the applications circuit in Figure 17, the
12 dB SINAD performance is –95 dBm for a fmod = 1.0 kHz
sinusoidal waveform and fdev ±40 kHz.
This circuit is breadboarded using the evaluation PC board
shown in Figures NO TAG and NO TAG. The RF ground is
VCC and path lengths are minimized. High quality surface
mount components were used except where specified. The
absolute values of the components used will vary with layout
placement and component parasitics.
RSSI Response
Figure 24 shows the full RSSI response in the application
circuit. The 10.7 MHz, 110 kHz wide bandpass ceramic filters
(recommended sources are TOKO part # SK107M5–AO–10X
or Murata Erie SFE10.7MHY–A) provide the correct
bandpass insertion loss to linearize the curve between the
limiter and IF portions of RSSI. Figure 23 shows that limiting
occurs at an input of –100 dBm. As shown in Figure 24, the
RSSI output linear from –100 dBm to –30 dBm.
The RSSI rise and fall times for various RF input signal
levels and R20 values are measured at Pin 20 without 10 nF
filter capacitor. A 10 kHz square wave pulses the RF input
signal on and off. Figure 25 shows that the rise and fall times
are short enough to recover greater than 10 kHz ASK data;
with a wider IF bandpass filters data rates up to 50 kHz may
be achieved. The circuit used is the application circuit in
Figure 17 with no RSSI output filter capacitor.
Figure 18. MC13156DW Application Circuit
fRF = 104 MHz; fLO = 93.30 MHz
5th Overtone Crystal Oscillator
(4)
0.135 µH
33
(2)
10 p
104 MHz
SMA
Mixer
120 p
RF Input
27 p
1
24
2
23
(1)
0.1 µ
3.0 p
1.0 µH
(3)
10 n
4.7 k
3
+
1.0 µ
VEE
30 p
5th OT
XTAL
22
10 n
To Filter
VCC
NOTES: 1. 0.1 µH Variable Shielded Inductor: Coilcraft part # M1283–A or equivalent.
2. Capacitors are Silver Mica.
3. 5th Overtone, Series Resonant, 25 PPM Crystal at 93.300 MHz.
4. 0.135 µH Variable Shielded Inductor: Coilcraft part # 146–05J08S or equivalent.
MOTOROLA RF/IF DEVICE DATA
13
MC13156
Figure 19. MC13156DW Application Circuit
fRF = 159 MHz; fLO = 148.30 MHz
7th Overtone Crystal Oscillator
(4)
76 nH
+
1.0 µ
33
(2)
5.0 p
159 MHz
RF Input
(1)
0.08 µH
SMA
27 p
Mixer
50 p
1
24
2
23
3
22
0.22 µH
47 p
10 n
4.7 k
470
VEE
(3)
7th OT
XTAL
10 n
VCC
To IF Filter
NOTES: 1. 0.08 µH Variable Shielded Inductor: Toko part # 292SNS–T1365Z or equivalent.
2. Capacitors are Silver Mica.
3. 7th Overtone, Series Resonant, 25 PPM Crystal at 148.300 MHz.
4. 76 nH Variable Shielded Inductor: Coilcraft part # 150–03J08S or equivalent.
Figure 20. MC13156DW Varactor Controlled LC Oscillator
4.7 k
MPS901
(1)
1:4 Transformer
6.8 p
47 k
VVCO
+
1.0 µ
(6)
318.5 to
320 MHz
RF Input
(2)
50 k
1.0 M
0.1 µ
Mixer
1
24
2
23
24 p
20 p
(4)
MMBV909L
SMA
12 k
24 p
(3)
18.5 nH
1.8 k
3
VEE
22
1.0 n
307.8–309.3 MHz
LC Varactor
Controlled Oscillator
VCC = 3.3 Vdc (Reg)
NOTES: 1. 1:4 Impedance Transformer: Mini–Circuits.
2. 50 k Potentiometer, 10 turns.
3. Spring Coil; Coilcraft A05T.
4. Dual Varactor in SOT–23 Package.
5. All other components are surface mount components.
6. Ferrite beads through loop of 24 AWG wire.
14
MOTOROLA RF/IF DEVICE DATA
MC13156
The 12 dB SINAD performance is –109 dBm for a fmod =
1.0 kHz and a fdev = ±4.0 kHz. The RSSI dynamic range is
approximately 80 dB of linear range (see Figure 22).
45 MHz Narrowband Receiver
The above application examples utilize a 10.7 MHz IF. In
this section a narrowband receiver with a 455 kHz IF will be
described. Figure 21 shows a full schematic of a 45 MHz
receiver that uses a 3rd overtone crystal with the on–chip
oscillator transistor. The oscillator configuration is similar to
the one used in Figure 17; it is called an impedance inversion
Colpitts. A 44.545 MHz 3rd overtone, series resonant crystal
is used to achieve an IF frequency at 455 kHz. The ceramic
IF filters selected are Murata Erie part # SFG455A3. 1.2 kΩ
chip resistors are used in series with the filters to achieve the
terminating resistance of 1.4 kΩ to the filter. The IF
decoupling is very important; 0.1 µF chip capacitors are used
at Pins 6, 7, 11 and 12. The quadrature detector tank circuit
uses a 455 kHz quadrature tank from Toko.
Receiver Design Considerations
The curves of signal levels at various portions of the
application receiver with respect to RF input level are shown
in Figure 26. This information helps determine the network
topology and gain blocks required ahead of the MC13156 to
achieve the desired sensitivity and dynamic range of the
receiver system. In the application circuit the input third order
intercept (IP3) performance of the system is approximately
–25 dBm (see Figure 27).
Figure 21. MC13156DW Application Circuit at 45 MHz
1.8 µH
+
1.0 µ
(6)
33 p
45 Hz
RF Input
SMA
(1)
0.33 µH
10 n
Mixer
1
24
56 p
10 n
470 k
23
2
39 p
1.2 k
3
(2) 455 kHz
Ceramic
Filter
4
VCC
10 k
10 n
21
Bias
RSSI
Output
10 n
47 k
IF Amp
6
VEE
0.1 µ
19
10 n
Data Slicer
Hold
18
7
Data
Slicer
1.2 k
10 k
17
Bias
8
Carrier
Detect
100 k
20
0.1 µ
(4) 3rd OT
XTAL
44.545
MHz
22
VEE
5
VCC
(5) 0.416 µH
180 p
Data
Output
(2) 455 kHz
Ceramic
Filter
9
VCC
VEE
100 n
15
10
0.1 µ
16
LIM Amp
100 k
14
11
1.0 n
0.1 µ
Audio To
C–Message
Filter and
Amp.
100 k
13
12
5.0 p
27 k
NOTES: 1. 0.33 µH Variable Shielded Inductor: Coilcraft part # 7M3–331 or equivalent.
2. 455 kHz Ceramic Filter: Murata Erie part # SFG455A3.
3. 455 kHz Quadrature Tank: Toko part # 7MC8128Z.
4. 3rd Overtone, Series Resonant, 25 PPM Crystal at 44.540 MHz.
5. 0.416 µH Variable Shielded Inductor: Coilcraft part # 143–10J12S.
6. 1.8 µH Molded Inductor.
MOTOROLA RF/IF DEVICE DATA
+
180 p
(3)
680 µH
VCC = 2.0 to 5.0 Vdc
1.0 µ
15
MC13156
Figure 23. S + N/N versus RF Input Signal Level
1.8
10
1.6
0
1.4
fRF = 45.00 MHz
VCC = 2.0 Vdc
12 dB SINAD @ –109 dBm
(0.8 µVrms)
(See Figure 21)
1.2
1.0
S + N, N (dB)
RSSI OUTPUT VOLTAGE (Vdc)
Figure 22. RSSI Output Voltage
versus Input Signal Level
0.8
S+N
VCC = 5.0 Vdc
fdev = ±75 kHz
fmod = 1.0 kHz
fin = 144.45 MHz
(See Figure 17)
–10
–20
–30
0.6
–40
0.4
–120
–50
–110 –100
N
–100
–80
–60
–40
–20
0
20
t r , t f , RSSI RISE AND FALL TIMES (µ s)
RSSI OUTPUT VOLTAGE (Vdc)
0.8
VCC = 5.0 Vdc
fc = 144.455 MHz
fLO = 133.755 MHz
Low Loss 10.7 MHz
Ceramic Filter
(See Figure 17)
0.6
0.4
35
30
25
20
15
10
5.0
0
–100
–80
–60
–40
–20
0
0
SIGNAL INPUT LEVEL (dBm)
Figure 26. Signal Levels versus
RF Input Signal Level
–40
–30
–20
É
É
É
É
É
É
ÇÇ
É
ÇÇ
É
ÇÇ
É
É
É
É
É
É
Ç
É
Ç
É
Ç
É
–20
É
É
É
É
É
ÇÇ
É
ÇÇ
É
ÇÇ
É
–40
tr
tf
tr
tf
tr
tf
@ 22 k
@ 22 k
@ 47 k
@ 47 k
@ 100 k
@ 100 k
É
ÇÇ
É
ÇÇ
É
ÇÇ
É
ÇÇ
É
ÇÇ
É
É ÇÇ
–60
–80
RF INPUT SIGNAL LEVEL (dBm)
10
LO Level = –2.0 dBm
(See Figure 17)
MIXER IF OUTPUT LEVEL (dBm)
IF Output
Limiter Input
–20
POWER (dBm)
–50
Figure 27. 1.0 dB Compression Pt. and Input
Third Order Intercept Pt. versus Input Power
0
–30
–40
–50
–60
–90
–80
–70
–60
–50
RF INPUT SIGNAL LEVEL (dBm)
16
–60
Figure 25. RSSI Output Rise and Fall Times
versus RF Input Signal Level
1.0
–70
–100
–70
Figure 24. RSSI Output Voltage
versus Input Signal Level
1.2
–10
–80
RF INPUT SIGNAL (dBm)
1.4
0.2
–120
–90
SIGNAL INPUT LEVEL (dBm)
–40
–30
0
–10
–20
VCC = 5.0 Vdc
fRF1 = 144.4 MHz
fRF2 = 144.5 MHz
fLO = 133.75 MHz
PLO = –2.0 dBm
(See Figure 17)
1.0 dB Comp. Pt.
= –37 dBm
IP3 = –25 dBm
–30
–40
–50
–60
–70
–100
–80
–60
–40
–20
0
RF INPUT POWER (dBm)
MOTOROLA RF/IF DEVICE DATA
MC13156
BER TESTING AND PERFORMANCE
• Data rate = 100 kbps
• Filter cutoff frequency set to 39% of the data rate or 39 kHz.
• Filter type is a 5 pole equal–ripple with 0.5° phase error.
• VCC = 4.0 Vdc
• Frequency deviation = ±32 kHz.
MOTOROLA RF/IF DEVICE DATA
10 –1
BER, BIT ERROR RATE
Description
The test setup shown in Figure 29 is configured so that the
function generator supplies a 100 kHz clock source to the bit
error rate tester. This device generates and receives a
repeating data pattern and drives a 5 pole baseband data
filter. The filter effectively reduces harmonic content of the
baseband data which is used to modulate the RF generator
which is running at 144.45 MHz. Following processing of the
signal by the receiver (MC13156), the recovered baseband
sinewave (data) is AC coupled to the data slicer. The data
slicer is essentially an auto–threshold comparator which
tracks the zero crossing of the incoming sinewave and
provides logic level data at its ouput. Data errors associated
with the recovered data are collected by the bit error rate
receiver and displayed.
Bit error rate versus RF signal input level and IF filter
bandwidth are shown in Figure 28. The bit error rate data was
taken under the following test conditions:
10 –3
Figure 28. Bit Error Rate versus RF
Input Signal Level and IF Bandpass Filter
VCC = 4.0 Vdc
Data Pattern = 2E09 Prbs NRZ
Baseband Filter fc = 50 kHz
fdev = ±32 kHz
IF Filter BW
110 kHz
IF Filter BW
230 kHz
10 –5
10 –7
–90
–85
–80
–75
–70
RF INPUT SIGNAL LEVEL (dBm)
Evaluation PC Board
The evaluation PCB is very versatile and is intended to be
used across the entire useful frequency range of this device.
The center section of the board provides an area for
attaching all SMT components to the circuit side and radial
leaded components to the component ground side (see
Figures NO TAG and NO TAG). Additionally, the peripheral
area surrounding the RF core provides pads to add
supporting and interface circuitry as a particular application
dictates.
17
MC13156
Figure 29. Bit Error Rate Test Setup
Function Generator
Bit Error Rate Tester
RF Generator
Wavetek Model No. 164
HP3780A or Equivalent
HP8640B
Clock
Out
Gen
Clock
Input
Rcr
Clock
Input
Rcr
Data
Input
Generator
Input
Modulation
Input
RF
Output
5 Pole
Bandpass
Filter
Data Slicer
Output
Mixer
Input
MC13156
UUT
18
MOTOROLA RF/IF DEVICE DATA
MC13156
OUTLINE DIMENSIONS
FB SUFFIX
PLASTIC QFP PACKAGE
CASE 873–01
ISSUE A
L
0.20 (0.008)
D
S
H A–B
V
M
B
0.20 (0.008)
L
0.05 (0.002) A–B
–B–
–A–
M
C A–B
S
D
S
16
S
17
24
25
B
32
P
B
DETAIL A
9
1
8
–A–, –B–, –D–
–D–
DETAIL A
A
0.20 (0.008)
M
C A–B
D
S
S
0.05 (0.002) A–B
S
0.20 (0.008)
M
H A–B
S
D
F
BASE
METAL
S
M
DETAIL C
N
J
C E
–H–
–C–
SEATING
PLANE
H
M
G
DATUM
PLANE
0.01 (0.004)
D
0.20 (0.008)
M
C A–B
S
D
S
SECTION B–B
VIEW ROTATED 90 _CLOCKWISE
U
T
R
–H–
DATUM
PLANE
K
X
DETAIL C
MOTOROLA RF/IF DEVICE DATA
Q
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: MILLIMETER.
3. DATUM PLANE –H– IS LOCATED AT BOTTOM OF
LEAD AND IS COINCIDENT WITH THE LEAD WHERE
THE LEAD EXITS THE PLASTIC BODY AT THE
BOTTOM OF THE PARTING LINE.
4. DATUMS –A–, –B– AND –D– TO BE DETERMINED AT
DATUM PLANE –H–.
5. DIMENSIONS S AND V TO BE DETERMINED AT
SEATING PLANE –C–.
6. DIMENSIONS A AND B DO NOT INCLUDE MOLD
PROTRUSION. ALLOWABLE PROTRUSION IS 0.25
(0.010) PER SIDE. DIMENSIONS A AND B DO
INCLUDE MOLD MISMATCH AND ARE DETERMINED
AT DATUM PLANE –H–.
7. DIMENSION D DOES NOT INCLUDE DAMBAR
PROTRUSION. ALLOWABLE DAMBAR PROTRUSION
SHALL BE 0.08 (0.003) TOTAL IN EXCESS OF THE D
DIMENSION AT MAXIMUM MATERIAL CONDITION.
DAMBAR CANNOT BE LOCATED ON THE LOWER
RADIUS OR THE FOOT.
DIM
A
B
C
D
E
F
G
H
J
K
L
M
N
P
Q
R
S
T
U
V
X
MILLIMETERS
MIN
MAX
6.95
7.10
6.95
7.10
1.40
1.60
0.273
0.373
1.30
1.50
0.273
–––
0.80 BSC
–––
0.20
0.119
0.197
0.33
0.57
5.6 REF
6_
8_
0.119
0.135
0.40 BSC
5_
10_
0.15
0.25
8.85
9.15
0.15
0.25
5_
11_
8.85
9.15
1.00 REF
INCHES
MIN
MAX
0.274
0.280
0.274
0.280
0.055
0.063
0.010
0.015
0.051
0.059
0.010
–––
0.031 BSC
–––
0.008
0.005
0.008
0.013
0.022
0.220 REF
6_
8_
0.005
0.005
0.016 BSC
5_
10_
0.006
0.010
0.348
0.360
0.006
0.010
5_
11_
0.348
0.360
0.039 REF
19
MC13156
OUTLINE DIMENSIONS
DW SUFFIX
PLASTIC PACKAGE
CASE 751E–04
(SO–24L)
ISSUE E
–A–
24
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: MILLIMETER.
3. DIMENSIONS A AND B DO NOT INCLUDE
MOLD PROTRUSION.
4. MAXIMUM MOLD PROTRUSION 0.15 (0.006)
PER SIDE.
5. DIMENSION D DOES NOT INCLUDE DAMBAR
PROTRUSION. ALLOWABLE DAMBAR
PROTRUSION SHALL BE 0.13 (0.005) TOTAL IN
EXCESS OF D DIMENSION AT MAXIMUM
MATERIAL CONDITION.
13
–B–
12X
P
0.010 (0.25)
1
M
B
M
12
24X
D
J
0.010 (0.25)
M
T A
S
B
S
F
R
C
–T–
SEATING
PLANE
M
22X
G
K
X 45 _
DIM
A
B
C
D
F
G
J
K
M
P
R
MILLIMETERS
MIN
MAX
15.25
15.54
7.40
7.60
2.35
2.65
0.35
0.49
0.41
0.90
1.27 BSC
0.23
0.32
0.13
0.29
0_
8_
10.05
10.55
0.25
0.75
INCHES
MIN
MAX
0.601
0.612
0.292
0.299
0.093
0.104
0.014
0.019
0.016
0.035
0.050 BSC
0.009
0.013
0.005
0.011
0_
8_
0.395
0.415
0.010
0.029
Motorola reserves the right to make changes without further notice to any products herein. Motorola makes no warranty, representation or guarantee regarding
the suitability of its products for any particular purpose, nor does Motorola assume any liability arising out of the application or use of any product or circuit, and
specifically disclaims any and all liability, including without limitation consequential or incidental damages. “Typical” parameters which may be provided in Motorola
data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including “Typicals”
must be validated for each customer application by customer’s technical experts. Motorola does not convey any license under its patent rights nor the rights of
others. Motorola products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other
applications intended to support or sustain life, or for any other application in which the failure of the Motorola product could create a situation where personal injury
or death may occur. Should Buyer purchase or use Motorola products for any such unintended or unauthorized application, Buyer shall indemnify and hold Motorola
and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees
arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that
Motorola was negligent regarding the design or manufacture of the part. Motorola and
are registered trademarks of Motorola, Inc. Motorola, Inc. is an Equal
Opportunity/Affirmative Action Employer.
Mfax is a trademark of Motorola, Inc.
How to reach us:
USA / EUROPE / Locations Not Listed: Motorola Literature Distribution;
P.O. Box 5405, Denver, Colorado 80217. 1–303–675–2140 or 1–800–441–2447
JAPAN: Nippon Motorola Ltd.: SPD, Strategic Planning Office, 141,
4–32–1 Nishi–Gotanda, Shagawa–ku, Tokyo, Japan. 03–5487–8488
Customer Focus Center: 1–800–521–6274
Mfax: [email protected] – TOUCHTONE 1–602–244–6609
ASIA/PACIFIC: Motorola Semiconductors H.K. Ltd.; 8B Tai Ping Industrial Park,
Motorola Fax Back System
– US & Canada ONLY 1–800–774–1848 51 Ting Kok Road, Tai Po, N.T., Hong Kong. 852–26629298
– http://sps.motorola.com/mfax/
HOME PAGE: http://motorola.com/sps/
20
◊
MC13156/D
MOTOROLA RF/IF DEVICE DATA