LANSDALE MC13175D

ML13175
ML13176
UHF FM/AM Transmitter
Legacy Device: Motorola MC13175, MC13176
The ML13175 and ML13176 are one chip FM/AM transmitter subsystems designed for AM/FM communication systems. they include a
Colpitts crystal reference oscillator, UHF oscillator, ÷ 8 (ML13175) or
÷ 32 (ML13176) prescaler and phase detector forming a versatile PLL
system. Targeted applications are in the 260 to 470MHz band and 902
to 982 MHz band covered by FCC Title 47; Part 15. Other applications include local oscillator sources in UHF and 900 MHz receivers,
UHF and 900 MHz video transmitters, RF Local Area Networks
(LAN), and high frequency clock drivers. ML13175/76 offer the following features;
• UHF Current Controlled Oscillator
• Uses Easily Available 3rd Overtone or Fundamental Crystals
for Reference
• Fewer External Parts Required
• Low Operating Supply Voltage (1.8 to 5.0 Vdc)
• Low Supply Drain Currents
• Power Output Adjustable (Up to + 10 dBm )
• Differential Output for Loop Antenna or Balun
Transformer Networks
• Power Down Feature
• ASK Modulated by Switching Output On and Off
• (ML13175) fo = 8 x fref, (ML13176) fo = 32 x fref
• Operating Temperature Range - TA = -40° to +85°C
ML13175-5P
PLASTIC PACKAGE
CASE 751B
(SO–16)
16
1
CROSS REFERENCE/ORDERING INFORMATION
PACKAGE
LANSDALE
MOTOROLA
SO 16
SO 16
MC13175D
MC13176D
Note: Lansdale lead free (Pb) product, as it
becomes available, will be identified by a part
number prefix change from ML to MLE.
PIN CONNECTIONS
Osc 1 1
Figure 1. Typical Application as 320 MHz AM Transmitter
AM Modulator
Osc
Tank
1
16
2
15
3
14
VEE
S2
(1)
(2)
VEE
0.1µ
150p
13
RFC1
5
12
6
11
7
10
8
9
1.0k
100p
(ML13176)
VCC
30p
(ML13175)
ML13175–30p
ML13176–180p
ML13175
Crystal
3rd Overtone
40.0000 MHz
NOTES: 1.
2.
2.
2.
3.
3.
3.
3.
Page 1 of 16
RFout
SMA
Z = 50Ω
f/N
Out
Gnd
NC 2
15
NC 3
14 Out 2
Osc 4 4
13 Out 1
VEE 5
12 VCC
150p
0.165µ
4
16 Imod
1.3k
0.01µ
Coilcraft
150–05J08
ML13175-5P
ML13176-5P
VCC
27k
ICont 6
11 Enable
PDout 7
10 Reg.
Gnd
Xtale 8
9 Xtalb
S1
VEE
0.1µ
0.01µ
0.82µ
1.0k
(3)
ML13176
Crystal
Fundamental
10 MHz
VCC
50 Ω coaxial balun, 1/10 wavelength at 320 MHz equals 1.5 inches.
Pins 5, 10 & 15 are ground and connected to VEE which is the component/DC ground plane
side of PCB. These pins must be decoupled to VCC; decoupling capacitors should be placed
as close as possible to the pins.
The crystal oscillator circuit may be adjusted for frequency with the variable inductor
(ML13175); recommended source is Coilcraft “slot seven” 7mm tuneable inductor , Part
#7M3–821. 1.0k resistor. Shunting the crystal prevents it from oscillating in the fundamental
mode.
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ML13175/ML13176
LANSDALE Semiconductor, Inc.
MAXIMUM RATINGS ( TA = 25 C, unless otherwise noted.)
Rating
Symbol
Value
Unit
Power Supply Voltage
VCC
7.0 (max)
Vdc
Operating Supply Voltage Range
VCC
1.8 to 5.0
Vdc
Junction Temperature
TJ
+150
C
Operating Ambient Temperature
TA
– 40 to + 85
C
Tstg
– 65 to +150
C
Storage Temperature
ELECTRICAL CHARACTERISTICS (Figure 2; V EE = – 3.0 Vdc, TA = 25 C, unless otherwise noted.)*
Pin
Symbol
Min
Supply Current (Power down: I 11 & I16 = 0)
–
IEE1
– 0.5
–
µA
Supply Current (Enable [Pin 11] to VCC thru 30 k, I16 = 0)
–
IEE2
–
–14
–18
mA
Total Supply Current (Transmit Mode)
(Imod = 2.0 mA; f o = 320 MHz)
–
IEE3
–
– 34
–39
mA
Differential Output Power (f o = 320 MHz; Vref [Pin 9]
= 500 mVp–p; f o = N x fref)
Imod = 2.0 mA (see Figure 7, 8)
Imod = 0 mA
13 & 14
Pout
Hold–in Range (± ∆fref x N)
ML13175 (see Figure 7)
ML13176 (see Figure 8)
13 & 14
Characteristic
Phase Detector Output Error Current
ML13175
ML13176
7
Oscillator Enable Time (see Figure 22b)
Typ
–
Max
Unit
dBm
2.0
–
+ 4.7
– 45
–
–
3.5
4.0
6.5
8.0
–
–
20
22
25
27
–
–
± ∆f H
MHz
µA
lerror
11 & 8
tenable
–
4.0
–
ms
16
BWAM
–
25
–
MHz
Spurious Outputs (Imod = 2.0 mA)
Spurious Outputs (Imod = 0 mA)
13 & 14
13 & 14
Pson
Psoff
–
–
– 50
– 50
–
–
dBc
Maximum Divider Input Frequency
Maximum Output Frequency
–
13 & 14
fdiv
fo
–
–
950
950
–
–
MHz
Amplitude Modulation Bandwidth (see Figure 24)
* For testing purposes, VCC is ground (see Figure 2).
Figure 2. 320 MHz Test Circuit
Imod
Osc
Tank
VEE
(1)
Coilcraft
150–03J0
8
15p
(ML13176)
16
2
15
0.1µ
0.1µ
3
0.098µ
4
14
f/N
5
0.1µ
27p
1
10k
10k
10p
(ML13175)
13
12
6
11
7
10
8
9
2.2k
ML13175–30p
ML13176–33p
VCC
51
0.01µ
51
0.01µ
RFout 1
RFout 2
(1)
30k
Ireg. enable
0.1µ
0.01µ
0.82µ
ML13175 1.0k
Crystal
3rd Overtone
40 MHz
(3)
ML13176
Crystal
Fundamental VCC
10 MHz
NOTES: 1. VCC is ground; while V EE is negative with respect to ground.
2. Pins 5, 10 and 15 are brought to the circuit side of the PCB via plated through holes.
They are connected together with a trace on the PCB and each Pin is decoupled to VCC (ground).
3. Recommended source is Coilcraft “slot seven inductor ” part number 7M3–821.
Page 2 of 16
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ML13175/ML13176
PIN FUNCTION DESCRIPTIONS
Pin
Symbol
1&4
Osc 1,
Osc 4
Internal Equivalent
Circuit
Description/External
Circuit Requirements
VCC
10k
10k
1
0sc 1
5
4
Osc 4
VEE
VEE
5
Subcon
VEE
6
ICont
Frequency Control
For VCC = 3.0 Vdc, the voltage at Pin 6 is approximately 1.55
Vdc. The oscillator is current controlled by the error current from
the phase detector. This current is amplified to drive the current
source in the oscillator section which controls the frequency of
the oscillator. Figures 9 and 10 show the ∆fosc versus ICont,
Figure 5 shows the ∆fosc versus ICont at – 40°C, + 25°C and
+ 85°C for 320 MHz. The CCO may be FM modulated as shown
in Figure 17, ML13176 320 MHz FM Transmitter. A detailed
discussion is found in the Applications Information section.
VCC
Reg
ICont
PDout
VCC
4.0k
4.0k
PDout
7
Page 3 of 16
Supply Ground (VEE)
In the PCB layout, the ground pins (also applies to Pins 10 and
15) should be connected directly to chassis ground. Decoupling
capacitors to VCC should be placed directly at
the ground returns.
VEE
6
7
CCO Inputs
The oscillator is a current controlled type. An external oscillator
coil is connected to Pins 1 and 4 which forms a parallel
resonance LC tank circuit with the internal capacitance of the
IC and with parasitic capacitance of the PC board. Three
base–emitter capacitances in series configuration form the
capacitance for the parallel tank. These are the base–emitters
at Pins 1 and 4 and the base–emitter of the differential amplifier.
The equivalent series capacitance in the differential amplifier is
varied by the modulating current from the frequency control
circuit (see Pin 6, internal circuit). A more thorough discussion
is found in the Applications Information section.
Phase Detector Output
The phase detector provides ± 30 µA to keep the CCO locked at
the desired carrier frequency. The output impedance of the
phase detector is approximately 53 kΩ. Under closed loop
conditions there is a DC voltage which is dependent upon the
free running oscillator and the reference oscillator frequencies.
The circuitry between Pins 7 and 6 should be selected for
adequate loop filtering necessary to stabilize and filter the loop
response. Low pass filtering between Pin 7 and 6 is needed so
that the corner frequency is well below the sum of the divider
and the reference oscillator frequencies, but high enough to
allow for fast response to keep the loop locked. Refer to the
Applications Information section regarding loop filtering and FM
modulation.
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ML13175/ML13176
LANSDALE Semiconductor, Inc.
PIN FUNCTION DESCRIPTIONS
Pin
Symbol
8
Xtale
9
Xtalb
Internal Equivalent
Circuit
8
Xtalb
8.0k
12k
4.0k
Xtale
Reg. Gnd
Regulator Ground
An additional ground pin is provided to enhance the stability of
the system. Decoupling to the VCC (RF ground) is essential; it
should be done at the ground return for Pin 10.
VCC
Reg
5.0pp
11
Enable
11
Enable
Subcon
8.0k
2.4k
10
Reg. Gnd
12
VCC
12
VCC
Out 1 and
Out 2
Differential Output
The output is configured differentially to easily drive a loop
antenna. By using a transformer or balun, as shown in the
application schematic, the device may then drive an unbalanced
low impedance load. Figure 6 shows how much the Output
Power and Free–Running Oscillator Frequency change with
temperature at 3.0 Vdc; Imod = 2.0 mA.
VCC
15
13
Out_Gnd
Out 1
16
Imod
15
Out_Gnd
Page 4 of 16
Device Enable
The potential at Pin 11 is approximately 1.25 Vdc. When Pin 11
is open, the transmitter is disabled in a power down mode and
draws less than 1.0 µA ICC if the MOD at Pin 16 is also open
(i.e., it has no current driving it). To enable the transmitter a
current source of 10 µA to 90 µA is provided. Figures 3 and 4
show the relationship between ICC, VCC and Ireg. enable. Note
that ICC is flat at approximately 10 mA for Ireg. enable = 5.0 to
100 µA (Imod = 0).
Supply Voltage (VCC)
The operating supply voltage range is from 1.8 Vdc to 5.0 Vdc.
In the PCB layout, the VCC trace must be kept as wide as
possible to minimize inductive reactances along the trace; it is
best to have it completely fill around the surface mount
components and traces on the circuit side of the PCB.
VCC
13 & 14
Crystal Oscillator Inputs
The internal reference oscillator is configured as a common
emitter Colpitts. It may be operated with either a fundamental
or overtone crystal depending on the carrier frequency and the
internal prescaler. Crystal oscillator circuits and specifications
of crystals are discussed in detail in the applications section.
With VCC = 3.0 Vdc, the voltage at Pin 8 is approximately 1.8
Vdc and at Pin 9 is approximately 2.3 Vdc. 500 to 1000 mVp–p
should be present at Pin 9. The Colpitts is biased at 200 µA;
additional drive may be acquired by increasing the bias to
approximately 500 µA. Use 6.2 k from Pin 8 to ground.
VCC
9
10
Description/External
Circuit Requirements
14
16
Out 2
Imod
Output Ground
This additional ground pin provides direct access for the output
ground to the circuit board VEE.
AM Modulation/Power Output Level
The DC voltage at this pin is 0.8 Vdc with the current source
active. An external resistor is chosen to provide a source
current of 1.0 to 3.0 mA, depending on the desired output power
level at a given VCC. Figure 23 shows the relationship of Power
Output to Modulation Current, Imod. At VCC = 3.0 Vdc, 3.5 dBm
power output can be acquired with about 35 mA ICC.
For FM modulation, Pin 16 is used to set the desired output
power level as described above.
For AM modulation, the modulation signal must ride on a
positive DC bias offset which sets a static (modulation off)
modulation current. External circuitry for various schemes is
further discussed in the Applications Information section.
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ML13175/ML13176
Figure 3. Supply Current
versus Supply Voltage
Figure 4. Supply Current versus
Regulator Enable Current
100
I CC , SUPPLY CURRENT (mA)
Ireg. enable = 90 µA
Imod = 0
8.0
6.0
4.0
2.0
1.0
2.0
3.0
VCC, SUPPLY VOLTAGE (Vdc)
4.0
10
1.0
0.1
0
0
VCC = 3.0 Vdc
Imod = 0
5.0
VCC = 3.0 Vdc
Imod = 2.0 mA
f = 320 MHz (ICont = 0; TA = 25 °C)
Free–Running Oscillator
0
– 40 °C
f ref , REFERENCE OSCILLATOR FREQUENCY (MHz)
– 5.0
25 °C
–10
–15
– 40
85 °C
60
– 20
0
20
40
ICont, OSCILLATOR CONTROL CURRENT (µA)
80
Figure 7. ML13175 Reference Oscillator
Frequency versus Phase Detector Current
41.0
Closed Loop Response:
VCC = 3.0 Vdc
fo = 8.0 x fref
Vref = 500 mVp–p
40.8
40.6
40.2
40.0
Imod = 2.0 mA
ICC = 36 mA
PO = 5.4 dBm
39.8
Page 5 of 16
4.0
3.0
5.5
∆fosc
PO
5.0
2.0
1.0
4.5
0
4.0
–1.0
VCC = 3.0 Vdc
Imod = 2.0 mA
f = 320 MHz (ICont = 0; TA = 25 °C)
Free–Running Oscillator
– 2.0
– 3.0
– 4.0
– 50
0
50
TA, AMBIENT TEMPERATURE (°C)
3.5
3.0
100
Figure 8. ML13176 Reference Oscillator
Frequency versus Phase Detector Current
10.3
Closed Loop Response:
VCC = 3.0 Vdc
fo = 32 x fref
Vref = 500 mVp–p
10.2
10.1
40.4
39.6
– 30
∆ f OSC , OSCILLATOR FREQUENCY (MHz)
10
5.0
1000
Figure 6. Change in Oscillator Frequency and
Output Power versus Ambient Temperature
f ref , REFERENCE OSCILLATOR FREQUENCY (MHz)
∆ f OSC , OSCILLATOR FREQUENCY (MHz)
Figure 5. Change Oscillator Frequency
versus Oscillator Control Current
1.0
10
100
Ireg. enable, REGULATOR ENABLE CURRENT (µA)
PO , OUTPUT POWER (dBm)
I CC , SUPPLY CURRENT (mA)
10
– 20
Imod = 1.0 mA
ICC = 25 mA
PO = – 0.2 dBm
–10
0
10
20
I7, PHASE DETECTOR CURRENT (µA)
30
Imod = 1.0 mA
ICC = 22 mA
PO = –1.1 dBm
10
Imod = 2.0 mA
ICC = 35.5 mA
PO = 4.7 dBm
9.9
9.8
– 30
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– 20
–10
0
10
20
I7, PHASE DETECTOR CURRENT (µA)
30
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ML13175/ML13176
LANSDALE Semiconductor, Inc.
Figure 10. Change in Oscillator Frequency
versus Oscillator Control Current
20
∆f OSC , OSCILLATOR FREQUENCY (MHz)
∆ f OSC , OSCILLATOR FREQUENCY (MHz)
Figure 9. Change in Oscillator Frequency
versus Oscillator Control Current
10
VCC = 3.0 Vdc
Imod = 2.0 mA
TA = 25 °C
fosc (ICont @ 0) 320 MHz
0
–10
– 20
– 30
– 40
–100
0
400
500
100
200
300
ICont, OSCILLATOR CONTROL CURRENT (µA)
600
20
10
VCC = 3.0 Vdc
Imod = 2.0 mA
TA = 25 °C
fosc (ICont @ 0) 450 MHz
0
–10
– 20
– 30
– 40
–100
400
500
0
100
200
300
ICont, OSCILLATOR CONTROL CURRENT (µA)
600
Legacy Applications Information
APPLICATIONS INFORMATION
EVALUATION PC BOARD
The evaluation PCB, shown in Figures 26 and 27, is very versatile
and is intended to be used across the entire useful frequency range
of this device. The center section of the board provides an area for
attaching all SMT components to the component ground side of the
PCB (see Figures 28 and 29). Additionally, the peripheral area surrounding the RF core provides pads to add supporting and interface
circuitry as a particular application requires. This evaluation board
will be discussed and referenced in this section.
CURRENT CONTROLLED OSCILLATOR (Pins 1 to 4)
It is critical to keep the interconnect leads from the CCO (Pins 1
and 4) to the external inductor symmetrical and equal in length.
With a minimum inductor, the maximum free running frequency
is greater than 1.0 GHz. Since this inductor will be small, it may
be either a microstrip inductor, an air wound inductor or a tuneable RF coil. An air wound inductor may be tuned by spreading
the windings, whereas tunable RF coils are tuned by adjusting
the position of an aluminum core in a threaded coilform. As the
aluminum core coupling to the windings is increased, the inductance is decreased. The temperature coefficient using an aluminum core is better than a ferrite core. The UniCoil™ inductors made by Coilcraft may be obtained with aluminum cores
(Part No. 51–129–169).
GROUND (Pins 5, 10 and 15)
GROUND RETURNS: It is best to take the grounds to a backside ground plane via plated through holes or eyelets at the pins.
The application PCB layout implements this technique. Note
that the grounds are located at or less than 100 mils from the
device pins.
DECOUPLING: Decoupling each ground pin to VCC isolates
each section of the device by reducing interaction between sections and by localizing circulating currents.
LOOP CHARACTERISTICS (Pins 6 and 7)
Figure 11 is the component block diagram of the ML1317x PLL
system where the loop characteristics are described by the gain
constants. Access to individual components of this PLL system
is limited, inasmuch as the loop is only pinned out at the phase
Page 6 of 16
detector output and the frequency control input for the CCO.
However, this allows for characterization of the gain constants
of these loop components. The gain constants Kp, Ko and Kn
are well defined in the ML13175 and ML13176.
PHASE DETECTOR (Pin 7)
With the loop in lock, the difference frequency output of the
phase detector is DC voltage that is a function of the phase difference. The sinusoidal type detector used in the IC has the following transfer characteristic:
le = A Sin θe
The gain factor of the phase detector, Kp (with the loop in lock)
is specified as the ratio of DC output current, le to phase error,
θe:
Kp = le/θe (Amps/radians)
Kp = A Sin θe/θe
Sin θe ~ θe for θe ≤ 0.2 radians;
thus Kp = A (Amps/radians)
Figures 7 and 8 show that the detector DC current is approximately 30 µA where the loop loses lock at θe = ±π/2 radians;
therefore Kp is 30 µA/radians.
CURRENT CONTROLLED OSCILLATOR, CCO (Pin 6)
Figures 9 and 10 show the non–linear change in frequency of
the oscillator over an extended range of control current for 320
and 450 MHz applications. Ko ranges from approximately
6.3x105 rad/sec/µA or 100 kHz/µA (Figure 9) to 8.8x105
rad/sec/µA or 140 kHz/µA (Figure 10) over a relatively linear
response of control current (0 to 100 µA). The oscillator gain
factor depends on the operating range of the control current
(i.e., the slope is not constant). Included in the CCO gain factor
is the internal amplifier which can sink and source at least
30µA of input current from the phase detector. The internal circuitry at Pin 6 limits the CCO control current to 50 µA of
source capability while its sink capability exceeds 200 µA as
shown in Figures 9 and 10. Further information to follow shows
how to use the full capabilities of the CCO by addition of an
external loop amplifier and filter (see Figure 15). This additional circuitry yields at Ko = 0.145 MHz/µA or 9.1x105
rad/sec/µA.
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ML13175/ML13176
Legacy Applications Information
Figure 11. Block Diagram of ML1317x PLL
θi(s)
fi = f ref
Pins 9,8
Phase
Detector
θe(s)
Kp = 30 µA/rad
Pin 7
fn = fo/N
Low Pass
Filter
Kf
θn(s) = θo(s)/N
Divider
θo(s)
Kn = 1/N
N = 8 : ML13175
N = 32 : ML13176
Pin 6
Amplifier and
Current Controlled
Oscillator
Phase detector gain constant in
µA/rad; Kp = 30 µA/rad
Filter transfer function
1/N; N = 8 for the MC13175 and
1/N; N = 32 for the MC13176
CCO gain constant in rad/sec/µA
9.1 x 105 rad/sec/µA
Ko = 0.91Mrad/sec/µA
fo = nfi
Pins 13,14
LOOP FILTERING
The fundamental loop characteristics, such as capture range,
loop bandwidth, lock–up time and transient response are controlled externally by loop filtering.
The natural frequency (ωn) and damping factor (L ) are
important in the transient response to a step input of phase or
frequency. For a givenL and lock time wn can be determined
from the plot shown in Figure 12.
Figure 12. Type 2 Second Order Response
1.9
1.8
Where: Kp =
=
Kf =
Kn =
Ko =
=
Ko =
For L = 0.707 and lock time = 1.0 ms;
then ω = 5.0/t = 5.0 krad/sec.
The loop filter may take the form of a simple low pass filter
or a lag–lead filter which creates an additional pole at origin
in the loop transfer function. This additional pole along with
2
that of the CCO provides two pure integrators (1/s ). In the
lag–lead low pass network shown in Figure 13, the values of
the low pass filtering parameters R1, R2 and C determine the
loop constants ωn and L. The equations t1=R1 C and t2=R2C
are related in the loop filter transfer functions
F(s) = 1 +
t2s/1 + (t1 +t2)s.
Figure 13. Lag–Lead Low Pass Filter
ζ = 0.1
1.7
Vin
1.6
0.2
VO
0.3
1.4
θ o (t), NORMALIZED OUTPUT RESPONSE
R2
C
1.5
The closed loop transfer function takes the form of a 2nd
order low pass filter given by,
H(s )= KvF(s)/s + KvF(s)
From control theory, if the loop filter characteristic has F(0) =
1, the DC gain of the closed loop, Kv is defined as,
Kv = KpKoKn
and the transfer function has a natural frequency,
ωn = Kv/t1 + t2)1/2
and a dampning factor,
L = (ωn/2) (t2 + 1Kv)
Rewriting the above equations and solving for the ML13176
with L = 0.707 and ωn = 5.0 k rad/sec.
Kv = KpKoKn = (30) (0.91 X 106)(1/32) = 0.853 X 106
t1 + t2 = Kv/ωn2 = 0.853 X 106/(25 X 106) = 34.1 ms
t2 = 2L /ωn = (2)(0.707)/(5 X 103) = 0.283 ms
t1 = (Kv/ωn2) –t2=(34.1–0.283) = 33.8 ms
0.4
1.3
1.2
0.5
1.1
0.6
1.0
0.7
0.9
0.8
1.0
0.8
1.5
0.7
2.0
0.6
0.5
0.4
0.3
0.2
0.1
0
R1
0
Page 7 of 16
1.0 2.0 3.0
4.0 5.0 6.0 7.0 8.0 9.0
ωnt
10
11
12 13
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Legacy Applications Information
For C = 0.47 µ:
then R1 = t1/C = 33.8 X 10 –3/0.47 X 10 –6 = 72 k
thus, R2 = t2/C = 0.283 X 10 – 3/0.47 X 10 –6 = 0.60k
In the above example, the following standard value components
are used,
C = 0.47 µf; R2 = 620 and R’1 = 72 kΩ – 53 kΩ ~ 18 kΩ
(R’1 is defined as R1 – 53 kΩ, the output impedance of the phase
detector.)
through 8 are a direct measurement of the hold–in range (i.e.
∆fref x N = ±∆fH x 2π). Since sin θe cannot exceed ±1.0, as θe
approaches ±π/2 the hold–in range is equal to the DC loop gain
Kv X N.
±∆ωH = ± Kv x N
where, Kv = KpKoKn.
In the above example,
±∆ωH = ±27.3 Mrad/sec
±∆fH = ±4.35 MHz
Since the output of the phase detector is high impedance (~50 k)
and serves as a current source, and the input to the frequency control, Pin 6 is low impedance (impedance of the two diode to
ground is approximately 500 Ω), it is imperative that the second
order low pass filter design above be modified. In order to minimize loading of the R2C shunt network, a higher impedance must
be established to Pin 6. A simple solution is achieved by adding a
low pass network between the passive second order network and
the input to Pin 6. This helps to minimize the loading effects on
the second order low pass while further suppressing the sideband
spurs of the crystal oscillator. A low pass filter with R3 = 1.0 k
and C2 - 1500 p has a corner frequency (fc) of 106 kHz; the reference sideband spurs are down greater than – 60 dBc.
EXTENDED HOLD–IN RANGE
The hold–in range of about 3.4% could cause problems over temperature in cases where the free–running oscillator drifts more
than 2 to 3% because of relatively high temperature coefficients
of the ferrite tuned CCO inductor. This problem might worsen for
lower frequency applications where the external tuning coil is
large compared to internal capacitance at Pins 1 and 4. To
improve hold–in range performance, it is apparent that the gain
factors involved must be carefully considered.
Kn = is either 1/8 in the ML13175 or 1/32 in the ML13176
Kp = is fixed internally and cannot be altered.
Ko = Figures 9 and 10 suggest that there is capability of
greater control range with more current swing. However,
this swing must be symmetrical about the center of the
dynamic response. The suggested zero current operating
point for ±100 µA swing of the CCO is at about + 70
µA offset point.
Ka = External loop amplification will be necessary since the
phase detector only supplies ±30 µA.
Figure 14. Modified Low Pass Loop Filter
Pin 7
18kΩ
1.0kΩ
R' 1
620
R2
0.47
µf
C
Pin 6
R3
C3
1500pf
VCC
In the design example in Figure 15, an external resistor (R5) of
15 kΩ to VCC (3.0 Vdc) provides approximately 100 µA of current boost to supplement the existing 50 µA internal source current. R4 (1.0 kΩ) is selected for approximately 0.1 Vdc across it
with 100 µA. R1, R2 and R3 are selected to set the potential at
Pin 7 and the base of 2N4402 at approximately 0.9 Vdc and the
emitter at 1.55 Vdc when error current to Pin 6 is approximately
zero µA. C1 is chosen to reduce the level of the crystal sidebands.
HOLD–IN RANGE
The hold–in range, also called the lock range, tracking range and
synchronization range, is the ability of the CCO frequency, fo to
track the input reference signal, fref • N as it gradually shifted
away from the free running frequency, ff. Assuming that the CCO
is capable of sufficient frequency deviation and that the internal
loop amplifier and filter are not overdriven, the CCO will track
until the phase error, θe approaches ±π/2 radians. Figures 5
Figure 15. External Loop Amplifier
VCC = 3.0Vdc
12
30µA
Phase
Detector
Output
R3
1000p
R1
68k
R2
33k
7
30µA
Page 8 of 16
C1
4.7k
R5
R4
1.0k
2N4402
50µA
15k
1.6V
6
Oscillator
Control
Circuitry
5, 10, 15
www.lansdale.com
Issue c
C
LANSDALE Semiconductor, Inc.
ML13175/ML13176
Legacy Applications Information
f ref , REFERENCE OSCILLATOR FREQUENCY (MHz)
Figure 16 Shows the improved hold–in range of the loop. The ∆fref
is moved 950 kHz with over 200 µA swing of control current for an
improved hold–in range of ±15.2 MHz or ±95.46 Mrad/sec.
Figure 16. ML13176 Reference Oscillator
Frequency versus Oscillator Control Current
10.6
10.4
10.2
10
Closed Loop Response:
fo = 32 x fref
VCC = 3.0 Vdc
ICC = 38 mA
Pout = 4.8 dB
Imod = 2.0 mA
Vref = 500 mVp–p
9.8
9.6
9.4
–150
–100
– 50
0
50
I6, OSCILLATOR CONTROL CURRENT (µA)
100
fc = 0.159/RC;
For R = 1.0 k + R7 (R7 = 53 k) and C = 390 pF
fc = 7.55 kHz or ωc = 47 krad/sec
The application example in Figure 17a of a 320 MHz FM transmitter
demonstrates the FM capabilities of the IC. A high value series resistor
(100 k) to Pin 6 sets up the current source to drive the modulation section of the chip. Its value is dependent on the peak to peak level of the
encoding data and the maximum desired frequency deviation. The data
input is AC coupled with a large coupling capacitor which is selected
for the modulating frequency. The component placements on the circuit
side and ground side of the PC board are shown in Figures 28 and 29
respectively.
For voice application using a dynamic or an electret microphone, an op
amp is used to amplify the microphone’s low level output . The microphone amplifier circuit is shown in Figure 19. Figure 17b shows an
application example for NBFM audio or direct FSK in which the reference crystal oscillator is modulated.
Figure 19. Microphone Amplifier
LOCK–IN RANGE/CAPTURE RANGE
If a signal is applied to the loop not equal to free running frequency,
ff, then the loop will capture or lock–in the signal by making fs = fo
(i.e. if the initial frequency difference is not too great). The lock–in
range can be expressed as ∆ωL ~ ± 2L ωn
FM MODULATION
Noise external to the loop (phase detector input) is minimized by narrowing the bandwidth. This noise if minimal in a PLL system since the
reference frequency is usually derived from a crystal oscillator. FM can
be achieved by applying a modulation current superimposed on the
control current of the CCO. The loop bandwidth must be narrow
enough to prevent the loop from responding to the modulation frequency components, thus, allowing the CCO to deviate in frequency. The
loop bandwidth is related to the natural frequency wn. In the lag–lead
design example where the natural frequency, ωn = 5.0 krad/sec and a
damping factor, L = 0.707, the loop bandwidth = 1.64 kHz.
Characterization data of the closed loop responses for both the
ML13175 and ML13176 at 320 MHz (Figures 7 and 8, respectively)
show satisfactory performance using only a simple low–pass loop filter
network. The loop filter response is strongly influenced by the high
output impedance of the push–pull current output of the phase detector.
Page 9 of 16
Data
Input
VCC
3.3k
Voice
Input
1.0k
100k 120k
3.9k
10k
10k
Electret
Microphone
VCC
1.0
MC33171
Data or
Audio
Output
LOCAL OSCILLATOR APPLICATION
To reduce internal loop noise, a relatively wide loop bandwidth is
needed so that the loop tracks out or cancels the noise. This is emphasized to reduce inherent CCO and divider noise or noise produced by
mechanical shock and environmental vibrations. In a local oscillator
application the CCO and divider noise should be reduced by proper
selection of the natural frequency of the loop. Additional low pass filtering of the output will likely be necessary to reduce the crystal sideband spurs to a minimal level.
www.lansdale.com
Issue c
ML13175/ML13176
LANSDALE Semiconductor, Inc.
Legacy Applications Information
Figure 17a. 320 MHz ML13176 FM Transmitter
RF Level Adjust
1.1k
Osc
Tank
5.0k
16
1
0.047µ
2
15
3
14
CW
Coilcraft
146–04J08
(1)
SMA
0.146µ
0.47µ
130k
(2)
VEE
15k
510p
13
f/32
0.1µ
9.1k
RFC1 (3)
5
12
6
11
VCC
18k
10
7
2N4402
0.47µ
100k
33k
VCC
27k
1.0k
620
RF Output
to Antenna
50Ω
4
VCC = 3.8 to
3.3 Vdc
VCC
9
8
Data Input
(1.6 Vp–p)
51p
VCC
51p
220p
6.8 (4)
Crystal
Fundamental
10 MHz
(5)
NOTES: 1. 50 Ω coaxial balun, 2 inches long.
2. Pins 5, 10 and 15 are grounds and connnected to VEE which is the component's side ground plane.
These pins must be decoupled to VCC; decoupling capacitors should be placed as close as possible to the pins.
3. RFC1 is 180 nH Coilcraft surface mount inductor or 190 nH Coilcraft 146–05J08.
4. Recommended source is a Coilcraft ™slot seven 7.0 mm tuneable inductor, part #7M3–682.
5. The crystal is a parallel resonant, fundamental mode calibrated with 32 pF load capacitance.
Figure 17b. 320 MHz NBFM Transmitter
RF Level Adjust
1.0k
Osc
Tank
5.0k
16
1
0.047µ
2
15
3
14
Coilcraft
146–04J08
SMA
130k
6.2k
0.1µ
9.1k
(2)
VEE
15k
f/32
470p
13
RFC1 (3)
5
12
6
11
VCC (3.6 Vdc – Lithium Battery)
VCC
27k
1.0k
15k
2N4402
7
10
8
9
0.47µ
33k
VCC
10p
External
Loop Amp
100p
180p
(6)
Crystal
Fundamental
10MHz
RFC2
(4)
www.lansdale.com
VCC
1.0k
10µ
RFC3
(5) MMBV432L
NOTES: 1. 50 Ω coaxial balun, 2 inches long.
2. Pins 5, 10 and 15 are grounds and connnected to VEE which is the component's side ground plane. These
pins must be decoupled to VCC; decoupling capacitors should be placed as close as possible to the pins.
3. RFC1 is 180 nH Coilcraft surface mount inductor.
4. RFC2 and RFC3 are high impedance crystal frequency of 10 MHz; 8.2 µH molded inductor gives XL > 1000 Ω..
5. A single varactor like the MV2105 may be used whereby RFC 2 is not needed.
6. The crystal is a parallel resonant, fundamental mode calibrated with 32 pF load capacitance.
Page 10 of 16
RF Output
to Antenna
UT–034
4
4700p
CW
(1)
0.146µ
VCC
VCC
+
0.01µ
Audio or
Data Input
Issue c
LANSDALE Semiconductor, Inc.
ML13175/ML13176
Legacy Applications Information
REFERENCE CRYSTAL OSCILLATOR
(Pins 8 and 9)
Selection of Proper Crystal: A crystal can operate in a number of mechanical modes. The lowest resonant frequency
mode is its fundamental while higher order modes are called
overtones. At each mechanical resonance, a crystal behaves
like a RLC series–tuned circuit having a large inductor and a
high Q. The inductor Ls is series resonance with a dynamic
capacitor, Cs determined by the elasticity of the crystal lattice
and a series resistance Rs, which accounts for the power dissipated in heating the crystal. This series RLC circuit is in parallel with a static capacitance, Cp which is created by the
crystal block and by the metal plates and leads that make contact with it.
Figure 20 is the equivalent circuit for a crystal in a signal resonant mode. It is assumed that other modes of resonance are
so far off frequency that their effects are negligible.
Series resonant frequency, fs is given by;
fs = 1/2π(LsCs)1/2
and parallel resonant frequency, fp is given by;
fp = fs(1 + Cs/Cp)1/2
Page 11 of 16
Figure 20. Crystal Equivalent Circuit
L3
Cp
R3
C3
the frequency separation at resonance is given by;
∆f = fp–fs = fs[1 – (1+ Cs/Cp)1/2]
Usually fp is less than 1% higher than fs, and a crystal
exhibits an extremely wide variation of the reactance with
frequency between fp and fs. A crystal oscillator circuit is
very stable with frequency. This high rate of change of
impedance with frequency stabilizes the oscillator, because
any significant change in oscillator frequency will cause a
large phase shift in the feedback loop keeping the oscillator
on frequency.
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Issue c
ML13175/ML13176
LANSDALE Semiconductor, Inc.
Legacy Applications Information
Manufacturers specify crystal for either series or parallel resonant operation. The frequency for the parallel mode is calibrated with a specified shunt capacitance called a “load
capacitance.” The most common value is 30 to 32 pF. If the
load capacitance is placed in series with the crystal, the
equivalent circuit will be series resonance at the specified
parallel–resonant frequency. Frequencies up to 20 MHz use
parallel resonant crystal operating in the fundamental mode,
while above 20 MHz to about 60 MHz, a series resonant
crystal specified and calibrated for operation in the overtone
mode is used.
APPLICATION EXAMPLES
Two types of crystal oscillator circuits are used in the applications circuits: 1) fundamental mode common emitter
Colpitts (Figures 1, 17a, 17b and 21) and 2) third overtone
impedance inversion Colpitts (also Figures 1 and 21).
The fundamental mode common emitter Colpitts uses a parallel resonant crystal calibrated with a 32 pf load capacitance.
The capacitance values are chosen to provide excellent frequency stability and output power of > 500 mVp–p at Pin 9.
In Figures 1 and 21, the fundamental mode reference oscillator is fixed tuned relying on the repeatability of the crystal
and passive network to maintain the frequency, while in the
circuit shown in Figure 17, the oscillator frequency can be
adjusted with the variable inductor for the precise operating
frequency.
The third overtone impedance inversion Colpitts uses a series
resonance crystal with a 25 ppm tolerance. In the application
examples (Figures 1 and 21), the reference oscillator operates
with the third overtone crystal at 40.0000 MHz. Thus, the
ML13175 is operated at 320 MHz (fo/8 = crystal; 320/8) =
40.0000 MHz. The resistor across the crystal ensures that the
crystal will operate in the series resonance mode. A tuneable
inductor is used to adjust the oscillation frequency; it forms a
parallel resonant circuit with the series and parallel combination of the external capacitors forming the divider and feedback network and the base–emitter capacitance of the
devices. If the crystal is shorted, the reference oscillator
should free–run at the frequency dictated by the parallel resonant LC network.
The reference oscillator can be operated as high as 60 MHz
with a third overtone crystal. Therefore, it is possible to use
the ML13175 up to at least 480 MHz and the ML13176 up to
950 MHz (based on the maximum capability of the divider
netowork).
ENABLER (Pin 11)
The enabling resistor at Pin 11 is calculated by:
Page 12 of 16
Reg. enable = VCC – 1.0 Vdc/lreg. enable
From Figure 4, lreg.enable is chosen to be 75µA. So, for a
VCC = 3.0 Vdc Rreg.enable = 26.6 kΩ, a standard value 27
kΩ resistor is adequate.
LAYOUT CONSIDERATIONS
Supply (Pin 12): In the PCB layout the VCC trace must be
kept as wide as possible to minimize inductive reactance
along the trace; it is best that VCC (RF ground) completely
fills around the surface mounted components and interconnect traces on the circuit side of the board. This technique is
demonstrated in the evaluation PC board.
BATTERY/SELECTION/LITHIUM TYPES
The device may be operated from a 3.0 V lithium battery.
Selection of a suitable battery is important. Because one of
the major problems for long life battery powered equipment
is oxidation of the battery terminals, a battery mounted in
clip–in socket is not advised. The battery leads or contact
post should be isolated from the air to eliminate oxide
build–up. The battery should have PC board mounting tabs
which can be soldered to the PCB. Consideration should be
given for the peak current capability of the battery. Lithium
batteries have current handling capabilities based on the composition of the lithium compound, construction and the battery size. A 1300 mA/hr rating can be achieved in the cylindrical cell battery. The Rayovac CR2/3A lithium–manganese
dioxide battery is a crimp sealed, spiral wound 3.0 Vdc, 1300
mA/hr cylindrical cell with PC board mounting tabs. It is an
excellent choice based on capacity and size (1.358” long by
0.665” in diameter).
DIFFERENTIAL OUTPUT (Pins 13, 14)
The availability of micro–coaxial cable and small baluns in
surface mount and radial–leaded components allows for simple interface to the output ports. A loop antenna may be
directly connected with bias via RFC or 50 Ω resistors.
Antenna configuration will vary depending on the space
available and the frequency of operation.
AM MODULATION (Pin 16)
Amplitude Shift Key: The ML13175 and ML13176 are
designed to accommodate Amplitude Shift Keying (ASK).
ASK modulation is a form of digital modulation corresponding to AM. The amplitude of the carrier is switched between
two or more values in response to the PCM code. For the
binary case, the usual choice is On–Off Keying (often abbreviated OOK). The resultant amplitude modulated waveform
consists of RF pulses called marks, representing binary 1 and
spaces representing binary 0.
www.lansdale.com
Issue c
LANSDALE Semiconductor, Inc.
ML13175/ML13176
Legacy Applications Information
Figure 21. ASK 320 MHz Application Circuit
Rmod
3.3k
Osc
Tank
1
16
2
15
3
14
0.01µ
Coilcraft
150–05J08
VEE
(1)
0.165µ
0.1µ
150p
100p
(ML13176)
VCC
f/N
12
6
11
7
10
8
9
VCC
VEE
ML13175–30p
ML13176–180p
2. Pins 5, 10 and 15 are ground and connnected to VEE which is
the component/DC ground plane side of PCB. These pins must
be decoupled to VCC; decoupling capacitors should be placed
as close as possible to the pins.
3. The crystal oscillator circuit may be adjusted for frequency with
the variable inductor (MC13175); 1.0 k resistor shunting the
crystal prevents it from oscillating in the fundamental mode.
Recommended source is Coilcraft “slot seven” 7.0 mm tuneable
inductor, part #7M3–821.
Figure 21 shows a typical application in which the output
power has been reduced for linearity and current drain. The
current draw on the device is 16 mA ICC (average) and –22.5
dBm (average power output) using a 10 kHz modulating rate
for the on–off keying. This equates to 20 mA and –2.3 dBm
“on”, 13 mA and –41 dBm “Off ”. The crystal oscillator
27k S1
(5)
0.1µ
0.01µ
0.82µ
NOTES: 1. 50 Ω coaxial balun, 1/10 wavelength line (1.5" ) provides the best
match to a 50 Ω load.
RFOut
RFC1
5
ML13175
Crystal
1.0k
3rd Overtone
40.0000 MHz
Page 13 of 16
150p
13
1.0k
30p
(ML13175)
SM
A
Z = 50
4
(2)
VEE
(4)
On–Off Keyed Input
TTL Level 10 kHz
(3) ML13176
Crystal
Fundamental
10 MHz
VCC
4. The On–Off keyed signal turns the output of the transmitter off and on with
TTL level pulses through R mod at Pin 16. The "On" power and I CC is set
by the resistor which sets Imod = VTTL – 0.8 / Rmod. (see Figure 23).
5. S1 simulates an enable gate pulse from a microprocessor which will
enable the transmitter. (see Figure 4 to determine precise value of the
enabling resistor based on the potential of the gate pulse and the
desired enable.)
enable time is needed to set the acquisition timing. It takes
typically 4.0 msec to reach full magnitude of the oscillator
waveform. A square waveform of 3.0 V peak with a period
that is greater than the oscillator enable time is applied to the
Enable (Pin 11)
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Issue c
ML13175/ML13176
LANSDALE Semiconductor, Inc.
Legacy Applications Information
Figure 23. Power Output versus Modulation Current
10
PO, POWER OUTPUT (dBm)
5.0
0
– 5.0
VCC = 3.0 Vdc
f = 320 MHz
–10
–15
– 20
– 25
0.1
Page 14 of 16
1.0
Imod, MODULATION CURRENT (mA)
10
ANALOG AM
In analog AM applications, the output amplifier’s linearity
must be carefully considered. Figure 23 is a plot of Power
Output versus Modulation Current at 320 MHz, 3.0 Vdc. In
order to achieve a linear encoding of the modulating sinusoidal waveform on the carrier, the modulating signal must
amplitude modulate the carrier in the linear portion of its
power output response. When using a sinewave modulating
signal, the signal rides on a positive DC offset called Vmod
which sets a static (modulation off) modulation current,
Imod. Imod controls the power output of the IC. As the modulating signal moves around this static bias point the modulating current varies causing power output to vary or to be
AM modulated. When the IC is operated at modulation current levels greater than 2.0 mAdc the differential output stage
starts to saturate.
www.lansdale.com
Issue c
ML13175/ML13176
LANSDALE Semiconductor, Inc.
Legacy Applications Information
In the design example, shown in FIgure 24, the operating
point is selected as a tradeoff between average power output
and quality of the AM.
For VCC = 3.0 Vdc;ICC = 18.5 mA and Imod = 0.5 mAdc
and a static DC offset of 1.04 Vdc, the circuit shown in
Figure 24 completes the design.
Where Rmod = (VCC – 1.04 Vdc)/0.5 mA = 3.92 kΩ, use a
standard value resistor of 3.9 kΩ.
Page 15 of 16
www.lansdale.com
Figure 24. Analog AM Transmitter
3.9kΩ 1.04Vdc 560
VCC
16
R
3.0Vdc mod
0.8Vdc
Data
Input
800mVp–p
+
6.8µ
Issue c
ML13175/ML13176
LANSDALE Semiconductor, Inc.
OUTLINE DIMENSIONS
PLASTIC PACKAGE
(ML13175-5P, ML13176-5P)
CASE 751B
(SO–16)
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: MILLIMETER.
3. DIMENSION A AND B DO NOT INCLUDE MOLD
PROTRUSION.
4. MAXIMUM MOLD PROTRUSION 0.15 (0.006)
PER SIDE.
5. 751B–01 IS OBSOLETE, NEW STANDARD
751B–03.
–A
–
16
9
P
–B
–
1
0.25 (0.010)
M
B
M
8 PL
8
R X 45°
G
C
–T
–
SEATING
PLANE
D 16 PL
0.25 (0.010)
K
M
T B
S
A
M
F
S
J
DIM
A
B
C
D
F
G
J
K
M
P
R
MILLIMETERS
MIN
MAX
9.80 10.00
4.00
3.80
1.75
1.35
0.49
0.35
1.25
0.40
1.27 BSC
0.25
0.19
0.25
0.10
5.80
0.25
6.20
0.50
INCHES
MIN
MAX
0.386 0.393
0.150 0.157
0.054 0.068
0.014 0.019
0.016 0.049
0.050 BSC
0.008 0.009
0.004 0.009
0.229
0.010
0.244
0.019
Lansdale Semiconductor reserves the right to make changes without further notice to any products herein to improve reliability, function or design. Lansdale does not assume any liability arising out of the application or use of any product or circuit
described herein; neither does it convey any license under its patent rights nor the rights of others. “Typical” parameters which
may be provided in Lansdale data sheets and/or specifications can vary in different applications, and actual performance may
vary over time. All operating parameters, including “Typicals” must be validated for each customer application by the customer’s technical experts. Lansdale Semiconductor is a registered trademark of Lansdale Semiconductor, Inc.
Page 16 of 16
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Issue c