SC2308A
600kHz Step-Up Switching
Regulator with 2.2A, 45V Switch
POWER MANAGEMENT
Features
Description
Input Voltage Range: 2.6V to 20V
Boost and SEPIC Topologies
Up to 40V Output in Boost Topology
Integrated 2.2A/45V Switch
600kHz Constant Switching Frequency
Current-Mode Control Eases Compensation
Cycle-by-Cycle Current-Limiting
Internal Soft-Start
Thermal Shutdown Protection
Low Shutdown Current (<1µA)
8-Pin SO Lead-Free Package
Fully WEEE and RoHS Compliant
The SC2308A is a 600kHz current-mode switching regulator
with an integrated low-side 2.2A power transistor. The operating supply voltage of the SC2308A ranges from that of a
single Li-ion cell to various PC board power supplies. The
internal switch is rated at 45V, making the device suitable
for high voltage boost and SEPIC applications. The SC2308A
shuts down to less than 1µA of supply current.
The SC2308A uses peak current-mode PWM control for ease
of loop compensation and excellent transient response.
Cycle-by-cycle current limiting lowers power transistor
dissipation. An internal soft-start timer prevents output
overshoot and limits the input current during start-up.
Thermal shutdown prevents the chip from overheating.
Applications
Telecommunication Equipment
Point of Load DC-DC Converters
Portable Devices
Typical Application Circuit
L
5V
10µH
IN
OFF ON
C1
4.7µF
EN
D1
VOUT
SS22
SW
C3
C2
22µF
22µF
FB
C4
470pF
V IN = 5V
90
R1
100k
SC2308A
COMP
100
12V, 0.8A
GND
R2
11.3k
R3
200k
80
E fficiency (%)
VIN
E fficiency vs Load C urrent
VOUT = 12V
70
60
50
40
30
L: Coilcraft DO3316P103
C1: Murata GRM31CR61A475K
D1: ON SS22
C2,C3: Murata GRM31CR61C226K
20
0
200
400
600
800
Load Current (mA)
Figure 1. 5V to 12V Step-Up Converter
Rev. 2.0
SC2308A
Pin Configuration
Ordering Information
Top View
COMP 1
8 NC
FB 2
7 NC
EN 3
6 IN
5 SW
GND 4
θJA = 160°C/W
8-Lead SOIC
Marking Information
Top View
SC2308A
yyww
xxxxx
yyww - Date Code
xxxxx - Semtech Lot Number
Device
Package
SC2308ASTRT(1) (2)
SO-8
SC2308AEVB
Evaluation Board
Notes:
(1) Available in tape and reel only. A reel contains 2,500 devices.
(2) Available in lead-free package only. Device is WEEE and RoHS
compliant and halogen free.
SC2308A
Recommended Operating Conditions
Absolute Maximum Ratings
Junction Temperature Range………………… -40°C to +105°C
VIN ………………………………………………… 2.6V to 20V
IN to GND………………………………………… -0.3V to 24V
SW …………………………………………………-0.3V to 45V
EN ……………………………………………-0.3V to VIN + 0.3V
Thermal Information
FB …………………………………………… -0.3V to VIN + 0.3V
COMP ……………………………………… -0.3V to VIN + 0.3V
ESD Protection Level …………………………………
(1)
ΘJA, 8-Lead SOIC(2) …………………………………… 160°C/W
Maximum Junction Temperature …………………… +150 °C
Storage Temperature Range ……………… -65°C to +150°C
Peak IR Reflow Temperature (10s to 30s) …………… +260°C
3.5kV
Exceeding the above specifications may result in permanent damage to the device or device malfunction. Operation outside of the parameters
specified in the Electrical Characteristics section is not recommended.
Notes:
(1) Tested according to JEDEC standard JESD22-A114-B.
(2) Calculated from package in still air, mounted to 3” x 4.5”, 4 layer FR4 PCB with thermal vias under the exposed pad per JESD51 standards.
Electrical Characteristics
Unless otherwise noted: VIN = VEN = 3V, TJ = -40°C to 105°C. Typical values are at TJ = 25°C.
Parameter
Symbol
Conditions
Min
Typ
Max
Units
20
V
Input Supply
Maximum Operating VIN
VIN(MAX)
VIN Start Voltage
VIN Rising
2.45
2.6
V
VEN = 0
0.01
1
µA
VFB = 1.5V (Not Switching)
1.3
1.8
mA
1.22
1.24
V
VIN = 3V to 20V
0.002
0.005
%/V
-25
nA
Shutdown Supply Current
Quiescent Supply Current
IQ
Control Loop
Feedback Regulation Voltage
VREF
VREF Line Regulation
1.20
FB Pin Input Bias Current
IFB
FB in Regulation
-15
Error Amplifier Transconductance
gm
VCOMP = 1.1V, DICOMP = ± 0.5µA
47
µW-1
Error Amplifier Open-Loop Gain
AV
51
dB
4
A/V
3
ms
COMP to Switch Current Gain
Soft-Start
Soft-Start Time(3)
tSS
Notes:
(3) Time taken for the error amplifier soft-start input to rise from 0 to 1.22V.
SC2308A
Electrical Characteristics (continued)
Unless otherwise noted: VIN = VEN = 3V, TJ = -40°C to 105°C. Typical values are at TJ = 25°C.
Parameter
Symbol
Conditions
Min
Typ
Max
Units
500
630
750
kHz
Oscillator
Switching Frequency
fSW
Minimum Switch Off-Time
tOFF(MIN)
60
ns
Minimum Switch On-Time
tON(MIN)
200
ns
Minimum Duty Cycle
DMIN
0
%
Maximum Duty Cycle
DMAX
87
96
ILIM
2.2
2.9
3.7
A
%
Power Switch
Switch Current Limit(4)
Switch Saturation Voltage
Switch Leakage Current
VCESAT
ISW=2.2A
320
480
mV
ILK
VSW =12V
0.1
0.5
µA
Enable Pin
High Voltage Threshold
VIH
Low Voltage Threshold
VIL
Enable Pin Current
IEN
2
V
0.3
V
VEN =0V
0.01
0.1
VEN =2V
3.3
5.1
VEN =6V
13
25
Tj rising
160
O
C
12
O
C
µA
Over Temperature Protection
Thermal Shutdown Temperature
TSHDN
Hysteresis
THYST
Notes:
(4) Switch current limit does not vary with duty cycle.
SC2308A
Typical Characteristics
E fficiency vs Load C urrent
VOUT = 12V
Efficiency vs Load Current
VOUT = 5V
V IN = 3.3V
90
V IN = 5V
V IN = 3V
80
70
60
50
60
50
40
40
30
30
20
V IN = 3.3V
70
Voltage (V)
E fficiency (%)
1.21
1.20
20
0
200
400
600
800
1000
1.22
1200
0
200
400
600
-50
800
Load Current (mA)
Load Current (mA)
Percentage Frequency
Variation vs Temperature
Switc h Saturation
Voltage vs Switc h Curre nt
S aturation Voltage (mV)
0
-5
-10
-15
-25
0
25
50
75
100
75
100
125
105oC
400
2.8
25oC
300
-45oC
200
100
2.6
2.4
V IN = 3V
2.2
2.0
0.0
125
50
Switch Current Limit vs Temperature
0
-50
25
3.0
V IN = 3V
V IN = 3V
0
Temperature ( C)
500
5
-25
o
Current (A)
Efficiency (%)
1.23
90
80
Percenatge Variation (%)
Feedback Voltage vs Temperature
100
100
0.5
1.0
1.5
2.0
2.5
3.0
3.5
-50
-25
0
25
50
75
100
125
Temperature (oC)
S witch Current (A)
Temperature ( C)
IN Pin Curre nt v s Switc h Curre nt
Error Amplifier Open-Loop
Gain vs Temperature
Error Amplifier
Transconductance vs Temperature
80
o
55
50
V IN = 3V
V IN = 3V
V IN = 3V
Transconductance (P : -1)
50
Gain (dB)
IN P in Current (mA)
45
60
40
45
20
40
35
30
25
25oC
0
20
40
0.0
0.5
1.0
1.5
2.0
2.5
S witch Current (A)
3.0
3.5
-50
-25
0
25
50
75
o
Temperature ( C)
100
125
-50
-25
0
25
50
75
100
125
o
Temperature ( C)
SC2308A
Typical Characteristics (Cont.)
2.8
2.0
2.6
1.5
2.0
25oC
2.4
1.5
-40oC
105oC
Current (mA)
Current (mA)
Input Voltage (V)
VIN Quiescent Current
vs Temperature
VIN Quiescent Current vs VIN
Minimum VIN vs Temperature
1.0
1.0
2.2
0.5
0.5
2.0
0.0
0.0
V IN = 3V
V EN = 3V
-50
-25
0
25
50
75
100
0
125
5
10
15
Temperature ( C)
EN Pin Current vs VEN
Soft-Start Time (1) vs Temperature
V IN = 3V
-40oC
3.5
25oC
Time (ms)
Current (P A)
60
40
105oC
20
3.0
2.5
2.0
0
0
5
10
V EN (V)
15
20
-50
-25
0
25
50
75
Temperature (oC)
Notes:
(1) Time taken for the error amplifier soft-start input to rise from 0 to 1.22V.
-50
-25
0
25
50
75
Temperature (oC)
4.0
80
20
V IN (V)
o
100
125
100
125
SC2308A
Pin Descriptions
Pin #
Pin Name
Pin Function
1
COMP
2
FB
The Inverting Input of the Error Amplifier. Tie to an external resistive divider to set the output voltage.
3
EN
Enable Pin. Pulling this pin below 0.3V shuts down the SC2308A to less than 1mA of quiescent current. Applying
more than 2V at this pin enables the SC2308A. For normal operation, this pin can be tied to IN or driven from a
logic gate with VOH > 2V.
4
GND
5
SW
Collector of the Internal Power Transistor. Connect to a boost inductor and a freewheeling diode. The maximum
switching voltage spike at this pin should be limited to less than 45V.
6
IN
Power Supply Pin. Bypassed with capacitor close to the pin.
7
NC
No connection.
8
NC
No connection.
Error Amplifier Output. The voltage at this pin controls the peak switch current. A series RC network from this pin
to ground compensates the control loop.
Ground. Tie to the ground plane. The converter output capacitor must be closely bypassed to the ground pin.
Block Diagram
SW
IN
6
5
REF NOT READY
EN
3
VOLTAGE
REFERENCE
THERMAL
SHUTDOWN
T > 160°C
J
CLK
1.22V
SOFTSTART
FB
2
SS
INTERNAL
SUPPLY
+
+ EA
-
-
R
+
S
PWM
Q1
Q
OC
ILIM
+
-
18mV
6.3mΩ
COMP
1
S
600kHz
OSCILLATOR
+
+
SLOPE COMP
+
ISEN
4
GND
Figure 2. SC2308A Block Diagram
SC2308A
General Description and Operation
The SC2308A is a 600kHz peak current-mode switching regulator with an integrated 2.2A (minimum) low-side power
transistor. The voltage reference runs off the input supply
and is enabled by applying at least 2V at the EN pin, as
shown in the block diagram in Figure 2. The reference also
senses VIN and produces a lockout signal “REF NOT READY”.
This signal does not go low until there is enough VIN headroom for the reference to achieve regulation (typically VIN
= 2.45V). The “REF NOT READY” signal and the temperature
sensor control the internal regulator, which powers all of
the internal control circuits.
The error amplifier EA has two non-inverting inputs. The
non-inverting input with the lower voltage predominates.
One of the non-inverting inputs is biased to a precision
1.22V reference and the other non-inverting input is tied
to a soft-start timer. Before the internal regulator turns
on, the output SS of the soft-start timer is discharged to
ground. As the internal regulator turns on, it also releases
the timer. The soft-start timer generates a slow rising SS
ramp, which is fed into one of the non-inverting inputs of
the EA. During power-up, the SS voltage becomes the EA
effective non-inverting input voltage. In a boost converter,
the part starts switching as VSS exceeds the FB voltage. If
the soft-start ramp is sufficiently slow, then the FB voltage
(hence the output voltage) will track VSS and there will be
no output overshoot during start-up. It takes about 3ms to
charge VSS from ground to the nominal feedback voltage.
The end of charge VSS is significantly higher than 1.22V so
that it has no effect on the error amplifier. Soft-start also
reduces the input start-up current.
The clock CLK resets the latch and blanks the power transistor Q1 conduction. Q1 is switched on at the trailing edge of
the clock. The switch current is sensed with an integrated
6.3mW sense resistor. The sensed current summed with
the slope-compensating ramp is fed into the modulating
ramp input of the PWM comparator. The latch is set and
Q1 conduction is terminated when the modulating ramp
intersects the error amplifier output. If the switch current
exceeds 2.9A (the typical current limit), then the currentlimit comparator ILIM will set the latch and turn off Q1. Due
to separate pulse-width modulating and current limiting
paths, cycle-by-cycle current limiting is not affected by
slope compensation.
The current-mode switching regulator is a dual-loop feedback control system, designed to simplify loop compensation. In the inner current loop, the EA output controls the
peak inductor current. In the outer loop, the error amplifier
regulates the output voltage. The double reactive poles
of the output LC filter are reduced to a single real pole
by the inner current loop, easing loop compensation. A
simple, two-pole, single-zero compensator network connected from COMP to ground is adequate to stabilize the
converter.
SC2308A
Applications Information
Duty Cycle
can be derived:
The duty cycle is the ratio of the switch on-time to the
switching period. For a boost converter, the duty cycle in
continuous-conduction mode (CCM) is:
D
VOUT VD VIN
VOUT VD VCESAT
(1)
where VCESAT is the switch saturation voltage and VD is the
rectifying diode forward voltage.
Setting the Output Voltage
VOUT d
(4)
where DMAX is the maximum duty cycle.
Example: Determine the highest output voltage that can
be achieved from a 3V input using the SC2308A as a boost
regulator. Assuming VD= 0.5V, VCESAT= 0.3V and using DMAX=
0.87:
VOUT d
The converter output voltage is set with an external resistive divider. The center tap of the divider is tied to the FB
pin ( Figure 3).
VIN DMAX VCESAT
VD
1 DMAX
3 0.87 u 0.3
0.5 20.6 V
1 0.87
The transient headroom requirement further reduces the
maximum achievable output voltage to less than 20V.
Maximum Output Current
VOUT
IIN
R1
SC2308A
Inductor
Current
2 FB
ON
OFF
ON
0
Switch Current
Diode Current
R2
(1-D)TS
DTS
ON
IOUT
ON
OFF
ON
0
Figure 3. R1 and R2 Divider Sets VOUT
Figure 4. Current Waveforms in a Boost Converter
The expression for VOUT is:
§ R ·
1.220 ¨¨ 1 1 ¸¸
(2)
© R2 ¹
R1 can be calculated from the output voltage and R2 as
follows:
VOUT
§ V
·
R1 R 2 ¨ OUT 1¸
1
.
220
©
¹
OFF
(3)
Using large resistors for the FB voltage divider reduces
power consumption.
Minimum Off-Time Limitation
There is also a 100ns minimum switch off-time, which limits
the maximum duty cycle. This determines the maximum
attainable output voltage for a given VIN. Using Equation
(1), the maximum output voltage for a boost converter
In a boost converter, the inductor is connected to the
input. The inductor DC current is the regulator input
current. When the power switch is turned on, the inductor
current flows into the switch. When the power switch is
off, the inductor current flows through the rectifying
diode to the output. The output current is the average
diode current. The diode current waveform is trapezoidal
with a pulse width of (1–D)TS (Figure 4). The output current
available from a boost converter, therefore, depends on
the converter operating duty cycle. The power switch
current in the SC2308A is internally limited to at least
2.2A. This is also the maximum peak inductor or the peak
input current. If the inductor ripple current is low, then
the maximum regulator input current will be very close to
the switch current limit ILIM. By estimating the conduction
SC2308A
Applications Information (Continued)
losses in both the switch and the diode, an expression
for the maximum available output current of a boost
converter can be derived using Equation (5):
IOUT (MAX )
ILIM VIN ª D VD DVD VCESAT º
» (5)
«1
VIN
VOUT ¬ 45
¼
Since switching losses are excluded in the derivation, the
actual output current is over-estimated in Equation (5).
Nevertheless, this calculation still provides a useful initial
approximation.
Inductor Selection
The inductor must be able to handle the peak current ILIM.
First, the inductor should not saturate at ILIM. Second, the
inductor needs to have low core loss at the switching frequency. Inductors with ferrite cores are preferrable. Moreover, the inductor should have low DCR for low copper
loss. The inductance can be selected such that the inductor
ripple current is between 20% to 40% of its average current
for improved efficiency.
The inductance can be calculated using Equation (6):
L
DVIN VCESAT 'IL fSW
(6)
The Coilcraft DO3316P series and the Sumida CDRH8D38NP
series inductors perform well in boost converters. The
inductors selected must be suitable for a 750kHz switching frequency.
Input Capacitor Selection
or polymer capacitors can be used for output filtering. In
a buck converter, the inductor ripple current flows in the
output capacitor, whereas in a boost converter, the output
capacitor current is the difference between the rectifying
diode current and the output current (Figure 4). This current is discontinuous with high current amplitudes. For this
reason, the output ripple voltage of a boost converter is
always higher than that of a buck converter with the same
inductor current and the same output capacitor.
If tantalum or polymer capacitors are used at the converter
output, then the converter output ripple voltage will be
primarily determined by the capacitor ESR, due to the relatively high ESR of these capacitors. The output voltage
ripple is the product of the peak inductor current and the
output capacitor ESR. For example, if two Sanyo 6TPG100M
(100mF, ESR=70mW) polymer capacitors are used for output
filtering, then the output peak-to-peak ripple voltage will
be 70mV, assuming a 2A peak inductor current.
Tantalum capacitor voltage derating is generally 50%.
Multi-layer ceramic capacitors, due to their extremely low
ESR (<5mΩ), are particularly well suited for output filtering.
It is worth noting that the output ripple voltage resulting
from charging and discharging of a 10μF or a 22mF ceramic
capacitor is higher than the ripple voltage resulting from
the capacitor ESR. The output ripple voltage due to charging
and discharging effects is calculated using the following
equation:
'VOUT
DIOUT
fSW C OUT
(7)
X5R and X7R ceramic capacitors are the preferred types.
The input current in a boost converter is the inductor current, which is continuous with low RMS current ripples.
A 2.2mF~4.7mF ceramic capacitor is adequate for most
applications. Use X5R or better ceramic capacitors, since
they have stable temperature and voltage coefficients. The
voltage rating for the input capacitor should exceed the
maximum input voltage by 10% to 25%. Murata and TDK
are two ceramic capacitor suppliers.
Output Capacitor Selection
Ceramic and low equivalent series resistance (ESR) tantalum
10
Rectifying Diode
For high efficiency, Schottky barrier diodes should be
used as rectifying diodes for the SC2308A. These diodes
should have an average forward current rating at least
equal to the output current. The reverse blocking voltage
of the Schottky diode should be derated by 10%-20% for
reliability. The Schottky diode used in a 12V output stepup converter should have a reverse voltage rating of at
least 15V (20% derating).
SC2308A
Applications Information (Continued)
SS22 and SS24 from ON Semiconductor and 10BQ020
and 10BQ040 from International Rectifier are widely used
Schottky diodes.
Figure 1 and the 3.3V to 5V step-up converter in Figure 10.
Notice that the regulator does not switch until the internal
SS voltage exceeds the FB voltage.
Soft-Start
If the input power supply to a step-up converter is turned
on with the EN and the IN pins shorted, then the start-up
waveforms will depend on the input voltage ramp rate and
the output load. The internal 3ms soft-start interval may be
insufficient to keep the input start-up current below the
switch current limit, especially with heavy loads and slow
VIN ramp. Figure 6 shows the start-up waveforms of the stepup converters in Figure 1 and Figure 10 when powering on
using the Agilent 6652A DC power supply. Before VIN rises
above the input start voltage, there is no switching and
the converter output simply follows VIN. When starting into
an 800mA constant-current load, the 5V to 12V converter
reaches the cycle-by-cycle current limit and the output
voltage ramp becomes non-linear {Figure 6(c)}. There is,
however, very little output voltage overshoot.
The SC2308A comprises an internal soft-start timer. The
output (SS) of the soft-start timer (see Figure 2), which forms
the second non-inverting input of the feedback amplifier,
is reset to zero before VIN rises above its turn-on threshold.
The SS voltage is subsequently charged from zero to the
nominal feedback voltage (1.22V) in about 3ms.
If a step-up converter is enabled by stepping the EN input
while connected to a live power supply, then its output
voltage will rise linearly from approximately VIN to its set
voltage. The current drawn from the input power supply
will be less than the switch current limit and there will be
no output overshoot during start-up. Figure 5 shows the
start-up waveforms of the 5V to 12V step-up converter in
VIN
5V/div
VIN
5V/div
VEN
2V/div
VEN
2V/div
VOUT
5V/div
IL1
0.5A/div
VOUT
5V/div
IL1
1A/div
2ms/div
2ms/div
(a)
(b)
IN_EN_OUT_IL_5V to 12V@800mA_EN Start
IN_EN_OUT_IL_5V to 12V@10mA_EN Start
VIN
2V/div
VEN
2V/div
VOUT
2V/div
IL1
1A/div
1ms/div
(c)
IN_EN_OUT_IL_3.3V to [email protected]_EN Start
Figure 5. Boost Converter Start-Up Waveforms. EN is Stepped with Input Applied.
(a) 5V to 12 V Step-Up Regulator (Figure 1), IOUT = 10mA
(b) 5V to 12 V Step-Up Regulator, IOUT = 800mA
(c) 3.3V to 5V Step-Up Regulator (Figure 10), IOUT = 1.1A
11
SC2308A
Applications Information (Continued)
VIN
5V/div
VIN
5V/div
VOUT
5V/div
VOUT
5V/div
IL1
0.5A/div
IL1
1A/div
2ms/div
2ms/div
(a)
(b)
IN_OUT_IL_5V to 12V@650mA_IN=EN Start
IN_OUT_IL_5V to 12V@10mA_IN=EN Start
VIN
5V/div
VIN
5V/div
VOUT
5V/div
VOUT
5V/div
IL1
1A/div
IL1
1A/div
(d)
4ms/div
2ms/div
(c)
(d)
IN_OUT_IL_3.3V to [email protected]_IN=EN Start
IN_OUT_IL_5V to 12V@800mA_IN=EN Start
Figure 6. Boost Converter Start-Up Waveforms. EN is Tied to IN and the Regulator is
Powered on Using the Agilent 6652A Power Supply.
(a) 5V to 12 V Step-Up Regulator (Figure 1), IOUT = 10mA
(b) 5V to 12 V Step-Up Regulator, IOUT = 650mA
(c) 5V to 12 V Step-Up Regulator, IOUT = 800mA
(d) 3.3V to 5V Step-Up Regulator (Figure 10), IOUT = 1.1A
Frequency Compensation
Figure 7 shows the simplified equivalent model of a boost
converter using the SC2308A.
Due to current-mode control, the double reactive poles
attributed to the inductor are reduced to a single real pole.
This pole results from the output capacitor and is at frequency:
fp 2
1
S RL C OUT
(8)
where RL is the equivalent output load resistance and COUT
is the output capacitance.
12
IOUT
VIN
POWER
STAGE
PWM
Modulator
COMP
R3
C4
C6
R1
Fi
Gm
-
VOUT
ESR
RL
COUT
FB
+
VREF
R2
1.22V
Figure 7. The Simplified Model of a Boost Converter
SC2308A
Applications Information (Continued)
The power stage also has a right half plane (RHP) zero at:
fz 2
1 D 2 RL
The poles p1, p2 and the RHP zero z2 all increase phase shift
in the loop response. For stable operation, the overall loop
gain should cross 0dB with -20dB/decade slope. Due to
the presence of the RHP zero, it is suggested that the 0dB
fz 2
crossover frequency should not be more than
.
3
(9)
2S L
The ESR zero frequency is:
fz 3
1
2S R C C OUT
A simple two-pole, single-zero compensator network is
adequate. The loop is compensated with R3, C4 and C6 from
the COMP pin to ground. The compensating zero z1 provides phase boost beyond p2. In general, the converter will
be more stable if the filter pole p2 and the RHP zero z2 are
widely separated. The RHP zero moves to low frequency
when either the duty cycle D or the output current IOUT
increases. It is beneficial to use small inductors and larger
output capacitors, especially when stepping up from low
VIN to high VOUT. An optional second pole can be placed at
the power stage ESR zero to attentuate any high-frequency
noise.
(10)
where RC is the ESR of the output capacitor.
R3 and C4 form a zero at:
fz1
1
2S R 3C 4
(11)
With the assumption that C4>>C6 , R3 and C6 also form a
pole p3 at frequency:
fp 3
1
2S R 3C 6
(12)
There is also a low-frequency integrator pole p1 formed
by C4 and the equivalent output resistance of the transconductance amplifier. The corresponding bode plots are
shown in Figure 8.
Thermal Shutdown
Thermal shutdown turns off the power switch and the
control circuit as the junction temperature exceeds 160°C.
Switching resumes when the junction temperature falls
by 12°C.
Gain (dB)
Crossover Frequency, fC
fz1
fp3
fp2
Power-Stage
Compensator
fp3,4
fz2
f
fz3
Loop Gain
Figure 8. Bode Magnitude Plots of the Power Stage, the
Compensator, and the Overall Loop Gain
13
SC2308A
Applications Information (Continued)
EN
C4
C6
VIN
R1
R2
C3
C2
C1
R3
In a boost converter, the main power switch, the rectifying
diode, and the output filter capacitor carry pulse currents
with high di/dt. For jitter-free operation, the size of the
loop formed by these components should be minimized.
The main power switch is integrated inside the SC2308A.
Therefore, the output capacitor should be connected close
to the device ground pin. Shortening the trace at the SW pin
reduces the parasitic trace inductance. This decreases voltage ringing at the SW node. The input capacitor should be
placed close to the input and the GND pins. Figure 9 shows
an example of external component placement around the
SC2308A.
GND
JP
Board Layout Considerations
pin1
L1
U1
D1
SW
VOUT
Figure 9. Suggested PCB Layout for the SC2308A
14
SC2308A
Typical Application Circuits
VIN
L
3.3V
10µH
IN
OFF ON
D1
VOUT
5V, 1.1A
SS22
R1
180k
SW
EN
C3
SC2308A
100µF
FB
C1
COMP
4.7µF
GND
R2
C4
220pF
C6
10pF
C2
100µF
59k
R3
200k
L: Coilcraft DO3316P103
C1: Murata GRM31CR61A475K
D1: ON SS22
C2,C3: Sanyo 6TPG100M
Figure 10. 3.3V to 5V Step-Up Converter
VIN
L
12V
22mH
IN
OFF ON
D1
36V, 0.4A
SS26
SW
EN
VOUT
R1
102k
SC2308A
C6
C5
C3
C2
10mF
10mF
10mF
10mF
FB
C1
COMP
4.7mF
C4
220pF
GND
R2
3.57k
R3
487k
L: Cooper CD1-220
C1: Murata GRM31CR61C475K
D1: ON SS26
C2,C3 ,C5,C6: Murata GRM31CR61H106K
Figure 11. 12V to 36V Step-Up Converter
15
SC2308A
Outline Drawing — SOIC-8
A
D
e
N
DIM
E1 E
1
2
ccc C
2X N/2 TIPS
e/2
B
D
aaa C
SEATING
PLANE
.053
.069
.004
.010
.049
.065
.012
.020
.007
.010
.189 .193 .197
.150 .154 .157
.236 BSC
.050 BSC
.010
.020
.016 .028 .041
(.041)
8
0°
8°
.004
.010
.008
A
A1
A2
b
c
D
E1
E
e
h
L
L1
N
01
aaa
bbb
ccc
2X E/2
DIMENSIONS
INCHES
MILLIMETERS
MIN NOM MAX MIN NOM MAX
h
A2 A
C
bxN
bbb
1.35
1.75
0.10
0.25
1.25
1.65
0.31
0.51
0.17
0.25
4.80 4.90 5.00
3.80 3.90 4.00
6.00 BSC
1.27 BSC
0.25
0.50
0.40 0.72 1.04
(1.04)
8
0°
8°
0.10
0.25
0.20
A1
h
H
C A-B D
c
GAGE
PLANE
0.25
SEE DETAIL
SIDE VIEW
L
(L1)
A
DETAIL
01
A
NOTES:
1.
CONTROLLING DIMENSIONS ARE IN MILLIMETERS (ANGLES IN DEGREES).
2. DATUMS -A- AND -B- TO BE DETERMINED AT DATUM PLANE -H3. DIMENSIONS "E1" AND "D" DO NOT INCLUDE MOLD FLASH, PROTRUSIONS
OR GATE BURRS.
4. REFERENCE JEDEC STD MS-012, VARIATION AA.
Land Pattern – SOIC-8
X
DIM
(C)
G
Z
Y
C
G
P
X
Y
Z
DIMENSIONS
INCHES
MILLIMETERS
(.205)
.118
.050
.024
.087
.291
P
NOTES:
1.
THIS LAND PATTERN IS FOR REFERENCE PURPOSES ONLY.
CONSULT YOUR MANUFACTURING GROUP TO ENSURE YOUR
COMPANY'S MANUFACTURING GUIDELINES ARE MET.
2. REFERENCE IPC-SM-782A, RLP NO. 300A.
16
(5.20)
3.00
1.27
0.60
2.20
7.40
SC2308A
© Semtech 2012
All rights reserved. Reproduction in whole or in part is prohibited without the prior written consent of the copyright
owner. The information presented in this document does not form part of any quotation or contract, is believed to be
accurate and reliable and may be changed without notice. No liability will be accepted by the publisher for any consequence of its use. Publication thereof does not convey nor imply any license under patent or other industrial or intellectual property rights. Semtech assumes no responsibility or liability whatsoever for any failure or unexpected operation
resulting from misuse, neglect improper installation, repair or improper handling or unusual physical or electrical stress
including, but not limited to, exposure to parameters beyond the specified maximum ratings or operation outside the
specified range.
SEMTECH PRODUCTS ARE NOT DESIGNED, INTENDED, AUTHORIZED OR WARRANTED TO BE SUITABLE FOR USE IN LIFESUPPORT APPLICATIONS, DEVICES OR SYSTEMS OR OTHER CRITICAL APPLICATIONS. INCLUSION OF SEMTECH PRODUCTS IN SUCH APPLICATIONS IS UNDERSTOOD TO BE UNDERTAKEN SOLELY AT THE CUSTOMER’S OWN RISK. Should a
customer purchase or use Semtech products for any such unauthorized application, the customer shall indemnify and
hold Semtech and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs damages and attorney fees which could arise.
Notice: All referenced brands, product names, service names and trademarks are the property of their respective
owners.
Contact Information
Semtech Corporation
Power Management Products Division
200 Flynn Road, Camarillo, CA 93012
Phone: (805) 498-2111 Fax: (805) 498-3804
www.semtech.com
17