QProx™ QT118H lQ Less expensive than many mechanical switches Projects a ‘touch button’ through any dielectric 100% autocal for life - no adjustments required No active external components Piezo sounder direct drive for ‘tactile’ click feedback LED drive for visual feedback 2.5 ~ 5V single supply operation 10µ µA at 2.5V - very low power drain Toggle mode for on/off control (via option pins) 10s or 60s auto-recalibration timeout (via option pins) Pulse output mode (via option pins) Gain settings in 3 discrete levels Simple 2-wire operation possible HeartBeat™ health indicator on output Pb-Free package Vdd 1 Out 2 Opt1 3 Opt2 4 QT118H CHARGE-TRANSFER TOUCH SENSOR 8 Vss 7 Sns2 6 Sns1 5 Gain APPLICATIONS Light switches Industrial panels Appliance control Security systems Access systems Pointing devices Elevator buttons Toys & games The QT118H charge-transfer (“QT’”) touch sensor is a self-contained digital IC capable of detecting near-proximity or touch. It will project a sense field through almost any dielectric, like glass, plastic, stone, ceramic, and wood. It can also turn small metal-bearing objects into intrinsic sensors, making them respond to proximity or touch. This capability coupled with an ability to self calibrate continuously can lead to entirely new product concepts. The device is designed specifically for human interfaces, like control panels, appliances, toys, lighting controls, or anywhere a mechanical switch or button may be found; it may also be used for some material sensing and control applications provided that the presence duration of objects does not exceed the recalibration timeout interval. A piezo element can also be connected to create a feedback click sound. The IC requires only a common inexpensive capacitor in order to function. Average power consumption is under 20µA in most applications, allowing battery operation. The QT118H employs digital signal processing techniques pioneered by Quantum, designed to make it survive real-world challenges, such as ‘stuck sensor’ conditions and signal drift. Sensitivity is digitally determined for the highest possible stability. No external active components are required for operation. The device includes several user-selectable built in features. One, toggle mode, permits on/off touch control, for example for light switch replacement. Another makes the sensor output a pulse instead of a DC level, which allows the device to 'talk' over the power rail, permitting a simple 2-wire twisted-pair interface. Quantum’s unique HeartBeat™ signal is also included, allowing a host controller to continuously monitor the health of the device. By using the charge transfer principle, the IC delivers a level of performance clearly superior to older technologies in a highly cost-effective package. TA 00C to +700C -400C to +850C lq AVAILABLE OPTIONS SOIC QT118H-ISG 8-PIN DIP QT118H-DG - ©1999-2004 Quantum Research Group R1.08 / 0405 1 - OVERVIEW Figure 1-1 Standard mode options The QT118H is a digital burst mode charge-transfer (QT) sensor designed specifically for touch controls; it includes all hardware and signal processing functions necessary to provide stable sensing under a wide variety of changing conditions. Only a few low cost, non-critical discrete external parts are required for operation. +2.5 ~ +5 1 2 Figure 1-1 shows the basic QT118H circuit using the device, with a conventional output drive and power supply connections. Figure 1-2 shows a second configuration using a common power/signal rail which can be a long twisted pair from a controller; this configuration uses the built-in pulse mode to transmit the output state to the host controller. 3 Vdd OUT SNS2 OPT1 GAIN RE 7 5 Rs Cs 4 OUTPUT = DC TIMEOUT = 10 Secs TOGGLE = OFF GAIN = HIGH 1.1 BASIC OPERATION The QT118H employs short, low duty cycle bursts of QT cycles to acquire capacitance. Burst mode permits power consumption in the low microamp range, dramatically reduces RF emissions, lowers susceptibility to EMI, and yet permits excellent response time. Internally the signals are digitally processed to reject impulse noise, using a 'consensus' filter which requires four consecutive confirmations of a detection before the output is activated. OPT2 SNS1 Vss SENSING ELECTRODE Cx 6 2nF - 500nF 8 Option pins allow the selection or alteration of several special features and sensitivity. 1.2 ELECTRODE DRIVE The internal ADC treats Cs as a floating transfer capacitor; as a direct result, the sense electrode can in theory be connected to either SNS1 or SNS2 with no performance difference. However, the noise immunity of the device is improved by connecting the electrode to SNS2, preferably via a series resistor Re (Figure 1-1) to roll off higher harmonic frequencies, both outbound and inbound. The QT switches and charge measurement hardware functions are all internal to the QT118H (Figure 1-3). A single-slope switched capacitor ADC includes both the required QT charge and transfer switches in a configuration that provides direct ADC conversion. The sensitivity depends on the values of Cs, Cx, and to a smaller degree, Vdd. Vdd is used as the charge reference voltage. In order to reduce power consumption and to assist in discharging Cs between acquisition bursts, a 470K series resistor Rs should always be connected across Cs (Figure 1-1). Higher values of Cs increase gain; higher values of Cx load reduce it. The value of Cs can thus be increased to allow larger values of Cx to be tolerated (Figures 4-1 and 4-2, page 10). The rule Cs >> Cx must be observed for proper operation. Normally Cx is on the order of 10pF or so, while Cs might be 10nF (10,000pF), or a ratio of about 1:1000. Piezo sounder drive: The QT118H can drive a piezo sounder after a detection for feedback. The piezo sounder replaces or augments the Cs capacitor; this works since piezo sounders are also capacitors, albeit with a large thermal drift coefficient. If Cpiezo is in the proper range, no additional capacitor. If Cpiezo is too small, it can simply be ‘topped up’ with a ceramic capacitor in parallel. The QT118H drives a ~4kHz signal across SNS1 and SNS2 to make the piezo (if installed) sound a short tone for 75ms immediately after detection, to act as an audible confirmation. It is important to minimize the amount of unnecessary stray capacitance Cx, for example by minimizing trace lengths and widths and backing off adjacent ground traces and planes so as keep gain high for a given value of Cs, and to allow for a larger sensing electrode size if so desired. The PCB traces, wiring, and any components associated with or in contact with SNS1 and SNS2 will become touch sensitive and should be treated with caution to limit the touch area to the desired location. Figure 1-2 2-wire operation, self-powered + 3.5 - 5.5V CMOS LOGIC 1K Twisted pair n-ch Mosfet 10µF 1 1N4148 2 Vdd OUT SNS2 7 3 OPT1 GAIN 5 SNS1 6 4 OPT2 RE SENSING ELECTRODE Cs Rs Cx Vss 8 lq 2 QT118H R1.08 / 0405 1.3 ELECTRODE DESIGN E LE C TRO DE R esult S tart S NS 2 Single-Slo pe 14-bit Switched Capacitor ADC There is no restriction on the shape of the electrode; in most cases common sense and a little experimentation can result in a good electrode design. The QT118H will operate equally well with long, thin electrodes as with round or square ones; even random shapes are acceptable. The electrode can also be a 3-dimensional surface or object. Sensitivity is related to electrode surface area, orientation with respect to the object being sensed, object composition, and the ground coupling quality of both the sensor circuit and the sensed object. Burst Controller 1.3.1 ELECTRODE GEOMETRY AND SIZE Figure 1-3 Internal Switching & Timing Do ne Cs Cx S NS 1 C ha rge Amp 1.3.2 KIRCHOFF’S CURRENT LAW Like all capacitance sensors, the QT118H relies on Kirchoff’s Current Law (Figure 1-5) to detect the change in capacitance of the electrode. This law as applied to capacitive sensing requires that the sensor’s field current must complete a loop, returning back to its source in order for capacitance to be sensed. Although most designers relate to Kirchoff’s law with regard to hardwired circuits, it applies equally to capacitive field flows. By implication it requires that the signal ground and the target object must both be coupled together in some manner for a capacitive sensor to operate properly. Note that there is no need to provide actual hardwired ground connections; capacitive coupling to ground (Cx1) is always sufficient, even if the coupling might seem very tenuous. For example, powering the sensor via an isolated transformer will provide ample ground coupling, since there is capacitance between the windings and/or the transformer core, and from the power wiring itself directly to 'local earth'. Even when battery powered, just the physical size of the PCB and the object into which the electronics is embedded will generally be enough to couple a few picofarads back to local earth. 1.3.3 VIRTUAL CAPACITIVE GROUNDS crumpled into a ball. Virtual ground planes are more effective and can be made smaller if they are physically bonded to other surfaces, for example a wall or floor. ‘Ground’ as applied to capacitive fields can also mean power wiring or signal lines. The capacitive sensor, being an AC device, needs only an AC ground return. 1.3.5 SENSITIVITY ADJUSTMENT 1.3.5.1 Gain Pin The QT118H can be set for one of 3 gain levels using option pin 5 (Table 1-1). This sensitivity change is made by altering the internal numerical threshold level required for a detection. Note that sensitivity is also a function of other things: like the values of Cs and Cx, electrode size, shape, and orientation, the composition and aspect of the object to be sensed, the thickness and composition of any overlaying panel material, and the degree of ground coupling of both sensor and object. The Gain input should never be connected to a pullup or pulldown resistor or tied to anything other than SNS1 or SNS2, or left unconnected (for high gain setting). When detecting human contact (e.g. a fingertip), grounding of the person is never required. The human body naturally has several hundred picofarads of ‘free space’ capacitance to the local environment (Cx3 in Figure 1-4), which is more than two orders of magnitude greater than that required to create a return path to the QT118H via earth. The QT118H's PCB however can be physically quite small, so there may be little ‘free space’ coupling (Cx1 in Figure 1-4) between it and the environment to complete the return path. If the QT118H circuit ground cannot be earth grounded by wire, for example via the supply connections, then a ‘virtual capacitive ground’ may be required to increase return coupling. Figure 1-4 Kirchoff's Current Law CX2 S e nse E le ctro de A ‘virtual capacitive ground’ can be created by connecting the QT118H’s own circuit ground to: S EN SO R - A nearby piece of metal or metallized housing; - A floating conductive ground plane; - Another electronic device (to which its might be connected already). CX 1 Free-floating ground planes such as metal foils should maximize exposed surface area in a flat plane if possible. A square of metal foil will have little effect if it is rolled up or lq Su rro und in g e nv iro nm en t 3 CX3 QT118H R1.08 / 0405 a slow rate, but only while there is no detection in effect. The rate of adjustment must be performed slowly, otherwise legitimate detections could be ignored. The QT118H drift compensates using a slew-rate limited change to the reference level; the threshold and hysteresis values are slaved to this reference. Table 1-1 Gain Strap Options Gain High Medium Low Tie Pin 5 to: Leave open Pin 6 Pin 7 1.3.5.2 Changing Cs, Cx The values of Cs and Cx have a dramatic effect on sensitivity, and Cs can be easily increased in value to improve gain. Sensitivity is directly proportional to Cs and inversely proportional to Cx: S= k$C S CX Where ‘k’ depends on a variety of factors including the gain pin setting (see prior section), Vdd, etc. Sensitivity plots are shown in Figures 4-1 and 4-2, page 10. 1.3.5.3 Electrode / Panel Adjustments Sensitivity can often be increased by using a bigger electrode, or reducing overlying panel thickness. Increasing electrode size can have a diminishing effect on gain, as the attendant higher values of Cx will start to reduce sensor gain. Also, increasing the electrode's surface area will not substantially increase touch sensitivity if its diameter is already much larger in surface area than the object being detected. The panel or other intervening material can be made thinner, but again there are diminishing rewards for doing so. Panel material can also be changed to one having a higher dielectric constant, which will help propagate the field through to the front. Locally adding some conductive material to the panel (conductive materials essentially have an infinite dielectric constant) will also help; for example, adding carbon or metal fibers to a plastic panel will greatly increase frontal field strength, even if the fiber density is too low to make the plastic bulk-conductive. 1.3.5.3 Ground Planes Grounds around and under the electrode and its SNS trace will cause high Cx loading and destroy gain. The possible signal-to-noise ratio benefits of ground area are more than negated by the decreased gain from the circuit, and so ground areas around electrodes are discouraged. Keep ground, power, and other signals traces away from the electrodes and SNS wiring 2 - QT118H SPECIFICS Once an object is sensed, the drift compensation mechanism ceases since the signal is legitimately high, and therefore should not cause the reference level to change. The QT118H's drift compensation is 'asymmetric': the reference level drift-compensates in one direction faster than it does in the other. Specifically, it compensates faster for decreasing signals than for increasing signals. Increasing signals should not be compensated for quickly, since an approaching finger could be compensated for partially or entirely before even touching the sense pad. However, an obstruction over the sense pad, for which the sensor has already made full allowance for, could suddenly be removed leaving the sensor with an artificially elevated reference level and thus become insensitive to touch. In this latter case, the sensor will compensate for the object's removal very quickly, usually in only a few seconds. 2.1.2 THRESHOLD AND HYSTERESIS The internal signal threshold level can be set to one of three settings (Table 1-1). These are fixed with respect to the internal reference level, which in turn moves in accordance with the drift compensation mechanism. The QT118H employs a hysteresis dropout below the threshold level of 17% of the delta between the reference and threshold levels. 2.1.3 MAX ON-DURATION If an object or material obstructs the sense pad the signal may rise enough to create a detection, preventing further operation. To prevent this, the sensor includes a timer which monitors detections. If a detection exceeds the timer setting, the timer causes the sensor to perform a full recalibration. This is known as the Max On-Duration feature. After the Max On-Duration interval, the sensor will once again function normally, even if partially or fully obstructed, to the best of its ability given electrode conditions. There are two timeout durations available via strap option: 10 and 60 seconds. 2.1.4 DETECTION INTEGRATOR It is desirable to suppress detections generated by electrical noise or from quick brushes with an object. To accomplish this, the QT118H incorporates a detect integration counter 2.1 SIGNAL PROCESSING Figure 2-1 Drift Compensation The QT118H digitally processes all signals using a number of algorithms pioneered by Quantum. The algorithms are specifically designed to provide for high survivability in the face of all kinds of adverse environmental changes. 2.1.1 DRIFT COMPENSATION ALGORITHM S ig na l T hre sh old R efer ence Signal drift can occur because of changes in Cx and Cs over time. It is crucial that drift be compensated for, otherwise false detections, non-detections, and sensitivity shifts will follow. Drift compensation (Figure 2-1) is performed by making the reference level track the raw signal at lq H ys te res is Ou tpu t 4 QT118H R1.08 / 0405 that increments with each detection until a limit is reached, after which the output is activated. If no detection is sensed prior to the final count, the counter is reset immediately to zero. The required count is 4. The Detection Integrator can also be viewed as a 'consensus' filter, that requires four detections in four successive bursts to create an output. As the basic burst spacing is 95ms, if this spacing was maintained through 4 consecutive bursts the sensor would be very slow to respond. In the QT118H, after an initial detection is sensed, the remaining three bursts are spaced only about 2ms apart, so that the slowest reaction time possible is the fastest possible. 2.1.5 FORCED SENSOR RECALIBRATION The QT118H has no recalibration pin; a forced recalibration is accomplished only when the device is powered up. However, the supply drain is so low it is a simple matter to treat the entire IC as a controllable load; simply driving the QT118H's Vdd pin directly from another logic gate or a microprocessor port (Figure 2-2) will serve as both power and 'forced recal'. The source resistance of most CMOS gates and microprocessors is low enough to provide direct power without any problems. Almost any CMOS logic gate can directly power the QT118H. A 0.01uF minimum bypass capacitor close to the device is essential; without it the device can break into high frequency oscillation. Max On-Duration expires, whichever occurs first. If the latter occurs first, the sensor performs a full recalibration and the output becomes inactive until the next detection. In this mode, two nominal Max On-Duration timeouts are available: 10 and 60 seconds. 2.2.2 TOGGLE MODE OUTPUT This makes the sensor respond in an on/off mode like a flip flop. It is most useful for controlling power loads, for example in kitchen appliances, power tools, light switches, etc. Max On-Duration in Toggle mode is fixed at 10 seconds. When a timeout occurs, the sensor recalibrates but leaves the output state unchanged. Table 2-1 Output Mode Strap Options Tie Pin 3 to: Tie Pin 4 to: Max OnDuration DC Out Vdd Vdd 10s DC Out Vdd Gnd 60s Toggle Gnd Gnd 10s Pulse Gnd Vdd 10s 2.2.3 PULSE MODE OUTPUT Option strap configurations are read by the QT118H only on powerup. Configurations can only be changed by powering the QT118H down and back up again; a microcontroller can directly alter most of the configurations and cycle power to put them in effect. This generates a positive pulse of 95ms duration with every new detection. It is most useful for 2-wire operation (see Figure 1-2), but can also be used when bussing together several devices onto a common output line with the help of steering diodes or logic gates, in order to control a common load from several places. 2.2 OUTPUT FEATURES Max On-Duration is fixed at 10 seconds if in Pulse output mode. The QT118H is designed for maximum flexibility and can accommodate most popular sensing requirements. These are selectable using strap options on pins OPT1 and OPT2. All options are shown in Table 2-1. OPT1 and OPT2 should never be left floating. If they are floated, the device will draw excess power and the options will not be properly read on powerup. Intentionally, there are no pullup resistors on these lines, since pullup resistors add to power drain if the pin(s) are tied low. 2.2.1 DC MODE OUTPUT The output of the device can respond in a ‘DC mode’, where the output is active-high upon detection. The output will remain active for the duration of the detection, or until the Figure 2-2 Powering From a CMOS Port Pin P ORT X. m 0.01µ F CMOS m ic ro c o n tr o lle r Vdd P ORT X. n OUT Q T11 8 The piezo beeper drive does not operate in Pulse mode. 2.2.4 HEARTBEAT™ OUTPUT The output has a full-time HeartBeat™ ‘health’ indicator superimposed on it. This operates by taking 'Out' into a tri-state mode for 350µs once before every QT burst. This output state can be used to determine that the sensor is operating properly, or, it can be ignored using one of several simple methods. Since Out is normally low, a pullup resistor will create positive HeartBeat pulses (Figure 2-3) when the sensor is not detecting an object; when detecting an object, the output will remain active for the duration of the detection, and no HeartBeat pulse will be evident. If the sensor is wired to a microcontroller as shown in Figure 2-4, the controller can reconfigure the load resistor to either ground or Vcc depending on the output state of the device, so that the pulses are evident in either state. Electromechanical devices will usually ignore this short pulse. The pulse also has too low a duty cycle to visibly activate LED’s. It can be filtered completely if desired, by adding an RC timeconstant to filter the output, or if interfacing directly and only to a high-impedance CMOS input, by doing nothing or at most adding a small non-critical capacitor from Out to ground (Figure 2-5). V ss lq 5 QT118H R1.08 / 0405 Figure 2-3 Figure 2-4 Getting HB pulses with a pullup resistor when not active Using a micro to obtain HB pulses in either output state 2.2.5 PIEZO ACOUSTIC DRIVE A piezo drive signal is generated for use with a piezo sounder immediately after a detection is made; the tone lasts for a nominal 95ms to create a ‘tactile feedback’ sound. The sensor drives the piezo using an H-bridge configuration for the highest possible sound level. The piezo is connected across pins SNS1 and SNS2 in place of Cs or in addition to a parallel Cs capacitor. The piezo sounder should be selected to have a peak acoustic output in the 3.5kHz to 4.5kHz region. Since piezo sounders are merely high-K ceramic capacitors, the sounder will double as the Cs capacitor, and the piezo's metal disc can even act as the sensing electrode. Piezo transducer capacitances typically range from 6nF to 30nF in value; at the lower end of this range an additional capacitor should be added to bring the total Cs across SNS1 and SNS2 to at least 10nF, or possibly more if Cx is above 5pF. Piezo sounders have very high, uncharacterized thermal coefficients and should not be used if fast temperature swings are anticipated, especially at high gains. They are also generally unstable at high gains; even if the total value of Cs is largely from an added capacitor the piezo can cause periodic false detections. The burst acquisition process induces a small but audible voltage step across the piezo resonator, which occurs when SNS1 and SNS2 rapidly discharge residual voltage stored on the resonator. The resulting slight clicking sound can be greatly reduced by placing a 470K resistor Rs in parallel with the resonator; this acts to slowly discharge the resonator, attenuating of the harmonic-rich audible step (Figure 2-6). Note that the piezo drive does not operate in Pulse mode. 2.2.6 OUTPUT DRIVE The QT118H’s output is active high and it can source or sink 1mA of non-inductive current. Care should be taken when the IC and the load are both powered from the same supply, and the supply is minimally regulated. The device derives its internal references from the power supply, and sensitivity shifts can occur with changes in Vdd, as happens when loads are switched on. This can induce detection ‘cycling’, whereby an object is detected, the load is turned on, the supply sags, the detection is no longer sensed, the load is turned off, the supply rises and the object is reacquired, ad infinitum. To prevent this occurrence, the output should only be lightly loaded if the device is operated from an unregulated supply, e.g. batteries. Detection ‘stiction’, the opposite effect, can occur if a load is shed when Out is active. 3 - CIRCUIT GUIDELINES 3.1 SAMPLE CAPACITOR When used for most applications, the charge sampler Cs can be virtually any plastic film or good quality ceramic capacitor. The type should be relatively stable in the anticipated Figure 2-5 Eliminating HB Pulses Figure 2-6 Piezo Sounder Circuit +2.5 ~ +5 GATE O R M ICR O IN P UT 1 OUT S NS 2 OPT 1 GA IN 2 7 Co 1 00pF 3 4 lq OPT 2 S NS 1 3 5 4 6 Vdd OUT SNS1 OPT1 GAIN OPT2 SNS2 RE 7 5 6 Piezo Sounder 10-30nF 2 C M OS Rs SENSING ELECTRODE Cx Vss 8 6 QT118H R1.08 / 0405 temperature range. If fast temperature swings are expected, especially with higher sensitivities, more stable capacitors be required, for example PPS film. In most moderate gain applications (ie in most cases), low-cost X7R types will work fine. 3.2 ELECTRODE WIRING See also Section 3.4. The wiring of the electrode and its connecting trace is important to achieving high signal levels and low noise. Certain design rules should be adhered to for best results: 1. Use a ground plane under the IC itself and Cs and Rs but NOT under Re, or under or closely around the electrode or its connecting trace. Keep ground away from these things to reduce stray loading (which will dramatically reduce sensitivity). 2. Keep Cs, Rs, and Re very close to the IC. 3. Make Re as large as possible. As a test, check to be sure that an increase of Re by 50% does not appreciably decrease sensitivity; if it does, reduce Re until the 50% test increase has a negligible effect on sensitivity. Without this capacitor the part can break into high frequency oscillation, get physically hot, stop working, or become damaged. PCB Cleanliness: All capacitive sensors should be treated as highly sensitive circuits which can be influenced by stray conductive leakage paths. QT devices have a basic resolution in the femtofarad range; in this region, there is no such thing as ‘no clean flux’. Flux absorbs moisture and becomes conductive between solder joints, causing signal drift and resultant false detections or temporary loss of sensitivity. Conformal coatings can trap existing amounts of moisture which will then become highly temperature sensitive. The designer should strongly consider ultrasonic cleaning as part of the manufacturing process, and in more extreme cases, the use of conformal coatings after cleaning and baking. 3.3.1 SUPPLY CURRENT Measuring average power consumption is a challenging task due to the burst nature of the device’s operation. Even a good quality RMS DMM will have difficulty tracking the relatively slow burst rate, and will show erratic readings. 4. Do not route the sense wire near other ‘live’ traces containing repetitive switching signals; the trace will pick up noise from external signals. The easiest way to measure Idd is to put a very large capacitor, such as 2,700µF across the power pins, and put a 220 ohm resistor from there back to the power source. Measure the voltage across the 220 resistor with a DMM and compute the current based on Ohm’s law. This circuit will 3.3 POWER SUPPLY, PCB LAYOUT The power supply can range from 2.5 to 5.0 volts. At 2.5 volts average out current to provide a much smoother reading. current drain averages less than 10µA with Cs = 10nF, To reduce the current consumption the most, use high or low provided a 470K Rs resistor is used (Figure 1-1). Sample Idd gain pin settings only, the smallest value of Cs possible that curves are shown in Figure 4-3. works, and a 470K resistor (Rs) across Cs (Figure 1-1). Rs Higher values of Cs will raise current drain. Higher Cx values can actually decrease power drain. Operation can be from batteries, but be cautious about loads causing supply droop (see Output Drive, Section 2.2.6) if the batteries are unregulated. As battery voltage sags with use or fluctuates slowly with temperature, the IC will track and compensate for these changes automatically with only minor changes in sensitivity. If the power supply is shared with another electronic system, care should be taken to assure that the supply is free of digital spikes, sags, and surges which can adversely affect the device. The IC will track slow changes in Vdd, but it can be affected by rapid voltage steps. if desired, the supply can be regulated using a conventional low current regulator, for example CMOS LDO regulators that have nanoamp quiescent currents. Care should be taken that the regulator does not have a minimum load specification, which almost certainly will be violated by the QT118's low current requirement. Furthermore, some LDO regulators are unable to provide adequate transient regulation between the quiescent and acquire states, creating Vdd disturbances that will interfere with the acquisition process. This can usually be solved by adding a small extra load from Vdd to ground, such as 10K ohms, to provide a minimum load on the regulator. Conventional non-LDO type regulators are usually more stable than slow, low power CMOS LDO types. Consult the regulator manufacturer for recommendations. For proper operation a 100nF (0.1uF) ceramic bypass capacitor must be used between Vdd and Vss; the bypass cap should be placed very close to the device’s power pins. lq acts to help discharge capacitor Cs between bursts, and its presence substantially reduces power consumption. 3.3.2 ESD PROTECTION In cases where the electrode is placed behind a dielectric panel, the IC will be protected from direct static discharge. However even with a panel transients can still flow into the electrode via induction, or in extreme cases via dielectric breakdown. Porous materials may allow a spark to tunnel right through the material. Testing is required to reveal any problems. The device has diode protection on its terminals which will absorb and protect the device from most ESD events; the usefulness of the internal clamping will depending on the dielectric properties, panel thickness, and rise time of the ESD transients. The best method available to suppress ESD and RFI is to insert a series resistor Re in series with the electrode as shown in Figure 1-1. The value should be the largest that does not affect sensing performance. If Re is too high, the gain of the sensor will decrease. Because the charge and transfer times of the QT118 are relatively long (~2µs), the circuit can tolerate a large value of Re, often more than 10k ohms in most cases. Diodes or semiconductor transient protection devices or MOV's on the electrode trace are not advised; these devices have extremely large amounts of nonlinear parasitic capacitance which will swamp the capacitance of the electrode and cause false detections and other forms of instability. Diodes also act as RF detectors and will cause serious RF immunity problems. 7 QT118H R1.08 / 0405 3.4 EMC AND RELATED NOISE ISSUES External AC fields (EMI) due to RF transmitters or electrical noise sources can cause false detections or unexplained shifts in sensitivity. In brief summary, the following design rules should be adhered to for best ESD and EMC results: 1. Use only SMT components. 2. Keep Cs, Rs, Re and Vdd bypass cap close to the IC. The influence of external fields on the sensor is reduced by means of the Rseries described in Section 3.2. The Cs capacitor and Rseries (see Figure 1-1) form a natural low-pass filter for incoming RF signals; the roll-off frequency of this network is defined by 1 F R = 2✜R series Cs 3. Maximize Re to the limit where sensitivity is not noticeably affected. 4. Do not place the electrode or its connecting trace near other traces, or near a ground plane. 5. Do use a ground plane under and around the QT118 itself, back to the regulator and power connector (but not beyond the Cs capacitor). If for example Cs = 22nF, and Rseries = 10K ohms, the rolloff frequency to EMI is 723Hz, vastly lower than any credible external noise source (except for mains frequencies i.e. 50 / 60 Hz). However, Rseries and Cs must both be placed very close to the body of the IC so that the lead lengths between them and the IC do not form an unfiltered antenna at very high frequencies. 6. Do not place an electrode (or its wiring) of one QT device near the electrode or wiring of another device, to prevent cross interference. 7. Keep the electrode (and its wiring) away from other traces carrying AC or switched signals. 8. If there are LEDs or LED wiring near the electrode or its wiring (ie for backlighting of the key), bypass the LED wiring to ground on both its ends. PCB layout, grounding, and the structure of the input circuitry have a great bearing on the success of a design to withstand electromagnetic fields and be relatively noise-free. 9. Use a voltage regulator just for the QT118 to eliminate power noise coupling from other switching sources. Make sure the regulator’s transient load stability provides for stable voltage just before each burst commences. For further tips on construction, PCB design, and EMC issues browse the application notes and faq at www.qprox.com lq 8 QT118H R1.08 / 0405 4.1 ABSOLUTE MAXIMUM SPECIFICATIONS Operating temp. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . as designated by suffix Storage temp. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -55OC to +125OC VDD. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.5 to +6.5V Max continuous pin current, any control or drive pin. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±20mA Short circuit duration to ground, any pin. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . infinite Short circuit duration to VDD, any pin. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . infinite Voltage forced onto any pin. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.6V to (Vdd + 0.6) Volts 4.2 RECOMMENDED OPERATING CONDITIONS VDD. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +2.5 to 5.0V Short-term supply ripple+noise. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±5mV Long-term supply stability. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±100mV Cs value. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10nF to 500nF Cx value. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0 to 100pF Rs value. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 470K✡ 4.3 AC SPECIFICATIONS Vdd = 3.0, Cs = 10nF, Rs = 470K, Cx = 10pF, Gain = High, Ta = 20OC, unless otherwise noted. Parameter Description Min Typ Max 550 Units Notes TRC Recalibration time TQ Charge, transfer duration 2 ms µs TBS Burst spacing interval 75 95 ms ms @ 5.0V Vdd @ 3.3V Vdd TBL Burst length ms Depends on Cs, Cx TR Response time FP Piezo drive frequency TP Piezo drive duration 75 ms TPO Pulse output width on Out 75 ms 0.5 50 129 3.6 ms 4 4.4 kHz THB Heartbeat pulse width 300 µs FQ Burst frequency 165 kHz 4.4 SIGNAL PROCESSING Vdd = 3.0, Cs = 10nF, Rs = 470K, Cx = 10pF, Gain = High, Ta = 20OC, unless otherwise noted. Description Min Typ Threshold differential Hysteresis Detect integrator filter length Positive drift compensation rate Negative drift compensation rate Post-detection recalibration timer duration (typical min/max) Max Notes 6, 12, or 24 counts 1 17 % 2 4 samples 750 ms/level 4 ms/level 4 75 10 Units 60 secs 3, 4 Note 1: Pin options Note 2: Percentage of signal threshold Note 3: Pin option Note 4: Cs, Cx dependent lq 9 QT118H R1.08 / 0405 4.5 DC SPECIFICATIONS Vdd = 3.0, Cs = 10nF, Rs = 470K, Cx = 10pF, Gain = High, Ta = 20OC Unless otherwise noted. Parameter Description Min Typ Max Units VDDL Guaranteed min Vdd 2.45 V -I suffix VDDL Guaranteed min Vdd 2.95 V -E suffix µA @ Vdd = 5.0V @ Vdd = 3.3V @ Vdd = 2.5V V/s Required for proper startup IDD Supply current VDDS 30 10 8 Supply turn-on slope VIL Low input logic level VHL High input logic level VOL Low output voltage VOH 100 0.8 High output voltage IIL Input leakage current AR Acquisition resolution S Sensitivity range V OPT1, OPT2 V OPT1, OPT2 0.6 V OUT, 4mA sink V OUT, 1mA source ±1 µA OPT1, OPT2 14 bits 28 fF 2.2 Vdd-0.7 9 1,000 Figure 4-1 - Typical Threshold Sensitivity vs. Cx, High Gain, at Selected Values of Cs; Vdd = 3.0 Note 2 Figure 4-2 - Typical Threshold Sensitivity vs. Cx, Medium Gain, Selected Values of Cs; Vdd = 3.0 10.00 10.00 1.00 Detection Threshold, pF Detection Threshold, pF Notes 10nF 20nF 50nF 100nF 200nF 500nF 0.10 0.01 0 10 20 30 1.00 10nF 20nF 50nF 100nF 200nF 500nF 0.10 0.01 40 0 Cx Load, pF 10 20 30 40 Cx Load, pF Figure 4-3 Typical Supply Current Vs Vdd Rs = 470K, Cx = 10pF, Gain = High 40 Idd, Microamperes 35 30 Cs = 20nF 25 20 . .. 15 Cs = 10nF 10 5 2.5 3 3.5 4 4.5 5 Vdd lq 10 QT118H R1.08 / 0405 4.6 MECHANICAL Package type: 8pin Dual-In-Line SYMBOL Min Millimeters Max a A M m Q P L L1 F R r S S1 Aa x Y 6.096 7.62 9.017 7.62 0.889 0.254 0.355 1.397 2.489 3.048 0.381 3.048 7.62 8.128 0.203 7.112 8.255 10.922 7.62 0.559 1.651 2.591 3.81 3.556 4.064 7.062 9.906 0.381 Notes Typical BSC Typical BSC Min Inches Max 0.24 0.3 0.355 0.3 0.035 0.01 0.014 0.055 0.098 0.12 0.015 0.12 0.3 0.32 0.008 0.28 0.325 0.43 0.3 0.022 0.065 0.102 0.15 0.14 0.16 0.3 0.39 0.015 Notes Typical BSC Typical BSC Package type: 8pin SOIC SYMBOL Min Millimeters Max M W Aa H h D L E e ß Ø 4.800 5.816 3.81 1.371 0.101 1.27 0.355 0.508 0.19 0.381 0º 4.979 6.198 3.988 1.728 0.762 1.27 0.483 1.016 0.249 0.762 8º Notes Min Inches Max BSC 0.189 0.229 0.15 0.054 0.004 0.050 0.014 0.02 0.007 0.229 0º 0.196 0.244 0.157 0.068 0.01 0.05 0.019 0.04 0.01 0.03 8º Notes BSC 5 - ORDERING INFORMATION PART TEMP RANGE QT118H-D 0 - 70C QT118H-ISG -40 - 85C lq PACKAGE PDIP Pb-Free SO-8 Pb-Free 11 MARKING QT118H-G QT1 + T + G or QT118H-IG QT118H R1.08 / 0405 6 - SOIC MARKING DIAGRAMS VERSION ‘A’ Lot Code Lot code (last letter varies) ©QT1 T F 0214HB6.G QPROX C 'G' ending indicates Pb-free package Pin 1 Dimple VERSION ‘B’ Lot Code QT118H-IG 0214HB6C 'G' ending indicates Pb-free package ©QPROX Pin 1 Dimple lq 12 QT118H R1.08 / 0405 NOTES lq 13 QT118H R1.08 / 0405 lQ Copyright © 1999 - 2004 QRG Ltd. All rights reserved. Patented and patents pending Corporate Headquarters 1 Mitchell Point Ensign Way, Hamble SO31 4RF Great Britain Tel: +44 (0)23 8056 5600 Fax: +44 (0)23 80565600 [email protected] www.qprox.com North America 651 Holiday Drive Bldg. 5 / 300 Pittsburgh, PA 15220 USA Tel: 412-391-7367 Fax: 412-291-1015 This device covered under one or more of the following United States and international patents: 5,730,165, 6,288,707, 6,377,009, 6,452,514, 6,457,355, 6,466,036, 6,535,200. Numerous further patents are pending which may apply to this device or the applications thereof. The specifications set out in this document are subject to change without notice. All products sold and services supplied by QRG are subject to our Terms and Conditions of sale and supply of services which are available online at www.qprox.com and are supplied with every order acknowledgement. QProx, QTouch, QMatrix, QLevel, and QSlide are trademarks of QRG. QRG products are not suitable for medical (including lifesaving equipment), safety or mission critical applications or other similar purposes. Except as expressly set out in QRG's Terms and Conditions, no licenses to patents or other intellectual property of QRG (express or implied) are granted by QRG in connection with the sale of QRG products or provision of QRG services. QRG will not be liable for customer product design and customers are entirely responsible for their products and applications which incorporate QRG's products.