LTC5584 30MHz to 1.4GHz IQ Demodulator with IIP2 and DC Offset Control Description Features I/Q Bandwidth of 530MHz or Higher n High IIP3: 31dBm at 450MHz, 28dBm at 900MHz n High IIP2: 70dBm at 450MHz, 65dBm at 900MHz n User Adjustable IIP2 to >80dBm n User Adjustable DC Offset Null n High Input P1dB: 13.1dBm at 900MHz n Image Rejection: 45dB at 900MHz n Noise Figure: 9.9dB at 450MHz 10dB at 900MHz n Conversion Gain: 5.4dB at 450MHz 5.7dB at 900MHz n Shutdown Mode n Operating Temperature Range (T ): –40°C to 105°C C n 24-Lead 4mm × 4mm QFN Package The LTC®5584 is a direct conversion quadrature demodulator optimized for high linearity receiver applications in the 30MHz to 1.4GHz frequency range. It is also usable in the 10MHz to 30MHz and 1.4GHz to 2GHz ranges with reduced performance. It is suitable for communications receivers where an RF signal is directly converted into I and Q baseband signals with bandwidth of 530MHz or higher. The LTC5584 incorporates balanced I and Q mixers, LO buffer amplifiers and a precision, high frequency quadrature phase shifter. In addition, the LTC5584 provides four analog control voltage interface pins for IIP2 and DC offset correction, greatly simplifying system calibration. n The high linearity of the LTC5584 provides excellent spurfree dynamic range for the receiver. This direct conversion demodulator can eliminate the need for intermediate frequency (IF) signal processing, as well as the corresponding requirements for image filtering and IF filtering. These I/Q outputs can interface directly to channel-select filters (LPFs) or to baseband amplifiers. Applications n n n n n LTE/W-CDMA/TD-SCDMA Base Station Receivers Wideband DPD Receivers Point-To-Point Broadband Radios High Linearity Direct Conversion I/Q Receivers Image Rejection Receivers L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Typical Application Direct Conversion Receiver with IIP2 and DC Offset Calibration 5V LNA BPF RF INPUT VCC RF+ I+ LPF VGA I– IIP2 vs IP2I, IP2Q Trim Voltage A/D 130 RF– IP2 AND DC OFFSET CAL LO INPUT LO IP2 ADJUST 120 D/A 100 LTC5584 IP2 AND DC OFFSET CAL D/A DC OFFSET EN Q+ Q– LPF Q, –40°C Q, 25°C Q, 85°C Q, 105°C fRF = 450MHz 90 80 70 IP2 ADJUST D/A ENABLE I, –40°C I, 25°C I, 85°C I, 105°C 110 DC OFFSET 0° 90° D/A IIP2 (dBm) BPF 60 50 VGA A/D 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 IP2I, IP2Q (V) 5584 TA01b 5584 TA01a 5584f 1 LTC5584 Pin Configuration VCC Supply Voltage.................................... –0.3V to 5.5V VCAP Voltage..................................................VCC ±0.05V I–, I+, Q+, Q –, CMI, CMQ Voltage.........2.5V to VCC + 0.3V Voltage on Any Other Pin..................–0.3V to VCC + 0.3V LO+, LO –, RF+, RF – Input Power............................20dBm RF+, RF – Input DC Voltage......................... –0.3V to 2.7V Maximum Junction Temperature (TJMAX).............. 150°C Operating Temperature Range (TC) (Note 3)................................................... –40°C to 105°C Storage Temperature Range................... –65°C to 150°C CMI Q– Q+ I– I+ REF TOP VIEW 24 23 22 21 20 19 IP2Q 1 18 CMQ DCOQ 2 17 VCAP DCOI 3 IP2I 4 RF + 16 LO– 25 GND 15 LO+ 5 14 GND RF– 6 13 GND EIP2 EDC 9 10 11 12 VCC 8 VBIAS 7 EN (Note 1) GND Absolute Maximum Ratings UF PACKAGE 24-LEAD (4mm × 4mm) PLASTIC QFN TJMAX = 150°C, θJC = 7°C/W EXPOSED PAD (PIN 25) IS GND, MUST BE SOLDERED TO PCB Order Information LEAD FREE FINISH TAPE AND REEL PART MARKING PACKAGE DESCRIPTION TEMPERATURE RANGE LTC5584IUF#PBF LTC5584IUF#TRPBF 5584 24-Lead (4mm x 4mm) Plastic QFN –40°C to 105°C Consult LTC Marketing for parts specified with wider operating temperature ranges. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ Electrical Characteristics C = 25°C, VCC = 5V, EN = 5V, EDC = EIP2 = 0V, REF = IP2I = IP2Q = DCOI = DCOQ T = 0.5V, PRF = –5dBm (–5dBm/tone for 2-tone IIP2 and IIP3 tests), PLO = 6dBm, unless otherwise noted. (Notes 2, 3, 5, 6, 9) SYMBOL PARAMETER CONDITIONS RF Input Frequency Range (Note 12) fRF(RANGE) LO Input Frequency Range (Note 12) fLO(RANGE) LO Input Power Range (Note 12) PLO(RANGE) fRF1 = 140MHz, fRF2 = 141MHz, fLO = 130MHz, L6 = 68nH, C19 = 8.0pF, L5 = 82nH RF Input Frequency Range Return Loss > 10dB fRF(MATCH) LO Input Frequency Range Return Loss > 10dB fLO(MATCH) Voltage Conversion Gain Loaded with 100Ω Pull-Up (Note 8) GV NF Noise Figure Double-Side Band (Note 4) Noise Figure Under Blocking Conditions Double-Side Band, PRF = 0dBm (Note 7) NFBLOCKING IIP3 Input 3rd Order Intercept IIP2 Input 2nd Order Intercept Unadjusted, EIP2 = 0V Optimized Input 2nd Order Intercept EIP2 = 5V, IP2I, IP2Q Adjusted for Minimum IM2 IIP2OPT P1dB Input 1dB Compression DC Offset at I/Q Outputs Unadjusted, EDC = 0V (Note 13) DCOFFSET I/Q Gain Mismatch ∆G ∆φ I/Q Phase Mismatch MIN TYP 30 to 1400 30 to 1400 0 to 10 MAX UNITS MHz MHz dBm 95 to 190 105 to 180 5.7 9.9 15.5 33 70 80 12 1.5 0.02 MHz MHz dB dB dB dBm dBm dBm dBm mV dB 0.2 Deg 5584f 2 LTC5584 Electrical Characteristics C = 25°C, VCC = 5V, EN = 5V, EDC = EIP2 = 0V, REF = IP2I = IP2Q = DCOI = DCOQ T = 0.5V, PRF = –5dBm (–5dBm/tone for 2-tone IIP2 and IIP3 tests), PLO = 6dBm, unless otherwise noted. (Notes 2, 3, 5, 6, 9) SYMBOL PARAMETER CONDITIONS IRR Image Rejection Ratio (Note 10) LO-RF LO to RF Leakage RF-LO RF to LO Isolation fRF1 = 450MHz, fRF2 = 451MHz, fLO = 440MHz, L6 = 15nH, C19 = 1.0pF, L5 = 12nH, C14 = 4.0pF RF Input Frequency Range Return Loss > 10dB fRF(MATCH) LO Input Frequency Range Return Loss > 10dB fLO(MATCH) Voltage Conversion Gain Loaded with 100Ω Pull-Up (Note 8) GV NF Noise Figure Double-Side Band (Note 4) Noise Figure Under Blocking Conditions Double-Side Band, PRF = 0dBm (Note 7) NFBLOCKING IIP3 Input 3rd Order Intercept IIP2 Input 2nd Order Intercept Unadjusted, EIP2 = 0V Optimized Input 2nd Order Intercept EIP2 = 5V, IP2I, IP2Q Adjusted for Minimum IM2 IIP2OPT P1dB Input 1dB Compression DC Offset at I/Q Outputs Unadjusted, EDC = 0V (Note 13) DCOFFSET I/Q Gain Mismatch ∆G MIN I/Q Phase Mismatch ∆φ IRR Image Rejection Ratio (Note 10) LO-RF LO to RF Leakage RF-LO RF to LO Isolation fRF1 = 900MHz, fRF2 = 901MHz, fLO = 940MHz, C17 = 1.5pF, L6 = 5.6nH, C13 = 2.2pF, L5 = 3.9nH RF Input Frequency Range Return Loss > 10dB fRF(MATCH) LO Input Frequency Range Return Loss > 10dB fLO(MATCH) Voltage Conversion Gain Loaded with 100Ω Pull-Up (Note 8) GV NF Noise Figure Double-Side Band (Note 4) Noise Figure Under Blocking Conditions Double-Side Band, PRF = 0dBm (Note 7) NFBLOCKING IIP3 Input 3rd Order Intercept IIP2 Input 2nd Order Intercept Unadjusted, EIP2 = 0V Optimized Input 2nd Order Intercept EIP2 = 5V, IP2I, IP2Q Adjusted for Minimum IM2 IIP2OPT P1dB Input 1dB Compression DC Offset at I/Q Outputs Unadjusted, EDC = 0V (Note 13) DCOFFSET I/Q Gain Mismatch ∆G I/Q Phase Mismatch ∆φ IRR Image Rejection Ratio LO-RF LO to RF Leakage RF-LO RF to LO Isolation Power Supply and Other Parameters Supply Voltage VCC Supply Current ICC Supply Current ICC(LOW) Shutdown Current ICC(OFF) Turn-On Time tON Turn-Off Time tOFF EN, EDC, EIP2 Input High Voltage (On) VEH EN, EDC, EIP2 Input Low Voltage (Off) VEL (Note 10) EDC = EIP2 = VCC EDC = EIP2 = 0V EN < 0.3V EN Transition from Logic Low to High (Note 14) EN Transition from Logic High to Low (Note 15) 4.75 180 174 TYP 53 –85 74 MAX UNITS dB dBm dB 300 to 600 310 to 590 5.4 9.9 13.6 31 70 80 12.6 2 0.02 MHz MHz dB dB dB dBm dBm dBm dBm mV dB 0.25 Deg 52 –80 70 dB dBm dB 630 to 1200 470 to 1100 5.7 10 14.7 28 65 80 13.1 2.5 0.01 MHz MHz dB dB dB dBm dBm dBm dBm mV dB 0.7 Deg 45 –75 65 dB dBm dB 5.0 200 194 11 0.2 0.8 5.25 220 214 900 2.0 0.3 V mA mA μA µs µs V V 5584f 3 LTC5584 Electrical Characteristics C = 25°C, VCC = 5V, EN = 5V, EDC = EIP2 = 0V, REF = IP2I = IP2Q = DCOI = DCOQ T = 0.5V, PRF = –5dBm (–5dBm/tone for 2-tone IIP2 and IIP3 tests), PLO = 6dBm, unless otherwise noted. (Notes 2, 3, 5, 6, 9) SYMBOL IENH IEDCH IEIP2H VREF VREF(RANGE) ZREF VCM ZOUT BWBB PARAMETER EN Pin Input Current EDC Pin Input Current EIP2 Pin Input Current REF Pin Voltage REF Pin Voltage Range REF Input Impedance DCOI, DCOQ, IP2I, IP2Q Pin Voltage DCOI, DCOQ, IP2I, IP2Q Voltage Range DCOI, DCOQ, IP2I, IP2Q Impedance DCOI, DCOQ, IP2I, IP2Q Settling Time DC Offset Adjustment Range CONDITIONS EN = 5.0V EDC = 5.0V EIP2 = 5.0V With REF Pin Unloaded When Driven with External Source (Note 11) Unloaded When Driven with External Source (Note 11) For Step Input, Output with 90% of Final Value DCOI, DCOQ Swept from 0V to 1V, EDC = 5V DC Offset Drift Over Temperature I+, I–, Q+, Q– Common Mode Voltage I+, I–, Q+, Q– Output Impedance I+, I–, Q+, Q– Output Bandwidth Unadjusted, EDC = 0V Single Ended 100Ω External Pull-Up, –3dB Corner Frequency Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: Tests are performed with the test circuit of Figure 1. Note 3: The LTC5584 is guaranteed to be functional over the –40°C to 105°C case temperature operating range. Note 4: DSB noise figure is measured at the baseband frequency of 15MHz with a small-signal noise source without any filtering on the RF input and no other RF signal applied. Note 5: Performance at the RF frequencies listed is measured with external RF and LO impedance matching, as shown in the table of Figure 1. Note 6: The complementary outputs (I+, I– and Q+, Q–) are combined using a 180° phase-shift combiner. Note 7: Noise figure under blocking conditions (NFBLOCKING) is measured at an output noise frequency of 60MHz with an RF input blocking signal at fLO + 1MHz. Both RF and LO input signals are appropriately filtered, as well as the baseband output. NFBLOCKING measured at fLO of 160MHz, 460MHz and 885MHz. MIN TYP 52 33 50 0.5 0.4 to 0.7 2||1 0.5 0 to 2VREF 8||1 20 ±20 20 VCC – 1.5 100||6 530 MAX UNITS μA μA μA V V kΩ||pF V V kΩ||pF ns mV μV/°C V Ω||pF MHz Note 8: Voltage conversion gain is calculated from the average measured power conversion gain of the I and Q outputs using the test circuit shown in Figure 1. Power conversion gain is measured with a 100Ω differential load impedance on the I and Q outputs. Note 9: Baseband outputs have a 100Ω external pull-up resistor to VCC as shown in the test circuit shown in Figure 1. Note 10: Image rejection is calculated from the measured gain error and phase error using the method listed in the appendix. Note 11: The DCOI, DCOQ, IP2I, IP2Q pins have an 8k internal resistor to ground. The REF pin has a 2k internal resistor to ground. If unconnected, these pins will float up to 500mV through internal current sources. A low output resistance voltage source is recommended for driving these pins. Note 12: This is the recommended operating range, operation outside the listed range is possible with degraded performance to some parameters. Note 13: DC offset measured differentially between I+ and I– and between Q+ and Q–. The reported value is the mean of the absolute values of the characterization data distribution. Note 14: Baseband amplitude is within 10% of final value. Note 15: Baseband amplitude is at least 30dB down from its on state. 5584f 4 LTC5584 DC Performance Characteristics EN = 5V, EDC = 0V and EIP2 = 0V. Test circuit shown in Figure 1 Supply Current vs Supply Voltage 260 TC = –40°C TC = 25°C TC = 85°C TC = 105°C 250 240 230 540 220 210 200 190 535 530 525 520 515 180 510 170 505 160 4.75 VCC = 4.75V VCC = 5V VCC = 5.25V 545 REF VOLTAGE (mV) SUPPLY CURRENT (mA) REF Voltage vs Temperature 550 500 –40 5.25 5 SUPPLY VOLTAGE (V) –20 40 20 0 60 TEMPERATURE (°C) 80 100 5584 G02 5585 G01 Typical Performance Characteristics 140MHz application. VCC = 5V, EN = 5V, EDC = 0V, EIP2 = 0V, REF = 0.5V, TC = 25°C, PLO = 6dBm, fLO = 130MHz, fRF1 = 140MHz, fRF2 = 141MHz, fBB = 10MHz, PRF1 = PRF2 = –5dBm, DC Blocks and Mini-Circuits PSCJ-2-1 180° combiner at baseband outputs de-embedded from measurement unless otherwise noted. Test circuit with RF and LO ports impedance matched as in Figure 1. IIP3 and P1dB vs Supply Voltage (VCC) IIP3 and P1dB vs Temperature (TC) IIP3, P1dB (dBm) 40 I, –40°C I, 25°C I, 85°C I, 105°C Q, –40°C Q, 25°C Q, 85°C Q, 105°C 45 30 25 20 TC = 25°C 45 35 30 25 100 120 140 160 180 200 LO FREQUENCY (MHz) 10 80 TC = 25°C 35 30 25 15 P1dB 100 120 140 160 180 200 LO FREQUENCY (MHz) 5584 G03 Q, 0dBm Q, 6dBm Q, 10dBm 20 15 P1dB I, 0dBm I, 6dBm I, 10dBm 40 IIP3 20 15 10 80 Q, 4.75V Q, 5.0V Q, 5.25V 40 IIP3 35 I, 4.75V I, 5.0V I, 5.25V IIP3 vs LO Power 50 IIP3 (dBm) 45 50 IIP3, P1dB (dBm) 50 10 80 100 120 140 160 180 200 LO FREQUENCY (MHz) 5584 G04 5584 G05 5584f 5 LTC5584 Typical Performance Characteristics 140MHz application. VCC = 5V, EN = 5V, EDC = 0V, EIP2 = 0V, REF = 0.5V, TC = 25°C, PLO = 6dBm, fLO = 130MHz, fRF1 = 140MHz, fRF2 = 141MHz, fBB = 10MHz, PRF1 = PRF2 = –5dBm, DC Blocks and Mini-Circuits PSCJ-2-1 180° combiner at baseband outputs de-embedded from measurement unless otherwise noted. Test circuit with RF and LO ports impedance matched as in Figure 1. 2-Tone IIP3 vs RF Power TC = 25°C fRF1 = 140MHz fRF2 = 141MHz fLO = 130MHz 45 130 Q I IIP2 (dBm) IIP3 (dBm) 35 30 25 –8 –6 Q, –40°C Q, 25°C Q, 85°C Q, 105°C –2 –4 0 RF POWER (dBm) 2 90 120 70 70 80 100 160 140 120 LO FREQUENCY (MHz) 60 200 180 110 fRF = 140MHz EIP2 = 5V TC = 25°C fRF1 = 140MHz fLO = 130MHz 100 110 I (UNCALIBRATED) I (NULLED AT 1MHz) Q (UNCALIBRATED) Q (NULLED AT 1MHz) 180 200 Q I TC = 25°C fLO = 140MHz 100 90 IIP2 (dBm) 80 160 140 120 LO FREQUENCY (MHz) 2x2 Half-IF IIP2 vs RF to LO Tone Spacing 90 90 100 5584 G08 IIP2 vs RF Tone Spacing 100 80 5584 G07 IIP2 (dBm) IIP2 (dBm) 110 Q, –40°C Q, 25°C Q, 85°C Q, 105°C 90 80 IIP2 vs IP2I, IP2Q Trim Voltage I, –40°C I, 25°C I, 85°C I, 105°C TC = 25°C 100 80 5584 G06 130 Q, 0dBm Q, 6dBm Q, 10dBm 110 100 60 4 I, 0dBm I, 6dBm I, 10dBm 120 110 40 20 –10 I, –40°C I, 25°C I, 85°C I, 105°C 120 Uncalibrated IIP2 vs LO Power 130 IIP2 (dBm) 50 Uncalibrated IIP2 vs Temperature (TC) 80 70 80 70 70 60 60 50 0 50 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 IP2I, IP2Q (V) 60 ZFSCJ-2-1 BB COMBINER 1 10 100 RF TONE SPACING (MHz) 18 16 Noise Figure and Conversion Gain vs LO Power 20 Q, –40°C Q, 25°C Q, 85°C Q, 105°C 16 Q, 0dBm Q, 6dBm Q, 10dBm 12 10 8 GAIN 6 12 8 16 14 13 4 12 2 11 0 0 140 160 120 LO FREQUENCY (MHz) 180 200 5584 G12 80 100 140 160 120 LO FREQUENCY (MHz) 180 200 5584 G13 TC = 25°C fRF = 160MHz fLO = 159MHz fNOISE = 60MHz EIP2 = 5V 15 2 100 Q, –20dBm Q, 0dBm 17 4 80 I, –20dBm I, 0dBm 19 18 GAIN 6 20 TC = 25°C NF 10 1000 Noise Figure vs RF Power and IP2I, IP2Q Trim Voltage 14 NF, GAIN (dB) 14 NF, GAIN (dB) I, 0dBm I, 6dBm I, 10dBm 18 NF 10 100 RF TO LO SPACING (MHz) 5584 G11 DSB NOISE FIGURE (dB) Noise Figure and Conversion Gain vs Temperature (TC) I, –40°C I, 25°C I, 85°C I, 105°C 1 5584 G10 5585 G09 20 50 1000 ZFSCJ-2-1 BB COMBINER 10 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 IP2I, IP2Q TRIM VOLTAGE (V) 5584 G14 5584f 6 LTC5584 Typical Performance Characteristics 140MHz application. VCC = 5V, EN = 5V, EDC = 0V, EIP2 = 0V, REF = 0.5V, TC = 25°C, PLO = 6dBm, fLO = 130MHz, fRF1 = 140MHz, fRF2 = 141MHz, fBB = 10MHz, PRF1 = PRF2 = –5dBm, DC Blocks and Mini-Circuits PSCJ-2-1 180° combiner at baseband outputs de-embedded from measurement unless otherwise noted. Test circuit with RF and LO ports impedance matched as in Figure 1. DC Offset vs Temperature (TC) 10 10 9 8 7 6 5 4 3 2 1 0 –1 –2 –3 –4 –5 I, –40°C I, 25°C I, 85°C I, 105°C 80 Q, –40°C Q, 25°C Q, 85°C Q, 105°C 100 160 140 180 120 LO FREQUENCY (MHz) L-R, –40°C L-R, 25°C L-R, 85°C L-R, 105°C 40 20 10 0 –2.5 –1.5 0.5 –0.5 DC OFFSET (mV) 200 DC Offset Distribution, Q-Side 30 1.5 40 fLO = 100MHz 30 20 10 0 2.5 TC = –40°C TC = 25°C TC = 85°C TC = 105°C –2.5 –1.5 0.5 –0.5 DC OFFSET (mV) 5584 G19 1.5 2.5 5584 G20 Image Rejection vs Temperature 100 R-L, –40°C R-L, 25°C R-L, 85°C R-L, 105°C 90 –50 –60 –70 –80 –90 –100 TC = 25°C 160 140 180 120 LO FREQUENCY (MHz) 100 90 fLO = 100MHz IMAGE REJECTION (dB) –40 80 Q, 0dBm Q, 6dBm Q, 10dBm 5584 G17 PERCENTAGE DISTRIBUTION (%) PERCENTAGE DISTRIBUTION (%) 0.8 0.9 1.0 TC = –40°C TC = 25°C TC = 85°C TC = 105°C LO to RF Leakage and RF to LO Isolation –30 I, 0dBm I, 6dBm I, 10dBm DC Offset Distribution, I-Side 90 5584 G18 LEAKAGE (dBm), –ISOLATION (dB) DC OFFSET (mV) Q, –40°C Q, 25°C Q, 85°C Q, 105°C 200 10 9 8 7 6 5 4 3 2 1 0 –1 –2 –3 –4 –5 5584 G16 5584 G15 DC Offset vs DCOI, DCOQ Control Voltage 40 I, –40°C f = 140MHz 35 LO I, 25°C EDC = 5V I, 85°C 30 I, 105°C 25 20 15 10 5 0 –5 –10 –15 –20 –25 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 DCOI, DCOQ (V) DC Offset vs LO Power DC OFFSET (mV) DC OFFSET (mV) DSB NOISE FIGURE (dB) Noise Figure vs RF Input Power 24 PLO = 0dBm 23 PLO = 6dBm 22 PLO = 10dBm 21 T = 25°C C 20 fLO = 159MHz 19 fRF = 160MHz 18 fNOISE = 60MHz 17 16 15 14 13 12 11 10 9 0 –20 –5 5 –15 –10 RF INPUT POWER (dBm) 80 TC = –40°C TC = 25°C TC = 85°C TC = 105°C 70 60 50 40 30 80 100 160 140 120 LO FREQUENCY (MHz) 180 200 20 80 100 120 140 160 180 200 LO FREQUENCY (MHz) 5584 G21 5584 G22 5584f 7 LTC5584 Typical Performance Characteristics 450MHz application. VCC = 5V, EN = 5V, EDC = 0V, REF = 0.5V, EIP2 = 0V, TC = 25°C, PLO = 6dBm, fLO = 440MHz, fRF1 = 450MHz, fRF2 = 451MHz, fBB = 10MHz, PRF1 = PRF2 = –5dBm, DC Blocks and Mini-Circuits PSCJ-2-1 180° combiner at baseband outputs de-embedded from measurement unless otherwise noted. Test circuit with RF and LO ports impedance matched as in Figure 1. IIP3 and P1dB vs Supply Voltage (VCC) IIP3 and P1dB vs Temperature (TC) IIP3, P1dB (dBm) 40 Q, –40°C Q, 25°C Q, 85°C Q, 105°C 35 I, 4.75V I, 5.0V I, 5.25V 45 Q, 4.75V Q, 5.0V Q, 5.25V TC = 25°C IIP3 30 25 35 IIP3 30 25 10 200 300 400 600 500 LO FREQUENCY (MHz) P1dB 10 200 700 300 400 600 500 LO FREQUENCY (MHz) TC = 25°C fRF1 = 450MHz fRF2 = 451MHz fLO = 440MHz 130 Q I 25 –2 –4 0 RF POWER (dBm) 2 120 110 Q, –40°C Q, 25°C Q, 85°C Q, 105°C 90 TC = 25°C 90 80 70 70 300 600 500 400 LO FREQUENCY (MHz) 60 200 700 300 600 500 400 LO FREQUENCY (MHz) 110 TC = 25°C fRF1 = 450MHz fLO = 440MHz 100 700 5584 G28 2x2 Half-IF IIP2 vs RF to LO Tone Spacing IIP2 vs RF Tone Spacing 110 I (UNCALIBRATED) I (NULLED AT 1MHz) Q (UNCALIBRATED) Q (NULLED AT 1MHz) TC = 25°C fLO = 450MHz 100 Q I 90 IIP2 (dBm) IIP2 (dBm) 80 Q, 0dBm Q, 6dBm Q, 10dBm 100 80 90 90 I, 0dBm I, 6dBm I, 10dBm 120 5584 G27 fRF = 450MHz EIP2 = 5V 700 110 60 200 4 100 400 600 500 LO FREQUENCY (MHz) Uncalibrated IIP2 vs LO Power 130 100 IIP2 vs IP2I, IP2Q Trim Voltage I, –40°C I, 25°C I, 85°C I, 105°C 300 5584 G25 Q, –40°C Q, 25°C Q, 85°C Q, 105°C 5584 G26 130 10 200 700 IIP2 (dBm) IIP2 (dBm) IIP3 (dBm) 30 –6 25 15 110 35 –8 I, –40°C I, 25°C I, 85°C I, 105°C 120 40 20 –10 30 Uncalibrated IIP2 vs Temperature (TC) 2-Tone IIP3 vs RF Power 45 35 5584 G24 5584 G23 50 TC = 25°C 20 15 P1dB Q, 0dBm Q, 6dBm Q, 10dBm 40 20 15 I, 0dBm I, 6dBm I, 10dBm 45 40 20 IIP2 (dBm) IIP3 vs LO Power 50 IIP3 (dBm) I, –40°C I, 25°C I, 85°C I, 105°C 45 50 IIP3, P1dB (dBm) 50 80 70 80 70 70 60 60 50 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 IP2I, IP2Q (V) 5584 G29 50 60 ZFSCJ-2-1 BB COMBINER 1 10 100 RF TONE SPACING (MHz) 1000 5584 G30 50 ZFSCJ-2-1 BB COMBINER 1 10 100 RF TO LO SPACING (MHz) 1000 5584 G31 5584f 8 LTC5584 Typical Performance Characteristics 450MHz application. VCC = 5V, EN = 5V, EDC = 0V, REF = 0.5V, EIP2 = 0V, TC = 25°C, PLO = 6dBm, fLO = 440MHz, fRF1 = 450MHz, fRF2 = 451MHz, fBB = 10MHz, PRF1 = PRF2 = –5dBm, DC Blocks and Mini-Circuits PSCJ-2-1 180° combiner at baseband outputs de-embedded from measurement unless otherwise noted. Test circuit with RF and LO ports impedance matched as in Figure 1. Noise Figure and Conversion Gain vs Temperature (TC) Noise Figure vs RF Input Power 19 17 I, –40°C I, 25°C I, 85°C I, 105°C 18 16 11 –15 16 –10 –5 0 RF INPUT POWER (dBm) 5 10 8 GAIN 2 2 300 500 600 400 LO FREQUENCY (MHz) –2.5 300 600 500 400 LO FREQUENCY (MHz) 700 40 35 30 25 20 15 10 5 0 –5 –10 –15 –20 –25 I, –40°C I, 25°C I, 85°C I, 105°C Q, –40°C Q, 25°C Q, 85°C Q, 105°C 0 L-R, –40°C L-R, 25°C L-R, 85°C L-R, 105°C TC = 25°C 2.5 0 300 600 500 400 LO FREQUENCY (MHz) 5584 G36 700 5584 G37 Image Rejection vs Temperature 100 R-L, –40°C R-L, 25°C R-L, 85°C R-L, 105°C 90 –50 –60 –70 –80 –90 –100 200 Q, 0dBm Q, 6dBm Q, 10dBm 5.0 –5.0 200 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 DCOI, DCOQ (V) LO to RF Leakage and RF to LO Isolation –40 7.5 I, 0dBm I, 6dBm I, 10dBm –2.5 5584 G35 –30 700 DC Offset vs LO Power 10.0 fLO = 450MHz EDC = 5V IMAGE REJECTION (dB) –5.0 200 500 600 400 LO FREQUENCY (MHz) 5584 G34 DC OFFSET (mV) DC OFFSET (mV) 0 300 5584 G33 Q, –40°C Q, 25°C Q, 85°C Q, 105°C 2.5 GAIN 0 200 700 DC Offset vs DCOI, DCOQ Control Voltage 5.0 LEAKAGE (dBm), –ISOLATION (dB) DC OFFSET (mV) 7.5 8 4 DC Offset vs Temperature (TC) I, –40°C I, 25°C I, 85°C I, 105°C TC = 25°C NF 10 6 5584 G32 10.0 12 4 0 200 10 Q, 0dBm Q, 6dBm Q, 10dBm 14 NF 12 6 13 I, 0dBm I, 6dBm I, 10dBm 18 14 15 9 –20 20 Q, –40°C Q, 25°C Q, 85°C Q, 105°C NF, GAIN (dB) DSB NOISE FIGURE (dB) 21 20 PLO = 0dBm PLO = 6dBm PLO = 10dBm TC = 25°C fLO = 460MHz fRF = 461MHz fNOISE = 60MHz NF, GAIN (dB) 23 Noise Figure and Conversion Gain vs LO Power 80 TC = –40°C TC = 25°C TC = 85°C TC = 105°C 70 60 50 40 30 300 600 500 400 LO FREQUENCY (MHz) 700 20 200 300 400 500 600 700 LO FREQUENCY (MHz) 5584 G38 5584 G39 5584f 9 LTC5584 Typical Performance Characteristics 900MHz application. VCC = 5V, EN = 5V, EDC = 0V, EIP2 = 0V, REF = 0.5V, TC = 25°C, PLO = 6dBm, fLO = 890MHz, fRF1 = 900MHz, fRF2 = 901MHz, fBB = 10MHz, PRF1 = PRF2 = –5dBm, DC Blocks and Mini-Circuits PSCJ-2-1 180° combiner at baseband outputs de-embedded from measurement unless otherwise noted. Test circuit with RF and LO ports impedance matched as in Figure 1. IIP3 and P1dB vs Supply Voltage (VCC) IIP3 and P1dB vs Temperature (TC) IIP3, P1dB (dBm) 40 Q, –40°C Q, 25°C Q, 85°C Q, 105°C Q, 4.75V Q, 5.0V Q, 5.25V TC = 25°C IIP3 25 P1dB 15 10 400 600 35 IIP3 30 25 1200 1000 LO FREQUENCY (MHz) P1dB 10 400 1400 600 800 1200 1000 LO FREQUENCY (MHz) TC = 25°C fRF1 = 900MHz fRF2 = 901MHz fLO = 890MHz 120 Q I 25 –2 –4 0 RF POWER (dBm) 2 120 80 80 TC = 25°C 80 70 60 60 600 1200 1000 800 LO FREQUENCY (MHz) 50 400 1400 600 1200 1000 800 LO FREQUENCY (MHz) 2x2 Half-IF IIP2 vs RF to LO Tone Spacing IIP2 vs RF Tone Spacing 110 TC = 25°C fRF1 = 900MHz fLO = 890MHz 100 1400 5584 G45 110 I (UNCALIBRATED) I (NULLED AT 1MHz) Q (UNCALIBRATED) Q (NULLED AT 1MHz) TC = 25°C fLO = 900MHz 100 Q I 90 IIP2 (dBm) 90 Q, 0dBm Q, 6dBm Q, 10dBm 90 70 90 100 I, 0dBm I, 6dBm I, 10dBm 110 5584 G44 fRF = 900MHz EIP2 = 5V 1400 100 50 400 4 IIP2 (dBm) IIP2 (dBm) 110 Q, –40°C Q, 25°C Q, 85°C Q, 105°C 800 1200 1000 LO FREQUENCY (MHz) Uncalibrated IIP2 vs LO Power 120 90 IIP2 vs IP2I, IP2Q Trim Voltage I, –40°C I, 25°C I, 85°C I, 105°C 600 5584 G42 Q, –40°C Q, 25°C Q, 85°C Q, 105°C 5584 G43 130 10 400 1400 IIP2 (dBm) IIP2 (dBm) IIP3 (dBm) 30 –6 25 15 100 35 –8 I, –40°C I, 25°C I, 85°C I, 105°C 110 40 20 –10 30 Uncalibrated IIP2 vs Temperature (TC) 2-Tone IIP3 vs RF Power 45 35 5584 G41 5584 G40 50 TC = 25°C 20 15 800 Q, 0dBm Q, 6dBm Q, 10dBm 40 20 20 I, 0dBm I, 6dBm I, 10dBm 45 40 35 30 I, 4.75V I, 5.0V I, 5.25V 45 IIP3 vs LO Power 50 IIP3 (dBm) I, –40°C I, 25°C I, 85°C I, 105°C 45 50 IIP3, P1dB (dBm) 50 80 70 80 70 70 60 60 50 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 IP2I, IP2Q (V) 5585 G46 50 60 ZFSCJ-2-1 BB COMBINER 1 10 100 RF TONE SPACING (MHz) 1000 5584 G47 50 ZFSCJ-2-1 BB COMBINER 1 10 100 RF TO LO SPACING (MHz) 1000 5584 G48 5584f 10 LTC5584 Typical Performance Characteristics 900MHz application. VCC = 5V, EN = 5V, EDC = 0V, EIP2 = 0V, REF = 0.5V, TC = 25°C, PLO = 6dBm, fLO = 890MHz, fRF1 = 900MHz, fRF2 = 901MHz, fBB = 10MHz, PRF1 = PRF2 = –5dBm, DC Blocks and Mini-Circuits PSCJ-2-1 180° combiner at baseband outputs de-embedded from measurement unless otherwise noted. Test circuit with RF and LO ports impedance matched as in Figure 1. Noise Figure and Conversion Gain vs Temperature (TC) Noise Figure vs RF Input Power 19 17 I, –40°C I, 25°C I, 85°C I, 105°C 18 16 11 –15 16 –10 –5 0 RF INPUT POWER (dBm) 5 10 8 GAIN 4 2 600 1000 1200 800 LO FREQUENCY (MHz) 10.0 Q, –40°C Q, 25°C Q, 85°C Q, 105°C I, 0dBm I, 6dBm I, 10dBm 7.5 0 600 1200 1000 800 LO FREQUENCY (MHz) Q, 0dBm Q, 6dBm Q, 10dBm TC = 25°C 5.0 2.5 0 –5.0 400 1400 600 1200 1000 800 LO FREQUENCY (MHz) 5584 G52 R-L, –40°C R-L, 25°C R-L, 85°C R-L, 105°C 90 –50 –60 –70 –80 –90 –100 400 80 TC = –40°C TC = 25°C TC = 85°C TC = 105°C 70 60 50 40 30 600 1200 1000 800 LO FREQUENCY (MHz) 1400 20 400 600 800 1000 1200 1400 LO FREQUENCY (MHz) 5584 G55 I, –40°C I, 25°C I, 85°C I, 105°C 0 Q, –40°C Q, 25°C Q, 85°C Q, 105°C fLO = 900MHz EDC = 5V 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 DCOI, DCOQ (V) Conversion Gain Distribution 70 PERCENTAGE DISTRIBUTION (%) L-R, –40°C L-R, 25°C L-R, 85°C L-R, 105°C 1400 5584 G54 Image Rejection vs Temperature 100 IMAGE REJECTION (dB) –40 1400 40 35 30 25 20 15 10 5 0 –5 –10 –15 –20 –25 5584 G53 LO to RF Leakage and RF to LO Isolation –30 1000 1200 800 LO FREQUENCY (MHz) DC Offset vs DCOI, DCOQ Control Voltage –2.5 –5.0 400 600 5584 G51 DC OFFSET (mV) 2.5 –2.5 LEAKAGE (dBm), –ISOLATION (dB) 0 400 1400 DC Offset vs LO Power 5.0 GAIN 5584 G50 DC OFFSET (mV) DC OFFSET (mV) 7.5 NF 8 2 DC Offset vs Temperature (TC) I, –40°C I, 25°C I, 85°C I, 105°C TC = 25°C 10 6 5584 G49 10.0 12 4 0 400 10 Q, 0dBm Q, 6dBm Q, 10dBm 14 NF 12 6 13 I, 0dBm I, 6dBm I, 10dBm 18 14 15 9 –20 20 Q, –40°C Q, 25°C Q, 85°C Q, 105°C NF, GAIN (dB) DSB NOISE FIGURE (dB) 21 20 PLO = 0dBm PLO = 6dBm PLO = 10dBm TC = 25°C fLO = 884MHz fRF = 885MHz fNOISE = 60MHz NF, GAIN (dB) 23 Noise Figure and Conversion Gain vs LO Power 5584 G56 TC = –40°C TC = 25°C TC = 85°C TC = 105°C 60 50 40 30 20 10 0 4.7 4.9 5.1 5.3 5.5 5.7 CONVERSION GAIN (dB) 5.9 5584 G57 5584f 11 LTC5584 Typical Performance Characteristics 900MHz application. VCC = 5V, EN = 5V, EDC = 0V, EIP2 = 0V, REF = 0.5V, TC = 25°C, PLO = 6dBm, fLO = 890MHz, fRF1 = 900MHz, fRF2 = 901MHz, fBB = 10MHz, PRF1 = PRF2 = –5dBm, DC Blocks and Mini-Circuits PSCJ-2-1 180° combiner at baseband outputs de-embedded from measurement unless otherwise noted. Test circuit with RF and LO ports impedance matched as in Figure 1. 70 60 50 40 30 20 10 70 60 50 40 30 20 10 0 26.5 27 27.5 28 28.5 29 29.5 30 IIP3 (dBm) 70 TC = –40°C TC = 25°C TC = 85°C TC = 105°C 27 27.5 28 28.5 29 29.5 TC = 85°C TC = 105°C 40 30 20 10 9 9.4 9.8 10.2 80 70 60 50 40 30 20 0 10.6 69 69.5 70 70.5 71 IIP2 (dBm) 71.5 5584 G61 90 50 40 30 20 10 –0.04 –0.02 80 70 0.02 0.04 0 GAIN ERROR (dB) 0.06 5584 G64 70 60 50 TC = –40°C TC = 25°C TC = 85°C TC = 105°C 50 40 30 20 0 72 64.5 65.5 67.5 66.5 IIP2 (dBm) 68.5 5584 G63 50 TC = –40°C TC = 25°C TC = 85°C TC = 105°C 30 20 10 0.4 0.5 0.6 0.7 0.8 0.9 PHASE ERROR (DEGREES) 69.5 Image Rejection Distribution (Note 10) 40 0 11 60 Phase Error Distribution PERCENTAGE DISTRIBUTION (%) PERCENTAGE DISTRIBUTION (%) 60 0 100 5584 G62 Gain Error Distribution TC = –40°C TC = 25°C TC = 85°C TC = 105°C 10.2 10.6 9.4 9.8 DSB NOISE FIGURE (dB) 9 10 DSB NOISE FIGURE (dB) 70 10 5584 G60 10 0 20 IIP2 Distribution, Q-Side TC = –40°C TC = 25°C TC = 85°C TC = 105°C 90 50 30 IIP2 Distribution, I-Side 100 PERCENTAGE DISTRIBUTION (%) PERCENTAGE DISTRIBUTION (%) TC = –40°C TC = 25°C 40 5584 G59 DSB Noise Figure Distribution, Q-Side 60 50 IIP3 (dBm) 5584 G58 70 TC = –40°C TC = 25°C TC = 85°C TC = 105°C 60 0 30 PERCENTAGE DISTRIBUTION (%) 0 80 PERCENTAGE DISTRIBUTION (%) TC = –40°C TC = 25°C TC = 85°C TC = 105°C PERCENTAGE DISTRIBUTION (%) PERCENTAGE DISTRIBUTION (%) 80 DSB Noise Figure Distribution, I-Side IIP3 Distribution, Q-Side PERCENTAGE DISTRIBUTION (%) IIP3 Distribution, I-Side 1.0 5584 G65 40 TC = –40°C TC = 25°C TC = 85°C TC = 105°C 30 20 10 0 41 42 46 47 43 44 45 IMAGE REJECTION (dB) 48 5584 G66 5584f 12 LTC5584 Pin Functions IP2Q, IP2I (Pin 1, Pin 4): IIP2 Adjustment Analog Control Voltage Input for Q and I Channel. A decoupling capacitor is recommended on this pin. A low output resistance voltage source is recommended for driving these pins. These pins should be left floating if unused. DCOQ, DCOI (Pin 2, Pin 3): DC Offset Analog Control Voltage Input for Q and I Channel. A decoupling capacitor is recommended on this pin. A low output resistance voltage source is recommended for driving these pins. These pins should be left floating if unused. RF+, RF– (Pin 5, Pin 6): RF Differential Inputs. An external balun transformer with matching is used to obtain good return loss across the RF input frequency range. The RF pin should be DC-blocked with a 0.01µF coupling capacitor. GND (Pins 8, 13, 14, Exposed Pad Pin 25): Ground. These pins must be soldered to the RF ground plane on the circuit board. The backside exposed pad ground connection should have a low inductance connection and good thermal contact to the printed circuit board ground plane using many through-hole vias. See Figures 2 and 3. EN (Pin 7): Enable Pin. When the voltage on the EN pin is a logic high, the chip is completely turned on; the chip is completely turned off for a logic low. An internal 200k pull-down resistor ensures the chip remains disabled if there is no connection to the pin (open-circuit condition). VBIAS (Pin 9): This pin can be pulled to ground through a resistor to lower the current consumption of the chip. See Applications Information. VCC (Pin 10): Positive Supply Pin. This pin should be bypassed with shunt 0.01µF and 1µF capacitors. EDC (Pin 11): DC Offset Adjustment Mode Enable Pin. When the voltage on the EDC pin is a logic high, the DC offset control circuitry is enabled. The circuitry is disabled for a logic low. An internal 200k pull-down resistor ensures the circuitry remains disabled if there is no connection to the pin (open-circuit condition). EIP2 (Pin 12): IP2 Offset Adjustment Mode Enable Pin. When the voltage on the EIP2 pin is a logic high, the IP2 adjustment circuitry is enabled. The circuitry is disabled for a logic low. An internal 200k pull-down resistor ensures the circuitry remains disabled if there is no connection to the pin (open-circuit condition). LO+,LO– (Pin 15, Pin 16): LO Inputs. External matching is required to obtain good return loss across the LO input frequency range. Can be driven single ended or differentially with an external transformer. The LO pins should be DC-blocked with 0.01µF coupling capacitors. VCAP, CMQ, CMI (Pin 17, Pin 18, Pin 19): Common Mode Bypass Capacitor Pins. It is recommended that CMI and CMQ be connected to VCAP through 0.1µF capacitors. Nothing else should be connected to VCAP since it is connected to VCC inside the chip. I+, I–, Q+, Q– (Pin 23, Pin 22, Pin 21, Pin 20): Differential Baseband Output Pins for the I Channel and Q Channel. The DC bias point is VCC – 1.5V for each pin. These pins must have an external 100Ω or an inductor pull-up to VCC. REF (Pin 24): Voltage Reference Input for Analog Control Voltage Pins. A decoupling capacitor is recommended on this pin. A low output resistance voltage source is recommended for driving this pin. This pin should be left floating if unused. 5584f 13 LTC5584 Block Diagram 5 6 10 VCC RF+ 17 VCAP CMI I+ I– RF– IP2 AND DC OFFSET CAL 15 16 0° LO– EIP2 90° REF 7 VBIAS EN IP2I EDC LO+ IP2 AND DC OFFSET CAL 9 DCOI IP2Q DCOQ Q+ Q– BIAS CMQ GND 8 GND 13 GND 14 EXPOSED PAD 25 19 23 22 3 4 11 12 24 1 2 21 20 18 5584 BD 5584f 14 LTC5584 Test Circuit 0.015" 0.062" RF GND DC GND NELCO N4000-13 0.015" C29 R9 C21 R11 C22 R13 C30 R14 I– OUT Q+ OUT I+ OUT Q– OUT C10 REF C32 C33 5 C35 C34 6 RF L6 6 C17 C19 4 • T2 C40 • 1 2 C41 Q– Q+ I– REF CMI LO– LTC5584IUF IP2I LO+ + RF RF– 7 8 9 EIP2 IP2I DCOI EDC 4 VCAP VCC DCOI DCOQ VBIAS 3 CMQ GND 2 DCOQ IP2Q EN 1 IP2Q I+ 24 23 22 21 20 19 C31 18 C11 17 16 15 C38 GND 13 GND 25 GND T1 3 2 14 C39 1 4 • • 6 10 11 12 LO C13 C14 EIP2 EDC 3 EN L5 C15 C16 VCC 4.75V TO 5.25V 5584 F01 RF MATCH FREQUENCY RANGE C17 140MHz 450MHz 900MHz REF DES VALUE 1.5pF SIZE LO MATCH L6 C19 68nH 8.0pF 15nH 1.0pF 5.6nH VENDOR C13 L5 C14 82nH 12nH 2.2pF 4.0pF 3.9nH REF DES VALUE SIZE VENDOR C10, C11, C31-C35 0.1μF 0402 Murata L5, L6 See Table 0402 Murata C15, C38-C41 0.01µF 0402 Murata R9, R11, R13, R14 100Ω 0402 Vishay C13, C14, C17, C19 See Table 0402 Murata T1, T2 1:1 AT224-1 Mini-Circuits TC1-1-13M+ C16, C21, C22, C29, C30 1μF 0402 Murata Figure 1. Test Circuit Schematic 5584f 15 LTC5584 Test Circuit Figure 2. Component Side of Evaluation Board Figure 3. Bottom Side of Evaluation Board Applications Information The LTC5584 is an IQ demodulator designed for high dynamic range receiver applications. It consists of RF transconductance amplifiers, I/Q mixers, quadrature LO amplifiers, IIP2 and DC offset correction circuitry, and bias circuitry. Operation As shown in the Block Diagram for the LTC5584, the RF signal is applied to the inputs of the RF transconductor V-to-I converters and is then demodulated into I/Q baseband signals using quadrature LO signals which are internally generated by a precision 90° phase shifter. The demodulated I/Q signals are lowpass filtered on-chip with a –3dB bandwidth of 530MHz. The differential outputs of the I-channel and Q-channel are well matched in amplitude and their phases are 90° apart. RF Input Port Figure 4 shows the demodulator’s differential RF input which consists of high linearity transconductance amplifiers (V-I converters). External DC voltage should not be applied to the RF input pins. DC current flowing into the pins may cause damage to the transconductance amplifiers. Series DC blocking capacitors should be used to couple the RF input pins to the RF signal source. The RF input port can be externally matched over the operating frequency range with simple L-C matching. An input return loss greater than 10dB can be obtained over a fractional bandwidth of greater than 66% with this method. Figure 5 shows the RF input return loss for various matching component values. Table 1 shows the differential and single-ended S parameters for the RF input without using any external matching components. The input transmission line length and balun are de-embedded from the measurement. 5584f 16 LTC5584 Applications Information RF INPUT (MATCHED) L6 C17 T2 MINI-CIRCUITS TC1-1-13M+ 6 • • 1 2 C19 4 3 C40 0.01µF LTC5584 BIAS RF+ RF– C41 0.01µF Larger bandwidths can be obtained by using more elements. For example Figure 6 shows an L-C match having a bandwidth of about 98% where return loss is >10dB. Figure 7 shows the RF input return loss for the wide bandwidth match. 0 –5 5584 F04 RETURN LOSS (dB) GND Figure 4: Simplified Schematic of the RF Interface Table 1. RF Input S Parameters S11 (SINGLE ENDED) MAG ANGLE(°) MAG ANGLE(°) 10 0.5657 –2.416 0.3253 –5.287 20 0.55 –2.674 0.3055 –5.761 40 0.5391 –2.288 0.2938 –4.499 80 0.5349 –2.268 0.2984 –4.517 140 0.5336 –2.946 0.3097 –9.805 200 0.5329 –3.836 0.2989 –16.34 300 0.5317 –5.453 0.2732 –21.46 400 0.5301 –7.128 0.2614 –24.35 450 0.5292 –7.975 0.2583 –25.79 500 0.5282 –8.826 0.2562 –27.29 600 0.5258 –10.54 0.2536 –30.43 700 0.523 –12.25 0.2523 –33.66 800 0.5199 –13.97 0.2517 –36.88 900 0.5164 –15.7 0.2519 –39.97 1000 0.5124 –17.43 0.2529 –42.85 1100 0.5082 –19.17 0.2556 –45.49 1200 0.5035 –20.91 0.2609 –48.02 1300 0.4985 –22.66 0.2693 –50.73 1400 0.4931 –24.42 0.2804 –53.98 1500 0.4873 –26.19 0.2925 –57.96 1600 0.4812 –27.97 0.3035 –62.52 1700 0.4747 –29.77 0.3122 –67.36 1800 0.4678 –31.58 0.3187 –72.19 1900 0.4606 –33.41 0.3235 –76.87 2000 0.453 –35.26 0.3271 –81.36 Note: Differential S parameters measured with 1:1 balun and singleended S parameters measured with 50Ω termination on unused port. –15 –20 TC = 25°C L6 = 68nH, C19 = 8.0pF L6 = 15nH, C19 = 1.0pF C17 = 1.5pF, L6 = 5.6nH NO MATCHING –25 –30 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 FREQUENCY (GHz) 5584 F05 Figure 5. RF Input Return Loss RF INPUT 650MHz TO 950MHz L7 5.6nH L6 7.5nH C17 2.7pF T2 MINI-CIRCUITS TC1-1-13M+ 6 • • 1 2 4 3 C40 0.01µF RF+ LTC5584 BIAS RF– C41 0.01µF 5584 F06 GND Figure 6. Wide Bandwidth RF Input Match 0 TC = 25°C C17 = 2.7pF L6 = 7.5nH L7 = 5.6nH –5 RETURN LOSS (dB) FREQUENCY (MHz) S11 (DIFFERENTIAL) –10 –10 –15 –20 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 FREQUENCY (GHz) 5584 F07 Figure 7. RF Input Return Loss for Wideband Match 5584f 17 LTC5584 Applications Information Broadband Performance LO Input Port To get an idea of the broadband performance of the LTC5584, a 6dB pad can be put on the RF and LO ports, and the ports can be left unmatched. The measured RF performance for this configuration is shown in Figures 8, 9, 10 and 11 with the 6dB pad de-embedded. The RF tone spacing is 1MHz, and fLO is 10MHz lower than fRF. The conversion gain is lower than under the impedance matched condition, and correspondingly the P1dB, IIP3, and NF are higher. As shown, the part can be used at frequencies outside its specified operating range with reduced conversion gain and higher NF. The demodulator’s LO input interface is shown in Figure 12. The input consists of a high precision quadrature phase shifter which generates 0° and 90° phase shifted LO signals for the LO buffer amplifiers to drive the I/Q mixers. DC blocking capacitors are required on the LO+ and LO– inputs. 50 I, –40°C I, 25°C I, 85°C I, 105°C 45 20 Q, –40°C Q, 25°C Q, 85°C Q, 105°C 16 IIP3 30 25 NF 12 10 8 GAIN 6 20 4 P1dB 15 0 2 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 LO FREQUENCY (GHz) 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 LO FREQUENCY (GHz) 5584 F08 5584 F10 Figure 8. Broadband IIP3 and IP1dB I, –40°C I, 25°C I, 85°C I, 105°C 130 120 100 Q, –40°C Q, 25°C Q, 85°C Q, 105°C TC = –40°C TC = 25°C TC = 85°C TC = 105°C 90 110 IIP2 (dBm) Figure 10. Broadband NF and Gain IMAGE REJECTION (dB) 140 100 90 80 70 60 80 70 60 50 40 30 50 40 Q, –40°C Q, 25°C Q, 85°C Q, 105°C 14 35 10 I, –40°C I, 25°C I, 85°C I, 105°C 18 NF, GAIN (dB) IIP3, P1dB (dBm) 40 The differential and single-ended LO input S parameters with the input transmission lines and balun de-embedded are listed in Table 2. 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 LO FREQUENCY (GHz) 5584 F09 Figure 9. Broadband IIP2 20 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 LO FREQUENCY (GHz) 5584 F11 Figure 11. Broadband Image Rejection 5584f 18 LTC5584 Applications Information VCC LTC5584 LO INPUT (MATCHED) L5 C14 C13 MINI-CIRCUITS TC1-1-13M+ 6 • • 1 2 4 3 C39 0.01µF TO IDENTICAL Q-CHANNEL LO+ LO– PHASE SHIFTER C38 0.01µF 5584 F12 GND Figure 12. Simplified Schematic of LO Input Interface with External Matching Components FREQUENCY (MHz) S11 (DIFFERENTIAL) MAG ANGLE(°) S11 (SINGLE ENDED) MAG ANGLE(°) 10 0.8138 –1.736 0.7869 –1.896 20 0.8485 –6.615 0.8127 –6.425 40 0.7857 –18.67 0.7382 –16.33 80 0.6608 –25.61 0.6356 –20.1 140 0.5968 –33.93 0.5801 –25.43 200 0.5515 –42.29 0.5395 –30.5 300 0.4932 –54.56 0.4911 –37.49 400 0.4538 –65.21 0.4606 –43.5 450 0.4396 –70.18 0.4498 –46.36 500 0.4283 –75.01 0.441 –49.17 600 0.412 –84.37 0.4278 –54.67 700 0.4018 –93.45 0.4187 –60.04 800 0.3958 –102.3 0.4124 –65.26 900 0.3928 –110.9 0.4083 –70.32 1000 0.3921 –119.2 0.4059 –75.21 1100 0.3931 –127.2 0.405 –79.94 1200 0.3955 –135 0.4052 –84.52 1300 0.399 –142.4 0.4064 –88.94 1400 0.4035 –149.5 0.4084 –93.23 1500 0.4088 –156.3 0.411 –97.37 1600 0.4148 –162.9 0.4143 –101.4 1700 0.4213 –169.1 0.4181 –105.3 1800 0.4283 –175.1 0.4224 –109.1 1900 0.4357 –180.8 0.4271 –112.8 2000 0.4435 –186.2 0.4322 –116.4 Figure 13 shows LO input return loss using the MiniCircuits TC1-1-13M+ 1:1 balun with various matching component values. For optimum IIP2 and large-signal NF performance the LO inputs should be driven differentially with a 1:1 balun such as the Mini-Circuits TC1-1-13M+ or M/A Com ETC1-1-13. As shown in Figure 14, the LO input can also be driven single-ended from either the LO+ or LO– input. The unused port should be DC-blocked and terminated with a 50Ω load. Figure 15 compares the uncalibrated IIP2 performance of single ended versus differential LO drive. 0 –5 RETURN LOSS (dB) Table 2. LO Input S-Parameters –10 –15 –20 –25 –30 TC = 25°C L5 = 82nH L5 = 12nH, C14 = 4.0pF C13 = 2.2pF, L5 = 3.9nH NO MATCHING 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 FREQUENCY (GHz) 5584 F13 Figure 13. LO Input Return Loss Note: Differential S parameters measured with 1:1 balun and singleended S parameters measured with 50Ω termination on unused port. 5584f 19 LTC5584 Applications Information VCC LTC5584 LO INPUT (MATCHED) C39 0.01µF L5 C13 50Ω LO+ LO– C38 0.01µF PHASE SHIFTER 5584 F14 GND Figure 14. Recommended Single-Ended LO Input Configuration 100 90 IIP2 (dBm) 80 60 40 30 when the output port is terminated by RLOAD(SE). For instance, the gain is reduced by 6dB when each output pin is connected to a 50Ω load (or 100Ω differentially). The output should be taken differentially (or by using differential-tosingle-ended conversion) for best RF performance, including NF and IIP2. When no external filtering or matching components are used, the output response is determined by the loading capacitance and the total resistance loading the outputs. The –3dB corner frequency, fC, is given by the following equation: fC = [2π(RLOAD(SE)||100Ω||RPULL-UP) (6pF)]–1 70 50 1 50Ω 20Log10 + dB 2 RPULL-UP ||RLOAD(SE) TC = 25°C SINGLE-ENDED LO, I-SIDE DIFFERENTIAL LO, I-SIDE SINGLE-ENDED LO, Q-SIDE DIFFERENTIAL LO, Q-SIDE 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 LO FREQUENCY (GHz) 5584 F15 Figure 15. Broadband IIP2 with Differential and Single-Ended LO Drive I-Channel and Q-Channel Outputs The phase relationship between the I-channel output signal and the Q-channel output signal is fixed. When the LO input frequency is higher (or lower) than the RF input frequency, the Q-channel outputs (Q+, Q–) lead (or lag) the I-channel outputs (I+, I–) by 90°. Each of the I-channel and Q-channel outputs is internally connected to VCC through a 100Ω resistor. In order to maintain an output DC bias voltage of VCC – 1.5V, external 100Ω pull-up resistors or equivalent 15mA DC current sources are required. Each single-ended output has an impedance of 100Ω in parallel with a 6pF internal capacitor. With an external 100Ω pull-up resistor this forms a lowpass filter with a –3dB corner frequency at 530MHz. Figure 16 shows the actual measured output response with various load resistances. Figure 17 shows a simplified model of the I, Q outputs with a 100Ω differential load and 100Ω pull-ups. The –1dB bandwidth in this configuration is about 520MHz, or about twice the –1dB bandwidth with no load. Figure 18 shows a simplified model of the I, Q outputs with a L-C matching network for bandwidth extension. Capacitor CS serves to filter common mode LO switching noise immediately at the demodulator outputs. Capacitor CC in combination with inductor LS is used to peak the CONVERSION GAIN (dB) C14 TO IDENTICAL Q-CHANNEL The outputs can be DC coupled or AC coupled to external loads. The voltage conversion gain is reduced by the external load by: 7 6 5 4 3 2 1 0 –1 –2 –3 –4 –5 –6 –7 –8 TC = 25°C RLOAD(DIFF) = 100Ω, BW = 850MHz RLOAD(DIFF) = 200Ω, BW = 630MHz RLOAD(DIFF) = 400Ω, BW = 530MHz RLOAD(DIFF) = 1kΩ, BW = 460MHz 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 BASEBAND FREQUENCY (GHz) Figure 16. Conversion Gain Baseband Output Response with RLOAD(DIFF) = 100Ω, 200Ω, 400Ω and 1k and RPULL-UP = 100Ω 5584f 20 LTC5584 Applications Information output response to give greater bandwidth of 650MHz. In this case, capacitor CC was chosen as a common mode capacitor instead of a differential mode capacitor to increase rejection of common mode LO switching noise. with a 5V supply, will ensure optimum performance. Each output can sink no more than 30mA when the outputs are connected to an external load with a DC voltage higher than VCC – 1.5V. When AC output coupling is used, the resulting highpass filter’s –3dB roll-off frequency, fC, is defined by the R-C constant of the external AC coupling capacitance, CAC, and the differential load resistance, RLOAD(DIFF): In order to achieve the best IIP2 performance, it is important to minimize high frequency coupling among the baseband outputs, RF port, and LO port. Although it may increase layout complexity, routing the baseband output traces on the backside of the PCB can improve uncalibrated IIP2 performance. Figure 19 shows the alternate layout having the baseband outputs on the backside of the PCB. fC = [2π • RLOAD(DIFF) • CAC]–1 Care should be taken when the demodulator’s outputs are DC coupled to the external load to make sure that the I/Q mixers are biased properly. If the current drain from the outputs exceeds about 6mA, there can be significant degradation of the linearity performance. Keeping the common mode output voltage of the demodulator above 3.15V, VCC VCC LTC5584 6pF 100Ω 100Ω 1k 30mA AC CURRENT SOURCE 6pF PACKAGE PARASITICS 1.5nH I+ 1.5nH I– 0.2pF 0.2pF RPULL-UP 100Ω RPULL-UP 100Ω RLOAD(DIFF) 100Ω –1dB BW = 520MHz 30mA 5584 F17 GND Figure 17. Simplified Model of the Baseband Output Figure 19. Alternate Layout of PCB with Baseband Outputs on the Backside VCC VCC LTC5584 6pF 100Ω 100Ω 1k 30mA DC AC CURRENT SOURCE 6pF PACKAGE PARASITICS 1.5nF I 1.5nF I– 0.2pF 0.2pF + 30mA DC CS 2pF LS 10nH LS 10nH CS 2pF CC 4pF RPULL-UP 100Ω 6mA MAX DC CC 4pF RPULL-UP 100Ω RLOAD(DIFF) 100Ω LOWPASS –1dB BW = 650MHz 5584 F18 GND Figure 18. Simplified Model of the Baseband Output Showing Bandwidth Extension with External L, C Matching 5584f 21 LTC5584 Applications Information Analog Control Voltage Pins As shown in Figure 21, the REF pin is similar to the DCOI pin, but the bias current source is 250µA, and the internal resistance is 2k. If this pin is left disconnected, it will self-bias to 500mV. A low impedance voltage source with a source resistance of less than 200Ω is recommended to drive this pin. The control voltage range of the DCOI, DCOQ, IP2I and IP2Q pins is set by the REF pin. This range is equal to 0V to twice the voltage on the REF pin, whether internally or externally applied. It is recommended to decouple any AC noise present on the signal lines that connect to the analog control-voltage inputs. A shunt capacitor to ground placed close to these pins can provide adequate filtering. For instance, a value of 1000pF on the DCOI, DCOQ, IP2I and IP2Q pins will provide a corner frequency of around 6 to 7MHz. A similar corner frequency can be obtained on the REF pin with a value of 3900pF. Using larger capacitance values such as 0.1µF is recommended on these pins unless a faster control response is needed. Figure 22 shows the input response –3dB bandwidth for the pins versus shunt capacitance when driven from a 50Ω source. DC Offset Adjustment Circuitry Any sources of LO leakage to the RF input of a direct conversion receiver will contribute to the DC offsets of its baseband outputs. The LTC5584 features DC offset adjustment circuitry to reduce such effects. When the EDC pin is a logic high the circuitry is enabled and the resulting DC offset adjustment range is typically ±20mV. In a typical direct conversion receiver application, DC offset calibration will be done periodically at a time when no receive data is present and when the receiver DC levels have sufficiently settled. LTC5584 62.5µA DCOI, DCOQ, IP2I, IP2Q 8k 5584 F20 GND Figure 20. Simplified Schematic of the Interface for the DCOI, DCOQ, IP2I and IP2Q Pins VCC LTC5584 250µA REF 2k 5584 F21 GND Figure 21. Simplified Schematic of the REF Pin Interface 0 TC = 25°C –1 –2 RESPONSE (dB) Figure 20 shows the equivalent circuit for the DCOI, DCOQ, IP2I, and IP2Q pins. Internal temperature compensated 62.5μA current sources keep these pins biased at a nominal 500mV through 8k resistors. A low impedance voltage source with a source resistance of less than 200Ω is recommended to drive these pins. VCC –3 –4 –5 –6 –7 –8 DCOI, DCOQ; C = 470pF DCOI, DCOQ; C = 1000pF IP2I, IP2Q; C = 100pF –9 –10 0 2 4 6 8 10 12 14 16 18 20 5584 F22 FREQUENCY (MHz) Figure 22. Input Response Bandwidth for the DCOI, DCOQ, IP2I and IP2Q Pins 5584f 22 LTC5584 Applications Information DC Offset Adjustment Example Figure 23 shows a typical direct conversion receive path having a DSP feedback path for DC offset adjustment. Any sources of LO leakage to the RF input of the LTC5584 demodulator will contribute to the DC offset of the receiver. This includes both static and dynamic DC offsets. If the coupling is static in nature due to fixed board-level leakage paths, the resulting DC offset does not typically need to be adjusted at a high repetition rate. Dynamic DC offsets due to transmitter transient leakage or antenna reflection can be much harder to correct for and will require a faster update rate from the DSP. LO leakage into the RF port of the demodulator causes a DC offset at the baseband outputs which is then multiplied by the gain in the baseband path. The usable ADC voltage window will be reduced by the amplified DC offset, resulting in lower dynamic range. Using DSP, this DC offset value can be averaged and sampled at a given update rate and then a 1D minimization algorithm can be applied before a new DCOI or DCOQ control signal is generated to minimize the offset. The 1-D minimization algorithm can be implemented in many ways such as golden-section search, backtracking, or Newton’s method. limit of about 200MHz. Any IM2 component that falls in this frequency range can be minimized. Beyond this frequency, the gain of the IM2 correction amplifier falls off appreciably and the circuit no longer improves IP2 performance. The lower baseband frequency limit of the IM2 adjustment circuitry is set by the common mode reference decoupling capacitor at the CMI and CMQ pins. Below this frequency the circuit can not minimize the IM2 component. Figure 24 shows the CMI (and identical CMQ) pin interface. These pins have an internal 40pF decoupling capacitance to VCC, to provide a reference for the IP2 adjustment circuitry. The lower 3dB frequency limit, fC, of the circuitry is set by the following equation: fC = [2π • 500(40pF + CCM(EXT))]–1 Without any external capacitor on the CMI or CMQ pin the lower limit is 8MHz. By adding a 0.1μF capacitor, CCM(EXT), between the CMI and CMQ pins to VCAP, the lower –3dB frequency corner can be reduced to 3kHz. Figure 25 shows IIP2 as a function of RF frequency spacing versus common mode decoupling capacitance values of 0.1µF and 1500pF. There is effectively no limit on the size of this capacitor, VCC LTC5584 VCAP IM2 Adjustment Circuitry The LTC5584 also contains circuitry for the independent adjustment of IM2 levels on the I and Q channels. When the EIP2 pin is a logic high, this circuitry is enabled and the IP2I and IP2Q analog control voltage inputs are able to adjust the IM2 level. The IM2 level can be effectively minimized over a large range of the baseband bandwidth. The circuitry has an effective baseband frequency upper 40pF CMI OR CMQ 5584 F24 GND Figure 24. Equivalent Circuit of the CMI and CMQ Pin Interfaces DSP DAC DCOI BPF LNA ADC DC AVERAGING LOWPASS FILTER 1-D MINIMIZATION ALGORITHM SAMPLE AND HOLD LTC5584 fLO = 900MHz 5585 F23 Figure 23. Block Diagram of a Receiver with a DSP Feedback Loop for DC Offset Adjustment 5584f 23 LTC5584 Applications Information other than the impact it has on enable time for the IM2 circuitry to be operational. When the chip is disabled, there is no current in the I or Q mixers, so the common mode output voltage will be equal to VCC (if no DC common mode current is being drawn by external baseband circuitry such as a baseband amplifier). When the chip is enabled, the off-chip common mode decouping capacitor must charge up through a 500Ω resistor. The time constant for this is essentially 500Ω times the common mode decoupling capacitance value. For example, with a 0.01µF capacitor this wait time is approximately 30μs. Figure 26 shows the pulsed enable response of the common-mode output voltage with 0.01µF on the CMI and CMQ pins. 130 0.1µF (UNCALIBRATED) 0.1µF (NULLED IP2I = 0.6V) 1500pF (UNCALIBRATED) 1500pF (NULLED IP2I = 0.6V) 110 T = 25°C C fRF1 = 1000MHz 100 f = 960MHz LO IIP2 (dBm) 120 90 80 70 60 0.1 1 10 RF FREQUENCY SPACING (MHz) 5584 F25 Figure 25. IIP2 vs Common Mode Decoupling Capacitance VCM (V) 6 5 EN PULSE ON 0 5 –5 CMI, CMQ 4 3 ENABLE VOLTAGE (V) EN PULSE OFF –10 0 A simplified schematic of the EN pin is shown in Figure 28. The enable voltage necessary to turn on the LTC5584 is 2V. To disable or turn off the chip, this voltage should be below 0.3V. If the EN pin is not connected, the chip is disabled. Figures 29 and 30 show the simplified schematics for the EDC and EIP2 pins. 10 TC = 25°C CCMI,Q = 0.01µF 7 IM2 adjustment circuitry can be used in a typical transceiver loop-back application as shown in Figure 27. In this example a 2-tone SSB training source of f1 = 20MHz and f2 = 21MHz is generated in DSP and upconverted by the LTC5588-1 quadrature modulator to RF tones at 870MHz and 871MHz using an LO source at 850MHz. A narrowband RF filter is required to remove the IM2 component generated by the LTC5588-1. During the loopback test these RF tones are routed through high isolation switches and an attenuation pad to the LTC5584 demodulator input. The tones are then downconverted by the same LO source at 850MHz to produce two tones at the baseband outputs of 20MHz and 21MHz plus an IM2 impairment signal at 1MHz. After baseband channel filtering and amplification the output of the ADC is filtered by a 1MHz bandpass filter in DSP to isolate the IM2 tone. The power in this tone is calculated in DSP and then a 1-D minimization algorithm is applied to calculate the correction signal for the IP2I control voltage pin. The 1-D minimization algorithm can be implemented in many ways such as golden-section search, backtracking or Newton’s method. Enable Interface 50 0.01 8 IM2 Suppression Example BASEBAND OUTPUTS –15 10 20 30 40 50 60 70 80 90 100 TIME (µs) 5584 F26 Figure 26. Common Mode Output Voltage with a Pulsed Enable It is important that the voltage applied to the EN, EDC and EIP2 pins should never exceed VCC by more than 0.3V. Otherwise, the supply current may be sourced through the upper ESD protection diode connected at the pin. Under no circumstances should voltage be applied directly to the enable pins before the supply voltage is applied to the VCC pin. If this occurs, damage to the IC may result. A 1k resistor in series with the enable pin can be used to limit current. Reducing Power Consumption Figure 31 shows the simplified schematic of the VBIAS interface. The VBIAS pin can be used to lower the mixer 5584f 24 LTC5584 Applications Information DSP 1-D MINIMIZATION ALGORITHM DAC 1MHz BPF IP2I LNA RMS DETECTION ADC LTC5584 LOOPBACK fLO = 850MHz f1 = 20MHz + DAC PA f2 = 21MHz LTC5588-1 5584 F27 Figure 27. Block Diagram for a Direct Conversion Transceiver with IM2 Adjustment. Only the I-Channel Is Shown VCC VCC LTC5584 LTC5584 EIP2 EN 100k 100k 10k 100k 100k 5584 F28 5584 F30 GND GND Figure 28. Simplified Schematic of the EN Pin Interface Figure 30. Simplified Schematic of the EIP2 Pin Interface VCC VCC LTC5584 LTC5584 EDC VBIAS OPTIONAL R TO REDUCE CURRENT 100k 100Ω EN COPT 100k EN 5584 F29 GND 5584 F31 GND Figure 29. Simplified Schematic of the EDC Pin Interface Figure 31. Simplified Schematic of the VBIAS Pin Interface core bias current and total power consumption for the chip. For example, adding 487Ω from the VBIAS pin to GND will lower the DC current to 169mA, at the expense of reduced IIP3 performance. Figure 32 shows IIP3 and P1dB performance versus DC current and resistor value. An optional capacitor, COPT in Figure 31, has minimal effect on improving PSRR and IIP2. 5584f 25 LTC5584 Applications Information 50 45 P1dB, IIP3 (dBm) 40 35 I, 194mA I, 169mA, 487Ω I, 145mA, 294Ω TC = 25°C fRF = 900MHz where P0 is the input noise power and –174dBm is the input thermal noise power in a 1Hz bandwidth. A measured 2-tone output spectrum at 890MHz is shown in Figure 36. IIP3 is calculated from the 2-tone IM3 levels: Q, 194mA Q, 169mA, 487Ω Q, 145mA, 294Ω IIP3 30 IIP3 = (–6.929 – (–88.33))/2 – 15.4 25 IIP3 = 25.3dBm 20 P1dB 15 10 5 0 400 600 1200 1000 800 RF FREQUENCY (MHz) 1400 5584 F32 Figure 32. IIP3 and P1dB vs DC Current and VBIAS Resistor Value For this example, receiver noise floor is approximated by a measurement from 28MHz to 36MHz offset frequency, where adequate filtering for RF and LO signals was possible. Using the test data from Figure 36, the receiver noise figure for the I-channel (Ch 1) is calculated using the –8.4dBm input power, 15kHz bin width, 40MHz bandwidth, and –108dBFS measured in-band noise floor: SNRIN = PIN – P0 900MHz Receiver Application SNRIN = –8.4 – (–174 + 76) = 89.6dB Figure 33 shows a typical receiver application consisting of the chain of LNA, demodulator, lowpass filter, ADC driver, and ADC. Total DC power consumption is about 2.1W. Fullscale power at the RF input is –8.4dBm. The Chebychev lowpass filter with unequal terminations is designed using the method shown in the appendix. Filter component values are then adjusted for the best overall response and available component values. A positive voltage gain slope with frequency is necessary to compensate for the roll-off contributed by the ADC Driver and Anti-Alias Filter. From the chain analysis shown in Figure 34, the IIP3-NF dynamic range figure of merit (FOM) is 5.3dB at the LNA input, 11.3dB at the demodulator input, and 16.8dB at the ADC driver amp input. SNROUT = –10 Log10(BinW/BW) – Floor SNROUT = –43.3 + 108 = 73.7dB NF = SNRIN – SNROUT NF = 89.6 – 73.7 = 15.9dB Finally, the receiver spurious free dynamic range can be calculated using the measured data at 890MHz: SFDR = 2(IIP3 – NF – P0)/3 SFDR = 2(25.3 – 15.9 – (–174 + 76))/3 SFDR = 73dB The measured 6th order lowpass baseband response is shown in Figure 35. The receiver spurious free dynamic range (SFDR) in terms of FOM can be calculated using the following equations: FOM = IIP3 – NF SFDR = 2/3(FOM – P0) P0 = –174dBm + 10Log10(BW|Hz) 5584f 26 RF INPUT 800MHz TO 1000MHz C1 0.01µF C5 100pF L2 33nH C3 4.7µF R1 0Ω C2 0.01µF LO INPUT 881MHz 6dBm C25 0.01µF C24 0.01µF 5V 200mA 6 1 2 RF– LO+ RF+ L7 180nH L8 180nH C13 150pF L5 470nH L6 470nH C12 47pF T1 MINI-CIRCUITS TC1-1-13M+ C10 0.01µF C14 150pF C9 47pF C17 1µF C15 150pF C16 150pF R4 110Ω 40MHz LOWPASS FILTER C18 0.1µF R7 20Ω R6 20Ω R5 110Ω R8 402Ω C20 0.5pF + – R9 402Ω VOCN LTC6409 – + 5V 52mA C19 0.5pF L9 270nH R15 27.4Ω R10 100Ω R13 243Ω R16 27.4Ω R11 36.5Ω L10 270nH R12 36.5Ω R14 243Ω R18 86.6Ω C22 39pF L12 180nH L11 180nH C21 39pF R17 86.6Ω 40MHz ANTI-ALIAS FILTER Figure 33. Simplified Schematic of 900MHz Receiver, (Only I-Channel Is Shown) 4 3 I– LO– VCC + LTC5584 I C8 2.2pF C11 0.01µF C7 1.5pF L3 5.6nH C4 33pF C6 4.7µF L4 3.9nH L2 33nH T2 MINI-CIRCUITS TC1-1-13M+ 6 • • 1 2 4 3 LNA BIAS AVAGO MGA-633P8 R3 10Ω 5V 48mA • • R2 5.6k 5585 F33 R19 150Ω R20 150Ω C23 1µF AIN– VCM AIN+ CONTROL LTC2185 ADC VDD 1.8V 206mA D13 • • • D0 LTC5584 Applications Information 5584f 27 LTC5584 Applications Information 900MHz Receiver Chain Analysis G = 34.2dB NF = 1.7dB IIP3 = 7dBm FOM = 5.3dB MGA-633P8 G = 18dB NF = 0.37dB OIP3 = 37dBm G = 16.2dB NF = 14.2dB IIP3 = 25.5dBm FOM = 11.3dB G = 16.5dB NF = 11.8dB IIP3 = 28.6dBm FOM = 16.8dB LTC5584 40MHz LPF G = –0.3dB NF = 10.4dB IIP3 = 27.9dBm G = –0.3dB NF = 0.3dB G = 16.8dB NF = 11.5dB IIP3 = 28.3dBm FOM = 16.8dB LTC6409 G = –1.2dB NF = 24.3dB IIP3 = 48.7dBm FOM = 24.4dB 40MHz AAF G = 18dB NF = 10dB OIP3 = 50dBm G = –1.2dB NF = 1.2dB G = 0dB NF = 23.1dB IIP3 = 47.5dBm FOM = 24.4dB LTC2185 G = 0dB NF = 23.1dB IP3 = 47.5dBm 5584 F34 Figure 34. 900MHz Receiver Chain Analysis 20 TC = 25°C 10 0 GAIN (dB) –10 –20 –30 –40 –50 –60 –70 –80 0 20 40 60 80 100 120 140 160 5584 F35 FREQUENCY (MHz) Figure 35. Baseband Gain Response without LNA Figure 36. fRF = 889MHz and 890MHz 2-Tone Receiver Test, fLO = 881MHz. Ch.1 Is the I Channel and Ch.2 Is the Q Channel. Tested without LNA 28 5584f LTC5584 Appendix Chebychev Filter Synthesis with Unequal Terminations To synthesize Chebychev filters with unequal terminations, two equally terminated filters are synthesized at the two different impedance levels and the resulting networks are joined using the Impedance Bisection Theorem[1]. This method only works with symmetrical odd-order filters. The general lowpass prototype element values are generated by the method shown [2]: L | β =In coth Ar dB 17.37 β γ = sinh 2n ak = sin π ( 2k – 1) , k = 1,2,...,n 2n 2 2 πk b = γ + sin , k = 1,2,...,n k n where LAr|dB is the passband ripple in dB, and n is the filter order. The prototype element values will be: g1 = gk = 2a 1 γ 4a k a k–1 b k−1g k−1 , k = 1,2,...,n where RIN is the input impedance and the terminating impedance ROUT is equal to RIN for the n odd case but is scaled by the gn+1 prototype value for the n even case. The Impedance Bisection Theorem can be applied to symmetrical networks by dividing the element values along the networks’ plane of symmetry, and then adding the two networks together. The filter response is preserved. For example, if LAr|dB = 0.2dB, fC = 40MHz, RIN = 100Ω, ROUT = 20Ω and n = 5, the prototype element values and resulting scaled filter values are listed: Filter 1: RIN = ROUT = 100Ω g1 = 1.339 → C1 = 53.3pF g2 = 1.337 → L1 = 531.98nH g3 = 2.166 → C2 = 86.19pF g4 = 1.337 → L2 = 531.98nH g5 = 1.339 → C3 = 53.3pF Filter 2: RIN = ROUT = 20Ω g1 = 1.339 → C1 = 266.48pF g2 = 1.337 → L1 = 106.4nH g3 = 2.166 → C2 = 430.93pF g4 = 1.337 → L2 = 106.4nH g5 = 1.339 → C3 = 266.48pF The Impedance Bisection Theorem can be applied at the plane of symmetry about C2 such that a new value of C2 can be computed with half the values of the two filters. g n+1 = 1 for n odd β g n+1 = coth for n even 4 2 Assuming the first element is a capacitor, we can scale the filter capacitor prototype values up to our desired cutoff frequency fC: Ck = gk , k = 1,3,...,n 2π • fC •RIN The filter inductor values can be scaled with: LK = gk •RIN , k = 2,4,...,n 2π • fC C2→ 86.19pF 430.93pF + = 258.56pF 2 2 The final unequally-terminated filter design values are shown in Figure 37. [1] A.C. Bartlett, “An Extension of a Property of Artificial Lines,” Phil. Mag., vol.4, p.902, November 1927. [2] G. Matthaei, L. Young, and E.M.T. Jones, Microwave Filters, Impedance-Matching Networks, and Coupling Structures, p.99, 1964. RIN 100Ω + – L1 531.98nH C1 53.3pF L2 106.4nH C2 258.56pF C3 266.48pF ROUT 20Ω 5585 F37 Figure 37. Final Design Schematic 5584f 29 LTC5584 Appendix Image Rejection Calculation The image rejection ratio (IRR) can then be written as: Image rejection can be calculated from the measured gain and phase error responses of the demodulator. Consider the signal diagram of Figure 38: where: Written in terms of AERR and φERR as: RF(t) = sin(ωLO + ωBB)t + sin(ωLO – ωIM)t IRR|dB = 10log IRR|dB = 10log |desired|2 |image|2 |1+ AERR 2 + 2AERR cos ( φERR ) | |1+ AERR 2 − 2AERR cos ( φERR ) | LOI(t) = cos(ωLOt + φERR) LOQ(t) = sin(ωLOt) Figure 39 shows image rejection as a function of amplitude and phase errors for a demodulator. ωLO + ωBB is the desired sideband frequency and ωLO – ωIM is the image frequency. The total phase error of the I and Q channels is lumped into the I-channel LO source as φERR. The total gain error is represented by AERR, and is lumped into a gain multiplier in the I-channel. AERR LOQ(t) AERR sin ( ω BB t – φERR ) – sin ( ωIM t + φERR ) 2 1 Q(t) = cos ( ω BB t ) + cos ( ωIM t ) 2 Shifting the Q channel by –90° can be accomplished by replacing sine with cosine such that the shifted Q-channel signal is: 1 Q –90 (t) = sin ( ω BB t ) + sin ( ωIM t ) 2 We combine I(t) + Q–90(t) and choose terms containing ωBB as the desired signal: 1 A desired = sin ( ω BB t ) + ERR sin ( ω BB t – φERR ) 2 2 Similarly, we choose terms containing ωIM as the image signal: Q(t) 5585 F38 Figure 38. Signal Diagram for a Demodulator 70 AERR = 0dB AERR = 0.05dB AERR = 0.1dB AERR = 0.2dB AERR = 0.3dB AERR = 0.5dB AERR = 1dB 60 IMAGE REJECTION (dB) I(t) = LOI(t) RF(t) After lowpass filtering the I and Q signals can be written as: I(t) 50 40 30 20 10 0 1 2 3 4 5 6 7 PHASE ERROR (DEG) 8 9 10 5585 F39 Figure 39. Image Rejection as a Function of Gain and Phase Errors 1 A image = sin ( ωIM t ) – ERR sin ( ωIM t + φERR ) 2 2 5584f 30 LTC5584 Package Description Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings. UF Package 24-Lead Plastic QFN (4mm × 4mm) (Reference LTC DWG # 05-08-1697 Rev B) 0.70 ±0.05 4.50 ±0.05 2.45 ±0.05 3.10 ±0.05 (4 SIDES) PACKAGE OUTLINE 0.25 ±0.05 0.50 BSC RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS 4.00 ±0.10 (4 SIDES) BOTTOM VIEW—EXPOSED PAD R = 0.115 TYP 0.75 ±0.05 PIN 1 NOTCH R = 0.20 TYP OR 0.35 × 45° CHAMFER 23 24 PIN 1 TOP MARK (NOTE 6) 0.40 ±0.10 1 2 2.45 ±0.10 (4-SIDES) (UF24) QFN 0105 REV B 0.200 REF 0.00 – 0.05 0.25 ±0.05 0.50 BSC NOTE: 1. DRAWING PROPOSED TO BE MADE A JEDEC PACKAGE OUTLINE MO-220 VARIATION (WGGD-X)—TO BE APPROVED 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE, IF PRESENT 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 5584f Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 31 LTC5584 Typical Application Simplified Schematic of 900MHz Receiver, (Only I-Channel Is Shown) RF INPUT 800MHz TO 1000MHz C2 0.01µF L3 5.6nH C7 1.5pF T2 MINI-CIRCUITS TC1-1-13M+ 6 • • 1 2 4 3 C17 1µF 5V 200mA C24 0.01µF C9 47pF VCC + LTC5584 I RF+ RF– LO+ I– LO– C25 0.01µF C11 0.01µF C14 150pF C16 150pF L5 470nH L7 180nH L6 470nH L8 180nH C12 47pF C13 150pF R6 20Ω R7 20Ω C15 150pF 40MHz ANTI-ALIAS FILTER (AAF) R8 402Ω R5 110Ω 5V 52mA VOCM LTC6409 + – C18 0.1µF • R14 243Ω R12 36.5Ω – + R9 402Ω C20 0.5pF R15 27.4Ω L9 270nH R11 36.5Ω L10 270nH R16 27.4Ω R13 243Ω R10 100Ω 3 R17 86.6Ω 1.8V 206mA C21 39pF AIN+ L11 180nH VCM L12 180nH C22 39pF R19 150Ω AIN– R20 150Ω C23 1µF R18 86.6Ω VDD LTC2185 ADC D13 • • • D0 CONTROL 5584 TA02 T1 MINI-CIRCUITS TC1-1-13M+ • L4 3.9nH R4 110Ω C10 0.01µF 1 2 LO INPUT 900MHz 6dBm C19 0.5pF 40MHz LOWPASS FILTER 6 4 C8 2.2pF Related Parts PART NUMBER Infrastructure LTC5569 LT5527 LT5557 LTC6409 LTC6412 LTC554X LT5554 LTC5585 DESCRIPTION COMMENTS 300MHz to 4GHz Dual Active Downconverting Mixer 400MHz to 3.7GHz, 5V Downconverting Mixer 400MHz to 3.8GHz, 3.3V Downconverting Mixer 10GHz GBW Differential Amplifier 31dB Linear Analog VGA 600MHz to 4GHz Downconverting Mixer Family Ultralow Distortion IF Digital VGA 700MHz to 3GHz IQ Demodulator 2dB Gain, 26.7dBm IIP3 and 11.7dB NF at 1950MHz, 3.3V/180mA Supply 2.3dB Gain, 23.5dBm IIP3 and 12.5dB NF at 1900MHz, 5V/78mA Supply 2.9dB Gain, 24.7dBm IIP3 and 11.7dB NF at 1950MHz, 3.3V/82mA Supply DC-Coupled, 48dBm OIP3 at 140MHz, 1.1nV/√Hz Input Noise Density 35dBm OIP3 at 240MHz, Continuous Gain Range –14dB to 17dB 8dB Gain, >25dBm IIP3, 10dB NF, 3.3V/200mA Supply 48dBm OIP3 at 200MHz, 2dB to 18dB Gain Range, 0.125dB Gain Steps >530MHz IQ Bandwidth, 25.7dBm IIP3, IIP2 Adjustable to >80dBm, DC Offset Null Adjustment 8.7dB Gain, 26dBm IIP3, 9.7dB Noise Figure 8.5dB Gain, 26.2dBm IIP3, 9.9dB Noise Figure 8.3dB Gain, 27.3dBm IIP3, 9.8dB Noise Figure LTC5590 Dual 600MHz to 1.7GHz Downconverting Mixer LTC5591 Dual 1.3GHz to 2.3GHz Downconverting Mixer LTC5592 Dual 1.6GHz to 2.7GHz Downconverting Mixer RF PLL/Synthesizer with VCO LTC6946-1 Low Noise, Low Spurious Integer-N PLL with Integrated VCO LTC6946-2 Low Noise, Low Spurious Integer-N PLL with Integrated VCO LTC6946-3 Low Noise, Low Spurious Integer-N PLL with Integrated VCO ADCs LTC2145-14 14-Bit, 125Msps 1.8V Dual ADC LTC2185 16-Bit, 125Msps 1.8V Dual ADC LTC2158-14 14-Bit, 310Msps 1.8V Dual ADC, 1.25GHz Full-Power Bandwidth 373MHz to 3.74GHz, –157dBc/Hz WB Phase Noise Floor, –100dBc/Hz Closed-Loop Phase Noise 513MHz to 4.9GHz, –157dBc/Hz WB Phase Noise Floor, –100dBc/Hz Closed-Loop Phase Noise 640MHz to 5.79GHz, –157dBc/Hz WB Phase Noise Floor, –100dBc/Hz Closed-Loop Phase Noise 73.1dB SNR, 90dB SFDR, 95mW/Ch Power Consumption 76.8dB SNR, 90dB SFDR, 185mW/Channel Power Consumption 68.8dBFS SNR, 88dB SFDR, 362mW/Ch Power Consumption, 1.32VP-P Input Range 5584f 32 Linear Technology Corporation LT 0412 • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com LINEAR TECHNOLOGY CORPORATION 2012