MIC45205-1

MIC45205-1/-2
26V/6A DC-to-DC Power Module
General Description
Features
Micrel’s MIC45205 is a synchronous step-down regulator
module, featuring a unique adaptive ON-time control
architecture. The module incorporates a DC-to-DC
controller, power MOSFETs, bootstrap diode, bootstrap
capacitor, and an inductor in a single package; simplifying
the design and layout process for the end user.
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This highly-integrated solution expedites system design
and improves product time-to-market. The internal
MOSFETs and inductor are optimized to achieve high
efficiency at a low output voltage. The fully-optimized
design can deliver up to 6A current under a wide input
voltage range of 4.5V to 26V, without requiring additional
cooling.
®
The MIC45205-1 uses Micrel’s HyperLight Load (HLL)
MIC45205-2 uses Micrel’s Hyper Speed Control™
architecture which enables ultra-fast load transient
response, allowing for a reduction of output capacitance.
The MIC45205 offers 1% output accuracy that can be
adjusted from 0.8V to 5.5V with two external resistors.
Additional features include thermal shutdown protection,
input undervoltage lockout, adjustable current limit, and
short circuit protection. The MIC45205 allows for safe
start-up into a pre-biased output.
Datasheet and other support documentation can be found
on Micrel’s web site at: www.micrel.com.
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No compensation required
Up to 6A output current
>93% peak efficiency
Output voltage: 0.8V to 5.5V with ±1% accuracy
Adjustable switching frequency from 200kHz to 600kHz
Enable input and open-drain power good output
Hyper Speed Control (MIC45205-2) architecture
enables fast transient response
HyperLight Load (MIC45205-1) improves light load
efficiency
Supports safe startup into pre-biased output
CISPR22, Class B compliant
–40°C to +125°C junction temperature range
Thermal-shutdown protection
Short-circuit protection with hiccup mode
Adjustable current limit
Available in 52-pin 8mm × 8mm × 3mm QFN package
Applications
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High power density point-of-load conversion
Servers, routers, Networking, and base stations
FPGAs, DSP, and low-voltage ASIC power supplies
Industrial and medical equipment
Typical Application
Efficiency (VIN = 12V)
vs. Output Current (MIC45205-1)
100
5.0VOUT
90
3.3VOUT
2.5VOUT
EFFICIENCY (%)
80
1.8VOUT
70
1.5VOUT
1.2VOUT
60
1.0VOUT
50
0.8VOUT
40
30
fSW = 600kHz
20
10
0
1
2
3
4
5
6
7
8
9
OUTPUT CURRENT (A)
Hyper Speed Control is a trademark of Micrel, Inc.
HyperLight Load is a registered trademark of Micrel, Inc.
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
January 21, 2015
Revision 2.0
Micrel, Inc.
MIC45205
Ordering Information(1)
Switching
Frequency
Features
Junction
Temperature
Range
Package
Lead
Finish
MIC45205-1YMP
200kHz to 600kHz
Hyper Light Load
–40°C to +125°C
52-pin
8mm × 8mm × 3mm QFN
Pb-Free
MIC45205-2YMP
200kHz to 600kHz
Hyper Speed Control
–40°C to +125°C
52-pin
8mm × 8mm × 3mm QFN
Pb-Free
Part Number
Note:
1. Devices are ESD sensitive. Handling precautions are recommended.
Pin Configuration
52-Pin 8mm × 8mm × 3mm QFN
(Top View)
January 21, 2015
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Micrel, Inc.
MIC45205
Pin Description
MIC45205
Pin Number
Pin Name
1
GND
Analog Ground. Connect bottom feedback resistor to GND. GND and PGND should be
connected together at a low impedance point.
2, 3
5VDD
Internal +5V Linear Regulator Output. Powered by VIN, 5VDD is the internal supply bus for the
device. In the applications with VIN < +5.5V, 5VDD should be tied to VIN to by-pass the linear
regulator.
4, 5
PVDD
PVDD. Supply input for the internal low-side power MOSFET driver.
6 − 8,
45, 52
PGND
Power Ground. PGND is the return path for the step-down power module power stage. The
PGND pin connects to the sources of internal low-side power MOSFET, the negative terminals of
input capacitors, and the negative terminals of output capacitors.
SW
The SW pin connects directly to the switch node. Due to the high-speed switching on this pin, the
SW pin should be routed away from sensitive nodes. The SW pin also senses the current by
monitoring the voltage across the low-side MOSFET during OFF time.
10 − 12,
31 − 34
Pin Function
14 −19
PVIN
Power Input Voltage. Connection to the drain of the internal high side power MOSFET. Connect
an input capacitor from PVIN to PGND.
21 − 29
VOUT
Output Voltage. Connected to the internal inductor, the output capacitor should be connected
from this pin to PGND as close to the module as possible.
36 − 38
RIA
Ripple Injection Pin A. Leave floating, no connection.
39
RIB
Ripple Injection Pin B. Connect this pin to FB.
40, 41
ANODE
42 − 44
BST
46
FB
Feedback. Input to the transconductance amplifier of the control loop. The FB pin is referenced to
0.8V. A resistor divider connecting the feedback to the output is used to set the desired output
voltage. Connect the bottom resistor from FB to GND.
47
PG
Power Good. Open drain output. If used, connect to an external pull-up resistor of at least
10kohm between PG and the external bias voltage.
48
EN
Enable. A logic signal to enable or disable the step-down regulator module operation. The EN pin
is TTL/CMOS compatible. Logic high = enable, logic low = disable or shutdown. EN pin has an
internal 1MΩ (typical) pull-down resistor to GND. Do not leave floating
49
VIN
Internal 5V Linear Regulator Input. A 1μF ceramic capacitor from VIN to GND is required for
decoupling.
50
FREQ
51
ILIM
9, 13,
20, 30, 35
KEEPOUT
Depopulated pin positions.
−
PVIN ePAD
PVIN Exposed Pad. Internally connected to PVIN pins. Please see PCB Layout
Recommendations section.
−
VOUT ePAD
VOUT Exposed Pad. Internally connected to VOUT pins. Please see PCB Layout
Recommendations section.
January 21, 2015
Anode Bootstrap Diode. Anode connection of internal bootstrap diode, this pin should be
connected to the PVDD pin.
Connection to the internal bootstrap circuitry and high-side power MOSFET drive circuitry.
Connect all three BST pins together.
Switching Frequency Adjust. Use a resistor divider from VIN to GND to program the switching
frequency. Connecting FREQ to VIN sets frequency = 600kHz.
Current Limit. Connect a resistor between ILIM and SW to program the current limit.
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Micrel, Inc.
MIC45205
Absolute Maximum Ratings(2)
Operating Ratings(3)
VPVIN, VVIN to PGND ....................................... −0.3V to +30V
VPVDD, V5VDD, VANODE to PGND......................... −0.3V to +6V
VSW , VFREQ, VILIM, VEN to PGND ............ −0.3V to (VIN +0.3V)
VBST to VSW ........................................................ −0.3V to 6V
VBST to PGND .................................................. −0.3V to 36V
VPG to PGND .................................. −0.3V to (5VDD + 0.3V)
VFB, VRIB to PGND .......................... −0.3V to (5VDD + 0.3V)
PGND to GND .............................................. −0.3V to +0.3V
Junction Temperature .............................................. +150°C
Storage Temperature (TS) ......................... −65°C to +150°C
Lead Temperature (soldering, 10s) ............................ 260°C
Supply Voltage (VPVIN, VVIN) .............................. 4.5V to 26V
Output Current ................................................................. 6A
Enable Input (VEN) .................................................. 0V to VIN
Power Good (VPG) ............................................. 0V to 5VDD
Junction Temperature (TJ) ........................ −40°C to +125°C
(4)
Junction Thermal Resistance
8mm × 8mm × 3mm QFN-52 (θJA) ................ 21.7°C/W
8mm × 8mm × 3mm QFN-52 (θJC) .................. 5.0°C/W
Electrical Characteristics(5)
VIN = VEN = 12V, VOUT = 3.3V, VBST − VSW = 5V, TJ = +25ºC. Bold values indicate −40ºC < TJ < +125ºC, unless otherwise noted.
Parameter
Condition
Min.
Typ.
Max.
Units
26
V
Power Supply Input
4.5
Input Voltage Range (VPVIN, VIN)
Quiescent Supply Current
(MIC45205-1)
VFB = 1.5V
0.35
0.75
mA
Quiescent Supply Current
(MIC45205-2)
VFB = 1.5V
2.1
3
mA
Operating Current
VPVIN = VIN = 12V, VOUT = 1.8V, IOUT = 0A
fSW = 600kHz (MIC45205-2)
31
Shutdown Supply Current
SW = unconnected, VEN = 0V
0.1
10
µA
mA
5VDD Output
5VDD Output Voltage
VIN = 7V to 26V, I5VDD = 10mA
4.8
5.1
5.4
V
5VDD UVLO Threshold
V5VDD rising
3.8
4.2
4.6
V
5VDD UVLO Hysteresis
V5VDD falling
LDO Load Regulation
I5VDD = 0 to 40mA
400
mV
0.6
2
3.6
%
TJ = 25°C
0.792
0.8
0.808
−40°C ≤ TJ ≤ 125°C
0.784
0.8
0.816
5
500
Reference
Feedback Reference Voltage
FB Bias Current
VFB = 0.8V
V
nA
Notes:
2. Exceeding the absolute maximum rating may damage the device.
3. The device is not guaranteed to function outside operating range.
4. θJA and θJC were measured using the MIC45205 evaluation board.
5. Specification for packaged product only.
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Micrel, Inc.
MIC45205
Electrical Characteristics(5) (Continued)
VIN = VEN = 12V, VOUT = 3.3V, VBST − VSW = 5V, TJ = +25ºC. Bold values indicate −40ºC < TJ < +125ºC, unless otherwise noted.
Parameter
Condition
Min.
Typ.
Max.
Units
Enable Control
1.8
EN Logic Level High
V
0.6
EN Logic Level Low
EN Hysteresis
EN Bias Current
200
VEN = 12V
V
mV
5
10
600
750
µA
Oscillator
Switching Frequency
400
VFREQ = VIN, IOUT = 2A
VFREQ = 50% VIN, IOUT = 2A
350
Maximum Duty Cycle
Minimum Duty Cycle
VFB = 1V
Minimum Off-Time
140
kHz
85
%
0
%
200
260
ns
Soft-Start
Soft-Start Time
5
ms
Short-Circuit Protection
Current-Limit Threshold
VFB = 0.79V
−30
−14
0
mV
Short-Circuit Threshold
VFB = 0V
−23
−7
9
mV
Current-Limit Source Current
VFB = 0.79V
55
70
85
µA
Short-Circuit Source Current
VFB = 0V
25
35
45
µA
SW, BST Leakage Current
10
µA
FREQ Leakage Current
10
µA
95
% VOUT
Leakage
Power Good (PG)
85
PG Threshold Voltage
Sweep VFB from Low-to-High
90
PG Hysteresis
Sweep VFB from High-to-Low
6
% VOUT
PG Delay Time
Sweep VFB from Low-to-High
100
µs
PG Low Voltage
VFB < 90% × VNOM, IPG = 1mA
70
TJ Rising
160
°C
15
°C
200
mV
Thermal Protection
Overtemperature Shutdown
Overtemperature Shutdown Hysteresis
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Micrel, Inc.
MIC45205
Typical Characteristics
10
480
360
240
VIN = 12V
VOUT = 1.8V
IOUT = 0A
fSW = 600kHz
120
2.0
VIN = 12V
VOUT = 1.8V
IOUT = 0A
8
6
4
2
0
25
50
75
100
125
-50
-25
0
TEMPERATURE (°C)
FEEBACK VOLTAGE (V)
VIN = 12V
VOUT = 1.8V
IOUT = 0A
6
4
2
50
75
100
0
25
50
75
FALLING
0.8
VIN = 12V
VOUT = 1.8V
0.4
-50
125
-25
100
2.1
1.0
2.0
0.9
0.8
0.7
VIN = 12V
VOUT = 1.8V
IOUT = 0A
50
75
100
125
1.9
1.8
1.7
VIN = 12V
VOUT= 1.8V
IOUT = 0A
1.6
1.5
-50
125
25
Output Voltage
vs. Temperature
1.1
0.6
0
TEMPERATURE (°C)
0.5
0
-25
1.2
Feedback Voltage
vs. Temperature
10
-50
RISING
TEMPERATURE (°C)
EN Bias Current
vs. Temperature
8
25
OUTPUT VOLTAGE (V)
0
-25
1.6
0.0
0
-50
EN BIAS CURRENT (µA)
ENABLE THRESHOLD (V)
VDD SUPPLY VOLTAGE (V)
SUPPLY CURRENT (µA)
600
-25
0
25
50
75
100
125
TEMPERATURE (°C)
TEMPERATURE (°C)
Switching Frequency
vs. Temperature
-50
-25
0
25
50
75
100
125
TEMPERATURE (°C)
Output Peak Current Limit
vs. Temperature
20
900
18
800
16
CURRENT LIMIT (A)
SWITCHING FREQUENCY (kHz)
Enable Threshold
vs. Temperature
VDD Supply Voltage
vs. Temperature
VIN Operating Supply Current
vs. Temperature (MIC45205-1)
700
600
500
VIN =12V
VOUT = 1.8V
fSW = 600kHz
12
10
8
6
4
VIN = 12V
VOUT = 1.8V
IOUT = 2A
400
14
2
300
0
-50
-25
0
25
50
75
TEMPERATURE (°C)
January 21, 2015
100
125
-50
-25
0
25
50
75
100
125
TEMPERATURE (°C)
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Micrel, Inc.
MIC45205
Typical Characteristics (Continued)
Efficiency (VIN = 5V)
vs. Output Current (MIC4205-1)
Efficiency (VIN = 12V)
vs. Output Current (MIC45205-1)
100
1.8VOUT
60
1.2VOUT
1.5VOUT
1.0VOUT
50
2.5VOUT
0.8VOUT
40
EFFICIENCY (%)
70
3.3VOUT
80
EFFICIENCY (%)
1.8VOUT
70
1.5VOUT
1.2VOUT
60
1.0VOUT
50
0.8VOUT
40
fSW = 600kHz
20
10
1
2
3
4
5
6
7
8
9
5.0VOUT
80
3.3VOUT
2.5VOUT
70
1.8VOUT
1.5VOUT
60
1.2VOUT
50
1.0VOUT
40
0.8VOUT
20
fSW = 600kHz
10
10
0
90
30
30
fSW = 600kHz
20
0
1
2
4
3
5
6
7
8
0
9
1
2
3
4
5
6
7
8
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Efficiency (VIN = 5V)
vs. Output Current (MIC45205-2)
Efficiency (VIN = 12V)
vs. Output Current (MIC45205-2)
Efficiency (VIN = 24V)
vs. Output Current (MIC45205-2)
100
100
90
3.3VOUT
90
3.3VOUT
90
80
2.5VOUT
80
2.5VOUT
80
70
1.8VOUT
1.5VOUT
2.5VOUT
70
1.8VOUT
1.5VOUT
70
60
1.2VOUT
1.0VOUT
1.8VOUT
1.5VOUT
60
1.2VOUT
1.0VOUT
60
50
0.8VOUT
1.2VOUT
1.0VOUT
EFFICIENCY (%)
50
0.8VOUT
40
40
fSW = 600kHz
1
2
3
4
5
6
7
8
0
9
fSW = 600kHz
10
10
0
0.8VOUT
40
20
20
10
50
30
fSW = 600kHz
20
5.0VOUT
3.3VOUT
30
30
9
100
5.0VOUT
EFFICIENCY (%)
EFFICIENCY (%)
2.5VOUT
5.0VOUT
90
3.3VOUT
80
30
EFFICIENCY (%)
100
100
90
1
2
3
4
5
6
7
8
9
Line Regulation
0
1
2
3
4
5
6
7
8
9
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Output Voltage
vs. Input Voltage
2.1
1.0%
0.8%
VOUT = 1.8V
IOUT = 0A
0.6%
2.0
OUTPUT VOLTAGE (V)
TOTAL REGULATION (%)
Efficiency (VIN = 24V)
vs. Output Current (MIC45205-1)
0.4%
0.2%
0.0%
-0.2%
-0.4%
-0.6%
VOUT = 1.8V
IOUT = 4A
1.9
1.8
1.7
1.6
-0.8%
1.5
-1.0%
5.5
9.8
14.1
18.4
INPUT VOLTAGE (V)
January 21, 2015
22.7
4.5
8.8
13.1
17.4
21.7
26.0
INPUT VOLTAGE (V)
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Micrel, Inc.
MIC45205
Typical Characteristics (Continued)
IC Power Dissipation (VIN = 5V)
vs. Output Current
3
4.5
2.5
2.5VOUT
1.8VOUT
2
1.5VOUT
1.2VOUT
3.3VOUT
1.5
1.0VOUT
0.8VOUT
1
0.5
VIN = 12V
fSW = 600kHz
3
5.0VOUT
3.3VOUT
2.5
2.5VOUT
1.8VOUT
2
1.5VOUT
1.2VOUT
1.5
0.8VOUT
1
0.5
1.0VOUT
0
1
2
3
4
5
6
7
OUTPUT CURRENT (A)
January 21, 2015
8
9
VIN = 12V
fSW = 600kHz
4
5.0VOUT
3.3VOUT
3.5
2.5VOUT
3
1.8VOUT
2.5
1.5VOUT
2
1.2VOUT
1.5
0.8VOUT
1.0VOUT
1
0.5
0
0
0
IC POWER DISSIPATION (W)
3.5
VIN = 5V
fSW = 600kHz
IC POWER DISSIPATION (W)
IC POWER DISSIPATION (W)
IC Power Dissipation (VIN = 24V)
vs. Output Current
IC Power Dissipation (VIN = 12V)
vs. Output Current
0
1
2
4
3
5
6
7
OUTPUT CURRENT (A)
8
8
9
0
1
2
3
4
5
6
7
8
9
OUTPUT CURRENT (A)
Revision 2.0
Micrel, Inc.
MIC45205
Functional Characteristics
January 21, 2015
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Micrel, Inc.
MIC45205
Functional Characteristics (Continued)
January 21, 2015
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Micrel, Inc.
MIC45205
Functional Characteristics (Continued)
January 21, 2015
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Micrel, Inc.
MIC45205
Functional Diagram
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Micrel, Inc.
MIC45205
Functional Description
The MIC45205 is an adaptive on-time synchronous buck
regulator module built for high-input voltage to low-output
voltage conversion applications. The MIC45205 is
designed to operate over a wide input voltage range,
from 4.5V to 26V, and the output is adjustable with an
external resistor divider. An adaptive on-time control
scheme is employed to obtain a constant switching
frequency in steady state and to simplify the control
compensation. Hiccup mode over-current protection is
implemented by sensing low-side MOSFET’s RDS(ON). The
device features internal soft-start, enable, UVLO, and
thermal shutdown. The module has integrated switching
FETs, inductor, bootstrap diode, resistor, and capacitor.
At the end of the ON-time period, the internal high-side
driver turns off the high-side MOSFET and the low-side
driver turns on the low-side MOSFET. The OFF-time
period length depends upon the feedback voltage in most
cases. When the feedback voltage decreases and the
output of the gm amplifier falls below 0.8V, the ON-time
period is triggered and the OFF-time period ends. If the
OFF-time period determined by the feedback voltage is
less than the minimum OFF-time tOFF(MIN), which is about
200ns, the MIC45205 control logic will apply the tOFF(MIN)
instead. tOFF(MIN) is required to maintain enough energy in
the boost capacitor (CBST) to drive the high-side
MOSFET.
Theory of Operation
As shown in Figure 1 (in association with Equation 1), the
output voltage is sensed by the MIC45205 feedback pin
(FB) via the voltage divider RFB1 and RFB2 and compared
to a 0.8V reference voltage (VREF) at the error comparator
through a low-gain transconductance (gm) amplifier. If
the feedback voltage decreases, and the amplifier output
falls below 0.8V, then the error comparator will trigger the
control logic and generate an ON-time period. The ONtime period length is predetermined by the “Fixed tON
Estimator” circuitry:
The maximum duty cycle is obtained from the 200ns
tOFF(MIN):
DMAX =
t S − t OFF(MIN)
tS
= 1−
200ns
tS
Eq. 2
Where:
tS = 1/fSW . It is not recommended to use MIC45205 with
an OFF-time close to tOFF(MIN) during steady-state
operation.
The adaptive ON-time control scheme results in a
constant switching frequency in the MIC45205 during
steady state operation. The actual ON-time and resulting
switching frequency will vary with the different rising and
falling times of the MOSFETs. Also, the minimum tON
results in a lower switching frequency in high VIN to VOUT
applications. During load transients, the switching
frequency is changed due to the varying OFF-time.
To illustrate the control loop operation, we will analyze
both the steady-state and load transient scenarios. For
easy analysis, the gain of the gm amplifier is assumed to
be 1. With this assumption, the inverting input of the error
comparator is the same as the feedback voltage.
Figure 2 shows the MIC45205 control loop timing during
steady-state operation. During steady-state, the gm
amplifier senses the feedback voltage ripple, which is
proportional to the output voltage ripple plus injected
voltage ripple, to trigger the ON-time period. The ON-time
is predetermined by the tON estimator. The termination of
the OFF-time is controlled by the feedback voltage. At the
valley of the feedback voltage ripple, which occurs when
VFB falls below VREF, the OFF period ends and the next
ON-time period is triggered through the control logic
circuitry.
Figure 1. Output Voltage Sense via FB Pin
t ON(ESTIMATED) =
VOUT
VIN × fSW
Eq. 1
where VOUT is the output voltage, VIN is the power stage
input voltage, and fSW is the switching frequency.
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MIC45205
Unlike true current-mode control, the MIC45205 uses the
output voltage ripple to trigger an ON-time period. The
output voltage ripple is proportional to the inductor
current ripple if the ESR of the output capacitor is large
enough.
In order to meet the stability requirements, the MIC45205
feedback voltage ripple should be in phase with the
inductor current ripple and are large enough to be sensed
by the gm amplifier and the error comparator. The
recommended feedback voltage ripple is 20mV~100mV
over full input voltage range. If a low ESR output
capacitor is selected, then the feedback voltage ripple
may be too small to be sensed by the gm amplifier and
the error comparator. Also, the output voltage ripple and
the feedback voltage ripple are not necessarily in phase
with the inductor current ripple if the ESR of the output
capacitor is very low. In these cases, ripple injection is
required to ensure proper operation. Please refer to
“Ripple Injection” subsection in the Application
Information section for more details about the ripple
injection technique.
Figure 2. MIC45205 Control Loop Timing
Figure 3 shows the operation of the MIC45205 during a
load transient. The output voltage drops due to the
sudden load increase, which causes the VFB to be less
than VREF. This will cause the error comparator to trigger
an ON-time period. At the end of the ON-time period, a
minimum OFF-time tOFF(MIN) is generated to charge the
bootstrap capacitor (CBST) since the feedback voltage is
still below VREF. Then, the next ON-time period is
triggered due to the low feedback voltage. Therefore, the
switching frequency changes during the load transient,
but returns to the nominal fixed frequency once the
output has stabilized at the new load current level. With
the varying duty cycle and switching frequency, the
output recovery time is fast and the output voltage
deviation is small. Note that the instantaneous switching
frequency during load transient remains bounded and
cannot increase arbitrarily. The minimum is limited by tON
+ tOFF(MIN) .Since the variation in VOUT is relatively limited
during load transient, tON stays virtually close to its
steady-state value.
Discontinuous Mode (MIC45205-1 only)
In continuous mode, the inductor current is always
greater than zero; however, at light loads, the MIC452051 is able to force the inductor current to operate in
discontinuous mode. Discontinuous mode is where the
inductor current falls to zero, as indicated by trace (IL)
shown in Figure 4. During this period, the efficiency is
optimized by shutting down all the non-essential circuits
and minimizing the supply current as the switching
frequency is reduced. The MIC45205-1 wakes up and
turns on the high-side MOSFET when the feedback
voltage VFB drops below 0.8V.
The MIC45205-1 has a zero crossing comparator (ZC)
that monitors the inductor current by sensing the voltage
drop across the low-side MOSFET during its ON-time. If
the VFB > 0.8V and the inductor current goes slightly
negative, then the MIC45205-1 automatically powers
down most of the IC circuitry and goes into a low-power
mode.
Once the MIC45205-1 goes into discontinuous mode,
both DL and DH are low, which turns off the high-side
and low-side MOSFETs. The load current is supplied by
the output capacitors and VOUT drops. If the drop of VOUT
causes VFB to go below VREF, then all the circuits will
wake up into normal continuous mode. First, the bias
currents of most circuits reduced during the
discontinuous mode are restored, and then a tON pulse is
triggered before the drivers are turned on to avoid any
possible glitches. Finally, the high-side driver is turned
on. Figure 4 shows the control loop timing in
discontinuous mode.
Figure 3. MIC45205 Load Transient Response
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MIC45205
Figure 5. MIC45205 Current-Limiting Circuit
In each switching cycle of the MIC45205, the inductor
current is sensed by monitoring the low-side MOSFET in
the OFF period. The sensed voltage VILIM is compared
with the power ground (PGND) after a blanking time of
150ns. In this way the drop voltage over the resistor R15
(VCL) is compared with the drop over the bottom FET
generating the short current limit. The small capacitor
(C15) connected from ILIM pin to PGND filters the
switching node ringing during the off-time allowing a
better short-limit measurement. The time constant
created by R15 and C6 should be much less than the
minimum off time.
Figure 4. MIC45205-1 Control Loop Timing
(Discontinuous Mode)
During discontinuous mode, the bias current of most
circuits is substantially reduced. As a result, the total
power supply current during discontinuous mode is only
about 350µA, allowing the MIC45205-1 to achieve high
efficiency in light load applications.
Soft-Start
Soft-start reduces the input power supply surge current at
startup by controlling the output voltage rise time. The
input surge appears while the output capacitor is charged
up.
The VCL drop allows programming of short limit through
the value of the resistor (R15). If the absolute value of the
voltage drop on the bottom FET becomes greater than
VCL, and the VILIM falls below PGND, an overcurrent is
triggered causing the IC to enter hiccup mode. The
hiccup sequence including the soft-start reduces the
stress on the switching FETs and protects the load and
supply for severe short conditions.
The MIC45205 implements an internal digital soft-start by
making the 0.8V reference voltage VREF ramp from 0 to
100% in about 5ms with 9.7mV steps. Therefore, the
output voltage is controlled to increase slowly by a staircase VFB ramp. Once the soft-start cycle ends, the related
circuitry is disabled to reduce current consumption.
PVDD must be powered up at the same time or after VIN
to make the soft-start function correctly.
The short-circuit current limit can be programmed by
using Equation 3.
Current Limit
The MIC45205 uses the RDS(ON) of the low-side MOSFET
and external resistor connected from ILIM pin to SW
node to set the current limit.
R15 =
(ICLIM − DIL (PP) × 0.5) × R DS(ON) + VCL
ICL
Eq. 3
Where:
ICLIM = Desired current limit
RDS(ON) = On-resistance of low-side power MOSFET,
16mΩ typically.
VCL = Current-limit threshold (typical absolute value is
14mV per the Electrical Characteristics table).
ICL = Current-limit source current (typical value is 70µA,
per the Electrical Characteristics table).
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MIC45205
ΔIL(PP) = Inductor current peak-to-peak, since the inductor
is integrated use Equation 4 to calculate the inductor
ripple current.
The peak-to-peak inductor current ripple is:
∆IL(PP) =
VOUT × (VIN(max) − VOUT )
VIN(max) × fsw × L
Eq. 4
The MIC45205 has a 1.0µH inductor integrated into the
module. In case of a hard short, the short limit is folded
down to allow an indefinite hard short on the output
without any destructive effect. It is mandatory to make
sure that the inductor current used to charge the output
capacitance during soft-start is under the folded short
limit; otherwise the supply will go in hiccup mode and
may not finish the soft-start successfully.
The MOSFET RDS(ON) varies 30% to 40% with
temperature; therefore, it is recommended to add a 50%
margin to ICLIM in Equation 3 to avoid false current limiting
due to increased MOSFET junction temperature rise.
With R15 = 1.37kΩ and C15 = 15pF, the typical output
current limit is 8A.
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MIC45205
Application Information
Setting the Switching Frequency
The MIC45205 switching frequency can be adjusted by
changing the value of resistors R1 and R2.
The switching frequency also depends upon VIN, VOUT
and load conditions as MIC45205 uses and adaptive ONtime architecture as explained in the “Theory of
Operation” subsection in the Functional Description.
Output Capacitor Selection
The type of the output capacitor is usually determined by
the application and its equivalent series resistance
(ESR). Voltage and RMS current capability are two other
important factors for selecting the output capacitor.
Recommended capacitor types are MLCC, OS-CON and
POSCAP. The output capacitor’s ESR is usually the main
cause of the output ripple. The MIC45205 requires ripple
injection and the output capacitor ESR affects the control
loop from a stability point of view.
The maximum value of ESR is calculated as in Equation
6:
Figure 6. Switching Frequency Adjustment
Equation 5 gives the estimated switching frequency:
ESR COUT ≤
fSW = fO ×
R2
R1 + R2
ΔVOUT(PP)
ΔIL(PP)
Eq. 6
Eq. 5
Where:
ΔVOUT(PP) = Peak-to-peak output voltage ripple
Where:
fO = 600kHz (typical per the Electrical Characteristics
table)
ΔIL(PP) = Peak-to-peak inductor current ripple
(5)
R1= 100kΩ is recommended.
R2 needs to be selected in order to set the required
switching frequency.
Switching Frequency
800
700
VOUT = 5V
VIN = 12V
SW FREQ (kHz)
600
500
400
300
200
R1 = 100kΩ
100
0
10.00
100.00
1000.00
10000.00
R2 (kΩ)
Figure 7. Switching Frequency vs. R2
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MIC45205
Input Capacitor Selection
The input capacitor for the power stage input PVIN
should be selected for ripple current rating and voltage
rating. The input voltage ripple will primarily depend on
the input capacitor’s ESR. The peak input current is equal
to the peak inductor current, so:
The total output ripple is a combination of the ESR and
output capacitance. The total ripple is calculated in
Equation 7:
2
ΔIL(PP)


 + ΔIL(PP) × ESR C
ΔVOUT(PP) = 
OUT

 C OUT × fSW × 8 
(
)2
∆VIN = IL(pk ) × ESR CIN
Eq. 10
Eq. 7
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming the
peak-to-peak inductor current ripple is low:
Where:
D = Duty cycle
COUT = Output capacitance value
fsw = Switching frequency
ICIN(RMS) ≈ IOUT(MAX) × D × (1 − D)
As described in the “Theory of Operation” subsection in
the Functional Description, the MIC45205 requires at
least 20mV peak-to-peak ripple at the FB pin to make the
gm amplifier and the error comparator behave properly.
Also, the output voltage ripple should be in phase with
the inductor current. Therefore, the output voltage ripple
caused by the output capacitors value should be much
smaller than the ripple caused by the output capacitor
ESR. If low-ESR capacitors, such as ceramic capacitors,
are selected as the output capacitors, a ripple injection
method should be applied to provide enough feedback
voltage ripple. Please refer to “Ripple Injection”
subsection in the Application Information section for more
details.
Eq.11
The power dissipated in the input capacitor is:
PDISS(CIN) = ICIN(RMS) 2 × ESR CIN
Eq. 12
The general rule is to pick the capacitor with a ripple
current rating equal to or greater than the calculated
worst-case RMS capacitor current.
Equation 13 should be used to calculate the input
capacitor. Also it is recommended to keep some margin
on the calculated value:
The output capacitor RMS current is calculated in
Equation 8:
CIN ≈
ICOUT (RMS) =
ΔIL(PP)
12
IOUT(MAX) × (1 − D)
fSW × dV
Eq. 13
Eq. 8
Where:
dV = The input ripple
The power dissipated in the output capacitor is:
fSW = Switching frequency
2
PDISS(COUT ) = ICOUT (RMS) × ESR COUT
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MIC45205
Output Voltage Setting Components
The MIC45205 requires two resistors to set the output
voltage as shown in Figure 8:
Table 1. VOUT Programming Resistor Look-Up
The output voltage is determined by Equation 14:
Eq. 14
0.8V
40.2kΩ
1.0V
20kΩ
1.2V
11.5kΩ
1.5V
8.06kΩ
1.8V
4.75kΩ
2.5V
3.24kΩ
3.3V
1.91kΩ
5.0V
1. Enough ripple at the feedback voltage due to the
large ESR of the output capacitors:
VFB = 0.8V
As shown in Figure 9, the converter is stable without
any ripple injection.
A typical value of RFB1 used on the standard evaluation
board is 10kΩ. If R1 is too large, it may allow noise to be
introduced into the voltage feedback loop. If RFB1 is too
small in value, it will decrease the efficiency of the power
supply, especially at light loads. Once RFB1 is selected,
RFB2 can be calculated using Equation 15:
VFB × R FB1
VOUT − VFB
Eq. 15
For fixed RFB1 = 10kΩ, output voltage can be selected by
RFB2. Table 1 provides RFB2 values for some common
output voltages.
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OPEN
The applications are divided into two situations according
to the amount of the feedback voltage ripple:
Where:
R FB2 =
VOUT
Ripple Injection
The VFB ripple required for proper operation of the
MIC45205 gM amplifier and error comparator is 20mV to
100mV. However, the output voltage ripple is generally
too small to provide enough ripple amplitude at the FB
pin and this issue is more visible in lower output voltage
applications. If the feedback voltage ripple is so small that
the gM amplifier and error comparator cannot sense it,
then the MIC45205 will lose control and the output
voltage is not regulated. In order to have some amount of
VFB ripple, a ripple injection method is applied for low
output voltage ripple applications.
Figure 8. Voltage-Divider Configuration


R
VOUT = VFB ×  1 + FB1 
R
FB2 

RFB2
Figure 9. Enough Ripple at FB from ESR
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MIC45205
The feedback voltage ripple is:
The injected ripple is:
ΔVFB(PP) = VIN × K div × D × (1 - D) ×
ΔVFB(PP) =
R FB2
× ESR C OUT × ΔIL(PP)
R FB1 + R FB2
1
fSW × τ
Eq.17
Eq. 16
K div =
Where:
ΔIL(PP) = The peak-to-peak value of the inductor current
ripple
R FB1//R FB2
R INJ + R FB1//R FB2
Eq.18
Where:
VIN = Power stage input voltage
2. Virtually no or inadequate ripple at the FB pin voltage
due to the very-low ESR of the output capacitors,
such is the case with ceramic output capacitor. In this
case, the VFB ripple waveform needs to be generated
by injecting suitable signal. MIC45205 has provisions
to enable an internal series RC injection network, RINJ
and CINJ as shown in Figure 10 by connecting RIB to
FB pin. This network injects a square-wave current
waveform into FB pin, which by means of integration
across the capacitor (C14) generates an appropriate
sawtooth FB ripple waveform.
D = Duty cycle
fSW = Switching frequency
τ = (RFB1//RFB2//RINJ) × C14
RINJ= 10kΩ
CINJ = 0.1µF
In Equations 18 and 19, it is assumed that the time
constant associated with C14 must be much greater than
the switching period:
1
T
= << 1
fSW × τ τ
Eq. 19
If the voltage divider resistors RFB1 and RFB2 are in the kΩ
range, then a C14 of 1nF to 100nF can easily satisfy the
large time constant requirements.
Figure 10. Internal Ripple Injection at FB via RIB Pin
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MIC45205
Thermal Measurements and Safe Operating Area
(SOA)
Measuring the IC’s case temperature is recommended to
ensure it is within its operating limits. Although this might
seem like a very elementary task, it is easy to get
erroneous results. The most common mistake is to use
the standard thermal couple that comes with a thermal
meter. This thermal couple wire gauge is large, typically
22 gauge, and behaves like a heatsink, resulting in a
lower case measurement.
Two methods of temperature measurement are using a
smaller thermal couple wire or an infrared thermometer. If
a thermal couple wire is used, it must be constructed of
36-gauge wire or higher (smaller wire size) to minimize
the wire heat-sinking effect. In addition, the thermal
couple tip must be covered in either thermal grease or
thermal glue to make sure that the thermal couple
junction is making good contact with the case of the IC.
Omega brand thermal couple (5SC-TT-K-36-36) is
adequate for most applications.
Wherever possible, an infrared thermometer is
recommended. The measurement spot size of most
infrared thermometers is too large for an accurate
reading on a small form factor ICs. However, an IR
thermometer from Optris has a 1mm spot size, which
makes it a good choice for measuring the hottest point on
the case. An optional stand makes it easy to hold the
beam on the IC for long periods of time.
The safe operating area (SOA) of the MIC45205 is shown
in Figure 11, Figure 12, Figure 13, Figure 14, and Figure
15. These thermal measurements were taken on
MIC45205 evaluation board. Since the MIC45205 is an
entire system comprised of switching regulator controller,
MOSFETs and inductor, the part needs to be considered
as a system. The SOA curves will give guidance to
reasonable use of the MIC45205.
SOA curves should only be used as a point of reference.
SOA data was acquired using the MIC45205 evaluation
board. Thermal performance depends on the PCB layout,
board size, copper thickness, number of thermal vias,
and actual airflow.
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MIC45205
7
MAXIMUM OUTPUT CURRENT (A)
MAXIMUM OUTPUT CURRENT (A)
7
6
5
0 LFM
200 LFM
400 LFM
4
3
85
90
95
100
105
110
115
120
6
5
0 LFM
200 LFM
400 LFM
4
3
80
125
90
95
100
105
110
115
120
AMBIENT TEMPERATURE (°C)
AMBIENT TEMPERATURE(°C)
Figure 11. MIC45205 Power Derating vs. Airflow
(5VIN to 1.5VOUT)
Figure 12. MIC45205 Power Derating vs. Airflow
(12VIN to 1.5VOUT)
7
MAXIMUM OUTPUT CURRENT (A)
7
MAXIMUM OUTPUT CURRENT (A)
85
6
5
0 LFM
200 LFM
400 LFM
4
3
80
85
90
95
100
105
110
115
6
5
4
0 LFM
200 LFM
400 LFM
3
70
120
75
80
85
90
95 100 105 110 115 120
AMBIENT TEMPERATURE (°C)
AMBIENT TEMPERATURE (°C)
Figure 13. MIC45205 Power Derating vs. Airflow
(12VIN to 3.3VOUT)
Figure 14. MIC45205 Power Derating vs. Airflow
(24VIN to 1.5VOUT)
MAXIMUM OUTPUT CURRENT (A)
7
6
5
4
0 LFM
200 LFM
400 LFM
3
60
65
70
75
80
85
90
95 100 105 110
AMBIENT TEMPERATURE (°C)
Figure 15. MIC45205 Power Derating vs. Airflow
(24VIN to 3.3VOUT)
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MIC45205
PCB Layout Guidelines
Input Capacitor
Warning: To minimize EMI and output noise, follow
these layout recommendations.
PCB layout is critical to achieve reliable, stable and
efficient performance. A ground plane is required to
control EMI and minimize the inductance in power, signal
and return paths.
Figure 16 is optimized from a small form factor point of
view shows top and bottom layer of a four layer PCB. It is
recommended to use mid layer 1 as a continuous ground
plane.
•
Place the input capacitors on the same side of the
board and as close to the IC as possible.
•
Place several vias to the ground plane close to the
input capacitor ground terminal.
•
Use either X7R or X5R dielectric input capacitors. Do
not use Y5V or Z5U type capacitors.
•
Do not replace the ceramic input capacitor with any
other type of capacitor. Any type of capacitor can be
placed in parallel with the ceramic input capacitor.
•
If a non-ceramic input capacitor is placed in parallel
with the input capacitor, it must be recommended for
switching regulator applications and the operating
voltage.
•
In “Hot-Plug” applications, an Electrolytic bypass
capacitor must be used to limit the over-voltage spike
seen on the input supply with power is suddenly
applied. If hot-plugging is the normal operation of the
system, using an appropriate hot-swap IC is
recommended.
RC Snubber (Optional)
•
Depending on the operating conditions, a RC
snubber on the same side of the board can be used.
Place the RC and as close to the SW pin as possible
if needed.
SW Node
• Do not route any digital lines underneath or close to
the SW node.
• Keep the switch node (SW) away from the feedback
(FB) pin.
Output Capacitor
•
Use a wide trace to connect the output capacitor
ground terminal to the input capacitor ground
terminal.
•
Phase margin will change as the output capacitor
value and ESR changes.
•
The feedback trace should be separate from the
power trace and connected as close as possible to
the output capacitor. Sensing a long high-current load
trace can degrade the DC load regulation.
Figure 16. Top And Bottom Layer of a Four-Layer Board
The following guidelines should be followed to insure
proper operation of the MIC45205 module:
IC
•
The analog ground pin (GND) must be connected
directly to the ground planes. Place the IC close to
the point-of-load (POL).
•
Use thick traces to route the input and output power
lines.
•
Analog and power grounds should be kept separate
and connected at only one location with a low
impedance.
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Micrel, Inc.
MIC45205
PCB Layout Recommendations
Top − Copper Layer 1
Copper Layer 2
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MIC45205
PCB Layout Recommendations (Continued)
Copper Layer 3
Bottom − Copper Layer 4
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MIC45205
Simplified PCB Design Recommendations
Periphery I/O Pad Layout and Large Pad for Exposed
Heatsink
The board design should begin with copper/metal pads
that sit beneath the periphery leads of a mounted QFN.
The board pads should extend outside the QFN package
edge a distance of approximately 0.20mm per side:
After completion of the periphery pad design, the larger
exposed pads will be designed to create the mounting
surface of the QFN exposed heatsink. The primary transfer
of heat out of the QFN will be directly through the bottom
surface of the exposed heatsink. To aid in the transfer of
generated heat into the PCB, the use of an array of plated
through-hole vias beneath the mounted part is
recommended. The typical via hole diameter is 0.30mm to
0.35mm, with center-to-center pitch of 0.80mm to 1.20mm.
Total pad length = 8.00mm + (0.20mm per side × 2 sides)
= 8.40mm
Note:
Exposed metal trace is “mirror image” of package bottom view.
Figure 17. Package Bottom View vs. PCB Recommended Exposed Metal Trace
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MIC45205
Solder Paste Stencil Design (Recommend Stencil
Thickness = 112.5 ±12.5µm)
The solder stencil aperture openings should be smaller
than the periphery or large PCB exposed pads to reduce
any chance of build-up of excess solder at the large
exposed pad area which can result to solder bridging.
The suggested reduction of the stencil aperture opening is
typically 0.20mm smaller than exposed metal trace.
Note: A critical requirement is to not duplicate land pattern
of the exposed metal trace as solder stencil opening as the
design and dimension values are different.
Note:
Cyan-colored shaded pad indicate exposed trace keep out area.
Figure 18. Solder Stencil Opening
Figure 19. Stack-Up of Pad Layout and Solder Paste Stencil
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MIC45205
Evaluation Board Schematic
Bill of Materials
Item
C1
Part Number
EEE-FK1V221P
Manufacturer
Panasonic
(6)
C1X, C6, C9,
C10, C7, C13
C3
C3216X5R1H106M160AB
(7)
TDK
(8)
Description
Qty.
220µF/35V, ALE Capacitor (optional)
1
Open
6
10uF/50V, 1206, X5R, 10%, MLCC
1
0.1µF/50V, X7R, 0603, 10%, MLCC
3
C2, C4, C8
GRM188R71H104KA93D
Murata
C5
C3216X5R0J107M160AB
TDK
100µF/6.3V, X5R, 1206, 20%, MLCC
1
C12
C1608C0G1H222JT
TDK
2.2nF/50V, NP0, 0603, 5%, MLCC
1
C11
GRM1885C1H150JA01D
Murata
15pF/50V, NP0, 0603, 5%, MLCC
3
CON1, CON2,
CON3, CON4
8174
15A, 4-Prong Through-Hole Screw Terminal
4
(9)
Keystone
Notes:
6. Panasonic: www.panasonic.com.
7. TDK: www.TDK.com.
8. Murata: www.murata.com.
9. Keystone: www.keyelco.com.
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MIC45205
Bill of Materials (Continued)
Item
J1
J2, J3, J4, TP3 − TP5
Part Number
M50-3500742
90120-0122
Manufacturer
Harwin
(10)
(11)
Molex
JPx1, JPx2
R1, R10
CRCW0603100K0FKEA
Vishay Dale
(12)
R2, R12, R13, R16
Description
Qty.
Header 2x7
1
Header 2
6
Open
2
100kΩ, 1%, 1/10W, 0603, Thick Film
2
Open
4
R3
CRCW060340K2FKEA
Vishay Dale
40.2kΩ, 1%, 1/10W, 0603, Thick Film
1
R4
CRCW06020K0FKEA
Vishay Dale
20kΩ, 1%, 1/10W, 0603, Thick Film
1
R5
CRCW060311K5FKEA
Vishay Dale
11.5kΩ, 1%, 1/10W, 0603, Thick Film
1
R6
CRCW06038K06FKEA
Vishay Dale
8.06kΩ, 1%, 1/10W, 0603, Thick Film
1
R7
CRCW06034K75FKEA
Vishay Dale
4.75kΩ, 1%, 1/10W, 0603, Thick Film
1
R8
CRCW06033K24FKEA
Vishay Dale
3.24kΩ, 1%, 1/10W, 0603, Thick Film
1
R9
CRCW06031K91FKEA
Vishay Dale
1.91kΩ, 1%, 1/10W, 0603, Thick Film
1
R11
CRCW060349K9FKEA
Vishay Dale
49.9kΩ, 1%, 1/10W, 0603, Thick Film
1
R14
CRCW060310K0FKEA
Vishay Dale
10kΩ, 1%, 1/10W, 0603, Thick Film
1
R15
CRCW06031K37FKEA
Vishay Dale
1.37kΩ, 1%, 1/10W, 0603, Thick Film
1
R17, R18, R19
RCG06030000Z0EA
Vishay Dale
0Ω Resistor, 1%, 1/10W, 0603, Thick Film
3
TP6 − TP9, JPx3, JPx4
1502-2
Single-End, Through-Hole Terminal
6
26V/6A DC-to-DC Power Module
1
U1
MIC45205-1YMP
MIC45205-2YMP
Keystone
(13)
Micrel, Inc.
Notes:
10. Harwin: http://www.harwin.com
11. Molex: www.molex.com.
12. Vishay-Dale: www.vishay.com.
13. Micrel: www.micrel.com.
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Micrel, Inc.
MIC45205
Package Information and Recommended Landing Pattern(14)
52-Pin 8mm × 8mm QFN (MP)
Note:
14. Package information is correct as of the publication date. For updates and most current information, go to www.micrel.com.
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MIC45205
Package Information and Recommended Landing Pattern(14) (Continued)
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MIC45205
Package Information and Recommended Landing Pattern(14) (Continued)
January 21, 2015
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MIC45205
Thermally-Enhanced Landing Pattern
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Micrel, Inc.
MIC45205
Thermally-Enhanced Landing Pattern (Continued)
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MIC45205
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
Micrel, Inc. is a leading global manufacturer of IC solutions for the worldwide high-performance linear and power, LAN, and timing & communications
markets. The Company’s products include advanced mixed-signal, analog & power semiconductors; high-performance communication, clock
management, MEMs-based clock oscillators & crystal-less clock generators, Ethernet switches, and physical layer transceiver ICs. Company
customers include leading manufacturers of enterprise, consumer, industrial, mobile, telecommunications, automotive, and computer products.
Corporation headquarters and state-of-the-art wafer fabrication facilities are located in San Jose, CA, with regional sales and support offices and
advanced technology design centers situated throughout the Americas, Europe, and Asia. Additionally, the Company maintains an extensive network
of distributors and reps worldwide.
Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this datasheet. This
information is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry,
specifications and descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual
property rights is granted by this document. Except as provided in Micrel’s terms and conditions of sale for such products, Micrel assumes no liability
whatsoever, and Micrel disclaims any express or implied warranty relating to the sale and/or use of Micrel products including liability or warranties
relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright, or other intellectual property right.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical
implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
© 2014 Micrel, Incorporated.
January 21, 2015
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