MIC2164/-2/-3/C Synchronous Buck Controllers featuring Adaptive On-Time Control 28V Input, Constant Frequency Hyper Speed Control™ Family General Description The Micrel MIC2164/-2/-3/C are constant-frequency, synchronous buck controllers featuring adaptive on-time control. The MIC2164/-2/-3/C are the first products in the new Hyper Speed Control™ family of buck controllers introduced by Micrel. The MIC2164/-2/-3/C controllers operate over an input supply range of 3V to 28V, and are independent of the IC supply voltage. The devices are capable of supplying 25A output current. While the MIC2164 operates at 300kHz, the MIC2164-2 operates at 600kHz, and the MIC2164-3 operates at 1MHz. A unique Hyper Speed Control architecture allows for ultra-fast transient response while reducing the output capacitance and also makes High VIN/Low VOUT operation possible. The MIC2164/-2/-3/C controllers utilizes an architecture which is adaptive TON ripple controlled. A UVLO feature is provided to ensure proper operation under power-sag conditions to prevent the external power MOSFET from overheating. A soft start feature is provided to reduce the inrush current. Foldback current limit and “hiccup” mode short-circuit protection ensure FET and load protection. The MIC2164/-2/-3/C controllers are available in a 10-pin MSOP (MAX1954A-compatible) package with a junction operating range from –40°C to +125°C. Datasheets and support documentation are available on Micrel’s web site at: www.micrel.com. Hyper Speed Control™ Features Hyper Speed Control architecture enables high delta V operation (VHSD = 28V and VOUT = 0.8V) and smaller output capacitors than competitors 3V to 28V input voltage Any Capacitor™ stable (Zero ESR to high ESR) 25A output current capability 300kHz/600kHz/1MHz switching frequency Adaptive on-time mode control Adjustable output from 0.8V to 5.5V with ±1% (MIC2164/-2/-3) or ±3% (MIC2164C) FB accuracy Up to 95% efficiency Foldback current-limit, “hiccup” mode short-circuit protection, thermal shutdown, and safe pre-bias startup 6ms Internal soft start –40°C to +125°C junction temperature range Available in 10-pin MSOP package Applications Set-top box, gateways and routers Printers, scanners, graphic cards and video cards Telecommunication, PCs and servers Typical Application MIC2164 12V to 3.3V Efficiency 100 95 EFFICIENCY (%) 90 85 80 75 70 65 60 55 VIN=5V 50 MIC2164/-2/-3/C Synchronous Controllers Featuring Adaptive On-Time Control 0 4 8 12 16 20 OUTPUT CURRENT (A) Hyper Speed Control and Any Capacitor are trademarks of Micrel, Inc. Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com February 12, 2015 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Ordering Information Part Number Voltage(1) Switching Frequency Accuracy Junction Temperature Range Package Lead Finish MIC2164YMM Adjustable 300kHz ±1% –40° to +125°C 10-pin MSOP Pb-Free MIC2164-2YMM Adjustable 600kHz ±1% –40° to +125°C 10-pin MSOP Pb-Free MIC2164-3YMM Adjustable 1MHz ±1% –40° to +125°C 10-pin MSOP Pb-Free MIC2164CYMM Adjustable 270kHz ±3% –40° to +125°C 10-pin MSOP Pb-Free Note: 1. Other voltages are available. Contact Micrel for details. Pin Configuration 10-Pin MSOP (MM) Pin Description Pin Number Pin Name 1 HSD High-Side N-MOSFET Drain Connection (input): Power to the drain of the external high-side N-channel MOSFET. The HSD operating voltage range is from 3V to 28V. Input capacitors between HSD and the power ground (PGND) are required. 2 EN Enable (input): A logic level control of the output. The EN pin is CMOS-compatible. Logic high or floating = enable, logic low = shutdown. In the off state, supply current of the device is greatly reduced (typically 0.8mA). 3 FB Feedback (input): Input to the transconductance amplifier of the control loop. The FB pin is regulated to 0.8V. A resistor divider connecting the feedback to the output is used to adjust the desired output voltage. 4 GND 5 IN Input Voltage (input): Power to the internal reference and control sections of the MIC2164/-2/-3. The IN operating voltage range is from 3V to 5.5V. A 1µF and 0.1µF ceramic capacitors from IN to GND are recommended for clean operation. 6 DL Low-Side Drive (output): High-current driver output for external low-side MOSFET. The DL driving voltage swings from ground-to-IN. 7 8 February 12, 2015 PGND DH Pin Function Signal ground. GND is the ground path for the device input voltage VIN and the control circuitry. The loop for the signal ground should be separate from the power ground (PGND) loop. Power Ground. PGND is the ground path for the MIC2164/-2/-3 buck converter power stage. The PGND pin connects to the sources of low-side N-Channel MOSFETs, the negative terminals of input capacitors, and the negative terminals of output capacitors. The loop for the power ground should be as small as possible and separate from the Signal ground (GND) loop. High-Side Drive (output): High-current driver output for external high-side MOSFET. The DH driving voltage is floating on the switch node voltage (LX). It swings from ground to VIN minus the diode drop. Adding a small resistor between DH pin and the gate of the high-side N-channel MOSFETs can slow down the turn-on and turn-off time of the MOSFETs. 2 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Pin Description (Continued) Pin Number Pin Name 9 10 February 12, 2015 LX BST Pin Function Switch Node and Current Sense input: High current output driver return. The LX pin connects directly to the switch node. Due to the high speed switching on this pin, the LX pin should be routed away from sensitive nodes. LX pin also senses the current by monitoring the voltage across the low-side MOSFET during OFF time. In order to sense the current accurately, connect the low-side MOSFET drain to LX using a Kelvin connection. Boost (output): Bootstrapped voltage to the high-side N-channel MOSFET driver. A Schottky diode is connected between the IN pin and the BST pin. A boost capacitor of 0.1μF is connected between the BST pin and the LX pin. Adding a small resistor in series with the boost capacitor can slow down the turn-on time of high-side N-Channel MOSFETs. 3 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Absolute Maximum Ratings(2) Operating Ratings(3) IN, FB, EN to GND .......................................... –0.3V to +6V BST to LX ........................................................ –0.3V to +6V BST to GND .................................................. –0.3V to +37V DH to LX ............................................ –0.3V to (VBST + 0.3V) DL, COMP to GND .............................. –0.3V to (VIN + 0.3V) HSD to GND .................................................... –0.3V to 31V PGND to GND .............................................. –0.3V to +0.3V Junction Temperature .............................................. +150°C Storage Temperature (TS) ......................... –65°C to +150°C Lead Temperature (soldering, 10sec) ........................ 260°C Input Voltage (VIN) ............................................ 3.0V to 5.5V Supply Voltage (VHSD) ....................................... 3.0V to 28V Operating Temperature Range ................. –40°C to +125°C Junction Temperature (TJ) ........................ –40°C to +125°C Junction Thermal Resistance MSOP (θJA) ......................................................... 130.5°C/W Continuous Power Dissipation (TA = 70°C).............. 421mW (derate 5.6mW/°C above 70°C) Electrical Characteristics(5) VBST – VLX = 5V; TA = 25°C, bold values indicate –40°C≤ TA ≤ +125°C, unless noted. Parameter Condition Min. Typ. Max. Units General Operating Input Voltage (VIN)(6) 3.0 5.5 V HSD Voltage Range (VHSD) 3.0 28 V Quiescent Supply Current (VFB = 1.5V, output switching but excluding external MOSFET gate current) 1.4 3.0 mA Standby Supply Current(7) VIN = VBST = 5.5V, VHSD = 28, LX = unconnected, EN = GND 0.8 2 mA 2.7 3 V 2.4 Under-Voltage Lockout Trip Level UVLO Hysteresis 50 mV DC-DC Controller Output-Voltage Adjust Range (VOUT) 0.8 5.5 0°C ≤ TJ ≤ 85°C -1 1 −40°C ≤ TJ ≤ 125°C -2 2 TJ = 25°C (MIC2164C) -3 3 V Error Amplifier FB Regulation Voltage FB Input Leakage Current Current-Limit Threshold 5 500 VFB = 0.8V 103 130 162 VFB = 0V 19 48 77 VFB = 0.8V (MIC2164C) 95 130 170 VFB = 0V (MIC2164C) 15 48 80 % nA mV Notes: 2. Exceeding the absolute maximum ratings may damage the device. 3. The device is not guaranteed to function outside its operating ratings. 4. Devices are ESD sensitive. Handling precautions are recommended. Human body model, 1.5kΩ in series with 100pF. 5. Specification for packaged product only. 6. The application is fully functional at low IN (supply of the control section) if the external MOSFETs have enough low voltage VTH. 7. The current will come only from the internal 100kΩ pull-up resistor sitting on the EN Input and tied to IN. 8. Measured in test mode. 9. Measured at DH. The maximum duty cycle is limited by the fixed mandatory off time TOFF of 363ns typical. February 12, 2015 4 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Electrical Characteristics(5) (Continued) VBST – VLX = 5V; TA = 25°C, bold values indicate –40°C≤ TA ≤ +125°C, unless noted. Parameter Condition Min. Typ. Max. Units Soft-Start Soft-Start Period 6 ms Oscillator Switching Frequency(8) 0.202 0.225 0.45 0.75 MIC2164C MIC2164 MIC2164-2 MIC2164-3 0.27 0.3 0.6 1 0.338 0.375 0.75 1.25 MHz Maximum Duty Cycle MIC2164/ MIC2164C MIC2164-2 MIC2164-3 87 74 66 % Minimum Duty Cycle Measured at DH, VFB = 1V 0 % (9) FET Drives DH, DL Output Low Voltage ISINK = 10mA DH, DL Output High Voltage ISOURCE = 10mA 0.1 VIN − 0.1V or VBST − 0.1V V V DH On-Resistance, High State 2.1 3.3 Ω DH On-Resistance, Low State 1.8 3.3 Ω DL On-Resistance, High State 1.8 3.3 Ω DL On-Resistance, Low State 1.2 2.3 Ω LX Leakage Current VLX = 28V, VIN = 5.5V,VBST = 33.5V 50 µA HSD Leakage Current VLX = 28V, VIN = 5.5V,VBST = 33.5V 20 µA Thermal Protection Over-Temperature Shutdown 155 °C Over-Temperature Shutdown Hysteresis 10 °C 0.8 V Shutdown Control EN Logic Level Low 3V < VIN <5.5V EN Logic Level High 3V < VIN <5.5V 0.4 0.9 EN Pull-Up Current February 12, 2015 50 5 1.2 V µA Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Typical Characteristics 100 100 95 95 95 90 90 90 85 80 75 70 EFFICIENCY (%) 100 85 80 75 70 65 65 60 60 VIN=5V 55 80 75 70 65 60 VIN=5V 4 8 12 16 20 50 0 4 8 OUTPUT CURRENT (A) 12 16 0 20 95 90 90 90 75 70 EFFICIENCY (%) 100 95 EFFICIENCY (%) 100 95 80 85 80 75 70 80 75 70 65 60 VIN=5V 55 60 VIN=5V 50 3 6 9 12 15 50 0 2 4 OUTPUT CURRENT (A) 6 8 0 10 Feedback Voltage vs. Input Voltage 0.84 0.81 0.80 0.79 0.78 VHSD=12V VIN=5V 0.76 FEEDBACK VOLTAGE (V) 0.85 0.84 FEEDBACK VOLTAGE (V) 0.85 0.82 0.83 0.82 0.81 0.80 0.79 0.78 0.77 VHSD=12V 0.75 4 8 12 16 3.5 Feedback Voltage vs. Temperature 0.80 0.79 0.78 4 4.5 5 5.5 3 0.804 0.802 0.800 0.798 0.796 0.794 VIN=5V 0.792 -40 -20 0 20 40 60 80 TEMPERATURE (°C) 100 120 18 23 28 MIC2164-2 Switching Frequency vs. Load 700 340 330 320 310 300 290 280 VHSD=12V VIN=5V 270 680 660 640 620 600 580 560 540 VHSD=12V VIN=5V 520 250 500 0 4 8 12 OUTPUT CURRENT (A) February 12, 2015 13 MIC2164 Switching Frequency vs. Load 260 0.790 8 HSD VOLTAGE (V) SWITCHING FREQUENCY (kHz) SWITCHING FREQUENCY (kHz) 0.806 VIN=5V 0.77 350 0.808 FEEDBACK VOLTAGE (V) 0.81 INPUT VOLTAGE (V) OUTPUT CURRENT (A) 0.810 0.82 0.75 3 20 0.83 0.76 0.76 0.75 10 Feedback Voltage vs. HSD Voltage 0.84 0.83 8 6 OUTPUT CURRENT (A) 0.85 0 4 2 OUTPUT CURRENT (A) Feedback Voltage vs. Load 0.77 VIN=5V 55 55 50 15 85 65 65 12 MIC2164-3 12V to 3.3V Efficiency MIC2164-3 12V to 1.5V Efficiency 85 9 OUTPUT CURRENT (A) 100 0 6 3 OUTPUT CURRENT (A) MIC2164-2 12V to 3.3V Efficiency 60 VIN=5V 55 50 0 EFFICIENCY (%) 85 55 50 FEEDBACK VOLTAGE (V) MIC2164-2 12V to 1.5V Efficiency MIC2164 12V to 3.3V Efficiency EFFICIENCY (%) EFFICIENCY (%) MIC2164 12V to 1.5V Efficiency 6 16 20 0 3 6 9 12 15 OUTPUT CURRENT (A) Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Typical Characteristics (Continued) MIC2164-3 Switching Frequency vs. Load MIC2164 Switching Frequency vs. VIN 1120 1090 1060 1030 1000 970 940 VHSD=12V VIN=5V 910 880 700 SWITCHING FREQUENCY (kHz) 350 SWITCHING FREQUENCY (kHz) SWITCHING FREQUENCY (kHz) 1150 MIC2164-2 Switching Frequency vs. VIN 340 330 320 310 300 290 280 VHSD=12V 270 260 850 2 4 6 8 10 3.5 4 4.5 5 1030 1000 970 VHSD=12V 910 880 330 VIN=5V 320 310 300 290 280 270 260 5 3 5.5 8 13 23 VIN=5V 1030 1000 970 940 910 880 620 600 580 560 540 520 3 28 330 320 310 300 290 280 270 VIN=5V 640 620 600 580 560 540 VIN=5V 500 -40 -20 0 20 40 60 80 100 120 -40 135 1000 970 940 VIN=5V 120 105 90 75 60 45 30 880 15 850 0 20 40 60 80 TEMPERATURE (°C) February 12, 2015 100 120 CURRENT LIMIT THRESHOLD (mV) 150 135 CURRENT LIMIT THRESHOLD (mV) 150 0 0 20 40 60 80 100 120 Current Limit Threshold vs. Temperature 1120 -20 -20 TEMPERATURE (°C) 1150 1030 28 660 Current Limit Threshold vs. Feedback Voltage Percentage 1060 23 680 TEMPERATURE (°C) 1090 18 520 MIC2164-3 Switching Frequency vs. Temperature -40 13 700 340 HSD VOLTAGE (V) 910 8 MIC2164-2 Switching Frequency vs. Temperature 250 23 VIN=5V VOUT=2.5V 640 HSD VOLTAGE (V) 260 18 5.5 500 28 SWITCHING FREQUENCY (kHz) SWITCHING FREQUENCY (kHz) SWITCHING FREQUENCY (kHz) 1090 850 SWITCHING FREQUENCY (kHz) 18 350 1120 5 660 MIC2164 Switching Frequency vs. Temperature 1150 4.5 680 HSD VOLTAGE (V) MIC2164-3 Switching Frequency vs. VHSD 13 4 700 340 INPUT VOLTAGE (V) 8 3.5 MIC2164-2 Switching Frequency vs. VHSD 250 850 3 VHSD=12V 540 INPUT VOLTAGE (V) SWITCHING FREQUENCY (kHz) SWITCHING FREQUENCY (kHz) SWITCHING FREQUENCY (kHz) 1060 1060 560 3 350 1090 4.5 580 MIC2164 Switching Frequency vs. VHSD 1120 4 600 5.5 1150 3.5 620 INPUT VOLTAGE (V) MIC2164-3 Switching Frequency vs. VIN 3 640 500 3 OUTPUT CURRENT (A) 940 660 520 250 0 680 VIN=5V 120 105 90 VFB=0.8V 75 VFB=0V 60 45 30 15 0 0 10 20 30 40 50 60 70 80 90 Feedback Voltage Percentage (%) 7 100 -40 -20 0 20 40 60 80 100 120 TEMPERATURE (°C) Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Typical Characteristics (Continued) Quiescent Supply Current vs. Input Voltage 2 QUIESCENT SUPPLY CURRENT (mA) 1.8 1.6 1.4 1.2 1 0.8 0.6 0.4 0.2 0 3 3.5 4 4.5 5 5.5 INPUT VOLTAGE (V) February 12, 2015 8 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Functional Characteristics February 12, 2015 9 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Functional Diagram (Continued) February 12, 2015 10 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Functional Characteristics (Continued) February 12, 2015 11 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Functional Diagram Figure 1. MIC2164/-2/-3 Block Diagram February 12, 2015 12 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Functional Description The MIC2164/-2/-3 are parts of an adaptive on-time synchronous buck controller family built for low cost and high performance. They are designed for a wide input voltage range, from 3V to 28V, and high output power buck converters. An estimated-ON-time method is applied to the MIC2164/-2/-3 to obtain a constant switching frequency and to simplify the control compensation. The over-current protection is implemented without the use of an external sense resistor. It includes an internal soft-start function which reduces the power supply input surge current at start-up by controlling the output voltage rise time. The maximum duty cycle is obtained from the 363ns TOFF(min): Dmax = VOUT VHSD × f sw = 1− 363ns TS Eq.2 It is not recommended to use MIC2164/-2/-3 with an OFF time close to TOFF(min) at the steady state. Also, as VOUT increases, the internal ripple injection will increase and reduce the line regulation performance. Therefore, the maximum output voltage of the MIC2164/-2/-3 should be limited to 5.5V. If a higher output voltage is required, use the MIC2176 instead. Please refer to “Setting Output Voltage” subsection in Application Information for more details. The estimated-ON-time method results in a constant switching frequency in MIC2164/-2/-3. The actual ON time is varied with the different rising and falling time of the external MOSFETs. Therefore, the type of the external MOSFETs, the output load current, and the control circuitry power supply VIN will modify the actual ON time and the switching frequency. Also, the minimum TON results in a lower switching frequency in the high VHSD and low VOUT applications, such as 24V to 1.0V MIC2164-3 application. The minimum TON measured on the MIC2164 evaluation board is about 138ns. During the load transient, the switching frequency is changed due to the varying OFF time. Eq. 1 where VOUT is the output voltage, VHSD is the power stage input voltage, and fSW is the switching frequency (300kHz for MIC2164, 600kHz for MIC2164-2, and 1MHz for MIC2164-3). To illustrate the control loop, the steady-state scenario and the load transient scenario are analyzed. For easy analysis, the gain of the gm amplifier is assumed to be 1. With this assumption, the inverting input of the error comparator is the same as the FB voltage. Figure 2 shows the MIC2164/-2/-3 control loop timing during the steady-state. During the steady-state, the gm amplifier senses the FB voltage ripple, which is proportional to the output voltage ripple and the inductor current ripple, to trigger the ON-time period. The ON time is predetermined by the estimation. The ending of OFF time is controlled by the FB voltage. At the valley of the FB voltage ripple, which is below than VREF, OFF period ends and the next ON-time period is triggered through the control logic circuitry. When the MIC2164/-2/-3 enters the OFF-time period, the DH pin becomes logic low and the DL pin is logic high. In most cases, the OFF-time period length is dependent on the FB voltage. When the FB voltage decreases and the output voltage of the gm amplifier drops below 0.8V, the ON-time period is triggered and the OFF-time period ends. If the OFF-time period, determined by the FB voltage, is less than the minimum OFF time (TOFF(min)), which is about 363ns typical, then the MIC2164/-2/-3 control logic will apply the TOFF(min) instead. TOFF(min) is required by the BST charging. February 12, 2015 TS where TS = 1/fSW. Theory of Operation Figure 1 illustrates the block diagram for the control loop. The output voltage variation will be sensed by the MIC2164/-2/-3 feedback pin FB via the voltage divider R1 and R2, and compared to a 0.8V reference voltage VREF at the error comparator through a low gain transconductance (gm) amplifier, the amplifier improves the MIC2164/-2/-3 converter output voltage regulation. If the FB voltage decreases and the output of the gm amplifier is below 0.8V, The error comparator will trigger the control logic and generate an ON-time period, in which DH pin is logic high and DL pin is logic low. The ON-time period length is predetermined by the “FIXED TON ESTIMATION” circuitry: TON(estimated) = TS − TOFF(min) 13 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C The MIC2164/-2/-3 family has its own stability concern. the FB voltage ripple should be in phase with the inductor current ripple and large enough to be sensed by the gm amplifier and the error comparator. The recommended minimum FB voltage ripple is 20mV. If a low ESR output capacitor is selected, then the FB voltage ripple may be too small to be sensed by the gm amplifier and the error comparator. Also, the output voltage ripple and the FB voltage ripple are not in phase with the inductor current ripple if the ESR of the output capacitor is very low. Therefore, the ripple injection is required for a low ESR output capacitor. Please refer to “Ripple Injection” subsection in “Application Information” for more details about the ripple injection. Soft-Start Soft-start reduces the power supply input surge current at startup by controlling the output voltage rise time. The input surge appears while the output capacitor is charged up. A slower output rise time will draw a lower input surge current. Figure 2. MIC2164/-2/-3 Control Loop Timing Figure 3 shows the load transient scenario of the MIC2164/-2/-3 converter. The output voltage will drop due to a sudden load increase, which causes the FB voltage to be less than VREF. This will cause the error comparator to the trigger ON-time period. At the end of the ON-time period a minimum OFF time, TOFF(min), is generated to charge the BST since the FB voltage is still below the VREF. The next ON-time period is triggered due to the low FB voltage. The switching frequency changes during the load transient. With the varying duty cycle and switching frequency, the output recovery time is fast and the output voltage deviation is small in the MIC2164/-2/-3 converter. MIC2164/-2/-3 implements an internal digital soft-start by making the 0.8V reference voltage VREF ramp from 0 to 100% in about 6ms with a 9.7mV step. The output voltage is controlled to increase slowly by a stair-case VREF ramp. Once the soft-start ends, the related circuitry is disabled to reduce the current consumption. To make the soft-start function behavior correctly, the VIN should not be powered up before VHSD. Current Limit The MIC2164/-2/-3 uses the RDS(ON) of the low-side power MOSFET to sense over-current conditions. The lowerside MOSFET is used because it displays much lower parasitic oscillations during switching then the high-side MOSFET. Using the low-side MOSFET RDS(ON) as a current sense is an excellent method for circuit protection. This method will avoid adding cost, board space and power losses taken by discrete current sense resistors. In each switching cycle of the MIC2164/-2/-3 converter, the inductor current is sensed by monitoring the low-side MOSFET in the OFF period. The sensed voltage is compared with a current-limit threshold voltage VCL after a blanking time of 150ns. If the sensed voltage is over VCL, which is 130mV typical at 0.8V feedback voltage, the MIC2164/-2/-3 turns off the high-side MOSFET and a soft-start sequence is trigged. This mode of operation is called the “hiccup mode” and its purpose is to protect the downstream load in case of a hard short. The current limit threshold VCL has a fold back characteristics related to the FB voltage. Please refer to the Typical Characteristics for the curve of VCL vs. FB voltage. The circuit in Figure 4 illustrates the MIC2164/-2/-3 current limiting circuit. Figure 3. MIC2164/-2/-3 Load-Transient Response Unlike the current-mode control, MIC2164/-2/-3 uses the output voltage ripple, which is proportional to the inductor current ripple if the ESR of the output capacitor is large enough, to trigger an ON-time period. The predetermined ON time makes the MIC2164/-2/-3 control loop have an advantage as the adaptive on-time mode control. The slope compensation, which is necessary for the currentmode control, is not required in the MIC2164/-2/-3 family. February 12, 2015 14 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C MOSFET Gate Drive The MIC2164/-2/-3 high-side drive circuit is designed to switch an N-Channel MOSFET. Figure 1 shows a bootstrap circuit, consisting of D1 (a Schottky diode is recommended) and CBST. This circuit supplies energy to the high-side drive circuit. The CBST capacitor is charged while the low-side MOSFET is on and the voltage on the LX pin is approximately 0V. When the high-side MOSFET driver is turned on, energy from CBST is used to turn the MOSFET on. As the high-side MOSFET turns on, the voltage on the LX pin increases to approximately VHSD. Diode D1 is reversed biased and the CBST floats high while continuing to keep the high-side MOSFET on. The bias current of the high-side driver is less than 10mA so a 0.1μF to 1μF capacitor is sufficient to hold the gate voltage with minimal droop for the power stroke, highside switching cycle, (i.e. ΔBST = 10mA x 3.33μs/0.1μF = 333mV) for MIC2164. When the low-side MOSFET is turned back on, CBST is recharged through D1. A small resistor (RG), which is in series with CBST, can slow down the turn-on time of the high-side N-channel MOSFET. Figure 4 MIC2164/-2/-3 Current Limiting Circuit Using the typical VCL value of 130mV, the current limit value is roughly estimated as: ICL ≈ 130mV R DS(ON) Eq. 3 The drive voltage is derived from the supply voltage (VIN). The nominal low-side gate drive voltage is VIN and the nominal high-side gate drive voltage is approximately VIN – VDIODE, where VDIODE is the voltage drop across D1. There is a delay for approximately 30ns between the high-side and low-side driver transitions, which used to prevent current from simultaneously flowing unimpeded through both MOSFETs. For designs where the current ripple is significant compared to the load current (IOUT), or for low duty cycle operation, calculating the current limit (ICL) should take into account that one is sensing the peak inductor current and that there is a blanking delay of approximately 150ns. ICL = V * TDLY ΔIL(pp) 130mV + OUT − R DS(ON) L 2 ΔIL(pp) = VOUT ⋅ (1 − D) f SW ⋅L Eq. 4 Eq. 5 where VOUT = The output voltage TDLY = Current limit blanking time, 150ns typical ΔIL(pp) = Inductor current ripple peak-to-peak value D = Duty Cycle fSW = Switching frequency The MOSFET RDS(ON) varies 30 to 40% with temperature; therefore, it is recommended to add a 50% margin to ICL in the above equation to avoid false current limiting due to increased MOSFET junction temperature rise. It is also recommended to connect LX pin directly to the drain of the low-side MOSFET to accurately sense the MOSFETs RDS(ON). February 12, 2015 15 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C The low-side MOSFET is turned on and off at VDS = 0 because an internal body diode or external freewheeling diode is conducting during this time. The switching loss for the low-side MOSFET is usually negligible. Also, the gate-drive current for the low-side MOSFET is more accurately calculated using CISS at VDS = 0 instead of gate charge. Application Information MOSFET Selection The MIC2164/-2/-3 controller operates between power stage input voltages of 3V to 28V, and has an external 3V to 5.5V VIN to turn the external N-Channel which powers the MOSFETs for the high- and low-side switches. For applications where VIN < 5V, it is necessary that the power MOSFETs used are sub-logic level and are in full conduction mode for a 2.5V VGS. For applications where VIN > 5V, the logic-level MOSFETs with a VGS of 4.5V must be used. For the low-side MOSFET: IG[low -side] (avg) = C ISS × VGS × f SW Since the current from the gate drive comes from the VIN, the power dissipated in the MIC2164/-2/-3 due to gate drive is: There are different criteria for choosing the high-side and low-side MOSFETs. These differences are more significant at lower duty cycles such as 12V to 1.8V conversion. In such an application, the high-side MOSFET is required to switch as quickly as possible to minimize transition losses, whereas the low-side MOSFET can switch slower, but must handle larger RMS currents. When the duty cycle approaches 50%, the current carrying capability of the high-side MOSFET starts to become critical. PGATEDRIVE = VIN .(IG[high-side] (avg) + IG[low -side] (avg)) Eq. 8 A convenient figure of merit for switching MOSFETs is the on resistance times the total gate charge RDS(ON) × QG. Lower numbers translate into higher efficiency. Low gate-charge logic-level MOSFETs are a good choice for use with the MIC2164/-2/-3. Also, the RDS(ON) of the lowside MOSFET will determine the current limit value. Please refer to “Current Limit” subsection is “Functional Description” for more details. It is important to note that the on-resistance of a MOSFET increases with increasing temperature. A 75°C rise in junction temperature will increase the channel resistance of the MOSFET by 50% to 75% of the resistance specified at 25°C. This change in resistance must be accounted for when calculating MOSFET power dissipation and in calculating the value of current limit. Total gate charge is the charge required to turn the MOSFET on and off under specified operating conditions (VDS and VGS). The gate charge is supplied by the MIC2164/-2/-3 gate-drive circuit. At 300kHz switching frequency and above, the gate charge can be a significant source of power dissipation in the MIC2164/2/-3. At low output load, this power dissipation is noticeable as a reduction in efficiency. The average current required to drive the high-side MOSFET is: IG[high-side] (avg) = Q G × fSW Eq. 7 Parameters that are important to MOSFET switch selection are: • Voltage rating • On-resistance • Total gate charge The voltage ratings for the high-side and low-side MOSFETs are essentially equal to the power stage input voltage VHSD. A safety factor of 20% should be added to the VDS(max) of the MOSFETs to account for voltage spikes due to circuit parasitic elements. Eq. 6 The power dissipated in the MOSFETs is the sum of the conduction losses during the on-time (PCONDUCTION) and the switching losses during the period of time when the MOSFETs turn on and off (PAC). where: IG[high-side](avg) = Average high-side MOSFET gate current PSW = PCONDUCTION + PAC QG = Total gate charge for the high-side MOSFET taken from the manufacturer’s data sheet for VGS = VIN. Eq. 9 2 fSW = Switching Frequency PCONDUCTION = ISW(RMS) * R DS(ON) Eq. 10 PAC = PAC(off ) + PAC(on) Eq. 11 where: RDS(ON) = on-resistance of the MOSFET switch D = Duty Cycle = VOUT / VHSD February 12, 2015 16 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Making the assumption that the turn-on and turn-off transition times are equal; the transition times can be approximated by: tT = The peak-to-peak inductor current ripple is: DIL(PP ) = C ISS × VIN + C OSS × VHSD IG VOUT × ( VHSD(max) − VOUT ) VHSD(max) × f SW × L Eq. 15 Eq. 12 The peak inductor current is equal to the average output current plus one half of the peak-to-peak inductor current ripple. where: CISS and COSS are measured at VDS = 0 IL(PK) = IOUT(max) + 0.5 × ∆IL(PP) Eq. 16 IG = gate-drive current The total high-side MOSFET switching loss is: PAC = (VHSD + VD ) × IPK × t T × f SW The RMS inductor current is used to calculate the I2R losses in the inductor. Eq. 13 IL(RMS) = IOUT(max) 2 + where: tT = Switching transition time VD = Body diode drop (0.5V) The high-side MOSFET switching losses increase with the switching frequency and the input voltage VHSD. The low-side MOSFET switching losses are negligible and can be ignored for these calculations. ( ) VHSD(max) × f SW × 20% × IOUT(max) 2 PINDUCTORCu=IL(RMS) × RWINDING Eq. 18 The resistance of the copper wire, RWINDING, increases with the temperature. The value of the winding resistance used should be at the operating temperature. Eq. 14 RWINDING = RWINDING(20°C`) × (1 + 0.0042 × (TH – T20°C)) Eq. 19 where: where: TH = temperature of wire under full load T20°C = ambient temperature RWINDING(20°C) = room temperature winding resistance (usually specified by the manufacturer) fSW = switching frequency 20% = ratio of AC ripple current to DC output current VHSD(max) = maximum power stage input voltage February 12, 2015 Eq. 17 Low-cost iron powder cores may be used, but the increase in core loss will reduce the efficiency of the power supply. This is especially noticeable at low output power. The winding resistance decreases efficiency at the higher output current levels. The winding resistance must be minimized even at the expense of a larger inductor. The power dissipated in the inductor is equal to the sum of the core and copper losses. At higher output loads, the core losses are usually insignificant and can be ignored. At lower output currents, the core losses can be a significant contributor. Core loss information is usually available from the magnetics vendor. Copper loss in the inductor is calculated by the equation below: Inductor Selection Values for inductance, peak, and RMS currents are required to select the output inductor. The input and output voltages and the inductance value determine the peak-to-peak inductor ripple current. Generally, higher inductance values are used with higher input voltages. Larger peak-to-peak ripple currents will increase the power dissipation in the inductor and MOSFETs. Larger output ripple currents will also require more output capacitance to smooth out the larger ripple current. Smaller peak-to-peak ripple currents require a larger inductance value and therefore a larger and more expensive inductor. A good compromise between size, loss and cost is to set the inductor ripple current to be equal to 20% of the maximum output current. The inductance value is calculated by the equation below: VOUT × VHSD(max) − VOUT 12 Maximizing efficiency requires both the proper selection of core material and the minimizing of the winding resistance. The high frequency operation of the MIC2164/-2/-3 requires the use of ferrite materials for all but the most cost sensitive applications. fSW = Switching Frequency L= ΔIL(PP)2 17 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Output Capacitor Selection The type of the output capacitor is usually determined by its ESR (equivalent series resistance). Voltage and RMS current capability are two other important factors for selecting the output capacitor. Recommended capacitors are tantalum, low-ESR aluminum electrolytic, OS-CON and POSCAPS. The output capacitor’s ESR is usually the main cause of the output ripple. The output capacitor ESR also affects the control loop from a stability point of view. The maximum value of ESR is calculated: ESR COUT ≤ ΔVOUT(pp) PDISS(COUT) = ICOUT(RMS) 2 × ESR COUT Input Capacitor Selection The input capacitor for the power stage input VHSD should be selected for ripple current rating and voltage rating. Tantalum input capacitors may fail when subjected to high inrush currents, caused by turning the input supply on. A tantalum input capacitor’s voltage rating should be at least two times the maximum input voltage to maximize reliability. Aluminum electrolytic, OS-CON, and multilayer polymer film capacitors can handle the higher inrush cu +rrents without voltage de-rating. The input voltage ripple will primarily depend upon the input capacitor’s ESR. The peak input current is equal to the peak inductor current, so: Eq. 20 ΔIL(PP) Eq. 23 where: ΔVOUT(pp) = peak-to-peak output voltage ripple ΔIL(PP) = peak-to-peak inductor current ripple ∆VIN = IL(PK ) × ESR CIN The total output ripple is a combination of the ESR and output capacitance. The total ripple is calculated below: 2 ΔIL(PP) + ΔIL(PP) ⋅ ESR COUT ΔVOUT(pp) = C ⋅ f ⋅ 8 OUT SW ( )2 Eq. 24 The input capacitor must be rated for the input current ripple. The RMS value of input capacitor current is determined at the maximum output current. Assuming the peak-to-peak inductor current ripple is low: Eq. 21 where: D = Duty cycle COUT = Output capacitance value fSW = Switching frequency ICIN(RMS) ≈ IOUT(max) × D × (1 − D) Eq. 25 The power dissipated in the input capacitor is: As described in the “Theory of Operation” subsection in Functional Description, MIC2164/-2/-3 requires at least 20mV peak-to-peak ripple at the FB pin to make the gm amplifier and the error comparator to behavior properly. Also, the output voltage ripple should be in phase with the inductor current. Therefore, the output voltage ripple caused by the output capacitor COUT should be much smaller than the ripple caused by the output capacitor ESR. If low ESR capacitors are selected as the output capacitors, such as ceramic capacitors, a ripple injection method is applied to provide the enough FB voltage ripples. Please refer to the “Ripple Injection” subsection for more details. PDISS(CIN) = ICIN(RMS)2×ESRCIN Eq. 26 The voltage rating of the capacitor should be twice the output voltage for a tantalum and 20% greater for aluminum electrolytic or OS-CON. The output capacitor RMS current is calculated below: ICOUT(RMS) = ΔIL(PP) Eq. 22 12 The power dissipated in the output capacitor is: February 12, 2015 18 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C is so small that the gm amplifier and error comparator could not sense it, then the MIC2164/-2/-3 will lose control and the output voltage will not be regulated. In order to have some amount of FB voltage ripple, the ripple injection method is applied for low output voltage ripple applications. External Schottky Diode (Optional) An external freewheeling diode, which is generally not necessary, can be used to keep the inductor current flow continuous while both MOSFETs are turned off. This dead time prevents current from flowing unimpeded through both MOSFETs and is typically 30ns. The diode conducts twice during each switching cycle. Although the average current through this diode is small, the diode must be able to handle the peak current. ID(avg) = IOUT ⋅ 2 ⋅ 30ns ⋅ f SW The applications are divided into three situations according to the amount of the FB voltage ripple: 1. Enough ripple at the FB voltage due to the large ESR of the output capacitors. As shown in Figure 5, the converter is stable without any adding in this situation. The FB voltage ripple is: Eq. 27 The reverse voltage requirement of the diode is: ΔVFB(pp) = R2 ⋅ ESR COUT ⋅ ΔIL (pp) R1 + R2 Eq. 29 VDIODE(rrm) = VHSD where ΔIL(pp) is the peak-to-peak value of the inductor current ripple. The power dissipated by the Schottky diode is: PDIODE = ID(avg) × VF 2. Inadequate ripple at the FB voltage due to the small ESR of the output capacitors. Eq. 28 The output voltage ripple is fed into the FB pin through a feedforward capacitor Cff in this situation, as shown in Figure 6. The typical Cff value is between 1nF to 100nF. With the feedforward capacitor, the FB voltage ripple is very close to the output voltage ripple: where, VF = forward voltage at the peak diode current. The external Schottky diode is not necessary for the circuit operation since the low-side MOSFET contains a parasitic body diode. The external diode will improve efficiency and decrease the high frequency noise. If the MOSFET body diode is then used, it must be rated to handle the peak and average current. The body diode has a relatively slow reverse recovery time and a relatively high forward voltage drop. The power lost in the diode is proportional to the forward voltage drop of the diode. As the high-side MOSFET starts to turn on, the body diode becomes a short circuit for the reverse recovery period, dissipating additional power. The diode recovery and the circuit inductance will cause ringing during the high-side MOSFET turn-on. ΔVFB(pp) ≈ ESR ⋅ ΔIL (pp) Eq. 30 3. Invisible ripple at the FB voltage is due to the very low ESR of the output capacitors. An external Schottky diode conducts at a lower forward voltage preventing the body diode in the MOSFET from turning on. The lower forward voltage drop dissipates less power than the body diode. The lack of a reverse recovery mechanism in a Schottky diode causes less ringing and less power loss. Depending upon the circuit components and operating conditions, an external Schottky diode will give a ½ to 1% improvement in efficiency. Ripple Injection The minimum FB voltage ripple requested by the MIC2164/-2/-3 gm amplifier and error comparator is 20mV (100mV maximum). However, the output voltage ripple is generally designed as 1% to 2% of the output voltage. For a low output voltage, such as 1V output, the output voltage ripple is only 10mV to 20mV, and the FB voltage ripple is less than 20mV. If the FB voltage ripple February 12, 2015 19 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C In the formula (29) and (30), it is assumed that the time constant associated with Cff must be much greater than the switching period: 1 T = << 1 fsw × τ τ If the voltage divider resistors R1 and R2 are in the kΩ range, a Cff of 1nF to 100nF can easily satisfy the large time constant consumption. Also, a 100nF injection capacitor Cinj is used in order to be considered as short for a wide range of the frequencies. Figure 5. Enough Ripple at FB The process of sizing the ripple injection resistor and capacitors is: Step 1. Select Cff to feed all output ripples into the feedback pin and make sure the large time constant assumption is satisfied. Typical choice of Cff is 1nF to 100nF if R1 and R2 are in kΩ range. Figure 6. Inadequate Ripple at FB Step 2. Select Rinj according to the expected feedback voltage ripple. According to Equation 30: K div = In this situation, the output voltage ripple is less than 20mV. Therefore, additional ripple is injected into the FB pin from the switching node LX via a resistor Rinj and a capacitor Cinj, as shown in Figure 7. The injected ripple is: K div = VHSD ⋅ f SW ⋅ τ D ⋅ (1 − D) Eq. 32 Then the value of Rinj is obtained as: Figure 7. Invisible Ripple at FB ΔVFB(pp) DVFB(pp ) R inj = (R1 // R2) ⋅ ( 1 − 1) K div Eq. 33 Step 3. Select Cinj as 100nF, which could be considered as short for a wide range of the frequencies. 1 = VHSD × K div × D × (1- D) × f SW × τ R1//R2 Rinj + R1//R2 (30) Eq. 31 where VHSD = Power stage input voltage at HSD pin D = Duty Cycle fSW = switching frequency τ = (R1// R2 // Rinj) ⋅ Cff February 12, 2015 20 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Setting Output Voltage The MIC2164/-2/-3 requires two resistors to set the output voltage, as shown in Figure 8: Figure 9. Internal Ripple Injection Figure 8. Voltage-Divider Configuration The output voltage is determined by the equation: VOUT = VREF × (1 + R1 ) R2 Eq. 34 where VREF = 0.8V. A typical value of R1 can be between 3kΩ and 10kΩ. If R1 is too large, it may allow noise to be introduced into the voltage feedback loop. If R1 is too small, it will decrease the efficiency of the power supply, especially at light loads. Once R1 is selected, R2 can be calculated using: R2 = VREF × R1 VOUT − VREF Eq. 35 In addition to the external ripple injection added at the FB pin, internal ripple injection is added at the inverting input of the comparator inside the MIC2164/-2/-3, as shown in Figure 7. The inverting input voltage VINJ is clamped to 1.2V. As VOUT is increased, the swing of VINJ will be clamped. The clamped VINJ reduces the line regulation because it is reflected back as a DC error on the FB terminal. Therefore, the maximum output voltage of the MIC2164/-2/-3 should be limited to 5.5V to avoid this problem. If a higher output voltage is required, use the MIC2176 instead. February 12, 2015 21 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Inductor PCB Layout Guideline • Keep the inductor connection to the switch node (LX) short. • Do not route any digital lines underneath or close to the inductor. • Keep the switch node (LX) away from the feedback (FB) pin. • The LX pin should be connected directly to the drain of the low-side MOSFET to accurate sense the voltage across the low-side MOSFET. • To minimize noise, place a ground plane underneath the inductor. Warning!!! To minimize EMI and output noise, follow these layout recommendations. PCB Layout is critical to achieve reliable, stable and efficient performance. A ground plane is required to control EMI and minimize the inductance in power, signal and return paths. The following guidelines should be followed to insure proper operation of the MIC2164/-2/-3 converter. IC • Place the IC and MOSFETs close to the point of load (POL). • Use fat traces to route the input and output power lines. • Signal and power grounds should be kept separate and connected at only one location. Output Capacitor • Use a wide trace to connect the output capacitor ground terminal to the input capacitor ground terminal. • Phase margin will change as the output capacitor value and ESR changes. Contact the factory if the output capacitor is different from what is shown in the BOM. • The feedback trace should be separate from the power trace and connected as close as possible to the output capacitor. Sensing a long high current load trace can degrade the DC load regulation. Input Capacitor • Place the HSD input capacitor next. • Place the HSD input capacitors on the same side of the board and as close to the MOSFETs as possible. • Keep both the HSD and PGND connections short. • Place several vias to the ground plane close to the HSD input capacitor ground terminal. • Use either X7R or X5R dielectric input capacitors. Do not use Y5V or Z5U type capacitors. • Do not replace the ceramic input capacitor with any other type of capacitor. Any type of capacitor can be placed in parallel with the input capacitor. • If a Tantalum input capacitor is placed in parallel with the input capacitor, it must be recommended for switching regulator applications and the operating voltage must be derated by 50%. • In “Hot-Plug” applications, a Tantalum or Electrolytic bypass capacitor must be used to limit the over-voltage spike seen on the input supply with power is suddenly applied. • An additional Tantalum or Electrolytic bypass input capacitor of 22uF or higher is required at the input power connection. • The 1µF and 0.1µF capacitors, which connect to the VIN terminal, must be located right at the IC. The VIN terminal is very noise sensitive and placement of the capacitor is very critical. Connections must be made with wide trace. February 12, 2015 Schottky Diode (Optional) • Place the Schottky diode on the same side of the board as the MOSFETs and HSD input capacitor. • The connection from the Schottky diode’s Anode to the input capacitors ground terminal must be as short as possible. • The diode’s cathode connection to the switch node (LX) must be keep as short as possible. RC Snubber • Place the RC snubber on the same side of the board and as close to the MOSFETs as possible. 22 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Evaluation Board Schematic Schematic of MIC2164 20A Evaluation Board February 12, 2015 23 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Bill of Materials Item C1 C2,C3 Part Number EPCOS 222215095001E3 Vishay(11) 1210YD226KAT2A AVX(12) GRM32ER61C226KE20L 06035C104KAT2A GRM188R71H104KA93D C1608X7R1H104K 0805ZD105KAT2A C7 C8 C11 C12 GRM219R61A105KC01D Murata(13) Murata 2 0.1µF Ceramic Capacitor, X7R, Size 0603, 50V 5 AVX Murata AVX Murata C2012X7R1A225K TDK 06035C102KAT2A AVX Murata C1608X7R1H102K TDK 06035C223KAZ2A AVX GRM188R71H223K Murata C1608X7R1H223K TDK C13, C15 C17 16ME1000WG SANYO(15) SD103BWS-7 Diodes Inc(16) SD103BWS 22µF Ceramic Capacitor, X5R, Size 1210, 16V TDK 12106D107MAT2A D1 1 AVX 0805ZC225MAT2A GRM32ER60J107ME20L 220µF Aluminum Capacitor, SMD, 35V TDK TDK GRM188R71H102KA01D Qty. (14) C2012X5R1A105K GRM21BR71A225KA01L Description (10) B41125A7227M C3225X5R1C226K C6, C9, C10, C14, C16 Manufacturer AVX Murata Vishay 1µF Ceramic Capacitor, X5R, Size 0805, 10V 1 2.2µF Ceramic Capacitor, X7R, Size 0805, 10V 1 1nF Ceramic Capacitor, X7R, Size 0603, 50V 1 22nF Ceramic Capacitor, X7R, Size 0603, 50V 1 100µF Ceramic Capacitor, X5R, Size 1210, 6.3V 2 1000µF Aluminum Capacitor, 16V 1 Small Signal Schottky Diode 1 L1 CDEP147NP-1R5M Sumida(17) 1.5µH Inductor, 27.2A Saturation Current 1 Q1, Q4 FDMS7672 Fairchild(18) 30V N-Channel MOSFET 6.9mΩ RDS(ON) @ 4.5V 2 Q2, Q3 FDS8672S Fairchild 30V N-Channel MOSFET 7mΩ RDS(ON) @ 4.5V 2 Notes: 10. EPCOS: www.epcos.com. 11. Vishay: www.vishay.com. 12. AVX: www.avx.com. 13. Murata: www.murata.com. 14. TDK: www.tdk.com. 15. Sanyo: www.sanyo.com. 16. Diodes Inc: www.diodes.com. 17. Sumida: www.sumida.com. 18. Fairchild: www.fairchildsemi.com. February 12, 2015 24 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Bill of Materials (Continued) Item Part Number R1 CRCW06032R21FKEA Vishay/Dale 2.21Ω Resistor, Size 0603, 1% 1 R2 CRCW06031R21FKEA Vishay/Dale 1.21Ω Resistor, Size 0603, 1% 1 R3,R4 CRCW060310K0FKEA Vishay/Dale 10kΩ Resistor, Size 0603, 1% 2 R5 CRCW060320R0FKEA Vishay/Dale 20Ω Resistor, Size 0603, 1% 1 R6 CRCW06033K24FKEA Vishay/Dale 3.24kΩ Resistor, Size 0603, 1% 1 U1 (20) U2 MIC2164YMM MIC5233-5.0YM5 Manufacturer (19) MICREL INC MICREL INC Description Qty. 300kHz Buck Controller 1 LDO 1 Notes: 19. Micrel, Inc.: www.micrel.com. 20. Optional: Required if 5V supply is not available in the system. February 12, 2015 25 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C PCB Layout Recommendations MIC2164 20A Evaluation Board Top Layer MIC2164 20A Evaluation Board Bottom Layer February 12, 2015 26 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C PCB Layout Recommendations (Continued) MIC2164 20A Evaluation Board Mid-Layer 1 MIC2164 20A Evaluation Board Mid-Layer 2 February 12, 2015 27 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Application Schematics MIC2164 12V to 3.3V @ 20A Buck Converter February 12, 2015 28 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Bill of Materials (MIC2164 12V to 3.3V @ 20A) Item Part Number C1, C8, C17, C19 06035C104KAT AVX 0.1µF Ceramic Capacitor, X7R, Size 0603, 50V 4 C2 0805ZD225MAT AVX 2.2µF Ceramic Capacitor, X5R, Size 0805, 10V 1 C3 222215095001 220µF Aluminum Capacitor, SMD, 35V 1 C4, C5, C6 1210YD226MAT AVX 22µF Ceramic Capacitor, X5R, Size 1210, 16V 3 C9 0805ZD105KAT AVX 1µF Ceramic Capacitor, X5R, Size 0805, 10V 1 C10 06035C223KAT AVX 22nF Ceramic Capacitor, X7R, Size 0603, 50V 1 C11 16ME1000WGL Sanyo 1000µF Aluminum Capacitor, 16V 1 C12 12106D107MAT AVX 100µF Ceramic Capacitor, X5R, Size 1210, 6.3V 1 C15 06035C102KAT AVX 1nF Ceramic Capacitor, X7R, Size 0603, 50V 1 D1 SD103BWS Vishay Small Signal Schottky Diode 1 L1 CDEP147NP-1R5M Sumida 1.5µH Inductor, 27.2A Saturation Current 1 Q1, Q4 FDMS7672 Fairchild 30V N-Channel MOSFET 6.9mΩ RDS(ON) @ 4.5V 2 Q2, Q3 FDS8672S Fairchild 30V N-Channel MOSFET 7mΩ RDS(ON) @ 4.5V 2 R1 CRCW06032R21FKEY3 Vishay Dale 2.21Ω Resistor, Size 0603, 1% 1 R5 CRCW06031R21FKEY3 Vishay Dale 1.21Ω Resistor, Size 0603, 1% 1 R6, R9 CRCW06031002FKEY3 Vishay Dale 10k Resistor, Size 0603, 1% 2 R15 CRCW06033241FKEY3 Vishay Dale 3.24k Resistor, Size 0603 1% 1 U1 MIC2164YMM Micrel. Inc. 300kHz Buck Controller 1 U2 MIC5233-5.0YM5 Micrel. Inc. LDO 1 February 12, 2015 Manufacturer Vishay Description 29 Qty. Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C MIC2164 12V to 1.8V @ 10A Buck Converter February 12, 2015 30 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Bill of Materials (MIC2164 12V to 1.8V @ 10A) Item Part Number C1, C8, C17, C19 06035C104KAT AVX 0.1µF Ceramic Capacitor, X7R, Size 0603, 50V 4 C2 0805ZD225MAT AVX 2.2µF Ceramic Capacitor, X5R, Size 0805, 10V 1 C3 222215095001 220µF Aluminum Capacitor, SMD, 35V 1 C4, C5 1210YD106MAT AVX 10µF Ceramic Capacitor, X5R, Size 1210, 16V 2 C9 0805ZD105KAT AVX 1µF Ceramic Capacitor, X5R, Size 0805, 10V 1 C10 06035C223KAT AVX 22nF Ceramic Capacitor, X7R, Size 0603, 50V 1 C11 6SEPC560MX 560µF OSCON Capacitor, 6.3V 1 C12 12106D107MAT AVX 100µF Ceramic Capacitor, X5R, Size 1210, 6.3V 1 C15 06035C102KAT AVX 1nF Ceramic Capacitor, X7R, Size 0603, 50V 1 D1 SD103BWS Vishay Small Signal Schottky Diode 1 L1 CDEP105-2R0MC-32 Sumida 2.0µH Inductor, 15.8A Saturation Current 1 Q1, Q2 FDS7764A Fairchild 30V N-Channel MOSFET 7.5mΩ RDS(ON) @ 4.5V 2 R1 CRCW06032R21FKEY3 Vishay Dale 2.21Ω Resistor, Size 0603, 1% 1 R5 CRCW06031R21FKEY3 Vishay Dale 1.21Ω Resistor, Size 0603, 1% 1 R6, R9 CRCW06031002FKEY3 Vishay Dale 10k Resistor, Size 0603, 1% 2 R15 CRCW06038061FKEY3 Vishay Dale 8.06k Resistor, Size 0603, 1% 1 U1 MIC2164YMM Micrel. Inc. 300kHz Buck Controller 1 U2 MIC5233-5.0YM5 Micrel. Inc. LDO 1 February 12, 2015 Manufacturer Vishay Sanyo 31 Description Qty. Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C MIC2164 12V to 1.0V @ 5A Buck Converter February 12, 2015 32 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Bill of Materials (MIC2164 12V to 1.0V @ 5A) Item Part Number Manufacturer Description Qty. C1, C8, C17, C19 06035C104KAT AVX 0.1µF Ceramic Capacitor, X7R, Size 0603, 50V 4 C2 0805ZD225MAT AVX 2.2µF Ceramic Capacitor, X5R, Size 0805, 10V 1 C3 222215095001 220µF Aluminum Capacitor, SMD, 35V 1 C4 1210YD106MAT AVX 10µF Ceramic Capacitor, X5R, Size 1210, 16V 1 C9 0805ZD105KAT AVX 1µF Ceramic Capacitor, X5R, Size 0805, 10V 1 C10 06035C223KAT AVX 22nF Ceramic Capacitor, X7R, Size 0603, 50V 1 C11, C12, C13 12106D107MAT AVX 100µF Ceramic Capacitor, X5R, Size 1210, 6.3V 3 C15 06035C102KAT AVX 1nF Ceramic Capacitor, X7R, Size 0603, 50V 1 D1 SD103BWS Vishay Small Signal Schottky Diode 1 L1 CDRH104RNP-3R8 Sumida 3.8µH Inductor, 6A Saturation Current 1 Q1 FDS6910 Fairchild Dual 30V N-Channel MOSFET 17mΩ RDS(ON) @ 4.5V 1 R1 CRCW06032R21FKEY3 2.21Ω Resistor, Size 0603, 1% 1 Vishay Vishay Dale R5 CRCW06031R21FKEY3 Vishay Dale 1.21Ω Resistor, Size 0603, 1% 1 R6, R9 CRCW06031002FKEY3 Vishay Dale 10k Resistor, Size 0603, 1% 2 R15 CRCW06034022FKEY3 Vishay Dale 40.2k Resistor, Size 0603, 1% 1 U1 MIC2164YMM Micrel. Inc. 300kHz Buck Controller 1 U2 MIC5233-5.0YM5 Micrel. Inc. LDO 1 February 12, 2015 33 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C MIC2164-2 12V to 3.3V @ 15A Buck Converter February 12, 2015 34 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Bill of Materials (MIC2164-2 12V to 3.3V @ 15A) Item Part Number Manufacturer Description Qty. C1, C8, C17, C19 06035C104KAT AVX 0.1µF Ceramic Capacitor, X7R, Size 0603, 50V 4 C2 0805ZD225MAT AVX 2.2µF Ceramic Capacitor, X5R, Size 0805, 10V 1 C3 222215095001 220µF Aluminum Capacitor, SMD, 35V 1 C4, C5 1210YD226MAT AVX 22µF Ceramic Capacitor, X5R, Size 1210, 16V 2 C9 0805ZD105KAT AV) 1µF Ceramic Capacitor, X5R, Size 0805, 10V 1 C10 06035C472KAT AVX 4.7nF Ceramic Capacitor, X7R, Size 0603, 50V 1 C11 16ME1000WGL Sanyo 1000µF Aluminum Capacitor, 16V 1 C12 12106D107MAT AVX 100µF Ceramic Capacitor, X5R, Size 1210, 6.3V 1 C15 06035C102KAT AVX 1nF Ceramic Capacitor, X7R, Size 0603, 50V 1 D1 SD103BWS Small Signal Schottky Diode 1 1.0µH Inductor, 29A DC Current 1 Vishay Vishay (21) L1 HCP1305-1R0 Q1 FDMS7672 Fairchild 30V N-Channel MOSFET 6.9mΩ RDS(ON) @ 4.5V 1 Q2, Q3 FDS8874 Fairchild 30V N-Channel MOSFET 7.0mΩ RDS(ON) @ 4.5V 2 R1 CRCW06032R21FKEY3 Vishay Dale 2.21Ω Resistor, Size 0603, 1% 1 R5 CRCW06031R21FKEY3 Vishay Dale 1.21Ω Resistor, Size 0603, 1% 1 R6 CRCW06031002FKEY3 Vishay Dale 10k Resistor, Size 0603, 1% 1 R9 CRCW06034021FKEY3 Vishay Dale 4.02k Resistor, Size 0603, 1% 1 R15 CRCW06033241FKEY3 Vishay Dale 3.24k Resistor, Size 0603 1% 1 U1 MIC2164-2YMM Micrel. Inc. 600kHz Buck Controller 1 U2 MIC5233-5.0YM5 Micrel. Inc. LDO 1 Cooper Bussmann Note: 21. Cooper Bussman: www.cooperindustries.com. February 12, 2015 35 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C MIC2164-3 12V to 1.8V @ 10A Buck Converter February 12, 2015 36 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Bill of Materials (MIC2164-3 12V to 1.8V @ 10A) Item Part Number Manufacturer Description Qty. C1, C8, C17, C19 06035C104KAT AVX 0.1µF Ceramic Capacitor, X7R, Size 0603, 50V 4 C2 0805ZD225MAT AVX 2.2µF Ceramic Capacitor, X5R, Size 0805, 10V 1 C3 222215095001 220µF Aluminum Capacitor, SMD, 35V 1 C4 1210YD106MAT AVX 10µF Ceramic Capacitor, X5R, Size 1210, 16V 1 C9 0805ZD105KAT AVX 1µF Ceramic Capacitor, X5R, Size 0805, 10V 1 C10 06035C222KAT AVX 2.2nF Ceramic Capacitor, X7R, Size 0603, 50V 1 C11 6SEPC560MX 560µF OSCON Capacitor, 6.3V 1 C12 12106D107MAT AVX 100µF Ceramic Capacitor, X5R, Size 1210, 6.3V 1 C15 06035C102KAT AVX 1nF Ceramic Capacitor, X7R, Size 0603, 50V 1 D1 SD103BWS Small Signal Schottky Diode 1 L1 HCF1305-1R0 1.0µH Inductor, 20A Saturation Current 1 Q1, Q2 FDS7764A 30V N-Channel MOSFET 7.5mΩ Rds(on) @ 4.5V 2 R1 CRCW06032R21FKEY3 Vishay Dale 2.21Ω Resistor, Size 0603, 1% 1 R5 CRCW06031R21FKEY3 Vishay Dale 1.21Ω Resistor, Size 0603, 1% 1 R6 CRCW06031002FKEY3 Vishay Dale 10k Resistor, Size 0603, 1% 1 R9 CRCW06032001FKEY3 Vishay Dale 2k Resistor, Size 0603, 1% 1 R15 CRCW06038061FKEY3 Vishay Dale 8.06k Resistor, Size 0603, 1% 1 U1 MIC2164-3YMM Micrel. Inc. 1MHz Buck Controller 1 U2 MIC5233-5.0YM5 Micrel. Inc. LDO 1 February 12, 2015 Vishay Sanyo Vishay Cooper Bussmann Fairchild 37 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C Package Information 10-Pin MSOP (MM) February 12, 2015 38 Revision 4.1 Micrel, Inc. MIC2164/-2/-3/C MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com Micrel, Inc. is a leading global manufacturer of IC solutions for the worldwide high performance linear and power, LAN, and timing & communications markets. The Company’s products include advanced mixed-signal, analog & power semiconductors; high-performance communication, clock management, MEMs-based clock oscillators & crystal-less clock generators, Ethernet switches, and physical layer transceiver ICs. Company customers include leading manufacturers of enterprise, consumer, industrial, mobile, telecommunications, automotive, and computer products. Corporation headquarters and state-of-the-art wafer fabrication facilities are located in San Jose, CA, with regional sales and support offices and advanced technology design centers situated throughout the Americas, Europe, and Asia. Additionally, the Company maintains an extensive network of distributors and reps worldwide. Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this datasheet. This information is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry, specifications and descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual property rights is granted by this document. Except as provided in Micrel’s terms and conditions of sale for such products, Micrel assumes no liability whatsoever, and Micrel disclaims any express or implied warranty relating to the sale and/or use of Micrel products including liability or warranties relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright, or other intellectual property right. Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully indemnify Micrel for any damages resulting from such use or sale. © 2009 Micrel, Incorporated. February 12, 2015 39 Revision 4.1