V22N1 - APRIL

April 2012
I N
T H I S
I S S U E
2.5MHz, dual monolithic
supply with integrated 3A
power switches 12
digital power manager
sequences any number of
supplies 28
dual monolithic ideal diode
extends battery life 34
supercapacitor-based
power supply backup 36
µModule® DC/DC converter
for isolated supplies 37
Volume 22 Number 1
High Voltage, High Current Battery
Charger Works with All Converter
Topologies, Any Battery Configuration
Eko Lisuwandi
The market for rechargeable batteries in consumer electronics has
reached a level of stable maturity, where designing a battery charger
requires little more effort than dropping a purpose-built battery
charger IC into the design. This is because the batteries in consumer
electronics follow well-worn standards, with popular configurations,
float voltages, charge currents, output voltages and charge
algorithms. Even so, there is an ever-growing demand for batteries
that don’t fit these standard molds. Much of this demand is driven by
industrial green initiatives, coupled with
a general move to portable equipment
in medical and other specialized fields.
Dedicated charger ICs can’t keep pace with the current explosion in application diversity. The growing variety of battery setups is simply too extensive,
ranging from kilowatt-powered indoor forklifts and
isolated medical equipment to micropower energy
harvesting industrial sensors. Many applications
have unique requirements for optimal energy storage, which cannot be met by existing charger ICs.
The LTC®4000 takes on battery charger jobs that dedicated charger ICs can’t handle. It
pairs with just about any DC/DC converter to produce a complete, feature-rich battery
Caption solution—forget cobbling together discrete components.
charger
w w w. li n ea r.com
For example, there are no dedicated charger ICs on
the market that can charge battery stacks with 30V or
higher float voltage, provide 10A charging current, and
support efficient charging in a buck-boost, boost or
flyback topology. As a result, designers have turned to
(continued on page 4)
Linear in the News
In this issue...
COVER STORY
High Voltage, High Current Battery Charger Works with
All Converter Topologies, Any Battery Configuration
Eko Lisuwandi
1
DESIGN FEATURES
Dual Output Monolithic Supply with
Integrated 3A Power Switches and Operation
to 2.5MHz in a 7mm × 4mm DFN
Mehdi Alimadadi
12
2MHz Dual DC/DC Controller Halves Settling
Time of Load Release Transients, Features
0.67% Differential VOUT Accuracy and
Is Primed for High Step-Down Ratios
Shuo Chen and Terry Groom
19
Current Mode Switching Supply with Ultralow Inductor
DCR Sensing for High Efficiency and High Reliability
Jian Li, Haoran Wu and Gina Le
24
Digital Power Management Reduces Energy
Costs While Improving System Performance
Andy Gardner
28
DESIGN IDEAS
What’s New with LTspice IV?
Gabino Alonso
32
Dual Monolithic Ideal Diode Extends
Battery Run Time and Prioritizes Power
Sources with Glitch-Free Switchover
Joshua Yee
34
Single-IC Supercapacitor-Based
Power Supply Backup Solution
Ashish Kirtania
36
µModule Converters Take the Hassle Out
of Designing Isolated Power Supplies
At the annual ACE Awards presentation in March, the editors of EE Times
and EDN magazines presented Linear Technology the Power product
ACE award for the LTC6803 battery stack monitor for hybrid/electric vehicles.
The ACE Awards showcase the best in today’s electronics industry, the hottest products, and celebrate the promise of talent and innovation.
The LTC6803 is a second-generation high voltage battery monitor for hybrid/electric vehicles (HEVs), electric vehicles (EVs) and other high voltage, high performance
battery systems. The product is a complete battery-measuring IC that includes a
12-bit ADC, a precision voltage reference, a high voltage input multiplexer and
a serial interface. Each LTC6803 can measure up to 12 individual battery cells in
series. The device’s proprietary design enables multiple LTC6803s to be stacked in
series without opto-couplers or isolators, permitting precision voltage monitoring of every cell in long strings of series-connected batteries. The LTC6803 offers
an extended cell measurement range from –300mV to 5V, enabling the LTC6803 to
monitor a wide range of battery chemistries, as well as supercapacitors. It ensures
accurate, safe, reliable and error-free operation in harsh automotive environments.
Also presented at the ACE Awards was the first award in a new category, the Jim
Williams Contributor of the Year Award (see sidebar). This new, annual award
was presented by the editors of EDN and EE Times to honor a contributor of the
year to the electronics industry. This initial award for 2012 was presented to Jim
Williams, in his memory, to show appreciation for his many contributions over
the years that stand as examples of how to communicate, educate and mentor a
community. The award was accepted by Jim’s wife, Siu Williams, along with his
friend and colleague of many years, Linear CTO Bob Dobkin. Jim Williams served
as Staff Scientist at Linear for nearly thirty years, and was a prolific inventor,
writer and mentor, both within Linear and in the broader electronics community.
EE TIMES CHINA PRESENTS ACE AWARD
David Ng
37
product briefs
39
back page circuits
40
2 | April 2012 : LT Journal of Analog Innovation
EE TIMES/EDN PRESENT AWARDS
Based on voting by electronic design engineers, Linear’s LTC4000 high voltage
controller and power manager was selected by EE Times China for a 2012 China
ACE Award in the Power Management category. This unique device converts virtually any externally compensated DC/DC power supply into a full-featured battery
charger. The LTC4000 is capable of driving typical DC/DC converter topologies,
including buck, boost, buck-boost, SEPIC and flyback. The device offers precision
input and charge current regulation and operates across a wide 3V to 60V input
and output voltage range, compatible with a variety of different input voltage
sources, battery stacks and chemistries. Applications include high power battery
charger systems, high performance portable instruments, battery backup systems,
industrial battery-equipped devices and notebook/subnotebook computers.
Linear in the news
CONFERENCES & EVENTS
IDTechEx Energy Harvesting & Storage Europe 2012,
Estrel Hotel and Convention Centre, Berlin, Germany,
May 15–16, Booth 3—Linear will showcase its
energy harvesting products, along with
its Dust Networks wireless sensor networking products. Joy Weiss, President
of Linear Technology’s Dust Networks
product group, will present “Low Power
Wireless Sensor Networks Made Practical”
at the conference. More at www.idtechex.
com/energy-harvesting-europe/eh.asp
BATTERY STACK MONITORS
FOR HYBRID AND
ELECTRIC VEHICLES
At the annual ACE Awards
presentation in March, the
editors of EE Times and
EDN magazines presented
Linear Technology the Power
products ACE Award for the
LTC6803 battery stack monitor
for hybrid/electric vehicles.
The LTC6803 is a second
generation high voltage battery
monitor for hybrid/electric
vehicle (HEVs), electric vehicles
(EVs) and other high voltage,
high performance battery
systems.
Jim Williams Honored
Sensors Expo & Conference, Donald E. Stephens
Convention Center, Rosemont, IL, June 6-7, Booths
1020 & 1022—Linear will exhibit energy
harvesting and Dust Networks wireless
sensor networking products. More info
at www.sensorsmag.com/sensors-expo
COMPUTER HISTORY MUSEUM
EXTENDS “ANALOG LIFE” EXHIBIT
The Computer History Museum in
Mountain View, California has just
announced that, due to significant interest, they have decided to extend through
September 15 the exhibit, “An Analog
Life: Remembering Jim Williams.” The
centerpiece of the exhibit is the engineering workbench of Jim Williams, who was
Staff Scientist at Linear Technology until
his untimely death in June of last year.
The museum painstakingly transported
Williams’ bench from his lab at Linear to
the museum, where visitors can now view
the bench as well as a video discussing
Jim’s many contributions to the world of
analog technology. For more information on the exhibit, visit www.computerhistory.org/highlights/analoglife/ n
INSPIRING, EXPLAINING, DESIGNING: JIM WILLIAMS AND ANALOG CIRCUITS*
This is a new category for the ACE Awards and was
conceived a means to recognize the accomplishments
of the people behind the extraordinary contributions to
EDN and EE Times over the past year. Their work has
broadened our understanding of the rapid advances
in engineering and design, while inspiring countless
engineers to reach ever higher.
It is with this in mind that the inaugural Award goes,
posthumously, to Jim Williams, the consummate
contributor, both in the specific content as well as in
the spirit in which it was given.
At a high level, Jim was a superb analog circuit
designer, one of the best. But behind his knowledge
of analog circuit behavior was his ability and desire
to explain these circuits and to inspire others with
the marvels analog circuits could perform. For Jim,
circuit design was a form of art. His first article ran in
EDN in 1975, and his articles immediately became a
source of information and inspiration for circuit design
engineers, continuing through 2011 with his last two
articles appearing after his death in June.
Did you know there’s a popular blog called “Reading
Jim Williams”?1 Jim’s friend Dr. Kent Lundberg started
this blog as a tribute to Jim’s 62 application notes
for Linear Technology. Kent is reading each one and
commenting on Jim’s circuit design decisions, as well
as including Jim’s “remarkable quotes.” For most
of us, when we think of “app note” we tend to think
“company commercial.” Jim seemed to approach all
of his writings, both his company app notes as well
as his many contributed articles for EDN, as teaching
moments. He ended one of his app notes with a
paraphrase of Einstein, “everything should be as
simple as possible, but not too simple.”
As EDN editor Paul Rako said of Jim’s writing,2 “[His]
articles always stressed understanding. Jim did not
condescend and write down to us. He never tried to
impress you with his math or his intellect. He didn’t
make things complicated so you would think he was
smart. He made things look simple. That is why he
was brilliant. Anyone can learn a bunch of jargon and
a few tricks and secrets and try to act smarter than
you. Jim was the exact opposite. He took the trouble
to describe the basic principles of what was going on.
Then he showed you how to achieve the goals your
designs needed to achieve.”
That is why Jim is the first winner of the Contributor
of the Year Award, and that is also why it has been
renamed the, “EETimes/EDN: Jim Williams Memorial
‘Contributor of the Year’ Award.” Jim will forever be
an example of what contributing to the engineering
community is all about.
Notes
*This article is courtesy of EDN magazine.
1http://readingjimwilliams.blogspot.com
2http://www.edn.com/article/518496-Analog_guru_Jim_Williams_dies_after_stroke.php
April 2012 : LT Journal of Analog Innovation | 3
The LTC4000 battery charger fills the gap between applications supported
by easy-to-use dedicated charger ICs and those that would otherwise
require complex discrete solutions. The LTC4000 uses a 2-IC model to
bring single-IC simplicity to a wide range of charger solutions.
(LTC4000, continued from page 1)
relatively cumbersome discrete component solutions, essentially returning to
the pre-charger-IC dark ages. Although
discrete solutions can satisfy many
charger requirements, they cannot match
the ease-of-use and compact nature of
dedicated charger ICs. Designers demand
a solution that retains the simplicity of
a dedicated charger IC with the versatility of discrete component solutions.
Linear’s LTC4000 battery charger fills
the gap between applications supported
by easy-to-use dedicated charger ICs
and those that would otherwise require
complex discrete solutions. The LTC4000
retains the simplicity of a dedicated
single-IC charger, but uses a 2-IC model
to match the applications versatility of
discrete solutions. It can be paired with
any DC/DC or AC/DC converter topology, including but not limited to buck,
boost, buck-boost, SEPIC and flyback.
The LTC4000’s wide input voltage range
(3V–60V) and virtually unlimited current
capability produces efficient, high performance, full-featured battery chargers that
rival the performance of dedicated charger
ICs. Figure 1 shows a typical application:
the LTC4000 paired with the LTC3786 to
create a 5A, 5-cell Li-ion battery charger.
FEATURE SUMMARY
The LTC4000 also includes intelligent
PowerPath™ control via low loss external PFETs. One external PFET is used to
prevent reverse current from the battery or system output going back to the
input. Another PFET is used to control
battery charging and discharging.
The LTC4000 converts virtually any
Linear Technology externally compensated DC/DC power supply into
a battery charger featuring:
•wide input and output voltage
range of 3V to 60V
•accurate (±0.25%) resistor
programmable battery float voltage
In this case, the low loss nature of the
PFETs is crucial for systems requiring
high charge current for high capacity
batteries. This second PFET also facilitates an instant-on feature that provides
immediate downstream system power
even when connected to a heavily discharged or short faulted battery.
•pin-selectable timer or
current termination
•temperature qualified charging
using an NTC thermistor
•automatic recharge
•C/10 trickle charge for
deeply discharged cells
•bad battery detection and
status indicator outputs
•precision current sense enables low sense
voltages in high current applications
6V TO 18V
SENSE+
SS
3.3mΩ
LT3786
ITH
PowerPath control preferentially provides
power to the system load. When input
power is limited, the system load is always
prioritized over charging. Furthermore,
if the system load requires more power
than the input can support, the battery
150µF
0.1µF
28.7k
RST
CLN
IN
ITH
1.87M
22nF
CC
383k
100k
BGATE
Si7135DP
BAT
OFB
107k
LTC4000
ENC
CHRG
FLT
FBG
107k
BFB
IIMON
10nF
10nF
1.87M
NTC
IBMON
CX
22.1k
GND
CL
10µF
22.1k
10k
TMR BIAS
10k
1µF
NTHS0603
N02N1002J
4 | April 2012 : LT Journal of Analog Innovation
10mΩ
IID IGATE CSP
CSN
1µF
VM
Figure 1. 6V to 21V at 5A boost converter
charger for five Li-ion cells
VOUT
12V
15A
Si7135DP
VBIAS
VBAT
21V FLOAT
5A MAX
CHARGE
CURRENT
5-CELL
Li-ION
BATTERY
PACK
design features
At the core of the LTC4000 are four internal error amplifiers,
whose outputs combine to drive the external DC/DC converter
control loop. In this way, it can control almost any battery
charging cycle, regardless of chemistry and float voltage.
The LTC4000 is available in low
profile 28-lead 4mm × 5mm
QFN and SSOP packages.
FOUR CONTROL LOOPS KEEP THE
BATTERY CHARGED AND OUTPUT IN
REGULATION
At the core of the LTC4000 are four
internal error amplifiers, whose outputs
combine to drive the external DC/DC converter control loop. In this way, it can
control almost any battery charging cycle,
regardless of chemistry and float voltage.
Figure 2 shows a simplified block diagram of the four internal error amplifiers
(A4-A7). Each of the four input transconductance amplifiers is responsible
for a different regulation loop: input
current, charge current, battery float
voltage and output voltage. The output
transconductance amplifier (A10) ensures
that the loop requiring the lowest voltage on the ITH pin for regulation controls the external DC/DC converter.
The input current regulation loop (A4 in
Figure 2) prevents the input current from
exceeding the resistor programmable input
current limit. This input current limit
prevents the overall system from overloading the source, allowing for more predictable and reliable behavior. Furthermore,
this adds an extra layer of protection
to extend the life of the power components of the DC/DC converter and any
sources that lack overcurrent protection.
ITH
LTC4000 CORE
A10
gm–
A4
IIMON
IL
1V
+
–gm
–
consume battery current when the battery
(connected to BAT pin) is the only available
power source. For VIN ≥ 3.0V, the typical
resistance from the FBG pin to GND is 100Ω.
CC
A5
–
1V
gm+
–
A7
+
–
CL
OFB
gm
A6
+
–
gm
IBMON
OFB
1.193V
BFB
BFB
1.136V
Figure 2. Simplified block diagram of the LTC4000
core—four error amplifiers with combined output
The other current regulation loop is the
charge current regulation loop (A5). This
loop controls the constant current phase
of the charging cycle, ensuring that the
charge current sensed through the charge
current sense resistor does not exceed the
resistor programmable full charge current.
The constant current regulation loop
controls charging until the battery reaches
its float voltage. At this point, the battery voltage regulation loop (A6) takes
over, the charge current begins to drop
and the charger enters the constant
voltage phase of the charging cycle.
The float voltage is programmed using
the feedback resistor divider between
the BAT pin and the FBG pin. The FBG pin
disconnects the resistor divider load when
VIN is not present. This ensures that the
float voltage resistor divider does not
When the battery is not being charged, nor
supplying power to the load, the external
PFET connected to the battery is turned
off (Figure 4). In this scenario, the output
voltage regulation loop (A7 in Figure 2)
controls the external DC/DC converter.
The output voltage regulation loop is
similar to the battery voltage regulation
loop. This loop regulates the voltage at
the CSP pin based on the feedback resistor divider between the CSP pin and the
FBG pin. This output voltage regulation is
important to ensure that the system output
voltage remains well regulated when the
battery is disconnected from the load.
Figure 3. Battery charging phases for 3 series
LiFePO4 cells with the circuit shown in Figure 1
12
CC TO CV
10
8
IBAT (A)
is used to provide additional power to
satisfy the total system output load.
6
PRECHARGE
4
CONSTANT
CURRENT
2
0
CONSTANT
VOLTAGE
TERMINATION
6
7
8
9
VBAT (V)
10
11
12
April 2012 : LT Journal of Analog Innovation | 5
The LTC4000’s PowerPath control feature consists of two functions: the
input ideal diode control, providing a low loss ideal diode function from
DC/DC converter to the output; and the battery PowerPath control, providing
a smart PowerPath route between the system output and the battery.
POWERPATH CONTROL
The other important feature of the
LTC4000 is PowerPath control, whichconsists of two functions: the input
ideal diode control, providing a
low loss ideal diode function from
DC/DC converter to the output; and the
battery PowerPath control, providing a smart PowerPath route between
the system output and the battery.
The input ideal diode feature provides
low loss conduction from the output of
the DC/DC converter (IID pin—anode) to
the system output (CSP pin—cathode).
Low loss conduction is important for
efficiency and heat management in high
current systems. This feature also prevents
reverse current from the system output to
the DC/DC converter. Such reverse current
causes unnecessary drain on the battery
and in some cases may result in undesirable DC/DC converter behavior. This ideal
diode behavior is achieved by controlling an external PFET (M1) whose gate is
connected to the IGATE pin (Figure 4).
The PowerPath controller of the external PFET connected to the BGATE pin is
similar to the input ideal diode controller
driving the IGATE pin (Figure 4). When
not charging, the PMOS behaves as an
ideal diode between the BAT (anode) and
the CSN (cathode) pins. The ideal diode
behavior allows the battery to provide
current to the system load when the
DC/DC output is in current limit or the
DC/DC is slow to react to an immediate
load increase at the output. This feature
ensures a stable system output voltage.
In addition to the ideal diode behavior,
BGATE allows current to flow from the
CSN pin to the BAT pin during charging.
There are two regions of operation when
current is flowing from the CSN pin to the
BAT pin. The first is when charging into a
heavily discharged battery (battery voltage is below the INSTANT ON threshold,
VBAT(INST ON)). In this region of operation,
the controller (A11 in Figure 4) regulates the voltage at the system output to
approximately 86% of the final float
voltage level. This feature provides a
system output voltage significantly higher
than the battery voltage when charging
into a heavily discharged battery. This
INSTANT ON feature allows the LTC4000 to
provide sufficient voltage at the system
output independent of the battery voltage.
The second region of operation
occurs when the battery feedback
voltage is greater than or equal to
the INSTANT ON threshold. In this
region, the BGATE pin is driven low
to allow the PMOS to turn completely
on, reducing any power dissipation due to the charge current.
FROM DC/DC
OUTPUT
TO SYSTEM
LOAD
M1
IID
Figure 4. Input ideal diode and
battery PowerPath controller
8mV
IGATE
CSP
+
–
A1
BATTERY POWERPATH CONTROLLER
A2
A11
RCS
BATTERY
CHARGE
CURRENT
SENSE
RESISTOR
INPUT IDEAL
DIODE DRIVER
ANALOG
CONTROLLER
CSN
BGATE
gm
M2
0.974V
BAT
OFB
+
–
BATTERY PACK
6 | April 2012 : LT Journal of Analog Innovation
design features
The LTC4000 has broad applications versatility—it
can be paired with a DC/DC converter to produce
a battery charger for any battery configuration.
APPLICATIONS
The LTC4000 has broad applications versatility—it can be paired with a DC/DC converter to produce a battery charger for
any battery configuration. The following
applications illustrate this versatility.
High Voltage, High Current Charger
Building a complete charging system
with the LTC4000 and a DC/DC converter
is as easy as using a dedicated charger
IC. Figure 5 shows the LTC4000 controlling an LT3845A buck converter in a
charger designed for a 3S LiFePO4 battery pack (3S refers to three cells in a
series configuration). The LT3845A buck
converter is selected for its simplicity
and high, 60V input voltage capability.
Each of the LiFePO4 cells has a typical float voltage of 3.6V, resulting in
an overall float voltage of 10.8V. The
10.8V float voltage is set by RBFB2 = 133k
and RBFB1 = 1.13M. Once the float voltage is set, the value of ROFB1 and ROFB2 are
Figure 5. 48V to 10.8V at 10A buck converter
charger for 3-series LiFePO4 battery pack
determined—this sets the output voltage
when charging is terminated. Here, ROFB2
is set at 127k and ROFB1 at 1.15M to set
the output regulation voltage at 12V.
level, but as long as it is less than 200k, it
affects the regulated trickle charge current
level. In this example, the 24.9k value is
chosen to set the trickle charge current
level at 1.25A. Trickle charging can occur
at the beginning of a charge cycle when
the voltage at the battery is less than 68%
of the float voltage. This trickle charge
feature is especially important for lithiumion batteries, as they require a smaller
current (typically <20% of full charge
current) to safely and gradually bring
the battery voltage higher before supplying them with the full charge current.
After setting the float and output voltages,
set the full charge current for the battery.
In this particular example, the full charge
current is set to 10A using an RCS value
of 5mΩ and an RCL value of 24.9k. The
regulated sense voltage across RCS should
be as large as possible for the highest
accuracy. However, a larger sense voltage causes RCS to dissipate more power.
Since the charge current regulation error
amplifier has a maximum regulation
level of 1V, this means that the regulated sense voltage across RCS is limited
to a maximum of 50mV (=1V/20). For a
10A charge current, the maximum power
dissipation on this sense resistor is 0.5W.
The only other regulation loop with
a set point is the input current regulation loop. Using a similar method to
setting RCS, in this example RIS is set to
5mΩ and the IL pin is left floating (internally pulled to a voltage above 1V) to set
a maximum input current limit of 10A.
Any value of RCL that is larger than 20k
will not affect the full charge current
15V TO 60V
5mΩ
IN
LT3845A
BIAS
LTspice IV
100µF
14.7k
1M
circuits.linear.com/549
RST
CLN
IN
ITH
ROFB1
1.15M
47nF
CC
IID IGATE CSP
BAT
OFB
LTC4000
ENC
CHRG
FLT
RBFB2
133k
BFB
RBFB1 1.13M
NTC
IBMON
10nF
ROFB2
127k
FBG
IIMON
10nF
SYSTEM
LOAD
BGATE
VM
3V
RCS
5mΩ
Si7135DP
CSN
1µF
1.10M
100k
Si7135DP
OUT
VC
SHDN
TMR
IL
0.1µF
CX
CL
RCL
24.9k
10k
GND BIAS
22.1k
10k
1µF
10.8V FLOAT
10A MAX
CHARGE
CURRENT
3-CELL
Li-Ion
BATTERY
PACK
NTHS0603
N02N1002J
April 2012 : LT Journal of Analog Innovation | 7
Building a complete charging system with the LTC4000 and a DC/DC
converter is nearly as easy as using a dedicated charger IC. Just a few
resistors and capacitors are needed to set the float voltage, charge current,
input current limit and charge termination (current level or timer termination).
TR1
PA1277NL
VIN
18V TO 72V
2.2µF
×2
221k
•
FDC2512
221k
ISENSE
15.8k
VSYS
4.4V
2.5A
PDS1040
309k
100k
150k
68Ω
GATE
RUN
SiA923EDJ
1µF
100µF
×3
BAS516
681Ω
1µF
VCC
20mΩ
•
•
MMBTA42
VCC
PDZ6.8B
6.8V
VOUT
150pF
3.01k
LTC3805-5
0.04Ω VCC
13.6k
SSFLT
IN
VM
CLN
IID
IGATE
CSP
VOUT
25mΩ
1.5k
CSN
ITH
0.1µF
FS
SYNC GND FB OC
75k
ISO1
PS2801-1-K
SiA923EDJ
BGATE
47nF
BAS516
10nF
CC
BAT
LTC4000
24.9k
BAS516
OFB
ITH
115k
RST
BFB
FLT
115k
CHRG
FBG
ENC
NTC
IIMON IBMON IL CL
10nF
10nF
Figure 6. 18V–72V VIN to 4.2V at 2A isolated single-cell Li-ion battery charger
The four simple steps described here
are sufficient to customize an LTC4000
charging solution to charge many generic
battery configurations. To customize
the solution further, a few other component values can be chosen to program the charge termination algorithm.
LTC4000 offers both timer termination
and charge current level termination.
With charge current level termination, the charging process is terminated
when the charge current level drops
8 | April 2012 : LT Journal of Analog Innovation
309k
TMR
CX
10k
GND BIAS
RNTC
100nF
22.1k
22.1k
1µF
NTHS0603
N02N1002J
SINGLE-CELL
Li-Ion
BATTERY
PACK
(in the constant voltage mode) to the
level programmed at the CX pin.
at which point the charge status indicator pin (CHRG) assumes a high-Z state.
With timer termination, the charging
process continues in the constant voltage
mode until a time period programmed
with a capacitor on the TMR pin expires.
In this example, the LTC4000 is set up
with a timer termination period of 2.9h
using a 0.1µF capacitor connected to the
TMR pin. The 22.1k resistor connected to
the CX pin sets a 1A charge current level,
LTC4000 offers temperature-qualified
charging via the NTC pin. A negative
temperature coefficient (NTC) resistor, thermally coupled to the battery, is
connected in a resistor divider network
between the BIAS, NTC and GND pins. This
NTC resistor allows charging to be paused
when the battery temperature is outside
a particular range. In this example, the
battery temperature range is set between
design features
With the LTC4000, the task of designing an isolated
charger is reduced to selecting the appropriate
isolated converter, choosing PFETs and determining
the values of some resistors and capacitors.
–1.5°C to 41.5°C. Temperature-qualified
charging protects batteries from hazardous charging conditions, such as extreme
hot or cold, which can potentially damage the batteries and shorten their life.
The only remaining components that may
need to be customized are the series resistor and capacitor compensation network
between the CC and ITH pins, as well as
the resistor divider network connected to
the VM pin. As starting values, the compensation network can be set to a 10k
resistor in series with a 100nF capacitor.
It can then be optimized by looking at
the time domain response to small signal
perturbation for each of the four regulation loops. In this example, the final
optimized values are 14.7k and 47nF.
The VM pin is an input to a comparator
with a threshold set at 1.193V. When the
voltage at this pin is below the threshold, the RST pin is driven low. When
it is above the threshold, the RST pin
assumes a high-Z state. By connecting the
RST pin to the DC/DC RUN or SHDN pin,
this comparator provides a simple and
accurate UVLO (undervoltage lockout)
signal that can be used to start the
external converter. In this example, the
input UVLO level is set to 14.3V. Setting a
minimum voltage ensures that the input
to the converter is within its operating
range before it is allowed to start up.
This in turn allows for a more consistent and predictable power-up behavior of the overall charging solution.
A discrete solution with similar features
to the 10A/3-cell LiFePO4 battery charger
would have required at least two high
side current sense amplifiers, four operational amplifiers as well as two high
voltage ideal diode controllers. Each of
these would need to be tested and qualified separately to ensure compatibility
of their specifications such as common
mode range, speed and input supply
voltage range. Furthermore a discrete
solution would require a microprocessor to handle charging algorithm.
As shown in the example, the LTC4000
eliminates these components and
the need to test them. Design is simplified to choosing an appropriate
DC/DC converter for the voltage and
power requirement, and a few passive
components—mostly resistors to set the
important charger system parameters.
Isolated Battery Charger
Figure 6 shows the LTC4000 paired
with the LTC3805-5 to build an isolated
single cell Li-ion charger with 2A charging current. This application shows the
power of the LTC4000 to create a unique
battery charger solution using readily
available DC/DC converters of practically any topology. This simple LTC4000based solution eliminates the need to
design a complex discrete solution.
With the LTC4000, the task of designing
an isolated charger is reduced to selecting the appropriate isolated converter,
choosing PFETs and determining the
values of some resistors and capacitors.
For the application shown in Figure 6,
we use the LTC3805-5 isolated flyback
converter with a high input voltage
capability. Two relatively low voltage
PFETs are used for PowerPath control
since only voltages less than 6V appear
on the secondary side. The only unique
connection in this particular application
is the use of the opto-coupler to deliver
the ITH feedback signal from the LTC4000
on the secondary side to the ITH pin of
the LTC3805-5 on the primary side.
The resulting charger is capable of charging a single cell Li-ion battery (4.2V float)
at 2A in an isolated environment. The
system has a wide input range of 18V to
72V with a 2.9h charging termination time
as well as a 220m A trickle charge current.
The overall solution limits the total system
output current to 2.5A in a controlled
manner. By preventing current overload
of the primary, the input current limit
provides an extra level of protection
for the power components and provides greater overall system reliability.
High Voltage Buck-Boost
Battery Charger
Another unique, but commonly requested
battery charger solution is a buckboost battery charger. Again, there is
no dedicated IC solution currently available. Figure 7 shows the LTC4000 paired
with the LTC3789 to create a full-featured
buck-boost 12V lead acid battery charger.
The buck-boost topology allows the battery to be charged from a voltage lower
or higher than its float voltage, easing
the battery and input voltage choice in
the system design. The number of battery
cells in series can then be optimized for
other system parameters or perhaps the
April 2012 : LT Journal of Analog Innovation | 9
5.6Ω
IHLP6767GZ
ER4R7M01
4.7µH
390pF
3.6Ω
B240A
VIN
6V TO 36V
12.5A MAX
4mΩ
Q2
270µF
Q4
B240A
Q5
0.22µF
TG1
SW1
0.01Ω
Q3
Si7135DP
0.22µF
0.01Ω
1.24k
3.3µF
×5
1800pF
VOUT
15V
5A
330µF
×2
1.24k
BG1 SENSE+ SENSE– BG2
BOOST1
SW2
TG2
BOOST2
DFLS160
DFLS160
INTVCC
INTVCC
10µF
MODE/PLLIN
VIN
1µF
100k
LTC3789
VINSNS
PGOOD
IOSENSE+
IOSENSE–
VOUTSNS
5.6V, BZT52C5V6
121k
FREQ
EXTVCC
ILIM
RUN
154k
VFB
ITH
SS
SGND
10µF
PGND1
8.06k
0.01µF
10mΩ
14.7k
RST
CLN
IN
365k
100k
ITH
100nF
CC
IID IGATE CSP
CSN
BGATE
1µF
VM
3V
BZX84C3VO
LTC4000
ENC
CHRG
187k
FBG
FLT
Q2: SiR422DP
Q3: SiR496DP
Q4: SiR4840BDY
Q5: SiR496DP
IIMON
10nF
IBMON
10nF
IL
CL
TMR
CX
GND BIAS
10k
18.2k
1µF
22.1k
Figure 7. 6V–36V VIN to 14.4V at 4.5A buck-boost 6-cell lead acid battery charger
pricing and availability of such battery
packs. Similarly, the flexibility and simplicity of programming the charge current
by setting the values of two resistors
(RCS and RCL) also further ease the battery
capacity choice in the system design.
The overall charging solution of the
LTC4000 and LTC3789 pairing shown above
is capable of charging a 12V lead acid
battery (14.4V absorption and 13.4V float)
10 | April 2012 : LT Journal of Analog Innovation
at 4.5A from an input source voltage that
can range from 6V to 36V. The system is
programmed with an input current limit
of 12.5A, allowing load sharing between
the input and the battery if a system load
demands more that 12.5A from the input.
This feature is especially important at the
lower end of the source voltage range,
where input current increases rapidly to
meet increasing output power demands.
1µF
OFB
15k
BFB
NTC
162k
NTHS0603
N02N1002J
RNTC
6-CELL
LEAD ACID BATTERY
14.4V ABSORPTION
13.4V FLOAT
4.5A MAX CHARGE CURRENT
The charger solution shown here provides no termination, allowing continuous constant voltage charging at the final
float voltage of 13.4V. Connecting the
CHRG pin to the BFB pin through the 187k
resistor implements a 2-stage charging
algorithm (absorption and float) common for lead acid batteries. The overall
charging algorithm first charges to an
absorption level of 14.4V until the charge
current drops to 500m A. At this point
design features
The LTC4000’s wide input voltage range (3V–60V)
and virtually unlimited current capability allow it to be
combined with just about any power converter to form
an efficient and high performance full-featured battery
charger typically occupying less than 3.6cm2.
Figure 8. Demonstration
circuit showing a complete
battery charger formed by
pairing the LTC4000 and
LTC3789
the CHRG pin assumes a high-Z state,
changing the feedback resistor network
connected to the BFB pin. In this way
the battery charger enters final float
constant voltage mode with a final float
target of 13.4V. If the battery drops below
13.1V (recharge threshold), the CHRG pin
turns low impedance again and the battery charger is again set to charge the
battery to the absorption level of 14.4V.
Because this is a buck-boost charger
setup, a battery stack with any float
voltage between 3V to 36V can be supported with a simple adjustment of the
resistor dividers and the PFET choice.
Similar changes allow the battery
charge current to be programmed from
a few milliamps to tens of amps.
Figure 8 shows a demo board of the
LTC4000 and LTC3789 pairing. Note
that the required space occupied by the
LTC4000 and its passive components
is small, occupying an area less than
3.6cm2. This allows for a compact charging solution for virtually any battery.
CONCLUSION
Increases in demand for alternative energy
sources, coupled with an explosion in
portable industrial and medical applications, have resulted in the need for a wide
variety of rechargeable-battery-powered
systems. Many of these systems have
requirements that dedicated battery charger ICs —geared to specific battery chemistries/configurations and input/output
voltages—cannot meet. Discrete solutions
can satisfy the needs of these systems, but
such solutions are more difficult to implement, occupy considerably more PC board
space and require significantly more
design time than dedicated IC solutions.
The LTC4000 battery charger fills the
gap between applications supported
by easy-to-use dedicated charger ICs
and those supported by more complex
discrete solutions. The LTC4000’s wide
input voltage range (3V–60V) and virtually unlimited current capability enable
pairing with any DC/DC or AC/DC converter topology, including buck, boost,
buck-boost, SEPIC and flyback. When
paired with the right power converter,
the LTC4000 forms an efficient and high
performance full-featured battery charger
typically occupying less than 3.6cm2. n
April 2012 : LT Journal of Analog Innovation | 11
Dual Output Monolithic Supply with Integrated 3A Power
Switches and Operation to 2.5MHz in a 7mm × 4mm DFN
Mehdi Alimadadi
There is no shortage of ICs to help designers build switching DC/DC switching power
supplies. Choices range from versatile controllers requiring a number of external
components, to fully integrated, monolithic solutions that benefit from a low external
parts count to minimize overall solution size. The LT8582 dual-channel converter offers
the versatility of a controller IC in a complete, monolithic dual-channel solution.
The LT8582 integrates two complete,
independent converters, including high
power 3A, 42V power switches. It can
operate up to 2.5MHz, and with its tiny
7mm × 4mm DFN package, fits into the
smallest spaces. It includes several features
that give designers the ability to optimize
the converter, such as soft-start, single-pin
feedback, single-resistor frequency setting,
master/slave power switches, separate
maximum commanded and fault current
limits, external PFET control for output
or input disconnect, FAULT protection,
PG pin for power supply sequencing, and
CLKOUT signal for out-of-phase synchronizing and die temperature monitoring.
FLEXIBILITY AND SIMPLICITY
Each channel of the LT8582 can be
independently configured in a boost,
SEPIC, inverting or flyback topology.
Figure 1 shows a few common combinations that could be used in commercial or industrial applications, such
as local power supplies, LCD/E-ink displays, and engine control units (ECU).
The LT8582 is rugged, with solid performance. Even with all of its advanced
features, it is easy to use—designers
can choose to apply features to fit a
variety of applications. Its wide input
operating voltage of 2.5V to 22V, and
12 | April 2012 : LT Journal of Analog Innovation
CLKOUT PIN, SYNCHRONIZING AND
TEMPERATURE MONITORING
the 3A, 42V switches on each channel
add to the versatility of the chip.
HIGH SWITCHING FREQUENCY
The LT8582’s constant frequency oscillator,
programmable from 200kHz to 2.5MHz
using one resistor, employs frequency
foldback to better control the inductor
current during converter start-up. This
wide frequency range allows the switching noise to be placed so that sensitive
frequencies are avoided. While lower
switching frequencies offer better efficiency, higher switching frequencies help
reduce the size of passive components. The
switching frequency can be synchronized
to an external clock by connecting a clock
signal to the SYNC pin. Grounding the
SYNC pin enables the internal oscillator.
The LT8582 has two CLKOUT signals, one
for each channel. CLKOUT for channel 1
has a fixed 50% duty cycle and is 180°
out of phase with the power switch.
This can be used to sychronize channel
2 antiphase to channel 1, reducing the
converter’s overall input current ripple.
The CLKOUT signal for channel 2 features
a duty cycle that varies with die temperature (3% per 10°C) and is in phase
with the power switch. This can be used
for monitoring the die temperature.
FAULT PROTECTION AND
THE GATE PIN
The LT8582 has internal circuitry to detect
switch overcurrent, VIN overvoltage and
die overtemperature (> ~165°C). The chip’s
Figure 1. Common dual power topology combinations
VIN
D1
L1
VOUT1
VIN
LT8582
INVERTING
D2
L2A
C4
D1
VOUT1
LT8582
BOOST
VIN
L1A
C3
L2B
C1
SEPIC
C2
INVERTING
VOUT2
VIN
L1B
C1
D2
L2A
C3
C2
L2B
VOUT2
design features
The LT8582 integrates two complete, independent converters, including
high power 3A, 42V power switches. It can operate up to 2.5MHz, and with
its tiny 7mm × 4mm DFN package, fits into the smallest spaces.
VIN
L1
M1
M2
R1
M2
VIN
VOUT
M1
R1
R2
GATEx
LT8582 CHx
L1
GATEx
GATEx
VIN
LT8582 CHx
LT8582 CHx
Figure 2. Controlling external PFETs for input disconnect (left) and output disconnect (right)
When a fault is detected, the LT8582 stops
switching and the GATE pin becomes high
impedance. The external PFET is then
turned off by the external RGATE resistor. The RGATE resistor must be selected
so that sufficient VGS is available for the
PFET to fully enhance into triode under
normal operation. When the fault is
removed, the LT8582 enters a timeout
period, allowing components to cool
down before a restart sequence begins.
Figure 4. A 1.5MHz
+5V to ±12V dual
converter using one
LT8582
VIN
5V
this feature allows converter start-up to
be very smooth, even for hot-plug events.
Figure 4 illustrates how the GATE pin
provides short-circuit protection for
a boost converter. The circuit produces ±12V output from 5V input supply by utilizing channel 1 of the chip
as a boost converter and channel 2 as
a dual inductor, inverting converter.
L1
4.7µH
CIN1
4.7µF
215k
LTspice IV
circuits.linear.com/538
100k
D1
SWA1
FBX1
VIN1
PG1
LT8582
*MAX TOTAL OUTPUT POWER: 14.4W
130k
SS1
CLKOUT1
RT1
PG2
6.49k
53.6k
CIN2
4.7µF
4.7nF
53.6k
RT2
SS2
VIN2
VC2
2.2nF
0.1µF
47pF
14.7k
COUT3
10µF
25V
X7R
1206
143k
FBX2
L2
4.7µH
COUT2
10µF
25V
X7R
1206
GND
SHDN2
SWA2
47pF
VOUT1
12V
550mA*
0.1µF
GATE2
CIN1, CIN2: 4.7µF, 16V, X7R, 1206
C1: 2.2µF, 25V, X7R, 1206
D1, D2: DIODES INC. PD3S230H
L1: COILCRAFT XAL6060-472ML
L2, L3: COILCRAFT MSD7342-472
M1: FAIRCHILD FDMC510P
6.04k
VC1
SYNC1
SYNC2
215k
COUT1
10µF
25V
X7R
1206
GATE1
SHDN1
CLKOUT2
100k
SWB1
M1
SWB2
C1
2.2µF
L3
4.7µH
•
Reverse input voltage protection and
output short circuit protection can be
achieved, as shown in Figure 3, using
two external PFETs and the GATE pin. At
start-up, the channel’s supply voltage is
provided through the body diode of M2
while M1 keeps the power path disconnected. When the GATE pin is pulled down,
both PFETs turn on. If the input voltage
is reversed, the channel and the power
path are disconnected from the input
supply by M2. If the output is shorted,
the power path is disconnected from
the input supply by M1. The GATE pin
can be left floating when not in use.
Another use of the GATE pin is to limit
the converter start-up current. During
start-up, the GATE pin current increases
linearly with SS pin voltage, to a maximum current of ~1m A when the SS voltage
exceeds 500mV. This allows the external
PFET to slowly turn on and gradually
ramp up the output voltage. Together
with frequency foldback and soft-start,
•
GATE pin is a pull-down current source and
can control an external PFET during the
fault. The external PFET can disconnect the
input or the output, as shown in Figure 2.
Figure 3. Reverse battery and output short
protection
D2
VOUT2
–12V
550mA*
April 2012 : LT Journal of Analog Innovation | 13
Each channel of the LT8582 can be independently
configured in a boost, SEPIC, inverting or flyback topology.
A common weak point of the boost
topology is that it has a direct DC path
from input to output through the inductor and diode. An output short can result
in an uncontrolled increase of current
through the converter, likely destroying
one or more components in the DC path
and the power switch if it switches during this time. The LT8582 addresses this
issue by disconnecting the DC path if the
part senses an overcurrent condition.
For the dual inductor inverting and
SEPIC topologies, because of the series
capacitor in the power path, there is no
direct DC path between input and output
and the external PFET is not required.
The circuit in Figure 4 is running at a high
switching frequency of 1.5MHz. If thermal
issues arise, using larger ground planes and
better air flow helps remove extra heat.
D6
Figure 5. High voltage VFD
and filament bias supplies
D4
VOUT2
66V
C5
120mA*
2.2µF
D3
C3
2.2µF
L1
22µH
D2
CIN1
4.7µF
D1
CIN1, CIN2: 4.7µF, 25V, X7R, 1206
C1 TO C6: 2.2µF, 50V, X7R, 1206
C7: 2.2µF, 25V, X7R, 0805
C8: 10µF, 25V, X7R, 1210
D1 TO D6: CENTRAL SEMI CMMSH2-40
D7: 10V, CENTRAL SEMI CMHZ5240B
D8: CENTRAL SEMI CTLSH5-40M833
D9: CENTRAL SEMI CTLSH2-40M832
L1: WÜRTH 744771122
L2, L3: WÜRTH 744870100
M1: VISHAY SI7611DN
M1**
D7**
C1
2.2µF
D8**
8.06k**
SWA1 SWB1
576k
100k
FBX1
VIN1
GATE1
SHDN1
PG1
LT8582
SS1
CLKOUT1
RT1
SYNC2
SHDN2
VIN2
21k
80.6k
1.5nF
GND
80.6k
1.5nF
RT2
2.2µF
SS2
VC2
C8
10µF
×2
113k
C7
2.2µF
D9
•
SWB2
47pF
11.8k
FBX2
SWA2
47pF
2.2µF
GATE2
L2
10µH
*CHANNEL 1 MAX OUTPUT POWER 8W
**OPTIONAL FOR OUTPUT SHORT PROTECTION
CIN2
4.7µF
•
L3
10µH
VOUT3
10.5V
0.85A
90
2.0
80
1.6
70
1.2
60
0.8
50
0
2
6
4
OUTPUT POWER (W)
8
10
POWER LOSS (W)
576k
PG2
C2
2.2µF
VC1
SYNC1
CLKOUT2
100k
383k
EFFICIENCY (%)
VIN
9V TO 16V
Each channel of the LT8582 incorporates
a master and a slave switch, which are
rated at 1.7A and 1.3A, respectively. The
switches are driven in phase and only
the current through the master switch
is sensed by the internal current comparator. Normally, these switches are tied
together; when separated, they can be used
for building high voltage charge pumps,
as shown in Figure 5. The charge pump
VOUT1
100V
C6
80mA*
2.2µF
D5
C4
2.2µF
MASTER/SLAVE POWER SWITCH
0.4
Figure 6. Charge pump efficiency vs output power
14 | April 2012 : LT Journal of Analog Innovation
design features
Normally, the master and slave switches of each channel are tied together;
when separated, they can be used for building high voltage charge pumps.
The high output voltage can be used for low current loads such as vacuum
fluorescent displays (VFDs). In this case, the second channel of the LT8582
can be configured as a SEPIC converter to bias the filament of the VFD.
Figure 7. Charging/discharging supercapacitors in a
backup power supply
GATE1
VIN
CH1
SEPIC
VIN1
VOUT1
LT8582
VIN2
CH2
BOOST
VOUT2
VOUT
SUPERCAPS
circuit generates output voltages that are
higher than what the IC can tolerate.
not need series resistors that are typically
used to limit the capacitive current spikes.
The first stage of the charge pump circuit
is based on boost topology and uses the
channel’s master switch. The channel’s
slave switch is used to drive the other
charge pump stages, multiplying the
output voltage of the boost stage. The
benefit of this configuration is that the
master switch is immune from capacitive current spikes, allowing the LT8582
to sense the inductor current distinctly.
Moreover, the charge pump diodes do
The high output voltage can be used
for low current loads such as vacuum
fluorescent displays (VFDs). In this case,
the second channel of the LT8582 can be
configured as a SEPIC converter to bias
the filament of the VFD. Here, the master
and slave switches of channel 2 can be
tied together to increase output current.
Figure 6 shows the efficiency of the charge
pump circuit at various power levels.
Figure 8. Backup power supply using supercapacitors
VOUT
VIN (VIN > 11.4V)
11V (VIN < 11.4V)
M1
L1
5µH
6.04k
D1
VOUT1
10V
•
VIN
12V ±5%
C1
2.2µF
CIN1
4.7µF
•
SWA1
100k
73.2k
FBX1
PG1
GATE1
SHDN1
LT8582
11k
CLKOUT1
CLKOUT2
SYNC2
100k
L2
5µH
130k
COUT2
10µF
VC1
SS1
SYNC1
VOUT1
SWB1
VIN1
15.4k
80.6k
1.2k
1/4W
CS1
60F
1.2k
1/4W
CS2
60F
1.2k
1/4W
CS3
60F
1.2k
1/4W
CS4
60F
1nF
GND
80.6k
RT2
SS2
SHDN2
VC2
100pF
12.7k
GATE2
SWB2
3.3nF
0.47µF
105k
FBX2
SWA2
100pF
0.47µF
RT1
PG2
VIN2
L3
2.2µH
COUT1
4.7µF
D2
COUT3
22µF
×2
CIN1, CIN2: 4.7µF, 16V, X7R, 1206
COUT1: 4.7µF, 25V, X7R, 1206
COUT2: 10µF, 25V, X7R, 1210
COUT3: 22µF, 16V, X7R, 1210
C1: 2.2µF, 25V, X7R, 0805
CS1 TO CS4: 60F, 2.5V, COOPER HB1840-2R5606-R
D1, D2: CENTRAL SEMI CTLSH5-40M833
L1, L2: COOPER CTX5-1A
L3: COOPER HCM0703-2R2
M1: VISHAY SI7123DN
CIN2
4.7µF
April 2012 : LT Journal of Analog Innovation | 15
The VC current limit feature can be used in situations
where the load voltage may be low for an extended
period of time, such as when charging supercapacitors.
FAULT AND VC CURRENT LIMITS
The LT8582 has two distinct current limits:
the VC current limit, which is the maximum current that can be commanded, and
the FAULT current limit, which is the maximum current in case of converter overcurrent. The FAULT current limit is internally
set higher than the VC current limit. When
the FAULT current limit is reached, the chip
goes into fault mode and stops switching. However, when the VC current limit
is reached, the chip reduces the switch
duty cycle, reducing the output voltage.
The VC current limit feature can be used
in situations where the load voltage may
be low for an extended period of time,
such as when charging supercapacitors.
Figure 7 demonstrates how the VC current
limit along with the GATE pin can be used
to build a backup power supply using one
LT8582 and a bank of four supercapacitors.
The actual circuit is shown in Figure 8.
Here, channel 1 of the LT8582 is configured
as a SEPIC converter and is used to charge
the supercapacitor bank when VIN is
IL1 + IL2
2A/DIV
present. At this time, the GATE pin of channel 1 is enabled and the external PFET provides a path for the load current from the
input to the output. Once the input supply
is disconnected, channel 2 of the LT8582
which is configured as a boost converter,
provides voltage to the load without any
delay, while the external PFET disconnects the input from the output, preventing energy from going back into VIN .
The complete backup power supply
circuit is shown in Figure 8. With the
component values shown, the supercapacitor bank is charged to 10V when
VIN is above ~11.4V. Once VIN falls below
~11.2V, the circuit holds up VOUT at
11V for about 90 seconds with 500m A of
load current. The waveforms of interest during charging/discharging the
supercapacitors are shown in Figure 9.
PG PIN AND EVENT-BASED
SEQUENCING
The PG pin is an open drain active high
pin that indicates the output voltage is
close to regulation. For most applications
IL1 + IL2
2A/DIV
IL3
2A/DIV
VOUT ≈ VIN
5V/DIV
VOUT1
5V/DIV
IL3
2A/DIV
VOUT ≈ VIN
5V/DIV
VOUT1
5V/DIV
20s/DIV
20s/DIV
Figure 9. Waveforms of interest during charging (left) and discharging (right) the supercapacitors
16 | April 2012 : LT Journal of Analog Innovation
LT8582
CH1
MASTER
SHDN1
CH2
SLAVE
PG1
SHDN2
VIN
RUVLO1
RUVLO2
SHDNSYS
10k
SET RUVLO1 AND RUVLO2 SUCH THAT
VIN1UVLO < VIN2UVLO
SEE CONFIGURABLE UNDERVOLTAGE LOCKOUT
SECTION FOR DETAILS
Figure 10. Sequenced power supplies
this corresponds to an output voltage 8% from the target output voltage.
The SHDN pin is used to enable/disable
the channel. Driving the SHDN pin to
ground disables the channel while driving SHDN above 1.3V enables the channel.
Figure 10 shows how these two pins can
be used to turn on power supplies in
sequence as may be required in systems
with multiple voltage levels. When channel
1’s output voltage is close to regulation,
the PG pin of channel 1 releases channel
2’s SHDN pin, which enables channel 2.
To ensure that the status of channel 1’s
PG pin is valid while it is being sensed
by channel 2, channel 1 has to become
active first, i.e., VIN1 UVLO should be
set lower than VIN2 UVLO. To provide a
global shutdown signal for the system,
the SHDNSYS signal drives two NFETs that
disable both channels when it is high.
The complete circuit diagram
and start-up waveforms are presented in Figures 11 and 12.
design features
C1
2.2µF
L1
8.2µH
D1
CIN1
10µF
SWA1
SWB1
•
L2
8.2µH
VIN1
SHDNSYS
10k
10k
130k
FBX1
SHDN1
M1
VOUT1
12V
0.3A (VIN = 3V)
0.5A (VIN = 5V)
1A (VIN = 12V)
•
VIN
3V to 19V
115k
PG1
M2
GATE1
SYNC1
SS1
CLKOUT1
RT1
CLKOUT2
SYNC2
100k
COUT1
10µF
×2
VC1
LT8582
20k
107k
107k
1.5nF
RT2
SHDN2
SS2
VIN2
VC2
0.1µF
IL1 + IL2
2A/DIV
C2
2.2µF
•
D2
CIN2
10µF
•
VOUT2
2V/DIV
IL3 + IL4
2A/DIV
COUT2
22µF
×2
45.3k
FBX2
SWB2
47pF
14.7k
GATE2
SWA2
1.5nF
GND
PG2
L3
6.8µH
VOUT1
5V/DIV
47pF
0.1µF
L4
6.8µH
2ms/DIV
Figure 12. Start-up waveforms at VIN = 12V
VOUT2
5V
0.7A (VIN = 3V)
1A (VIN = 5V)
1.45A (VIN = 12V)
CIN1, CIN2: 10µF, 25V, X7R, 1210
COUT1: 10µF, 25V, X7R, 1210
COUT2: 22µF, 16V, X7R, 1210
C1,C2 : 2.2µF, 25V, X7R, 0805
D1, D2: CENTRAL SEMI CTLSH2-40M832
L1, L2: COOPER DRQ125-8R2
L3, L4: COOPER DRQ125-6R8
M1, M2: 2N7002
Figure 11. Sequenced 12V and 5V dual outputs
FB PIN AND SINGLE RESISTOR
VOLTAGE FEEDBACK
The LT8582 needs only one feedback pin
for both positive and negative output
voltages. In addition, only one external resistor from VOUT to FB is required
Figure 13. Tracking power supplies using one extra
resistor
VOUT1
CH1
BOOST
RFB1
+
C1
FBX1
LT8582
FBX2
CH2
INVERTING
RFB2
This feedback structure can be used to
design simple tracking power supplies
without using a tracking controller chip.
As shown in Figure 13, only one extra
resistor connected between the two feedback pins of LT8582 is needed for this.
RFB1, RFB12 and RFB2 form a resistor voltage divider. The more current through
them, the better the tracking. Thus,
the current through the connecting
RFB12
+
to set the output voltage. The internal feedback circuitry automatically
selects the correct reference voltage,
1.204V or 7mV for topologies with positive or negative outputs, respectively.
C2
resistor RFB12 must be relatively higher
than the FB1 and FB2 currents, so:
IFB12 =
1.204 – 7m
>> 83.3µ
RFB12
After selecting RFB12, the feedback resistors
RFB1 and RFB2 can be calculated as follows:
RFB1 =
RFB2 =
VOUT1 – 1.204
1.197
83.3µ +
RFB12
7m − VOUT 2
1.197
83.3µ +
RFB12
For the circuit shown in Figure 14,
plotting the output voltages vs
load currents yields Figure 15.
VOUT2
April 2012 : LT Journal of Analog Innovation | 17
The LT8582 only needs one feedback pin for both positive and negative output voltages.
In addition, only one external resistor from VOUT to FB is needed to set the output
voltage. The internal feedback circuitry automatically selects the correct reference
voltage, 1.204V or 7mV for topologies with positive or negative outputs, respectively.
L1
10µH
VIN
2.7V TO 5.5V
CIN1
10µF
D1
SWA1
PG1
6.04k
FBX1
LT8582
SS1
CLKOUT1
RT1
SYNC2
PG2
SHDN2
VIN2
COUT1
10µF
×2
VC1
SYNC1
CLKOUT2
6.65k
107k
107k
RT2
100pF
6.65k
SS2
VC2
COUT2
10µF
×2
53.6k
SWB2
C1
4.7µF
L3
15µH
•
CIN2
10µF
6.8nF
0.1µF
VOUT2
–15V
0.27A(VIN = 2.7V)
0.37A(VIN = 3.6V)
0.46A(VIN = 4.5V)
0.54A(VIN = 5.5V)
•
Figure 14. Dual tracking power supplies
using one LT8582
CIN1, CIN2: 10µF, 16V, X7R, 1206
COUT1, COUT2: 10µF, 25V, X7R, 1210
C1: 4.7µF, 50V, X7R, 1206
D1, D2: DIODES INC. PD3S230H
L1: COILCRAFT XAL6060-103ME
L2, L3: COILCRAFT MSD1260-153
6.8nF
GND
FBX2
SWA2
100pF
0.1µF
GATE2
L2
15µH
FBX2
GATE1
SHDN1
100k
49.9k
SWB1
VIN1
VOUT1
15V
0.3A(VIN = 2.7V)
0.42A(VIN = 3.6V)
0.56A(VIN = 4.5V)
0.69A(VIN = 5.5V)
D2
CONCLUSION
18 | April 2012 : LT Journal of Analog Innovation
The LT8582 is easy to use and robust.
Because of its high switching frequency
and monolithic structure, it can be used
to fit power converters into the tightest
spaces. The LT8582 is available in a tiny
24-pin 7mm × 4mm DFN package. n
Figure 15. Tracking output voltages vs load current
(load between the two outputs)
15.30
15.25
15V
15.20
MAGNITUDE VOUT (V)
The LT8582 is a dual independent monolithic converter with two 3A, 42V power
switches. In addition to popular features
such as soft-start, single-pin feedback and
single-resistor oscillator, it includes unique
features such as the master/slave power
switches, separate maximum commanded
and fault current limits, external PFET control for output or input disconnect,
FAULT protection, PG pin for power supply
sequencing, and CLKOUT signal for out-ofphase synchronizing and die temperature
monitoring. These features enable the
LT8582 to be used in a variety of applications, from typical dual rail voltage regulators to supercapacitor backup supplies.
15.15
15.10
–15V
15.05
15.10
14.95
14.90
0 0.05 0.1 0.15 0.2 0.25 0.3 0.35 0.4
LOAD CURRENT (A)
design features
2MHz Dual DC/DC Controller Halves Settling Time of
Load Release Transients, Features 0.67% Differential VOUT
Accuracy and Is Primed for High Step-Down Ratios
Shuo Chen and Terry Groom
Electrical conditions once considered extreme are now
the norm. Modern electronic systems demand high
currents and very low voltages that can appear to a
DC/DC converter as an intermittent electrical short. It is
not uncommon for sub-0.9V power supply rails to demand
25A or more. In this environment, tight total differential
regulation accuracy is critical to achieve the demanding
voltage tolerances required to power core processors
and large ASICs. In addition, PCB and component
size constraints have driven up converter operating
frequencies to enable the use of smaller components.
VIN
4.5V TO 38V
+
CIN1
100µF
CIN2
10µF
×3
LTspice IV
circuits.linear.com/546
VIN
0.1µF
3.57k
+
COUT2
330µF
×2
SENSE1–
SENSE2–
SENSE1+
SENSE2+
BOOST1
BOOST2
0.1µF
TG1
MT1
3.57k
TG2
MT2
DB2
SW1
1µF
15k
0.1µF
DB1
2.2Ω
COUT1
100µF
×2
LTC3838
0.1µF
15k
VOUT1
1.2V
15A
In addition to architectural advantages,
the proprietary detect transient release
(DTR) feature improves the transient
2.2Ω
1µF
L1
0.56µH
The LTC3838 and LTC3839 controllers are
designed to meet the needs of the most
demanding low output voltage, high load
current applications. Both feature superior
differential regulation accuracy and fast
transient response. The controlled on-time
architecture yields minimum on-times as
low as 30ns and is capable of switching
frequencies from 200kHz to 2MHz with
synchronization to an external clock.
4.7µF
DRVCC1
INTVCC
DRVCC2
EXTVCC
BG2
MB2
PGND
10k
VOUTSENSE1+
PGOOD1
0.01µF
Figure 1. 4.5V to 38V input, 1.2/15A,
1.5V/15A dual output, 350MHz step-down
converter. With the output sensed directly
through a resistor divider network, the
remote sensing scheme in channel 1
mimics the traditional feedback used in
channel 2. The LTC3838’s novel remote
sensing scheme eliminates the diff amp
output pin required in other parts.
40.2k
220pF
115k
COUT3
100µF
×2
10k
VOUTSENSE1–
PGOOD1
PGOOD2
PGOOD2
ITH1
DTR1
VRNG1
RT
SGND
RUN1
ITH2
DTR2
VRNG2
100k
0.01µF
TRACK/SS1 TRACK/SS2
22pF
COUT4 +
330µF
×2
15k
VFB2
10k
100k
VOUT2
1.5V
15A
SW2
BG1
MB1
L2
0.56µH
22pF
220pF
40.2k
CIN1: NICHICON UCJ1H101MCL165
CIN2: MURATA GRM32ER71H106K
COUT2, COUT4: SANYO 2R5TPE330M9
COUT1, COUT3: MURATA GRM31CR60J107ME39L
DB1, DB2: DIODES INC. SDM10K45
L1, L2: TOKO FDA1055-R56M
MT1, MT2: INFINEON BSC093N04LSG
MB1, MB2: INFINEON BSC035N04LSG
PHASMD
MODE/PLLIN
CLKOUT
RUN2
April 2012 : LT Journal of Analog Innovation | 19
The LTC3838 and LTC3839 controllers are designed to meet
the needs of the most demanding low output voltage, high
load current applications. Both feature superior differential
regulation accuracy and fast transient response.
ILOAD
10A/DIV
PLLIN
5V/DIV
ILOAD
10A/DIV
VOUT
50mV/DIV
AC-COUPLED
VOUT
50mV/DIV
AC-COUPLED
SW1
10V/DIV
IL
10A/DIV
SW2
10V/DIV
IL
10A/DIV
0°
180°
CLKOUT
5V/DIV
5µs/DIV
LOAD STEP = 0A TO 15A
VIN = 12V
VOUT = 1.2V
FORCED CONTINUOUS MODE
5µs/DIV
LOAD RELEASE = 15A TO 0A
VIN = 12V
VOUT = 1.2V
FORCED CONTINUOUS MODE
60°
500ns/DIV
VIN = 12V
VOUT1 = 5V, VOUT2 = 3.3V
LOAD = 0A
MODE/PLLIN = 333kHz EXTERNAL CLOCK
PHASMD = GND
Figure 2. Switching frequency is constant and phase locked during steady state, but fast transient performance is achieved
by momentarily adjusting the switching frequency: increasing it on a load step; decreasing it on a load release.
performance in high step-down ratio,
low output voltage applications. This
enables the LTC3838/LTC3839 to maintain accuracy and respond to load transients faster than other topologies.
In high output current supplies applications, it is important that overall regulation accuracy is well understood. To this
end, the LTC3838 and LTC3839 internally
combine the output differential amplifier and error amplifier and specify DC,
line and load regulation output voltage
errors as a single lumped parameter.
This allows the LTC3838 and LTC3839 to
achieve a level of total differential accuracy unavailable in other controllers.
The LTC3838 and LTC3839 make high frequency switching practical in a high input
voltage, low output voltage converter.
Both devices can produce high step-down
ratios at high switching frequencies while
maintaining high efficiency at heavy load
20 | April 2012 : LT Journal of Analog Innovation
currents—previously challenging due
to greater switching losses and limitations inherent in other architectures.
For instance, in the typical 12V input
to 3.3V/25A output application shown
in Figure 3, the LTC3838/LTC3839 delivers a peak efficiency of 93% at 2MHz.
FLEXIBLE DUAL/SINGLE OUTPUT,
HIGH ACCURACY REMOTE SENSE
The LTC3838’s dual channels can be
configured for either dual- or singleoutput applications, whereas the LTC3839
is dedicated for single-output applications. Both convert an input of 4.5V to
38V (40V abs max) down to outputs of
0.6V to 5.5V (6V abs max) in applications
with per-channel currents up to 25A.
Their remotely sensed differential feedback has a voltage regulation accuracy
of ±0.67%—where the remote power
ground can deviate as much as ±500mV.
The LTC3838’s second channel can provide an independent ±1% output, or
together with the first channel, serve as
one of the PolyPhase® channels for a
single-output, higher current application.
For higher load currents, or to maximize
efficiency, multiple LTC3838s and LTC3839s
can be paralleled for up to 12-phases.
FAST TRANSIENT PERFORMANCE,
CONSTANT FREQUENCY
The LTC3838 and LTC3839 employ the new
controlled on-time, valley current mode
architecture, primed for fast transient
performance. This architecture retains the
benefits of a constant on-time controller: it responds to sudden load increases
by a sequence of consecutive on-time
pulses with a very short 90ns off-time in
between, without having to wait until the
next switching cycle like that of a fixed
frequency controller. During a load release,
the LTC3838/LTC3839 delays the turn-on of
the top FET until inductor current drops
design features
The controlled on-time architecture yields minimum on-times
as low as 30ns and makes high frequency switching practical
in a high input voltage, low output voltage converter, while
maintaining high efficiency at heavy load currents.
CIN2
22µF
×4
2.2Ω
90
LTC3839
VIN
10Ω
SENSE1–
1nF
10Ω
1nF
SENSE1+
0.1µF
SENSE2+
BOOST1
VOUT
RS1
0.004Ω
MT1
L1
0.3µH
1µF
MB1
TG2
SW1
SW2
DRVCC1
INTVCC
BG2
VOUT
3.3V
25A
8
6
80
FORCED
CONTINUOUS
MODE
70
60
RS2
0.004Ω
DISCONTINUOUS
MODE
50
LOSS
FORCED
CM
0.1
4
2
LOSS
DCM
0
100
1
10
LOAD CURRENT (A)
MB2
PGND
45.3k
100
VOUTSENSE+
10k
VOUTSENSE–
PGOOD
0.01µF
PHASMD
TRACK/SS
MODE/PLLIN
150pF
ITH
DTR
18.7k
LTspice IV
CIN1: SANYO 16SVP180MX
CIN2: MURATA GRM32ER61C226KE20L
COUT1, COUT2: MURATA GRM31CR60J107ME39L
DB1, DB2: CENTRAL CMDSH-3
L1, L2: WÜRTH 7443340030
MT1, MT2: INFINEON BSC050NE2LS
MB1, MB2: INFINEON BSC032NE2LS
VRNG
CLKOUT
SGND
RUN
10
VIN = 5V
8
FORCED
CONTINUOUS
MODE
80
6
4
70
60
50
RT
circuits.linear.com/547
90
DISCONTINUOUS
MODE
LOSS
FORCED CM
0.1
1
10
LOAD CURRENT (A)
POWER LOSS (W)
PGOOD
33.2k
L2
0.3µH
DRVCC2
EXTVCC
BG1
100k
MT2
DB2
4.7µF
10Ω
0.1µF
BOOST2
TG1
DB1
2.2Ω
COUT1
100µF
×6
10Ω
SENSE2–
10
VIN = 12V
POWER LOSS (W)
1µF
EFFICIENCY (%)
CIN1
180µF
100
+
EFFICIENCY (%)
VIN
4.5V TO 14V
2
LOSS
DCM
0
100
Figure 3. A 2MHz, 3.3V/25A step-down converter. The LTC3838/LTC3839 can operate at switching frequencies above the AM radio band (fSW > 1.8MHz). The high
switching frequency permits the use of inductors of very small footprint, so that the entire circuit can fit within a 0.9in2 area with both sides populated. The peak
efficiency is 95%, and full load efficiency well above 90% at 25A, even at a frequency of 2MHz.
to desired value, preventing overcharging
the output capacitor. Once the transient
condition subsides, the switching frequency quickly returns to the programmed
nominal or external clock frequency.
Meanwhile, the on-time is adjusted
(hence controlled on-time) so that the
switching frequency is constant during
steady-state operation, synchronized to
its internal programmable or an external
clock, to mimic a fixed frequency controller with predictable switching noise.
HIGH AND WIDE STEP-DOWN RATIO,
SWITCHING FREQUENCY
The LTC3838/LTC3839’s 30ns minimum
on-time (60ns effective on-time with
dead-time delays) enables low duty cycles
for high VIN to low VOUT applications,
even while the part operates at high
frequency. The 90ns minimum off-time
helps achieve high duty cycle operation
and avoid output dropout when VIN is
only slightly above the regulated VOUT.
The LTC3838 and LTC3839 are capable of
a full decade programmable switching
frequency from 200kHz to 2MHz. They
can be synchronized to external clocks
of ±30% of the programmed frequency.
April 2012 : LT Journal of Analog Innovation | 21
In addition to the LTC3838/LTC3839’s architectural advantages,
the proprietary detect transient release (DTR) feature
improves the transient performance in low output voltage
applications. This enables these parts to maintain accuracy
and respond to load transients faster than other topologies.
LTC3838/LTC3839
EA
VREF
VFB
SW
5V/DIV
INTVCC
1/2 INTVCC
+
–
ITH
+
–
DTR
LOAD
RELEASE
DETECTION
TO LOGIC
CONTROL
DTR
1V/DIV
CITH2
(OPTIONAL)
BOTTOM MOSFET GATE
TURNS BACK ON, INDUCTOR
CURRENT (IL) GOES NEGATIVE
IL
10A/DIV
INTVCC
CITH1
BG
5V/DIV
RITH2
RITH1
DTR DETECTS LOAD RELEASE,
TURNS OFF THE BOTTOM MOSFET GATE
FOR FASTER INDUCTOR CURRENT (IL) DECAY
5µs/DIV
Figure 4. Transient detection is done through the detect-transient (DTR) pin, which is DC-biased slightly above ½ INTVCC, and AC-coupled to ITH pin through the
compensation capacitor CITH1. The equivalent compensation resistance RITH = RITH1 || RITH2 .
NOVEL TRANSIENT DETECTION
REDUCES LOAD-RELEASE VOUT
OVERSHOOT
As the output voltage becomes lower
and the VIN -to-VOUT step-down ratio
increases, a major challenge is to limit
the overshoot in VOUT during a fast
load current drop. An innovative feature of the LTC3838/LTC3839 is to detect
“load-release” transients indirectly by
monitoring the ITH negative slew rate.
The detection is done through the detecttransient (DTR) pin that is coupled to
ITH pin through the compensation capacitor. At steady state, the DTR pin remains
slightly higher than the detection threshold (half of the voltage on INTVCC pin)
with a voltage divider of the compensation resistors from INTVCC to SGND.
In the event of a sudden drop of load
current, the output voltage overshoots
and ITH slews down quickly. If the
DTR pin drops below half of INTVCC , the
22 | April 2012 : LT Journal of Analog Innovation
LTC3838/LTC3839 temporarily turns off the
bottom MOSFET, and the inductor current flows through the body diode of the
bottom MOSFET. This increases the reverse
voltage drop across the inductor, allowing
the inductor current to drop to zero faster,
lowering the VOUT overshoot by reducing
overcharging of the output capacitor.
Once the inductor current reaches
zero, the bottom MOSFET turns back
on to pull the inductor current to
negative, discharging the output
capacitor to recover regulation.
Figure 5. Load-release detect transient (DTR) feature significantly reduces VOUT overshoot and time to recover
regulation. (Shades are obtained with infinite persistence on oscilloscope triggered at load release steps.)
VSW
3V/DIV
VSW
3V/DIV
VOUT
50mV/DIV
AC-COUPLED
VOUT
50mV/DIV
AC-COUPLED
ITH
1V/DIV
ITH
1V/DIV
IL
10A/DIV
IL
10A/DIV
5µs/DIV
FIGURE 1 CIRCUIT, CHANNEL 1 MODIFIED:
• RFB2 = 0Ω, VRNG2 = SGND, CITH1 = 120pF, CITH2 = 0pF,
• FROM DTR1 PIN: RITH1/2 = 46.4k TO SGND, 42.2k TO INTVCC
VIN = 5V, LOAD RELEASE = 15A TO 5A, VOUT = 0.6V
5µs/DIV
• CONNECTION FROM RITH1/2 AND CITH1
TO DTR1 PIN REMOVED
• DTR1 PIN TIED TO INTVCC
design features
The LTC3838 and LTC3839 are
based on and have all features of the
single-channel controller LTC3833.
For a full discussion of the features
shared with LTC3833, refer to the cover
article, “Fast, Accurate Step-Down DC/
DC Controller Converts 24V Directly
to 1.8V at 2MHz” in the LT Journal
of Analog Innovation, October 2011
(Volume 21 Number 3). Download
at cds.linear.com/docs/LT%20
Journal/LTJournal-V21N3-2011-10.pdf
For More Information
VIN
4.5V TO 14V
+
CIN2
22µF
×4
CIN1
180µF
2.2Ω
1µF
VIN
LTC3839
SENSE1–
SENSE2–
SENSE1+
SENSE2+
BOOST1
BOOST2
0.1µF
0.1µF
2.55k
L1
0.33µH
VOUT
1.2V
50A
MT1
TG1
+
COUT2
330µF
×2
4.7µF
MB1
BG2
PGOOD
0.01µF
The LTC3838 and LTC3839 are high performance, feature-rich, 2-phase, synchronous
step-down DC/DC controllers that excel
at meeting the performance demands of
high current, low voltage loads, in either
dual or single output applications.
Their controlled on-time architecture
retains the fast response and low on-time
of traditional constant on-time controllers, and allows for constant frequency
and external clock synchronization.
Other unique features include
novel remote output sensing, which
VOUTSENSE–
PGOOD
PHASMD
TRACK/SS
47pF
CONCLUSION
MB2
COUT3 +
330µF
×2
COUT4
100µF
×2
VOUTSENSE+
10k
47.5k
VOUT
DRVCC2
EXTVCC
PGND
10k
Figure 6. The LTC3839 in a single 1.2V/50A
output, 2-phase, 300kHz, DCR sense, step-down
converter, with the detect transient load-release
(DTR) feature enabled for VOUT overshoot
reduction. The LTC3838 can also be used here.
The LTC3838/LTC3839 is ideal for powering low
voltage, high current, fast slew rate loads such as
with a microprocessor.
L2
0.33µH
SW2
DRVCC1
INTVCC
BG1
100k
2.55k
MT2
DB2
SW1
1µF
0.1µF
TG2
DB1
2.2Ω
COUT1
100µF
×2
0.1µF
470pF
41.2k
137k
ITH
MODE/PLLIN
DTR
VRNG
CIN1: SANYO 16SVP180MX
CIN2: MURATA GRM32ER61C226KE20L
COUT1, COUT4: MURATA GRM31CR60J107ME39L
COUT2, COUT3: SANYO 2R5TPE330M9
DB1, DB2: CENTRAL SEMI CMDSH-4ETR
L1, L2: VISHAY IHLP5050CEERR33M01
MT1, MT2: INFINEON BSC050NE2LS
MB1, MB2: INFINEON BSC010NE2LS
CLKOUT
RT
SGND
RUN
allows for a ±500mV remote ground,
and load-release transient detection for overshoot reduction.
In addition, LTC3838 and LTC3839
include popular features, such as:
•external VCC power pin for loss
reduction in the controller
•continuously programmable range
of current limits for flexibility with
either RSENSE or inductor DCR sensing
•selectable light load operating modes:
discontinuous operation (similar to
Burst Mode® operation) for higher
efficiency, or forced continuous
operation for constant frequency
•overvoltage protection and
current limit foldback
•soft-start/rail tracking, PGOOD,
and RUN pins for each output.
The LTC3838 is offered in 38-pin
QFN (5mm × 7mm) and TSSOP packages. The LTC3839 is offered in a 32-pin
QFN (5mm × 5mm). All packages
have exposed pads for enhanced
thermal performance. n
April 2012 : LT Journal of Analog Innovation | 23
Current Mode Switching Supply with Ultralow Inductor DCR
Sensing for High Efficiency and High Reliability
Jian Li, Haoran Wu and Gina Le
Current mode switching supplies have several advantages over voltage-mode switching
supplies: (1) high reliability with fast, cycle-by-cycle current sensing and protection;
(2) simple and reliable loop compensation—stable with all ceramic output capacitors;
(3) easy and accurate current sharing in high current PolyPhase supplies. In high
current applications, power losses in the current sensing component are a concern,
so the resistance of the sense component must be as low as possible. The problem
is that low resistance sensing elements produce reduced signal-to-noise ratios, such
that switching jitter becomes an issue in high current, high density applications.
The LTC3866 solves this problem by making it possible to build reliable current
mode switching supplies with <0.5mΩ current sensing resistance. This single-phase
synchronous buck controller drives all
N-channel power MOSFET switches with
onboard gate drivers. It employs a unique
architecture that enhances the signal-tonoise ratio of the current sense signal,
allowing the use of a very low DC resistance (DCR) power inductor or low value
current sensing resistor to maximize
efficiency in high current applications.
This feature reduces the switching jitter
commonly found in low DCR applications.
The controller has a wide 4.5V–38V input
range, remote output voltage sensing with
accurate 0.5% reference, programmable
and temperature-compensated current
limit when using inductor DCR sensing, short-circuit soft recovery without
overshoot, and chip thermal shutdown.
The LTC3866 facilitates the design of high
efficiency, high power density and high
reliability solutions for telecom systems, industrial and medical instruments,
and DC power distribution systems. The
controller is available in a low thermal
24 | April 2012 : LT Journal of Analog Innovation
Figure 1. LTC3866 current sensing
scheme with ultralow inductor DCR.
High current paths are shown with
thicker lines.
VIN
INTVCC
VIN
BOOST
INDUCTOR
LTC3866
RITEMP
ITEMP
TG
DCR
VOUT
BG
PGND
RS
22.6k
SNSD+
SNS–
RNTC
100k
L
SW
RP
90.9k
SNSA+
R1
R2
C1
C2
SGND
PLACE C1, C2 NEXT TO IC
PLACE R1, R2 NEXT TO INDUCTOR
R1C1 = 5 • R2C2
Figure 2. High efficiency,
1.5V/30A step-down
converter with very low
DCR sensing
100k
0.1µF
FREQ
MODE/PLLIN
RUN
PGOOD
TK/SS
ITEMP
30.1k
220pF
20k
10k
1.5nF
C1
220nF
C2
220nF
4.7µF
EXTVCC
ITH
VFB
220µF
VIN
4.5V TO 20V
LTC3866
VIN
DIFFOUT
INTVCC
DIFFP
BOOST
DIFFN
TG
SNSD+
SW
SNS–
SNSA+
ILIM
0.1µF
0.33µH
DCR = 0.32mΩ
BG
PGND
CLKOUT
SGND
R2
931Ω
R1
4.64k
COUT
470µF
×2
VOUT
1.5V
30A
design features
The LTC3866 employs a unique architecture that
enhances the signal-to-noise ratio of the current sense
signal, allowing the use of a very low DC resistance (DCR)
power inductor or low value current sensing resistor
to maximize efficiency in high current applications.
3
Burst Mode® OPERATION
PULSE-SKIPPING
MODE
CCM
90
EFFICIENCY (%)
80
70
TOP OF BOARD
BOTTOM OF BOARD
60
50
40
INDUCTOR
TOP FET
BOTTOM FET
30
20
VIN = 12V
VOUT = 1.5V
FSW = 400kHz
10
0
0.01
0.1
1
ILOAD (A)
10
100
LTC3866
12V INPUT
1.5V/30A OUTPUT
NO AIRFLOW
Figure 3. Efficiency of the circuit in Figure 2
Figure 4. Thermal test of the circuit in Figure 2
impedance 24-lead 4mm × 4mm QFN and
24-lead exposed pad FE packages.
It is especially well suited to low voltage, high current supplies because of a
unique architecture that enhances the
signal-to-noise ratio of the current sense
circuit. This allows it to operate with
the small sense signals produced by very
low DCR, 1mΩ or less, inductors, which
improve power efficiency in high current
supplies. The improved SNR minimizes
FEATURES
The LTC3866 uses a constant frequency
peak current mode control architecture, guaranteeing cycle-by-cycle
peak current limit and current sharing
between different power supplies.
Figure 5. Switching node jitter comparison at 12V input, 1.5V/25A output
STANDARD DCR SENSING
160ns
jitter due to switching noise, which could
corrupt the signal. The LTC3866 can sense
a DCR value as low as 0.2mΩ with careful PCB layout, though in this extreme
situation, the additional PCB and solder resistance should be considered.
As shown in Figure 1, the LTC3866 comprises two positive sense pins, SNSD+ and
SNSA+, to acquire signals and processes
Figure 6. Short circuit test
LTC3866 ENHANCED DCR SENSING
60ns
VOUT
1V/DIV
VSW
10V/DIV
VSW
10V/DIV
IL
10A/DIV
VIN = 12V
VOUT = 1.5V
ILOAD = 25A
100ns/DIV
VIN = 12V
VOUT = 1.5V
ILOAD = 25A
100ns/DIV
500µs/DIV
April 2012 : LT Journal of Analog Innovation | 25
APPLICATIONS
R2 • C2 = R1 • C1/5. An additional,
optional temperature compensation
circuit guarantees the accurate current
limit over a wide temperature range,
especially important in DCR sensing.
them internally to provide a 14dB (5×) signal-to-noise ratio improvement in response
to low voltage sense signals. The current
limit threshold is still a function of the
inductor peak current and its DCR value,
and can be accurately set from 10mV to
30mV in a 5mV steps with the ILIM pin.
The part-to-part current limit error is only
about 1mV over the full temperature range.
Figure 2 shows a high efficiency,
1.5V/30A step-down converter with
very low DCR sensing. An inductor with DCR = 0.32mΩ is used in this
design to maximize efficiency.
The LTC3866 also features a precise
0.6V reference with a guaranteed limit
of ±0.5% that provides an accurate
output voltage from 0.6V to 3.5V. Its
differential remote VOUT sensing amplifier makes the LTC3866 ideal for low
voltage, high current applications.
The filter time constant, R1 • C1, of the
SNSD+ path should equal the L/DCR of
the output inductor, while the filter at
SNSA+ path should have a bandwidth
five times larger than SNSD+, namely
The efficiency in different operation modes
is shown in Figure 3. The full load efficiency is as high as 90.3% at 12V input
voltage. It is about 1.4% improvement
over the supply using a 1mΩ sense resistor with the same power stage design.
The hot spot (bottom MOSFET) temperature rise is only 39.6°C without any
Figure 7. A high efficiency, 1.5V/80A power supply based on parallel LTC3866s and power blocks
100k
100k
INTVCC1
ITH
ITEMP
PGOOD
VIN
100k
ITH
VFB
PGOOD
ITEMP
INTVCC
PGND
CLKOUT
ILIM
SNSA+
SNS–
SGND
VGATE
TEMP–
GND
CS–
GND
CS+
BG
INTVCC2
+
330µF
+
330µF
GND
10Ω
10Ω
4.75k
10µF
VOUT
VIN
4.7µF
VIN1
VOUT1
VIN2
VOUT2
100µF
PWMH
CMDSH-3
BOOST
TG
100µF
VIN
VIN
SW
26 | April 2012 : LT Journal of Analog Innovation
TEMP+
470µF
INTVCC2
SNSD+
47nF
PWML
ACBEL POWER BLOCK
VRA001-4C1G
2.2Ω
DIFFP
47nF
VOUT2
INTVCC2
EXTVCC
LTC3866EUF
VIN2
VOUT
1.5V
80A
GND
1µF
MODE/PLLIN
FREQ
RUN
TK/SS
100k
DIFFN
0.1µF
INTVCC1
47nF
VOUT1
GND
BG
PGND
CLKOUT
SW
DIFFOUT
CMDSH-3
BOOST
SNSD+
VIN1
PWMH
INTVCC
TG
47nF
VOUT
4.7µF
EXTVCC
10µF
VIN
INTVCC1
DIFFP
ILIM
20k
DIFFN
SNSA+
30.1k
470µF
2.2Ω
LTC3866EUF
SGND
220pF
DIFFOUT
SNS–
5.36k
VFB
MODE/PLLIN
RUN
FREQ
TK/SS
0.1µF
1500pF
VIN
1µF
0.1µF
PWML
TEMP+
VGATE
TEMP–
GND
CS–
GND
CS+
ACBEL POWER BLOCK
VRA001-4C1G
330µF
+
330µF
GND
GND
GND
+
4.75k
design features
2.2Ω
LTspice IV
circuits.linear.com/548
10µF
×2
1µF
VIN
VIN
180µF 12V
×2
MODE/PLLIN
FREQ
20k
RUN
TK/SS
0.1µF
VOUT
500mV/DIV
ILOAD
50A/DIV
DIFFP
BOOST
DIFFN
TG
28.7k
SNSD+
SW
100pF
SNS–
SNSA+
PGND
ILIM
D1: CMDSH-3
M1: BSC024NE2LS
M2: BSC010NE2LS
D1
INTVCC
C1
220nF
4.7µF
VOUT
LTC3866
DIFFOUT
1nF
120k
ITEMP
EXTVCC
ITH
VFB
IL1 & IL2
10A/DIV
PGOOD
L1
1µH
DCR = 1.3mΩ
M1
M2
BG
R1
3.48k
CLKOUT
SGND
100µF
×2
330µF
×2
VOUT
5V
25A
10µs/DIV
Figure 8. Current sharing performance of the
1.5V/80A supply in Figure 7
Figure 9. High efficiency power supply, 12V
input to 5V/25A output
airflow, as shown in Figure 4, where the
ambient temperature is about 23.8°C.
output inductor signal and connect to the
SNSA+ pin. If the RC filter is used, its time
constant, R • C, is set equal to L/DCR of
the output inductor. In these applications, the current limit, VSENSE(MAX), is five
times larger for the specified ILIM, and
the operating voltage range of SNSA+ and
SNS– is 0V to 5.25V. Without using the
internal differential amplifier, the output
voltage of 5V can be generated as shown
in Figure 9. The thermal test shows that
the hot spot (the inductor) temperature
is about 57.3°C at full load without any
airflow, as shown in Figure 10, where
the ambient temperature is 25°C.
The unique design improves the efficiency, as well as the noise sensitivity.
The worst case switching node jitter is
reduced by 60%, as shown in Figure 5,
with a very low 0.32mΩ inductor DCR.
Another unique feature of LTC3866 is
short-circuit soft recovery. The internal
soft recovery circuit guarantees that
there is no overshoot when the power
supply recovers from a short-circuit
condition as shown in Figure 6.
The LTC3866 can be used with a power
block for a more compact design and very
high current. Figure 7 shows a dual-phase,
high efficiency, 1.5V/80A power supply
based on a 2× parallel LTC3866 + power
block scheme. Although the DCR of the
inductor in the power block is only
0.53mΩ, the current sharing performance
is excellent in both DC and transient
conditions, as shown in Figure 8.
In applications where higher value
DCR inductor or RSENSE is used, the LTC3866
can be used like any typical current mode
controller by disabling the SNSD+ pin,
shorting it to ground. An RSENSE resistor or a RC filter can be used to sense the
R3 R2
20k 147k
CONCLUSION
The LTC3866 delivers an outsized set of
features for its small 4mm × 4mm 24-pin
QFN package. The unique, ultralow
DCR current sensing with current mode
control makes the LTC3866 a good fit for
low voltage, high current applications
with high efficiency and high reliability.
Tracking, strong on-chip drivers, multichip
operation and external sync capability
fill out its menu of features. The LTC3866
is ideal for computer and telecom systems, industrial and medical instruments,
and DC power distribution systems. n
Figure 10. Thermal test of the circuit in Figure 9
TOP OF BOARD
INDUCTOR
BOTTOM FET
TOP FET
BOTTOM OF BOARD
LTC3866
13V INPUT
5V/25A OUTPUT
NO AIRFLOW
April 2012 : LT Journal of Analog Innovation | 27
Digital Power Management Reduces Energy Costs While
Improving System Performance
Andy Gardner
Today’s designers of networking equipment are expected to
push the limits of performance and add functionality under
the pressure of vanishingly short development times and tight
cost constraints. Increasing network system functionality
adds ASICs and processors, each requiring several voltage
rails, resulting in line cards with dozens of rail voltages.
The challenge with so many rails is to optimize hardware
utilization so that overall power consumption is minimized.
To meet this need, digital power management is fast emerging as a key
requirement in complex high reliability
applications. Digital power management allows complex multirail systems
to be efficiently debugged via PC-based
software tools, avoiding time-consuming
hardware changes. Software-based
in-circuit testing (ICT) and board bringup is much easier than in a traditional
hardware ECN approach since firmware
changes can be made on a PC, without
touching the board. Digital power management gives designers real-time telemetry data and fault logs, enabling fast
diagnosis of power system failures and
implementation of corrective action.
Perhaps most significantly, DC/DC converters with digital management functionality allow designers to develop green
power systems that optimize energy
usage while meeting system performance
targets (compute speed, data rate, etc.).
Optimization can be implemented at
the point of load, at the board, rack and
even at installation levels, reducing both
infrastructure costs and the total cost of
ownership over the life of the product.
This article shows how performance,
reliability and energy efficiency are
improved in network switches and routers, base stations and servers, as well
as industrial and medical equipment
through the use of the LTC2974 quadchannel digital power management IC.
The LTC2974 simplifies the sequencing
of any number of supplies. By using a
time-based algorithm, users can dynamically sequence supplies on and off in
any order. Sequencing across multiple
LTC2974s is also possible using the 1-wire
share-clock bus and one or more of the
bidirectional fault pins (see Figure 2). This
approach greatly simplifies system design
because channels can be sequenced in
any order, regardless of which LTC2974
provides control. Additional LTC2974s can
be added at any time without concern
for system constraints, such as a limited
supply of daughter-card connector pins.
Power-up sequencing can be triggered in
response to a variety of conditions. For
example, the LTC2974s can auto-sequence
when the downstream DC/DC POL converters’ intermediate bus voltages exceed a
particular turn-on voltage. Alternatively,
VIN
4.5V < VIBUS < 15V
I–
I+
VIN_SNS
VPWR
ISENSEP0
AUXFAULTB
PMBus
INTERFACE
SDA
LTC2974*
SCL
ALERTB
FAULTB0
FAULTB1
R10
VSENSEM0
SGND
VOUT_EN0
MMBT3906
TSENSE0
PWRGD
GND
VFB
LOAD
SHARE_CLK
WP
BG
R20
ASEL1
Figure 1. Quad power supply controller
with EEPROM. One channel is shown.
R30
VSENSEP0
CONTROL0
TO/FROM
OTHER
DEVICES
SW
VDAC0
VDD33**
DC/DC
CONVERTER
TG
ISENSEM0
OV
ASEL0
28 | April 2012 : LT Journal of Analog Innovation
SEQUENCE ANY NUMBER OF
SUPPLIES; ADD SUPPLIES AT WILL
WDI/RESETB
TO µP
RESETB
INPUT
RUN/SS
GND
0.1µF
WATCHDOG
TIMER INTERRUPT
*SOME DETAILS OMITTED FOR CLARITY
ONLY ONE OF FOUR CHANNELS SHOWN
**LTC2974 MAY ALSO BE POWERED
DIRECTLY FROM EXTERNAL 3.3V SUPPLY
design features
By using a time-based algorithm, users can dynamically sequence supplies on and off
in any order. Sequencing across multiple LTC2974s is also possible using the 1-wire
share-clock bus and one or more of the bidirectional fault pins. This approach greatly
simplifies system design since channels can be sequenced in any order, regardless of
which LTC2974 provides control. Additional LTC2974s can be added at any time.
LTC2974 #1
SHARE_CLK
FAULT
An integrated watchdog timer is available
for supervising external microcontrollers.
Two timeout intervals are available:
the first watchdog interval and subsequent intervals. This makes it possible to
specify a longer timeout interval for the
microcontroller just after the assertion
of the power good signal. If a watchdog
fault occurs, the LTC2978 can be configured to reset the microcontroller for a
predetermined amount of time before
reasserting the power good output.
INDIVIDUAL
MARGINING
FOR ALL
SUPPLIES
SEQUENCE
SUPPLIES
DOWN IN
ANY ORDER
LTC2974 #1
SHARE_CLK
FAULT
ROBUST SYSTEMS REQUIRE
VERSATILE FAULT MANAGEMENT
The bidirectional fault pins can be used
to establish fault response dependencies between channels. For instance, on
sequencing can be aborted for one or more
channels in the event of short-circuit. The
overvalue and undervalue limit thresholds
and response times of the voltage and current supervisors are all programmable. In
addition, input voltage, die temperature,
and four external diode temperatures are
also monitored. If any of these quantities
exceed their over- or undervalue limits, the
LTC2974 can be set to respond in a number
of ways, including immediate latchoff,
deglitched latchoff, and latchoff with retry.
SEQUENCE
SUPPLIES
UP IN
ANY ORDER
VOUT
0.5V/DIV
(AC-COUPLED)
LTC2974 #1
SHARE_CLK
FAULT
200ms/DIV
Figure 2. Multiple LTC2974s can be cascaded seamlessly using only two connections.
IMPROVE MANUFACTURING
YIELDS WITH ACCURATE VOLTAGE
MONITORING
As voltages drop below 1.8V, many offthe-shelf modules have trouble meeting
output voltage accuracy requirements
over temperature. Absolute accuracy
requirements of less than ±10mV are
now common, making it necessary to
trim the output voltage in manufacturing, a time-consuming process.
OEMs must margin test to ensure that
they ship dependable systems in the face
of drifting rail voltages, which can result
in significant manufacturing yield fallout. A far better solution to this problem embraces the reality of inaccurate
power modules, and enables the system
to self-trim in the field. The LTC2974’s
digital servo loop minimizes rail-voltage
drift by externally trimming the module’s
output voltage to better than ±0.25%
accuracy over temperature (see Figure 3).
In addition to improving manufacturing
yields, the digital servo loop makes it
easier to source power modules by avoiding the limitation of module accuracy.
ROBUST SYSTEMS A RESULT OF
EASY MARGINING
The LTC2974’s digital servo loop 10-bit
DACs allow users to margin power supplies
over a wide range while maintaining high
resolution for applications such as Shmoo
Figure 3. The LTC2974 offers excellent voltage servo
accuracy over temperature.
0.07
0.06
0.05
0.04
ERROR (%)
on sequencing can be initiated in
response to the rising- or falling-edge
of the control pin input. Immediate
turn-off or off-sequencing in response
to a fault condition is also available.
Sequencing can also be initiated by a
simple I2C command. The LTC2974 supports any combination of these conditions.
0.03
0.02
0.01
0
–0.01
–0.02
THREE TYPICAL PARTS
–0.03
–25
0
25
–50
50
TEMPERATURE (°C)
75
100
April 2012 : LT Journal of Analog Innovation | 29
LTC3601
INDUCTOR
TEMPERATURE SENSOR
Figure 4. Thermal image of a DC/DC converter showing the difference between the actual inductor temperature and the temperature sensing point.
plotting. Margining is controlled over the
I2C interface with a single command, and
the margin DAC outputs are connected to
the feedback nodes or trim inputs of the
DC/DC converters via a resistor. The value
of this resistor sets a hardware limit on the
range over which the output voltage can
be margined, an important safeguard for
power supplies under software control.
requirements. Accurate real-time telemetry greatly simplifies this task.
ACCURATE, TEMP COMPENSATED,
DCR LOAD CURRENT MONITORING
With the trend to lower and lower core
voltages, accurate measurement of load
currents has become a challenge, since the
use of a precision current sense resistor
can lead to unacceptable power losses.
One option is to use the DC resistance of
the inductor (DCR) as a current shunt element. This has several advantages, including zero additional power loss, lower
To achieve the desired savings in power
consumption, it is necessary to characterize the loads during all modes of operation. FPGA users optimize their code to
minimize power while ASIC users adjust
core voltages depending on throughput
Using the LTC2974, system health can be
determined from the voltage, current and
temperature status registers, while the
multiplexed, 16-bit ∆∑ ADC monitors input
and output voltages, output currents, and
internal and external diode temperatures.
0.25
AVERAGE IOUT ERROR (FULL-SCALE %)
INDUCTOR SELF-HEATING (°C)
12
10
8
6
τ
ΘIS
4
2
0
0
500
1000
1500
TIME (s)
2000
2500
Figure 5. LTC2974 compensates for inductor
self-heating using thermal resistance and delay
parameters.
30 | April 2012 : LT Journal of Analog Innovation
0.2
0.1
0.05
0
20°C
30°C
40°C
50°C
60°C
70°C
80°C
90°C
–0.05
–0.1
–0.15
–0.2
–0.25
0
0.5
1
The LTC2974 makes accurate DCR sensing
possible with a patent-pending temperature compensation algorithm that compensates for the thermal gradient from
the sense diode to the inductor’s core, as
well as the time lag that occurs between
changes in inductor current and temperature (see Figure 5). This capability,
combined with the LTC2974’s low noise
16-bit ∆∑ ADC, enable accurate measurement of load currents using inductors
with vanishingly small DCR (see Figure 6).
PC-BASED DESIGN AND
FAULT DIAGNOSTICS
–40°C
–30°C
–20°C
–10°C
0°C
10°C
0.15
circuit complexity and cost. However,
the strong temperature dependence of
the inductor resistance and the difficulty
in measuring the exact inductor core
temperature invariably introduces errors
in current measurement (see Figure 4).
1.5 2 2.5 3 3.5 4
CURRENT SET POINT (A)
4.5
5
Figure 6. Total current measurement error of the
LTC2974 for a DC/DC converter across the full range
of temperature and output current.
When used in conjunction with
LTpowerPlay™ software, the LTC2974’s
fault and warning registers allow designers
(and field users) to determine the status
of their power infrastructure at a glance
(see Figure 7). Status information, uptime,
and the last 500ms of ADC telemetry are
available in a data log. In the event a
channel is disabled in response to a fault,
the LTC2974’s data log can be dumped into
protected EEPROM. This 255-byte block
of data is held in non-volatile memory
until it is cleared with an I2C command.
design features
Figure 7 shows the data log contents
viewed in LTpowerPlay’s LTC2974 interface. In this way, the LTC2974 provides
a complete snapshot of the state of the
power system immediately preceding
the critical fault, thus making it possible
to isolate root cause well after the fact.
This is an invaluable feature for debugging both prerelease characterization and
in-field failures in high reliability systems.
STANDALONE OPERATION
The easy-to-use PC-based LTpowerPlay
software allows users to configure the
LTC2974 via a USB interface and a dongle
card. LTpowerPlay software, which is free
and downloadable, takes much of the
coding out of the development process and
improves time-to-market by allowing the
designer to configure all device parameters within an intuitive framework.
Once a device configuration has been
finalized, the designer can save the
parameters to a file and upload it to the
Linear Technology factory. Linear can use
the file to preprogram parts, thus allowing customers to bring up their boards
with maximum ease. When the onboard
EEPROM has been configured, the LTC2974
is capable of complete autonomous operation without the need for custom software.
Furthermore, the addition of one tiny
connector allows LTpowerPlay software to
communicate with the LTC2974 in-system,
providing field users access to telemetry,
system status and the fault log as needed.
CONCLUSION
The LTC2974 digital power manager brings
unprecedented parametric accuracy, a rich
feature set and an expandable modular
architecture to high availability systems.
Design of complex multirail systems is
simplified with the LTC2974. It uses an
industry-standard PMBus interface, it
interfaces directly with high powered, free
PC-based LTpowerPlay control software,
and it includes an integrated EEPROM for
complete customization. Design your
application with the LTpowerPlay design
tool and simply upload the configuration to the Linear Technology factory.
Linear can use your custom configuration to produce pre-programmed devices
ready-to-use in your application. n
Figure 7. LTpowerPlay software
allows the designer to plug
a PC into the system via a
tiny connector, enabling the
power-management system to
be completely configured and
controlled without writing a
single line of code.
April 2012 : LT Journal of Analog Innovation | 31
What’s New with LTspice IV?
Gabino Alonso
Stepping Parameters
in LTspice IV
NEW HOW-TO VIDEOS
Stepping Parameters
video.linear.com/103
There are two ways to examine a circuit
by changing the value of a parameter.
You can manually enter each value,
then resimulate the circuit, or you can
use the .STEP command to sweep across
a range of values in a single simulation
run for a side-by-side comparison. This
video provides an overview of the basic
steps for using the .STEP command to
perform repeated analysis of a circuit.
FULLY DIFFERENTIAL OPERATIONAL
AMPLIFIER DEMO CIRCUITS
(from Tyler Hutchison)
Analyzing and interfacing with fully
differential op amps in simulation may
prove more complicated than with
familiar, single-ended output op amps.
These demo circuits provide examples of
interfacing to fully differential amplifiers
What is LTspice IV?
LTspice® IV is a high performance SPICE
simulator, schematic capture and waveform
viewer designed to speed the process of power
supply design. LTspice IV adds enhancements
and models to SPICE, significantly reducing
simulation time compared to typical SPICE
simulators, allowing one to view waveforms for
most switching regulators in minutes compared
to hours for other SPICE simulators.
LTspice IV is available free from Linear
Technology at www.linear.com/LTspice. Included
in the download is a complete working version of
LTspice IV, macro models for Linear Technology’s
power products, over 200 op amp models, as
well as models for resistors, transistors and
MOSFETs.
32 | April 2012 : LT Journal of Analog Innovation
Follow @LTspice on Twitter for
up-to-date information on models, demo circuits,
events and user tips: www.twitter.com/LTspice
including single-ended and differential
impedance matching, noise analyses,
mixer and high speed ADC signal chain
models, and gain modification of fixedgain amplifiers with external resistors.
• LTC6405: Fully differential ADC driver
with simplified mixer and ADC models
www.linear.com/LTC6405
• LTC6400-20: Single-ended impedance
matching for fully differential amplifiers www.linear.com/LTC6400-20
• LTM®8048: Isolated µModule
DC/DC converter with LDO post regulator
(4V–30V to 6V at 100m A & 5V at
100m A) www.linear.com/LTM8048
• LTC6400-20: Differential impedance
matching for fully differential amplifiers www.linear.com/LTC6400-20
• LTC6400-20: Noise simulation
for fully differential amplifiers
www.linear.com/LTC6400-20
• LTC6401-8: Unity gain for fully differential
amplifiers with internal feedback
resistors www.linear.com/LTC6401-8
• LTC6404-1: Impedance matching and
noise measurements for fully differential
amplifiers www.linear.com/LTC6404
NEW DEMO CIRCUITS
µModule Regulators
• LTM8052: 36V, 5A, 2-quadrant CVCC stepdown µModule regulator (6V–36V to
2.5V at ±5A) www.linear.com/LTM8052
Switching Regulators
• LT3758: High efficiency SEPIC (8V–72V to
5V at 2A) www.linear.com/LT3758
• LT3759: Boost with low input voltage
range (1.8V–4.5V to 5V at 2A)
www.linear.com/LT3759
• LT3759: SEPIC with wide input voltage
range (2.8V–36V to 12V at 1A)
www.linear.com/LT3759
Download this demo circuit showing impedance matching and noise
measurements for fully differential amplifiers www.linear.com/LTC6404
design ideas
• LTC4000/LTC3891: High voltage high
current step-down, PowerPath
controller & Li-ion battery
charger (24V–60V to 16.8V at 6A)
www.linear.com/LTC4000
Amplifiers, References,
Comparators and TimerBlox ® IC s
• LT6108/LTC6994: Energy-tripped
circuit breaker with
automatic delayed retry
(5V–80V input, 500m A threshold)
www.linear.com/LT6108
PARAMETRIC PLOTS
Plotting results in LTspice IV is as easy as clicking
on a node to show voltage, or a component to show
current—the trace is then displayed in the waveform
viewer. In transient analysis, the horizontal axis
defaults to showing time, but you can always change
the horizontal axis to show other quantities (such as
current) to validate model parameters.
To change the default settings of the x-axis:
1.Click on a node/component to plot its voltage/
current in the waveform viewer.
2.Move the cursor to the horizontal axis of the
waveform viewer (the cursor will turn into a ruler)
and left-click.
3.In the Horizontal Axis dialog, enter an expression for
the “Quantity Plotted.”
4.Click OK.
Happy simulations!
NEW DEVICE
MACRO MODELS
Switching Regulators
• LT8582: Dual 3A boost/inverting/
SEPIC DC/DC converter with fault
protection www.linear.com/LT8582
• LTC3103: 1.8µ A quiescent current, 15V, 300m A synchronous
step-down DC/DC converter
www.linear.com/LTC3103
• LTC3115V–1: 40V, 2A synchronous
buck-boost DC/DC converter
www.linear.com/LTC3115-1
• LTC3613: 24V, 15A monolithic
step-down regulator with
differential output sensing
www.linear.com/LTC3613
• LTC3839: Fast, accurate, 2-phase,
single-output step-down
DC/DC controller with differential output
sensing www.linear.com/LTC3839
• LTC3866: Current mode synchronous
controller for sub-mΩ DCR sensing
www.linear.com/LTC3866
• LTC3876: Dual DC/DC controller
for DDR power with differential
VDDQ sensing and ±50m A VTT reference
http://www.linear.com/LTC3876
This parametric plot compares the
instantaneous power dissipation
(Alt + click on component) of
an LTC4358 5A ideal diode to a
Schottky diode. In this example,
the quantity plotted on the
horizontal axis has been changed
from the default of “time” to
“load current.” The resulting
plot confirms the advantage
of the LTC4358 as a low loss
replacement to Schottky diodes
in high current diode applications.
More information is available at
www.linear.com/LTC4358.
• LTC4366: High voltage surge stopper
www.linear.com/LTC4366 n
LTC4358-BASED, LOW LOSS,
IDEAL DIODE SOLUTION
Power User Tip
Download the LTspice IV demonstration circuit for this energy-tripped
circuit breaker with automatic delayed retry at www.linear.com/LT6108
R3
3.66k
LOAD SUPPLY
5V TO 80V DC
R4
0.667V/A 40.2k
LT6108-2
CURRENT
SENSE AMP
R1
100Ω
RSENSE
10mΩ
–
+
5V
+
60ms TO 350ms
DELAY TIMER
LT1783
IN
R6
64.9k
R3
6.34k
400mV
–
+
LRELAY
Q1
NPN
R2
1.62k
5V
–
COMPARATOR
CIRCUIT BREAKER
Hot Swap™ Controllers and
Surge Stoppers
• LTC4232: 5A integrated Hot Swap
controller www.linear.com/LTC4232
SCHOTTKY DIODE
(B530C)
R14
15k
5
0
CURRENT SENSE
COMPARATOR
500mA TRIP
THRESHOLD
OUT
LTC6994-2
GND
SET
5V
R7
301k
R10
10k
ROPTIONAL
931k
V+
5V
DIV
R8
172k
R9
100k
CLOSED
OPEN
LOAD
April 2012 : LT Journal of Analog Innovation | 33
Dual Monolithic Ideal Diode Extends Battery Run Time and
Prioritizes Power Sources with Glitch-Free Switchover
Joshua Yee
Mobile devices increasingly
rely on several power
sources that must be
interchangeable on the fly,
such as a wall adapter and
a backup battery, where
switchover between sources
must be transparent and
immediate. The simplest
scheme for switching
between two power sources
is a simple diode-OR, but
at higher current levels,
power losses in the diode
are a problem. To minimize
power losses and maximize
battery run time, replace the
Schottkys in a traditional
diode-OR with the LTC4415
dual monolithic ideal diode.
IN1
100k
21.5k
R1
R2
IN2
+
OUT1
4.7µF
D2
TO
LOAD
470k
470k
470k
470k
OUT2
GND
Figure 1. Automatic ideal diode switchover between wall adapter and battery
FEATURES
Figure 2 shows that the LTC4415 conducts with a regulated forward voltage
drop of 15mV when the load current
is below 500m A, an improvement of
>28mV compared to the LTC4413. Once the
load current climbs above that, LTC4415
operates with an on-resistance of 50mΩ, a
140mΩ improvement over the LTC4413 in
this region. At 4A, the LTC4415 produces
a mere 200mV forward drop, a reduction of over 50% compared to a typical
Schottky diode, which would produce
a forward drop greater than 400mV.
Because of its low forward voltage drop,
LTC4415 dissipates less than half the
power of a typical, low reverse leakage Schottky diode at 4A and less than
a quarter at 2A, as shown in Figure 3.
Another useful feature of the LTC4415
is short-circuit protection. The current
limit of LTC4415 can be adjusted up to
4A with R1 and R2. Eliminating R1 and
R2, and grounding CLIM1 and CLIM2,
2
CONSTANT CURRENT
1.8
ILIM
1.6
3
2
RON = 50mΩ
SCHOTTKY
DIODE
MBRS410E
CONSTANT
RESISTANCE
1
0
POWER LOSS (W)
LTC4415
1.4
1.2
1
SCHOTTKY
0.8
POWER
SAVINGS
0.6
0.4
CONSTANT
VOLTAGE
0
100
300
400
200
FORWARD VOLTAGE DROP (mV)
500
Figure 2. The LTC4415 I-V curve and regions of
operation vs a typical Schottky diode
34 | April 2012 : LT Journal of Analog Innovation
D1
LTC4415
EN1
CLIM1
STAT1
CLIM2
WARN1
WARN2
STAT2
EN2
SECONDAY
POWER
SOURCE
(BAT)
4
LOAD CURRENT (A)
The LTC4415 efficiently and intelligently switches between two
power sources in an input voltage
range of 1.7V–5.5V. When used in
the diode-OR application as shown
in Figure 1, LTC4415 delivers load
current through ideal diode D1 from
the wall adapter until its voltage
drops below the 4.5V switchover
threshold set by the resistor divider
on EN1/EN2. Once D1 is disabled, the
battery delivers load current through
D2 instead. STAT1 and STAT2 indicate
which ideal diode is conducting.
PRIMARY
POWER
SOURCE
(ADAPTER)
LTC4415
0.2
0
0
0.5
1
1.5
2 2.5
ILOAD (A)
3
3.5
4
Figure 3. The LTC4415 dissipates only 800mW at
4A, which is over 50% lower than the 1700mW
dissipated by a Schottky diode
4.5
design ideas
The LTC4415 is an easy-to-use, high performance
ideal-diode-OR solution for instantaneous power
supply switchover. It requires as few as four
external resistors and one output capacitor.
LTC4415
IOUTX
1000
CLIMX
R CLIMX = 1000 •
OUTX
INX
0 . 5V
I LIM
RCLIMX
124Ω
PX
UVLOX
ENX OUTX
GATE
DRIVER
UP TO 4A
CURRENT
LIMIT
TO WARNX
RLOAD
Figure 4. Current limit detection inside the
LTC4415 for each channel
instead triggers the internal limit at 6A.
Current limit is accomplished by a novel
approach that allows the LTC4415 to both
detect the load current and simultaneously produce a scaled analog voltage
for load current monitoring across the
same current limit resistor. Figure 4 shows
a simplified block diagram. This eliminates series losses, and saves board space
and BOM costs associated with a current
sense resistor and amplifier circuit.
LTC4415 P-channel MOSFETs are optimized for minimal on-resistance with
rapid switchover between sources
VIN2 = 4.6V
VIN1,2
1V/DIV
VIN1 = 3.6V
VIN2 = 2.6V
VOUT1,2
1V/DIV
STAT1
5V/DIV
STAT2
5V/DIV
without any appreciable load droop.
Figure 5 shows the LTC4415 switching
between input sources of different voltages, with only a 200mV transient dip
and recovery within 20µs. Note that
transient voltage spikes are usually
caused by inductive connections. This
can be reduced with short leads, proper
layout technique, and input and output
bypass capacitors with appropriate ESR.
For status monitoring purposes, the
active-low signals of STATx and WARNx
provide feedback to a digital controller/processor. STATx reflects conduction
VOUT
2V/DIV
OUTPUT SHORTED
TO GND
IOUT
2A/DIV
3.55V
20µs/DIV
COUT = 47µF
RLOAD = 3.6Ω
VOUT1 = VOUT2 (SHORTED)
Figure 5. Rapid path switchover with only 5%
transient voltage dip
STAT
5V/DIV
WARN
5V/DIV
THERMAL
SHUTDOWN
VIN = 3.6V
RCLIM = 124Ω
COUT = 4.7µF
RESTART DUE TO
THERMAL HYSTERESIS
10ms/DIV
Figure 6. Current limit warning and thermal
shutdown on output short circuit
status of a given channel. It can also be
used to detect failure of a source. WARNx
serves the dual purpose of indicating if
a path is in current limit—when STATx
is also low—or in thermal shutdown.
Thermal shutdown is triggered when
die temperature exceeds 160°C. Figure 6
shows how these two signals reflect the
system behavior when a path transitions
back and forth between current limit
and thermal shutdown. For about 25ms
after the output is shorted, the current
limit is active and WARNx stays low. Then
STATx goes high as thermal shutdown is
triggered. Restart occurs as the device
cools below 140°C, but shuts down
repeatedly due to the persistent short.
CONCLUSION
The LTC4415 is an easy-to-use, high
performance ideal-diode-OR solution for
instantaneous power supply switchover.
It requires as few as four external resistors and one output capacitor. The
low power loss and status monitoring makes LTC4415 an obvious choice
in applications requiring dual diodes
with built-in protection features.
LTC4415 is offered in both 3mm × 5mm
16-pin DFN and MSOP packages. n
April 2012 : LT Journal of Analog Innovation | 35
Single-IC Supercapacitor-Based Power Supply Backup
Solution
Ashish Kirtania
Supercapacitors are used in an increasing number of applications that require a ready
source of backup energy that can be called on to provide short-term power when regular
input power is lost. In these applications, supercapacitors have a number of advantages
over traditional energy storage devices such as batteries, including low maintenance
requirements, virtually unlimited cycle life, and low effective series resistance. The
LTC3226 simplifies the design of supercapacitor-powered backup application with a
single-IC solution that charges the supercapacitor when input power is available, and then
delivers energy from the supercapacitor to the load when nominal input power fails.
DESCRIPTION
Figure 1 shows a typical 3.3V backup supply application in which the main power
path from the input source to the load
goes through the external PFET. As long
as input power is available, the LTC3226
maintains the supercapacitor stack at a
full 5V charge. If the input voltage falls
below 3.15V, the 1.2F supercapacitor stack
becomes the supply, supporting a 2A load
at 3.3V for 600ms (See Figure 2). Achieving
a seamless transition from main supply
to backup storage requires four principal
circuit components: a dual mode (1×/2×)
charge pump with automatic cell balance and cell voltage clamp, an LDO to
supply the load current during backup,
an ideal diode controller to prevent the
LDO from back-driving the input supply, and a power-fail comparator to
detect the input voltage threshold below
which a backup needs to be initiated.
The dual-mode constant-frequency
(900kHz) low noise charge pump charges
the supercapacitor stack to an externally
programmed target voltage. The input current to the charge pump is programmed by
an external resistor between the PROG pin
and GND. At the beginning of a charge
cycle, when the CPO pin voltage is less than
VIN, the charge pump operates in 1× mode,
acting like a pass element, and the charge
current is approximately equal to the
programmed input current. As the CPO pin
voltage rises to within 200mV of VIN, the
charge pump enters 2× mode (voltage
doubler) and the charge current drops to
half of the programmed input current.
(continued on page 38)
Figure 1. 3.3V backup supply
MPEXT
LTC3226
VIN
LDO
2.2µF
1.96M
2.2µF
VOUT
CPO
VIN
C–
PROG
33.2k
5
255k
RST_FB
+
–
C+
1.21M
GATE
LDO_FB
1.2V
PFI
VMID
CHARGE
PUMP
PFO
RST
CAPGOOD
36 | April 2012 : LT Journal of Analog Innovation
80.6k
5V
4
VOUT
3
VIN
2
BACKUP
BACKUP MODE
MODE
(LDO IN
(LDO IN
REGULATION) DROPOUT)
1
CSC
1.2F
3.83M
1.21M
GND
COUT
47µF
–1
CSC = 1.2F
COUT = 47µF
ILOAD = 2A
CPO
0
CPO_FB
EN_CHG
6
VOLTAGE (V)
VIN
3.3V
Figure 2. 3.3V backup supply timing diagram
TO LOAD
(2A)
PFO (2V/DIV)
0
0.4
0.8
1.2
TIME (SECONDS)
1.6
2.0
design ideas
µModule Converters Take the Hassle Out of Designing
Isolated Power Supplies
David Ng
Sometimes a system needs a little bit of isolated power, but designing an isolated power
supply is rarely easy. The nature of isolated supplies makes them complicated and touchy,
resulting in late nights and long weekends spent on design and debug. The LTM8047
and LTM8048 µModule converters take the hassle out of designing isolated power
supplies, placing a flyback regulator in a compact 9mm × 11.25mm × 4.92mm BGA
RoHS compliant package. The controller, power switching and rectification elements,
as well as transformer and isolated feedback circuitry are all integrated. Both parts
operate from 3.1V to 32V inputs and produce over 1W of isolated power. The LTM8048
is identical to the LTM8047, but adds an integrated 300mA linear post regulator.
The linear post regulator integrated
into the LTM8048 is a high performance 300m A device, boasting a low
dropout of less than 450mV at room
temp, full load. As shown in Figure 4,
the output noise and ripple of the post
LTM8047
VIN
2.2µF
RUN
BIAS
4.7µF
7.15k
LTspice IV
ADJ
circuits.linear.com/553
VOUT
5V
280mA
(15VIN)
VOUT
ISOLATION BARRIER
VIN
3.1V TO 29V
SS
GND
22µF
VOUT–
725VDC ISOLATION
Figure 2. The LTM8048 is the
LTM8047 with the addition of
an LDO post regulator.
LTM8048
VIN
3.1V TO 30V
2.2µF
VIN
VOUT1
RUN
VOUT2
BIAS
4.7µF
LTspice IV
circuits.linear.com/550
6.19k
ADJ1
SS
GND
ISOLATION BARRIER
As is the case with most flyback converters, the output voltage can be above
or below the input, accommodating
a wide range of operating conditions.
And, as is nature of flyback converters,
the amount of current that the LTM8047
and LTM8048 can deliver depends on
the input voltage. Figure 3 shows the
load capability of a typical LTM8047 at
2.5V, 3.3V and 5V outputs. The LTM8048
features the same load capability.
Figure 1. The LTM8047 only requires four
additional components to implement an
isolated 5V power supply that accepts an
3.1V–29V input.
5.7V
VOUT2
5V
BYP
22µF
ADJ2
162k
10µF
VOUT–
725VDC ISOLATION
500
BIAS = VIN IF VIN ≤ 5V
450 BIAS = 5V IF VIN > 5V
MAXIMUM VOUT1 LOAD (mA)
Designing with the LTM8047 and LTM8048
is easy. Figure 1 shows a complete
LTM8047-based isolated power supply,
requiring only the addition of capacitors for input, output, and biasing, and
a resistor to set the output voltage. The
LTM8048 requires only one more component: a resistor to set the voltage of
the LDO output, as shown in Figure 2.
400
350
VOUT
500µV/DIV
(AC-COUPLED)
300
250
200
2.5VOUT1
3.3VOUT1
5VOUT1
150
100
0
5
10
15
VIN (V)
20
25
30
Figure 3. Maximum load capability of the LTM8047
and LTM8048 depends on the input voltage.
1µs/DIV
Figure 4. The output noise of the LTM8048 post
regulator is less than 1mV.
April 2012 : LT Journal of Analog Innovation | 37
The LTM8047 and LTM8048 are two flyback µModule converters
that can be used to produce more than 1W of isolated power from
a small, easy-to-use, 9mm × 11.25mm × 4.92mm BGA package.
VIN
RUN
BIAS
4.7µF
The LTM8047 and LTM8048 are two flyback µModule converters that can be used to produce more than 1W of isolated
power from a small, easy-to-use, 9mm × 11.25mm × 4.92mm
BGA package. The LTM8048 is nearly identical to the LTM8047,
but with an integrated high performance post regulator. n
ADJ
SS
1µF
22µF
VOUT–
GND
725VDC ISOLATION
22µF
LTM8047
VIN
2.2µF
RUN
BIAS
4.7µF
CONCLUSION
7.15k
5V
280mA
(15VIN)
VOUT
ISOLATION BARRIER
The LTM8047 and LTM8048 both integrate a transformer that is rated for 725VDC isolation. Every isolated
µModule converter is factory tested for 100% reliability, with 725V applied in one direction for one second,
followed by the reverse voltage for one second.
For flexibility, there is no circuitry connected between the
primary and secondary, so if a safety capacitor or other elements are required for a system, they can be added. This flexibility allows various configurations of the output. As shown
in Figure 5, for example, two LTM8047s can be combined to
deliver individually regulated positive and negative outputs.
LTM8047
VIN
3.5V TO 31V
2.2µF
7.15k
ADJ
SS
1µF
VOUT
ISOLATION BARRIER
regulator is less than 1mV. These measurements were
taken using a 150MHz HP-461A differential amplifier.
GND
22µF
VOUT–
725VDC ISOLATION
–5V
280mA
(15VIN)
Figure 5. Use two LTM8047 converters to produce ±5V from a 3.5V–31V input.
LTC3226, from page 38
One of the limitations of supercapacitors
is low cell voltage, typically 2.7V, requiring a series connection of two cells for
5V applications. Since supercapacitors
have more self-discharge due to leakage than most batteries, they require cell
balancing to prevent overcharging of
one of the series capacitors. The LTC3226
charge pump is equipped with an active
balancer circuit, thus eliminating the
need for external balancing resistors.
However, since this balancer has limited
source and sink capability, the charge
pump is equipped with voltage clamp
circuitry which constantly monitors cell
38 | April 2012 : LT Journal of Analog Innovation
voltages during the charging process and
prevents the cells from overcharging.
A fast comparator detects when the
input voltage falls unacceptably low
and enables the LDO which powers the
load from the supercapacitors. This
power-fail threshold is programmed
by an external resistor divider via the
PFI pin. The output of the PFI comparator drives an open-drain output on the
PFO pin to indicate the status of the input
source. An external resistor divider to the
LDO_FB pin sets the LDO output voltage.
CONCLUSION
The LTC3226 enables seamless supercapacitor-based power backup solutions by integrating the functions of
a charge pump, an LDO and an ideal
diode controller in a compact low profile 3mm × 3mm 16-pin QFN package. Its
low 50µ A quiescent current and small
footprint make it particularly suitable
for battery powered applications, as
well as 3.3V systems that require protection from short power interruptions. n
product briefs
Product Briefs
MULTIPHASE CURRENT MODE STEPUP DC/DC CONTROLLER SUPPLIES
10V GATE DRIVE, RIDES THROUGH
COLD CRANK
The LTC3862-2 is a high power multiphase
current mode step-up DC/DC controller.
Like its predecessors, the LTC3862 and
LTC3862-1, the -2 uses a constant frequency, peak current mode architecture
with two channels operating 180° out of
phase. It retains popular features, including adjustable slope compensation gain,
max duty cycle and leading edge blanking,
programmable frequency with a external
resistor (75kHz to 500kHz) or SYNC to
an external clock with a phase-lockable
fixed frequency of 50kHz to 650kHz.
The PHASEMODE control pin enables
2-, 3-, 4-, 6-, or 12- phase operation.
The internal LDO regulates to 10V, the same
as the LTC3861-1, optimizing the gate drive
for most automotive and industrial grade
power MOSFETs. But unlike the LTC3861-1,
the undervoltage lockout (UVLO) falling threshold is reduced to 4V from the
original 7V. UVLO shuts off the circuit
when there is not enough gate drive.
Lowering it provides compatibility with
the most efficient 10V gate drive MOSFETs,
while allowing the part to regulate
even when the input voltage dips below
10V (as when an engine is turned on).
The LTC3862-2 also has improved
current sense matching, channel-tochannel and chip-to-chip. Maximum
current sense threshold matching is
reduced from ±10mV to ±7.5mV. This
allows sharing of thermal dissipation more evenly between phases.
3.5A LiFePO 4 CHARGING SOLUTION
WITH ADVANCED TELEMETRY
The LTC4156 is a 3.5A high power, high
efficiency, monolithic charging solution
designed for lithium iron phosphate cells.
In addition to the rich feature set found on
all Linear Technology lithium chargers, the
LTC4156 includes extensive telemetry and
configurability via a 2-wire I2C/SMBus port.
The LTC4156 features four I2C programmable float voltages between 3.45V and
3.8V tailored to the LiFePO4 chemistry. The
input current limit can be programmed
via I2C or resistor to a USB-compliant
value of 100m A, 500m A or 900m A, or
up to 3A for high power wall adapter
sources. A second circuit monitors the
input voltage to the LTC4156 and reduces
input current as necessary to maintain sufficient voltage when connected
through undersized resistive cabling.
Charge current is programmed independently from input current, also
though I2C or a resistor. For deeply
discharged batteries, the LTC4156 supports instant-on operation to provide
3.2V to the system power rail immediately on application of external power.
The LTC4156 autonomously suspends
charging when the cell temperature
is beyond a fixed safety limit. For the
LTC4156, the range has been extended to
0°C–60°C to take advantage of LiFePO4
chemistry’s increased tolerance to extreme
operating conditions. The actual cell temperature is constantly measured by an integrated ADC converter and may be queried
at any time via the I2C interface. Charge
status, input power status, numerous fault
conditions, and many other telemetry
items are also constantly available to the
system, either by polling or via an integrated programmable interrupt system.
Dual input power connectors, overvoltage protection, reverse voltage
protection, and USB On-The-Go are
all supported by the LTC4156 with
minimal external components.
The LTC4156 is fully pin, component
and I2C compatible with the related
LTC4155 Li-ion/Li-polymer charger
to facilitate painless cell chemistry
changes without expensive retooling and requalification. The LTC4156
is available in a 28-lead 4mm × 5mm
QFN surface mount package. n
The LTC2960 is a nano-current high voltage monitor that provides supervisory reset generation and
undervoltage or overvoltage detection. Low quiescent current (0.85μA) and a wide operating voltage
range of 2.5V to 36V make the LTC2960 useful in multicell battery applications.
DEVICE OPTION
OUTPUT TYPE
INPUTS
RESET TIMEOUT PERIOD
LTC2960-1
36V Open Drain
ADJ/IN+
15ms/200ms
LTC2960-2
36V Open Drain
ADJ/IN-
15ms/200ms
LTC2960-3
Active Pull-up
ADJ/IN+
200ms
LTC2960-4
Active Pull-up
ADJ/IN-
200ms
April 2012 : LT Journal of Analog Innovation | 39
highlights from circuits.linear.com
SDM20E-40C
SOLAR-POWERED 2.2V SUPPLY WITH Li-ION
BATTERY BACKUP AND RUN THRESHOLD
SET TO BATTERY MINIMUM VOLTAGE
The LTC3103 is a high efficiency, monolithic
synchronous step-down converter using a current
mode architecture capable of supplying 300mA
of output current. Additionally, the LTC3103
includes an accurate RUN comparator, thermal
overload protection, a power good output and
an integrated soft-start feature to guarantee that
the power system start-up is well controlled.
circuits.linear.com/524
R4
3.09M
+
+
3.6V TADIRAN
AA LITHIUM
BATTERY
4.8V, 0.5W
SOLAR PANEL
MPT4.8-150
(6.5VOC)
CBULK
100µF
BST
VIN
3.2V RUN
THRESHOLD
MODE PGOOD
CIN
10µF
L1
15µH
VOUT
2.2V
SW
RUN
R3
715k
+
CBST
22nF
LTC3103
R2
1.78M
FB
VCC
CFF
22pF
GND
C1
1µF
COUT
22µF
R1
665k
LTspice IV
circuits.linear.com/524
L1: COILCRAFT LP54018
L1
10µH
CBST1
0.1µF
INDUSTRIAL 12V 1MHZ REGULATOR WITH CUSTOM
INPUT UNDERVOLTAGE LOCKOUT THRESHOLDS
The LTC3115-1 is a high voltage monolithic synchronous
buck-boost DC/DC converter. Its wide 2.7V to 40V input and
output voltage ranges make it well suited to a wide variety
of automotive and industrial applications. A proprietary low
noise switching algorithm optimizes efficiency with input
voltages that are above, below or equal to the output voltage,
ensuring seamless transitions between operational modes.
circuits.linear.com/552
BST1 SW1
10V TO
40V
ENABLED WHEN VIN
REACHES 10.6V
DISABLED WHEN VIN
FALLS BELOW 8.7V
CBST2
0.1µF
CIN
10µF
R1
2M
SW2 BST2
PVIN
VIN
PVOUT
LTC3115-1
VC
RUN
R2
255k
RT
35.7k
LTspice IV
circuits.linear.com/552
CO
22µF
CFB
RFB 820pF
40.2k
FB
PWM/SYNC
PVCC
VCC
RT
GND
PGND
12V
1.4A
RTOP
1M
CFF
33pF
RFF
10k
RBOT
90.9k
C1
4.7µF
CIN: MURATA GRM55DR61H106K
CO: TDK CKG57NX5R1H226M
L1: WÜRTH 744065100
±5A, 2.5V (2-QUADRANT) µMODULE VOLTAGE REGULATOR
The LTM8052 is a 36VIN, 5A, 2-quadrant constant-voltage, constant-current
step-down μModule regulator. Included in the package are the switching
controller, power switches, inductor and support components. Operating
over an input voltage range of 6V to 36V, the LTM8052 supports an output
voltage range of 1.2V to 24V. The LTM8052 is able to sink or source current to
maintain voltage regulation up to the positive and negative current limits.
circuits.linear.com/554
LTspice IV
circuits.linear.com/554
VIN*
6V TO 36V
10µF
510k
VIN
RUN
VREF
100µF
SYNC
CTL_I
COMP
CTL_T
GND ADJ
+
SS
OPTIONAL
INPUT
PROTECTION
VOUT
2.5V
±5A
LTM8052
VOUT
RT
90.9k
330µF
9.09k
*INPUT VOLTAGE PROTECTION MAY BE NECESSARY WHEN THE
LTM8052 IS SINKING CURRENT (SEE APPLICATIONS INFORMATION)
L, LT, LTC, LTM, Linear Technology, the Linear logo, LTspice, Burst Mode, PolyPhase, TimerBlox and µModule are registered trademarks, and Hot Swap, LTpowerPlay and PowerPath are trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
© 2012 Linear Technology Corporation/Printed in U.S.A./55.3K
Linear Technology Corporation
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(408) 432-1900
www.linear.com
Cert no. SW-COC-001530