V22N3 - OCTOBER

October 2012
I N
T H I S
I S S U E
1A, low noise buck-boost
converter with 1.8V–5.5V
input voltage range 9
surge stopper with ideal
diode protects input and
output 12
digital power system
management with analog
control loop for ±0.5% VOUT
accuracy 17
LED PWM dimming
simplified 34
Volume 22 Number 3
Real Power Density:
26A µModule Regulator
Keeps Cool in Tight Spaces
Eddie Beville
Each generation of high end processors, FPGAs and ASICs
demands more power, but even as power supplies are expected to
carry significantly heavier loads, they are given less board space to
do so. It is now common for POL (point-of-load) supplies to produce
multiple voltage rails at tens of amps to over a hundred amps at low,
≤1V, output voltages, and in less space than the generation before.
Supplies that must support high load currents and fit in
tight spaces are often judged primarily on their power
density, or watts/cm2. Indeed, many of the latest packaged power supplies and discrete solutions proclaim
impressively high power densities—power supply manufacturers seem to be able to squeeze more and more
power from smaller packages. Unfortunately, there is a
big problem lurking behind amazing increases in power
density. That problem is heat (see sidebar, page 4).
The LTM®4620 µModule® regulator enables high current power supplies to fit tight spaces.
Thermal management is built into the package to prevent hot spots on the board, a
common problem with POL power supplies.
Caption
w w w. li n ea r.com
Heat dissipation is a significant problem at high currents and low voltages. In many systems, cranking up
the power density actually compounds the problem,
because more power in less space also pumps up the
density of power losses—more heat in less space. It is
not enough to simply squeeze a high power supply onto
a board—the solution must also be carefully evaluated
with regard to power loss and thermal resistance—
two parameters that can make or break an otherwise
good regulator. Claims of high power density can be
(continued on page 4)
Linear in the News
In this issue...
COVER STORY
Real Power Density:
26A µModule Regulator
Keeps Cool in Tight Spaces
Eddie Beville
1
DESIGN FEATURES
Multiple Power Sources No Problem for
1A, Low Noise Buck-Boost Converter with
1.8V–5.5V Input Voltage Range
Genesia Bertelle
9
Surge Stopper with Ideal Diode
Protects Input and Output
Zhizhong Hou
DC/DC Controller Combines Digital Power
System Management with Analog Control
Loop for ±0.5% VOUT Accuracy
Hellmuth Witte
12
17
60V, 4-Switch Synchronous Buck-Boost Controller
Regulates Voltage from Wide Ranging Inputs and
Charges Batteries at 98.5% Efficiency at 100W+
Keith Szolusha
22
100V Micropower No-Opto Isolated Flyback
Converter in 5-Lead TSOT-23
Min Chen
27
DESIGN IDEAS
What’s New with LTspice IV?
Gabino Alonso
30
100V Surge Stopper Protects Components from
300V Transients
Hamza Salman Afzal
32
Accurate PWM LED Dimming without External
Signal Generators, Clocks or µControllers
Keith Szolusha
34
Eliminate Opto-Isolators and Isolated Power Supply
from Power over Ethernet Power Sourcing Equipment
Heath Stewart
36
product briefs
38
back page circuits
40
2 | October 2012 : LT Journal of Analog Innovation
LINEAR ON MARS
The recent groundbreaking mission to Mars, launched by NASA’s Jet Propulsion
Laboratory (JPL), was powered in part by Linear Technology products. Linear’s
high performance analog semiconductors are used in the Mars Science Laboratory
(Curiosity) rover to enable collection of vast amounts of data, including detailed
visual images of the Martian landscape and precise readings to assist scientists in assessing the geology and history of Mars. NASA stated that the goal of
the mission is to help determine whether Mars ever had conditions favorable
for supporting life and whether life could have existed on the red planet.
The Linear Technology devices were selected for the Mars program based on
their performance, precision and reliability, as well as their ability to survive
the harsh environment in flight and on the Martian surface. Linear Technology
products are used in the Mars Curiosity rover and in the spacecraft that delivered it to Mars. These include power switching regulators to deliver power to
the rover instruments, analog to digital converters for camera motion control
to allow the rover to “see” the Martian landscape and digitize the images for
their long journey back to earth, and operational amplifiers to amplify signals to
precise levels for accurate delivery of data on the composition of the red planet.
“We are proud to continue our 20-year partnership with the NASA Space program
with the current Mars mission,” stated Linear CEO Lothar Maier. “With over
200 Linear Technology devices in the current Mars expedition, we continue to
provide analog products with the highest performance and reliability, whatever
the operating conditions or application. As the breathtaking images and valuable
data come back from Mars, we are honored to contribute to this historic effort.”
In addition to the numerous Linear Technology devices in the current Mars
Curiosity rover, Linear products were part of the Spirit and Opportunity rovers, which landed on Mars in 2004, as well as Mars Global Surveyor, Mars
Pathfinder, Cassini, Deep Space 1, and Mars Odyssey. Linear Technology provides
NASA/JPL with analog integrated circuits that demonstrate the highest performance, precision, and reliability in extremely small packages. Linear products
were delivered in radiation hardened versions. Linear’s expertise in high precision analog circuits provides enabling technology for the sophisticated scientific
instruments and communications systems in the Mars rover and spacecraft.
Linear in the news
WIRELESS SENSOR NETWORK
PRODUCTS ANNOUNCED
This month Linear is making the first formal announcement of products from the
company’s acquisition of Dust Networks®,
a provider of low power wireless sensor
network (WSN) technology. Dust Networks
pioneered SmartMesh™ networks that
comprise a self-forming mesh of nodes,
or “motes,” which collect and relay data,
and a network manager that monitors
and manages network performance and
sends data to the host application. This
technology is now the basis for a number
of seminal networking standards. The
hallmark of Dust Networks’ technology
is that it combines low power, standardsbased radio technology, time diversity,
frequency diversity and physical diversity—to assure reliability, scalability,
wire-free power source flexibility and
ease-of-use. All motes in a SmartMesh
network—even the routing nodes—are
designed to run on batteries for years,
allowing the ultimate flexibility in placing
sensors exactly where they need to go with
low cost “peel and stick” installations.
Dust Networks’ customers range from
the world’s largest industrial process
automation and control providers such
as GE and Emerson, to innovative, green
companies such as Vigilent and Streetline
Networks. Dust Networks’ technology can
be found in a variety of monitoring and
control solutions, including data center
energy management, renewable energy,
remote monitoring and transportation.
LINEAR TECHNOLOGY DEVICES PROBE MARS
Linear Technology devices used in the Mars Curiosity rover include power switching regulators to deliver
power to the rover instruments, ADCs for camera motion control to allow the rovers to “see” the Martian
landscape and digitize the images for their long journey back to earth, and operational amplifiers to amplify
signals to precise levels for accurate delivery of data on the composition of the red planet.
Joy Weiss, President of Linear Technology’s
Dust Networks product group, stated,
“Our primary goal is to enable our
customers to confidently place sensors
anywhere data needs to be gleaned,
cost effectively and requiring minimum
background in wireless networks. The
advent of SmartMesh systems featuring
Eterna™ and the addition of IP-enabled
wireless sensor networks reflect Linear’s
continued commitment to that goal.”
CONFERENCES & EVENTS
Energy Harvesting & Storage Conference, Hyatt
Regency Crystal City at Ronald Reagan Washington
Electronica 2012, Messe München, Munich, Germany,
November 13-16, 2012, Hall A4, Booth 538—Linear
will exhibit its broad range of analog
products, with particular focus on industrial and automotive applications. More
info at www.electronica.de/en/home
Wireless Congress Systems & Applications,
International Congress Center, Munich, Germany,
November 14–15, 2012—Joy Weiss, President
of Linear’s Dust Networks product
group speaking on “Low Power Wireless
Sensing” in Session 7a, Wireless Sensor
Networks at 11:15 am, November 15.
More info at www.wireless-congress.com
National Airport, Washington, DC, November 7-8,
The Battery Show 2012, Suburban Collection
2012, Booths 4 & 9—Linear will showcase its
Showplace, Novi, Michigan, November 13-15,
Dust Networks wireless sensor network
products as well as energy harvesting
products. Linear’s Joy Weiss will present “Low Power WSN Made Practical”
and Jim Noon will speak on the topic,
“Untapped Potential: Energy Harvesting
Solutions.” More info at www.idtechex.
com/energy-harvesting-usa/eh.asp
2012, Booth B664—Linear will show its
battery stack monitor and power
management products. Presentation
by Mike Kultgen, “The Key Battery
Management Electronics for Maximum
Pack Performance,” Sapphire/Ruby
Ballroom, 3:30 pm, November 14. More
info at www.thebatteryshow.com n
October 2012 : LT Journal of Analog Innovation | 3
Claims of high power density can be impressive, but high power density is
meaningless if the heat produced by the supply is not effectively managed. The
LTM4620 solves the real power density problem by squeezing a complete dual
output regulator in a package uniquely designed to simplify thermal management.
(LTM4620, continued from page 1)
The LTM4620 solves the real power density
problem by squeezing a complete dual
output regulator into a 15mm × 15mm
× 4.41mm LGA package that is uniquely
designed to minimize thermal resistance
and thus simplify thermal management. The package includes an internal
heat sink and other cutting edge features that yield effective top and bottom heat sinking, allowing it to run
4.41mm
15mm
impressive, but the promises made by these
claims are empty if the heat produced by
the supply is not effectively managed.
Figure 1. The LTM4620 LGA package
includes thermal contacts on the top
and bottom that connect to a unique
internal heat sink, which keeps internal
components cool by minimizing the
internal thermal resistance.
at maximum load currents, even in
elevated temperature environments.
Figure 1 shows the LTM4620
15mm × 15mm × 4.41mm LGA package.
15mm
A single device can deliver two independent outputs at 13A (Figure 4) or a
single output at 26A (Figure 5). Multiple
LTM4620s can be combined to produce
from 50A to more than 100A (Figure 7).
The Real Cost of Power Density
PAY ATTENTION TO THE HEAT
Unwanted heat is a major challenge facing designers
of high performance electronics systems. Modern
processors, FPGAs, and custom ASICs dissipate
increasing amounts of power as their temperature
increases. To compensate for these power losses,
power supplies must increase their power output. This,
in turn, increases the power dissipation of the power
supplies, contributing additional heat to an already
hot system, and so on. Unless the heat is evacuated
fast enough, the temperature of the entire system can
elevate to the point where most components must be
derated to compensate.
System and thermal engineers expend significant
time and energy modeling and evaluating complex
electronic systems to remove unwanted heat from the
system. Fans, cold plates, heat sinks and even cooling
bath submersion are all strategies that engineers
have implemented to overcome the heat. Cooling size,
weight, maintenance and cost become a significant
portion of the engineering and manufacturing budget.
As systems add features and performance, the heat
can only rise. Most processors and power supplies run
about as efficiently as they can, and cooling systems
are expensive mitigation. So simplification and cost
savings must be found by improving power dissipation
4 | October 2012 : LT Journal of Analog Innovation
at the component level. The problem is that most
compact packaged power solutions either dissipate
too much power or their thermal resistance is too
high—there is no way to effectively remove enough
heat to operate them at elevated temperature without
significant derating.
POWER DENSITY NUMBERS NOT AS IMPRESSIVE AS
THEY APPEAR
The term high power density DC/DC regulator is
misleading because it does not address the behavior
of the device with respect to temperature. System
designers often look to satisfy a watts/cm2 requirement,
and power supply manufacturers are happy to oblige
with impressive power density numbers. Even so,
hidden in any device’s data sheet are temperaturerelated values that can be more important than the
quoted power density.
For example, consider a 2cm × 1cm DC/DC regulator
that delivers 54W to a load. This calculates to an
impressive rated power density of 27W/cm2. This
number should satisfy the power and size requirements
of some designers. What’s often forgotten, though, is
power dissipation, which translates into rising board
temperature. The key piece of information is listed
in the data sheet as the DC/DC regulator’s thermal
impedance, including the values for the package’s
junction-to-case, junction-to-air and junction-to-PCB
thermal impedances.
To continue with this example, this regulator has
another attractive attribute: it operates at an impressive
efficiency of 90%. Even at such high efficiency, it
dissipates 6W while delivering 54W to the output
in a package with 20ºC/W junction-to-air thermal
impedance. Multiply 6W by 20ºC/W and the result is a
120ºC rise above the ambient temperature. At a 45ºC
ambient temperature, the junction temperature of the
package of this DC/DC regulator rises to 165ºC. This is
far above the typical maximum temperature specified
for most silicon ICs, which is roughly 120ºC. Using this
power supply at its maximum rating would require
extensive cooling to keep the junction temperature at a
value below 120ºC.
Even if a DC/DC regulator addresses all of the electrical
and power requirements of the system, if it fails to meet
the basic thermal guidelines, or proves too costly when
heat-mitigation measures are taken into account, all the
impressive electrical specifications are moot. Evaluating
the thermal performance of a DC/DC regulator can
be as important as judging it on volts, amps and
centimeters.
design features
The internal power MOSFETs are stacked in a proprietary lead
frame to produce high power density, low interconnect resistance,
and high thermal conductivity to both the top and bottom of the
device. Everything is topped off with a proprietary internal heat sink.
EFFECTIVE TOPSIDE HEAT SINKING
POWER INDUCTORS
EFFECTIVE BOTTOM HEAT SINKING
POWER MOSFET STACK
Figure 2. LTM4620 side view rendering and photo of an unmolded LTM4620 showing topside heat sink
UNIQUE PACKAGE DESIGN
ACHIEVES TRUE HIGH POWER
DENSITY
The LTM4620 is designed from the
ground up to produce dual or single
outputs at high power density with
easy-to-manage thermal characteristics.
Unlike other high power density solutions, it is truly self-contained, requiring
no unwieldy heat sinks or liquid cooling to run at maximum load current.
Figure 2 shows a side view rendering
and a top view photo of an unmolded
LTM4620. The package consists of a highly
thermal conductive BT substrate with
adequate copper layers for current carrying capacity and low thermal resistance
to the system board. The internal power
MOSFETs are stacked in a proprietary lead
frame to produce high power density,
low interconnect resistance, and high
LOAD CURRENT (A)
Figure 3. LTM4620 thermal image and derating curve
26
24
22
20
18
16
14
12
10
8
6
4
2
0
400LFM
200LFM
0LFM
0
80
20
40
60
100
AMBIENT TEMPERATURE (°C)
120
See it in Action
Go to video.linear.com/126 to see the impressive
performance of the LTM4620. The videos here
show real lab bench setup and measurement of
short-circuit protection, thermal behavior and
temperature rise at 26A and 100A, heat sink
attachment and precision current sharing at
start-up, steady state and shutdown.
thermal conductivity to both the top
and bottom of the device. Everything is
topped off with a proprietary internal heat
sink that attaches directly to the power
MOSFET stacks and the power inductors for effective topside heat sinking.
The construction of the heat sink and
the mold encapsulation keeps the part
running cool even when thermal management is simply forced air flow across the
top of the package. For a more robust
solution, an external heat sink can be
attached to the topside exposed metal
for even better thermal management.
October 2012 : LT Journal of Analog Innovation | 5
PGOOD1
LTspice IV
TRACK1
circuits.linear.com/586
VIN
CSS1
VIN
VIN
= 100µA
RT
RT
VIN
CIN2
22µF
25V
×2
1µF
GND
TEMP
MTOP1
SW1
CLKOUT
0.33µH
RUN1
MODE_PLLIN
VOUT1
2.2µF
MBOT1
PHASMD
+
GND
VOUT1
1.5V/13A
COUT1
VOUTS1
COMP1
60.4k
VFB1
INTERNAL
COMP
SGND
RFB1
40.2k
POWER
CONTROL
PGOOD2
TRACK2
CSS2
VIN
INTVCC
CIN4
22µF
25V
×2
1µF
4.7µF
GND
EXTVCC
MTOP2
SW2
0.33µH
RUN2
VOUT2
2.2µF
MBOT2
GND
+
VOUT2
1.2V/13A
COUT2
VOUTS2
60.4k
COMP2
fSET
+ –
RFB2
60.4k
INTERNAL
COMP
Rf(SET)
121k
SGND
VFB2
INTERNAL
FILTER
DIFFOUT
Figure 4. Block diagram of the LTM4620 in a dual
output, 1.5V/13A and 1.2V/13A, application
Figure 3 shows a LTM4620 thermal
image and a derating curve for a 12V to
1V at 26A design. The temperature rise
is only 35°C above ambient with no
heat sink and 200LFM of airflow. The
derating curve shows that maximum
load is available out to ~80°C, well
beyond the 65°C that the thermal image
shows for the full-running part.
This result reveals the real merits of a
thermally enhanced high density power
regulator solution. The unique package
design allows the part to not only produce
6 | October 2012 : LT Journal of Analog Innovation
DIFFN
DIFFP
high power in tight spots, but it can do so
without contributing significantly to the
heat problem or requiring derating. Few, if
any other high power density solutions can
make this claim without adding expensive
heat-mitigating components and strategies.
DUAL 13A REGULATOR
Figure 4 shows a simplified block diagram of the LTM4620 µModule regulator
in a dual output design. Its two internal
high performance synchronous buck
regulators produce 1.2V and 1.5V rails,
each with 13A load current capability.
The input voltage range is 4.5V to 16V.
The output voltage range of the LTM4620
is 0.6V to 2.5V, and 0.6V to 5.5V for
the LTM4620A. Total output accuracy
is ±1.5%, with 100% factory-tested
accurate current sharing, fast transient
response, multiphase parallel operation
with self-clocking and programmable
phase shift, frequency synchronization,
and an accurate remote sense amplifier. Protection features include output
design features
INTVCC
0.1µF
VCC
D+
470pF
VREF
LTC2997
D–
GND
VPTAT
µC
1.8V
4mV/K
A/D
4.7µF
MODE_PLLIN
VIN
5V TO 16V
INTERMEDIATE BUS
10k*
CLKOUT
INTVCC PGOOD1
VOUT1
VIN
22µF
25V
×4
TEMP
COUT1
100µF
6.3V
VOUTS1
SW1
RUN1
RUN2
D1*
5.1V ZENER
EXTVCC
5k
TRACK
VFB2
LTM4620
+
COUT2
470µF
6.3V
40.2k
COMP1
TRACK2
COMP2
VOUTS2
fSET
VOUT2
COUT1
100µF
6.3V
SW2
PHASMD
PGOOD2
121k
SGND
COUT2
470µF
6.3V
VFB1
TRACK1
0.1µF
+
GND
DIFFP
DIFFN
VOUT
1.5V AT 26A
DIFFOUT
* PULL-UP RESISTOR AND ZENER ARE OPTIONAL
Figure 5. The two outputs of the LTM4620 can be tied together to produce a 2-phase, 2-parallel-channel, design that yields 1.5V at 26A. Internal diode temperature
monitoring is provided through the LTC2997.
Figure 5 shows a 1.5V at 26A solution that
combines the LTM4620’s two output channels in a parallel 2-phase design. The RUN,
TRACK, COMP, VFB, PGOOD and VOUT pins are
tied together to implement parallel operation. This design also features a LTC®2997
temperature sensor that monitors the
LTM4620’s internal temperature diode.
Figure 6 shows the 1.5V efficiency for
the 2-phase parallel output and the current sharing of the two channels. 86%
efficiency is very good for such a high
density, high step-down ratio solution, and
thermal results are as good or better than
the 1V solution shown in Figure 3. The
Figure 6 shows the well-balanced current
sharing of VOUT1 and VOUT2 . The LTM4620’s
internal controller is accurately trimmed
and tested for output current sharing.
Figure 6. Efficiency and current sharing of 2-phase, single output 26A design shown in Figure 5
90
14
85
12
PER CHANNEL CURRENT (A)
1.5V AT 26A IN 15mm 2 WITH EASY
THERMAL MANAGEMENT
temperature rise is well controlled due to
the low θJA thermal resistance after board
mount. Effective top and bottom heat
sinking enables the LTM4620 to operate
at full power with low temperature rise.
EFFICIENCY (%)
overvoltage protection feedback referred,
foldback overcurrent protection, and
internal temperature diode monitoring.
80
75
70
65
60
VOUT = 1.5V
fSW = 550kHz
0 2 4 6 8 10 12 14 16 18 20 22 24 26
OUTPUT CURRENT (A)
10
8
6
4
2
0
IOUT1
IOUT2
0 2 4 6 8 10 12 14 16 18 20 22 24 26
TOTAL OUTPUT CURRENT (A)
October 2012 : LT Journal of Analog Innovation | 7
The LTM4620 µModule regulator is a true high density
power solution. It differentiates itself in a crowded field of
“high power density” regulators because it manages heat,
the fatal flaw for many proclaimed high density “solutions.”
30
PER µModule CURRENT (A)
25
20
15
10
IOUT(µModule1)
IOUT(µModule2)
IOUT(µModule3)
IOUT(µModule4)
5
0
Figure 7. Four µModule regulators combined in an 8-phase parallel design support 100A
The LTM4620’s current mode architecture yields high efficiency and fast
transient response—top requirements
for low voltage core power supplies
for high performance processors, FPGAs
and custom ASICS. Outstanding initial
output voltage accuracy and the differential remote sensing result in accurate
DC voltage regulation at the load point.
The unique thermal capabilities of the
LTM4620 and its tight current sharing
capabilities make it possible to easily scale
the output above 100A (see Figure 7).
No external clock sources are needed
to set up multiphase operation—the
CLKIN and CLKOUT pins produce internal
programmable phase shifting for paralleled channels. The LTM4620 supports
either external frequency synchronization or internal onboard clocking.
8 | October 2012 : LT Journal of Analog Innovation
REAL POWER DENSITY: 100A IN
UNDER 50mm 2 WITH AIR COOLING
Figure 7 shows four µModule regulators combined in parallel to produce an
8-phase, 100A design. Figure 8 shows
the balanced current sharing for all four
regulators. As shown in Figure 7 the
entire 100A solution only takes about 1.95
square inches of board space. Even at this
high current, a simple heat sink and air
flow can be applied across the top of all
four modules to remove enough power
loss to require no derating. Releasing
heat out of the topside also helps keep
the system board cool to minimize the
heating effect on other components.
0
10 20 30 40 50 60 70 80 90 100
TOTAL OUTPUT CURRENT (A)
Figure 8. Current sharing for the four LTM4620s
combined in a 100A design shown in Figure 7
CONCLUSION
The LTM4620 µModule regulator is a true
high density power solution. It differentiates itself in a the field of high power
density regulators because it manages
heat, the fatal flaw for many proclaimed
high density solutions. It features two high
performance regulators housed inside a
superior thermal package, which makes
possible high power designs that fit into
tight spaces—with minimal external
cooling. Built-in multiphase clocking and
factory-tested accurate current sharing
allow easy scaling of the output current to
25A, 50A, and 100A+. The LTM4620’s unique
thermal properties allow full power operation at elevated ambient temperatures. n
design features
Multiple Power Sources No Problem for 1A, Low Noise
Buck-Boost Converter with 1.8V–5.5V Input Voltage Range
Genesia Bertelle
Users expect their portable devices to operate from a range of power sources including
USB, wall adapters and various types of batteries—alkaline, lithium-ion and LiFePO4.
The LTC®3536 monolithic synchronous buck-boost converter easily accommodates
a variety of power sources by efficiently operating in both buck and boost modes
from an input voltage range of 1.8V to 5.5V. No complicated topology is required
to accommodate power source inputs above, below or equal to the output.
DESIGN VERSATILITY
At 3.3V output, a load current of up
to 1A can be supported over the entire
lithium-ion input voltage range; 300m A of
load current is supported when the input
is 1.8V. The 1% accurate output voltage is programmable from 1.8V to
5.5V via an external resistor divider.
The switching frequency of the LTC3536 is
user programmable from 300kHz to 2MHz
via a single external resistor, allowing
the converter to be optimized to meet the
space and efficiency requirements of each
25
transient response regardless of inductor value and output capacitor size.
COMPETITIVE BUCK-BOOST
fSW = 1.3MHz
15
5
Depending on the application requirements, a designer can prioritize light load
efficiency or minimize supply noise by
choosing from two operating modes: Burst
Mode® operation and PWM operation,
which can be enabled via a dedicated pin.
–5
NOISE (dBm)
The LTC3536 utilizes a proprietary switching algorithm that provides seamless
transitions between buck and boost modes
while simultaneously optimizing efficiency
and minimizing noise over all operating
conditions. This advanced control algorithm uses only a single inductor, which
greatly simplifies the power supply design
and minimizes the total PCB footprint. As
a result, the LTC3536 easily fits lithiumion/polymer, 2-3 cell alkaline/NiMH and
lithium phosphate battery applications,
which often require a supply voltage that
is somewhere in the middle of the battery
voltage range. In such cases, the high efficiency and extended input operating range
of the LTC3536 offer greatly improved
battery run time and design versatility.
–15
–25
–35
–45
–55
–65
–75
LTC3536
fSW = 1MHz
0
0.2
0.4
0.6 0.8 1 1.2
FREQUENCY (MHz)
1.4
1.6
Figure 1. Worst-case spectral comparison of the
LTC3536 and typical competitor’s part. Note the
much lower noise floor exhibited by the LTC3536
and its lower integrated subharmonic noise.
particular application. The default frequency is set to 1.2MHz by tying the RT pin
to VIN. The switching frequency can also be
synchronized to an external clock applied
to the MODE/SYNC pin. In case of synchronization, the free running frequency of the
oscillator can be programmed slower or
faster than the external clock frequency.
External resistors and capacitors provide compensation of the feedback
loop, enabling the frequency response
to be adjusted to suit a wide array
of external components. This flexibility allows for rapid output voltage
Burst Mode operation is an efficient solution for low current conditions. It reduces
the amount of switching to the minimum
level required to support the load, thereby
minimizing the power supply switching
losses. Sometimes noise suppression is
of higher importance and PWM operation, though not as efficient as Burst
Mode operation at light loads, maintains
a steady frequency, making it easier to
reduce noise and RF interference. The
output current capability in Burst Mode
operation is lower than in PWM mode.
Therefore, higher load current applications
require that the MODE/SYNC pin be driven
externally to enter PWM mode operation.
The LTC3536 includes robust VOUT short
circuit protection. If VOUT is shorted to
ground, the inductor current decays very
slowly during a single switching cycle.
During a short-circuit condition, the
LTC3536 reduces its peak current limit
October 2012 : LT Journal of Analog Innovation | 9
The LTC3536 utilizes a proprietary switching algorithm that provides
seamless transitions between buck and boost modes while simultaneously
optimizing efficiency and minimizing noise over all operating conditions.
This advanced control algorithm uses only a single inductor, greatly
simplifying power supply design and minimizing the total PCB footprint.
to a safe level and forces the device to
enter PWM mode, guaranteeing a smooth
recovery when the output short is released.
NOISE PERFORMANCE
Many applications are sensitive to noise
generated by switching converters. The
LTC3536 uses a low noise switching
architecture to reduce unwanted subharmonic frequencies. Subharmonic noise or
jitter is difficult to filter and can interfere
with other sensitive circuitry, and is most
prominent when VIN and VOUT are approximately equal. Some buck-boost converters
operating in this region produce pulsewidth and frequency jitter. The LTC3536
employs Linear Technology’s latest
generation buck-boost PWM modulation
circuitry, which dramatically minimizes
jitter, satisfying the demanding requirements of noise-sensitive RF applications.
Remember, this test is specifically designed
to produce absolute worst-case subharmonic peaks for the LTC3536—if the
input voltage is slightly higher or lower,
the magnitudes of the subharmonics are
much smaller. In contrast, in the competing buck-boost, there is less reduction in
subharmonic content over a much wider
input voltage range. Also, it exhibits a
much higher noise floor than the LTC3536,
indicative of significant pulse-width jitter
and potential noise interference issues.
4.7µH*
Figure 2. Supercapacitor-based backup power supply
VSUPERCAP
1.8V TO 5.5V
10µF
SW1
VIN
SW2
VOUT
RT
VC
49.9k
DC/DC
22µF
845k
0.1µF
R2
*COILCRAFT XFL4020
VH
VL
330pF
GND
100k
circuits.linear.com/587
MAIN POWER
12V
47pF
MODE/SYNC FB
PWM BURST
This feature makes the LTC3536 ideal for
supercapacitor-powered backup power
supply systems, as shown in Figure 2. In
this application, two series of supercapacitors are charged to 5V during normal
operation to provide the needed backup
energy in case the primary power fails.
As long as the primary power is present, the LTC3536 remains in Burst Mode
operation with very low quiescent current,
minimizing drain on the backup storage
capacitor. The MODE pin is used to change
from Burst Mode to PWM mode operation when primary power is interrupted.
300mA FOR VIN ≥ 1.8V
1A FOR VIN ≥ 3V
VSYS
3.3V
6.49k
LTC3536
SHDN
10 | October 2012 : LT Journal of Analog Innovation
The LTC3536 includes a soft-start circuit
to minimize the inrush current transient
during power-up. If at start-up the output
voltage is already pre-charged, the internal
soft-start is skipped and the LTC3536
immediately enters the mode of operation that has been set on the MODE pin. If
the MODE pin is tied high and Burst Mode
operation is selected, the output voltage is
regulated smoothly to the target voltage
value without discharging the output.
The LTC3536 exhibits the expected
single large magnitude tone at its switching frequency of 1MHz, but note that
the total integrated noise is very low
when compared to the competition.
Figure 1 shows worst-case spectral
comparisons of the LTC3536 (switching frequency 1MHz) and a competitive
buck-boost converter (switching frequency 1.3MHz) that does not feature the
low noise architecture of the LTC3536.
LTspice IV
SUPERCAPACITOR BACKUP POWER
The worst-case condition is achieved by
placing a fixed 1A load on the output and
slowly increasing or decreasing the input
voltage until the highest harmonic content
in the converter spectrum was observed.
182k
10pF
6.04k
866k
UV
OV
LTC2912-2
R1
DIS
GND
20k
20k
VCC
TMR
CRT
design features
The LTC3536 maintains an accurate output voltage with
input voltages above or below the output. Its programmable
switching frequency and internal low RDS(ON) power switches, in
combination with low noise architecture, enable the LTC3536 to
offer high performance, compact and highly efficient solutions.
LTspice IV
Figure 3. Solar panel application
Figure 4. Efficiency of the solar panel application in Figure 3
circuits.linear.com/588
100
4.7µH*
98
PHOTOVOLTAIC
CELL
10µF
+
–
221k
60F
SW1
VIN
SW2
VOUT
6.49k
LTC3536
SHDN
MODE/SYNC FB
OFF ON
60F
RT
221k
2.9V
VC
1020k
47pF
49.9k
22µF
LED
220pF
GND
96
EFFICIENCY (%)
1.8V TO 5.5V
158k
10pF
100k
SOLAR-POWERED LED DRIVER
The power generated by a solar cell varies significantly with lighting conditions.
So a rechargeable storage device, such
a supercapacitor, is required to provide
continuous power when the solar cell
is insufficiently illuminated. Although it
has much less charge storage capacity
compared to a battery, a supercapacitor
requires significantly less maintenance,
94
92
90
88
86
ILED = 200mA
ILED = 150mA
ILED = 105mA
ILED = 70mA
84
2Ω
*COILCRAFT XFL4020
LTC3536 in backup mode can provide a
regulated 3.3V at a constant 1A load for
input voltages higher than 3V, and it can
operate down to VIN at 1.8V at a constant
300m A, while maintaining VOUT at 3.3V.
By covering the voltage range of the
supercapacitor, the supply maximizes
operating time, allowing systems enough
time to recover or perform housekeeping tasks before shutting down.
VOUT = 2.9V
is easy to charge and its cycle lifetime is
orders of magnitude longer than a battery.
The storage device can be combined with
the LTC3536 buck-boost DC/DC converter,
which is designed to simplify the task of
harvesting and managing energy from low
input voltage such as photovoltaic cells.
LTC3536 can work down to input voltages
as low as 1.8V, and provides very high
efficiency over a wide range of input voltages above or below the output voltage.
Figure 3 shows an application for an
LED driver supplied with a solar cell for
an emergency LED torch. When the torch
is off, the LTC3536 is in shutdown. The
quiescent current of less than 1µ A minimizes supercapacitor drain when the
ambient light is no longer available.
82
80
1.5
2
2.5
3
3.5
VIN (V)
4
4.5
5
When the LED torch is switched on, the
LTC3536 is turned on via the SHDN pin to
supply 105m A constant load current to
the LED. Figure 4 shows the high efficiency
of this supply, enabling a slow discharge
of the two series 60F capacitors down to
1.8V. The LTC3536 regulates the output
and the LED current, guaranteeing light for
14 minutes if the supercap is charged up
to 5V. It is possible to extend this time by
increasing the supercap value or using a
battery with an adequate charge control.
CONCLUSION
The LTC3536 maintains an accurate
output voltage with input voltages above
or below the output. Its programmable
switching frequency and internal low
RDS(ON) power switches, in combination with low noise architecture, enable
the LTC3536 to offer high performance,
compact and highly efficient solutions. n
October 2012 : LT Journal of Analog Innovation | 11
Surge Stopper with Ideal Diode Protects Input and Output
Zhizhong Hou
Power systems in automobile and industrial applications must cope with short duration
high voltage surges, maintaining regulation at the load, while protecting sensitive
circuitry from dangerous transients. One common protection scheme involves a series
iron core inductor and high value electrolytic bypass capacitor, augmented by a high
power transient voltage suppressor (TVS) and fuse. This heavy-handed approach takes
significant board real estate—the bulky inductor and capacitor are often the tallest
components in the system. Even this protection scheme cannot protect against reverse
input potentials or supply brownouts—possible scenarios in automotive environments.
To protect against these events and maintain the output voltage, designers add a
blocking diode, but the additional voltage drop in the diode increases power losses.
The LTC4364 is a complete control solution
for load protection and output holdup in a
small footprint, eliminating bulky components and undesirable voltage drops.
Figure 1 shows a functional block diagram of the LTC4364. The part drives two
back-to-back N-channel pass transistors:
one protects against voltage surges and
maintains a regulated voltage to the output (M1 in Figure 1), while the other acts
as an ideal diode for reverse input protection and output holdup (M2 in Figure 1).
The LTC4364 also guards against overloads
and short circuits, withstands output
voltage reversal, holds off the MOSFETs in
input undervoltage conditions and inhibits
turn-on or auto-retry in input overvoltage conditions. A shutdown mode reduces
the supply current to as low as 10µ A.
M1
INPUT
LTC4364
VCC
SOURCE
HGATE
10µA
CHARGE
PUMP
RSENSE
M2
12V
DGATE
SENSE
20µA
DA
VA
–
SHDN
UV
1.25V
–
1.25V
–
IA
+
–
+
–
+
+
–
30mV
50mV/
25mV
+
FB
+
TIMER
ENOUT
–
FLT
TMR
12 | October 2012 : LT Journal of Analog Innovation
OUT
12V
+
OV
Figure 1. Simplified block diagram of the LTC4364
OUTPUT
GND
design features
The LTC4364 is a complete control solution for load
protection and output holdup in a small footprint, eliminating
bulky components and undesirable voltage drops.
MAX DC:
100V/–24V VIN
MAX 1ms 12V
TRANSIENT:
200V
M1
FDB33N25
D4
SMAJ24A
D3
1.5KE200A
R1
383k
1%
R2
90.9k
1%
R3
10k
1%
ADVANCED SURGE STOPPER
WITHSTANDS HIGHER VOLTAGES
AND ENSURES SAFE OPERATION
Figure 2 shows a typical application of
the LTC4364. Under normal operating
conditions, the LTC4364 drives the surge
stopper N-channel MOSFET (M1) fully on
and regulates the VDS of the ideal diode
N-channel MOSFET (M2) to 30mV so that
the voltage drop from the input supply
to the load circuitry is minimized. Once
VOUT rises to 0.7V below VIN, the ENOUT pin
goes high to activate the load circuitry.
During an input voltage surge, the LTC4364
regulates the HGATE pin, clamping the
output voltage through MOSFET M1 and a
resistive divider so that the FB pin voltage is maintained at 1.25V. The load
circuit continues to operate, with little
more than a modest increase in supply voltage as illustrated in Figure 3.
In the case of a current overload, the
LTC4364 limits the output current through
M1 so that the voltage across the SENSE and
OUT pins is maintained at 50mV (when
R4
2.2k
0.5W
M2
FDB3682
R5
10Ω
R6
100Ω
C1
0.1µF
D5
1N4148W
D1
CMZ5945B
68V
UV = 6V
OV = 60V
RSNS
10mΩ
+
CHG
0.1µF
VCC HGATE
SHDN
SOURCE DGATE SENSE
UV
OUT
FB
LTC4364
ENOUT
OV
GND
FLT
TMR
VOUT
4A
CLAMPED
AT 27V
COUT
22µF
R7
102k
1%
R8
4.99k
1%
ENABLE
FAULT
CTMR
47nF
OUT > 2.5V). For a severe output short
when OUT is below 1.5V, the current limit
sense voltage folds back to 25mV for additional protection of the MOSFET (Figure 4).
The timer capacitor ramps up whenever
output limiting occurs (either overvoltage as shown in Figure 5 or overcurrent).
If the condition persists long enough
for the TMR pin to reach 1.25V, the
Figure 3. The LTC4364 regulates output at 27V while
load circuit continues to operate in the face of a 92V
input spike.
92V INPUT SURGE
FAULT pin goes low to give early warning
to downstream circuitry of impending
power loss. At 1.35V the timer turns off
the MOSFETs and waits for a cooldown
interval before attempting to restart.
The LTC4364 monitors voltage across
the MOSFET and shortens the turn-off
timer interval in proportion to increasing
VCC – VOUT. In this way a highly stressful
Figure 4. A 2:1 foldback of current limit reduces
MOSFET stress upon severe output short.
60
CTMR = 6.8µF
ILOAD = 0.5A
VIN
20V/DIV
50
ΔVSNS (mV)
Figure 2. Surge stopper with
reverse current protection
withstands 200V/–24V transients
at VIN.
12V
40
30
27V CLAMP (ADJUSTABLE)
VOUT
20V/DIV
20
12V
50ms/DIV
10
0
0.5
1.0
1.5 2.0 2.5
VOUT (V)
3.0
3.5 4.0
October 2012 : LT Journal of Analog Innovation | 13
An important feature of the LTC4364 is that a current limiting device such
as a resistor can be placed between the input supply and the VCC pin. Now
supply transients at the VCC pin can be either filtered with a capacitor or
clamped by a Zener diode. If a proper MOSFET is selected, this scheme
makes it possible to withstand supply transients much higher than 100V.
1.25V
<1.25V
FB
TMR
OV < 1.25V CHECKED
1.35V
1.25V
0.15V
1st
2nd
31st
32nd
FLT
Figure 5. The LTC4364-2 auto-retry timer sequence
following an overvoltage fault provides a very long
cooldown period (0.1% duty cycle).
output short-circuit condition lasts for a
shorter time interval than a brief, minor
overload, helping ensure the MOSFET operates within its safe operating area.
The LTC4364 features a very low restart
duty cycle of about 0.1% in either
overvoltage or overcurrent conditions,
ensuring the MOSFET cools down before
restarting following a turn-off caused by
Figure 6. Input UV and OV monitors can be
configured to block start-up into an overvoltage
condition.
∆VHGATE
COOLDOWN PERIOD
fault. Figure 5 demonstrates the autoretry timer sequence of the LTC4364-2
following an overvoltage fault.
An important feature of the LTC4364 is
that a current limiting device such as a
resistor (R4 in Figure 2) can be placed
between the input supply and the VCC pin.
Now supply transients at the VCC pin can
Figure 7. LTC4364 input protection:
a. Upon an input short or brownout, the DGATE pin
pulls low, shutting down the ideal diode MOSFET
and holding up the output voltage.
VIN
INPUT SHORTED TO GND
0V
475k
UV = 6V
UV
100k
LTC4364
0V = 60V
1nF
b. In reverse input conditions, the DGATE pins pulls
to the SOURCE pin, keeping the ideal diode MOSFET
off and cutting off back feeding.
12V
383k
10nF
be either filtered with a capacitor (C1 in
Figure 2) or clamped by a Zener diode
(D1 in Figure 2). If a proper MOSFET M1 is
selected, this scheme makes it possible to
withstand supply transients much higher
than 100V. The circuit in Figure 2 can
withstand supply transients up to 200V.
OV
VIN
10V/DIV
DGATE
10V/DIV
τUV = (383k||100k) • 10nF
τOV = (475k||10k) •1nF
DGATE PULLS LOW
0V
INPUT FORCED TO –24V
–24V
DGATE
20V/DIV
0V
DGATE PULLS LOW
–24V
12V
OUTPUT HELD UP
CLOAD = 6300µF
ILOAD = 0.5A
14 | October 2012 : LT Journal of Analog Innovation
0V
16.5V
10k
VOUT
10V/DIV
VIN
20V/DIV
1ms/DIV
VOUT
20V/DIV
0V
CLOAD = 6300µF
ILOAD = 0.5A
OUTPUT HELD UP
1ms/DIV
0V
design features
The LTC4364 also guards against overloads and short
circuits, withstands output voltage reversal, holds off the
MOSFETs in input undervoltage conditions and inhibits turnon or auto-retry in input overvoltage conditions. A shutdown
mode reduces the supply current to as low as 10µA.
M1
FDB3632
VIN
12V
UV
6V
OV
60V
Figure 8. LTC4364 offers built-in output
port protection against overvoltage, short
or reverse voltage.
INPUT VOLTAGE MONITORING
PREVENTS UNWANTED TURN-ON
The LTC4364 detects input undervoltage
conditions such as low battery using the
UV pin, and keeps the MOSFETs off if the
UV pin voltage is below 1.25V. The LTC4364
also monitors input overvoltage conditions
and holds off the MOSFETs for start-up or
restart following an output fault condition.
Figure 9. LTC4364 output port protection:
a. When output is forced above input, the DGATE pin
pulls low to cut off back feeding.
VOUT
20V/DIV
24V
12V
CIN
10µF
R1
383k
1%
R2
90.9k
1%
R3
10k
1%
VCC
SHDN
CHG
6.8nF
RSNS
0.2Ω
M2
FDMS86101
R7
49.9k
1%
R5
10Ω
HGATE SOURCE DGATE SENSE
UV
LTC4364
GND
FLT
TMR
0.1µF
At power-up, if the OV pin voltage is
higher than 1.25V before the 100µs
power-on-reset delay expires, or before
the UV pin voltage rises above 1.25V,
the MOSFETs remain off until the OV pin
voltage drops below 1.25V. This feature
allows prevention of start-up when a
board is inserted into an overvoltage
supply by using two separate resistive
b. When output is forced below the GND potential,
the HGATE pin pulls to the SOURCE pin, cutting off
forward conduction and saving battery power at
input.
VOUT
20V/DIV
12V
OUTPUT FORCED TO –12V
–12V
OUTPUT FORCED TO 24V
VIN
20V/DIV
23V
DGATE PULLS LOW
HGATE
20V/DIV
12V
INPUT DISCONNECTED
FROM OUTPUT
1ms/DIV
HGATE PULLS LOW
12V
–12V
VIN
20V/DIV
D2
DDZ9702T
15V
10µF
50V
CER
VOUT*
CLAMPED
AT 18V
RESR
100mΩ
R8
4.99k
1%
ENOUT
OV
24V
DGATE
20V/DIV
OUT
FB
R9
16.9k
1%
10µF
50V
CER
12V
INPUT DISCONNECTED
FROM OUTPUT
1ms/DIV
*PROTECTED AGAINST BACKFEEDING
OR FORWARD CONDUCTING
FROM –20V TO 50V
dividers with appropriate filtering capacitors for the OV and UV pins (Figure 6).
After start-up, under normal conditions, a subsequent input overvoltage
condition does not turn off the MOSFETs,
but rather blocks auto-retry following an output fault. If the OV pin voltage is above 1.25V when the cooldown
timer cycle ends following a fault, the
MOSFETs remain off until the input
overvoltage condition is cleared.
IDEAL DIODE PROTECTS AGAINST
REVERSE INPUT AND BROWNOUT
WITH MINISCULE VOLTAGE DROP
To protect against reverse inputs, a
Schottky blocking diode is often included
in the power path of an electronic system.
This diode not only consumes power
but also reduces the operating voltage
available to the load circuitry, particularly significant with low input voltages,
such as during an automotive cold crank
condition. The LTC4364 eliminates the
conventional Schottky blocking diode and
its voltage and power losses by including
October 2012 : LT Journal of Analog Innovation | 15
The LTC4364 is a compact and complete solution
to limit and regulate voltage and current to protect
sensitive load circuitry against dangerous supply
transients, including those over 100V.
the DGATE pin to drive a second, reverseconnected MOSFET (M2 in Figure 2).
In normal operating conditions, the
LTC4354 regulates the forward voltage
drop (VDS of M2) to only 30mV. If the
load current is large enough to result
in more than a 30mV forward voltage drop, M2 is driven fully on and
its VDS is equal to RDS(ON) • ILOAD.
In the event of an input short or a power
supply failure, reverse current temporarily flows through M2. The LTC4364 detects
the reverse voltage drop and immediately
turns off M2, minimizing discharging of
the output reservoir capacitor and holding up the output voltage. Figure 7a shows
the result of a 12V input supply shorted
to ground. The LTC4364 responds to
this condition by pulling the DGATE pin
low, cutting off the reverse current path
so the output voltage is held up.
In a reverse battery connection, the
LTC4364 shorts the DGATE pin to the
SOURCE pin (that follows the input)
without the need of external components, keeping M2 off and disconnecting the load circuitry from the input
as shown in Figure 7b. The VCC , SHDN,
UV, OV, HGATE, SOURCE and DGATE pins
can all withstand up to 100V above
and 40V below the GND potential.
16 | October 2012 : LT Journal of Analog Innovation
BUILT-IN OUTPUT PORT
PROTECTION
When the output is on a connector as
shown in Figure 8, it could experience
overvoltage, short-circuit or reverse
voltage. The LTC4364 protects the load
circuitry and input supply against those
conditions with several features:
•If the output port is plugged into a
supply that is higher than the input,
the ideal diode MOSFET M2 turns
off to cut the back feeding path
open as shown in Figure 9a.
•If the output port is shorted to
ground, the HGATE pin first regulates the forward current to the
current limit and then turns off
MOSFET M1 if the fault times out.
•If a reverse supply is applied to the
output port, the LTC4364 turns off
the pass MOSFET M1 once the OUT pin
voltage drops below the GND potential, cutting the forward conducting current path open and avoiding
battery drainage at the input.
Figure 9b shows the result when a
–12V supply is applied to the output. The
LTC4364 immediately shorts the HGATE pin
to the SOURCE pin (that follows output),
turning MOSFET M1 off so the input supply
is disconnected from the faulty output.
The OUT and SENSE pins of the LTC4364
can withstand up to 100V above and
20V below the GND potential. For applications where the output port could be
forced below ground, ceramic bypass
capacitors with proper voltage ratings
should be used at the output to stabilize
the voltage and current limiting loops
and to minimize capacitive feedthrough
of input transients (see Figure 8). A
low leakage diode (D2 in Figure 8)
should be used to protect the FB pin.
CONCLUSION
The LTC4364 is a compact and complete
solution to limit and regulate voltage and
current to protect sensitive load circuitry
against dangerous supply transients,
including those over 100V. It is an easy-toimplement, high performance alternative
to the traditionally bulky protection circuits in automotive and industrial systems.
The LTC4364’s integrated ideal diode driver
holds up output voltage during input
short, supply brownout, or reverse input
while cutting the voltage loss associated
with blocking diodes. The built-in output
port protection is useful when the output
is on the connector side. Its feature set is
rounded out by input UV and OV monitoring and a low current shutdown mode. n
design features
DC/DC Controller Combines Digital Power System
Management with Analog Control Loop for
±0.5% VOUT Accuracy
Hellmuth Witte
The LTC3883/-1 is a versatile, single channel, PolyPhase® capable, buck controller
with digital power system management, high performance analog control loop,
on-chip drivers, remote output voltage sensing and inductor temperature sensing.
To minimize solution size and cost, the LTC3883/-1 features Linear’s patent pending
auto-calibration routine to measure the DC resistance of the inductor to obtain
accurate output current measurements when the cycle-by-cycle current is measured
across the inductor (lossless DCR sensing). The LTC3883/-1 is based on the popular
dual channel LTC3880/-1, described in the January 2012 issue of LT Journal.
DIGITAL POWER MANAGEMENT
In today’s data center systems the challenge is to “go green” by becoming as
efficient as possible at all levels of the
system—including the point of load, the
board, rack and even installation levels. For instance, overall system power
consumption can be reduced by routing
the workflow to as few servers as possible,
shutting down servers that are not needed
at the time. The only way to do this and
meet system performance targets (compute
speed, data rate, etc.) is to implement a
comprehensive digital power management system that monitors real-time
power-consumption data at all levels.
In the past, designers have cobbled
together digital power management solutions using a grab bag of ICs including
supervisors, sequencers, DACs and ADCs.
In addition to the inherent complexity of
such solutions, they lack easy expandability, requiring significant up front planning
for future system upgrades. The LTC3883/-1
eliminates this complexity by combining
all digital power management functions
in the DC/DC controller. The result is an
easy-to-use, robust and flexible point-ofload (POL) power management solution.
Figure 1. Digital power system management using the LTC3883.
The LTC3883/-1 can operate autonomously
or communicate via an industry-standard
I2C serial bus with a system host processor for command, control and to report
telemetry. This makes it possible to
monitor critical operating information
directly from the LTC3883/-1 such as realtime voltages, currents and temperatures,
which can used to dynamically optimize
system performance and reliability.
Access to this data makes it possible to
predict power system failures and take
preventive or mediation measures.
Important regulator parameters—such
as output voltages and current limits,
margining voltages, overvoltage and
undervoltage supervisory limits, startup characteristics and timing and fault
responses—can also all be directly
October 2012 : LT Journal of Analog Innovation | 17
The LTC3883/-1 can operate autonomously or communicate
via an industry-standard I2C serial bus with a system host
processor for command, control and to report telemetry.
programmed via the serial bus, instead
of using external components such as
resistors, sequencers, and monitoring ICs.
Digital power system management
makes it possible to quickly and efficiently develop complex multirail systems. Design is further simplified by
the LTpowerPlay™ software, which
enables PC-based board monitoring and
parametric adjustments. This allows a
designer to debug and perform in-circuit
testing (ICT) without any circuit board
rewiring or component modification.
5mΩ
VIN
6V TO 24V
10µF
100Ω
1µF
100Ω
The LTC3883/-1 features optimized gate
driver dead times to minimize switching losses and body diode conduction, thus maintaining high efficiency
in all operating conditions. It supports
a wide VIN range of 4.5V to 24V and a
VOUT range of 0.5V to 5.5V. The precision
reference, 12-bit DAC, and temperaturecompensated analog current mode
control loop yield ±0.5% DC output
voltage accuracy, and an integrated high
side input current sense amplifier allows
for accurate input current sensing and
auto-calibration of the inductor DCR.
18 | October 2012 : LT Journal of Analog Innovation
TG
LTC3883
BOOST
IIN_SNS
10nF
3Ω
10nF
VIN
10k
10k
PMBus
INTERFACE
10k
10k
10k
10k
10k
5k
VDD33
PGOOD
SDA
RUN
1.4k
1µF
VDD25
20k
24.9k
10k
20k
12.7k
9.09k
23.2k
17.8k
VOUT_CFG
1.4k
VTRIM_CFG
0.22µF
SHARE_CLK ASEL
GPIO
SYNC
WP
VDD25
VDD33
1.0µF
M2
FREQ_CFG
SCL
ALERT
M1
0.56µH
BG
PGND
10µF
0.1µF
22µF
50V
1µF
SW
VIN_SNS
FEATURE OVERVIEW
The LTC3883/-1 is a single output synchronous step-down DC/DC controller
with integrated power FET gate drivers
and an analog current mode control loop
that is PolyPhase capable to six phases.
The frequency can be set from 250kHz
to 1MHz, and if an external oscillator is
available, the internal phase-locked loop
enables the LTC3883/-1 to synchronize with
any frequencies within the same range.
D1
INTVCC
ISENSE+
ISENSE–
VSENSE+
VSENSE–
+
TSNS
GND
ITH
1.0µF
2200pF
100pF
VOUT
1.8V
20A
COUT
530µF
MMBT3906
4.99k
D1: CENTRAL CMDSH-3TR
M1: INFINEON BSC050N03LSG COUT: 330μH SANYO 4TPF330ML,
2× 100µF AVX 12106D107KAT2A
L: VISHAY IHLP-4040DZ-11 0.56µH M2: INFINEON BSC011N03LSI
Figure 2. High efficiency 500kHz 1.8V step-down converter with DCR sense
A 16-bit data acquisition system provides
digital read back of input and output
voltages and currents, duty cycle and
temperature. Peak values of important
measurements can be read back by the
user. Critical controller parameters can be
programmed via the PMBus. Fault logging capability includes an interrupt flag
and a black box recorder in nonvolatile
memory that stores the operating conditions immediately prior to a fault.
The hallmark of the LTC3883 is an on-chip
LDO regulator for increased integration,
whereas the LTC3883-1 is powered from
an external 5V bias voltage for increased
efficiency. Both parts are available in a
thermally enhanced 32-lead 5mm × 5mm
QFN package with either a –40°C to 105°C
operating junction temperature range
(E-grade) or a –40°C to 125°C operating
junction temperature range (I-grade).
design features
High performance PMBus controllers, such as the LTC3883/-1, and companion ICs,
such as the LTC2978, work together efficiently and seamlessly to meet the strict
digital power management requirements for today’s sophisticated circuit boards.
PMBus CONTROL
The LTC3883/-1 PMBus interface allows
digital programming of critical power
supply parameters as well as the read
back of important real-time conditions.
Parameter configurations can be downloaded to the internal EEPROM using Linear
Technology’s LTpowerPlay development
software. Figure 5 shows the PC-based
LTpowerPlay development platform
with a USB to I2C/SMBus/PMBus adapter.
Once a part is configured as desired,
it will operate autonomously without
control from the host, so no further
firmware or microcontroller is required.
•Overvoltage and undervoltage high
speed supervisor thresholds
•System status
•Output voltage on/off time delays
•Peak output voltage
•Output voltage rise/fall times
•Peak internal/external temperature
•Input voltage on/off thresholds
•Fault log status
•Output rail on/off
•Output rail margin high/margin low
•Responses to internal/external faults
•Fault propagation
The PMBus allow the user to monitor
the following power supply conditions:
•Output/input voltage
The PMBus enables programming of the
following power supply parameters:
•Output/input current
•Output voltage and margin voltages
•External inductor temperature
•Temperature-compensated current limit
threshold based on inductor temperature
•Part status
•Internal die temperature
•Fault status
•Switching frequency
Figure 3. Complete development platform
with LTpowerPlay software
USB
DC1613A
USB to PMBus
Controller
DC1890
Socketed
Programming Board
DC1778A
LTC3883
Demo Board
Demonstration
Kit
or
Socketed
Programming
Customer Board
with
LTC3883/LTC3883-1
or
•Peak output current
ANALOG CONTROL LOOP
The LTC3883/-1 is digitally programmable for numerous functions including the output voltage, current limit
set point and sequencing. The control
loop, though, remains purely analog,
offering optimum loop stability and
transient response without the quantization effect of a digital control loop.
Figure 4 compares the ramp curves of
a controller IC with an analog feedback
control loop to one with a digital feedback control loop. The analog loop has
a smooth ramp, whereas the digital loop
has discrete steps that can result in stability problems, slower transient response,
more required output capacitance in
some applications and higher output
ripple and jitter on the PWM control
signals due to quantization effects.
The current mode control loop produces
the best loop stability, cycle-by-cycle
current limit, and fast and accurate
responses to line and load transients.
The simple loop compensation is independent of operating conditions and
converter configuration. Continuous,
discontinuous, and Burst Mode® inductor current control are all supported.
In-Circuit Serial
Programming
October 2012 : LT Journal of Analog Innovation | 19
Digital power system management makes it possible to quickly and efficiently develop
complex multirail systems. Design is further simplified by LTpowerPlay software,
which enables PC-based board monitoring and parametric adjustments.
AUTO-CALIBRATION OF
INDUCTOR DCR
Using the DC resistance of the inductor
instead of a sense resistor to sense the
output current of a DC/DC converter has
several advantages, including reduced
power loss, and lower circuit complexity and cost. However, any difference
between the specified nominal inductor
DCR and the actual inductor DCR causes a
proportional error in the measured output
current, and in the peak current limit.
ANALOG CONTROL LOOP
ANALOG
CURRENT
WAVEFORM
FIXED fSW
DIGITAL CONTROL LOOP
DIGITAL
RAMP
FIXED fSW
The tolerance of the inductor DCR from
its nominal value can be measured and
compensated for by the LTC3883/-1
using Linear’s patent pending algorithm.
Just complete a simple 180ms calibration procedure via PMBus command
while the converter is in a steady state
condition with a large enough load
current to allow for accurate input
and output current measurements.
The inductor temperature is accurately
measured by the LTC3883/-1 to maintain
an accurate current read back over the
entire operating temperature range. The
LTC3883 dynamically models the temperature rise from the external temperature
sensor to the inductor core to account
for the self-heating effect of the inductor.
This patent pending algorithm simplifies
the placement requirements of the external temperature sensor and compensates
LTpowerPlay
LTpowerPlay software is available for free at
www.llinear.com/ltpowerplay
20 | October 2012 : LT Journal of Analog Innovation
Figure 4. The LTC3883’s analog control loop vs a
digital control loop. The analog loop has a smooth
ramp, whereas the digital loop has discrete steps
that can result in stability problems, slower transient
response, more required output capacitance in
some applications and higher output ripple and jitter
on the PWM control signals due to quantization
effects.
for the significant steady state and transient temperature error from the inductor core to the primary heat sink.
MULTIPLE IC SYSTEMS
Large multirail power boards are normally
comprised of an isolated intermediate
bus converter, which converts –48V from
the backplane to a lower intermediate
bus voltage (IBV), typically 12V, which is
distributed around a PC card. Individual
point-of-load (POL) DC-DC converters step
down the IBV to the required rail voltages,
which normally range from 0.5V to 5V with
output currents ranging anywhere from
0.5A to 120A. These boards are densely
populated and the digital power system
management circuitry cannot afford to
take up much PC board real estate.
High performance LTC PMBus controllers,
such as the LTC3883/-1, and companion
ICs, such as the LTC2978, work together
efficiently and seamlessly to meet the
strict digital power management requirements for today’s sophisticated circuit
boards. These include sequencing, voltage
accuracy, overcurrent and overvoltage
limits, margining, supervision and fault
control. Any combination of these devices
makes sequencing design an easy process
for any number of supplies. By using a
time-based algorithm, users can dynamically sequence rails on and off in any
order with a simple programmable delay.
Sequencing across multiple chips is made
possible using the 1-wire SHARE_CLK bus
and one or more of the bidirectional
general purpose IO (GPIO) pins.
LTPOWERPLAY SOFTWARE
LTpowerPlay software makes it easy to
control and monitor multiple Linear
PMBus-enabled devices simultaneously.
Modify the DC/DC controller configuration in real time by downloading system
parameters to the internal EEPROM of the
LTC3883/-1. This reduces design development time by allowing system configurations to be adjusted in software rather
than resorting to the draconian tasks
of swapping components and manually
rewiring boards. Figure 5 shows how
output voltage, OV/UV protection limits,
and on/off ramps are controlled. The
waveform displays the soft-start and
soft-stop of the output voltage. Warning
and fault conditions are also shown.
design features
LTpowerPlay software makes it easy to control and monitor multiple Linear PMBus
enabled devices simultaneously. Modify the DC/DC controller configuration in real
time by downloading system parameters to the LTC3883/-1 internal EEPROM.
Figure 5. Power systems simplified. LTpowerPlay puts complete power supply control at your fingertips.
CONCLUSION
The LTC3883/-1 combines high performance analog switching regulation
with precision data conversion and
a flexible digital interface. Multiple
LTC3883s can be used in parallel with
other devices to easily create optimized
multiple-rail digital power systems.
All Linear PMBus products are supported by the LTpowerPlay software
development system, which helps
board designers quickly debug systems.
LTpowerPlay can be used to monitor, control and adjust supply voltages,
limits and sequencing. Production
margin testing is easily performed
using a couple of standard PMBus commands. Combining the LTC3883/-1 with
other Linear PMBus products is the best
way to quickly bring to market digitally controlled power supplies. n
October 2012 : LT Journal of Analog Innovation | 21
60V, 4-Switch Synchronous Buck-Boost Controller Regulates
Voltage from Wide Ranging Inputs and Charges Batteries at
98.5% Efficiency at 100W+
Keith Szolusha
The LT®3791-1 is a 4-switch synchronous buck-boost DC/DC converter that regulates
both constant voltage and constant current at up to 98.5% efficiency using only a
single inductor. It can deliver well over a hundred watts and features a 60V input
and output rating, making it an ideal DC/DC voltage regulator and battery charger
when both step-up and step-down conversion is needed. In addition to the high
voltage, power and efficiency, it features short-circuit protection, a SYNC pin for
synchronization to an external clock, a CLKOUT pin for driving an external SYNC
pin or for parallel operation, OVLO (overvoltage lockout), SHORT output flag, C/10
detection and output flag for battery chargers, and a CCM pin for discontinuous or
continuous conduction mode. The inclusion of DCM (discontinuous conduction mode)
increases light load efficiency and prevents reverse current when it is undesirable.
120W, 24V 5A OUTPUT BUCK-BOOST
VOLTAGE REGULATOR
operates from an input voltage range
of 12V to 58V. Adjustable undervoltage and overvoltage lockout protect
the circuit. It has short-circuit protection and the SHORT output flag indicates
The buck-boost converter shown in
Figure 1 regulates 24V with 0A–5A load
at up to 98.5% efficiency (Figure 2). It
VIN
12V TO
58V
0.003Ω
VIN
1µF
51Ω
499k
INTVCC
0.1µF
IVINP
BST1
499k
TG1
EN/UVLO
27.4k
100k
18.7k
BG1
INTVCC
LT3791-1
200k
0.1µF
VOUT
24V
5A
715k
M3
13.7k
BG2
VC
1000pF
RT
15k
10nF
22 | October 2012 : LT Journal of Analog Innovation
SW2
TG2
ISP
ISN
FB
SGND
147k
200kHz
LTspice IV
38.3k
SNSN
PGND
SS SYNC
33nF
L1
10µH
M4
COUT
220µF
35V ROUT
×2 7.5mΩ
4.7µF
50V
×2
Figure 1. 120W 24V 5A output buckboost voltage regulator accepts a
12V–58V input
C1
47µF
80V
0.004Ω
PWM
100k
M2
0.1µF
+
SNSP
SHORT
C/10
CCM
IVINMON
ISMON
CLKOUT
CTRL
VREF
33nF
M1
SWI
OVLO
56.2k
+
BST2
IVINN
4.7µF
100V
4.7µF
D1 D2
470nF
10k
when there is a short circuit on the
output. It features DCM operation at
light load for lowest power consumption and reverse current protection.
ROUT limits the output current during both
D1, D2: NXP BAT46WJ
L1: COILCRAFT SER2915L-103KL 10µH
M1, M2: RENESAS RJK0651DPB 60Vds
M3, M4: RENESAS RJK0451DPB 40Vds
C1: NIPPON CHEMICON EMZA800ADA470MJAOG
COUT: SUNCON 35HVT220M 35V 220µF ×2
circuits.linear.com/589
design features
The LT3791-1 can regulate both constant voltage and constant current. Large
capacitive loads such as supercapacitors and batteries require constant current
charging until they are charged up to a termination voltage, at which point they
require constant voltage regulation. The LT3791-1 easily satisfies this requirement.
100
VIN = 14V
IIN = 8.87A
VOUT = 24V
IOUT = 5A
95
EFFICIENCY (%)
90
85
80
75
70
VIN = 12V
VIN = 24V
VIN = 54V
65
Figure 2. Efficiency and worst
case thermal results for the 24V
converter in Figure 1
60
0
1
2
3
LOAD CURRENT (A)
4
5
a short circuit and an overload situation, making this a robust application.
The 14V, 10A voltage regulator in Figure 3
takes a slightly different approach. It runs
in CCM throughout its entire load current range 0A-10A to provide the lowest
EMI at light load. It is still very efficient.
The circuit retains short-circuit protection
even though ROUT is replaced with a short.
The main switch sense resistor RSW limits
the short-circuit current at a higher level
than ROUT, but hiccup mode during shortcircuit limits the power consumption of
the IC, maintaining a low temperature
rise on the components during a short.
When DCM is not needed, ROUT may not
be necessary and removing ROUT slightly
increases circuit efficiency. OVLO is tied
to the output to limit the output voltage
transient during a 10A to 0A transition.
This protects both the output capacitors
and M3 and M4 switches from overvoltage.
VIN
9V TO
36V
0.002Ω
470nF
499k
IVINP
EN/UVLO
51Ω
CIN1
4.7µF
50V
1µF
+
IVINN VIN INTVCC
CCM
76.8k
CIN2
100µF
63V
×2
4.7µF
10V
D1 D2
BST2
0.1µF
BST1
C/10
IVINMON
CLKOUT
ISMON
TG1
BG1
INTVCC
LT3791-1
200k
SHORT
100k
22nF
VC
5.1k
M3
COUT1
4.7µF
50V
×2 +
COUT2
270µF
35V
×2
VOUT
14V
10A
100k
9.31k
BG2
CTRL
SS
10nF
Figure 3. 140W (14V, 10A) CCM buck-boost voltage regulator with
9V–56V input has output OVLO for transient protection.
SNSN
PGND
SYNC
100pF
L1
3.3µH
0.0025Ω
PWM
M5
M2
M4
SNSP
SHORT
VREF
0.1µF
M1
SWI
0.1µF
RT
SGND
147k
200kHz
SW2
TG2
FB
ISP
ISN
OVLO
499k
D1, D2: NXP BAT46WJ
L1: COILCRAFT SER2915H-332L 3.3µH 48A
M1: RENESAS RJK0652DPB 60Vds
M2: RENESAS RJK0651DPB 60Vds
M3, M4: INFINEON BSC0904NSI 30Vds
M5: NXP NX7002AK
COUT2: SUNCON 35HVT270M
CIN2: NIPPON CHEMICON EMZA630ADA101MJAOG
88.7k
October 2012 : LT Journal of Analog Innovation | 23
0.003Ω
VIN
12V TO
58V
499k
51Ω
470nF
499k
IVINP
EN/UVLO
1µF
27.4k
CCM
C/10
BST1
M1
TG1
SWI
SHORT
VREF
M2
BG1
LT3791-1
PWM
0.1µF
Figure 4. Parallel LT3791-1s in a 240W application
L1
10µH
4.7µF
50V
×2
0.015Ω
VOUT
24V
10A
M4
51Ω
M3
+
COUT1
220µF
35V
×2
0.47µF
SNSP
CTRL
0.004Ω
100k
SNSN
PGND
SS
33nF
4.7µF
10V
D1 D2
0.1µF
200k
0.1µF
C1
47µF
80V
BST2
INTVCC1
SHORT
+
INTVCC1
IVINN VIN INTVCC
OVLO
56.2k
4.7µF
100V
BG2
IVINMON
CLKOUT
ISMON
SYNC
SW2
VC
SGND
RT
715k
TG2
ISP
ISN
FB
13.7k
38.3k
147k
200kHz
3.3k
33nF
INTVCC1
+
LTC6240
–
45k
10k
10k
0.003Ω
VIN
499k
470nF
499k
10k
IVINP
EN/UVLO
51Ω
1µF
IVINN VIN INTVCC
27.4k
0.1µF
C/10
BST1
TG1
200k
SHORT
0.1µF
PWM
BG1
LT3791-1
M6
0.1µF
L2
10µH
4.7µF
50V
×2
0.015Ω
M8
M7
51Ω
+
0.22µF
SNSP
0.004Ω
100k
33nF
M5
SWI
SHORT
VREF
C2
47µF
80V
4.7µF
10V
D3 D4
BST2
INTVCC2
45k
+
INTVCC2
CCM
OVLO
56.2k
4.7µF
100V
SNSN
PGND
SS
BG2
CTRL
IVINMON
ISMON
CLKOUT
SYNC
VC
10k
1000pF
2.2k
RT
SW2
TG2
ISP
ISN
FB
SGND
147k
200kHz
22nF
24 | October 2012 : LT Journal of Analog Innovation
715k
13.7k
38.3k
D1–D4: NXP BAT46WJ
L1, L2: COILCRAFT SER2915L-103KL 10µH
M1, M2, M5, M6: RENESAS RJK0651DPB 60Vds
M3, M4, M7, M8: RENESAS RJK0451DPB 40Vds
COUT1, COUT2: SUNCON 35HVT220M ×2
C1, C2: NIPPON CHEMICON EMZA800ADA470MJAOG
COUT2
220µF
35V
×2
PARALLEL CONVERTERS FOR HIGH
POWER USING CLKOUT AND SYNC
The LT3791-1 has a CLKOUT output that
can be used to synchronize other converters to its own clock with a 180°
phase shift. By tying the CLKOUT of one
converter to the SYNC input of another,
the maximum output power is doubled
while the output ripple is reduced.
Figure 4 shows a 24V, 10A regulator
formed by running two LT3791-1s in parallel. By using two parallel circuits, the maximum temperature rise seen on any one
discrete component is only 28°C on the M3
and M7 MOSFETs at the lowest VIN . The top
converter (master) in Figure 4 commands
the current level provided by the bottom
(slave) converter. The ISMON output of the
master indicates how much current the
master is providing, and by connecting
ISMON to the CTRL input of the slave, the
slave is forced to follow the master. A single op amp is needed to provide the simple
200mV level shift needed to match the
CTRL input to the ISMON output levels. The
master converter runs in constant voltage regulation while the slave converter
is running in constant current regulation.
Note that the output voltage of the slave is
set slightly higher (28V) so that the voltage
feedback loop of the slave is not in regulation for it to be able to follow the master.
design features
The LT3791-1 features both continuous conduction mode (CCM) and
discontinuous conduction mode (DCM). CCM provides continuous switching
at light load and inductor current can be either positive or negative. When the
LT3791-1 enters DCM operation at light load, it prevents backward running current
(negative inductor current) and light load power dissipation is minimized.
100W+ 2.5A BUCK-BOOST 36V SLA
BATTERY CHARGER
In some battery charger applications,
once termination voltage is reached and
charge current tails off, a standby or
float voltage regulation level is needed
that is different from the charge voltage.
The C/10 detection level of the LT3791-1
provides this capability. In the circuit
in Figure 3 the C/10 function drops the
battery voltage from charging (44V)
to float (41V) when the battery is near
full charge. When the battery voltage
is then pulled down from an increased
load, the voltage feedback loop returns
the charger to its charge state of 44V.
The LT3791-1 can regulate both constant
voltage and constant current. Large
capacitive loads such as supercapacitors
and batteries require constant current
charging until they are charged up to a
termination voltage, at which point they
require constant voltage regulation. The
LT3791-1 easily satisfies this requirement.
As an example, the buck-boost converter
shown in Figure 5 charges a 36V 12Ah
SLA battery at 44V with 2.5A DC from a
9V-to-58V input. DCM operation prevents
reverse battery current when the output load is overcharged, protecting the
circuit from large negative currents.
Figure 5. SLA battery charger
regardless of the voltage relationship
between them. Furthermore, a microcontroller can be used to create a maximum
power point tracking (MPPT) charger
from a solar panel. The output diagnostics ISMON and IVINMON and current
control pin CTRL make it easy to create a
high power solar panel battery charger.
DCM INCREASES EFFICIENCY AND
PREVENTS REVERSE CURRENT
The LT3791-1 features both continuous
conduction mode (CCM) and discontinuous
conduction mode (DCM). Figure 6 shows
the difference between CCM and DCM. The
mode is selected by simply connecting
the CCM pin to either the INTVCC or C/10
pin. CCM provides continuous switching
The LT3791-1 can be tailored to charge
a range of battery chemistries and
capacities from a variety of input sources
PVIN
9V TO 57V
RIN
0.003Ω
1µF
50Ω
VIN
INTVCC
D1
IVINN
TG1
BG1
OVLO
LT3791-1
19.6k
M1
0.1µF
M4
M2
L1
10µH
M3
SWI
EN/UVLO
RBAT
0.04Ω
SNSN
SHORT
+
PGND
IVINMON
ISMON
CTRL
BG2
SW2
TG2
ISP
CLKOUT
SGND
100k
VREF
PWM
ISN
FB
CCM
SS
C/10
RT
SYNC VC
D1, D2: BAT46WJ
33nF
L1: COILCRAFT SER2915L-103K
M1-M4: RENESAS RJK0651DPB
M5: NXP NX7002AK
CIN2: ×2 NIPPON CHEMI-CON EMZA630ADA101MJA0G
COUT2: ×3 NIPPON CHEMI-CON EMZA630ADA101MJA0G
CIN2
100µF
63V
×2
COUT1
4.7µF
50V
×2
RSENSE
0.004Ω
200k
0.1µF
+
+
COUT2
100µF
63V
×3
SNSP
INTVCC
CHARGE CURRENT CONTROL
CIN1
4.7µF
100V
×2
0.1µF
BST1
IVINP
24.3k
D2
BST2
470nF
332k
4.7µF
3k
84.5k
300kHz
0.1µF
36V
SLA BATTERY
AGM TYPE
41V FLOAT
44V CHARGE
AT 25°C
50Ω
1.00M
INTVCC
10k
0.47µF
2.5A
CHARGE
10k
402k
30.1k
M5
22nF
October 2012 : LT Journal of Analog Innovation | 25
5000
90
1000
CCM RISING THRESHOLD
600
EFFICIENCY (%)
ILOAD (mA)
FOR CCM OPERATION
OVER ALL IOUT
DCM FALLING THRESHOLD
400
INTVCC
LT3791-1
LT3791-1
100k
CCM
CCM
CCCM
OPTIONAL
0
DCM (TG2 FOR M4 STAYS LOW)
18
12
24
30
36
VIN (V)
42
48
at light load and inductor current can
be either positive or negative. Although
zero-load inductor current in CCM is both
positive and negative and more power
is consumed than DCM, the switch node
ringing associated with DCM is eliminated for those that do not want it.
DCM operation at light load, the TG2 driver
for M4 stays low and M4 no longer runs
as a switch, but instead as a catch diode.
This prevents backward running current (negative inductor current) and light
load power dissipation is minimized.
When DCM is selected, the converter
remains in CCM until the load drops below
about 10% of the programmed maximum
output current. When the LT3791-1 enters
The LT3791-1 synchronous buck-boost
controller delivers over 100W at up to
98.5% efficiency to a variety of loads. Its
wide, 4.7V to 60V input range and 0V to
1.2
1
40
0.8
30
INTVCC
IVINN
470nF
0.01
0.1
IOUT (A)
1
TG1
EN/UVLO
OVLO
0.1µF
18.7k
100k
BG1
LT3791-1
200k
0.0125Ω
M2
L1
22µH
M4
M3
COUT1
4.7µF
100V
×2 +
COUT2
330µF
63V
×3
Figure 7. 48V application
VOUT
48V
3A
0.005Ω
SNSN
PGND
BG2
SW2
TG2
ISP
ISN
FB
SGND
PWM
10k
CIN2
100µF
80V
×2
SNSP
SHORT
C/10
CCM
IVINMON
ISMON
CLKOUT
VREF
0.1µF
M1
SWI
INTVCC
100k
+
BST1
0.1µF
CTRL SS SYNC V
C
RT
40k
1µF
22nF
26 | October 2012 : LT Journal of Analog Innovation
105k
250kHz
1000pF
0.4
0.2
10
0
60V output range make it powerful and
versatile, and its built-in short-circuit
capabilities make for robust solutions
in potentially hazardous environments.
CCM and DCM operation make it useful for highest efficiency or lowest noise
operation at light load. Its multiple control
loops make it ideal for regulating constant
voltage, constant current or both. This
feature-rich IC easily fulfills buck-boost
requirements where other topologies fail. n
0.1µF
IVINP
499k
28.7k
4.7µF
10V
D1 D2
BST2
CIN1
4.7µF
100V
0.6
DCM
DCM
CCM
CCM
c. DCM improves efficiency at light loads.
CONCLUSION
VIN
1.4
50
0.004Ω
1µF
51Ω
34.8k
60
1.6
POWER
LOSS
EFFICIENCY
0
0.001
54
b. DCM/CCM transition thresholds remain stable as
the LT3791-1 moves through boost, buck-boost and
buck modes of operation.
499k
70
10
a. DCM vs CCM setup
VIN
18V TO 55V
1.8
20
200
INTVCC
C/10
2
VIN = 24V
VOUT = 24V
80
800
FOR DCM OPERATION
AT IOUT < 10mV/ROUT
100
CCM
POWER LOSS (W)
Figure 6. Overview of continuous conduction
mode (CCM) for low noise and discontinuous
conduction mode (DCM) for light load efficiency
95.3k
2.15k
D1, D2: NXP BAT46WJ
L1: WÜRTH ELECTRONICS 74435572200 22µH 11A
M1, M2, M4: RENESAS RJK0651DPB 60Vds
M3: RENESAS RJK0652DPB 60Vds
COUT2: YAGEO ST 330µF 63V ×3
CIN2: NIPPON CHEMICON EMZA630ADA101MJAOG 100µF 63V ×2
2.49k
design features
100V Micropower No-Opto Isolated Flyback Converter in
5-Lead TSOT-23
Min Chen
The non-synchronous flyback topology is widely used
in isolated power supplies ranging from sub-watt power
levels to tens of watts. Linear’s no-opto isolated flyback
family dramatically simplifies isolated power supply design
with proprietary primary-side sensing, which requires
no opto-coupler or transformer third winding for output
regulation. The new LT8300, the first micropower part in
this family, significantly improves light load efficiency and
reduces no-load input standby current to about 200µA.
The LT8300 operates from an input
voltage range of 6V to 100V and delivers up to 2W of isolated output power.
The 150V integrated DMOS power switch
eliminates the need for a snubber in most
applications. By sampling the isolated
output voltage directly from the primaryside flyback waveform, the LT8300 requires
no opto-coupler or transformer third
winding for regulation. The output voltage is set with a single external resistor.
Internal loop compensation and soft start
further reduce external component count.
Boundary mode operation at heavy load
enables the use of small magnetics and
produces excellent load regulation. Low
ripple Burst Mode operation maintains
high efficiency at light load while minimizing output voltage ripple. All these features
are packed in a 5-lead TSOT-23 package
(Figure 1) with high voltage pin spacing
conforming to IPC-2221 requirement.
Figure 2. The LT8300 isolated
flyback converter solution size
is less than 1 inch by ½ inch in a
standard demo board DC1825A.
Figure 1. The LT8300 is available in a 5-lead TSOT-23
package with high voltage pin spacing between pins
4 and 5.
PERFORMANCE AND SIMPLICITY
A complete isolated flyback solution
fits into an area less than 1 by ½ inch,
as shown in Figure 2. Figure 3 shows a
typical LT8300 application, generating a
5V isolated output from a 36V-to-72V input.
The solution only requires five external
components (input capacitor, output
capacitor, transformer, feedback resistor and output diode) and two optional
undervoltage lockout resistors.
Although the LT8300 simplifies isolated
flyback converter design, it delivers
superior performance. Figure 4 shows
the power efficiency (85% peak) of the
5V application in Figure 3. Figure 5 shows
the load and line regulation (±0.5%) of
the 5V application in Figure 3. Figures
6 and 7 show its 50m A-to-250m A load
step transient and 1m A resistive load
start-up waveforms, respectively.
October 2012 : LT Journal of Analog Innovation | 27
By sampling the isolated output voltage directly from the
primary-side flyback waveform, the LT8300 requires no
opto-coupler or transformer third winding for regulation.
The output voltage is set with a single external resistor.
VIN
36V TO 72V
2.2µF
•
300µH
VIN
1M
VOUT–
RFB
GND
VIN = 36V
5.15
70
VIN = 72V
60
OUTPUT VOLTAGE (V)
80
VIN = 48V
50
40
30
20
5.10
5.05
5.00
4.95
4.90
VIN = 36V
VIN = 48V
VIN = 72V
4.85
10
0
50
100
150
200
LOAD CURRENT (mA)
250
Figure 4. Power efficiency of the 5V application in
Figure 3
300
4.80
ILPRI
100mA/DIV
VSW
50V/DIV
VSW
50V/DIV
VOUT
500mV/DIV
VOUT
5V/DIV
500µs/DIV
Figure 6. 50mA-to-250mA load step transient
waveforms of the 5V application in Figure 3
28 | October 2012 : LT Journal of Analog Innovation
0
50
100
150
200
LOAD CURRENT (mA)
250
300
Figure 5. Output load and line regulation of the 5V
application in Figure 3
IOUT
100mA/DIV
The output voltage in a typical LT8300
application can be expressed as
The first term in the VOUT equation does
not have temperature dependence, but
the output diode forward voltage VF has
a significant negative temperature coefficient (–1mV/°C to –2mV/°C). Such a negative
temperature coefficient produces approximately 200mV to 300mV voltage variation on the output across temperature.
5.20
90
POST REGULATOR ELIMINATES
OUTPUT TEMPERATURE VARIATION
R 
VOUT = 100µA •  FB  − VF
 NPS 
210k
100
EFFICIENCY (%)
47µF
SW
40.2k
Figure 3. A 5V/300mA micropower
isolated flyback converter from a
36V-to-72V input
19µH
•
LT8300
EN/UVLO
0
VOUT+
5V
1mA TO 300mA
4:1
500µs/DIV
Figure 7. 1mA resistive load start-up waveforms of
the 5V application in Figure 3
For relatively high voltage outputs, say
12V and 24V, the output diode temperature coefficient has a negligible effect on
the output voltage regulation. But for
lower voltage outputs, such as 3.3V and
5V, the output diode temperature coefficient contributes an additional 2%
to 5% output voltage regulation.
For designs requiring tight output voltage
regulation across temperature, a micropower low dropout linear regulator can be
added to post-regulate the LT8300 output.
The LT8300 should be programmed slightly
higher than the sum of the regulation
voltage and the LDO’s dropout voltage.
Figure 8 shows the LT8300 combined
with a LT3009-3.3 post-regulator to
generate a 3.3V/20m A isolated output from an 18V-to-32V input. The
no-load input standby current is less
than 250µ A as shown in Figure 9,
which conforms to DEF-STAN61-5.
design features
The LT8300 greatly simplifies the design of
isolated flyback converters, improves light load
efficiency and reduces no-load input standby
current when compared to traditional schemes.
400
VIN
18V TO 32V
1µF
VOUT+
3.3V
0mA TO 20mA
OUT
LT3009-3.3
150µH
•
LT8300
EN/UVLO
IN
•
150µH
VIN
1M
D1
Z1
1µF
SHDN
GND
300
1µF
IVIN (µA)
L1
1:1
VOUT–
SW
42.2k D1: DIODES INC. SBR0560S1-7
L1: DRQ73-151-R
Z1: CENTRAL CMDZ4L7
93.1k
RFB
GND
200
100
0
Figure 8. A 3.3V/20mA micropower isolated converter from an 18V-to-32V input conforming to DEF-STAN61-5
VARIOUS INPUT-REFERRED POWER
SUPPLIES
VOUT+
10V
50mA
The LT8300 greatly simplifies the design
of isolated flyback converters, improves
light load efficiency and reduces no-load
input standby current when compared
VOUT–
VIN
15V TO 80V
VOUT+
10V
100mA
1µF
Z1
L1
330µH
VIN
LT8300
EN/UVLO
D1
1M
SW
4.7µF
30
32
Z1
L1
330µH
VIN
LT8300
EN/UVLO
102k
118k
D1
SW
102k
118k
RFB
RFB
GND
L1: COILTRONICS DR73-331-R
D1: DIODES INC. SBR1U150SA
Z1: CENTRAL CMDZ12L
28
VOUT–
1µF
1M
24
26
VIN (V)
Figure 11. A VIN to (VIN – 10V) micropower converter
VIN
15V TO 80V
4.7µF
22
to traditional schemes. The high level
of integration and the use of boundary
and low ripple burst modes results in
a simple to use, low component count,
and high efficiency solution for isolated
power supplies, as well as various special nonisolated power supplies. n
CONCLUSION
Figure 10. A VIN to (VIN + 10V) micropower converter
20
Figure 9. No-load input standby current of the 3.3V
application in Figure 8
micropower converter. In both of these
converters, the LT8300’s unique feedback
sensing scheme is used to easily develop
an output voltage that tracks VIN .
In addition to isolated power supplies, the
LT8300 can be used in various nonisolated
applications. Two interesting applications
are input-referred positive and negative
power supplies often used for special gate
drivers. Figure 10 shows a simple VIN to
(VIN +10V) micropower converter, and
Figure 11 shows a simple VIN to (VIN – 10V)
18
GND
L1: COILTRONICS DR73-331-R
D1: DIODES INC. SBR1U150SA
Z1: CENTRAL CMDZ12L
October 2012 : LT Journal of Analog Innovation | 29
What’s New with LTspice IV?
Gabino Alonso
Follow @LTspice on Twitter for
up-to-date information on models, demo circuits,
events and user tips: www.twitter.com/LTspice
NEW DEMO CIRCUITS
NEW MODELS
Overvoltage Protection and Pushbutton
Controllers
Battery Chargers, Bipolar Supplies and
LED Drivers
High Speed Amplifiers and Resistor
Networks
• LT4363-2: Overvoltage
regulator with 250V surge protection (5.5V–250V to 16V clamped
output) www.linear.com/LT4363
• LT3796: Boost LED driver with
short-circuit protection and current
monitor (9V–60V to 85V LED string at
400m A) www.linear.com/LT3796
• LT5400: Quad matched resistor
network www.linear.com/LT5400
• LTC2955: Pushbutton on/off
control with auto turn-on at
12V (12V or battery backup to 3.3V at
20m A) www.linear.com/LTC2955
• LTC3260: Low noise ±12V power supply
from a single 15V Input (15V to ±12V at
50m A) www.linear.com/LTC3260
Step-Down Regulators
• LT3988: Dual 60V step-down regulator
(7V–60V to 5V at 1A and 3.3V at
1A) www.linear.com/LT3988
• LT3992: FMEA fault tolerant dual
converter (7V–60V to 5V at 2A and
3.3V at 2A) www.linear.com/LT3992
• LT8611: µ Power synchronous step-down
regulator with current sense (3.8V–42V to
3.3V at 2.5A) www.linear.com/LT8611
• LTM4620: High efficiency 8-phase
100A step-down regulator (4.5V–16V to
1V at 100A) www.linear.com/LTM4620
• LTM8062A: 2A, 4-cell Li-ion battery
charger (18V–32V to 16.4V at 2A)
www.linear.com/LTM8062
Currrent Sense Amplifiers
• LT1787: Bidirectional current sense
amplifier with offset bipolar
output www.linear.com/LT1787
• LT6105: Unidirectional current sense
amplifier for negative supplies
www.linear.com/LT6105
• LT6106: Single supply, unidirectional current sense amplifier
www.linear.com/LT6106
• LTM8026: Two 2.5V series super­capacitor
charger (7V–36V to 5V at 5.6A)
www.linear.com/LTM8026
What is LTspice IV?
LTspice® IV is a high performance SPICE simulator, schematic capture and waveform viewer designed to speed
the process of power supply design. LTspice IV adds enhancements and models to SPICE, significantly reducing
simulation time compared to typical SPICE simulators, allowing one to view waveforms for most switching
regulators in minutes compared to hours for other SPICE simulators.
LTspice IV is available free from Linear Technology at www.linear.com/LTspice. Included in the download is a
complete working version of LTspice IV, macro models for Linear Technology’s power products, over 200 op amp
models, as well as models for resistors, transistors and MOSFETs.
30 | October 2012 : LT Journal of Analog Innovation
• LTC6417: 1.6GHz low noise high
linearity differential buffer/16-bit
ADC driver www.linear.com/LTC6417
Energy Harvesting
• LTC3109: Auto-polarity, ultralow
voltage step-up converter and power
manager www.linear.com/LTC3109
• LTC3588-2: Piezoelectric energy harvesting
power supply with 14V minimum
VIN www.linear.com/LTC3588-2
Step-Down Regulators
• LT3975: 42V, 2.5A, 2MHz step-down
switching regulator with 2.7µ A quiescent
current www.linear.com/LT3975
• LT3976: 40V, 5A, 2MHz step-down
switching regulator with 3.3µ A quiescent
current www.linear.com/LT3976
• LTC3605A: 20V, 5A synchronous step-down
regulator www.linear.com/LTC3605A
• LTC3626: 20V, 2.5A synchronous
monolithic step-down regulator with
current and temperature monitoring
www.linear.com/LTC3626
• LTM4620: Dual 13A or single
26A DC/DC µModule regulator
www.linear.com/LTC4260
Multi-Topology Regulators
• LT3758A: HV, boost, flyback, SEPIC and
inverting controller with improved
transient www.linear.com/LT3758
design ideas
Power User Tip
PIECEWISE LINEAR FUNCTION FOR VOLTAGE OR CURRENT SOURCES
Piecewise linear (PWL) functions are used to construct a waveform from a series of
straight line segments connecting points defined by the user. Since PWL functions are
useful in creating custom waveforms, they are typically used in defining voltage or
current sources.
Other Forms of PWL Statement
To add a PWL function to a voltage or current source:
LTspice IV supports many other forms of PWL statement. To explore these you will
have to directly edit your statement by right-clicking on the text line with the PWL
statement (not the component symbol), in the schematic editor. Some examples of
alternate PWL forms:
1.Right-click on the symbol in the schematic editor
• Repeating data pairs a specified number of cycles, or forever:
• LT3957A:
Boost, flyback, SEPIC and
2.Click Advanced
inverting
converter
with
5A, 40V
switch
3.Select either
PWL(t1, v1,
t2, v2…)
or PWL
File:
www.linear.com
/
LT3957A
n
4.Depending on your choice in step 3, either enter the PWL values or choose a file.
If you choose to enter the values directly, the PWL statement will be built from
your values. The syntax of a PWL statement is a list of two-dimensional points that
represent time and value data pairs where the time value is in ascending order:
PWL (0 0 1m 1 2m 1 3m 0)
PWL REPEAT FOR 5 (0 0 1m 1 2m 1 3m 0) ENDREPEAT
PWL REPEAT FOREVER (0 0 1m 1 2m 1 3m 0) ENDREPEAT
• A trigger expression that turns the source on as long as the expression is true:
PWL (0 0 1m 1 2m 1 3m 0) TRIGGER V(n003)>1
• Scaled time or source values:
PWL TIME_SCALE_FACTOR=0.5 VALUE_SCALE_FACTOR=2 (0 0 1m 1 2m 1 3m 0)
Try using one of these forms of PWL expressions in your next simulation.
Happy simulations!
Time values can also be defined relative to the previous time value by prefixing the
time value with a + sign:
PWL (0 0 +1m 1 +1m 1 +1m 0)
Here’s an example of the nonrelative value pairs in the dialog:
PWL functions are an easy way to create a custom waveform, typically used
to define values for a voltage or current source
WAVEFORM PRODUCED BY THE PWL STATEMENT
The list of two-dimensional points that represent time and value data pairs can be
encapsulated in a file and used in a PWL statement:
PWL (file=data.txt)
SCHEMATIC EDITOR SHOWS COMPONENT
WITH ATTACHED PWL STATEMENT
FILE WITH PWL DATA
October 2012 : LT Journal of Analog Innovation | 31
100V Surge Stopper Protects Components from
300V Transients
Hamza Salman Afzal
High voltage transients in automotive and
industrial systems are common and can last
from microseconds to hundreds of milliseconds,
sending significant energy downsteam. Transient
causes include automotive load dumps, and
spikes caused by load steps and parasitic
inductance. To avoid the risk of failure, all
electronics in these systems must either be robust
enough to directly withstand the transient energy
spikes, or they must be protected from them.
The LT4356 surge stopper is a dramatic
performance upgrade over traditional,
passive clamp protection techniques. It
actively protects downstream components
from overvoltage by regulating the gate
of a pass MOSFET and it limits current
with the help of a standard sense resistor.
Figure 1 shows a typical 12V application.
The LT4356 has a rated maximum of
100V with an operating voltage range
of 4V to 80V, making it ideal for protecting downstream electronics in a
wide variety of industrial and automotive applications. Nevertheless, some
circuits require protection against
transients as high as 200V to 300V.
Figure 2 shows one way that the LT4356
can be made to suppress such high voltages, but at the cost of the current limiting
feature. In Figure 2 the VCC and SNS pins
are decoupled from the raw input voltage
and separately clamped to a safe value
below 100V. Since the VCC and SNS pins are
of necessity disconnected from the input
path, current sensing is not possible and
the circuit serves only as a voltage clamp.
32 | October 2012 : LT Journal of Analog Innovation
VIN
12V
10mΩ
VOUT
IRLR2908
10Ω
383k
VCC
SNS
GATE
SHDN
IN+
100k
UNDERVOLTAGE
102k
OUT
VCC
FB
4.99k
LT4356DE-1
EN
AOUT
GND
TMR
FLT
DC-DC
CONVERTER
SHDN GND
FAULT
0.1µF
Figure 1. 12V overvoltage regulator
It is possible to overcome this limitation
by cascading a second pre-regulating
MOSFET, Q2, as shown in Figure 3. Q2
clamps the VCC and SNS pins to a safe
level, restores the current limit feature and as an added benefit, shares
SOA (safe operating area) stress with Q1.
and sending power through to the output.
Thus R3 and D1 are critical to start-up.
Under normal operating conditions the
GATE pin limits itself to about 12.5V above
the output, so with 12V at the input,
Q1’s gate is biased to 24.5V and Q2’s gate
is biased slightly lower, about 24V.
When power is first applied, R3 and D1
pull up on the gate of Q2, which in turn
passes power through to the LT4356. The
GATE pin then pumps up the gates of Q1
and Q2, fully enhancing both MOSFETs
When the input is subjected to a high
voltage transient, R3 and D1 pull up on
the gate of Q2, which in turn is clamped
by D2 to approximately 80V. Acting as
a source follower, Q2’s source rises no
further than about 75V, keeping VCC and
SNS safely below their 100V maximum
rating. Unlike the shunt clamped application shown in Figure 2, the series clamped
topology of Figure 3 permits full use of
the LT4356’s current limiting feature. Q1
regulates in the normal way, limiting the
output voltage as prescribed by R1 and R2.
Figure 2. 24V application circuit capable of
withstanding 150V
VIN
24V
Q1
IRF640
1k
1W
VOUT
CLAMPED
AT 32V
10Ω
118k
SNS GATE
D2*
SMAT70A
OUT
VCC
FB
4.99k
LT4356DE-1
SHDN
FLT
EN
*DIODES INC.
GND
TMR
CTMR
0.1µF
An added benefit of the topology shown
in Figure 3 is that Q2 shares SOA stress
with Q1. For inputs in the range of
150V to 200V, the SOA stress is shared
equally between Q1 and Q2. In certain
applications this allows two inexpensive
design ideas
The LT4356 has a rated maximum of 100V with an operating
voltage range of 4V to 80V, but a little extra circuitry
enables it to protect against transients as high as 300V.
VIN
12V
Figure 3. Pre-regulator topology
extends protection range of the LT4356.
Figure 4 shows the complete circuit.
RSNS
Q2
Q1
VOUT
R3
D1
D3
D2
80V
VCC
GATE
SNS
OUT
R1
LT4356
FB
R2
GND
TMR
CTMR
MOSFETs to replace a single, and much
more costly, special high SOA device. As
the peak input voltage requirement rises
above 200V, the SOA becomes increasingly concentrated in Q2 and the series
connection offers no substantial relief.
described, Q2’s gate is clamped at 80V so
that with a 300V input, Q2 drops 225V,
while Q1 sees no more than 75V total. For
this reason a 250V device is specified for
Q2, and a 100V device suffices for Q1. It is
possible to withstand even higher input
voltages by appropriate selection of Q2.
Figure 4 shows a complete circuit based
on the new topology, designed to withstand up to 300V peak input. As previously
VIN
MAX RANGE: 0V–300V
OPERATING RANGE: 9V–16V
D1
1N4148
10Ω
RSNS
33mΩ
D4
1N756A
Q1
FQB55N10
100Ω
+
10Ω
Q3
2N3904
D3
1N4148
D2
SMAJ70A*
0.039µF
VCC
GATE
SNS
OUT
100µF
R1
178k
FB
SHDN
R2
15k
LT4356
AOUT
IN+
*DIODES INC.
VOUT
1.5A LOAD CURRENT
16V REGULATION
INPUT
50V/DIV
10k
0.1µF
Figure 5. Results of 300V spike on input of circuit in
Figure 4
CSNUB
0.01µF
Q2
FDB33N25
R3
10k
Figure 5 shows the results of the circuit subjected to a 300V spike. CTMR is
sized to ride through such excursions,
but longer duration surges will be
interrupted, thereby protecting the
MOSFETs from certain destruction. n
When designing circuits to withstand
such high input voltages, it is important
Figure 4. 16V overvoltage regulator capable of blocking 300V transients
RSNUB
51Ω
to recognize the potential for high dV/dt
at the input and resulting consequences.
Until the circuit can respond, current
arising from an instantaneously applied
high input voltage is limited only by
the parasitic inductance and the path
resistance to the output capacitor. While
most test waveforms specify some sufferable rise time, an infinite input slew
rate is not inconceivable, such as might
arise during bench testing. Q3 is added
to give the LT4356’s current limit loop
a head start under these conditions.
FLT
GND
OUTPUT
20V/DIV
2ms/DIV
EN
TMR
CTMR
0.1µF
October 2012 : LT Journal of Analog Innovation | 33
Accurate PWM LED Dimming without External Signal
Generators, Clocks or µControllers
Keith Szolusha
LEDs can be dimmed in two ways: analog and pulse-width modulation (PWM) dimming.
Analog dimming changes LED light output by simply adjusting the DC current in the string,
while PWM dimming acheives the same effect by varying the duty cycle of a constant
current in the string to effectively change the average current in the string. Despite its
attractive simplicity, analog dimming is inappropriate for many applications because it
loses dimming accuracy by about 25%+ at only 10:1 brightness levels, and it skews the
color of the LEDs. In contrast, PWM dimming can produce 3000:1 and higher dimming
ratios (at 100Hz) without any significant loss of accuracy, and no change in LED color.
The LT3761 combines the simplicity of
analog dimming with the accuracy of
PWM dimming by generating its own
PWM signal. High dimming ratios are
possible by adjusting a simple DC signal
at its dimming input—no additional
PWM-generating microcontrollers, oscillators or signal generators are required.
The LT3761’s internal PWM signal
can produce 25:1 dimming, while it
can still deliver up to 3000:1 dimming with an external PWM signal.
HIGH POWER LED DRIVER
The LT3761 is a high power LED driver
similar to the LT3755-2 and LT3756-2
family. It is a 4.5V-to-60V input to
0V-to-80V output single-switch controller IC that can be configured as a
boost, SEPIC, buck-boost mode or buck
mode LED driver. It has a 100kHz to
1MHz switching frequency range, open
LED protection, extra internal logic to
provide short-circuit protection, and can
be operated as a constant voltage regulator with current limit or as a constantcurrent SLA battery or supercap charger.
Figure 1 shows a 94% high efficiency 60V,
1A (60W) 350k Hz automotive headlamp
34 | October 2012 : LT Journal of Analog Innovation
application with PWM dimming. The
LT3761 uses the same high performance
PWM dimming scheme as the LT3755/LT3756
family, but with the additional feature
of the internally generated PWM dimming signal and no additional pins.
INTERNAL PWM DIMMING
GENERATOR
Unlike other high power LED drivers, the
LT3761 can generate its own PWM dimming signal to produce up to 25:1 dimming. This enables it to produce accurate
PWM dimming without the need for
external PWM-generating components. The
Figure 1. 94% efficient boost LED driver for automotive headlamp with 25:1 internal PWM dimming
L1
10µH
VIN
8V TO
60V
CIN
2.2µF
×2
100V
499k
90.9k
EN/UVLO
VREF
1M
VIN
140k
RSENSE
10mΩ
1M
RLED
0.25Ω
1A
16.9k
FB
60W
LED
STRING
ISP
OPENLED
ISN
DIM/SS
PWM
PWMOUT
VC
RT INTVCC
RDIM
124k
CSS
0.01µF
COUT
2.2µF
×4
GND
100k
DIM
M1
SENSE
LT3761
CTRL
INTVCC
GATE
D1
CPWM
47nF
300Hz
RC
5.1k
CC
4.7nF
RT
28.7k
350kHz
M1: INFINEON BSC123N08NS3-G
D1: DIODES INC PDS5100
L1: COILTRONICS HC9-100-R
M2: VISHAY SILICONIX Si2328DS
COUT, CIN: MURATA GRM42-2X7R225K100R
INTVCC
CVCC
1µF
M2
(CURRENT DERATED FOR VIN < 10V)
design ideas
The LT3761 generates its own PWM signal to achieve accurate PWM dimming,
but with the simple control of analog dimming. High dimming ratios are possible
by adjusting a simple DC signal at its dimming input—no additional PWMgenerating microcontrollers, oscillators or signal generators are required.
Figure 2. Internally generated PWM signal and LED
current for the application in Figure 1
VDIM = 7.7V
DCPWM = 96%
VDIM = 4V
DCPWM = 50%
ILED
1A/DIV
VDIM = 1.5V
DCPWM = 10%
VDIM = 0.4V
DCPWM = 4.3%
0.5ms/DIV
LT3761 requires only an external DC voltage, much like analog dimming control,
for high performance PWM dimming at
a chosen frequency. It can still receive a
PWM input signal to drive the LED string
with that signal in standard fashion.
the PWM pin to GND according to the
equation: fPWM = 14kHz • nF/CPWM. The
duty cycle of the signal at PWMOUT is set
by a µ A-scale current into the DIM/SS pin
as shown in Figure 3. Internally generated pull-up and pull-down currents
on the PWM pin are used to charge and
discharge its capacitor between the high
and low thresholds to generate the duty
cycle signal. These current signals on the
PWM pin are small enough so they can
be easily overdriven by a digital signal
The internal PWM dimming signal
generator features programmable
frequency and duty cycle. The frequency of the square wave signal at
PWMOUT is set by a capacitor CPWM from
Figure 3. Setting the duty cycle at the DIM/SS pin
takes a µA-scale signal. This pin can also be used
with an external PWM signal for very high dimming
ratios.
CONCLUSION
The high power and high performance
LT3761 LED driver has its own onboard
PWM dimming signal generator that
is both accurate and easy to use. n
Figure 4. Given a high speed PWM input signal, the
LT3761 still provides a high speed PWMOUT signal.
100
PWMOUT DUTY RATIO (%)
from a microcontroller to obtain very
high dimming performance. The practical minimum duty cycle using the internal signal generator is about 4% if the
DIM/SS pin is used to adjust the dimming
ratio. For 100% duty cycle operation,
the PWM pin can be tied to INTVCC .
CPWMOUT = 2.2nF
80
PWM
INPUT
60
PWMOUT
5V/DIV
40
20
0
–10
0
10
20
30
DIM/SS CURRENT (µA)
40
50
200ns/DIV
October 2012 : LT Journal of Analog Innovation | 35
Eliminate Opto-Isolators and Isolated Power Supply from
Power over Ethernet Power Sourcing Equipment
Heath Stewart
Power over Ethernet (PoE), is defined by the IEEE 802.3at specification to safely
deliver application power over existing Ethernet cabling. Implementation of Power
over Ethernet requires careful architecture and component selection to minimize
system cost, while maximizing performance and reliability. A successful design must
adhere to IEEE isolation requirements, protect the Hot Swap™ FET during shortcircuit and overcurrent events, and otherwise comply with the IEEE specification.
The IEEE standard also defines PoE terminology. A device that provides power to
the network is known as power sourcing equipment (PSE), while a device
that draws power from the network
is known as a powered device (PD).
Figure 1. Traditional PSE
isolation schemes require a
number of opto-isolators and
a cumbersome, and costly
isolated DC/DC converter. The
LT4271-LT4290 solution shown
in Figure 2 eliminates these
components.
The LTC4290/LTC4271 PSE controller chipset revolutionizes PSE architecture by deleting the customary digital isolation and
removing an entire isolated power supply.
Instead, the chipset employs a proprietary isolation protocol using a low cost
Ethernet transformer pair, leading to a significant reduction in bill of materials cost.
The LTC4290/LTC4271 fourth generation PSE controller supports fully
compliant IEEE 802.3at operation,
while minimizing heat dissipation
through the use of low RDS(ON) external
MOSFETs and 0.25Ω sense resistors.
1500V ISOLATION
PSE
PD
PHY
HOST
PHY
+
PD
CONTROLLER
PSE
CONTROLLER
ISOLATED
DC/DC
CONVERTER
36 | October 2012 : LT Journal of Analog Innovation
ISOLATED
DC/DC
CONVERTER
design ideas
1500V ISOLATION
PSE
PD
PHY
Figure 2. In contrast to the
traditional scheme shown
in Figure 1, a cleaner PSE
architecture incorporates the
LTC4290/LTC4271 chipset,
achieving isolation without
any opto-isolators and
eliminatinating the need for
a dedicated isolated DC/DC
converter.
PHY
HOST
SYSTEM ISOLATION REQUIREMENTS
The PoE specification clearly lays out
isolation requirements, guaranteeing
ground loops are broken, maintaining
Ethernet data integrity and minimizing noise in the PD application circuit.
Traditional PSE isolation architectures
isolate the digital interface and power
at the host-to-PSE controller interface.
Digital isolation elements such as optocouplers are inherently expensive and
unreliable. ICs capable of performing
the isolation function are cost-prohibitive or do not support fast I2C transfer
rates. In addition, isolated DC/DC converters needed to power the PSE logic
increase board space and system cost.
ISOLATION MADE EASY
The LTC4290/LTC4271 chipset takes a different approach to PSE isolation (Figure 2)
by moving all digital functions to the
host side of the isolation barrier. This
significantly reduces the cost and complexity of required components. There is no
longer the need for a separate, isolated
DC/DC power supply; the LTC4271 digital
controller can use the host’s logic supply.
The LTC4271 controls the LTC4290 using
a transformer-isolated communication
scheme. An inexpensive and ubiquitous
Ethernet transformer pair replaces six
LT4271
LT4290
opto-couplers. Intra-IC communication
including port management, reset and
fast port shutdown are encoded in a
protocol designed to minimize radiated
energy and provide 1500V of isolation.
ADVANCED FOURTH GENERATION
FEATURES
Linear’s PSE family incorporates a wealth
of PoE experience and expertise backed
by well over 100 million shipped ports.
This latest PSE generation adds features
to a proven, field-tested product line.
New features include field-upgradable
firmware, future-proofing platforms that
incorporate the LTC4290/LTC4271. Also
new is optional 1-second current averaging, which simplifies host power management. The highest grade LTC4290A analog
controller enables delivered PD power
of up to 90W using the new LTPoE++™
physical classification scheme.
PD
CONTROLLER
ISOLATED
DC/DC
CONVERTER
detection mechanism that ensures
immunity from false PD detection.
Advanced power management includes
prioritized fast shutdown, 12-bit per-port
voltage and current read back, 8-bit programmable current limits and 7-bit programmable overload current thresholds.
A 1MHz I2C interface allows a host
controller to digitally configure the
IC or query port readings. “C” libraries are available to reduce engineering
costs and improve time to market.
CONCLUSION
The LTC4290/LTC4271 builds on
an established, robust lineup of
Linear PSE solutions by slashing BOM costs while providing an
overall best-in-field solution. n
As with previous generations, a key benefit
of the LTC4290/LTC4271 chipset architecture is the lowest-in-industry power
dissipation, making thermal design significantly easier than designing with PSEs that
integrate more fragile and higher RDS(ON)
MOSFETs. System designers will appreciate
the robustness provided by 80V-tolerant
port pins. PD discovery is accomplished
using a proprietary dual-mode, 4-point
October 2012 : LT Journal of Analog Innovation | 37
Product Briefs
NANO-CURRENT HIGH VOLTAGE
MONITOR
The LTC2960 is a nano-current high voltage monitor that provides supervisory
reset generation and undervoltage or
overvoltage detection. Low quiescent
current (0.85µ A) and a wide operating voltage range of 2.5V to 36V make
the LTC2960 useful in multicell battery
applications. Status indicators RST and
OUT are available with 36V open-drains
or low voltage active pull-ups.
External resistive dividers configure
monitor thresholds for each of the two
comparator inputs. The LTC2960 monitors the ADJ input and pulls RST output
low when the voltage at the comparator input drops below the comparator
threshold. RST remains low until the
ADJ input rises 2.5% above the threshold.
A reset timeout period delays the return
of the RST output to a high state to allow
voltage settling, initialization time and/
or a microprocessor reset function. An
additional comparator with inverting or
noninverting input includes 5% hysteresis and is indicated on the OUT pin.
A manual reset (MR) input enables external activation of the RST output. Other
options include a selectable 15ms or
200ms reset timeout periods. A logic supply pin, DVCC, provides a power input for
the active pull-up circuits. The LTC2960 is
available in 8-lead 2mm × 2mm DFN and
TSOT-23 packages. Electrical specifications
are guaranteed from –45ºC to 125ºC.
38 | October 2012 : LT Journal of Analog Innovation
MULTIPHASE STEP-UP DC/DC
PROVIDES 10V GATE DRIVE, RIDES
THROUGH COLD CRANK
FULLY DIFFERENTIAL AMPLIFIER
DRIVES 18-BIT ADCs & CONSUMES
ONLY 5mW
The LTC3862-2 is a high power multiphase
current mode step-up DC/DC controller.
Like its predecessors, the LTC3862
and LTC3862-1, the LTC3862-2 uses
a constant frequency, peak current
mode architecture with two channels
operating 180° out of phase. It retains
popular features, including adjustable
slope compensation gain, max duty
cycle and leading edge blanking,
programmable frequency with a external
resistor (75kHz to 500kHz) or SYNC to
an external clock with a phase-lockable
fixed frequency of 50kHz to 650kHz.
The PHASEMODE control pin allows for
2-, 3-, 4-, 6-, or 12-phase operation.
Linear Technology announces the LTC6362,
a low power fully differential amplifier
that can drive high precision 16- and
18-bit SAR ADCs at only 1m A supply current. With 200µV max input offset voltage and 3.9nV/√Hz input-referred noise,
it is well suited for precision industrial
and data acquisition applications.
Like the LTC3861-1, the LTC3862-2’s
internal LDO regulates to 10V, optimizing
gate drive for most automotive and
industrial grade power MOSFETs. But
unlike the LTC3861-1, the LTC3862-2’s
undervoltage lockout (UVLO) falling
threshold is reduced to 4V from the
original 7V. UVLO shuts off the circuit
when there is not enough gate drive.
Lowering it provides compatibility with
the most efficient 10V gate drive MOSFETs,
while allowing the part to regulate
even when the input voltage dips below
10V (as when an engine is turned on).
The LTC3862-2 also has improved
current sense matching, channel-tochannel and chip-to-chip. This allows
thermal dissipation to be shared
more evenly between phases.
The LTC6362 has an output commonmode pin with a 0.5V to 4.5V range, and
18-bit settling time of 550ns with an
8VP–P output step, making it ideal for
driving ADCs such as the LTC2379-18
in multiplexed input and control loop
applications. This 18-bit SAR ADC features
digital gain compression, which sets its
full scale input range at 10% to 90% of
the reference voltage. Together with the
rail-to-rail output stage of the LTC6362,
this feature eliminates the need for a
negative supply rail, simplifying the circuit
and minimizing power consumption.
The flexible architecture of the LTC6362
can convert single-ended DC-coupled,
ground-referenced signals to differential,
or DC level shift differential input signals.
The low input bias current, low offset
voltage and rail-to-rail inputs of the
LTC6362 enable its use in a high impedance configuration to interface directly
to sensors early in the signal chain.
The LTC6362 is available in MSOP-8
and 3mm × 3mm DFN packages, with
fully guaranteed specifications over
the 0°C to 70°C, –40°C to 85°C and
–40°C to 125°C temperature ranges.
product briefs
The LTC2960 is a nano-current high voltage
monitor that provides supervisory reset generation
and undervoltage or overvoltage detection. Low
quiescent current (0.85μA) and a wide operating
voltage range of 2.5V to 36V make the LTC2960
useful in multicell battery applications.
60V SYNCHRONOUS BUCK-BOOST
LED DRIVER DELIVERS OVER 100W
OF LED POWER
The LT3791 is a synchronous buck-boost
DC/DC LED driver and voltage controller,
which can deliver over 100W of LED power.
Its 4.7V to 60V input voltage range makes
it ideal for a wide variety of applications, including automotive, industrial
and architectural lighting. Similarly, its
output voltage can be set from 0V to 60V,
enabling the LT3791 to drive a wide range
of LEDs in a single string. Its internal
4-switch buck-boost controller operates
from input voltages above, below or equal
to the output voltage, ideal for applications such as automotive, where the input
voltage can vary dramatically during stop/
start, cold crank and load dump scenarios.
Transitions between buck, pass-through
and boost operating modes are seamless, offering a well regulated output even
with wide variations of supply voltage.
The LT3791’s unique design utilizes three
control loops to monitor input current,
LED current and output voltage to deliver
optimal performance and reliability.
The LT3791 uses four external switching
MOSFETs and delivers from 5W to over
100W of continuous LED power with efficiencies up to 98.5%. LED current accuracy
of +6% ensures constant lighting while
±2% output voltage accuracy enables the
converter to operate as a constant voltage
source. The LT3791 utilizes either analog or PWM dimming as required by the
application. Furthermore, its switching
frequency can be programmed between
200kHz and 700kHz or synchronized to
DEVICE OPTION
OUTPUT TYPE
INPUTS
RESET TIMEOUT PERIOD
LTC2960-1
36V Open Drain
ADJ/IN+
15ms/200ms
LTC2960-2
36V Open Drain
ADJ/IN-
15ms/200ms
LTC2960-3
Active Pull-up
ADJ/IN+
200ms
LTC2960-4
Active Pull-up
ADJ/IN-
200ms
an external clock. Additional features
include output disconnect, input and
output current monitors, open and shorted
LED detection and integrated fault protection. The LT3791EFE is available in 38-lead
thermally enhanced TSSOP package.
30MHz TO 1.4GHZ WIDEBAND
I/Q DEMODULATOR WITH IIP2
OPTIMIZATION & DC OFFSET
CANCELLATION IMPROVES ZERO-IF
RECEIVER PERFORMANCE
The LTC5584 is an ultrawide bandwidth
direct conversion I/Q demodulator with
outstanding linearity of 31dBm IIP3 and
70dBm IIP2. The device offers best-in-class
demodulation bandwidth of over 530MHz,
supporting the latest generation of
LTE multimode, LTE Advanced receivers, as
well as digital predistortion (DPD) receivers. The I/Q demodulator operates over
a wide frequency range from 30MHz to
1.4GHz , covering a broad range of VHF and
UHF radios and the 450MHz /700MHz
LTE frequency bands. Unique to the
LTC5584 are two built-in calibration
features. One is advanced circuitry that
enables the system designer to optimize
the receiver’s IIP2 performance, increasing from a nominal 70dBm to an unprecedented 80dBm or higher. The other is
on-chip circuitry to null out the DC offset
voltages at the I and Q outputs. Combined
with a 9.9dB noise figure, these features
enhance the dynamic range performance in receivers. Moreover, the device
exacts P1dB of 12.6dBm, along with its
13.6d B noise figure under a 0d Bm in-band
blocker, ensuring robust receiver performance in the presence of interference.
To enhance its flexibility for use in low
IF receiver applications, the LTC5584
exhibits very low I/Q amplitude and phase
mismatch. The amplitude mismatch is
typically 0.02dB, while the phase error
is typically 0.25 degree, both specified
at 450MHz. This combination produces
receiver image rejection of 52dB.
With its wide bandwidth capability, the LTC5584 is ideal for multimode
LTE and CDMA DPD receivers as well as
other wideband receiver applications.
Particularly suited for DPD, these latest generation base stations are pushing
demodulation bandwidth of over 300MHz.
The LTC5584 exceeds these bandwidth
requirements while delivering better than
±0.5dB conversion gain flatness. Beyond
wireless infrastructure applications, the
LTC5584 is ideal for military receivers,
broadband communications, point-topoint microwave data links, image-reject
receivers and long-range RFID readers.
The LTC5584 is offered in a 24-lead
4mm × 4mm QFN package. The device
is specified for case operating temperature from –40°C to 105°C. Powered from
a single 5V supply, the LTC5584 draws
a total supply current of 200m A. The
device provides a digital input to enable
or disable the chip. When disabled, the
IC draws 11µ A of leakage current typical.
The demodulator’s fast turn-on time of
200ns and turn-off time of 800ns enables
it to be used in burst-mode receivers. n
October 2012 : LT Journal of Analog Innovation | 39
highlights from circuits.linear.com
–3.3V NEGATIVE CONVERTER WITH 1A OUTPUT CURRENT LIMIT
VIN
3.8V TO 38V
The LT8611 is a compact, high efficiency, high speed synchronous
monolithic step-down switching regulator that consumes only 2.5μA of
quiescent current. Top and bottom power switches are included with
all necessary circuitry to minimize the need for external components.
The built-in current sense amplifier with monitor and control pins
allows accurate input or output current regulation and limiting. Low
ripple Burst Mode® operation enables high efficiency down to very low
output currents while keeping the output ripple below 10mVP–P.
circuits.linear.com/585
4.7µF
VIN
0.1µF
BST
SW
SYNC
ISP
LT8611
60.4k
4.7µF
IMON
TR/SS
0.1µF
1µF
10pF
PG
INTVCC
circuits.linear.com/585
1µF
ISN
BIAS
ICTRL
LTspice IV
0.1µF
4.7µH
EN/UV
RT
PGND GND
60.4k
1M
FB
412k
47µF
VOUT
–3.3V
1A
0.05Ω
f = 700kHz
2A, 4-CELL LI-ION BATTERY CHARGER WITH C/10 TERMINATION
The LTM8062/LTM8062A are complete 32V VIN, 2A μModule power
tracking battery chargers. The LTM8062/ LTM8062A provide a constantcurrent/constant-voltage charge characteristic, a 2A maximum charge
current, and employ a 3.3V float voltage feedback reference, so any
desired battery float voltage up to 14.4V for the LTM8062 and up to
18.8V for the LTM8062A can be programmed with a resistor divider.
circuits.linear.com/584
VIN
22V TO 32VDC
LTM8062A
VINA
4.7µF
BAT
VIN
VINREG
CHRG
RUN
FAULT
TMR
ADJ
NTC
BIAS
GND
LTspice IV
+
(OPTIONAL
ELECTROLYTIC
CAPACITOR)
0.47µF
3.3µH
1.24M
4-CELL
Li-Ion
(4 × 4.1V)
BATTERY
PACK
312k
circuits.linear.com/584
EXTERNAL 3.3V
4.3V TO 42V INPUT, 3.3V, 5A OUTPUT STEP-DOWN CONVERTER
The LT3976 is an adjustable frequency monolithic buck switching regulator
that accepts a wide input voltage range up to 40V. Low quiescent current
design consumes only 3.3μA of supply current while regulating with no load.
Low ripple Burst Mode operation maintains high efficiency at low output
currents while keeping the output ripple below 15mV in a typical application.
circuits.linear.com/583
VIN
4.3V TO 42V
OFF ON
VIN
EN
BOOST
PG
10µF
SW
SS
LTspice IV
circuits.linear.com/583
10nF
PDS540
LT3976
RT
SYNC
130k
2Ω
470pF
OUT
FB
GND
1M
10pF
576k
VOUT
3.3V
5A
47µF
2
f = 400kHz
L, LT, LTC, LTM, Linear Technology, the Linear logo, LTspice, Burst Mode, Dust Networks, PolyPhase and µModule are registered trademarks, and Eterna, Hot Swap, LTpowerPlay, LTPoE++ and SmartMesh are trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
© 2012 Linear Technology Corporation/Printed in U.S.A./57K
Linear Technology Corporation
1630 McCarthy Boulevard, Milpitas, CA 95035
(408) 432-1900
www.linear.com
Cert no. SW-COC-001530