V23N3 - OCTOBER

October 2013
I N
T H I S
I S S U E
15V buck-boost converters
with ultralow 1.3µA IQ 15
inverting DC/DC controller
converts a positive input
to a negative output with a
single inductor 20
improve solar battery
Volume 23 Number 3
RF/IF Amplifiers with OIP3 of
47dBm/50dBm at 240MHz
Ease Implementation,
Guarantee Performance
Greg Fung
charger efficiency in
low light 24
replace halogen bulbs with
LEDs in 24VAC and 12VAC
lighting systems 28
Our communication infrastructure’s limited bandwidth is nearly
filled to capacity by our increasing thirst for data transmitted
via smartphone, TV, GPS and Wi-Fi. To quench this thirst,
communications architects define systems that pack increasingly
more data into limited bandwidth,
but data rate improvements
come at a price: the need for
increasingly higher fidelity transmit
and receive signal chains.
When it comes to amplifiers, low noise and high
linearity are required to faithfully reproduce a signal
without degrading the original signal. At low signal powers, undesired noise must be low enough to
allow the intended signal to rise above the noise floor.
At high signal levels, a linear amplifier must prevent
unwanted harmonics and intermodulation products
from masking the intended signal. The LTC®6431-15
and LTC6430-15 achieve both of these goals.
The LTC6431-15 and LTC6430-15 are two fixed gain
amplifiers that feature very high OIP3 (linearity) with
very low associated noise. The LTC6431-15 is a singleended radio frequency (RF)/intermediate frequency
The LT®3795 LED driver reduces peak EMI without incurring LED flicker. See page 12.
Caption
w w w. li n ea r.com
(continued on page 4)
Linear in the News
In this issue...
NEW ELECTRONIC AGE OF AUTOMOTIVE AND INDUSTRIAL PRODUCTS
COVER STORY
RF/IF Amplifiers with OIP3 of 47dBm/50dBm at
240MHz Ease Implementation, Guarantee Performance
Greg Fung
1
DESIGN FEATURES
LED Driver with Integrated Spread Spectrum
Reduces EMI without Adding Flicker
Keith Szolusha
12
15V Buck-Boost Converters with Ultralow 1.3µA
Quiescent Current are Tailored to Micropower
Applications and the Internet of Things
Dave Salerno
15
Inverting DC/DC Controller Converts a Positive
Input to a Negative Output with a Single Inductor
David Burgoon
20
DESIGN IDEAS
What’s New with LTspice IV?
Gabino Alonso
22
Solar Battery Charger Maintains
High Efficiency in Low Light
J. Celani
24
Meet Green Standards in 24VAC and 12VAC Lighting
Systems: Replace Halogen Bulbs with LEDs Driven by
High Power Factor, High Efficiency Converter
Keith Szolusha
28
back page circuits
32
We are now in the midst of a new electronic age in automotive and industrial products. This is the theme of the just-released Linear Technology 2013
Annual Profile. If you examine the history of industrial output, you can see
various trends—from an industrial cottage industry in the early 1800s, to the
industrial revolution in which mechanization overtook many tasks previously
performed by workers, to the current electronics revolution. The latter includes
implementation of such systems as wireless transmission of sensor measurements, electronically activated valves, digital x-ray machines and proliferation
of industrial robotics. To this, you can add smart manufacturing systems and
a new level of focus on energy efficient systems. Linear’s electronics are being
widely deployed across a range of industrial systems, including medical equipment, factory automation, industrial process control, manufacturing equipment, inventory control systems, industrial wireless sensor networks, security,
instrumentation, test and measurement, and renewable energy generation.
This new age is even more evident in automotive systems. Operations that
were previously purely mechanical, such as braking and steering, can now
be performed electronically. Valuable safety features, such as collision avoidance, lane departure and parking assistance are now a reality in many vehicles.
Stored, alternatively sourced energy now assists automotive acceleration.
These new electronic automotive and industrial systems demand exceptional
performance, quality, reliability and repeatability. And a large portion of the
electronic content is analog, given that signal clarity and power efficiency are
significant design considerations. We see these new electronics in a range of
automotive systems, including body electronics, exhaust systems, navigation
and entertainment, battery management systems, LED lighting, electronic braking, electronic steering and engine management. The electronic content is especially significant in the growing market for hybrid and all-electric vehicles.
Over the past several years, Linear Technology has introduced an array of high
performance analog products to meet the growing demands in automotive and
industrial electronics. Linear’s products have been designed to operate at lower
power and high voltages, and to perform flawlessly in harsh environmental
conditions.
2 | October 2013 : LT Journal of Analog Innovation
Linear in the news
The Linear Technology
2013 Annual Profile
focuses on the
increasingly important
role that electronics
play in automotive and
industrial systems.
A few of Linear’s innovations that
have impacted the growing industrial
and automotive markets include:
CONFERENCES & EVENTS
•Battery management systems
for hybrid/electric vehicles
Tokyo, Japan, October 30—Linear
Home of the Analog Gurus Seminar,
Tokyo Conference Center Shinagawa,
•Power systems management solutions
that provide control and monitoring
of power usage, voltages, sequencing,
margining and fault logging
•Low power ultra-precise SAR (successive
approximation register) analog-todigital converters (ADCs) that enable
more accurate product testing
•Enhanced Power over Ethernet
(LTPoE++™) solutions that enable delivery
of up to 90W of power over traditional
Ethernet cables
•µModule® power devices that combine
several functions into one integrated
solution
•Wireless sensor network solutions
that transmit sensor output from low
power sources and operate in harsh
environments
All told, Linear is providing designers with a broad range of products
that enable solutions for expanding
applications in automotive and industrial systems. To download Linear
Technology’s 2013 Annual Profile,
visit www.linear.com/docs/43732.
LINEAR PRODUCTS AWARDED
ElectroniqueS Electron d’Or Award for best power
and energy conversion product, LTC3300-1 multicell
active battery balancer—With the LTC3300-1,
applications such as electric vehicles
(EVs), plug-in hybrid EVs and large energy
storage systems using cells with mismatched capacities are no longer limited
by the lowest capacity cell in the stack.
Electronic Products China Top 10 Power Award,
LTC3300-1 multicell active battery balancer—The
LTC3300-1 goes beyond purely dissipative
passive balancing solutions, enhancing
battery performance by efficiently transferring charge to or from adjacent cells in
order to bring mismatched cells into stateof-charge (SoC) balance within the stack.
By redistributing charge throughout the
stack, the LTC3300-1 compensates for lost
capacity due to the weakest cells, enabling
faster charging and extending the run time
and usable lifetime of the battery stack.
Technology and co-sponsor
Nikkei Electronics will provide
an overview of today’s analog
design challenges. Speakers
include Linear CTO and cofounder Bob Dobkin, Steve
Pietkiewicz, Vice President,
Power Management Products for
Linear, and Prof. A. Matsuzawa
of Tokyo Institute of Technology. More
at ac.nikkeibp.co.jp/ne/ag1030/
Measurement and Control Show, Tokyo Bigsight,
Tokyo, Japan, November 6-8—Presenting
Linear’s high speed ADCs, power management and wireless sensor network
products. More at www.jemima.or.jp/
event/keisoku2013/en/index.html
Energy Harvesting & Storage Conference, Santa
Clara Convention Center, Santa Clara, California,
November 20-21, Booths S7-S8—Linear will
showcase its energy harvesting and
wireless sensor network products.
Speakers include Dave Loconto on
energy harvesting battery charging
and Ross Yu on wireless sensor networks. More at www.idtechex.com/
energy-harvesting-usa/eh.asp. n
October 2013 : LT Journal of Analog Innovation | 3
The LTC6431-15 boasts a typical OIP3 of 47dBm at 240MHz—
essentially hammering the intermodulation products (IM3) into the
noise floor so they don’t interfere with the intended signals.
(LTC6430/1-15 continued from page 1)
FUNDAMENTAL
SOURCE
x2
y = a1x + a2
x3
+ a3
SOURCE
LOAD
y = a1x + a2x2 + a3x3
FREQUENCY
0
–10
DESIRED TONE
DESIRED SIGNAL
–20
0
–40
IM3 PRODUCT
–30
–40
–50
–60
–70
–80
–120
–140
10 20 30
–30 –20 –10 0
INPUT POWER (dBm)
2ND ORDER
10
OIP3 IN dBm
–20
–80
UNDESIRABLE
IM3 PRODUCTS
For instance, if a single tone is injected
into a nonlinear amplifier, the result is the
desired tone plus its harmonics. Normally,
these harmonics can be filtered out, as
they are far enough in frequency from the
desired tone. If two tones are injected into
40
–60
3RD ORDER
of an issue, but an amplifier’s linearity becomes increasingly important.
Linearity limits the ability to isolate the
desired signal from unwanted signals
in the frequency domain. At high input
signal levels, the desired signal rises far
above the noise floor, so noise is less
20
AMPLITUDE
NONLINEAR
AMP
MULTIPLE
TONES
IMPRESSIVE OIP3 HAMMERS
DOWN IM PRODUCTS
80
2ND ORDER
OUTPUT
FREQUENCY
60
FUNDAMENTAL
FREQUENCY
INPUT
Figure 2. Two
tones at the input
of a nonlinear
device create
intermodulation
product at the output.
–100
–90
40
50
Figure 3. Output 3rd order intercept point (OIP3)
4 | October 2013 : LT Journal of Analog Innovation
LOAD
AMPLITUDE
AMPLITUDE
NONLINEAR
AMP
AMPLITUDE (dB)
Noise limits communication system sensitivity at low input signal levels. Noise
in a communication system is characterized by the noise figure (NF), which is the
signal-to-noise power ratio at the output
divided by the signal-to-noise power
ratio at the input expressed in decibels.
There is always noise at the input of an
amplifier and it is gained up along with
desired signal. The NF is an indicator of
how much unwanted noise the amplifier itself adds to the signal. Ideally, the
amplifier would have a NF of 0dB, but
any real amplifier adds noise, so the goal
is to minimize noise impairment. Typical
IF amplifiers have noise figures of 3dB to
12d B. The LTC6431-15 and LTC6430-15
both exhibit a 3.3dB NF at 240MHz.
OUTPUT
FREQUENCY
AMPLITUDE
LOW NF FOR LOW INPUT SIGNALS
INPUT
Figure 1. A single
tone at the input of
a nonlinear device
creates harmonics
at the output.
OUTPUT POWER (dBm)
(IF) gain block that can directly drive
a 50W load, whereas the LTC6430-15
is a differential RF/IF gain block with
higher power and an even wider linear
bandwidth. These gain blocks combine state-of-the-art performance with
ease of use—eliminating implementation difficulties by internally handling
of biasing, impedance matching, temperature compensation and stability.
–100
200 210 220 230 240 250 260 270 280
FREQUENCY (MHz)
Figure 4. The LTC6431-15 boasts an OIP3 of 47dBm
at 240MHz—essentially hammering the IM3 products
of a 2-tone signal into the noise floor so that they
don’t interfere with the intended signals.
design features
The single-ended LTC6431-15 excels as an IF amplifier to overcome filter
losses, or as an ADC driver when used with a balun transformer. With
its wide bandwidth, the LTC6431-15 can cover the entire CATV band.
Z1
Figure 5. Adding matching networks
to the input and output
Z2
Z1
INPUT
MATCH
Z2
OUTPUT
MATCH
INPUT
Z = 50Ω
OUTPUT
Z = 50Ω
TRADITIONAL
RF AMPLIFIER
f = 240MHz
a nonlinear amplifier, the result is a far
more complicated mix of the two desired
tones and a multitude of unwanted tones,
including harmonics of the two tones, the
sum and difference of the two input tones,
and other intermodulation products.
Figure 6. Single-ended IF amplifier
Intermodulation (IM3) products
(2f1 – f2 and 2f2 – f1) are a subset of these
unwanted tones and they are particularly
onerous. IM3 products can fall very close
to the intended signal’s frequency, making them nearly impossible to filter out.
5V
VCC = 5V
1000pF
Noise (characterized by NF) limits an
amplifier’s sensitivity at low input signal
amplitudes, while linearity (characterized
by OIP3) limits sensitivity at high input
amplitudes. Taken together, these two
metrics, NF and OIP3, define the amplifier’s useful dynamic range for a signal.
RF
CHOKE,
560nH
1000pF
LTC6431-15
RSOURCE
50Ω
Amplifier linearity is most often characterized by the 3rd order output
intercept point (OIP3)—the hypothetical point where the power of the IM3
products intersects the fundamental
power (Figure 3). The LTC6431-15 exhibits very small IM3 products and thus its
OIP3 is very good. Minimizing the IM3
product is especially important when a
blocker (interferer) or an adjacent channel is nearby. Figure 3 shows that IM3
products grow three times faster than the
desired tones. This limits the acceptable
output power, and therefore the input
power, that the amplifier can handle
without distorting the desired signal.
RLOAD
50Ω
20
54
15
50
10
MAGNITUDE (dB)
OIP3 (dBm)
46
42
38
34
0
–5
–10
–15
–20
30
26
S PARAMETER
S11
S21
S12
S22
5
–25
0
200
400
600
FREQUENCY (MHz)
800
LTC6431-15 OIP3 vs frequency
1000
–30
0
0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8
FREQUENCY (GHz)
2
LTC6431-15 S parameter vs frequency
Figure 7. LTC6431-15 100MHz–1700MHz single-ended evaluation circuit and performance
October 2013 : LT Journal of Analog Innovation | 5
The LTC6430-15 excels as an ADC driver for high speed, high resolution
ADCs. The challenge in these applications is to drive the unbuffered ADC
inputs to their required input voltage levels while preserving the signal-tonoise ratio (SNR) and spurious free dynamic range (SFDR) of the ADC.
Figure 8. Simplified schematic of
wideband differential 14-bit ADC driver
560nH
0402AF
60pF
GUANELLA
BALUN
1:1
1nF
VCC = 5V
150Ω
HIGH LINEARITY SOLVES THE
TOUGHEST COMMUNICATION
PROBLEMS
VCM
5V
49.9Ω
350Ω
•
•
The LTC6431-15 boasts a typical OIP3 of
47dBm at 240MHz —essentially hammering
the IM3 products into the noise floor so
that they don’t interfere with the intended
signals (Figure 4). Not to be outdone, the
LTC6430-15 features an OIP3 of 50dBm
at 240MHz. Both amplifiers offer a very
wide dynamic range when combined with
their 3.3dB NFs —addressing the high data
rate challenge by maintaining high fidelity at both high and low signal levels.
1nF
100nH
0402CS
LTC6430-15
LTC2158
200ps
EASY TO INSERT
Implementing an RF/IF gain stage has
not always been easy. Traditionally, the
designer must first consider circuit biasing. The LTC6431-15 has an internal bias
circuit that requires only 90m A from a
single 5V supply, while the LTC6430-15
draws 160m A from a single 5V supply.
Figure 9. LTC6430-15 driver
and LTC2158-14, dual
14-bit ADC combination
evaluation circuit
1000pF
60pF
0.1µF
1000pF
DNC
GND
DNC
T_DIODE
LTC6430-15
DNC
120nH
0402CS
–OUT
1000pF
560nH
348Ω
BALUN = MaCom 1:1 TRANSFORMER MABA-007159
6 | October 2013 : LT Journal of Analog Innovation
1000pF
AIN+
LTC2158-14
AIN–
DNC
DNC
–IN
GND
DNC
60pF
1000pF
1000pF
VCM
DNC
DNC
DNC
–IN
0.1µF
49.9Ω
+OUT
VCC
100Ω
DIFFERENTIAL
560nH
DNC
DNC
VCC
DNC
DNC
GND
+IN
1:1
BALUN
+IN
348Ω
GND
49.9Ω
The internal bias circuit optimizes the
device operating point for maximum linearity. A temperature compensation circuit
maintains performance over environmental changes and prevents current runaway at high temperature. These devices
also include an internal voltage regulator
to minimize performance changes due
to imperfections in the power supply.
0.1µF
VCC = 5V
GND
An RF/IF amplifier must also be impedance
matched at the input and the output to
maximize power transfer and minimize
reflections. This is traditionally a timeconsuming iterative task. Typically the
designer must add input and output networks to match the amplifier impedance
design features
Figure 10. 500MHz single
tone SFDR and SNR of
LTC6430-15 and LTC2158
driver/ADC combo board
(SNR = 61.5dB, SFDR =
75.7dB)
to the system impedance, normally
50Ω (Figure 5). These matching networks
in turn alter the amplifier’s NF and OIP3—
often compromising the NF and OIP3 to
achieve a reasonable impedance match.
The LTC6431-15 and LTC6430-15 amplifiers
internally match their input and output
impedance over the 20MHz –1700MHz
band, simplifying design while preserving their NF and OIP3. The single ended
LTC6431-15 is internally input and output
matched to 50W, whereas the LTC6430-15
is internally matched to 100W differential impedance at the input and the
output. This is allows the devices to be
easily inserted into various applications
without additional matching elements.
Figure 11. 500MHz 2-tone
measurement of IM3
products of LTC6430-15
and LTC2158 driver/ADC
combo board (IM3 low
= –101dBfs, IM3 high =
–102dBfs)
GUARANTEED STABILITY AND
PERFORMANCE
The LTC6431-15 and LTC6430-15 are
unconditionally stable when implemented with our applications circuits.
A-grade versions of the LTC6431-15
are individually characterized for OIP3
at 240MHz, guaranteeing a minimum
OIP3 of 44dBm. Similarly, A-grade versions of the LTC6430-15 are individually
characterized for OIP3 at 240MHz, guaranteeing a minimum OIP3 of 47dBm.
Table 1. Summary of results over frequency for ADC driver evaluation circuit
LTC6430/LT2158 COMBO CIRCUIT
LT2158 ADC ALONE
FREQ.(MHz)
1M
SFDR
SNR
1M
SFDR
SNR
250
–87
73.8
63.1
–95
78
66.5
300
–86
77.5
62.8
–94
78
65.5
A NEW BREED OF RF AMPLIFIER
400
–87
75.0
62.3
–92
78
64.5
Linear Technology has a long history of
producing superior op amp style amplifiers that handle low frequency signals
with minimal noise and distortion. While
the LTC6431-15 and LTC6430-15 are not
capable of amplifying DC signals like an
op amp, they are capable of amplifying
500
–101
75.7
61.5
–84
70
63.0
600
–88
72.0
60.7
–88
62.5
62.5
700
–92
67.5
60.0
–86
62.0
61.0
800
–94
84.0
59.5
–85
61.5
60.0
900
–82
73.0
58.6
–80
61.0
59.0
1000
–85
61.4
58.1
–83
60.5
58.0
October 2013 : LT Journal of Analog Innovation | 7
Using an appropriate pair of 2:1 balun transformers, the LTC6430-15 provides wideband
amplification with low noise and low distortion. In this balanced configuration, the amplifier is
matched to 50Ω at the input and output. The balanced configuration also has the advantage
of suppressing 2nd order distortion which is critical in multi-octave wideband applications.
BALUN_A = ADT2-1T FOR 50MHz TO 300MHz
BALUN_A = ADT2-1P FOR 300MHz TO 400MHz
BALUN_A = ADTL2-18 FOR 400MHz TO 1300MHz
ALL ARE MINI-CIRCUITS CD542 FOOTPRINT
signals up to 2GHz. Op amps typically
struggle to operate above 200MHz.
With an op amp, feedback typically needs
to be added to set the gain. Increasing the
gain of a voltage feedback op amp further
decreases its operational bandwidth. On
the other hand, our RF style amplifiers
offer a fixed power gain of 15dB. The
RF solution lacks the versatility of gain
adjustment, but the usable bandwidth far
exceeds that attainable from an op amp.
Op amps are designed to drive high
impedance loads, while the LTC6430/31
amplifiers can drive a 50Ω load and deliver
real power over a wide frequency range
(20MHz –1700MHz). Unlike an op amp,
this RF-focused design does not require
termination resistors at the input nor
at the output, as impedance matching
is done internally. Termination resistors at the input add noise and termination resistors at the output attenuate the
power delivered to the load. Therefore,
DNC
DNC
–OUT
R2
350Ω
C5
1000pF
OPTIONAL STABILITY
NETWORK
the RF amplifier solution results in better
overall noise and linearity. The LTC6430-15
and LTC6431-15 amplifiers offer a superior
solution for AC signal applications that
do not require DC-coupled performance.
LTC6431-15 SINGLE-ENDED 50Ω
AMPLIFIER
The single-ended LTC6431-15 is an ideal
solution for a number of applications.
It excels as an IF amplifier to overcome
filter losses, or as an ADC driver when
used with a balun transformer. With
its wide bandwidth, the LTC6431-15
can cover the entire CATV band.
Figure 6 shows a single-ended
IF amplifier, while Figure 7 shows an
LTC6431-15 100MHz –1700MHz evaluation board and performance.
DNC
GND
DNC
–IN
C2
1000pF
• •
100Ω
DIFFERENTIAL
C4
1000pF
BALUN_A
DNC
DNC
C3
1000pF
T2
2:1
T_DIODE
LTC6430-15
DNC
C8
60pF
VCC
GND
DNC
BALUN_A
8 | October 2013 : LT Journal of Analog Innovation
+OUT
DNC
DNC
100Ω
DIFFERENTIAL
RFIN
50Ω, SMA
L1
560nH
DNC
DNC
T1
1:2
VCC
PORT
INPUT
+IN
R1
350Ω
DNC
C7
60pF
GND
C1
1000pF
GND
Figure 12. 50Ω input/output balanced amplifier
PORT
OUTPUT
RFOUT
50Ω, SMA
L2
560nH
C6
0.1µF
VCC = 5V
LTC6430-15 DIFFERENTIAL
APPLICATIONS
The differentially configured inputs and
outputs of the LTC6430-15 lend themselves to a variety of system applications. In the following examples, the
LTC6430-15 linearity, low noise and
wideband performance are put to the test.
In the first example, its differential
outputs mate well to the differential
inputs of an ADC. The LTC6430-15 is
internally input/output matched to
100W differential impedance. 100W is a
convenient impedance for driving high
speed ADCs. Next, using 2:1 balun transformers in a balanced configuration, the
LTC6430-15 delivers wideband amplification with low distortion into 50W.
Finally, using 1.33:1 balun transformers, the LTC6430-15 can be matched to a
75W system to deliver wideband amplification across the entire CATV band.
design features
A single balun cannot cover the entire LTC6430-15 band of operation. Linear
offers several evaluation circuits that cover the amplifier’s intended bandwidth.
Conveniently transformed to 50Ω at the input and output(s) for ease of bench
characterization, these evaluation circuits also demonstrate the performance of the
LTC6430-15 when used in a purely differential application without the baluns.
Figure 13. Evaluation circuit of balanced amplifier shown in Figure 12: 50MHz–300MHz (ADT2-1T baluns)
54
15
10
50
MAGNITUDE (dB)
OIP3 (dBm)
46
42
38
34
0
–5
–10
–15
–20
30
26
S PARAMETER
S11
S21
S12
S22
5
–25
0
100
200
300
FREQUENCY (MHz)
400
–30
500
0
100
200 300 400 500
FREQUENCY (MHz)
600
700
Figure 14. Evaluation circuit of balanced amplifier shown in Figure 12: 300MHz–1100MHz (ADTL2 baluns)
20
50
15
46
10
MAGNITUDE (dB)
OIP3 (dBm)
42
38
34
0
–5
–10
–15
–20
30
26
S PARAMETER
S11
S21
S12
S22
5
–25
0
200
400
600
FREQUENCY (MHz)
800
–30
1000
0
0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8
FREQUENCY (GHz)
2
Figure 15. Evaluation circuit of balanced amplifier shown in Figure 12: 200MHz–1500MHz (TCM2-43X baluns)
50
15
10
46
5
MAGNITUDE (dB)
OIP3 (dBm)
42
38
34
–5
–10
–15
–20
30
26
S PARAMETER
S11
S21
S12
S22
0
–25
0
0.25
1
0.5
0.75
FREQUENCY (GHz)
1.25
1.5
–30
0
100
200 300 400 500
FREQUENCY (MHz)
600
700
October 2013 : LT Journal of Analog Innovation | 9
Cable TV offers unique challenges for an amplifier. A high channel count requires
excellent 3rd order linearity and due to the multiple octave environment, 2nd order
products must be suppressed as well. The LTC6430-15 meets these challenges using
a pair of 1.33:1 baluns to transform its inherent 100Ω differential impedance to 75Ω.
ADC Driver
Table 1 displays minimal degradation
of SNR and SFDR for this high speed,
high resolution ADC. The LTC6430-15’s
high linearity (Figures 10 and 11) and
low noise allow the designer to drive
the ADC with minimal filtering at the
ADC input. All measurements are taken
from a single application circuit without adjusting the matching networks.
This highlights the LTC6430-15 wide
bandwidth and linearity performance.
Balanced Amplifier Drives 50Ω Loads
Using an appropriate pair of 2:1 balun
transformers, the LTC6430-15 provides
wideband amplification with low noise
and low distortion (Figure 12). In this
balanced configuration, the amplifier is
matched to 50Ω at the input and output. The balanced configuration also
has the advantage of suppressing 2nd
order distortion which is critical in
multi-octave wideband applications.
10 | October 2013 : LT Journal of Analog Innovation
DNC
DNC
VCC
GND
DNC
•
100Ω
DIFFERENTIAL
C4
0.047µF
DNC
–OUT
DNC
GND
DNC
MINI-CIRCUITS 1:1.33
T2
1.33:1
DNC
DNC
BALUN_A = TC1.33-282+
FOR 50MHz TO 1000MHz
C3
0.047µF
T_DIODE
LTC6430-15
DNC
C2
0.047µF
DNC
+OUT
DNC
DNC
BALUN_A
GND
+IN
100Ω
DIFFERENTIAL
L1
560nH
DNC
VCC
RFIN
75Ω,
CONNECTOR
T1
1:1.33
GND
PORT
INPUT
–IN
The LTC6430-15 excels as an ADC driver for
high speed, high resolution ADCs (Figure 8).
The challenge in these applications is to
drive the unbuffered ADC inputs to their
required input voltage levels while preserving the signal-to-noise ratio (SNR) and
spurious free dynamic range (SFDR) of the
ADC. As shown by the performance results
for the evaluation circuit in Figure 9, the
LTC6430-15 is able to drive the LTC2158
(dual 14-bit, 310Msps ADC) over its full
input bandwidth with very little degradation in SFDR and SNR (Figure 10).
C1
0.047µF
C5
1000pF
•
BALUN_A
PORT
OUTPUT
RFOUT
75Ω,
CONNECTOR
L2
560nH
C6
0.1µF
VCC = 5V
Figure 16. 50MHz to 1000MHz CATV push-pull amplifier with 75Ω input and 75Ω output
Unfortunately, a single balun cannot cover
the entire LTC6430-15 band of operation.
Linear Technology offers a number of
evaluation circuits that cover the amplifier’s intended bandwidth (Figures 13–15).
Conveniently transformed to 50Ω at the
input and output(s) for ease of bench
characterization, these evaluation circuits
also demonstrate the performance of the
LTC6430-15 when used in a purely differential application without the baluns.
The results reveal the importance of
selecting the correct balun transformer
for the frequency of interest. Due to their
limited bandwidth, the balun transformers limit the LTC6430-15 performance.
Together, these three balanced circuits
demonstrate the linearity and wide bandwidth attainable with the LTC6430-15.
CATV Application
A CATV application circuit is the final
example of the LTC6430-15’s versatility (Figure 16). Cable TV offers unique
challenges for an amplifier. Often the
required frequency band covers more
than four octaves and the amplifier must
have flat gain and impedance matching
to a 75Ω environment. A high channel
count requires excellent 3rd order linearity
and due to the multiple octave environment, 2nd order products must be suppressed as well. The LTC6430-15 meets
these challenges using a pair of 1.33:1
baluns to transform its inherent 100Ω differential impedance to 75Ω (Figure 17).
Given its low noise, low 2nd and 3rd
order distortion, and flat gain, this circuit
can handle CATV demands while consuming only 800mW from a 5V supply.
design features
The LTC6431-15 and LTC6430-15 are manufactured using a high performance
SiGe BiCMOS process, compared to other RF gain blocks manufactured using
GaAs transistors. Using a silicon-based process yields better reproducibility over
comparable GaAs processes. A BiCMOS process also allows Linear to integrate
distortion cancellation, bias control and voltage regulator functions into the devices.
SILICON-BASED PROCESS FOR
BETTER REPRODUCIBILITY
The LTC6431-15 and LTC6430-15 are
manufactured using a high performance
SiGe BiCMOS process, compared to other
RF gain blocks manufactured using Ga As
transistors. Using a silicon-based process
yields better reproducibility over comparable Ga As processes. A BiCMOS process
also allows Linear to integrate distortion
cancellation, bias control and voltage
regulator functions into the devices.
Figure 17. LTC6430-15
50MHz–1000MHz CATV
evaluation circuit and
performance results
CONCLUSION
50
15
10
46
MAGNITUDE (dB)
OIP3 (dBm)
38
34
–10
–15
–25
0
0
200
400
600
FREQUENCY (MHz)
800
–30
1000
–20
NOISE FIGURE (dB)
–40
–50
–60
–70
–80
–90
HD2 AVG
HD3 AVG
0
200
400
600
FREQUENCY (MHz)
0.25
1
0.5
0.75
FREQUENCY (GHz)
1.25
1.5
5
–30
–100
0
6
VCC = 5V
T = 25°C
POUT = 8dBm/TONE
–10
HD2 & HD3 (dBc)
0
–5
–20
30
–110
S PARAMETER
S11
S21
S12
S22
5
42
26
To meet the demands of modern communications standards, and simplify
RF/IF designs, the LTC6431-15 and the
LTC6430-15 achieve best-in-class noise
and linearity at the lowest DC power
dissipation. They are easy to use, versatile, and guarantee performance
over a wide range of conditions. n
800
1000
4
3
2
VCC = 5V
T = 25°C
INCLUDES BALUN LOSS
1
0
0
200
400
600
FREQUENCY (MHz)
800
1000
October 2013 : LT Journal of Analog Innovation | 11
LED Driver with Integrated Spread Spectrum
Reduces EMI without Adding Flicker
Keith Szolusha
Automotive LED drivers should be compact, efficient and support flicker-free PWM
dimming. They should not produce significant conducted EMI at and around the AM
radio band. Unfortunately, low EMI is not in the nature of high power switch mode power
supplies—the constant switching frequency produces a significant EMI signature at a
number of frequencies, including the power supply’s fundamental operating frequency
and its harmonics. Odds are good that something will fall into the AM band.
One way to minimize EMI peaks is to
allow the switch mode power supply
(SMPS) operating frequency to cover a
range of values, namely spread spectrum
switching. The desired effect of spread
spectrum switching is to push down
VIN
8V TO 60V
100V TRANSIENT
L2
4.7µH
2.2µF
100V
OPTIONAL
EMI FILTER
the EMI peaks that would occur at the
SMPS fundamental operating frequency
and harmonics, spreading the EMI energy
over a range of frequencies instead.
RINSNS
12mΩ
5A MAXIMUM
CIN2
4.7µF
×2
100V
L1
22µH
1M
115k EN/UVLO V
IN
IVINP
IVINN
GATE
12.4k
10Ω
M1 13.3k
SENSE
RSNS
15mΩ
LT3795
GND
VREF
CTRL2
FB
10k
RLED
620mΩ
SS
ISN
0.1µF
OPENLED
OPENLED
SHORTLED
SHORTLED
100k
ISMON
INTVCC
PWM DIM
D2
LED
CURRENT
REPORTING
0.1µF
INTVCC
4.7µF
D1: DIODES INC PDS5100H
D2: VISHAY ES1B
L1: COOPER HC9-220-R
L2: WÜRTH 744071047 4.7µH
M1: RENESAS RJK1051DPB 100V
M2: VISHAY Si7113DN 100V
M2
TG
PWM
IVINCOMP
VC
RC
4.7k
CC
10nF
RT
RAMP
RT
17.4k
450kHz–300kHz
10nF
6.8nF
~1kHz
TRIANGLE
SPREAD
SPECTRUM
MODULATION
Figure 1. 80V, 400mA automotive LED driver with internal spread spectrum for low EMI
12 | October 2013 : LT Journal of Analog Innovation
To this end, the LT3795 generates its own
spread spectrum ramp signal and aligns
it with the lower frequency PWM dimming input with a patent pending technique. This eliminates the chance that
the spread spectrum frequency could
combine with the PWM signal to produce visible flicker in the LEDs —even
at the highest PWM dimming ratio.
HIGH POWER LED DRIVER
ISP
CTRL1
100k
COUT
2.2µF
×4
100V
499k
OVLO
ANALOG
DIMMING
CONTROL
D1
LED driver SPMSs have an additional
requirement: the frequency spreading should also be synchronized
with the PWM dimming (brightness
control) frequency to ensure that
there is no resulting LED flicker.
80V LED
400mA
The LT3795 is a high power LED driver that
uses the same high performance PWM dimming scheme as the LT3756/LT3796 family, but with the additional feature of
the internal spread spectrum ramp for
reduced EMI. It is a 4.5V-to-110V input to
0V-to-110V output single-switch controller
IC that can be configured as a boost, SEPIC,
buck-boost mode or buck mode LED driver.
It features a 100kHz to 1MHz switching
frequency range, open LED protection,
short-circuit protection, and can also be
operated as a constant voltage regulator
with current limit or as a constant current
SLA battery or supercapacitor charger.
design features
100
100
90
90
80
70
CISPR25 CLASS 5
(AM RADIO BAND)
LT3795
SPREAD SPECTRUM
ENABLED
60
50
40
30
20
10
0
SPREAD SPECTRUM DISABLED
0.5 0.7 0.9 1.1 1.3 1.5 1.7 1.9 2.1 2.3 2.5
FREQUENCY (MHz)
PEAK AMPLITUDE (dBµV)
PEAK AMPLITUDE (dBµV)
The LT3795 generates its own spread spectrum ramp signal and aligns it with the
lower frequency PWM dimming input with a patent pending technique. This eliminates
the chance that the spread spectrum frequency could combine with the PWM signal
to produce visible flicker in the LEDs—even at the highest PWM dimming ratio.
80
70
CISPR25 CLASS 5
(MW, AM)
(SW)
60
LT3795
SPREAD SPECTRUM
ENABLED
(CB)
50
40
30
20
10
0
150kHz
SPREAD SPECTRUM DISABLED
30MHz
FREQUENCY
Figure 2. Conducted peak EMI around the AM band
is reduced by 3dBµV–6dBµV when the LT3795’s
spread spectrum frequency modulation is used.
The CISPR25 Class 5 AM-band limit is provided for
reference.
Figure 3. Spectrum analyzer scan of the LT3795
150kHz–30MHz peak conducted EMI shows the
reduction in peak EMI over a broad frequency range.
Figure 1 shows a 92% high efficiency
80V, 400m A, 300kHz-450kHz automotive LED headlamp driver with
spread spectrum frequency modulation and short-circuit protection.
this reason, in many high end LED driver
applications, implementing spread
spectrum is not trivial. Without spread
spectrum, designers must rely upon bulky
EMI filters, gate resistors that slow down
switching edges (but reduce efficiency) and
snubbers on the switch and catch diode.
INTERNAL SPREAD SPECTRUM
SIMPLIFIES DESIGN
Unlike other high power LED drivers,
the LT3795 generates its own spread
spectrum ramp to produce 30% switching frequency modulation below the
programmed switching frequency. This
lowers its conducted EMI peaks, reducing
the need for costly and bulky EMI input
filter capacitors and inductors.
Using an external, or separate, spread
spectrum clock to produce the switching
frequency in an LED driver can produce
visible flicker during PWM dimming since
the spread spectrum frequency pattern is
not synchronized with the PWM period. For
Figure 2 shows a comparison of the
conducted EMI measurements of the
LT3795 LED driver around the AM band
when spread spectrum is enabled and
disabled. Normal (non-spread spectrum)
operation yields high energy peaks at
the switching frequency and its harmonics. These peaks can prevent the design
from passing stringent EMI requirements in EMI sensitive applications
such as automobiles. For reference,
the CISPR 25 class 5 automotive conducted EMI limits are shown in Figure 2.
Figure 3 shows the effect of spread spectrum over a wider frequency band.
Since there is no limit between 300kHz
and 580kHz, that is an excellent place
for the fundamental frequency to be
placed. In this application it is placed
at 450kHz and spread down to 300kHz.
Spread spectrum can be disabled by
simply grounding the RAMP pin.
Figure 4. Spread spectrum as implemented in the LT3795 has no discernable effect on LED brightness. The
1kHz spread spectrum sweep set in Figure 1 has a negligible effect on LED ripple current (b) when compared
to no spread spectrum (a) and is much too high a frequency to be detected by the human eye as flicker.
(a)
ILED
50mA/DIV
(AC-COUPLED)
(b)
ILED
50mA/DIV
(AC-COUPLED)
VRAMP
1V/DIV
VRAMP
1V/DIV
1ms/DIV
1ms/DIV
October 2013 : LT Journal of Analog Innovation | 13
The 6.8nF capacitor at the RAMP pin sets
the spread spectrum frequency modulation rate to a 1kHz triangle—that is,
the LT3795’s operating frequeny sweeps
from 300kHz to 450kHz and back every
millisecond. The addition of the triangular 1kHz spread spectrum signal
has a negligible effect on LED ripple
current, as shown in Figure 4.
The modulation frequency of 1kHz is chosen because it is low enough to be within
the LT3795’s bandwidth, yet high enough to
signicantly attenuate AM-band conducted
EMI peaks. Further reducing the modulation frequency degrades peak attenuation
in the AM band, where it may be most
important for classification. The choice of
spread spectrum modulation frequency
does not appear to affect EMI peak attenuation at higher frequencies. Nothing above
100Hz is perceived by the human eye.
FLICKER-FREE PWM DIMMING
It is possible to reduce EMI with a spread
spectrum source that is not synchronized with the PWM signal, but the
beat of the switching frequency and
PWM signal can produce visible flicker
in the LED. The spread spectrum ramp
generated inside the LT3795 synchronizes itself with the PWM period when
PWM dimming is used. This provides
repeatable, flicker-free PWM dimming,
even at high dimming ratios of 1000:1.
Figure 5 compares the PWM dimming
current waveforms of two spread spectrum solutions: one with the LT3795’s
patent-pending spread-spectrum-toPWM synchronization technique, and one
without. Both captures are produced with
infinite persist, showing an overlay of
a number of cycles of a 1% PWM dimming waveform. Figure 5(a) shows the
result of LT3795’s spread spectrum
operation on the PWM LED current. The
waveform is consistent cycle-to-cycle,
which results in flicker-free operation. Figure 5(b) shows the results of a
14 | October 2013 : LT Journal of Analog Innovation
ILED
200mA/DIV
ILED
200mA/DIV
IL1
2A/DIV
IL1
2A/DIV
20µs/DIV
INFINITE PERSIST CAPTURE
120Hz PWM DIMMING WITH 1% DUTY CYCLE
20µs/DIV
INFINITE PERSIST CAPTURE
120Hz PWM DIMMING WITH 1% DUTY CYCLE
(a)
(b)
Figure 5. Comparison of two spread spectrum LED driver solutions and the effect on PWM dimming.
The infinite persist scope captures show repeated and overlaid PWM LED current waveforms. In (a), the
patent-pending spread spectrum technique of the LT3795 produces consistent cycle-to-cycle LED PWM
on-time shape. The result is flicker-free operation at high dimming ratios. The waveforms in (b) show a
comparable, non-LT3795, spread spectrum LED driver result. In this case, without the spread-spectrum-toPWM synchronization of the LT3795, the LED current waveform is inconsistent cycle-to-cycle, producing
perceivable flicker at high PWM dimming ratios.
comparable, non-LT3795, spread spectrum
solution. The cycle-to-cycle variation
in on-time shape produces variation in
average LED current, which can be seen
as LED flicker at high dimming ratios.
Note that spread spectrum driver ICs
without the LT3795’s patented technique
might produce a clean spread spectrum
EMI reduction result, the flicker may still
be present. One has to observe the LEDs
or the LED current waveform to understand if flicker is present. In the case of the
LT3795, both the conducted EMI scan and
the scope shot of LED current are good.
SHORT-CIRCUIT PROOF BOOST
The LT3795 boost LED driver shown in
Figure 1 is short-circuit proof. The high
side PMOS disconnect is not only used
for PWM dimming, but also for shortcircuit protection when the LED+ terminal
is shorted to ground. Unique internal
circuitry monitors when the output
current is too high and the LED+ voltage is too low, turns off the disconnect
PMOS and reports a short LED fault.
Similarly, if the LED string is removed or
opened, the IC limits its maximum output
voltage and reports an open LED fault.
MULTITOPOLOGY SOLUTION
The LT3795 can be used to drive LEDs in
a boost setup as shown here, or it can be
used in buck mode, buck-boost mode,
SEPIC and flyback topologies when the
relationship of the LED string voltage and
input voltage ranges requires it. All topologies feature the same spread spectrum
and short-circuit protection. The LT3795
can even be configured as a constant
boost or SEPIC voltage regulator with
spread spectrum frequency modulation.
CONCLUSION
The LT3795 is a 110V, versatile LED driver
IC with built-in spread spectrum frequency
modulation to reduce EMI. This simplifies
the design of LED applications that must
pass stringent EMI testing. Spread spectrum
requires only a single capacitor, and unlike
external-clock-based spread spectrum
solutions, produces flicker-free LED operation during PWM dimming. Short-circuit
protection is available in all topologies,
making this IC a robust and powerful
solution for driving automotive LEDs. n
design features
15V Buck-Boost Converters with Ultralow 1.3µA Quiescent
Current are Tailored to Micropower Applications and the
Internet of Things
Dave Salerno
The proliferation of wireless sensors supporting the “Internet of Things” has increased
the need for small, efficient power converters tailored to untethered low power devices.
The new LTC3129 and LTC3129-1 are designed to satisfy this need. The LTC3129
and LTC3129-1 are monolithic buck-boost DC/DC converters with an input voltage
range of 2.42V to 15V. The LTC3129 has an output voltage range of 1.4V to 15.75V,
while the LTC3129-1 offers eight pin-selectable fixed output voltages between 1.8V
and 15V. Both parts can supply a minimum output current of 200mA in buck mode.
Low power sensors can take advantage
the LTC3129’s and LTC3129-1’s zero current
when disabled (on both VIN and VOUT),
and a quiescent current on VIN of just
1.3µ A when power saving Burst Mode®
operation is selected, making them ideal
for µ Power and energy harvesting applications, where high efficiency at extremely
light loads is crucial. Their buck-boost
architecture makes them well suited
to a wide variety of power sources.
PWM mode, an accurate RUN pin threshold to allow the UVLO threshold to be
programmed, a power good output and
an MPPC (maximum power point control)
function for optimizing power transfer
when operating from photovoltaic cells.
The compact 3mm × 3mm QFN package and the high level of integration ease
the LTC3129/LTC3129-1’s placement into
space-constrained applications. Only a
few external components and an inductor, which can be as small as 2mm × 3mm,
are required to complete the power
supply design. Internal loop compensation further simplifies the design process.
Other key features of the LTC3129 and
LTC3129-1 include a fixed 1.2MHz operating frequency, current mode control,
internal loop compensation, automatic
Burst Mode operation or low noise
Figure 1. 3.3V solar powered converter
operates from indoor light
22nF
VOC = 5V
UVLO = 3.5V
VIN
+
CIN
470µF
6.3V
BST1 SW1
4.7µF
SW2 BST2
VOUT
3.3V
VOUT
LTC3129-1
22µF
RUN
VCC
MPPC
VS2
2.26M
VCC
VS3
PWM
GND
PGND
•The RUN pin must exceed 1.22V (typical).
•The VIN pin must exceed 1.9V (typical).
•VCC (which is internally generated from
VIN but can also be supplied externally) must exceed 2.25V (typical).
PGOOD
VS1
10pF
The circuit in Figure 1 exploits the unique
ability of the LTC3129 and LTC3129‑1 to
start up and operate from an input power
source as weak as 7.5 microwatts—making them capable of operating from small
(less than 1in2), low cost solar cells with
indoor light levels less than 200-lux.
This enables such applications as indoor
light powered wireless sensors, where
the DC/DC converter must support an
extremely low average power requirement, due to a low duty cycle of operation, from very low available power, while
consuming as little power as possible.
To make this low current start-up possible,
the LTC3129 and LTC3129-1 draw a meager
two microamps of current (less in shutdown) until three conditions are satisfied:
22nF
VIN
4.22M
PV PANEL
SANYO
AM-1815
4.9cm × 5.8cm
L1
4.7µH
3.3V CONVERTER OPERATES FROM
INDOOR LIGHT USING A SMALL
SOLAR CELL
2.2µF
L1: Toko DEM2812C
Until all three of these conditions are satisfied, the part remains in a “soft-shutdown”
or standby state, drawing just 2µ A.
October 2013 : LT Journal of Analog Innovation | 15
The LTC3129 and LTC3129‑1 can start up and operate from an input power
source as weak as 7.5 microwatts—making them capable of operating from
small (less than 1in2), low cost solar cells with indoor light levels less than 200-lux.
This enables such applications as indoor light powered wireless sensors.
Figure 2. Solar powered converter
with coin cell backup
UVLO = 3.5V
VIN
L1
4.7µH
22nF
4.7µF
BST1 SW1
22nF
SW2 BST2
VOUT
VIN
LTC3129
4.22M
PV PANEL
SANYO AM-1815
VCC
6.3V
4.22M
S1
D1
2.43M
G1
D2
G2
VOUT
2.43M
MPPC
VOUT
3V TO 3.2V
S2
2.2µF
10pF
FB
CR2032
3V COIN CELL
PGOOD
2.26M
10pF
3.20V
22µF
RUN
+ 470µF
FDC6312P
DUAL PMOS
BAT54
VCC
PWM
GND
PGND
74LVC2G04
2.2µF
L1: Toko DEM2812C
This allows a weak input source to
charge the input storage capacitor until
the voltage is high enough to satisfy all
three previously mentioned conditions,
at which point the LTC3129/LTC3129-1
begins switching, and VOUT rises to regulation, provided the input capacitor has
sufficient stored energy. The input voltage at which the part exits UVLO can be
set anywhere from 2.4V to 15V using the
external resistive divider on the RUN pin.
With a RUN pin current of less than
1n A typical, high value resistors may be
used to minimize current draw on VIN .
In the application example shown in
Figure 1, the energy stored on CIN is
used to bring VOUT into regulation once
the converter starts. If the average
power demand on VOUT is less than the
power delivered by the solar cell, the
LTC3129/LTC3129-1 remains in Burst Mode
operation, and VOUT remains in regulation.
If the average output power demand
exceeds the input power available, then
VIN drops until UVLO is reached, at which
16 | October 2013 : LT Journal of Analog Innovation
point the converter reenters soft-shutdown. At this point, VIN begins recharging, allowing the cycle to repeat. In this
hiccup mode of operation, VIN is positioned hysteretically about the UVLO point,
with a VIN ripple of approximately
290mV in this example. This ripple is set
by the 100mV hysteresis at the RUN pin,
gained up by the UVLO divider ratio.
Note that by setting the converter’s
UVLO voltage to the MPP (maximum
power point) voltage for the chosen solar
cell (typically between 70% to 80% of
the open-circuit voltage), the cell always
operates near its maximum power transfer
voltage (unless the average load requirement is less than the power output of
the solar cell, in which case VIN climbs
and remains above the UVLO voltage).
To further optimize efficiency and eliminate unnecessary loading of VOUT, the
LTC3129/LTC3129-1 does not draw any
current from VOUT during soft-start or
at any time if Burst Mode operation is
selected. This prevents the converter from
discharging VOUT during soft-start, thereby
preserving charge on the output capacitor. In fact, when the LTC3129 is sleeping,
there is no current draw at all on VOUT. In
the case of the LTC3129-1, the VOUT current draw is sub-microamp, due to the
high resistance internal feedback divider.
ADDING A BATTERY BACKUP
In many solar powered applications, a
backup battery provides power when
solar power is insufficient. Figure 2 shows
an application where a primary lithium
coin cell and a few external components
have been added to the converter from
the previous example to provide backup
power to the output in the event that
the light source is unable to provide the
necessary power to maintain VOUT. The
LTC3129 is used in this case, allowing
VOUT to be programmed for 3.2V to better match the voltage of the coin cell.
In this example, the battery is used on
the output side of the converter, and the
LTC3129 is set to regulate VOUT slightly
design features
The LTC3129-1 can operate at high efficiency over a wide range of loads and input voltages,
with a minimal number of external components. The flexibility of running seamlessly from a
wide variety of power sources is an asset in critical field applications, such as military radios.
above the battery voltage. This assures
that there is no load on the battery whenever VOUT can be powered by the solar
input. In the event that VOUT droops due
to insufficient light to power the load,
the PGOOD output from the LTC3129 goes
low, switching the load from the converter output to the battery, thus holding VOUT at the battery voltage. During
this time, the converter’s input and
output capacitors are able to recharge
(if some light is available), enabling the
load to be periodically switched from
the battery back to the converter by the
PGOOD signal. In this manner the load is
powered by the solar input as much as
possible, and the battery is only used in
a time-shared manner, extending its life.
The diode connected from PGOOD to VCC is
used to hold PGOOD low during start-up,
before VCC (and therefore PGOOD) is valid.
CHOOSING WHERE TO PUT THE
BACKUP BATTERY
In the previous example, the backup
battery was placed on the output. For
light load applications, this has the
advantage of not exposing the battery—which may be a low capacity
battery with high internal resistance—to
relatively high converter start-up input
current bursts, causing significant battery droop and lossy internal power
dissipation, in turn reducing battery life.
The disadvantages of putting the backup
battery on the output of the converter
are that the battery voltage must be well
matched to the desired output voltage, and it must have a relatively flat
discharge curve so as to maintain reasonable regulation of VOUT. The 3V lithium
cell satisfies both of these requirements.
the VS1–VS3 pins, can be powered from a
5V USB input, a variety of battery options
or a 3V to 15V wall adapter. The flexibility
of running seamlessly from a wide variety
of power sources is an asset in critical
field applications, such as military radios.
Putting the backup battery on the input
side of the converter allows its voltage
to be different from the desired output
voltage, but it must be able to withstand
the higher currents that the converter
draws during start-up or load transients.
If used on the input side, a lithiumthionyl chloride battery is generally a
better choice for long life applications. It
can be diode-OR’d with the solar cell or
switched in and out with MOSFET switches,
in a similar manner to Figure 2.
The LTC3129-1’s low IQ of just 1.3µ A in
sleep mode, combined with a high resistance internal feedback divider, enables
it to maintain high efficiency over a wide
range of loads, as shown in Figure 4. At
a load current of just 100µ A, the efficiency is ~80% over nearly the entire
VIN range. This is an important feature
for extending battery life in applications that spend a large percentage
of the time in a low power state.
5V CONVERTER OPERATES
SEAMLESSLY FROM A VARIETY OF
INPUT SOURCES
The line step response (VIN is stepped
from 5V to 12V) is shown in Figure 5,
with VOUT measured under both heavy
and light load conditions. At a load of
200m A, the part is operating in PWM mode,
and VOUT overshoot is only 150mV (3%).
At a load of 10m A, the part is in Burst
Mode operation, with a burst ripple of
The ability of the LTC3129-1 to operate at
high efficiency over a wide range of loads
and input voltages with a minimal number
of external components is illustrated in
Figure 3. In this example, the output,
which has been programmed for 5V using
WALL ADAPTER
3V TO 15V
Figure 3. Multi-input 5V converter
L1
10µH
22nF
22nF
5V USB
VIN
1.8V TO 15V
BST1 SW1
SW2 BST2
VOUT
VIN
22µF
LTC3129-1
RUN
BATTERIES:
2–9 ALKALINE,
1–3 Li-ION,
OR Li-SOCl2
VCC
1M
PGOOD
MPPC
VOUT
5V
BAT54
OPTIONAL
PGOOD
VS1
10µF
VS2
VCC
VS3
PWM
GND
PGND
2.2µF
L1: Taiyo Yuden NR3015T
October 2013 : LT Journal of Analog Innovation | 17
The LTC3129 and LTC3129-1 include a maximum power point control (MPPC) feature
that allows the converter to servo VIN to a minimum voltage under load, as set by the
user. Regulating VIN maintains optimal power transfer in applications using higher current
solar cells or other sources with high internal resistance. This feature prevents the
converter from crashing the input voltage when operating from a current-limited source.
100
Burst Mode OPERATION
90
VOUT
200mV/DIV
(AC-COUPLED)
EFFICIENCY (%)
80
70
60
OUTDOOR SOLAR CONVERTER/
CHARGER WITH MPPC
ILOAD = 200mA (PWM MODE)
150mV
ILOAD = 10mA (Burst Mode OPERATION)
50
40
PWM
30
VIN = 2.5V
VIN = 3.6V
VIN = 5V
VIN = 10V
VIN = 15V
20
10
0
0.01
0.1
1
10
100
OUTPUT CURRENT (mA)
VOUT
200mV/DIV
(AC-COUPLED)
1000
Figure 4. Efficiency vs VIN and load of
the 5V converter in Figure 3
V
and less than 100mV of
VOUT overshoot due to the line step.
100m PK-PK (2%),
The VCC pin is the output of an internal
LDO that generates a nominal 3.9V from
VIN to power the IC. The LDO is designed
so that it can be externally back-driven up
to 5V. In this example, an optional bootstrap diode is shown from VOUT to VCC .
The addition of this external bootstrap
diode has two advantages. First, it
improves efficiency at low VIN and high
load current by providing a higher gate
drive voltage to the internal switches,
lowering their RDS(ON). Also, at high
VIN and light load, it improves efficiency by reducing the power lost in
the internal LDO used to generate VCC .
(Note that the VCC pin must not be
raised above 6V, so it cannot be diodeconnected to higher output voltages.)
100mV
VIN
5V/DIV
500µs/DIV
VIN = 5V TO 12V STEP
Figure 5. Line transient response of the 5V
converter in Figure 3
The MPPC control loop operates by
reducing the average inductor current commanded by the converter, thus
maintaining the minimum programmed
VIN voltage under load. This voltage is
set using an external resistor divider
connected to VIN and the MPPC pin, as
shown in the supercapacitor charging
example of Figure 6. The MPPC control
loop is designed to be stable with a
minimum input capacitance of 22µ F.
above its minimum value of 2.2V (by the
output voltage in this case), then the converter can operate at a lower input voltage,
down to 1.75V, where the fixed internal
VIN UVLO threshold is reached. This capability extends the usable voltage range
enough to make it possible to run from
two depleted alkaline batteries. Note that
if the battery voltage is below 2.4V and
the converter is shut down (or VOUT is
shorted), the IC is not be able to restart.
Figure 6. Outdoor solar powered supercapacitor
charger with maximum power point control
VMPPC = 6V
VIN
47µF
1M
The LTC3129 and LTC3129-1 include a
maximum power point control (MPPC)
feature that allows the converter to servo
VIN to a minimum voltage under load, as
set by the user. Regulating VIN maintains
optimal power transfer in applications
using higher current solar cells or other
sources with high internal resistance.
This feature prevents the converter from
crashing the input voltage when operating from a current-limited source.
L1
6.8µH
22nF
BST1 SW1
22nF
SW2 BST2
4.7µF
LTC3129
RUN
18 | October 2013 : LT Journal of Analog Innovation
2.8M
+
FB
MPPC
PowerFilm
MPT6-150
SOLAR
MODULE
1M
PGOOD
PWM
VCC
11.4cm × 15cm
The second advantage of adding a bootstrap diode is that it allows operation from
a lower VIN . After start-up, if VCC is held
VOUT
4.47V
VOUT
VIN
243k
GND
PGND
2.2µF
C1: Cooper Bussmann PB-5R0V105-R
L1: Coiltronics SD3118
C1
1F
5V
design features
The LTC3129 and LTC3129-1 monolithic buck-boost DC/DC converters offer
exceptional low power performance and power source flexibility demanded
by real-world wireless sensor and portable electronic instruments. The
ultralow 1.3µA quiescent current and high conversion efficiency can extend
battery lifetime indefinitely if used in concert with energy harvesting.
Note that reducing the inductor current under MPPC would cause the output
voltage to droop if it were driving a
conventional load. Therefore, most applications employing MPPC involve charging a large storage capacitor (or trickle
charging a battery) from a solar cell. The
MPPC feature assures that the capacitor or
battery is charged at the highest current
possible, while operating the solar cell
at its maximum power point voltage.
It is important to note that when the
LTC3129/LTC3129-1 is in MPPC control,
Burst Mode operation is inhibited, and
the VIN quiescent current is several milliamps, since the IC is switching continuously at 1.2MHz. Therefore, MPPC is not
appropriate for use with sources that
cannot supply a minimum of about
10m A. For applications requiring an
MPPC-like function with very weak input
sources, the accurate RUN pin should be
used to program a UVLO threshold, as
described in the example of Figure 1.
INTRINSIC SAFETY USING MPPC
INPUT CURRENT LIMIT USING MPPC
The MPPC feature can be used in other
applications, including those designed
for intrinsic safety, where the input
source has a series current limiting resistor between it and the DC/DC converter.
In this case, the MPPC loop prevents the
LTC3129/LTC3129-1 from drawing too
much current, especially during startup when the output capacitor is being
charged, and crashing the input voltage.
An example of this is shown in Figure 7,
where the input voltage is maintained at a
minimum of 3V, as set by the MPPC divider.
Note that the MPPC feature can be used to
set the maximum input current to a given
value. By choosing a series input resistor
value and setting the MPPC voltage to a
value below a fixed input source voltage,
the maximum input current is limited to:
In this case, because the input capacitor
value is limited to just 10µ F for safety
(less than the recommended minimum
value of 22µ F when using MPPC), an
additional RC compensation network
is added to the MPPC pin for improved
phase margin of the MPPC loop.
VMPPC = 3V
VIN
10Ω
L1
3.3µH
22nF
Figure 7. 3.3V Converter using MPPC
for intrinsic safety application
10µF
1.5M
BST1 SW1
1.5V
1.5V
RC
150k
CC
1nF
SW2 BST2
IOUT = 100mA
VOUT
VIN
10µF
LTC3129-1
VOUT
3.3V
VSOURCE − VMPPC
RSERIES
CONCLUSION
The LTC3129 and LTC3129-1 monolithic
buck-boost DC/DC converters offer exceptional low power performance and power
source flexibility demanded by real-world
wireless sensor and portable electronic
instruments. The ultralow 1.3µ A quiescent
current and high conversion efficiency
can extend battery lifetime indefinitely if
used in concert with energy harvesting.
A choice of maximum power point
control schemes allows optimization
of power performance over a wide
range of power sources. The expanding reach of wireless monitoring applications demands easy to use, efficient
and flexible DC/DC power converter
solutions. The LTC3129 and LTC3129-1
are ready to meet this challenge. n
RUN
MPPC
1.5V
22nF
IIN =
PGOOD
PWM
VCC
VS1
VCC
VS2
1M
VS3
GND
PGND
2.2µF
L1: Coilcraft EPL2014
NOTE: RC AND CC HAVE BEEN ADDED FOR IMPROVED MPPC LOOP STABILITY WHEN USING AN INPUT
CAPACITOR VALUE LESS THAN THE RECOMMENDED MINIMUM OF 22µF
October 2013 : LT Journal of Analog Innovation | 19
Inverting DC/DC Controller Converts a Positive Input to a
Negative Output with a Single Inductor
David Burgoon
There are a number of ways to produce a negative voltage from a positive voltage
source, including using a transformer or two inductors and/or multiple switches, but
none are as easy as using the LTC3863, which is elegant in its simplicity, has superior
efficiency at light loads and reduces parts count when compared to these solutions.
The LTC3863 can produce a –0.4V to
–150V negative output voltage from a
positive input range of 3.5V to 60V. It
uses a single-inductor topology with one
active P-channel MOSFET switch and one
diode. The high level of integration yields
a simple, low parts count solution.
The LTC3863 offers excellent light load
efficiency, drawing only 70µ A quiescent
current in user programmable Burst
Mode® operation. Its peak current mode,
constant frequency PWM architecture
provides positive control of inductor
current, easy loop compensation and
top-notch loop dynamics. The switching frequency can be programmed from
50kHz to 850kHz with an external resistor
and can be synchronized to an external
clock from 75kHz to 750kHz. The LTC3863
offers programmable soft-start or output
tracking. Safety features include overvoltage, overcurrent, and short-circuit
protection including frequency foldback.
–12V, 1A CONVERTER OPERATES
FROM 4.5V–16V SOURCE
The circuit shown in Figure 1 produces
a –12V, 1A output from a 4.5V–16V input.
Operation is similar to a flyback converter, storing energy in the inductor
when the switch is on and releasing it
through the diode to the output when
20 | October 2013 : LT Journal of Analog Innovation
the switch is off, except that with the
LTC3863, no transformer is required.
To prevent excessive current that can
result from minimum on-time when the
45.3k
Figure 1. Inverting
converter produces
–12V at 1A from a
4.5V–16V source
output is short-circuited, the controller
folds back the switching frequency when
the output is below half of nominal.
100k
0.47µF 16V
CLKIN
RUN
CAP
27nF
390pF
SENSE
SS
Q1 D1
GATE
14.7k
61.9k
LTC3863
L1
10µH
ITH
1.21M
FREQ
SGND
VFBN
68pF
PGND
Figure 2. Switch node voltage, inductor current and
ripple waveforms at 5V input and –12V output at 1A
VSW
10V/DIV
100µF
20V
VIN
4.5V TO 16V
16mΩ
PLLIN/MODE
0.1µF
10µF
25V
×2
VIN
+
VFB
33µF
16V
×2
Figure 3. Switch node voltage, inductor current and
ripple waveforms at 5V input and –12V output at
30mA in pulse-skipping mode
VOUT
50mV/DIV
(AC-COUPLED)
VOUT
50mV/DIV
(AC-COUPLED)
IL
1A/DIV
VIN = 5V
VOUT = –12V
IOUT = 1A
1µs/DIV
VOUT
–12V
150µF 1A
16V
×2
D1: DIODES PDS540
80.6k L1: TOKO 919AS-100M
Q1: VISHAY SI7129DN-T1-GE3
VSW
10V/DIV
IL
1A/DIV
+
ADVANCED CONTROLLER
CAPABILITIES
1µs/DIV
VIN = 5V
VOUT = –12V
IOUT = 30mA
PULSE-SKIPPING MODE
design features
The LTC3863 can produce a –0.4V to –150V negative output voltage
from a positive input range of 3.5V to 60V. It uses a single-inductor
topology with one active P-channel MOSFET switch and one diode.
The high level of integration yields a simple, low parts count solution.
The LTC3863 can be programmed to enter
either high efficiency Burst Mode operation or pulse-skipping mode at light loads.
In Burst Mode operation, the controller
directs fewer, higher current pulses and
then enters a low current quiescent state
for a period of time depending on load.
In pulse-skipping mode, the LTC3863
skips pulses at light loads. In this mode,
the modulation comparator may remain
tripped for several cycles and force the
external MOSFET to remain off, thereby
skipping pulses. This mode offers the
benefits of smaller output ripple, lower
audible noise, and reduced RF interference,
at the expense of lower efficiency when
compared to Burst Mode operation. This
circuit fits in about 0.5in2 (3.2cm2) with
components on both sides of the board.
Figure 2 shows switch node voltage,
inductor current, and ripple waveforms
at 5V input and –12V output at 1A.
The inductor is charged (current rises)
when the PMOSFET is on, and discharges
through the diode to the output when the
PMOS turns off. Figure 3 shows the same
waveforms at 30m A out in pulse-skipping
mode. Notice how the switch node rings
out around 0V when the inductor current
reaches zero. The effective period stops
when the current reaches zero. Figure 4
shows the same load condition with Burst
Mode operation enabled. Power dissipation drops by 36% at this operating point, and efficiency increases from
72% to 80%. Figure 5 shows waveforms
with the output shorted. The switching frequency is reduced to about 80kHz
in this condition to prevent excessive
current that could otherwise result.
Figure 4. Switch node voltage, inductor current and
ripple waveforms at 5V input and –12V output at
30mA in Burst Mode operation
Figure 5. Switch node voltage, inductor current and
ripple waveforms at 5V input with the output shorted
HIGH EFFICIENCY
Figure 6 shows efficiency curves for both
pulse-skipping and Burst Mode operation. Exceptional efficiency of 89.3% is
achieved at 1A load and 12V input. Notice
how Burst Mode operation dramatically improves efficiency at loads less
than 0.1A. Pulse-skipping efficiency at
light loads is still much higher than that
obtained from synchronous operation.
CONCLUSION
The LTC3863 simplifies the design of converters producing a negative output from a
positive source. It is elegant in its simplicity, high in efficiency, and requires only a
small number of inexpensive external components to form a complete converter. n
Figure 6. Efficiency in normal and Burst Mode
operation
90
VSW
10V/DIV
VSW
10V/DIV
85
80
EFFICIENCY (%)
IL
1A/DIV
VOUT
50mV/DIV
(AC-COUPLED)
VOUT
50mV/DIV
(AC-COUPLED)
75
70
65
60
55
50
IL
1A/DIV
500µs/DIV
VIN = 5V
VOUT = –12V
IOUT = 30mA
Burst Mode OPERATION
VIN = 5V
SHORTED OUTPUT
5µs/DIV
45
0.01
VIN = 5V, Burst Mode OPERATION
VIN = 12V, Burst Mode OPERATION
VIN = 5V, PULSE-SKIPPING MODE
VIN = 12V, PULSE-SKIPPING MODE
0.1
ILOAD (A)
1
October 2013 : LT Journal of Analog Innovation | 21
What’s New with LTspice IV?
Gabino Alonso
www.linear.com/blog/LTspice
LTspice BLOG
Check out the new LTspice blog
(www.linear.com/blog/LTspice) for
tech news, insider tips and interesting points of view regarding LTspice.
Here are just a few of the topics:
•Simulating Power Planes
•Parametric Plots
•Importing & Exporting Data
•Noise Simulations
•Adding Third-Party Models
Follow @LTspice on Twitter for
up-to-date information on models, demo circuits,
events and user tips: www.twitter.com/LTspice
• LT8697: 2MHz 5V step-down converter
with cable drop compensation (6V–42V to
5V at 2.1A) www.linear.com/LT8697
LED Driver
• LT3761: 94% efficient boost LED driver
for automotive headlamp with 25:1
PWM dimming (8V–60V to 60V LED string
at 1A) www.linear.com/LT3761
• LT3055: 5V supply with 497m A precision
current limit, 10m A IMIN (5.4V–45V to
5V at 497m A) www.linear.com/LT3055
• LT3081: Extended safe operating area
supply (2.7V–40V to 1.5V at 1.5A)
www.linear.com/LT3081
Buck Regulators
• LT3514: 36V triple buck regulator
(5.4V–36V to 5V at 1A, 3.3V at 2A and
1.8V at 1A) www.linear.com/LT3514
• LT3995: 3.3V step-down converter
(4.3V–60V to 3.3V at 3A)
www.linear.com/LT3995
• LT3030: Dual, µ Power, low noise linear
regulator (2.2V–20V to 1.8V at 750m A and
at 250m A) www.linear.com/LT3030
1.5V
TimerBlox ® Silicon Timing Devices
• LTC6995-1: Active low
power-on reset timer (1s POR)
www.linear.com/LTC6995–1
Supercapcitor Charger
Precision Amplifiers
• LTC3122: Dual supercapacitor backup
• LTC6090 and LT5400: Wide common mode
power supply (0.5V–5V to 5V at 50m A)
www.linear.com/LTC3122
SELECTED DEMO CIRCUITS
Linear Regulators
Linear Regulator
µModule Regulators
•
High efficiency 20A µModule
buck regulator (4.5V–20V to 1.2V at 20A)
www.linear.com/LTM4637
LTM®4637:
• LTM8028: Low output noise, 1.8V,
5A regulator (6V–36V to 1.8V at 5A)
www.linear.com/LTM8028
• LTM8045: –5V inverting converter
(2.8V–18V to –5V at 430m A)
www.linear.com/LTM8045
• LTM8050: 5V step-down
converter (7.5V–58V to 5V at 2A)
www.linear.com/LTM8050
range 10× gain instrumentation amplifier
www.linear.com/LTC6090
SELECTED MODELS
Buck Regulators
• LT3514: Triple step-down switching
regulator with 100% duty cycle
operation www.linear.com/LT3514
• LT3995: 60V, 3A, 2MHz step-down
switching regulator with 2.7µ A quiescent
current www.linear.com/LT3995
• LT8697: USB 5V 2.5A output, 42V input
synchronous buck with cable drop
compensation www.linear.com/LT8697
• LTC3374: 8-channel parallelable
1A
buck
DC/DCs www.linear.com/LTC3374
LED Driver
• LT3954: 40V input LED converter
What is LTspice IV?
LTspice® IV is a high performance SPICE simulator, schematic capture and waveform viewer designed to speed
the process of power supply design. LTspice IV adds enhancements and models to SPICE, significantly reducing
simulation time compared to typical SPICE simulators, allowing one to view waveforms for most switching
regulators in minutes compared to hours for other SPICE simulators.
LTspice IV is available free from Linear Technology at www.linear.com/LTspice. Included in the download is a
complete working version of LTspice IV, macro models for Linear Technology’s power products, over 200 op amp
models, as well as models for resistors, transistors and MOSFETs.
22 | October 2013 : LT Journal of Analog Innovation
with internal PWM generator
www.linear.com/LT3954
Inverting Regulators
• LTC3863:
60V low IQ inverting DC/DC controller
www.linear.com/LTC3863
design ideas
Power User Tip
Like us on Facebook at
facebook.com/LTspice
GENERATING A BODE PLOT OF A SWITCH MODE POWER SUPPLY IN LTspice IV
Determining the open loop gain from a closed loop switch mode power supply (SMPS) is best solved using
Middlebrook’s method, which appears in the International Journal of Electronics, Volume 38, Number 4, 1975. This
method injects test signals into the closed loop system to independently solve for the voltage and current gains so
that the loop remains closed and operating points undisturbed. Using the voltage gain portion of the Middlebrook
method is particularly useful in performing a frequency response analysis (FRA) of an SMPS in LTspice.
To perform a FRA of a switch mode power supply in LTspice:
µModule Regulators
• LTM4624: 14V input,
4A step-down DC/DC µModule regulator
www.linear.com/LTM4624
• LTM4630: Dual 18A or single
36A DC/DC µModule regulator
www.linear.com/product/LTM4630
• LTM4649: 10A step-down DC/DC µModule
regulator www.linear.com/LTM4649
• LTM4676: Dual 13A or single 26A µModule
regulator with digital power system
management www.linear.com/LTM4676
• LTM8050: 58V, 2A step-down µModule reg-
ulator www.linear.com/product/LTM8050
Linear Regulator
• LT3007 Series: 3µ A IQ, 20m A, 45V low
dropout fault tolerant linear regulators
www.linear.com/LT3007
• LT3030 : Dual 750m A /250m A low dropout,
low noise, micropower linear regulator
www.linear.com/LT3030
• LT3081: 1.5A single resistor rugged
linear regulator with monitors
www.linear.com/LT3081
• LT3055: 500m A, linear regulator with
• Insert a voltage source with a value of “SINE(0 1m {Freq})” in the SMPS feedback loop in series with the
feedback pin and label the nodes of this voltage source “A” and “B” as shown. The choice of amplitude (1mV)
impacts accuracy and the signal to noise ratio. Lower
amplitudes lower the signal to noise and the larger the
amplitude the less relevant the frequency response will
be. A good starting point is 1mV to 20mV.
• Paste the following .measure statements on the
schematic as a SPICE directive. These statements
perform the Fourier transform of nodes A and B,
compute the complex open loop gain of the SMPS,
resulting magnitude in dB and phase in degrees.
.measure
.measure
.measure
.measure
.measure
.measure
.measure
.measure
Aavg avg V(a)
Bavg avg V(b)
Are avg (V(a)-Aavg)*cos(360*time*Freq)
Aim avg -(V(a)-Aavg)*sin(360*time*Freq)
Bre avg (V(b)-Bavg)*cos(360*time*Freq)
Bim avg -(V(b)-Bavg)*sin(360*time*Freq)
GainMag param 20*log10(hypot(Are,Aim) / hypot(Bre,Bim))
GainPhi param mod(atan2(Aim, Are) - atan2(Bim, Bre)+180,360)-180
• Paste the following SPICE directive on the schematic. Parameter t0 is the length of time required for the system
to settle to steady state and also sets when the simulator starts saving data. The difference between start and
stop times in this case has been chosen as 25/freq so that the error from a non-integral number of switching
cycles is small, since many cycles are included.
.param t0=.2m
.tran 0 {t0+25/freq} {t0}
• Insert a .step command to set the frequency range over which you want to perform the analysis. In this
example, the simulation runs from 50kHz to 200kHz using five points per octave. Hint: Before stepping through
the entire frequency range, test at a couple of frequencies (e.g., insert “.param Freq = 125K”) and look at
V(A) and V(B) to ensure you have sufficient amplitude in your voltage source, and if possible, tighten up the
frequency range to minimize simulation time.
.step oct param freq 5K 500K 5
.save V(a) V(b)
.option plotwinsize=0 numdgt=15
• Run your simulation (see bottom left corner for status update).
• To view the Bode plot, open the SPICE Error Log (choose SPICE Error Log from the View menu) and right-click
on the log to select “Plot .step’ed .meas data”. Choose Visible Traces from the Plot Settings Menu. Select gain.
From this plot you can then determine the crossover frequency and phase margin of your SMPS design.
precision current limit and diagnostics
www.linear.com/LT3055
Precision Amplifiers
• LTC2057: High voltage, low noise
zero-drift operational amplifier
www.linear.com/LTC2057
Ideal Diode
• LT4320/-1: Ideal diode bridge controller
www.linear.com/LT4320
Further examples and documentation can be found in the educational examples (..\LTspiceIV\examples\
Educational\FRA\) and under the FAQ section of the Help Topics (press F1).
Happy simulations!
October 2013 : LT Journal of Analog Innovation | 23
Solar Battery Charger Maintains High Efficiency in Low Light
J. Celani
An important characteristic of any solar panel is that
it achieves peak power output at a relatively constant
operating voltage (VMP) regardless of illumination level (see
Figure 1). The LT3652 2A battery charger exploits this
characteristic to maintain a solar panel at peak operating
efficiency by implementing input voltage regulation (patent
pending). When available solar power is inadequate
to meet the power requirements of an LT3652 battery
charger, input voltage regulation reduces the battery
charge current. This reduces the load on the solar panel
to maintain the panel voltage at VMP, maximizing the
panel output power. This method of achieving peak panel
efficiency is called maximum power point control (MPPC).
While MPPC optimizes solar panel efficiency during periods of low illumination, the power conversion efficiency of
the battery charger suffers when power
levels are low, degrading the overall power
transfer efficiency from the panel to the
battery. This article shows how to improve
battery charger efficiency by applying
a simple PWM charging technique that
forces the battery charger to release energy
in bursts when power levels are low.
USING THE CURRENT MONITOR
STATUS PIN TO INDICATE LOW
POWER CONDITIONS
The CHRG current monitor status pin on
the LT3652 indicates the state of battery
charge current, and is used here to control
the PWM function. The pin is pulled low
when the charger output current is greater
than C/10, or 1/10 of the programmed
maximum current, and high impedance
when the output current is below C/10.
INCREASING ILLUMINATION
IPANEL (A)
PPANEL (W)
P VS V
VPANEL (V)
Figure 1. A solar panel produces maximum power at
a particular output voltage, VMP, which is relatively
independent of illumination level. The LT3652 2A
battery charger maximizes the output power of a
solar panel by regulating the input panel voltage at
VMP.
solar panel voltage that is higher than
the input regulation voltage (VIN(REG)).
The solar panel voltage climbs through
the UVLO hysteresis range in response
to the charger being disabled until the
UVLO rising threshold is achieved, when
the charger is re-enabled at full power. The
charger then provides charge current until
During periods of low illumination, the
input regulation loop can reduce the
output current of the charger to below
C/10, causing the CHRG pin to become
high impedance. This status pin changeof-state is used to disable the IC by
triggering an input undervoltage lockout
(UVLO) with the falling threshold at a
SOLAR PANEL INPUT
~25V OC VOLTAGE
VMP = 17V
D2
D1
R4
536k
C1
390µF
R6
1M
VIN
VIN_REG
SHDN
R5
100k
SW
LT3652
BAT
VFB
GND
C4
0.68µF
M1
RSENSE 0.05Ω
R1
280k
NTC
CHRG
TIMER
L1
10µH
SENSE
FAULT
R7
63.4k
D3
1µF
BOOST
R8
1M
PWM COMPONENTS
Figure 2. 17V VMP solar panel to 3-cell Li-ion (12.6V) 2A charger
24 | October 2013 : LT Journal of Analog Innovation
VMP
I VS V
R3
174k
R2
100k
D1,D2: CMSH3-40MA
D3: CMPSH1-4
L1: IHLP-2525CZ-11
M1: BSS123
C2
10µF
+
3-CELL
Li-ION
design ideas
While MPPC optimizes solar panel efficiency during periods of low illumination, the power
conversion efficiency of the battery charger suffers when power levels are low. This article
shows how to improve battery charger efficiency by applying a simple PWM charging
technique that forces the battery charger to release energy in bursts at low power levels.
100
TA = 25°C
CHARGER CONVERSION EFFICIENCY (%)
INPUT REGULATION VOLTAGE (V)
22
20
18
100% TO 98% PEAK POWER
16
98% TO 95% PEAK POWER
14
12
10
0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8
CHARGER OUTPUT CURRENT (A)
2
PWM CIRCUIT ENABLED
80
VIN
5V/DIV
60
40
20
VCHRG
10V/DIV
WITHOUT PWM CIRCUIT
VBAT = 12V
1
10
100
ICHARGE (mA)
1k
10k
5ms/DIV
Figure 3. Typical “12V system” (VMP = 17V) solar
panel efficiency
Figure 4. Efficiency for the circuit in Figure 2
Figure 5. Waveform of VIN during PWM for the
circuit in Figure 2
input voltage regulation again disables
the charger. This cycle repeats, generating a charger output that is a series of
high current bursts, which maximizes
the efficiency of the charger as well as
the efficiency of the entire solar charger system at any illumination level.
The LT3652’s CHRG pin is pulled low while
required charge current exceeds 1/10 of
the 2A programmed maximum charge
current, or 200m A. When charge current
is reduced by the input regulation loop
below the 200m A level, the CHRG pin
becomes high impedance, which allows
the gate of M1 to be pulled up to VBAT,
enabling the FET, M1. This FET pulls R7
to ground, engaging an input voltage
UVLO function using the SHDN pin and
the resistor divider made from R6 and R7.
The UVLO function is programmed with
that divider to have a falling threshold
of 18V and a rising threshold of 20V. The
falling threshold is the critical design
value, and must be programmed to a voltage that is higher than the input regulation voltage, and is 10% lower than the
rising threshold, as is dictated by the
LT3652 shutdown threshold hysteresis.
charge current, the LT3652’s input voltage regulation reduces the output charge
current until the charger input power is
equivalent to the available power provided by the panel. With input regulation
active, the panel voltage at VIN is held at
the programmed 17V peak power voltage, maximizing the power produced
from the panel. If the panel illumination
becomes low enough that the available
panel power corresponds to charge current
less than 200m A, the CHRG pin becomes
high impedance and the UVLO function is enabled via M1, R6 and R7.
HIGH EFFICIENCY LI-ION CHARGER
Figure 2 shows a solar panel to 3-cell
Li-Ion charger with low power PWM functionality. This charger employs a 17V input
regulation voltage (a common VMP for
“12V system” panels), programmed using
the resistor divider R4 and R5 at the
VIN_REG pin. Keeping the operating voltage
of a typical 12V system solar panel near its
17V rated VMP voltage yields panel efficiencies close to 100%, as shown in Figure 3.
The low power PWM function is implemented using M1, R6, R7 and R8. Figure 4
shows that the addition of the PWM circuitry significantly increases efficiency at
battery charge currents below 200m A.
During low illumination conditions,
when available panel power is insufficient for the LT3652 to provide required
Since VIN is at 17V, which is lower than the
UVLO falling threshold, the LT3652 shuts
down, disabling all of the battery charging functions. With the battery charger
disabled, virtually all of the panel output
current charges the input capacitor (C1),
increasing the voltage at VIN until the
20V UVLO rising threshold is achieved, reenabling the LT3652. The battery charger
is re-enabled with VIN well above the
October 2013 : LT Journal of Analog Innovation | 25
D2
D1
R4
536k
C1
390µF
VIN
R6
1M
LT3652
VIN_REG
SHDN
R5
100k
SW
D3
1µF
L1
10µH
BOOST
RSENSE 0.05Ω
SENSE
FAULT
BAT
R1
309k
NTC
CHRG
VFB
TIMER
GND
R3
174k
C2
10µF
R2
100k
100
+
6-CELL
LEAD
ACID
C4
4.7µF
R7
63.4k
M1
R8
1M
D5
C5
4.7µF
PWM COMPONENTS
D4
R9
1M
D1,D2: CMSH3-40MA
D3: CMPSH1-4
D4,D5: 1N914
L1: IHLP-2525CZ-11
M1: BSS123
FLOAT COMPONENTS
Figure 6. 17V VMP panel to 6-cell 2A lead-acid charger
17V
input regulation threshold, so full
charge current flows into the battery. The
CHRG status pin is pulled low in response
to the high battery charge current level,
which disables the UVLO function. As
long as the power required by the battery
charger remains less than that available
from the solar panel, the panel voltage will collapse until VIN is reduced to
17V, when the battery charge current is
reduced by input regulation to maintain
that voltage. When the charge current is
again reduced to 200m A, the CHRG pin
becomes high impedance, the UVLO circuit
is reengaged, and the disable/enable cycle
repeats, resulting in a string of charge
current ‘bursts’ that average to the battery charge current corresponding to the
available power from the solar panel.
Figure 5 shows the PWM operation of the
circuit in Figure 2. While the LT3652 is
disabled, the voltage on VIN ramps from
the input regulation threshold of 17V to
the shutdown threshold of 20V. The voltage on the LT3652 CHRG pin is low while
the charger is enabled and high while the
charger is disabled. While the charger
is disabled, the panel energy is stored in
the input capacitor, so the output power
from the panel remains continuous. The
26 | October 2013 : LT Journal of Analog Innovation
CHARGER CONVERSION EFFICIENCY (%)
SOLAR PANEL INPUT
~25V OC VOLTAGE
VMP = 17V
PWM CIRCUIT ENABLED
80
60
WITHOUT PWM CIRCUIT
40
20
VBAT = 13V
1
10
100
ICHARGE (mA)
1k
10k
Figure 7. Efficiency curve for circuit in Figure 6
efficiency of the solar panel corresponds
to the average voltage on the panel during
PWM operation, which is about 18.5V.
HIGH EFFICIENCY LEAD-ACID
CHARGER
Figure 6 shows a 6-cell lead-acid battery
charger with low current PWM functionality. The battery charger is designed for a
solar panel that has similar characteristics
to that used for the charger in Figure 2.
This lead-acid charger performs a 3-stage
lead-acid charging profile, employing 2A bulk mode charging, absorption
mode charging to 14.4V, and float charge
maintenance at 13.5V. The battery charger
provides up to 2A while charging with
CC/CV characteristics up to the absorption mode regulation voltage of 14.4V,
provided there is ample input power
available from the solar panel. As the
battery nears the 14.4V regulation voltage, charge current is reduced, completing absorption mode charging when the
charge current falls to 200m A, or 1/10
the maximum charge current (C/10).
When absorption mode charging is
completed, the CHRG pin becomes high
impedance in response to achieving the
C/10 charge current threshold, and float
mode maintenance charging begins.
The regulation voltage is reduced from
14.4V to 13.5V in float mode, achieved by
effectively removing R9 from the VFB summing node—accomplished by a diodeOR circuit (D4 and D5) when CHRG is pulled
high by R8, via the reverse-biased D4.
Float mode charging regulation is also
implemented if the LT3652 charger experiences inadequate input power due to low
solar panel illumination levels. If charge
current is reduced to less than 200m A via
input regulation and PWM operation
begins, the CHRG pin voltage becomes a
pulsed waveform. D5 and C5 implement a
peak-detect filter that maintains a continuous reverse-bias on D4, keeping the
charger in float mode (VCHARGE = 13.5V)
during PWM operation. Figure 7 shows
that the addition of the PWM circuitry
significantly increases efficiency at battery charge currents below 200m A.
During PWM operation, the input voltage
ramps from the input regulation threshold of 17V to the shutdown threshold of
20V during the period the IC is disabled,
as previously described for the battery
charger in Figure 2. The output power
from the solar panel corresponds to the
design ideas
SOLAR PANEL INPUT
~25V OC VOLTAGE
VMP = 17V
D2
D1
R4
1M
C1
390µF
VIN
R6
1M
VIN_REG
SHDN
R5
215k
SW
LT3652
FAULT
R7
73.2k
C6
4.7µF
R10
1.2M
M1
R8
1M
TIMER
D7
VMP REDUCTION
COMPONENTS
PWM COMPONENTS
L1
10µH
RSENSE 0.05Ω
SENSE
BAT
R1
309k
NTC
CHRG
D6
D3
1µF
BOOST
VFB
GND
R3
174k
C2
10µF
R2
100k
+
6-CELL
LEAD
ACID
C4
4.7µF
D5
C5
4.7µF
D4
R9
1M
FLOAT COMPONENTS
D1,D2: CMSH3-40MA
D3: CMPSH1-4
D4–D7: 1N914
L1: IHLP-2525CZ-11
M1: BSS123
Figure 8. 17V VMP panel to 6-cell 2A lead-acid charger with low current VMP tracking
average voltage of the panel, or about
18.5V. Figure 3 shows that this voltage is
within the optimum operational range
for higher output currents, but is above
that range at currents less than 200m A.
To maximize both solar panel output
efficiency and battery charger efficiency
in applications with extended low light
operation, the VIN(REG) and UVLO voltages
should be reduced during the burst period.
A method to do so is described below.
HIGH EFFICIENCY LEAD-ACID
CHARGER WITH LOW CURRENT
V MP TRACKING
The LT3652 lead-acid battery charger in
Figure 8 is similar to the battery charger in
Figure 6, but also lowers the input regulation voltage (VIN(REG)) while the charge current is below 200m A. This improves panel
efficiency by tracking the panel’s characteristic reduction in VMP at low currents.
Low current VMP tracking is implemented
by adding R10 to the input regulation
divider of R4 and R5. R10 is connected
to the input regulation summing node
through a diode-OR circuit (D6 and D7).
When the CHRG pin voltage is high,
R10 is effectively removed from the
summing node via the reverse-biased
D7, lowering VIN(REG) from 17V to 15V.
If the charger experiences inadequate
input power due to low illumination levels, charge current is reduced via the input
regulation loop to maintain the VMP solar
panel voltage of 17V. If charge current is
reduced to less than 200m A, the charger
begins PWM operation and the regulation
threshold is reduced for float charging, as
in the previous lead-acid battery charger
circuit. Additionally, this charger reduces
VIN(REG) to 15V, tracking the reduction
of the solar panel VMP at low currents.
D6 and C6 implement a peak-detect filter,
similar to the previously described D5
and C5. This filter maintains a continuous
reverse-bias on D7, keeping the charger
input regulation voltage at the 15V low
illumination level during PWM operation. The PWM control components (M1
and R6-R8) implement UVLO thresholds
of 16V (falling) and 17.5V (rising). During
PWM operation, the panel voltage at
VIN ramps from the 15V input regulation
voltage to the 17.5V UVLO rising threshold, yielding an average panel voltage of
about 16.25V. This charger maximizes
both charger conversion efficiency and
solar panel output power efficiency by
reducing the operational panel voltage while implementing PWM operation
during periods of low illumination.
CONCLUSION
The LT3652 battery charger IC features a
patent pending input voltage regulation
circuit that is used to maintain a solar
panel at its maximum power voltage,
VMP. While the power output efficiency
of a solar panel is optimized using this
technique, the efficiency of the battery
charger drops at low output currents.
The efficiency of a LT3652 solar-powered
battery charger can be greatly improved
during low illumination conditions with
a simple PWM technique, implemented
using only a few external components,
maximizing the operational efficiency of
both the charger and the solar panel. n
October 2013 : LT Journal of Analog Innovation | 27
Meet Green Standards in 24VAC and 12VAC Lighting
Systems: Replace Halogen Bulbs with LEDs Driven by
High Power Factor, High Efficiency Converter
Keith Szolusha
LEDs are increasingly used in 24VAC and 12VAC lighting systems as a robust,
energy efficient and high performance alternative to halogen lamps. Power
converters that drive the LEDs should have a high power factor (above 90% in order
to meet generally accepted green standards), should be efficient, use a minimal
number of components and should run cool. They do not need isolation.
One solution that meets these requirements combines a rectifier bridge and a
current-controlled synchronous step-up/
step-down converter. Specifically, a synchronous 4-switch buck-boost converter
can be paired with a 4-switch ideal diode
rectifier bridge for high power LEDs; lower
power solutions can use a standard diode
bridge. Both solutions are shown here.
The LT3791 60V 4-switch synchronous
buck-boost controller IC can drive constant current (either DC or pulsating) into
PVIN
24VRMS
PULSATING 120Hz
M5
RIN
0.003Ω
M6
CIN
1µF
50V
51Ω
0.1µF
TG2
24VAC
60Hz
TG1
IN1
LT4320
IN2
1µF
50V
100k
D3
OUTN
BG2
470nF
1M
OUTP
IVINP
CTRL
IVINN
VIN
INTVCC
TG1
22.6k
M1
SWI
BG1
INTVCC
200k
LT3791
M2
IVINMON
L1 7.8µH
M4
1M
M3
44.2k
RSENSE
0.008Ω
SHORTLED
OPENLED
0.1µF
OPENLED
SNSN
IVINMON
PGND
ISMON
ISMON
BG2
CLKOUT
CLKOUT
SW2
OVLO
TG2
SYNC
FB
SGND
ISP
0.1µF
SS
RT
VC
CSS
22nF
CC
22nF
45.3k
500kHz
D1, D2: NXP BAT46WJ
D3: SMAJ60A
L1: WÜRTH 744325780 7.8µH 8A
M1, M2: RENESAS RJK0651DPB 60VDS
M3, M4: RENESAS RJK0451DPB 40VDS
M5–M8: VISHAY Si7414DN 60VDS
Figure 1. 24V AC to 60W LED driver (600W halogen equivalent) features high power factor and high efficiency
28 | October 2013 : LT Journal of Analog Innovation
RLED
0.022Ω
PULSATING
LEDs
120Hz
15V–25V
0A–4.4A
ISN
PWMOUT
VREF
COUT
4.7µF
50V
×4
SNSP
200k
SHORTLED
CVCC
4.7µF
10V
0.1µF
BST1
EN/UVLO
M8
D1 D2
PWM
BST2
37.4k
BG1
M7
a string of high power LEDs. It features
an output current feedback loop used to
drive constant current through a string of
LEDs, and a CTRL dimming input pin that
can be tied to the 120Hz half-sine wave
design ideas
This eco-friendly 60W LED lighting solution is roughly
equivalent to 600W of halogen lighting without using
lead, mercury, argon, xenon or krypton gases.
IAC
2A/DIV
PVIN
10V/DIV
VLED
5V/DIV
IL1
2A/DIV
VAC
20V/DIV
ILED
2A/DIV
ILED
2A/DIV
5ms/DIV
5ms/DIV
5ms/DIV
Figure 2. 60Hz 24VAC input waveforms
Figure 3. 120Hz pulsating LED driver waveforms
Figure 4. 120Hz pulsating PVIN
output of a rectifier bridge to create a high
power factor pulsating LED current output.
output. When currents reach 5A and
higher, the diodes in a standard rectifier
bridge dissipate significant power and heat
up. The LT4320 helps high power AC applications run efficient and cool by driving
low resistance external N-channel FETs.
98.1% POWER FACTOR
The LT4320 is an ideal diode rectifier
bridge that drives four MOSFETs in place
of four typical rectifier diodes for highest efficiency conversion of the 60Hz
24VAC input to 24VRMS 120Hz pulsating
Figure 1 shows an LED driver that operates
with 98.1% power factor directly from
24VAC. It can drive up to 25V of LEDs with
120Hz pulsating power with LED current
peaking at 4.4A. At 120Hz, the pulsing of
the light is not detectable by the human
Figure 5. Components remain cool in the high efficiency LED driver shown in Figure 1. Note that the The LT4320 ideal driver remains cool at full LED current. The LT3791
high power buck-boost converter and supporting components rise less than 24°C while delivering 60W of LED power. The four ideal diode bridge MOSFETs on the back
of the board (inset) temperature rise less than 13°C (23°C ambient).
October 2013 : LT Journal of Analog Innovation | 29
PVIN
24VRMS
PULSATING 120Hz
D3
D4
D5
CIN
1µF
50V
51Ω
0.1µF
1µF
50V
D7
24VAC
60Hz
RIN
0.003Ω
470nF
1M
D6
100k
IVINP
CTRL
IVINN
VIN
INTVCC
D1 D2
PWM
BST2
37.4k
TG1
22.6k
M1
SWI
BG1
INTVCC
200k
LT3791
M2
IVINMON
L1 15µH
M4
1M
M3
44.2k
RSENSE
0.015Ω
SHORTLED
OPENLED
0.1µF
SNSP
200k
SHORTLED
COUT
4.7µF
50V
×4
0.1µF
BST1
EN/UVLO
CVCC
4.7µF
10V
OPENLED
SNSN
IVINMON
PGND
ISMON
ISMON
BG2
CLKOUT
CLKOUT
SW2
OVLO
TG2
SYNC
FB
SGND
ISP
RLED
0.05Ω
PULSATING
LEDs
120Hz
15V–25V
0A–2A
ISN
0.1µF
PWMOUT
VREF
SS
RT
VC
CSS
22nF
CC
22nF
45.3k
500kHz
D1, D2: NXP BAT46WJ
D3–D6: PDS360
D7: SMAJ60A
L1: WÜRTH 744071150 15µH
M1, M2: RENESAS RJK0651DPB 60VDS
M3, M4: RENESAS RJK0451DPB 40VDS
Figure 6. Alternate, 24W solution uses a standard diode rectifier for simplicity
eye and is seen as constant brightness.
The high power factor 24VAC input voltage and current waveforms are shown in
Figure 2. The 120Hz pulsating LED current waveforms are shown in Figure 3.
start-up is not harsh and inrush currents
do not affect the high power factor.
LED current foldback with the CTRL pin
voltage is used to achieve the high power
factor. The maximum LED current is set
by RLED at 4.5A, but the CTRL pin monitors
the post-rectifier 120Hz PVIN input voltage
(see Figure 4) and shapes the LED current waveform to match the input. When
the input drops below the shutdown pin
threshold, the IC goes into shutdown and
switching stops. The LED current trails off
as the output capacitors are discharged
and soon enough, the input rises above the
shutdown pin threshold and the LT3791
starts back up. With the CTRL pin folding back the LED current at low input,
The 24VAC pulsating LED driver converter in Figure 1 delivers approximately
60W of LED lighting at 94% efficiency.
This eco-friendly solution is roughly
equivalent to 600W of halogen lighting
replacement without using lead, mercury, argon, xenon or krypton gases. The
four synchronous switches of the LT3791
buck-boost converter and those of the
LT4320 ideal diode bridge are responsible
for the high efficiency. Figure 5 shows
the circuit components remaining cool
despite the 60W conversion. The components have less than 24°C temperature rise,
showing that there is plenty of room to
spare for even higher power applications.
30 | October 2013 : LT Journal of Analog Innovation
HIGH EFFICIENCY & HIGH POWER
FACTOR 60W PULSATING LED
DRIVER
A standard rectifier bridge would produce about a 50°C temperature rise and
run several efficiency points lower.
Total efficiency is calculated by measuring the input power, the power factor, and the delivered output power
separately. The values of 63.0W real input
power, 64.4W apparent input power
and 98.1% power factor are measured
with an HP 6812A AC power source.
Measurement of the output power is a
bit more complex. A current probe and
oscilloscope are used to capture the pulsing current and voltage waveforms at the
output of the converter. From these waveforms, the converter output RMS current
and voltage is calculated for the on-time
(tON) of the LED. The on-time output power
is POUT(ON) = VRMS(ON) • IRMS(ON). Output
power is zero during LED off-time, where
design ideas
The principals of the 24W circuit are the same as the 60W circuit
and the two operate in the same manner. Efficiency of the 24W
circuit is 90%, lower than the 94% achieved by the 60W circuit.
Nevertheless, this loss is acceptable due to the overall lower power.
Figure 7. Thermal performance of 24W solution
the current is zero. The output power of
60W is calculated via a simple duty cycle
equation: POUT = POUT(ON) • tON • 120Hz.
Overall efficiency = output power divided
by real input power.
HIGH EFFICIENCY & HIGH POWER
FACTOR 24W PULSATING
LED DRIVER
The circuit in Figure 6 is a high efficiency
and high power factor 24W pulsating
LED driver that operates from 24VAC input.
Because the power level here is less than
half of the 60W LED driver in Figure 1,
the rectifier bridge shown in Figure 8 is
made from four discrete Schottky diodes,
instead of ideal diodes. The trade-offs
for simplicity are slightly lower efficiency and additional heat dissipation.
The principals of the 24W circuit are the
same as the 60W circuit and the two operate in the same manner. Efficiency of the
24W circuit is 90%, lower than the 94%
achieved by the 60W circuit. Nevertheless,
this loss is acceptable due to the overall
lower power, making the temperature
rise in the discrete rectifier bridge components comparable between the two.
With the discrete diode rectifier bridge,
the components only heat up to 49°C as
shown in Figure 7, well within the requirements of most high power LED drivers.
CONCLUSION
The LT4320 and LT3791 synchronous
buck-boost pulsating LED driver combine to deliver 60W of LED power at
120Hz with 98.1% power factor and 94%
efficiency. This circuit can be used to
easily replace high power 24VAC halogen lighting with more robust and ecofriendly LEDs. At lower power levels,
the LT3791 can be used with a simple
discrete diode rectifier bridge—such as
in a 24W LED driver with 90% efficiency
and similarly high power factor. n
For higher efficiency, simply replace
the discrete rectifier with a LT4320based rectifier. In general, as power
levels and temperatures rise, the need
for synchronous rectification in both
the converter and rectifier goes up.
October 2013 : LT Journal of Analog Innovation | 31
highlights from circuits.linear.com
1nF
VCC
3.3V
0.56µF
WIDEBAND RECEIVER
The LTC5551 is a 2.5V to 3.6V mixer optimized for RF downconverting
mixer applications that require very high dynamic range. The LTC5551
covers the 300MHz to 3.5GHz RF frequency range with LO frequency
range of 200MHz to 3.5GHz. The LTC5551 provides very high IIP3 and
P1dB with low power consumption. A typical application is a base station
receiver covering 700MHz to 2.7GHz frequency range. The RF input can be
matched for a wide range of frequencies and the IF is usable up to 1GHz.
circuits.linear.com/644
22pF
470nH
470nH
475Ω
475Ω
IF+
C+
EN
4mA TO 20mA INPUT
>10V COMPLIANCE
1µF
220k
LTC3255
PGOOD
2.15M
SHUNT
0.1µF
OUTPUT
3.3V
7.4mA
VOUT
PGOOD
BIAS
–
C–
GND
10µF
FB
1.21M
+
3.9pF
LTC6946
SYNTH
LO
LO
1700MHz
BIAS
EN
VCC
0.56µF
22pF
7.4mA DC SUPPLY FROM 4mA TO 20mA CURRENT LOOP
The LTC3255 is a switched-capacitor step-down DC/DC converter
that produces a regulated output (2.4V to 12.5V adjustable) from
a 4V to 48V input. In applications where the input voltage exceeds
twice the output voltage, 2:1 capacitive charge pumping extends
output current capability beyond input supply current limits. At no
load, Burst Mode® operation cuts VIN quiescent current to 16μA.
With its integrated VIN shunt regulator, the LTC3255 excels in
4mA to 20mA current loop applications. The device enables
current multiplication; a 4mA input current can power a 7.4mA
load continuously. Alternatively, the LTC3255 serves as a higher
efficiency replacement for linear regulators and saves space.
circuits.linear.com/643
4V TO 19V
3V TO 19V
SOLAR
PANEL
AC1
AC2
VIN
SW
1µF
6.3V
10µF
25V
–
SOLAR & PIEZO ENERGY HARVESTER AND BATTERY LIFE EXTENDER
The LTC3330 integrates a high voltage energy harvesting power supply
plus a DC/DC converter powered by a primary cell battery to create
a single output supply for alternative energy applications. The energy
harvesting power supply, consisting of an integrated full-wave bridge
rectifier and a high voltage buck converter, harvests energy from
piezoelectric, solar or magnetic sources. The primary cell input powers a
buck-boost converter capable of operation down to 1.8V at its input. Either
DC/DC converter can deliver energy to a single output. The buck operates
when harvested energy is available, reducing the quiescent current draw
on the battery to essentially zero, thereby extending the life of the battery.
The buck-boost powers VOUT only when harvested energy goes away.
circuits.linear.com/642
LTC5551
LO
VCC
3.3V
VIN
ADC
1nF
7.5nH
1µF
LTC2208
IF
AMP
RF
RFIN
LTC6416
IF –
IF
2.2pF
EN
(0V/3.3V)
+
BPF
4.7µF, 6.3V
LTC3330
SWA
CAP
SWB
VIN2
VOUT
PIEZO
MIDE
V25W
22µH
22µH
1.8V TO 5V
50mA
LDO_IN
+
PRIMARY
CELL
1.8V TO 5.5V
4.7µF
6.3V
3
3
3
4
10mF
2.7V
SCAP
BAT
BAL
47µF
6.3V
10mF
2.7V
OUT[2:0]
EH_ON
LDO[2:0]
PGVOUT
IPK[2:0]
UV[3:0]
PGLDO
LDO_EN
LDO_OUT
GND
VIN3
OPTIONAL
1.2V TO 3.6V
50mA
1µF
6.3V
22µF
6.3V
L, LT, LTC, LTM, Linear Technology, the Linear logo, Burst Mode, LTspice, TimerBlox and µModule are registered trademarks, and LTPoE++ is a trademark of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
© 2013 Linear Technology Corporation/Printed in U.S.A./61.8K
Linear Technology Corporation
1630 McCarthy Boulevard, Milpitas, CA 95035
(408) 432-1900
www.linear.com
Cert no. SW-COC-001530