LT3761A - 60VIN LED Controller with Internal PWM Generator

LT3761A
60VIN LED Controller with
Internal PWM Generator
Features
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Description
3000:1 True Color PWM™ Dimming for LEDs
Wide VIN Range: 4.5V to 60V
Rail-to-Rail Current Sense Range: 0V to 80V
Programmable PWM Dimming Signal Generator
Constant Current (±3%) and Constant-Voltage
(±2%) Regulation
Analog Dimming
Drives LEDs in Boost, SEPIC, Inverting, Buck Mode,
Buck-Boost Mode, or Flyback Configuration
Output Short-Circuit Protected Boost
Open LED Protection and Reporting
Adjustable Switching Frequency: 100kHz to 1MHz
Programmable VIN UVLO with Hysteresis
C/10 Indication for Battery Chargers
Low Shutdown Current: <1µA
Thermally Enhanced 16-Lead MSOP Package
The LT®3761A is a DC/DC controller designed to operate as
a constant-current source and constant-voltage regulator.
It features a programmable internal PWM dimming signal.
The LT3761A is ideally suited for driving high current LEDs,
but also has features to make it suitable for charging batteries and supercapacitors. The fixed frequency, current
mode architecture results in stable operation over a wide
range of supply and output voltages. A voltage feedback
pin serves as the input for several LED protection features,
and also makes it possible for the converter to operate
as a constant-voltage source. A frequency adjust pin
allows the user to program the frequency from 100kHz
to 1MHz to optimize efficiency, performance or external
component size.
The LT3761A senses output current at the high side or at
the low side of the load. The PWM input can be configured
to self-oscillate at fixed frequency with duty ratio programmable from 4% to 96%. The LT3761A offers improved
accuracy of internal PWM dimming over LT3761. When
driven by an external signal, the PWM input provides LED
dimming ratios of up to 3000:1. The CTRL input provides
additional analog dimming capability.
Applications
High Voltage LED Strings >100V with Ground
Referred Current Sense
n Grounded Anode LEDs
n Battery and SuperCap Chargers
n Accurate Current Limited Voltage Regulators
n
L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks and
True Color PWM is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners. Protected by U.S. Patents including 7199560, 7321203.
Typical Application
94% Efficient Boost LED Driver for Automotive
Headlamp with 25:1 Internal PWM Dimming
VIN
8V TO
60V
2.2µF
×2
499k
90.9k
EN/UVLO
VREF
1M
VIN
140k
100k
2.2µF
×4
GATE
SENSE
LT3761A
CTRL
INTVCC
DIM
PWM Dimming Waveforms at
Various DIM Voltage Settings
10µH
10mΩ
0.01µF
47nF
300Hz
0.25Ω
1A
VDIM = 7.7V
DCPWM = 96%
16.9k
VDIM = 4V
DCPWM = 50%
GND
ILED
1A/DIV
FB
5.1k
4.7nF
28.7k
350kHz
VDIM = 1.5V
DCPWM = 10%
60W
LED
STRING
ISP
OPENLED
ISN
DIM/SS
PWM
PWMOUT
VC
RT INTVCC
140k
1M
VDIM = 0.4V
DCPWM = 4.3%
0.5ms/DIV
3761A TA01b
INTVCC
1µF
3761A TA01a
3761af
For more information www.linear.com/LT3761A
1
LT3761A
Absolute Maximum Ratings
Pin Configuration
(Note 1)
VIN, EN/UVLO.............................................................60V
ISP, ISN......................................................................80V
INTVCC....................................................VIN + 0.3V, 9.6V
GATE, PWMOUT................................................. (Note 2)
CTRL, OPENLED.........................................................15V
FB, PWM...................................................................9.6V
VC, VREF.......................................................................3V
RT, DIM/SS...............................................................1.5V
SENSE.......................................................................0.5V
Operating Ambient Temperature Range (Notes 3, 4)
LT3761AE............................................... –40 to 125°C
LT3761AI................................................ –40 to 125°C
Storage Temperature Range................... –65°C to 150°C
TOP VIEW
PWMOUT
FB
ISN
ISP
VC
CTRL
VREF
PWM
1
2
3
4
5
6
7
8
17
GND
16
15
14
13
12
11
10
9
GATE
SENSE
VIN
INTVCC
EN/UVLO
RT
DIM/SS
OPENLED
MSE PACKAGE
16-LEAD PLASTIC MSOP
θJA = 43°C/W, θJC = 4°C/W
EXPOSED PAD (PIN 17) IS GND, MUST BE SOLDERED TO PCB
Order Information
(http://www.linear.com/product/LT3761A#orderinfo)
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3761AEMSE#PBF
LT3761AEMSE#TRPBF
3761A
16-Lead Plastic MSOP
–40°C to 125°C
LT3761AIMSE#PBF
LT3761AIMSE#TRPBF
3761A
16-Lead Plastic MSOP
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ Some packages are available in 500 unit reels through
designated sales channels with #TRMPBF suffix.
Electrical
Characteristics
The
l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 24V, EN/UVLO = 24V, CTRL = 2V, PWM = 5V, unless otherwise noted.
PARAMETER
CONDITIONS
VIN Minimum Operating Voltage
VIN Tied to INTVCC
VIN Shutdown IQ
EN/UVLO = 0V, PWM = 0V
EN/UVLO = 1.15V, PWM = 0V
VIN Operating IQ (Not Switching)
PWM = 0V
VREF Voltage
–100µA ≤ IVREF ≤ 0µA
VREF Line Regulation
4.5V ≤ VIN ≤ 60V
VREF Pull-Up Current
VREF = 0V
SENSE Current Limit Threshold
SENSE Input Bias Current
Current Out of Pin, SENSE = 0V
DIM/SS Pull-Up Current
Current Out of Pin, DIM/SS = 0V
DIM/SS Voltage Clamp
IDIM/SS = 0µA
MIN
TYP
MAX
UNITS
4.5
V
0.1
1
6
µA
µA
1.8
2.2
mA
2.02
2.05
V
l
l
1.955
l
150
185
210
µA
l
98
105
118
mV
0.001
%/V
40
l
11
14
1.2
µA
17
µA
V
3761af
2
For more information www.linear.com/LT3761A
LT3761A
Electrical
Characteristics
The
l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 24V, EN/UVLO = 24V, CTRL = 2V, PWM = 5V, unless otherwise noted.
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Error Amplifier
Full-Scale ISP/ISN Current Sense Threshold
(VISP-ISN)
CTRL ≥ 1.2V, ISP = 48V
CTRL ≥ 1.2V, ISN = 0V
l
l
242
243
250
257
258
271
mV
mV
1/10th Scale ISP/ISN Current Sense Threshold
(VISP-ISN)
CTRL = 0.2V, ISP = 48V
CTRL = 0.2V, ISN = 0V
l
l
21
17
25
28
30
39
mV
mV
Mid-Scale ISP/ISN Current Sense Threshold
(VISP-ISN)
CTRL = 0.5V, ISP = 48V
CTRL = 0.5V, ISN = 0V
l
l
96
95
100
105
104
115
mV
mV
ISP/ISN Overcurrent Threshold
600
0
ISP/ISN Current Sense Amplifier Input Common Mode
Range (VISN)
ISP/ISN Input Bias Current High Side Sensing (Combined)
mV
80
V
PWM = 5V (Active), ISP = ISN = 48V
PWM = 0V (Standby), ISP = ISN = 48V
100
0.1
µA
µA
ISP/ISN Input Bias Current Low Side Sensing (Combined)
PWM = 5V, ISP = ISN = 0V
–230
µA
ISP/ISN Current Sense Amplifier gm (High Side Sensing)
VISP-ISN = 250mV, ISP = 48V
120
µS
ISP/ISN Current Sense Amplifier gm (Low Side Sensing)
VISP-ISN = 250mV, ISN = 0V
70
µS
CTRL Pin Range for Linear Current Sense Threshold
Adjustment
l
0
1.0
V
100
nA
CTRL Input Bias Current
Current Out of Pin
50
VC Output Impedance
0.9V ≤ VC ≤ 1.5V
15
VC Standby Input Bias Current
PWM = 0V
FB Regulation Voltage (VFB)
ISP = ISN = 48V, 0V
FB Amplifier gm
FB = VFB, ISP = ISN = 48V
500
FB Pin Input Bias Current
Current Out of Pin, FB = VFB
40
100
nA
FB Open LED Threshold
OPENLED Falling, ISP Tied to ISN
VFB  –
65mV
VFB –
50mV
VFB –
40mV
V
C/10 Inhibit for OPENLED Assertion (VISP-ISN)
FB = VFB, ISN = 48V, 0V
14
25
39
FB Overvoltage Threshold
PWMOUT Falling
VFB +
50mV
VFB +
60mV
VFB +
75mV
–20
l
l
1.225
VC Current Mode Gain (∆VVC /∆VSENSE)
MΩ
20
1.255
1.275
nA
V
µS
4
mV
V
V/V
Oscillator
Switching Frequency
RT = 95.3kΩ
RT = 8.87kΩ
GATE Minimum Off-Time
CGATE = 2200pF
160
ns
GATE Minimum On-Time
CGATE = 2200pF
180
ns
l
85
925
100
1000
115
1050
kHz
kHz
Linear Regulator
INTVCC Regulation Voltage
10V ≤ VIN ≤ 60V
l
INTVCC Maximum Operating Voltage
7.6
7.85
8.05
8.1
V
INTVCC Minimum Operating Voltage
4.5
Dropout (VIN – INTVCC)
IINTVCC = –10mA, VIN = 7V
INTVCC Undervoltage Lockout
EN/UVLO = 2V
INTVCC Current Limit
INTVCC = 6V, 8V ≤ VIN ≤ 60V
INTVCC Current in Shutdown
EN/UVLO = 0V, INTVCC = 8V
390
l
V
V
mV
3.9
4.1
4.4
V
30
36
42
mA
8
13
µA
3761af
For more information www.linear.com/LT3761A
3
LT3761A
Electrical
Characteristics
The
l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 24V, EN/UVLO = 24V, CTRL = 2V, PWM = 5V, unless otherwise noted.
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
1.18
1.220
1.26
V
Logic Inputs/Outputs
EN/UVLO Threshold Voltage Falling
l
EN/UVLO Rising Hysteresis
20
EN/UVLO Input Low Voltage
IVIN Drops Below 1µA
EN/UVLO Pin Bias Current Low
EN/UVLO = 1.15V
EN/UVLO Pin Bias Current High
EN/UVLO = 1.33V
OPENLED Output Low
IOPENLED = 1mA
l
1.6
mV
0.4
V
2.3
2.7
µA
10
100
nA
200
mV
V
PWM Pin Signal Generator
PWM Falling Threshold
l
0.78
0.83
0.88
PWM Threshold Hysteresis (VPWMHYS)
IDIM/SS = 0µA
0.35
0.4
0.6
V
PWM Pull-Up Current (IPWMUP)
PWM = 0.7V, IDIM/SS = 0µA
6
7.5
9
µA
PWM Pull-Down Current (IPWMDN)
PWM = 1.5V, IDIM/SS = 0µA
68
88
110
µA
PWM Fault Mode Pull-Down Current
INTVCC = 3.8V
PWMOUT Duty Ratio for PWM Signal Generator (Note 5)
IDIM/SS = –6.5µA
IDIM/SS = 0µA
IDIM/SS = 20µA
IDIM/SS = 44µA
3.2
7.2
42
93
3.8
7.9
50
95
4.4
8.6
58
97
%
%
%
%
PWMOUT Signal Generator Frequency
PWM = 47nF to GND, IDIM/SS = 0µA
215
300
400
Hz
15
mA
PWMOUT, Gate Pin Drivers
PWMOUT Driver Output Rise Time (tr)
CL = 560pF
35
ns
PWMOUT Driver Output Fall Time (tf)
CL = 560pF
35
ns
PWMOUT Output Low (VOL)
PWM = 0V
0.05
PWMOUT Output High (VOH)
INTVCC –
0.05
V
V
GATE Output Rise Time (tr)
CL = 3300pF
25
ns
GATE Output Fall Time (tf)
CL = 3300pF
25
ns
GATE Output Low (VOL)
0.1
GATE Output High (VOH)
INTVCC –
0.05
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: Do not apply a positive or negative voltage or current source to
GATE or PWMOUT pins, otherwise permanent damage may occur.
Note 3: The LT3761AE is guaranteed to meet performance specifications
from the 0°C to 125°C junction temperature. Specifications over the –40°C
to 125°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls. The
V
V
LT3761AI is guaranteed over the full –40°C to 125°C operating junction
temperature range. Operating lifetime is derated at junction temperatures
greater than 125°C.
Note 4: The LT3761A includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed the maximum operating junction temperature
when overtemperature protection is active. Continuous operation above
the specified maximum junction temperature may impair device reliability.
Note 5: PWMOUT Duty Ratio is calculated:
Duty = IPWMUP/(IPWMUP + IPWMDN)
3761af
4
For more information www.linear.com/LT3761A
LT3761A
Typical Performance Characteristics
VISP-ISN Threshold
vs CTRL Voltage
300
TA = 25°C, unless otherwise noted.
VISP-ISN Threshold
vs ISP Voltage
Full-Scale VISP-ISN Threshold
vs Temperature
260
260
FB < 1.18V
200
150
100
50
256
255
VISP -ISN (mV)
VISP-ISN THRESHOLD (mV)
VISP -ISN THRESHOLD (mV)
250
250
0.5
1
1.5
CTRL VOLTAGE (V)
240
2
0
20
3761A G01
250
ISP = 48V
300
1.265
250
1.260
200
1.250
1.245
0
3761A G03
VREF Source Current
vs Temperature
CTRL = 2V
150
CTRL = 0.5V
100
VREF Voltage vs Temperature
1.21
1.22 1.23 1.24
FB VOLTAGE (V)
1.25
190
180
170
160
150
–50 –25
1.26
2.05
415
2.01
2.00
1.99
1.98
–50 –25
800
600
500
400
25 50 75 100 125 150
TEMPERATURE (°C)
3761A G07
100
405
400
395
390
300
385
200
0
RT = 25.5k
410
700
FREQUENCY (kHz)
SWITCHING FREQUENCY (kHz)
420
900
2.02
25 50 75 100 125 150
TEMPERATURE (°C)
3761A G06
Switching Frequency
vs Temperature
Switching Frequency vs RT
1000
2.03
0
3761A G05
2.06
2.04
25 50 75 100 125 150
TEMPERATURE (°C)
200
0
1.20
25 50 75 100 125 150
TEMPERATURE (°C)
3761A G04
0
3761A G02
50
1.240
–50 –25
244
–50 –25
80
VISP-ISN Threshold vs FB Voltage
1.270
1.255
40
60
ISP VOLTAGE (V)
VREF SOURCE CURRENT (µA)
0
VISP-ISN (mV)
VFB (V)
252
246
FB Regulation Voltage (VFB)
vs Temperature
VREF (V)
ISN = 0
254
248
245
0
–50
CTRL = 2V
258
10
100
RT (kΩ)
3761A G08
380
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3761A G09
3761af
For more information www.linear.com/LT3761A
5
LT3761A
Typical Performance Characteristics
SENSE Current Limit Threshold
vs Temperature
EN/UVLO Hysteresis Current
vs Temperature
110
EN/UVLO Threshold
vs Temperature
2.8
1.27
100
95
90
–50 –25
0
EN/UVLO THRESHOLD (V)
2.6
105
EN/UVLO CURRENT (µA)
SENSE THRESHOLD (mV)
TA = 25°C, unless otherwise noted.
2.4
2.2
2.0
1.8
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
25 50 75 100 125 150
TEMPERATURE (°C)
3761A G11
0
1.25
RISING
1.23
FALLING
1.21
1.19
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3761A G10
3761A G12
INTVCC Current Limit vs
vs Temperature
INTVCC Dropout Voltage
vs Current, Temperature
40
VISP-ISN C/10 Threshold
vs Temperature
35
0
–0.2
34
–0.6
TA = 25°C
–0.8
–1.0
–1.2
–1.8
25 50 75 100 125 150
TEMPERATURE (°C)
VIN = 7V
0
5
10
15
20
LDO CURRENT (mA)
25
PWM Signal Generator Duty Ratio
vs DIM/SS Current
30
10
–50 –25
PWM Signal Generator Frequency
vs Duty Ratio
100
PWMOUT FREQUENCY (Hz)
70
60
50
40
30
20
25 50 75 100 125 150
TEMPERATURE (°C)
3761A G15
CPWMOUT = 2.2nF
CPWM = 47nF
80
0
PWMOUT Waveform
340
90
PWMOUT DUTY RATIO (%)
ISN = 0V
20
3761A G14
3761A G13
320
PWM
INPUT
5V/DIV
300
PWMOUT
5V/DIV
280
200ns/DIV
10
0
–10
25
15
–1.6
0
ISP = 24V
TA = 130°C
–1.4
32
30
–50 –25
VISP-ISN (mV)
36
30
TA = –45°C
–0.4
LDO DROPOUT (V)
INTVCC CURRENT LIMIT (mA)
38
0
10
20
30
DIM/SS CURRENT (µA)
40
50
3761A G16
260
0
20
40
60
DUTY RATIO (%)
80
3761AA G18
100
3761A G17
3761af
6
For more information www.linear.com/LT3761A
LT3761A
Typical Performance Characteristics
DIM/SS Voltage
vs Current, Temperature
ISP/ISN Input Bias Current
vs CTRL Voltage, ISP = 48V
1.30
0
INPUT BIAS CURRENT (µA)
1.25
TA = –45°C, 25°C
1.20
TA = 130°C
1.15
1.10
–10
0
10
20
30
DIM/SS CURRENT (µA)
40
80
60
40
0
50
ISN
0
0.5
1
CTRL (V)
1.5
3761 G19
PWMOUT Duty Ratio
vs Temperature, IDIM/SS = 0µA
55
CPWM = 47nF
9.0
7.5
0
25 50 75 100 125 150
TEMPERATURE (°C)
3761A G22
ISP
–160
0.5
1
CTRL (V)
1.5
2
3761 G21
800
CPWM = 47nF
700
51
49
45
–50 –25
0
VISP-ISN Overcurrent Threshold
vs Temperature
600
ISP = 24V
500
ISN = 0V
400
47
7.0
–120
3761 G20
VISP-ISN (mV)
DUTY RATIO (%)
8.0
ISN
–80
–200
2
53
8.5
–40
PWMOUT Duty Ratio
vs Temperature, IDIM/SS = 20µA
9.5
6.5
–50 –25
INPUT BIAS CURRENT (µA)
ISP
20
DUTY RATIO (%)
ISP/ISN Input Bias Current
vs CTRL Voltage, ISN = 0V
120
100
DIM/SS VOLTAGE (V)
TA = 25°C, unless otherwise noted.
0
25 50 75 100 125 150
TEMPERATURE (°C)
3761 G23
300
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3761 G24
3761af
For more information www.linear.com/LT3761A
7
LT3761A
Pin Functions
PWMOUT (Pin 1): Buffered Version of PWM Signal for
Driving LED Load Disconnect NMOS or Level Shift. This
pin also serves in a protection function for the FB overvoltage condition—will toggle if the FB input is greater
than the FB regulation voltage (VFB) plus 60mV (typical).
The PWMOUT pin is driven from INTVCC. Use of a FET
with gate cut-off voltage higher than 1V is recommended.
FB (Pin 2): Voltage Loop Feedback Pin. FB is intended for
constant-voltage regulation or for LED protection and open
LED detection. The internal transconductance amplifier with
output VC will regulate FB to 1.25V (nominal) through the
DC/DC converter. If the FB input exceeds the regulation
voltage, VFB, minus 50mV and the voltage between ISP
and ISN has dropped below the C/10 threshold of 25mV
(typical), the OPENLED pull-down is asserted. This action
may signal an open LED fault. If FB is driven above the FB
overvoltage threshold, the PWMOUT and GATE pins will be
driven low to protect the LEDs from an overcurrent event.
Do not leave the FB pin open. If not used, connect to GND.
ISN (Pin 3): Connection Point for the Negative Terminal
of the Current Feedback Resistor. The constant output
current regulation can be programmed by ILED = 250mV/
RLED when CTRL > 1.2V or ILED = (CTRL – 100mV)/(4 •
RLED). If ISN is greater than INTVCC, input bias current
is typically 20μA flowing into the pin. Below INTVCC,
ISN bias current decreases until it flows out of the pin.
ISP (Pin 4): Connection Point for the Positive Terminal of
the Current Feedback Resistor. Input bias current depends
upon CTRL pin voltage. When it is greater than INTVCC
it flows into the pin. Below INTVCC, ISP bias current
decreases until it flows out of the pin. If the difference
between ISP and ISN exceeds 600mV (typical), then an
overcurrent event is detected. In response to this event,
the GATE and PWMOUT pins are driven low to protect
the switching regulator, a 15mA pull-down on PWM and
a 9mA pull-down on the DIM/SS pin are activated for 4µs.
VC (Pin 5): Transconductance Error Amplifier Output Pin
Used to Stabilize the Switching Regulator Control Loop
with an RC Network. The VC pin is high impedance when
PWM is low. This feature allows the VC pin to store the
demand current state variable for the next PWM high
transition. Connect a capacitor between this pin and GND;
a resistor in series with the capacitor is recommended for
fast transient response.
CTRL (Pin 6): Current Sense Threshold Adjustment Pin.
Constant current regulation point VISP-ISN is one-fourth
VCTRL plus an offset for 0V ≤ CTRL ≤ 1V. For CTRL >
1.2V the VISP-ISN current regulation point is constant at
the full-scale value of 250mV. For 1V ≤ CTRL ≤ 1.2V, the
dependence of VISP-ISN upon CTRL voltage transitions from
a linear function to a constant value, reaching 98% of fullscale value by CTRL = 1.1V. Do not leave this pin open.
VREF (Pin 7): Voltage Reference Output Pin, Typically 2V.
This pin drives a resistor divider for the CTRL pin, either
for analog dimming or for temperature limit/compensation
of LED load. It can be bypassed with 10nF or greater, or
less than 50pF. Can supply up to 185µA (typical).
PWM (Pin 8): A signal low turns off switcher, idles the
oscillator and disconnects the VC pin from all internal
loads. PWMOUT pin follows the PWM pin, except in fault
conditions. The PWM pin can be driven with a digital
signal to cause pulse width modulation (PWM) dimming
of an LED load. The digital signal should be capable of
sourcing or sinking 200μA at the high and low thresholds.
During start-up when DIM/SS is below 1V, the first rising
edge of PWM enables switching which continues until
VISP-ISN ≥ 25mV or SS ≥ 1V. Connecting a capacitor from
PWM pin to GND invokes a self-driving oscillator where
internal pull-up and pull-down currents set a duty ratio
for the PWMOUT pin for dimming LEDs. The magnitude
of the pull-up/down currents is set by the current in the
DIM/SS pin. The capacitor on PWM sets the frequency of
the dimming signal. For hiccup mode response to output
short-circuit faults, connect this pin as shown in the application titled Boost LED Driver with Output Short-Circuit
Protection. If not used, connect the PWM pin to INTVCC
through a 1k resistor.
3761af
8
For more information www.linear.com/LT3761A
LT3761A
Pin Functions
OPENLED (Pin 9): An open-drain pull-down on this pin
asserts if the FB input is greater than the FB regulation
voltage (VFB) minus 50mV (typical) AND the difference
between current sense inputs ISP and ISN is less than
25mV. To function, the pin requires an external pull-up
resistor, usually to INTVCC. When the PWM input is low
and the DC/DC converter is idle, the OPENLED condition
is latched to the last valid state when the PWM input was
high. When PWM input goes high again, the OPENLED
pin will be updated. This pin may be used to report transition from constant current regulation to constant voltage
regulation modes, for instance in a charger or current
limited voltage supply.
EN/UVLO (Pin 12): Enable and Undervoltage Detect Pin.
An accurate 1.22V falling threshold with externally programmable hysteresis causes the switching regulator to
shut down when power is insufficient to maintain output
regulation. Above the 1.24V (typical) rising enable threshold
(but below 2.5V), EN/UVLO input bias current is sub-μA.
Below the 1.22V (typical) falling threshold, an accurate
2.3μA (typical) pull-down current is enabled so the user
can define the rising hysteresis with the external resistor
selection. An undervoltage condition causes the GATE
and PWMOUT pins to transition low and resets soft-start.
Tie to 0.4V, or less, to disable the device and reduce VIN
quiescent current below 1μA.
DIM/SS (Pin 10): Soft-Start and PWMOUT Dimming Signal
Generator Programming Pin. This pin modulates switching
regulator frequency and compensation pin voltage (VC)
clamp when it is below 1V. The soft-start interval is set
with an external capacitor and the DIM/SS pin charging
current. The pin has an internal 14μA (typical) pull-up
current source. The soft-start pin is reset to GND by an
undervoltage condition (detected at the EN/UVLO pin),
INTVCC undervoltage, overcurrent event sensed at ISP/
ISN, or thermal limit. After initial start-up with EN/UVLO,
DIM/SS is forced low until the first PWM rising edge. When
DIM/SS reaches the steady-state voltage (~1.17V), the
charging current (sum of internal and external currents) is
sensed and used to set the PWM pin charging and discharge
currents and threshold hysteresis. In this manner, the SS
charging current sets the duty cycle of the PWMOUT signal
generator associated with the PWM pin. This pin should
always have a capacitor to GND, minimum 560pF value,
when used with the PWMOUT signal generator function.
Place the PWM pin capacitor close to the IC.
INTVCC (Pin 13): Current limited, low dropout linear regulator regulates to 7.85V (typical) from VIN. Supplies internal
loads, GATE and PWMOUT drivers. Must be bypassed with
a 1µF ceramic capacitor placed close to the pin and to the
exposed pad GND of the IC.
RT (Pin 11): Switching Frequency Adjustment Pin. Set the
frequency using a resistor to GND (for resistor values, see
the Typical Performance curve or Table 2). Do not leave
the RT pin open. Place the resistor close to the IC.
VIN (Pin 14): Power Supply for Internal Loads and INTVCC
Regulator. Must be locally bypassed with a 0.22µF (or
larger) low ESR capacitor placed close to the pin.
SENSE (Pin 15): The Current Sense Input for the Switch
Control Loop. Kelvin connect the SENSE pin to the positive
terminal of the switch current sense resistor in the source
of the external power NFET. The negative terminal of the
switch current sense resistor should be Kelvin connected
to the exposed pad (GND) of the LT3761A.
GATE (Pin 16): N-channel FET Gate Driver Output. Switches
between INTVCC and GND. Driven to GND during shutdown,
fault or idle states.
GND (Exposed Pad Pin 17): Ground. This pin also serves
as current sense input for the control loop, sensing the
negative terminal of the current sense resistor. Solder the
exposed pad directly to the ground plane.
3761af
For more information www.linear.com/LT3761A
9
LT3761A
Block Diagram
185µA
VREF
1.3V
–
+A7
2.02V
0.8V + F3(IDIM/SS)
F1(IDIM/SS)
PWM
0.8V
F2(IDIM/SS)
100mV
+
–
CTRL
1V
CLAMP
ILED
RLED
×1/4
–
+
S
R
PWMINT
Q
PWM LATCH
+
A2
–
R
Q
S
A1
–
gm
+
CURRENT MODE
COMPARATOR
10µA AT
A1+ = A1–
ISENSE
A5
+
gm
–
1.25V
10µA AT
FB = 1.25V
CV EAMP
VC
IDIM_SS
DETECT
100kHz TO 1MHz
OSCILLATOR
A6
+
+
–
SHDN
PWMINT
FREQ
PROG
BANDGAP
REFERENCE
ISP > ISN + 0.6V
T > 165°C
GND
ISP
14µA
1.2V
FAULT
LOGIC
FB
DIM/SS
ISWITCH
RSNS
OPENLED
LOGIC
25mV
ISN
FAULT
2.3µA
+
–
SENSE
OPENLED
+
–
1.22V
105mV
RAMP
GENERATOR
1V
–
+
+
–
A4
CC EAMP
EN/UVLO
GATE
DRIVER
A3
ISN
FB
7.85V INTVCC
–
+
10µA
FAULT
– LDO
+A8
1.25V
OVFB
COMPARATOR
15mA
+
–
ISP
CTRL
BUFFER
VIN
PWMOUT
–
+
FB
+
–
–
+
RT
3761A BD
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LT3761A
Operation
The LT3761A is a constant-frequency, current mode controller with a low side NMOS gate driver. The GATE pin and
PWMOUT pin drivers and other chip loads are powered
from INTVCC, which is an internally regulated supply. In
the discussion that follows it will be helpful to refer to
the Block Diagram of the IC. In normal operation with the
PWM pin low, the GATE and PWMOUT pins are driven to
GND, the VC pin is high impedance to store the previous
switching state on the external compensation capacitor,
and the ISP and ISN pin bias currents are reduced to
leakage levels. When the PWM pin transitions high, the
PWMOUT pin transitions high after a short delay. At the
same time, the internal oscillator wakes up and generates a pulse to set the PWM latch, turning on the external
power MOSFET switch (GATE goes high). A voltage input
proportional to the switch current, sensed by an external
current sense resistor between the SENSE and GND input
pins, is added to a stabilizing slope compensation ramp
and the resulting switch current sense signal is fed into
the negative terminal of the PWM comparator. The current
in the external inductor increases steadily during the time
the switch is on. When the switch current sense voltage
exceeds the output of the error amplifier, labeled VC, the
latch is reset and the switch is turned off. During the
switch-off phase, the inductor current decreases. At the
completion of each oscillator cycle, internal signals such
as slope compensation return to their starting points and
a new cycle begins with the set pulse from the oscillator.
Through this repetitive action, the PWM control algorithm
establishes a switch duty cycle to regulate a current or
voltage in the load. The VC signal is integrated over many
switching cycles and is an amplified version of the difference between the LED current sense voltage, measured
between ISP and ISN, and the target difference voltage set
by the CTRL pin. In this manner, the error amplifier sets
the correct peak switch current level to keep the LED current in regulation. If the error amplifier output increases,
more current is demanded in the switch; if it decreases,
less current is demanded. The switch current is monitored
during the on-phase and the voltage across the SENSE
pin is not allowed to exceed the current limit threshold of
105mV (typical). If the SENSE pin exceeds the current limit
threshold, the SR latch is reset regardless of the output
state of the PWM comparator. The difference between
ISP and ISN is monitored to determine if the output is in
a short-circuit condition. If the difference between ISP
and ISN is greater than 600mV (typical), the SR latch will
be reset regardless of the PWM comparator. The DIM/SS
pin will be pulled down and the PWMOUT and GATE pins
forced low for at least 4µs. These functions are intended
to protect the power switch as well as various external
components in the power path of the DC/DC converter.
In voltage feedback mode, the operation is similar to that
described above, except the voltage at the VC pin is set by
the amplified difference of the internal reference of 1.25V
and the FB pin. If FB is lower than the reference voltage,
the switch current will increase; if FB is higher than the
reference voltage, the switch demand current will decrease.
The LED current sense feedback interacts with the FB
voltage feedback so that FB will not exceed the internal
reference and the voltage between ISP and ISN will not
exceed the threshold set by the CTRL pin. For accurate
current or voltage regulation, it is necessary to be sure that
under normal operating conditions the appropriate loop is
dominant. To deactivate the voltage loop entirely, FB can
be connected to GND. To deactivate the LED current loop
entirely, the ISP and ISN should be tied together and the
CTRL input tied to VREF.
Two LED specific functions featured on the LT3761A are
controlled by the voltage feedback pin. First, when the
FB pin exceeds a voltage 50mV lower (–4%) than the FB
regulation voltage, and the difference voltage between ISP
and ISN is below 25mV (typical), the pull-down driver on
the OPENLED pin is activated. This function provides a
status indicator that the load may be disconnected and
the constant-voltage feedback loop is taking control of the
switching regulator. The OPENLED pin de-asserts only when
PWM is high and FB drops below the voltage threshold. FB
overvoltage is the second protective function. When the
FB pin exceeds the FB regulation voltage by 60mV (plus
5% typical), the PWMOUT pin is driven low, ignoring the
state of the PWM input. In the case where the PWMOUT
pin drives a disconnect NFET, this action isolates the
LED load from GND, preventing excessive current from
damaging the LEDs.
3761af
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11
LT3761A
Applications Information
INTVCC Regulator Bypassing and Operation
The INTVCC pin requires a capacitor for stable operation
and to store the charge for the large GATE switching currents. Choose a 10V rated low ESR, X7R ceramic capacitor
for best performance. A 1μF capacitor will be adequate
for many applications. Place the capacitor close to the IC
to minimize the trace length to the INTVCC pin and also
to the IC ground.
allow the user to program the rising hysteresis. The following equations should be used to determine the value
of the resistors:
R1+R2
R2
= 2.3µA •R1 + VIN,FALLING
VIN,FALLING = 1.22 •
VIN,RISING
VIN
An internal current limit on the INTVCC output protects the
LT3761A from excessive on-chip power dissipation. The
minimum value of this current should be considered when
choosing the switching NMOS and the operating frequency.
IINTVCC can be calculated from the following equation:
IINTVCC = QG • fOSC
Careful choice of a lower QG FET will allow higher switching
frequencies, leading to smaller magnetics. The INTVCC pin
has its own undervoltage disable set to 4.1V (typical) to
protect the external FETs from excessive power dissipation caused by not being fully enhanced. If the INTVCC pin
drops below the UVLO threshold, the GATE and PWMOUT
pins will be forced to 0V and the soft-start pin will be reset.
If the input voltage, VIN, will not exceed 8V, then the INTVCC pin could be connected to the input supply. Be aware
that a small current (less than 13μA) will load the INTVCC
in shutdown. This action allows the LT3761A to operate
from VIN as low as 4.5V. If VIN is normally above, but
occasionally drops below the INTVCC regulation voltage,
then the minimum operating VIN will be close to 5V. This
value is determined by the dropout voltage of the linear
regulator and the INTVCC undervoltage lockout threshold
mentioned above.
Programming the Turn-On and Turn-Off Thresholds
with the EN/UVLO Pin
The power supply undervoltage lockout (UVLO) value can
be accurately set by the resistor divider to the EN/UVLO pin.
A small 2.3μA pull-down current is active when EN/UVLO
is below the threshold. The purpose of this current is to
LT3761A
R1
EN/UVLO
R2
3761A F01
Figure 1. Resistor Connection to Set
VIN Undervoltage Shutdown Threshold
LED Current Programming
The LED current is programmed by placing an appropriate
value current sense resistor, RLED, in series with the LED
string. The voltage drop across RLED is (Kelvin) sensed
by the ISP and ISN pins. A half watt resistor is usually
a good choice. To give the best accuracy, sensing of the
current should be done at the top of the LED string. If this
option is not available then the current may be sensed
at the bottom of the string, or in the source of the PWM
disconnect NFET driven by the PWMOUT signal. A unique
case of GND sensing is the inverting converter shown in
the applications where the LED current is sensed in the
cathode of the power Schottky rectifier. This configuration
allows the LED anode to be grounded for heat sinking. In
this case, it is important to lowpass filter the discontinuous current signal. Input bias currents for the ISP and ISN
inputs are shown in the typical performance characteristics
and should be considered when placing a resistor in series
with the ISP or ISN pins.
The CTRL pin should be tied to a voltage higher than 1.2V
to get the full-scale 250mV (typical) threshold across the
sense resistor. The CTRL pin can also be used to dim the
3761af
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LT3761A
Applications Information
LED current to zero, although relative accuracy decreases
with the decreasing voltage sense threshold. When the
CTRL pin voltage is less than 1V, the LED current is:
ILED =
VCTRL − 100mV
RLED • 4
50mV should not cause mis-operation, but may lead to
noticeable offset between the current regulation and the
user-programmed value.
Programming Output Voltage (Constant Voltage
Regulation) or Open LED/Overvoltage Threshold
When the CTRL pin voltage is between 1V and 1.2V the LED
current varies with CTRL, but departs from the previous
equation by an increasing amount as the CTRL voltage
increases. Ultimately, the LED current no longer varies for
CTRL ≥ 1.2V. At CTRL = 1.1V, the value of ILED is ~98% of
the equation’s estimate. Some values are listed in Table 1.
For a boost or SEPIC application, the output voltage can
be set by selecting the values of R3 and R4 (see Figure 2)
according to the following equation:
VOUT = 1.25 •
R3 + R4
R4
VOUT
Table 1. (ISP-ISN) Threshold vs CTRL
VCRTL (V)
(ISP-ISN) Threshold (mV)
1.0
225
1.05
236
1.1
244.5
1.15
248.5
1.2
250
When CTRL is higher than 1.2V, the LED current is regulated to:
ILED =
250mV
RLED
The CTRL pin should not be left open (tie to VREF if not
used). The CTRL pin can also be used in conjunction with
a thermistor to provide overtemperature protection for
the LED load, or with a resistor divider to VIN to reduce
output power and switching current when VIN is low.
The presence of a time varying differential voltage signal
(ripple) across ISP and ISN at the switching frequency
is expected. The amplitude of this signal is increased by
high LED load current, low switching frequency and/or a
smaller value output filter capacitor. Some level of ripple
signal is acceptable: the compensation capacitor on the
VC pin filters the signal so the average difference between
ISP and ISN is regulated to the user-programmed value.
Ripple voltage amplitude (peak-to-peak) in excess of
R3
LT3761A
FB
R4
3761A F02
Figure 2. Feedback Resistor Connection for
Boost or SEPIC LED Driver
For a boost type LED driver, set the resistor from the output
to the FB pin such that the expected voltage level during
normal operation will not exceed 1.17V. For an LED driver
of buck mode or a buck-boost mode configuration, the
output voltage is typically level-shifted to a signal with
respect to GND as illustrated in Figure 3. The output can
be expressed as:
VOUT = VBE + 1.25 •
R3
R4
R3
+
RSEN(EXT)
VOUT
–
LT3761A
100k
LED
ARRAY
COUT
3761A F03
FB
R4
Figure 3. Feedback Resistor Connection for
Buck Mode or Buck-Boost Mode LED Driver
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13
LT3761A
Applications Information
ISP/ISN Short-Circuit Protection Feature
The ISP/ISN pins have a protection feature independent
of their LED current sense feature. The purpose of this
feature is to prevent the development of excessive currents that could damage the power components or the
load. The action threshold (VISP-ISN > 600mV, typical) is
above the default LED current sense threshold, so that
no interference will occur with current regulation. This
feature acts in the same manner as switch current limit:
it prevents switch turn-on until the ISP/ISN difference
falls below the threshold. Exceeding the threshold also
activates a pull-down on the SS and PWM pins and causes
the GATE and PWMOUT pins to be driven low for at least
4µs. If an overcurrent condition is sensed at ISP/ISN and
the PWM pin is configured either to make an internal
dimming signal, or for always-on operation as shown in
the application titled Boost LED Driver with Output Short
Protection, then the LT3761A will enter a hiccup mode of
operation. In this mode, after the initial response to the
fault, the PWMOUT pin re-enables the output switch at an
interval set by the capacitor on the PWM pin. If the fault
is still present, the PWMOUT pin will go low after a short
delay (typically 7µs) and turn off the output switch. This
fault-retry sequence continues until the fault is no longer
present in the output.
PWM Dimming Control
There are two methods to control the current source for
dimming using the LT3761A. One method uses the CTRL
pin to adjust the current regulated in the LEDs. A second
method uses the PWM pin to modulate the current source
between zero and full current to achieve a precisely programmed average current. To make PWM dimming more
accurate, the switch demand current is stored on the VC
node during the quiescent phase when PWM is low. This
feature minimizes recovery time when the PWM signal
goes high. To further improve the recovery time, a disconnect switch may be used in the LED current path to
prevent the ISP node from discharging during the PWM
signal low phase.
The minimum PWM on or off time is affected by choice of
operating frequency and external component selection. The
data sheet application titled “Boost LED Driver for 30kHz
PWM Dimming” demonstrates regulated current pulses
as short as 3μs are achievable. The best overall combination of PWM and analog dimming capability is available if
the minimum PWM pulse is at least six switching cycles.
A low duty cycle PWM signal can cause excessive start-up
times if it were allowed to interrupt the soft-start sequence.
Therefore, once start-up is initiated by PWM > 1.3V, it will
ignore a logical disable by the external PWM input signal.
The device will continue to soft-start with switching and
PWMOUT enabled until either the voltage at SS reaches
the 1V level, or the output current reaches one-tenth of
the full-scale current. At this point the device will begin
following the dimming control as designated by PWM.
Disconnect Switch Selection
An NMOS in series with the LED string at the cathode is
recommended in most LT3761A applications to improve
the PWM dimming. The NMOS BVDSS rating should be as
high as the open LED regulation voltage set by the FB pin,
which is typically the same rating as the power switch of the
converter. The maximum continuous drain current ID(MAX)
rating should be higher than the maximum LED current.
A PMOS high side disconnect is needed for buck mode,
buck-boost mode or an output short circuit protected
boost. A level shift to drive the PMOS switch is shown
in the application schematic Boost LED Driver with Output Short Circuit Protection. In the case of a high side
disconnect follow the same guidelines as for the NMOS
regarding voltage and current ratings. It is important to
include a bypass diode to GND at the drain of the PMOS
switch to ensure that the voltage rating of this switch is
not exceeded during transient fault events.
PWM Dimming Signal Generator
The LT3761A features a PWM dimming signal generator
with programmable duty cycle. The frequency of the square
wave signal at PWMOUT is set by a capacitor CPWM from
the PWM pin to GND according to the equation:
fPWM = 14kHz • nF/CPWM
3761af
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LT3761A
Applications Information
The duty cycle of the signal at PWMOUT is set by a µA
scale current into the DIM/SS pin (see Figure 4 and the
Typical Performance Characteristics).
PWMOUT DUTY RATIO (%)
100
CPWM = 47nF
80
60
Internal PWM Oscillator Operation
40
The PWM oscillator operation is similar to a 555 timer
(astable multi-vibrator). However, the currents that charge
and discharge the capacitor are not directly proportional
to the controlling current.
20
0
digital signal from a microcontroller to obtain very high
dimming performance. The practical minimum duty cycle
using the internal signal generator is about 4% if the DIM/
SS pin is used to adjust the dimming ratio. Consult the
factory for techniques for and limitations of generating a
duty ratio less than 4% using the internal generator. For
always on operation, the PWM pin should be connected
as shown in the application Boost LED Driver with Output
Short Protection.
0
2
4
6
DIM VOLTAGE (V)
8
3761 F04
Figure 4. PWMOUT Duty Ratio vs DIM Voltage for RDIM = 140k
Internally generated pull-up and pull-down currents on
the PWM pin are used to charge and discharge its capacitor between the high and low thresholds to generate the
duty cycle signal. These current signals on the PWM pin
are small enough so they can be easily overdriven by a
IPULL-UP = F1(IDIM/SS) = 7.2μA • exp(0.062 • IDIM/SS)
IPULL-DOWN = F2(IDIM/SS) = 85μA • exp(–0.062 • IDIM/SS)
The negative sign in the exponential makes IPULL-DOWN
decrease when IDIM/SS increases.
0.8V + F3 (IDIM/SS)
F1 (IDIM/SS)
PWM
CPWM
0.8V
F2 (IDIM/SS)
FAULT
–
+
–
+
R
S
Q
PWMINT
15mA
VTH1 = 0.8 + F3 (IDIM/SS)
VPWM
dV = F3 (IDIM/SS)
VTH2 = 0.8V
t1
t2
dV/dt = IPULL-UP/CPWM
dV/dt = IPULL-DOWN /CPWM
VPWMINT
3761A F05
Figure 5. Internal PWM Oscillator Logic and Waveform
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15
LT3761A
Applications Information
Voltage on the external cap ramps up at dV/dt = IPULL-UP
/CPWM. When the PWM pin reaches the high threshold
(0.8V + F3(IDIM/SS)), the flip flop SETs and IPULL-UP goes
to zero and current IPULL-DOWN goes to F2(IDIM/SS).
T1
Duty Cycle =
T1+ T2
dV
T1=
⎛ IPULL−DOWN ⎞
⎜
⎟
⎝ CPWM ⎠
dV
T2 =
⎛ IPULL−UP ⎞
⎜
⎟
CPWM ⎠
⎝
⎛
Duty ⎞
⎟
IDIM/SS = 8.06 • ln ⎜⎜11.8 •
(1−Duty ) ⎟⎠
⎝
⎛
0.2 ⎞
= 8.06 • ln ⎜11.8 •
⎟ = 8.72µA
⎝
0.8 ⎠
VREF − 1.20
RDIM = −2.5kΩ +
IDIM/SS
After simplification, one can obtain the formula for duty
cycle of PWMOUT as a function of IDIM/SS:
Duty Cycle =
These equations can be worked in reverse starting with a
desired duty cycle using 20%, for example, and solving for
a resistor value, RDIM, placed between VREF and DIM/SS:
1
1+ 11.8 • exp(−0.124 •IDIM/SS )
To calculate the duty cycle of the internal PWM generator
given a voltage of the DIM signal, determine first the current
into the DIM/SS pin by the equation (referring to Figure 6):
= −2.5kΩ +
For some applications, a duty cycle lower than 3% is
desired. It is possible to achieve a discrete value of duty
cycle that is lower than range attainable using DIM/SS
current. A resistor, RPD, and switch driven by PWMOUT
can be added as shown in Figure 7.
LT3761A
DIM
RDIM
10nF
100%
Duty (in%) =
1+ 11.8 • exp(−0.124 •IDIM/SS )
LT3761A
RDIM
PWMOUT
DIM/SS
PWM
GND
10nF
PWM
RPD
47nF
300Hz
3761A F07
Knowing the IDIM/SS in μA , the duty cycle of the PWMOUT
pin can be calculated for the range –10μA < IDIM/SS < 55μA:
DIM
PWMOUT
DIM/SS
GND
V − 1.20V
IDIM/SS = DIM
in µA
R
+
2.5kΩ
DIM
2.015 − 1.20
= 90.9kΩ
0.00872
47nF
300Hz
3761A F06
Figure 6. Configuration of Dimming Resistor, RDIM
Figure 7. Configuration for Sub 4% PWM Dimming
The addition of this resistor increases the pull-down
current on PWM, thus decreasing the duration of the onphase of the switching regulator. Since PWM frequency
at low duty cycle is primarily determined by the pull-up
current, the additional pull-down current from RPD has
little effect on the PWM period, so frequency calculation
remains the same.
An example solving for RPD given a 1% duty cycle is provided below. For this example, the IDIM/SS current flowing
in RDIM is assumed zero, which normally provides an
~8% duty cycle. The average voltage on the PWM pin is
approximately 1.05V at this IDIM/SS setting.
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LT3761A
Applications Information
Duty =
=
by the fixed minimum on-time and the switching frequency
(fSW). The maximum duty cycle of the switch is limited
by the fixed minimum off-time and fSW. The following
equations express the minimum/maximum duty cycle:
IPULL−UP
IPULL−UP +IPULL−DOWN +IRPD
7.2
= 0.01
7.2+ 85+IRPD
IRPD = 629µA =
Min Duty Cycle = 220ns • fSW
1.05V
RPD
Max Duty Cycle = 1 – 170ns • fSW
300
250
Programming the Switching Frequency
200
The RT frequency adjust pin allows the user to program
the switching frequency (fSW) from 100kHz to 1MHz to
optimize efficiency/performance or external component
size. Higher frequency operation yields smaller component size but increases switching losses and gate driving
current, and may not allow sufficiently high or low duty
cycle operation. Lower frequency operation gives better
performance at the cost of larger external component
size. For an appropriate RT resistor value see Table 2. An
external resistor from the RT pin to GND is required—do
not leave this pin open.
Table 2. Switching Frequency (fSW) vs RT Value
fSW (kHz)
RT (kΩ)
100
95.3
200
48.7
300
33.2
400
25.5
500
20.5
600
16.9
700
14.3
800
12.1
900
10.7
1000
8.87
Duty Cycle Considerations
Switching duty cycle is a key variable defining converter
operation, therefore, its limits must be considered when
programming the switching frequency for a particular application. The minimum duty cycle of the switch is limited
TIME (ns)
Therefore, RPD ~ 1.65kΩ
CGATE = 3300pF
MINIMUM ON-TIME
MINIMUM OFF-TIME
150
100
50
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3761 F05
Figure 8. Typical Minimum On and Off
GATE Pulse Width vs Temperature
Besides the limitation by the minimum off-time, it is
also recommended to choose the maximum duty cycle
below 95%.
DBOOST =
VLED − VIN
VLED
DBUCK _ MODE =
VLED
VIN
DSEPIC , DCUK =
VLED
VLED + VIN
Thermal Considerations
The LT3761A is rated to a maximum input voltage of
60V. Careful attention must be paid to the internal power
dissipation of the IC at higher input voltages to ensure
that a junction temperature of 125°C is not exceeded. This
junction limit is especially important when operating at high
ambient temperatures. If LT3761A junction temperature
3761af
For more information www.linear.com/LT3761A
17
LT3761A
Applications Information
reaches 165°C, the GATE and PWMOUT pins will be driven
to GND and the soft-start (DIM/SS) and PWM pins will be
discharged to GND. Switching will be enabled after device
temperature is reduced 10°C. This function is intended to
protect the device during momentary thermal overload
conditions.
The majority of the power dissipation in the IC comes from
the supply current needed to drive the gate capacitance of
the external power MOSFET. This gate drive current can
be calculated as:
IGATE = fSW • QG
TJ = TA + [VIN (IQ + fSW • QG) • θJA]
For monitoring the LED string voltage, if the open LED
clamp voltage is programmed correctly using the FB
resistor divider then the FB pin should not exceed 1.18V
when LEDs are connected. If the OPENLED pull-down is
asserted and the PWM pin transitions low, the pull-down
will continue to be asserted until the next rising edge of
ILED
ISP
RLED
ISN
–
TJ= 85°C + [40V • (2mA + 400kHz • 20nC) • 43°C/W] =
102°C
The exposed pad on the bottom of the package must be
soldered to a ground plane. This ground should then be
connected to an internal copper ground plane with thermal
vias placed directly under the package to spread out the
heat dissipated by the IC.
Open LED Reporting – Constant Voltage Regulation
Status Pin
The LT3761A provides an open-drain status pin, OPENLED,
that pulls low when the FB pin is within 50mV of its 1.25V
regulated voltage AND output current sensed by VISP-ISN
has reduced to 25mV, or 10% of the full-scale value. The
10% output current qualification (C/10) is unique for an LED
driver but fully compatible with open LED indication – the
OPENLED
+
S
Q
1mA
R
1.2V
where TA is the ambient temperature, IQ is the quiescent
current of the part (maximum 2mA) and θJA is the package
thermal impedance (43°C/W for the MSE package). For
example, an application has TA(MAX) = 85°C, VIN(MAX) =
40V, fSW = 400kHz, and having a FET with QG = 20nC, the
maximum IC junction temperature will be approximately:
C10
COMPARATOR
25mV
+
–
A low QG power MOSFET should always be used when
operating at high input voltages, and the switching frequency should also be chosen carefully to ensure that
the IC does not exceed a safe junction temperature. The
internal junction temperature of the IC can be estimated by:
qualification is always satisfied since for an open load, zero
current flows in the load. The C/10 feature is particularly
useful in the case where OPENLED is used to indicate the
end of a battery charging cycle and terminate charging or
transition to a float charge mode.
FB
–
+
OPEN LED
COMPARATOR
PWM
1. OPENLED ASSERTS WHEN VISP-ISN < 25mV AND FB > 1.2V, AND IS LATCHED
2. OPENLED DE-ASSERTS WHEN FB < 1.19V, AND PWM LOGIC 1 = 1V
3. ANY FAULT CONDITION RESETS THE LATCH, SO LT3761 STARTS UP
WITH OPENLED DE-ASSERTED
3761 F06
Figure 9. OPENLED Logic Block Diagram
PWM even if FB falls below the OPENLED threshold.
Input Capacitor Selection
The input capacitor supplies the transient input current for
the power inductor of the converter and must be placed
and sized according to the transient current requirements.
The switching frequency, output current and tolerable input
voltage ripple are key inputs to estimating the capacitor
value. An X7R type ceramic capacitor is usually the best
choice since it has the least variation with temperature and
DC bias. Typically, boost and SEPIC converters require a
3761af
18
For more information www.linear.com/LT3761A
LT3761A
Applications Information
lower value capacitor than a buck mode converter. Assuming that a 100mV input voltage ripple is acceptable,
the required capacitor value for a boost converter can be
estimated as follows:
CIN (µF) = ILED (A) •
⎛ µF ⎞
VOUT
• t SW (µs) • ⎜
⎝ A • µs ⎟⎠
VIN
Therefore, a 10μF capacitor is an appropriate selection
for a 400kHz boost regulator with 12V input, 48V output
and 1A load.
With the same VIN voltage ripple of 100mV, the input capacitor for a buck converter can be estimated as follows:
⎛ µF ⎞
CIN (µF) = ILED (A) • t SW (µs) • 4.7 • ⎜
⎝ A • µs ⎟⎠
A 10μF input capacitor is an appropriate selection for a
400kHz buck mode converter with a 1A load.
In the buck mode configuration, the input capacitor has
large pulsed currents due to the current returned through
the Schottky diode when the switch is off. In this buck
converter case it is important to place the capacitor as close
as possible to the Schottky diode and to the GND return
of the switch (i.e., the sense resistor). It is also important
to consider the ripple current rating of the capacitor. For
best reliability, this capacitor should have low ESR and
ESL and have an adequate ripple current rating.
Table 3. Recommended Ceramic Capacitor Manufacturers
MANUFACTURER
WEB
TDK
www.tdk.com
Kemet
www.kemet.com
Murata
www.murata.com
Taiyo Yuden
www.t-yuden.com
Output Capacitor Selection
The selection of the output capacitor depends on the load
and converter configuration, i.e., step-up or step-down
and the operating frequency. For LED applications, the
equivalent resistance of the LED is typically low and the
output filter capacitor should be sized to attenuate the
current ripple. Use of X7R type ceramic capacitors is
recommended.
To achieve the same LED ripple current, the required filter
capacitor is larger in the boost and buck-boost mode applications than that in the buck mode applications. Lower
operating frequencies will require proportionately higher
capacitor values.
Soft-Start Capacitor Selection
For many applications, it is important to minimize the
inrush current at start-up. The built-in soft-start circuit
significantly reduces the start-up current spike and output
voltage overshoot. Connect a capacitor from the DIM/SS
pin to GND to use this feature. The soft-start interval is
set by the soft-start capacitor selection according to the
equation:
TSS = CSS •
1.2V
100µs
= CSS •
14µA
nF
provided there is no additional current supplied to the
DIM/SS pin for programming the duty cycle of the PWM
dimming signal generator. A typical value for the soft-start
capacitor is 10nF which gives a 1ms start-up interval. The
soft-start pin reduces the oscillator frequency and the
maximum current in the switch.
The soft-start capacitor discharges if one of the following events occurs: the EN/UVLO falls below its threshold;
output overcurrent is detected at the ISP/ISN pins; IC
overtemperature; or INTVCC undervoltage. During startup with EN/UVLO, charging of the soft-start capacitor is
enabled after the first PWM high period. In the start-up
sequence, after switching is enabled by PWM the switching
continues until VISP-ISN > 25mV or DIM/SS > 1V. PWM
pin negative edges during this start-up interval are not
processed until one of these two conditions are met so
that the regulator can reach steady state operation shortly
after PWM dimming commences.
Power MOSFET Selection
The selection criteria for the power MOSFET includes
the drain-source breakdown voltage (VDS), the threshold
voltage (VGS(TH)), the on-resistance (RDS(ON)), the gate
to source and gate to drain charges (QGS and QGD), the
maximum drain current (ID(MAX)) and the MOSFET’s
thermal resistances (RθJC, RθJA).
3761af
For more information www.linear.com/LT3761A
19
LT3761A
Applications Information
For applications operating at high input or output voltages,
the power switch is typically chosen for drain voltage VDS
rating and low gate charge QG. Consideration of switch
on-resistance, RDS(ON), is usually secondary because
switching losses dominate power loss. The INTVCC regulator on the LT3761A has a fixed current limit to protect the
IC from excessive power dissipation at high VIN, so the FET
should be chosen so that the product of QG at 7.85V and
switching frequency does not exceed the INTVCC current
limit. For driving LEDs be careful to choose a switch with
a VDS rating that exceeds the threshold set by the FB pin
in case of an open-load fault. The required power MOSFET
VDS rating of different topologies can be estimated using
the following equations plus a diode forward voltage, and
any additional ringing across its drain-to-source during
its off-time.
section on power MOSFET selection. If using the PWM
feature for dimming, it may be important to consider diode
leakage, which increases with the temperature, from the
output during the PWM low interval. Therefore, choose
the Schottky diode with sufficiently low leakage current.
Table 5 has some recommended component vendors. The
diode current and VF should be considered when selecting the diode to be sure that power dissipation does not
exceed the rating of the diode. The power dissipated by
the diode in a converter is:
Boost: VDS > VLED
MANUFACTURER
WEB
Buck Mode: VDS > VIN(MAX)
Vishay
www.vishay.com
Central Semiconductor
www.centralsemi.com
Diodes, Inc.
www.diodes.com
SEPIC, Inverting: VDS > VIN(MAX) + VLED
Since the LT3761A gate driver is powered from the 7.85V
INTVCC, the 6V rated MOSFET works well for all the
LT3761A applications.
It is prudent to measure the MOSFET temperature in
steady state to ensure that absolute maximum ratings
are not exceeded.
Several MOSFET vendors are listed in Table 4. The MOSFETs
used in the application circuits in this data sheet have been
found to work well with the LT3761A. Consult factory applications for other recommended MOSFETs.
Table 4. Recommended Power MOSFET Manufacturers
MANUFACTURER
WEB
Vishay Siliconix
www.vishay.com
Infineon
www.infineon.com
Renesas
www.renesas.com
PD = ID • VF • (1 – DMAX)
It is prudent to measure the diode temperature in steady
state to ensure that its absolute maximum ratings are not
exceeded.
Table 5. Schottky Rectifier Manufacturers
Sense Resistor Selection
The resistor, RSENSE, between the source of the external
NMOS FET and GND should be selected to provide adequate
switch current to drive the application without exceeding
the 105mV (typical) current limit threshold on the SENSE
pin of LT3761A. For a boost converter, select a resistor
value according to:
RSENSE,BOOST ≤
VIN • 0.07V
VLED • ILED
For buck-boost mode and SEPIC, select a resistor according to:
RSENSE,BUCK-BOOST ≤
VIN • 0.07V
( VIN + VLED )ILED
For buck mode, select a resistor according to:
Schottky Rectifier Selection
The power Schottky diode conducts current during the
interval when the switch is turned off. Select a diode
rated for the maximum SW voltage as described in the
RSENSE,BUCK ≤
0.07V
ILED
3761af
20
For more information www.linear.com/LT3761A
LT3761A
Applications Information
These equations provide an estimate of the sense resistor
value based on reasonable assumptions about inductor current ripple during steady state switching. Lower
values of sense resistor may be required in applications
where inductor ripple current is higher. Examples include
applications with current limited operation at high duty
cycle, and those with discontinuous conduction mode
(DCM) switching. It is always prudent to verify the peak
inductor current in the application to ensure the sense
resistor selection provides margin to the SENSE current
limit threshold.
The placement of RSENSE should be close to the source of
the NMOS FET and GND of the LT3761A. The SENSE input
to LT3761A should be a Kelvin connection to the positive
terminal of RSENSE. Verify the power on the resistor to
ensure that it does not exceed the rated maximum.
Inductor Selection
The inductor used with the LT3761A should have a saturation current rating appropriate to the maximum switch
current selected with the RSENSE resistor. Choose an inductor value based on operating frequency, input and output
voltage to provide a current mode ramp on SENSE during
the switch on-time of approximately 20mV magnitude.
The following equations are useful to estimate the inductor value for continuous conduction mode operation (use
the minimum value for VIN and maximum value for VLED):
LBUCK =
RSENSE • VLED ( VIN – VLED )
VIN • 0.02V • fOSC
LBUCK-BOOST =
LBOOST =
Table 6 provides some recommended inductor vendors.
Table 6. Recommended Inductor Manufacturers
MANUFACTURER
WEB
Coilcraft
www.coilcraft.com
Cooper-Coiltronics
www.cooperet.com
Wurth-Midcom
www.we-online.com
Vishay
www.vishay.com
Loop Compensation
The LT3761A uses an internal transconductance error
amplifier whose VC output compensates the control loop.
The external inductor, output capacitor and the compensation resistor and capacitor determine the loop stability.
The inductor and output capacitor are chosen based on
performance, size and cost. The compensation resistor
and capacitor at VC are selected to optimize control loop
response and stability. For typical LED applications, a 4.7nF
compensation capacitor at VC is adequate, and a series
resistor should always be used to increase the slew rate
on the VC pin to maintain tighter regulation of LED current
during fast transients on the input supply to the converter.
The DC-Coupling Capacitor Selection for SEPIC
LED Driver
The DC voltage rating of the DC-coupling capacitor CDC
connected between the primary and secondary inductors of
a SEPIC should be larger than the maximum input voltage:
VCDC > VIN(MAX)
RSENSE • VLED • VIN
( VLED + VIN ) • 0.02V • fOSC
RSENSE • VIN ( VLED – VIN )
VLED • 0.02V • fOSC
Use the equation for Buck-Boost when choosing an inductor value for SEPIC – if the SEPIC inductor is coupled,
then the equation’s result can be used as is. If the SEPIC
uses two uncoupled inductors, then each should have a
inductance double the result of the equation.
CDC has nearly a rectangular current waveform. During
the switch off-time, the current through CDC is IVIN, while
approximately –ILED flows during the on-time. The CDC
voltage ripple causes current distortions on the primary
and secondary inductors. The CDC should be sized to limit
its voltage ripple. The power loss on the CDC ESR reduces
the LED driver efficiency. Therefore, the sufficient low ESR
ceramic capacitors should be selected. The X5R or X7R
ceramic capacitor is recommended for CDC.
Short-Circuit Protection for a Boosted Output
The LT3761A has two features that provide protection
from a shorted circuit load on a boost. The first of these
3761af
For more information www.linear.com/LT3761A
21
LT3761A
Applications Information
is the ISP/ISN based overcurrent response. The second is
the FB overvoltage response. The primary mode of action
for both features is to drive the PWMOUT pin low, which
turns off the switch connecting the output to the load. The
ISP/ISN short-circuit protection also drives the PWM and
DIM/SS pins low for a brief period of time. For best protection, a PMOS disconnect switch M1 is placed as shown in
Figure 10. During an overcurrent event caused by a short
across the LED string, the current in RS increases until PNP
Q1 turns on and pulls up the gate of M1, throttling back
the current. In approximately 1µs, the ISP/ISN overcurrent
response will cause the PWMOUT pin to drive low, which
will turn off M1 altogether. If an external PWM signal is
used, then the circuit including Q3, the 1N4148 diode and
two resistors must be used to ensure the switch remains
off while the output is in a faulted state. This sub-circuit
drives the FB pin into the overvoltage state.
If the PWM pin is configured (with a capacitor load) as
shown in the application titled Boost LED Driver with
Output Short Protection, then the small circuit driving
FB may be omitted. In this case, the boost converter will
demonstrate a hiccup mode response, turning on M1 at an
interval determined by the PWM capacitor, then turning off
after ~1µs due to excessive current, until the fault clears.
Board Layout
The high speed operation of the LT3761A demands careful attention to board layout and component placement.
Figure 11 provides a simplified layout for the boost converter. The exposed pad of the package is the only GND
terminal of the IC and is also important for its thermal
management. It is crucial to achieve a good electrical and
thermal contact between the exposed pad and the ground
plane of the board. To reduce electromagnetic interference
(EMI), it is important to minimize the area of the high dV/
dt switching node between the inductor, switch drain and
anode of the anode of the Schottky rectifier. Use a ground
plane under the switching node to eliminate interplane
coupling to sensitive signals.
Proper layout of the power paths with high di/dt is essential to robust converter operation. The following high
di/dt loops of different topologies should be kept as tight
as possible to reduce inductive ringing:
1. In boost configuration, the high di/dt loop of each channel contains the output capacitor, the sensing resistor,
the power NMOS and the Schottky diode.
2. In buck mode configuration, the high di/dt loop of each
channel contains the input capacitor, the sensing resistor,
the power NMOS and the Schottky diode.
RS
0.5Ω
D1
COUT
PGND
LT3761A
1k
Q1
2k
ISP
M1
ISN
0.15nF
1M
PWMOUT
20k
Q2
FB
Q3
24.9k
GND
INTVCC
INTVCC
1µF
2.2k
27k
1N4148
1k
D2
3955 F10
Figure 10. Protection Circuit for Fault to Ground on LED Load. Includes Fast Level Shift for PWM Switch M1
3761af
22
For more information www.linear.com/LT3761A
LT3761A
Applications Information
3. In buck-boost mode configuration, the high di/dt loop of
each channel contains the capacitor connecting between
VOUT and GND, the sensing resistor, the power NMOS
and the Schottky diode.
switch being closest to the IC, along with the INTVCC bypass capacitor. The ground for the compensation network
(VC) and other DC control signals (e.g., FB, PWM, DIM/SS,
CTRL) should be star connected to the underside of the
IC. Do not extensively route high impedance signals such
as FB and VC, as they may pick up switching noise. In particular, avoid routing FB and PWMOUT in parallel for more
than a few millimeters on the board. Minimize resistance
in series with the SENSE input to avoid changes (most
likely reduction) to the switch current limit threshold. The
best overall results can be obtained using a 4-layer board,
and design files for a proven board design specifically for
this product can be downloaded from demo.linear.com.
4. In SEPIC configuration, the high di/dt loop contains the
power NMOS, sense resistor, output capacitor, Schottky
diode and the DC-coupling capacitor.
The ground terminal of the switch current sense resistor
should Kelvin connect to the GND of the LT3761A. Likewise,
the ground terminal of the bypass capacitor for the INTVCC
regulator should be placed near the GND of the switching
path. Typically this requirement will result in the external
PGND AGND
DIM
RDIM
VREF CTRL
OPENLED
CPWM
CSS
RT
R2
R1
L1
LAYER 2
GROUND
PLANE
SPLIT
CVCC
VIAS TO GROUND PLANES
8
7
6
5
4
3
2
1
9
10
11
12
13
14
15
16
SENSE VIA x
CC
RC
x VOUT VIA
R3
x
R4
x
AGND
PGND
5
4
1
6
3
M2
M1
7
2
8
1
3
LED–
2
RSENSE
COUT
COUT
D1
CIN
LED+
RLED
VIN
GND
3761A F07
COMPONENT DESIGNATIONS REFER TO BOOST LED DRIVER FOR AUTOMOTIVE HEADLAMP SCHEMATIC
Figure 11. Simplified 2-Layer Layout of the Boost LED Driver for Automotive Headlamp in the Typical Applications Section
3761af
For more information www.linear.com/LT3761A
23
LT3761A
Typical Applications
94% Efficient Boost LED Driver for Automotive Headlamp with 25:1 PWM Dimming
L1
10µH
CIN
2.2µF
×2
100V
R1
499k
R2
90.9k
EN/UVLO
VREF
1M
VIN
SENSE
RSENSE
10mΩ
LT3761A
RLED
0.25Ω
100
R4
16.9k
60W
LED
STRING
92
1.2
88
1.0
OUTPUT CURRENT
84
RT
28.7k
350kHz
RC
5.1k
CC
4.7nF
1.4
EFFICIENCY
OUTPUT CURRENT (A)
FB
1.6
PWM TIED TO INTVCC
96
OPENLED
DIM/SS
PWM
PWMOUT
VC
RT INTVCC
CPWM
47nF
300Hz
Boost Efficiency and Output Current vs VIN
1A
ISP
ISN
RDIM
140k
CSS
0.01µF
R3
1M
GND
140k
100k
DIM
COUT
2.2µF
×4
M1
GATE
CTRL
INTVCC
D1
EFFICIENCY (%)
VIN
8V TO
60V
0.8
INTVCC
CVCC
1µF
80
M2
0
10
20
30
VIN (V)
40
50
0.6
60
3761A TA02b
3761A TA02a
M1: INFINEON BSC123N08NS3-G
D1: DIODES INC PDS5100
L1: COILTRONICS HC9-100-R
M2: VISHAY SILICONIX Si2328DS
COUT, CIN: MURATA GRM42-2X7R225K100R
(CURRENT DERATED FOR VIN < 10V)
SEE SUGGESTED LAYOUT FIGURE 7
Boost LED Driver with Output Short-Circuit Protection with Externally Driven PWM
VIN
8V TO
60V
L1
10µH
2.2µF
×2
100V
499k
90.9k
INTVCC
EN/UVLO
VREF
1M
VIN
LT3761A
100k
10mΩ
Q1
2.4k
M2
150pF
1M
PWM
1A
PWMOUT
DIM/SS
10nF
1k
ISP
ISN
OPENLED
FB
RT
5.1k
M1
GND
140k
VC
0.25Ω
2.2µF
×4
100V
SENSE
CTRL
1k
GATE
D1
INTVCC
27k
INTVCC
16.9k
D2
60W
LED
STRING
1k
Q3
28.7k
350kHz
Q2
27k
4.7nF
1µF
2.2k
1N4148
3761A TA10
M1: INFINEON BSC123NO8NS3-G
D1: DIODES INC PDS5100
L1: COILTRONICS HC9-100-R
M2: VISHAY SILICONIX Si7113DN
D2: VISHAY 10BQ100
Q1, Q3: CENTRAL CMPT3906
Q2: ZETEX FMMT493
3761af
24
For more information www.linear.com/LT3761A
LT3761A
Typical Applications
Boost LED Driver with Output Short-Circuit Protection with Internally Generated PWM
L1
10µH
VIN
8V TO
60V
2.2µF
×2
100V
499k
90.9k
INTVCC
OPTION FOR
INTERNALLY GENERATED
PWM DIMMING
DIM
EN/UVLO
VREF
1M
VIN
GATE
D1
0.25Ω
2.2µF
×4
100V
M1
SENSE
LT3761A
CTRL
10mΩ
1k
100k
2.4k
ISP
ISN
OPENLED
M2
1M
140k
DIM/SS
10nF
150pF
RT
PWMOUT
INTVCC
28.7k
350kHz
5.1k
4.7nF
ILED
1A
LED+
FB
PWM
VC
22nF
640Hz
Q1
GND
140k
16.9k
D2
20k
Q2
INTVCC
1µF
1k
60W
LED
STRING
3761A TA09a
CURRENT DERATED
FOR VIN < 10V
M1: INFINEON BSC123NO8NS3-G
D1: DIODES INC PDS5100
L1: COILTRONICS HC9-100-R
M2: VISHAY SILICONIX Si7113DN
D2: VISHAY 10BQ100
Q1: CENTRAL CMPT3906
Q2: ZETEX FMMT493
1N4148 28k
OPTIONAL CIRCUIT
FOR ALWAYS-ON
OPERATION
High Side Disconnect Internally Generated
PWM Dimming Waveform
Output Short-Circuit Waveform Showing Hiccup
Mode Operation with Internally Generated PWM
PWMOUT
DIM = 8V
10V/DIV
VIN = 24V, VLED = 60V, DIM = 0V
PWMOUT
10V/DIV
ILED+
2A/DIV
52V
ILED
0.5A/DIV
VLED+
10µs/DIV
3761A TA09b
LED+ SHORT TO GND
0V
1ms/DIV
3761AA TA09c
3761af
For more information www.linear.com/LT3761A
25
LT3761A
Typical Applications
10W Grounded Anode Inverting LED Driver
3
100k
EN/UVLO
VIN
GATE
•
4.7µF
25V
•
VIN
5V TO
18V
2.2µF
×2
35V
L1
4.7µH
1:1
1
10Ω
2
LT3761A
0.01µF
10mΩ
OPENLED
ISN
FB
ISP
4.7µF
10V
–10V
CLAMP
10Ω
M2
0.47µF
DIM/SS
VREF
INTVCC
CTRL
0.1µF
91mΩ
GND
PWM
VC
1µF
D1
M1
SENSE
34k
1k
4
RT
59k
IN4148
LED–
2k
10nF
2.5A
10W
LED
PWMOUT
2k
28.7k
350kHz
Q1
20k
M1: VISHAY SILICONIX Si4162DY (30V)
D1: DIODES PDS1040 CTL
L1: WÜRTH 744870004
M2: VISHAY SILICONIX Si2312BDS (20V)
LED: CREE XLAMP XM-L
Q1: ZETEX FMMT593
3761A TA04a
PWM Dimming Waveform
PWM
10V/DIV VIN = 12V, VLED = –3.6V
ILED
1A/DIV
VLED–
2V/DIV
10µs/DIV
3761A TA04b
3761af
26
For more information www.linear.com/LT3761A
LT3761A
Typical Applications
40W SEPIC LED Driver
C1
2.2µF
×2
50V
•
VIN
8V TO
40V
C2
2.2µF
×2
50V
L1
10µH
1:1
3
499k
EN/UVLO
VREF
90.9k
1M
VIN
GATE
•2
C3
10µF
4 ×5
35V
M1
SENSE
LT3761A
CTRL
INTVCC
1
D1
1M
8mΩ
100k
1.67A
49.9k
GND
133k
0.15Ω
FB
1k
OPENLED
PWM
DIM/SS
VC
10nF
10k
40W
LED
STRING
ISP
ISN
RT
PWMOUT
INTVCC
28.7k
350kHz
INTVCC
1µF
4.7nF
M2
3761A TA05a
M1: INFINEON BSC123NO8
D1: DIODES INC PDS5100
L1: COILCRAFT MSD1278-103
M2: VISHAY SILICONIX Si2306BDS
LED: CREE XLAMP XM-L (×7)
C1, C2: KEMET C1210C225K5
C3: TAIYO YUDEN UMK325BJ106M
SEPIC Efficiency,
Output Current vs VIN
2.1
100
1.8
OUTPUT CURRENT
1.5
92
EFFICIENCY
88
PWM
5V/DIV
1.2
OUTPUT CURRENT (A)
EFFICIENCY (%)
96
ILED
0.5A/DIV
100µs/DIV
3761A TA05c
0.9
84
80
PWM Dimming Waveform
0
10
20
VIN (V)
30
0.6
40
3761A TA05b
3761af
For more information www.linear.com/LT3761A
27
LT3761A
Typical Applications
Boost LED Driver for 30kHz PWM Dimming
L1
0.82µH
VIN
8V TO
20V
C1
10µF
25V
187k
VIN
1k
VREF
CTRL
SENSE
LT3761A
6mΩ
PWM
0.25Ω
Boost PWM Dimming Waveform
1A
PWM
5V/DIV
17.8k
FB
100k
OPENLED
DIM/SS
VC
0.1µF
499k
VIN = 16V, VLED = 30V
GND
INTVCC
1µF
C2
10µF
×2
35V
M1
GATE
EN/UVLO
39.2k
D1
ILED
0.5A/DIV
ISP
ISN
PWMOUT
5µs/DIV
RT
20.5k
500kHz
3k
4.7nF
L1: VISHAY IHLP2525CZ
D1: DIODES PDS1040CTL
M1: INFINEON BSC059N04
M2: VISHAY Si2318CDS
3761A TA06b
M2
3761A TA06a
Buck Mode 5A LED Driver for 40kHz PWM Dimming
VIN
44V TO 60V
340k
10k
ISP
VIN
EN/UVLO
LT3761A
50mΩ
2W
UP TO
8 LEDS
4V TO 40V
ISN
VREF
1k
5A
0.22µF
CTRL
PWMOUT
M2
PWM
D2
6.2V
1M
C1
4.7µF
×4
50V
INTVCC
1µF
100k
2.2nF
FB
DIM/SS
VC
0.1µF
22k
200k
GATE
OPENLED
GND SENSE
RT
158k
VIN = 48V, VLED = 38V
ILED
2A/DIV
L1
1µH
Q1
10k
5µs/DIV
M1
VISHAY
PWM
10V/DIV
200k
16.9k
600kHz
4.7nF
Buck Mode PWM Dimming Waveform
5mΩ
0.5W
3761A TA07b
D1
C2
2.2µF
×2
100V
3761A TA07a
L1: WÜRTH 744331010
M1: INFINEON BSC123NO8NS3
M2: INFINEON BSC093NO4LS
D1: DIODES PDS5100
D2: CENTRAL SEMI CMSSH-3S
C1: TDK C4532X7R1H475
C2: TDK C3225X7R2A225
LED: CREE XLAMP XM-L (×7)
Q1: ZETEX FMMT593
3761af
28
For more information www.linear.com/LT3761A
LT3761A
Package Description
Please refer to http://www.linear.com/product/LT3761A#packaging for the most recent package drawings.
MSE Package
16-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1667 Rev F)
BOTTOM VIEW OF
EXPOSED PAD OPTION
2.845 ±0.102
(.112 ±.004)
5.10
(.201)
MIN
2.845 ±0.102
(.112 ±.004)
0.889 ±0.127
(.035 ±.005)
8
1
1.651 ±0.102
(.065 ±.004)
1.651 ±0.102 3.20 – 3.45
(.065 ±.004) (.126 – .136)
0.305 ±0.038
(.0120 ±.0015)
TYP
16
0.50
(.0197)
BSC
4.039 ±0.102
(.159 ±.004)
(NOTE 3)
RECOMMENDED SOLDER PAD LAYOUT
0.254
(.010)
0.35
REF
0.12 REF
DETAIL “B”
CORNER TAIL IS PART OF
DETAIL “B” THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
9
NO MEASUREMENT PURPOSE
0.280 ±0.076
(.011 ±.003)
REF
16151413121110 9
DETAIL “A”
0° – 6° TYP
3.00 ±0.102
(.118 ±.004)
(NOTE 4)
4.90 ±0.152
(.193 ±.006)
GAUGE PLANE
0.53 ±0.152
(.021 ±.006)
DETAIL “A”
1.10
(.043)
MAX
0.18
(.007)
SEATING
PLANE
0.17 – 0.27
(.007 – .011)
TYP
1234567 8
0.50
(.0197)
BSC
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL
NOT EXCEED 0.254mm (.010") PER SIDE.
0.86
(.034)
REF
0.1016 ±0.0508
(.004 ±.002)
MSOP (MSE16) 0213 REV F
3761af
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representaFor more
www.linear.com/LT3761A
tion that the interconnection
of itsinformation
circuits as described
herein will not infringe on existing patent rights.
29
LT3761A
Typical Application
HV Boost Efficiency and
LED Current vs VIN
100
80W High Voltage Boost LED Driver with 25:1 Internally Generated PWM Dimming
L1
22µH
187k
21k
EN/UVLO
VREF
590k
VIN
GATE
SENSE
LT3761A
CTRL
INTVCC
COUT
0.22µF
×8
250V
20k
1M
0805
80W
LED
STRING
135V MAX
8.66k
FB
0.1µF
10nF
1.4kHz
2k
0.6
LED CURRENT
88
0.4
84
0.2
80
PWMOUT
OPENLED
DIM/SS
ISP
PWM
ISN
VC
RT INTVCC
140k
92
12mΩ
GND
100k
DIM
0V TO
8V
M1
EFFICIENCY
EFFICIENCY (%)
2.2µF
×2
100V
0.8
96
D1
39.2k
250kHz
20
M2
M1, M2: INFINEON BSC520N15
D1: DIODES PDS4150
L1: COILTRONICS HC9-220-R
COUT: TDK C3225X7R2E224K
0.39Ω
3761A TA03a
50
0.6
60
3761A TA03b
PWM
(1V/DIV) DIM = 4V
VIN = 36V
VLED = 134V
650mA
CURRENT DERATED
FOR VIN < 35V
40
30
VIN (V)
Dimming Waveform
INTVCC
1µF
4.7nF
10
LED CURRENT (A)
VIN
12V TO
60V
1.0
PWM TIED TO INTVCC
ILED
(0.2A/DIV)
200µs/DIV
3761A TA03c
Related Parts
PART NUMBER
DESCRIPTION
COMMENTS
LT3761
High Side 60V, 1MHz LED Controller with 3000:1 PWM Dimming
VIN: 4.5V to 60V, VOUT(MAX) = 75V, 3000:1 PWM Dimming
ISD < 1μA, MSOP-16E Package
LT3755/LT3755-1/ High Side 40V, 1MHz LED Controller with 3000:1 PWM Dimming
LT3755-2
VIN: 4.5V to 40V, VOUT(MAX) = 75V, 3000:1 PWM Dimming
ISD < 1μA, 3mm × 3mm QFN-16 and MSOP-16E Packages
LT3756/LT3756-1/ High Side 100V, 1MHz LED Controller with 3000:1 PWM Dimming
LT3756-2
VIN: 6V to 100V, VOUT(MAX) = 100V, 3000:1 PWM Dimming
ISD < 1μA, 3mm × 3mm QFN-16 and MSOP-16E Packages
LT3796
High Side 100V, 1MHz LED Controller with 3000:1 PWM Dimming,
PMOS Disconnect FET Driver, Input Current Limit and Input/Output
Current Reporting
VIN: 6V to 100V, VOUT(MAX) = 100V, 3000:1 PWM Dimming
ISD < 1μA, TSSOP-28E Packages
LT3956
High Side 80V, 3.5A, 1MHz LED Driver with 3,000:1 PWM Dimming
VIN: 6V to 80V, VOUT(MAX) = 80V, PWM Dimming = 3000:1,
ISD < 1μA, 5mm × 6mm QFN-36 Package
LT3754
60V, 1MHz Boost 16-Channel 40mA LED Driver with 3000:1 PWM
Dimming and 2% Current Matching
VIN: 4.5V to 40V, VOUT(MAX) = 60V, PWM Dimming = 3000:1,
ISD < 1μA, 5mm × 5mm QFN-32 Package
LT3518
2.3A, 2.5MHz High Current LED Driver with 3000:1 Dimming with
PMOS Disconnect FET Driver
VIN: 3V to 30V, VOUT(MAX) = 45V, 3000:1 PWM Dimming,
ISD < 1μA, 4mm × 4mm QFN-16 and TSSOP-16E Packages
LT3478/LT3478-1
4.5A, 2MHz High Current LED Driver with 3000:1 Dimming
VIN: 2.8V to 36V, VOUT(MAX) = 40V, 3000:1 PWM Dimming,
ISD < 1μA, TSSOP-16E Package
LT3791/LT3791-1
60V, Synchronous Buck-Boost 700kHz LED Controller
VIN: 4.7V to 60V, VOUT Range: 0V to 60V, PWM,
Analog = 100:1, ISD < 1µA, TSSOP-38E Package
3761af
30
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
For more information www.linear.com/LT3761A
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com/LT3761A
LT 0516 • PRINTED IN USA
 LINEAR TECHNOLOGY CORPORATION 2016