LTC3536 - 1A Low Noise, Buck-Boost DC/DC Converter

LTC3536
1A Low Noise, Buck-Boost
DC/DC Converter
Features
Description
Regulated Output with Input Voltage Above, Below
or Equal to the Output Voltage
n 1.8V to 5.5V Input and Output Voltage Range
n 1A Continuous Output Current for
VIN ≥ 3V, VOUT = 3.3V
n ±1% Output Voltage Accuracy
n Low Noise Buck-Boost Architecture
n Up to 95% Efficiency
n Programmable Frequency from 300kHz to 2MHz
n Synchronizable Oscillator
n Burst Mode® Operation: 32µA I
Q
nInternal 1ms Soft-Start
n Output Disconnect in Shutdown
n Shutdown Current: 1µA
n Short-Circuit Protection
n Small Thermally Enhanced 12-Pin MSOP
and 10-Pin (3mm × 3mm) DFN Packages
The LTC®3536 is an extended VIN range, fixed frequency,
synchronous buck-boost DC/DC converter that operates
from input voltages above, below or equal to the regulated
output voltage. The topology incorporated in the LTC3536
provides low noise operation, making it ideal for RF and
precision measurement applications.
n
Applications
n
n
n
n
The device can produce up to 1A of continuous output
current, and it includes two N-channel and two P-channel
MOSFET switches. Switching frequencies up to 2MHz can
be programmed with an external resistor and the oscillator can be synchronized to an external clock. Quiescent
current is only 32µA in Burst Mode operation, maximizing
battery life in portable applications. Burst Mode operation
is user controlled and improves efficiency at light loads.
Other features include a 1µA shutdown current, internal
soft-start, overtemperature protection and current limit.
The LTC3536 is available in 12-pin thermally enhanced
MSOP and 10-pin (3mm × 3mm) DFN packages.
L, LT, LTC, LTM, Burst Mode, LTspice, Linear Technology and the Linear logo are registered
trademarks and PowerPath and No RSENSE are trademarks of Linear Technology Corporation. All
other trademarks are the property of their respective owners.
Wireless Inventory Terminals
Handheld Medical Instruments
Wireless Locators, Microphones
Supercapacitor Backup Power Supply
Typical Application
Efficiency vs Input Voltage
100
4.7µH
SW2
VOUT
47pF
LTC3536
10µF
PWM BURST
OFF ON
MODE/SYNC FB
SHDN
RT
VC
PGND SGND
VOUT
3.3V
1A FOR VIN ≥ 3V
6.49k
49.9k
1000k
220pF
22µF
90
EFFICIENCY (%)
SW1
VIN
VIN
1.8V TO 5.5V
ILOAD = 200mA
95
85
ILOAD = 1A
80
75
70
65
60
221k
55
3536 TA01a
50
1.5
2
2.5
3 3.5
4 4.5
INPUT VOLTAGE (V)
5
5.5
3536 TA01b
3536fa
1
LTC3536
Absolute Maximum Ratings (Note 1)
VIN, VOUT, (SVIN, PVIN) Voltage..................... –0.3V to 6V
SW1, SW2 Voltage
DC............................................................. –0.3V to 6V
Pulsed (<100ns).........................................–1.0V to 7V
VC, RT, FB, SHDN Voltage............................. –0.3V to 6V
MODE/SYNC Voltage.................................... –0.3V to 6V
Operating Junction Temperature Range
(Notes 2, 3)............................................. –40°C to 125°C
Storage Temperature Range................... –65°C to 150°C
Lead Temperature (Soldering, 10 sec)
MSE................................................................... 300°C
Pin Configuration
TOP VIEW
RT
1
SGND
2
MODE/SYNC
3
SW1
4
SW2
5
TOP VIEW
RT
SGND
MODE/SYNC
SW1
PGND
SW2
10 VC
11
PGND
9 FB
8 SHDN
7 VIN
6 VOUT
DD PACKAGE
10-LEAD (3mm × 3mm) PLASTIC DFN
TJMAX = 125°C, θJA = 39.7°C/W
EXPOSED PAD (PIN 11) IS PGND, MUST BE SOLDERED TO PCB
1
2
3
4
5
6
13
PGND
12
11
10
9
8
7
VC
FB
SHDN
VIN
VIN
VOUT
MSE PACKAGE
12-LEAD PLASTIC MSOP
TJMAX = 125°C, θJA = 40°C/W
EXPOSED PAD (PIN 13) IS PGND MUST BE SOLDERED TO PCB
FOR RATED THERMAL PERFORMANCE AND LOAD REGULATION
Order Information
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3536EDD#PBF
LTC3536EDD#TRPBF
LFZD
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LTC3536IDD#PBF
LTC3536IDD#TRPBF
LFZD
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LTC3536EMSE#PBF
LTC3536EMSE#TRPBF
3536
12-Lead Plastic MSOP
–40°C to 125°C
LTC3536IMSE#PBF
LTC3536IMSE#TRPBF
3536
12-Lead Plastic MSOP
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
Electrical Characteristics
The l denotes the specifications which apply over the full operating junction
temperature range, otherwise specifications are at TA = 25°C (Note 2). VIN = 3.3V, VOUT = 3.3V, RT = 100kΩ unless otherwise noted.
PARAMETER
CONDITIONS
Input Operating Range
MIN
l
1.8
TYP
MAX
UNITS
5.5
V
l
1.8
5.5
V
Undervoltage Lockout Threshold
VIN Ramping Down
VIN Ramping Up
l
l
1.6
1.67
1.75
1.8
V
V
Feedback Voltage
0°C < TJ < 85°C (Note 5)
–40°C < TJ < 125°C
l
0.594
0.591
0.6
0.6
0.606
0.609
V
V
Output Voltage Adjust Range
Feedback Pin Input Current (FB)
VFB = 0.6V in Servo Loop, VMODE/SYNC = 0V
50
nA
Quiescent Current, Burst Mode Operation
VFB = 0.7V, VMODE/SYNC = VIN
32
42
µA
Quiescent Current, Shutdown (IVIN)
VSHDN = 0V
0.1
1
2
µA
3536fa
LTC3536
Electrical
Characteristics
The l denotes the specifications which apply over the full operating junction
temperature range, otherwise specifications are at TA = 25°C (Note 2). VIN = 3.3V, VOUT = 3.3V, RT = 100kΩ unless otherwise noted.
PARAMETER
CONDITIONS
Quiescent Current, Active (IVIN)
VFB = 0.7V, VMODE/SYNC = 0V
Input Current Limit
VMODE/SYNC = 0V (Note 4)
l
MIN
TYP
2
2.5
3.4
MAX
UNITS
800
µA
A
Peak Current Limit
VMODE/SYNC = 0V (Note 4)
Burst Mode Peak Current Limit
VMODE/SYNC = VIN (Note 4)
Reverse Current Limit
(Note 4)
NMOS Switch Leakage
Switch B, C: SW1 = SW2 = 5.5V, VIN = 5.5V, VOUT = 5.5V
0.1
1
µA
PMOS Switch Leakage
Switch A, D: SW1 = SW2 = 0V, VIN = 5.5V, VOUT = 5.5V
0.1
1
µA
NMOS Switch On-Resistance
Switch B (From SW1 to GND) (Note 6)
Switch C (From SW2 to GND) (Note 6)
0.11
0.1
Ω
Ω
PMOS Switch On-Resistance
Switch A (From VIN to SW1) (Note 6)
Switch D (From VOUT to SW2) (Note 6)
0.12
0.145
Ω
Ω
Frequency Accuracy
RT = 100k
l
0.8
1
1.2
MHz
l
0.96
1.2
1.44
MHz
0.6
0.9
1.2
ms
88
91
l
Frequency Accuracy Default
RT = VIN
Internal Soft-Start Time
VFB from 0.06V to 0.54V
Maximum Duty Cycle
Percentage of Period SW2 is Low in Boost Mode
l
Minimum Duty Cycle
Percentage of Period SW1 is High in Buck Mode
l
4
A
0.4
0.6
A
0.3
0.55
A
%
0
Error Amplifier AVOL
%
90
dB
Error Amplifier Sink Current
FB = 1.3V, VC = 1V
250
300
µA
Error Amplifier Source Current
FB = 0.3V, VC = 0V
400
480
µA
MODE/SYNC Input Logic Threshold
Disable Burst Mode Operation
0.3
MODE/SYNC External Synchronization
SYNC Level High
SYNC Level Low
MODE/SYNC Synchronization Frequency
MODE/SYNC Input Current
1.2
l
0.3
VMODE/SYNC = 5.5V = VIN
SHDN Input Logic Threshold
SHDN Input Current
l
l
l
VSHDN = 5.5V = VIN
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3536 is tested under pulsed load conditions such that
TJ ≈ TA. The LTC3536E is guaranteed to meet specifications
from 0°C to 125°C junction temperature. Specifications over the
–40°C to 125°C operating junction temperature range are assured by
design, characterization and correlation with statistical process controls.
The LTC3536I is guaranteed over the full –40°C to 125°C operating
junction temperature range. Note that the maximum ambient temperature
consistent with these specifications is determined by specific operating
conditions in conjunction with board layout, the rated package thermal
impedance and other environmental factors. The junction temperature
(TJ, in °C) is calculated from the ambient temperature (TA, in °C) and
power dissipation (PD, in watts) according to the formula:
TJ = TA + (PD • θJA),
where θJA (in °C/W) is the package thermal impedance.
0.3
1
V
0.4
V
V
2
MHz
1
µA
1
V
1
µA
Note 3: This IC includes overtemperature protection that is intended to
protect the device during momentary overload conditions. The maximum
rated junction temperature will be exceeded when this protection is active.
Continuous operation above the specified absolute maximum operating
junction temperature may impair device reliability or permanently damage
the device.
Note 4: Current measurements are performed when the LTC3536 is
not switching. The current limit values measured in operation will be
somewhat higher due to the propagation delay of the comparators.
Note 5: Guaranteed by design characterization and correlation with
statistical process controls.
Note 6: Guaranteed by correlation and design.
3536fa
3
LTC3536
Typical Performance Characteristics
Efficiency Li-Ion (3V, 3.7V, 4.2V)
to 3.3V Output
90
80
80
70
70
EFFICIENCY (%)
100
90
60
50
VIN = 1.8V
VIN = 2.5V
VIN = 5.5V
VIN = 1.8V BURST
VIN = 2.5V BURST
VIN = 5.5V BURST
30
20
10
0
0.001
0.01
0.1
LOAD CURRENT (A)
60
60
50
40
VIN = 3V
VIN = 3.7V
VIN = 4.2V
VIN = 3V BURST
VIN = 3.7V BURST
VIN = 4.2V BURST
30
20
10
0
0.001
1
0.01
0.1
LOAD CURRENT (A)
3536 G01
INPUT CURRENT LIMIT (A)
INPUT QUIESCENT CURRENT (mA)
10
8
6
4
2
0
1.5
2
2.5
3 3.5 4 4.5 5
INPUT VOLTAGE (V)
5.5
6
2.0
1.9
1.8
1.7
1.6
1.5
1.4
1.3
1.2
1.1
1.0
0.9
0.8
0.7
0.6
0.5
1.5
2
2.5
3 3.5 4 4.5 5
INPUT VOLTAGE (V)
240
220
5.5
6
180
160
140
3.1
3.0
2.9
2.8
2.7
2.6
2.5
2.4
2.3
2.2
2.1
2.0
1.9
1.8
1.7
1.6
1.5
180
160
140
80
60
4
2.5
3 3.5 4 4.5 5
INPUT VOLTAGE (V)
5.5
6
–45°C
0°C
25°C
85°C
125°C
140
60
2
5.5
180
100
3536 G07
3 3.5 4 4.5 5
INPUT VOLTAGE (V)
160
80
1.5
6
200
80
6
2.5
220
200
120
5.5
2
240
100
3 3.5 4 4.5 5
INPUT VOLTAGE (V)
5.5
125°C
85°C
25°C
–40°C
260
100
2.5
3 3.5 4 4.5 5
INPUT VOLTAGE (V)
RDS(ON) for N-Channel Switch C
120
2
2.5
280
120
1.5
2
3536 G06
RDS(ON) (mΩ)
200
VOUT = 1.8V
VOUT = 3.3V
VOUT = 5.5V
25
3536 G03
–45°C
0°C
25°C
85°C
125°C
260
RDS(ON) (mΩ)
RDS(ON) (mΩ)
220
30
RDS(ON) for N-Channel Switch B
280
–45°C
0°C
25°C
85°C
125°C
240
35
3536 G05
RDS(ON) for P-Channel Switch A
260
40
Input Current Limit vs Supply
Voltage, VOUT = 3.3V
125°C
85°C
25°C
–40°C
3536 G04
280
45
Input Current Limit vs Supply
Voltage, VOUT = GND
VOUT = 3.3V
VOUT = 1.8V
VOUT = 5.5V
12
50
3536 G02
No-Load Quiescent Current in
PWM Mode Operation
14
55
20
1.5
1
INPUT CURRENT LIMIT (A)
EFFICIENCY (%)
100
No-Load Quiescent Current in Burst
Mode Operation (MODE = VIN)
INPUT QUIESCENT CURRENT (µA)
Efficiency 3.3V vs Load Current
40
TA = 25°C, VIN = VOUT = 3.3 V unless otherwise noted.
6
3536 G08
60
1.5
2
2.5
3 3.5 4 4.5 5
INPUT VOLTAGE (V)
5.5
6
3536 G09
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LTC3536
Typical Performance Characteristics
RDS(ON) for P-Channel Switch D
240
220
200
CHANGE IN VOLTAGE FROM 25°C (%)
–45°C
0°C
25°C
85°C
125°C
260
180
160
140
120
100
80
60
1.5
2
2.5
3 3.5 4 4.5 5
INPUT VOLTAGE (V)
5.5
Oscillator Frequency vs RT
2.0
0.8
1.8
0.6
0.4
0.2
0
–0.2
–0.4
–0.6
3536 G10
MAXIMUM LOAD CURRENT (mA)
MAXIMUM LOAD CURRENT (mA)
500
0
VOUT = 1.8V
VOUT = 3.3V
VOUT = 5.5V
1.5
2
2.5
3 3.5 4 4.5 5
INPUT VOLTAGE (V)
5.5
6
0.6
0.4
40
90
140
190
240
RT (kΩ)
250
200
150
100
0
VOUT = 1.8V
VOUT = 3.3V
VOUT = 5.5V
1.5
2
2.5
3 3.5 4 4.5 5
INPUT VOLTAGE (V)
3536 G13
5.5
6
VOUT PULLED UP TO 3.6V
VIN = 1.8V
L = 4.7µH
–0.5
–1.0
–1.5
–2.0
0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8 2 2.2 2.4
OSCILLATOR FREQUENCY (MHz)
3536 G15
3536 G14
Change in Output Voltage vs Load
Current for 3.3V Output and 3.3V
Input
340
Negative Inductor Current
vs Oscillator Frequency
0
50
290
3536 G12
300
1000
1.0
0.8
Maximum Load Current in Burst
Mode Operation vs Input Voltage
2500
1500
1.2
3536 G11
Maximum Load Current in PWM Mode
vs Input Voltage 1MHz Switching
Frequency, 4.7µH Inductor Value
2000
1.4
0
–1.0
– 50 – 30 – 10 10 30 50 70 90 110 130 150
TEMPERATURE (°C)
6
1.6
0.2
–0.8
REVERSE CURRENT LIMIT (A)
RDS(ON) (mΩ)
Feedback Voltage
1.0
OSCILLATOR FREQUENCY (MHz)
280
TA = 25°C, VIN = VOUT = 3.3 V unless otherwise noted.
Load Step 0A to 1A
Load Step 0mA to 300mA
OUTPUT VOLTAGE REGULATION (%)
1
VOUT
200mV/DIV
VOUT
200mV/DIV
ILOAD
500mA/DIV
ILOAD
100mA/DIV
0
–1
VIN = 3.2V
VOUT = 3.3V
100µs/DIV
3536 G17
VIN = 1.8V
VOUT = 3.3V
100µs/DIV
3536 G18
–2
–3
0.001
PWM
BURST
0.01
0.1
1
LOAD CURRENT (mA)
10
3536 G16
3536fa
5
LTC3536
Typical Performance Characteristics
Output Voltage Ripple in
PWM Mode
Output Voltage Ripple in Burst
Mode Operation
VOUT
20mV/DIV
VIN = 2.5V
VOUT
20mV/DIV
VIN = 3.3V
VOUT
20mV/DIV
VIN = 5V
TA = 25°C, VIN = VOUT = 3.3 V unless otherwise noted.
Burst Mode Operation to PWM
Mode Transient
VOUT
50mV/DIV
VIN = 5.5V
VOUT
50mV/DIV
INDUCTOR
CURRENT
200mA/DIV
VOUT = 3.3V
ILOAD = 0.5A
2µs/DIV
VOUT = 3.3V
ILOAD = 4mA
3536 G19
50µs/DIV
3536 G20
VOUT
50mV/DIV
VIN = 3.3V
VOUT
50mV/DIV
VIN = 1.8V
Start-Up in Burst Mode Operation
with Output Precharged
Soft-Start
VOUT
100mV/DIV
VOUT
500mV/DIV
SHDN
2V/DIV
ILOAD
500mA/DIV
SHDN
1V/DIV
SHDN
1V/DIV
6
200µs/DIV
3536 G22
VIN = 3V
VOUT = 3.3V
COUT = 22µF
ILOAD = 1mA
5ms/DIV
3536 G23
100µs/DIV
3536 G21
Start-Up in PWM Mode Operation
with Output Precharged
VOUT
1V/DIV
VIN = 3V
VOUT = 3.3V
COUT = 22µF
VOUT = 3.3V
ILOAD = 25mA
COUT = 22µF
L = 4.7µH
VIN = 3V
VOUT = 3.3V
COUT = 22µF
ILOAD = 20mA
500µs/DIV
3536 G24
3536fa
LTC3536
Pin Functions
(DFN/MSOP)
RT (Pin 1/Pin 1): Oscillator Frequency Programming
Input. Connect a resistor from RT to GND to program the
internal oscillator frequency. The frequency is given by:
fOSC (MHz) = 100/RT (kΩ)
where RT is in kΩ and fOSC is between 0.3MHz and 2MHz.
Tying the RT pin to VIN enables the internal 1.2MHz default
oscillator frequency.
SGND (Pin 2/Pin 2): Ground Connection for the LTC3536.
A ground plane is highly recommended. Sensitive analog
components terminated at ground should connect to the
GND pin with a Kelvin connection, separated from the
high current path.
MODE/SYNC (Pin 3/Pin 3): Pulse Width Modulation/Burst
Mode Selection and Synchronization Input. Driving MODE
to a logic 0 state programs fixed frequency, low noise
PWM operation. Driving MODE to logic 1 state programs
Burst Mode operation for highest efficiency at light loads.
In Burst Mode operation, the output current capability is
significantly less than what is available in PWM operation.
Refer to the Applications Information section of this data
sheet for details. Frequency synchronization is achieved
if a clock pulse is applied to MODE/SYNC. The external
clock pulse amplitude must have an amplitude equal or
higher than 1.2V and duty cycle from 10% and 90%. The
free-running frequency of the LTC3536 oscillator can be
programmed slower or faster than the synchronization
clock frequency.
SW1 (Pin 4/Pin 4): Switch Pin. Connect to internal power
switches A and B. Connect one side of the buck-boost
inductor to SW1. Provide a short wide PCB trace from the
inductor to SW1 to minimize voltage transients and noise.
SW2 (Pin 5/Pin 6): Switch Pin. Connect to internal power
switches C and D. Connect one side of the buck-boost
inductor to SW2. Provide a short wide PCB trace from the
inductor to SW2 to minimize voltage transients and noise.
VOUT (Pin 6/Pin 7): Output Voltage. This pin is the power
output for the regulator. A low ESR capacitor should be
placed between this pin and the ground plane. The capacitor should be placed as close to this pin as possible and
have a short return path to ground.
VIN (Pin 7/Pins 8, 9): Power Input for the Converter. A low
ESR 10µF or larger bypass capacitor should be connected
between this pin and ground. The capacitor should be
placed as close to this pin as possible and have a short
return path to ground.
SHDN (Pin 8/Pin 10): Enable Input. A logic 1 on SHDN
activates the buck-boost regulator. A logic 0 on SHDN
deactivates the buck-boost regulator.
FB (Pin 9/Pin 11): Output Voltage Programming Feedback
Divider Input. The regulator output voltage is programmed
by the voltage divider connected to FB. The buck-boost
output is given by the following equation:
VOUT = 0.6V • (1 + RTOP/RBOT) (V)
where RBOT is a resistor connected between FB and ground
and RTOP is a resistor connected between FB and VOUT.
The buck-boost output voltage can be adjusted from 1.8V
to 5.5V.
VC (Pin 10/Pin 12): Error Amplifier Output. Frequency
compensation components are connected between VC
and FB to provide stable operation of the converter. Refer
to the Applications Information section of this data sheet
for design details.
PGND (Exposed Pad Pin 11/Pin 5, Exposed Pad Pin 13):
Power Ground. The exposed pad must be soldered to the
PCB and electrically connected to ground through the
shortest and lowest impedance connection possible.
3536fa
7
LTC3536
Block Diagram
L
SW1
VIN
1.8V TO 5.5V
SW2
SWA
VOUT
1.8V TO 5.5V
SWD
–0.4A
SWB
–
CIN
GATE
DRIVERS
AND
ANTICROSS
CONDUCTION
+
+
REVERSE
CURRENT
LIMIT
SWC
PGND
+
3.4A
–
+
1.75V
+
PEAK
CURRENT
LIMIT
UVLO
ERROR
AMP
RTOP
COUT
–
2.5A
PWM
AND
OUTPUT
PHASING
CURRENT
LIMIT
+
+
–
SOFT-START
0.6V
FB
VC
–
CFB
RT
OSC
SYNC
RT
SLEEP
1 = BURST
0 = PWM
RBOT
Burst Mode
CONTROL
RUN LOGIC
MODE/SYNC
SHDN
1 = ON
0 = OFF
SGND
3635 BD
8
3536fa
LTC3536
Operation
Introduction
The LTC3536 is a monolithic buck-boost converter that
can operate with input and output voltages from as low
as 1.8V to as high as 5.5V. A proprietary switch control
algorithm allows the buck-boost converter to maintain
output voltage regulation with input voltages that are
above, below or equal to the output voltage. Transitions
between these operating modes are seamless and free of
transients and subharmonic switching.
The LTC3536 can be configured to operate over a wide
range of switching frequencies, from 300kHz to 2MHz,
allowing applications to be optimized for board area and
efficiency. The LTC3536 has an internal fixed-frequency
oscillator with a switching frequency that is easily set by
a single external resistor. In noise sensitive applications,
the converter can also be synchronized to an external clock
via the MODE/SYNC pin. The operating frequency defaults
to 1.2MHz when RT is connected to VIN eliminating the
external resistor.
The LTC3536 has been optimized to reduce input current
in shutdown and standby for applications that are sensitive to quiescent current draw, such as battery-powered
devices. In Burst Mode operation, the no-load standby
current is only 32µA and in shutdown the total supply
current is reduced to less than 1µA.
PWM Mode Operation
With the MODE/SYNC pin forced low or driven by an external clock, the LTC3536 operates in a fixed-frequency
pulse-width modulation (PWM) mode using a voltage mode
control loop. This mode of operation maximizes the output
current that can be delivered by the converter, reduces output voltage ripple, and yields a low noise fixed-frequency
switching spectrum. A proprietary switching algorithm
provides seamless transitions between operating modes
and eliminates discontinuities in the average inductor current, inductor current ripple, and loop transfer function
throughout all regions of operation. These advantages
result in increased efficiency, improved loop stability, and
lower output voltage ripple in comparison to the traditional
4-switch buck-boost converter.
Figure 1 shows the topology of the LTC3536 power stage
which is comprised of two P-channel MOSFET switches
and two N-channel MOSFET switches and their associated
gate drivers. In response to the error amplifier output, an
internal pulse-width modulator generates the appropriate
switch duty cycles to maintain regulation of the output
voltage.
VIN
VOUT
PMOS A
SW1
PMOS D
L
SW2
NMOS B
NMOS C
3536 F01
Figure 1. Power Stage Schematic
When the input voltage is significantly greater than the
output voltage, the buck-boost converter operates in
buck mode. Switch D turns on continuously and switch
C remains off. Switch A and B are pulse-width modulated
to produce the required duty cycle to support the output
regulation voltage. As the input voltage decreases, switch A
remains on for a larger portion of the switching cycle.
When the duty cycle reaches approximately 90% the
switch pair AC begins turning on for a small fraction of the
switching period. As the input voltage decreases further,
the AC switch pair remains on for longer durations and
the duration of the BD phase decreases proportionally. At
this point, switch A remains on continuously while switch
pair CD is pulse-width modulated to obtain the desired
output voltage. At this point, the converter is operating
solely in boost mode.
Oscillator and Phase-Locked Loop
The LTC3536 operates from an internal oscillator with a
switching frequency that can be configured by a single
external resistor between RT and ground. Tying RT to VIN
sets the default internal operating frequency to typically
1.2MHz. If the RT pin is driven externally to a level higher
than VIN, a current limiting resistor should be used. 1M for
6V on the RT pin limits the current to 6µA. Also, a Schottky
3536fa
9
LTC3536
Operation
Whether operating from its internal oscillator or when
synchronized to an external clock signal, the LTC3536 is
able to operate with a switching frequency from 300kHz
to 2MHz, providing the ability to minimize the size of the
external components and optimize the power conversion
efficiency.
Error Amplifier
The LTC3536 has an internal high gain operational amplifier which provides frequency compensation of the control
loop that maintains output voltage regulation. To ensure
stability of this control loop, an external compensation
network must be installed in the application circuit. A
Type III compensation network as shown in Figure 2 is
recommended for most applications since it provides the
flexibility to optimize the converter’s transient response
while simultaneously minimizing any DC error in the output
voltage. Details on designing the compensation network
in LTC3536 applications can be found in the Applications
Information section of this data sheet.
VOUT
LTC3536
RFF
RTOP
RBOT
0.6V
FB
CFF
CFB
RFB
+
–
PWM
VC
CPOLE
GND
3536 F02
Figure 2. Error Amplifier and Compensation Network
Input and Peak Current Limits
The LTC3536 has two current limit circuits that are designed to limit the peak inductor current to ensure that
the switch currents remain within the capabilities of the
IC during output short-circuit or overload conditions. The
10
input current limit operates by injecting a current into the
feedback pin, which is proportional to the extent that the
inductor current exceeds the input current limit threshold
(typically 2.5A). Due to the high gain of the feedback loop,
this injected current forces the error amplifier output to
decrease until the average current through the inductor
is approximately reduced to the current limit threshold.
For this current limit feature to be most effective, the
Thevenin resistance (RBOT//RTOP) from FB to ground
should exceed 100kΩ.
CURRENT FB PIN (µA)
diode from the RT pin to VIN can be used in addition to
current limiting resistor. For noise sensitive applications,
an internal phase-locked loop allows the LTC3536 to
be synchronized to an external clock signal applied to
the MODE/SYNC pin. The free-running frequency of the
oscillator can be programmed slower or faster than the
synchronization clock frequency.
15
14
13
12
11
10
9
8
7
6
5
4
3
2
1
0
0
0.5
1 1.5 2 2.5 3 3.5 4
CURRENT OUT OF SW1 PIN (A)
4.5
3536 F03
Figure 3. FB Current for Input Current Limitation
Since this input current limit circuit maintains the error
amplifier in an active state it ensures a smooth recovery
and minimal overshoot once the current limit fault condition is removed. On a hard output short, it is possible for
the inductor current to increase substantially beyond the
current limit threshold before the input current limit has
time to react and reduce the inductor current. For this
reason, there is a second current limit circuit (peak current limit), which turns off power switch A if the current
through switch A exceeds the approximately 3.4A limit
threshold. This provides additional protection in the case of
an instantaneous hard output short and provides time for
the primary current limit to react. When the input voltage
is lower than 2.4V, the input and peak current limit thresholds are gradually decreased. For minimum input voltage
(1.8V) they are typically 1.7A and 2.3A respectively. See
the Typical Performance Characteristics and the Inductor
Selection section for information about the inductor value
for maximum output current capability.
3536fa
LTC3536
Operation
Reverse Current Limit
Burst Mode OPERATION
In PWM mode operation the LTC3536 has the ability to
actively conduct current away from the output if that is
necessary to maintain regulation. If the output is held above
regulation, this could result in large reverse currents. This
situation can occur if the output of the LTC3536 is held
up momentarily by another supply as may occur during a
power-up or power-down sequence. To prevent damage to
the part under such conditions, the LTC3536 has a reverse
current comparator that monitors the current entering
power switch D from the load. If this current exceeds 0.55A
(typical) switch D is turned off for the remainder of the
switching cycle in order to prevent the reverse inductor
current from reaching unsafe levels.
When MODE/SYNC is held high, the buck-boost converter
operates in Burst Mode operation using a variable frequency
switching algorithm that minimizes the no-load input
quiescent current and improves efficiency at light load by
reducing the amount of switching to the minimum level
required to support the load. The output current capability in Burst Mode operation is substantially lower than in
PWM mode and is intended to support light stand-by loads.
Curves showing the maximum Burst Mode load current
as a function of the input and output voltage can be found
in the Typical Performance Characteristics section of this
data sheet. If the converter load in Burst Mode operation
exceeds the maximum Burst Mode current capability, the
output will lose regulation.
For no-load current application, the inductor current ripple
must be lower than double the minimum reverse current
limit (0.3A • 2 = 0.6A maximum inductor current ripple).
See the Inductor Selection section for information about
how to calculate the inductor current ripple.
Output Current Capability
The maximum output current that can be delivered by
the LTC3536 is dependent upon many factors, the most
significant being the input and output voltages. For VOUT
= 3.3V and VIN ≥ 3V, the LTC3536 is able to support a
1A load continuously. For VOUT = 3.3V and VIN =1.8V, the
LTC3536 is able to support a 300mA load continuously.
Typically, the output current capability is greatest when
the input voltage is approximately equal to the output
voltage. At larger step-up voltage ratios, the output current capability is reduced because the lower duty cycle of
switch D results in a larger inductor current being needed
to support a given load. Additionally, the output current
capability generally decreases at large step-down voltage
ratios due to higher inductor current ripple which reduces
the maximum attainable inductor current.
The output current capability can also be affected by inductor characteristics. An inductor with large DC resistance
will degrade output current capability, particularly in boost
mode operation. In addition, larger value inductors generally maximize output current capability by reducing inductor
current ripple. See the Typical Performance Characteristics
and the Inductor Selection section for information.
Each Burst Mode cycle is initiated when switches A and
C turn on producing a linearly increasing current through
the inductor. When the inductor current reaches the Burst
Mode peak current limit (0.6A typically), switches B and D
are turned on, discharging the energy stored in the inductor into the output capacitor and load. Once the inductor
current reaches zero, all switches are turned off and the
cycle is complete. Current pulses generated in this manner
are repeated as often as necessary to maintain regulation
of the output voltage. In Burst Mode operation, the error
amplifier is used as burst comparator. If the MODE pin
is driven externally to a level higher than VIN, a current
limiting resistor should be used. 1M for 6V on the MODE
pin limits the current to 6µA. Also, a Schottky diode from
the MODE pin to VIN can be used in addition to current
limiting resistor.
Soft-Start
To minimize input current transients on power-up, the
LTC3536 incorporates an internal soft-start circuit with a
nominal duration of 0.9ms. The soft-start is implemented
by a linearly increasing ramp of the error amplifier reference voltage during the soft-start duration. As a result,
the duration of the soft-start period is largely unaffected
by the size of the output capacitor or the output regulation voltage. Given the closed-loop nature of the soft-start
implementation, the converter is able to respond to load
transients that occur during the soft-start interval. The
3536fa
11
LTC3536
Operation
soft-start period is reset by thermal shutdown and UVLO
events on VIN and the mode of operation is always PWM.
In case the output voltage at start -up is already precharged
above 90% (typically) of the target value, the internal softstart is skipped and the LTC3536 immediately enters the
mode of operation that has been set on the MODE pin.
If the MODE pin is tied high and Burst Mode operation is
selected, the output voltage is regulated smoothly to the
target voltage value. Instead if the MODE pin is tied low and
PWM mode is selected, the error amplifier needs to charge
up the VC pin and the output voltage might be pulled to
lower voltage values for a short period of time, proportional
to the value of the main compensation capacitor.
Undervoltage Lockout
To ensure proper operation, the LTC3536 incorporates
internal undervoltage lockout (UVLO) circuitry. The converter is disabled if VIN falls below its respective UVLO
threshold (typical 1.67V). If the input voltage falls below
this level all switching is disabled until the input voltage
rises above 1.75V (nominal).
Output Disconnect
The LTC3536 is designed to allow true output disconnect
by opening both P-channel MOSFET rectifiers. This allows
VOUT to go to zero volts during shutdown, drawing no
current from the input source.
Thermal Considerations
The power switches in the LTC3536 are designed to operate
continuously with currents up to the internal current limit
thresholds. However, when operating at high current levels
there may be significant heat generated within the IC. As a
result, careful consideration must be given to the thermal
environment of the IC in order to optimize efficiency and
ensure that the LTC3536 is able to provide its full-rated
output current. Specifically, the exposed pad of both the
DD and MSOP packages shall be soldered to the PC board
and the PC board should be designed to maximize the
conduction of heat out of the IC package.
If the die temperature exceeds approximately 165°C, the
IC will enter overtemperature shutdown and all switching
will be inhibited. The part will remain disabled until the
die cools by approximately 10°C. The soft-start circuit
is reinitialized in overtemperature shutdown to provide
a smooth recovery when the fault condition is removed.
If the SHDN pin is driven externally to a level higher than
VIN, a current limiting resistor should be used. 1M for 6V
on the SHDN pin limits the current to 6µA. Also, a Schottky
diode from the SHDN pin to VIN can be used in addition
to current limiting resistor.
Applications Information
The standard LTC3536 application circuit is shown as the
Typical Application on the front page of this data sheet. The
appropriate selection of external components is dependent
upon the required performance of the IC in each particular
application given considerations and trade-offs such as
PCB area, cost, output and input voltage, allowable ripple
voltage, efficiency and thermal considerations. This section
of the data sheet provides some basic guidelines and considerations to aid in the selection of external components
and the design of the application circuit.
12
Inductor Selection
The choice of inductor used in LTC3536 application circuits
influences the maximum deliverable output current, the
magnitude of the inductor current ripple, and the power
conversion efficiency. The inductor must have low DC
series resistance or output current capability and efficiency
will be compromised. Larger inductance values reduce
inductor current ripple and will therefore generally yield
greater output current capability. For a fixed DC resistance,
a larger value of inductance will yield higher efficiency by
3536fa
LTC3536
Applications Information
reducing the peak current to be closer to the average output current and therefore minimize resistive losses due to
high RMS currents. However, a larger inductor within any
given inductor family will generally have a greater series
resistance, thereby counteracting this efficiency advantage.
An inductor used in LTC3536 applications should have a
saturation current rating that is greater than the worst-case
average inductor current plus half the ripple current. The
peak-to-peak inductor current ripple for each operational
mode can be calculated from the following formula, where
f is the switching frequency in MHz, L is the inductance
in µH.
 V –V 
V
∆IL(P-P)(BUCK) = OUT  IN OUT 
f •L 
V

IN
∆IL(P-P)(BOOST) =
accordingly in order to have the same current ripple
(2.2µH for 2MHz, 15µH for 300kHz).
Different inductor core materials and styles have an impact
on the size and price of an inductor at any given current
rating. Shielded construction is generally preferred as it
minimizes the chances of interference with other circuitry.
The choice of inductor style depends upon the price, sizing,
and EMI requirements of a particular application. Table 1
provides a small sampling of inductors that are well suited
to many LTC3536 applications.
Table 1. Recommended Inductors
VENDOR
PART/STYLE
Coilcraft
847-639-6400
www.coilcraft.com
LPO2506
LPS4012, LPS4018
MSS6122
MSS4020
MOS6020
DS1605, DO1608
XPL4020
XAL4040
XFL4020
Coiltronics
www.cooperet.com
SD52, SD53
SD3114, SD311B
Murata
714-852-2001
www.sumida.com
LQH55D
Sumida
847-956-0666
www.sumida.com
CDH40D11
Taiyo Yuden
www.t-yuden.com
NP04S8
NR3015
NR4018
TDK
847-803-6100
www.component.tdk.com
VLP, LTF
VLF, VLCF
Würth Elektronik
201-785-8800
www.we-online.com
WE-TPC Type S, M, MH
VIN  VOUT – VIN 
f •L  VOUT 
In addition to its influence on power conversion efficiency,
the inductor DC resistance can also impact the maximum
output capability of the buck-boost converter particularly
at low input voltages. In buck mode, the output current of
the buck-boost converter is limited only by the inductor
current reaching the current limit threshold. However, in
boost mode, especially at large step-up ratios, the output
current capability can also be limited by the total resistive
losses in the power stage. These include switch resistances, inductor resistance and PCB trace resistance. Use
of an inductor with high DC resistance can degrade the
output current capability from that shown in the Typical
Performance Characteristics section of this data sheet.
As a guideline, in most applications the inductor DC resistance should be significantly smaller than the typical
power switch resistance of 120mΩ.
The minimum inductor value must guarantee that the
worst-case average input current plus half the ripple
current don’t reach the input current limit threshold.
For a switching frequency of 1MHz the recommended
typical inductor value is 4.7µH. For a higher and lower
switching frequency the inductor value should be changed
Output Capacitor Selection
A low ESR output capacitor should be utilized at the buckboost converter output in order to minimize output voltage
ripple. Multilayer ceramic capacitors are an excellent option
as they have low ESR and are available in small footprints.
The capacitor value should be chosen large enough to
reduce the output voltage ripple to acceptable levels.
3536fa
13
LTC3536
Applications Information
Neglecting the capacitor ESR and ESL, the peak-to-peak
output voltage ripple can be calculated by the following
formulas, where f is the frequency in MHz, COUT is the
capacitance in µF and ILOAD is the output current in amps.
∆V(P-P)(BUCK) =
2
VOUT
8 • f •L • COUT
∆V(P-P)(BOOST) =
 VIN – VOUT 


V
IN
ILOAD  VOUT – VIN 
f • COUT  VOUT 
Given that the output current is discontinuous in boost
mode, the ripple in this mode will generally be much larger
than the magnitude of the ripple in buck mode.
In addition to output voltage ripple generated across the
output capacitance, there is also output voltage ripple
produced across the internal resistance of the output
capacitor. The ESR-generated output voltage ripple is
proportional the series resistance of the output capacitor.
Input Capacitor Selection
The PVIN pin carries the full inductor current and provides
power to internal control circuits in the IC. To minimize
input voltage ripple and ensure proper operation of the IC,
a low ESR bypass capacitor with a value of at least 10µF
should be located as close to this pin as possible. The
traces connecting this capacitor to PVIN and the ground
plane should be made as short as possible. The SVIN pin
provides power to the internal circuitry. In every application, the SVIN and PVIN must be connected together on
the PC Board.
Recommended Input and Output Capacitors
The capacitors used to filter the input and output of the
LTC3536 must have low ESR and must be rated to handle
the large AC currents generated by switching converters.
This is important to maintain proper functioning of the IC
and to reduce output voltage ripple.
The choice of capacitor technology is primarily dictated
by a trade-off between cost, size and leakage current.
Ceramic capacitors are often utilized in switching converter applications due to their small size, low ESR and
14
low leakage currents. However, many ceramic capacitors
designed for power applications experience significant
loss in capacitance from their rated value with increased
DC bias voltages. For example, it is not uncommon for
a small surface mount ceramic capacitor to lose more
than 50% of its rated capacitance when operated near its
rated voltage. As a result, it is sometimes necessary to
use a larger value capacitance or a capacitor with a higher
voltage rating than required in order to actually realize the
intended capacitance at the full operating voltage. To ensure
that the intended capacitance is realized in the application
circuit, be sure to consult the capacitor vendor’s curve of
capacitance versus DC bias voltage.
The capacitors listed in Table 2 provide a sampling of small
surface mount ceramic capacitors that are well suited to
LTC3536 application circuits. All listed capacitors are either
X5R or X7R dielectric in order to ensure that capacitance
loss over temperature is minimized.
Table 2. Representative Bypass and Output Capacitors
VALUE
(µF)
VOLTAGE
(V)
SIZE (mm)
L × W × H (FOOTPRINT)
AVX
12066D106K
12066D226K
12066D476K
10
22
47
6.3
6.3
6.3
3.2 × 1.6 × 0.5 (1206)
3.2 × 1.6 × 0.5 (1206)
3.2 × 1.6 × 0.5 (1206)
Kemet
C0603C106K9P
C0805C226K9P
C0805C476K9P
10
22
47
6.3
6.3
6.3
1.6 × 0.8 × 0.8 (0603)
2.0 × 1.25 × 1.25 (0805)
2.0 × 1.25 × 1.25 (0805)
Murata
GRM21
GRM21
10
22
10
6.3
2.0 × 1.25 × 1.25 (0805)
2.0 × 1.25 × 1.25 (0805)
TDK
C2102X5R0J
C2102X5R0J
22
47
6.3
6.3
2.0 × 1.25 × 0.85 (0805)
2.0 × 1.25 × 1.25 (0805)
Taiyo Yuden
JMK212BJ
JMK212BJ
22
47
6.3
6.3
2.0 × 1.25 × 0.85 (0805)
2.0 × 1.25 × 0.85 (0805)
PART NUMBER
Small-Signal Model
The LTC3536 uses a voltage mode control loop to maintain
regulation of the output voltage. An externally compensated error amplifier drives the VC pin to generate the
appropriate duty cycle of the power switches. Use of an
external compensation network provides the flexibility for
optimization of closed-loop performance over the wide
3536fa
LTC3536
Applications Information
VIN
The denominator of the buck mode transfer function
exhibits a pair of resonant poles generated by the LCOUT
filtering of the power stage. The resonant frequency of
the power stage, fO, is given by the following expression
where L is the value of the inductor in henries.
VOUT
COUT
A
D
L
VC
PWM
RC
RS
B
RLOAD
C
3536 F04
Figure 4. Small-Signal Model
variety of output voltages, switching frequencies, and
external component values supported by the LTC3536.
VIN is the input supply voltage, VOUT the programmed
output voltage, L is the external buck-boost inductor, COUT
the output capacitor, RS the series resistance in the power
path (it can be approximated as twice the average power
switch resistance plus the DC resistance of the inductor)
and RC is the output capacitor ESR.
Buck Mode
The small-signal transfer function of the buck-boost
converter is different in the buck and boost modes of operation and care must be taken to ensure stability in both
operating regions. When stepping down from a higher
input voltage to a lower output voltage, the converter
will operate in buck mode and the small-signal transfer
function from the error amplifier output, VC, to the converter output voltage is given by the following equation:
VOUT
= 2.64 • VIN •
(s)
VC
Buck Mode
1+ sRCCOUT
 s 
1+
+
ω OQ  ω O 
s
2
This transfer function has a single zero created by the
output capacitor ESR and a resonant pair of poles. In most
applications, an output capacitor with a very low ESR is
utilized in order to reduce the output voltage ripple to acceptable levels. Such low values of capacitor ESR result
in a very high frequency zero and as a result the zero is
commonly too high in frequency to significantly impact
compensation of the feedback loop.
ωO =
1
1
, fO =
LCOUT
2π LCOUT
The quality factor, Q, has a significant impact on compensation of the voltage loop since a higher Q factor produces
a sharper loss of phase near the resonant frequency. The
quality factor is inversely related to the amount of damping
in the power stage and is substantially influenced by the
average series resistance of the power stage, RS. Lower
values of RS will increase the Q and result in a sharper
loss of phase near the resonant frequency and will require
more phase boost or lower bandwidth to maintain an
adequate phase margin.
Q=
LCOUT
COUT (RC +RS ) +
L
RLOAD
Boost Mode
When stepping up from a lower input voltage to a higher
output voltage, the buck-boost converter will operate in
boost mode where the small-signal transfer function from
control voltage, VC, to the output voltage is given by the
following expression:

VOUT
= 2.64 • G
(s)
VC
Boost Mode
(1+ sRCCOUT )  1– ωs
1+
 s 
+
ω OQ  ω O 
s
2
Z


In boost mode operation, the transfer function is characterized by a pair of resonant poles and a zero generated by
the ESR of the output capacitor as in buck mode. However,
in addition there is a right-half plane zero which generates
3536fa
15
LTC3536
Applications Information
increasing gain and decreasing phase at higher frequencies. As a result, the crossover frequency in boost mode
operation generally must be set lower than in buck mode
in order to maintain sufficient phase margin.
R
G = VIN • LOAD •
RS
2
RLOAD
V 
•  OUT 
 V 
R
1+ LOAD
RS
 V 
•  IN 
 VOUT 
2
1–
RS
IN
Buck-Boost Mode
When the converter operates in buck-boost mode and the
small-signal transfer function from control voltage, VC, to
the output voltage is given by the following expression:
VOUT
=
(s)
VC
Buck-Boost Mode

17.62 • G
2
1+
 V 
RS +RLOAD  IN 
 VOUT 
ωO =
LCOUT (RLOAD +RC )
In boost mode operation, the frequency of the right-half
plane zero, fZ, is given by the following expression. The
frequency of the right half plane zero decreases at higher
loads and with larger inductors.
2
2
 VIN 
 VIN 
R
–
R
LOAD
S
 V 
 V  RLOAD – RS
OUT
ωZ =
, fZ = OUT
L
2πL
Q=
 V 
RS +RLOAD  IN 
 VOUT 
2
2
 V 
L + COUTRLOADRC  IN  +RSCOUT (RLOAD +RC )
 VOUT 
G=
Z
(
0.15 • VOUT RLOAD • ε 2 • 1.85 – RS • (1.85 – ε )
(
ε • (1.85 – ε ) • RS +RLOAD • ε 2
)
)
where the variable ε is defined:
ε=
ωO =
VIN • 1.85
VOUT + VIN
RS +RLOAD • ε 2
LCOUT (RLOAD +RC )
In buck-boost mode operation, the frequency of the righthalf plane zero, fZ, is given by the following expression.
The frequency of the right-half plane zero decreases at
higher loads and with larger inductors.
16
 s 
+
ω OQ  ω O 
s
2


Also in buck-boost mode operation, the transfer function
is characterized by a pair of resonant poles and a zero
generated by the ESR of the output capacitor as in buck
mode and a right half plane zero.
Finally, the magnitude of the quality factor of the power
stage in boost mode operation is given by the following
expression:
LCOUT (RLOAD +RC )
(1+ sRCCOUT )  1– ωs
ωZ =
1.85 • ε 2RLOAD – RS • (1.85 – ε )
L • (1.85 – ε )
3536fa
LTC3536
Applications Information
Finally, the magnitude of the quality factor of the power
stage in buck-boost mode operation is given by the following expression:
Q=
LCOUT (RLOAD +RC ) RS +RLOAD • ε 2
L + COUTRLOADRC • ε 2 +RSCOUT (RLOAD +RC )
Compensation of the Voltage Loop
The small-signal models of the LTC3536 reveal that the
transfer function from the error amplifier output, VC, to
the output voltage is characterized by a set of resonant
poles and a possible zero generated by the ESR of the
output capacitor as shown in the Bode plot of Figure 5.
In boost mode operation, there is an additional right-half
plane zero that produces phase lag and increasing gain at
higher frequencies. Typically, the compensation network
is designed to ensure that the loop crossover frequency
is low enough that the phase loss from the right-half
plane zero is minimized. The low frequency gain in buck
mode is a constant, but varies with both VIN and VOUT in
boost mode.
low enough that the resultant crossover frequency of the
control loop is well below the resonant frequency.
In most applications, the low bandwidth of the Type I compensated loop will not provide sufficient transient response
performance. To obtain a wider bandwidth feedback loop,
optimize the transient response, and minimize the size of
the output capacitor, a Type III compensation network as
shown in Figure 7 is required.
VOUT
0.6V
FB
RTOP
RBOT
C1
+
–
VC
GND
3536 F06
Figure 6. Error Amplifier with Type I Compensation
VOUT
LTC3536
RFF
RTOP
RBOT
GAIN
LTC3536
0.6V
FB
CFF
CFB
RFB
CPOLE
+
–
VC
GND
3536 F07
–40dB/DEC
–20dB/DEC
0°
PHASE
–90°
BUCK MODE
–180°
BOOST MODE
–270°
fO
fRHPZ
f
3536 F05
Figure 5. Buck-Boost Converter Bode Plot
For charging or other applications that do not require an
optimized output voltage transient response, a simple
Type I compensation network as shown in Figure 6 can
be used to stabilize the voltage loop. To ensure sufficient
phase margin, the gain of the error amplifier must be
Figure 7. Error Amplifier with Type III Compensation
A Bode plot of the typical Type III compensation network
is shown in Figure 8. The Type III compensation network
provides a pole near the origin which produces a very high
loop gain at DC to minimize any steady-state error in the
regulation voltage. Two zeros located at fZERO1 and fZERO2
provide sufficient phase boost to allow the loop crossover
frequency to be set above the resonant frequency, fO, of
the power stage. The Type III compensation network also
introduces a second and third pole. The second pole, at
frequency fPOLE2, reduces the error amplifier gain to a
zero slope to prevent the loop crossover from extending
too high in frequency. The third pole at frequency fPOLE3
provides attenuation of high frequency switching noise.
3536fa
17
LTC3536
Applications Information
In most applications the compensation network is designed
so that the loop crossover frequency is above the resonant
frequency of the power stage, but sufficiently below the
boost mode right-half plane zero to minimize the additional
phase loss. Once the crossover frequency is decided upon,
the phase boost provided by the compensation network
is centered at that point in order to maximize the phase
margin. A larger separation in frequency between the
zeros and higher order poles will provide a higher peak
phase boost but may also increase the gain of the error
amplifier which can push out the loop crossover to a
higher frequency.
GAIN
–20dB/DEC
–20dB/DEC
90°
0°
–90°
PHASE
fPOLE2 fPOLE3
fZERO1
f
3536 F08
fZERO2
Figure 8. Type III Compensation Bode Plot
The transfer function of the compensated Type III error
amplifier from the input of the resistor divider to the output
of the error amplifier, VC, is:
VC(S)
VOUT(S)
= GEA

s
 1+ 2πf
ZERO1

s
  1+ 2πf
ZERO2





s
s
1+
s  1+


 2πfPOLE1   2πfPOLE2 
The error amplifier gain is given by the following equation.
The simpler approximate value is sufficiently accurate in
most cases since CFB is typically much larger in value
than CPOLE.
GEA =
1
1
≈
R TOP (CFB + CPOLE ) R TOPCFB
The pole and zero frequencies of the Type III compensation
network can be calculated from the following equations
where all frequencies are in Hz, resistances are in ohms,
and capacitances are in farads.
fZERO1 =
1
1
1
≈
2π (R TOP +RFF ) CFF 2πR TOPCFF
fPOLE2 =
CFB + CPOLE
1
≈
2πCFBCPOLERFB 2πCPOLERFB
18
However, with higher losses in the power stage (larger RS)
the Q factor will be lower and the phase loss will occur
more gradually. As a result, the power stage phase will
not be as close to –180° at the crossover frequency and
less phase boost is required of the compensation network.
The LTC3536 error amplifier is designed to have a fixed
maximum bandwidth in order to provide rejection of
switching noise to prevent it from interfering with the
control loop. From a frequency domain perspective, this
can be viewed as an additional single pole as illustrated
in Figure 9. The nominal frequency of this pole is 400kHz.
For typical loop crossover frequencies below about 40kHz
the phase contributed by this additional pole is usually
2πRFBCFB
fZERO2 =
fPOLE3 =
The Q of the power stage can have a significant influence
on the design of the compensation network because it
determines how rapidly the 180° of phase loss in the power
stage occurs. For very low values of series resistance, RS,
the Q will be higher and the phase loss will occur sharply.
In such cases, the phase of the power stage will fall rapidly
to –180° above the resonant frequency and the total phase
margin must be provided by the compensation network.
0.6V
FB
VC
+
–
LTC3536
RFILT
CFILT
INTERNAL
VC
3536 F09
Figure 9. Internal Loop Filter
1
2πCFFRFF
3536fa
LTC3536
Applications Information
Loop Compensation Example
This section provides an example illustrating the design of
a compensation network for a typical LTC3536 application
circuit. In this example a 3.3V regulated output voltage is
generated with the ability to supply 300mA load from an
input power source ranging from 1.8V to 5.5V. To optimize
efficiency 1MHz switching frequency has been chosen. In
this application the maximum inductor current ripple will
occur at the highest input voltage. An inductor value of
4.7µH has been chosen to limit the worst-case inductor
current ripple. A low ESR output capacitor with a value
of 22µF is specified to yield a worst-case output voltage
ripple of approximately 10mV (occurring at the worst-case
step-up ratio and maximum load current). In summary, the
key power stage specifications for this LTC3536 example
application are given below:
f = 1MHz
VIN = 1.8V to 5.5V
VOUT = 3.3V at 300mA
COUT = 22µF,
RC = 10mΩ
L = 4.7µH,
RL = 60mΩ
The first step in designing the compensation network
is to determine the target crossover frequency for the
compensated loop. This example will be designed for a
60° phase margin to ensure adequate performance over
parametric variations and varying operating conditions. As
a result, the target crossover frequency, fC, will be the point
In this case, the phase reaches –180° at 37.8kHz making
fC = 37.8kHz the target crossover frequency for the compensated loop. From the Bode plot of Figure 9 the gain of
the power stage at the target crossover frequency is –2dB.
At this point in the design process, there are three constraints that have been established for the compensation
network. It must have +2dB gain at fC = 37.8kHz, a peak
phase boost of 60° and the phase boost must be centered
at fC = 37.8kHz.
An analytical approach can be used to design a compensation network with the desired phase boost, center frequency
and gain. In general, this procedure can be cumbersome
due to the large number of degrees of freedom in a Type III
compensation network. However the design process can
be simplified by assuming that both compensation zeros
VO/VC
30
0
24
–20
18
–40
–60
12
GAIN
6
–80
PHASE
0
–100
–6
–120
–12
–140
–18
–160
–24
–180
–30
–200
–36
–220
–42
1
10
100
1k
10k
100k
FREQUENCY (Hz)
1M
10M
PHASE (DEG)
With the power stage parameters specified, the compensation network can be designed. A reasonable approach
is to design the compensation network at this worst-case
corner and then verify that sufficient phase margin exists
across all other operating conditions. In this example application, at VIN = 1.8V and the full 300mA load current,
the right-half plane zero will be located at 100kHz and this
will be a dominant factor in determining the bandwidth of
the control loop.
at which the phase of the buck-boost converter reaches
–180°. It is generally difficult to determine this frequency
analytically, because it is significantly impacted by the Q
factor of the resonance in the power stage. As a result,
it is best determined from a Bode plot of the buck-boost
converter as shown in Figure 10. This Bode plot is for
the LTC3536 buck-boost converter using the previously
specified power stage parameters and was generated from
the small signal model equations using LTspice® software.
GAIN (dB)
negligible (for 40kHz is around –5.7°). However, for loops
with higher crossover frequencies this additional phase
lag should be taken into account when designing the
compensation network.
–240
100M
3536 F10
Figure 10. Converter Bode Plot, VIN = 1.8V, ILOAD = 300mA
3536fa
19
LTC3536
Applications Information
occur at the same frequency, fZ, and both higher order poles
(fPOLE2 and fPOLE3) occur at the common frequency, fP .
This is a good starting point for determining the compensation network. However the Bode plot for the complete
loop should be checked overall operating conditions and
for variations in components values to ensure that sufficient phase margin and gain margin exists in all cases.
A reasonable choice is to pick the frequency of the poles,
fP, to be about 50 times higher than the frequency of the
zeros, fZ, which provides a peak phase boost of approximately ΦMAX = 60° as was assumed previously. Next, the
phase boost must be centered so that the peak phase
occurs at the target crossover frequency. The frequency
of the maximum phase boost, fC, is the geometric mean
of the pole and zero:
2
fC = fP • fZ = 50 • fZ = 7 • fZ
Therefore, in order to center the phase boost given a factor
of 50 separation between the pole and zero frequencies,
the zeros should be located at one-seventh of the crossover frequency and the poles should be located at seven
times the crossover frequency as given by the following
equations:
1
1
fZ = • fC = • ( 37.8kHz ) = 5.4kHz
7
7
fP = 7 • fC = 7 • ( 37.8kHz ) = 264.6kHz
Assuming a multiple of 50 separation between the pole
frequencies and zero frequencies this can be simplified
to the following expression:


50
GCENTER = 20log 

 ( 2πfC ) (R TOPCFB ) 
The first step in defining the compensation component
values is to pick a value for RTOP that provides an acceptably low quiescent current through the resistor divider.
A value of RTOP = 845k is a reasonable choice. Next, the
value of CFB can be found:
GCENTER = 2dB
CFB
50
2π • ( 37.8kHz )
2dB
• 845kΩ • 10 20
= 198pF ≈ 180pF
The compensation poles can be set at 264.6kHz and the
zeros at 5.4kHz by using the expressions for the pole and
zero frequencies given in the previous section. Setting the
frequency of the first zero, fZERO1, to 5.4kHz results in the
following value for RFB:
RFB =
1
= 163kΩ ≈ 162kΩ
2π • (180pF ) • 5.4kHz
This leaves the free parameter, CPOLE, to set the frequency
fPOLE1 to the common pole frequency of 264.6kHz as given:
1
= 3.71pF ≈ 3.9pF
2π • (162kΩ ) • 264.6kHz
This placement of the poles and zeros will yield a peak phase
boost of 60° that is centered at the crossover frequency,
fC. Next, in order to produce the desired target crossover
frequency, the gain of the compensation network at the
point of maximum phase boost, GCENTER, must be set to
+2dB. The gain of the compensated error amplifier at the
point of maximum phase gain is given by:
Next, CFF can be chosen to set the second zero, fZERO2, to
the common zero frequency of 5.4kHz.


2πfP


GCENTER = 10log
 ( 2πf )3 (R C )2 
Z
TOP FB 

Finally, the resistor value RFF can be chosen to place the
second pole at 264.6kHz:
20
CPOLE =
CFF =
RFF =
1
= 34.9pF ≈ 33pF
2π • ( 845kΩ ) • 5.4kHz
1
= 18.2kΩ
2π • ( 33pF ) • 264.6kHz
3536fa
LTC3536
Applications Information
A Bode plot of the error amplifier with the designed compensation component values is shown in Figure 11. The
Bode plot confirms that the peak phase occurs at 37.8kHz
and the phase boost at that point is about 60°. In addition,
the gain at the peak phase frequency is 2dB which is close
to the design target.
50
63
40
54
30
PHASE
20
10
27
0
18
–10
GAIN
9
–20
0
–30
–9
–40
–18
–50
–27
–60
–36
–70
–45
–80
–54
1
10
100
1k
10k
100k
FREQUENCY (Hz)
1M
10M
PHASE (DEG)
36
120
–60
100
–80
80
GAIN (dB)
72
45
GAIN (dB)
60
–40
–100
PHASE
60
–120
40
–140
–160
20
GAIN
0
–180
–20
–200
–40
–220
–60
–240
–80
–260
–100
1
10
100
1k
10k
100k
FREQUENCY (Hz)
1M
10M
PHASE (DEG)
VO/VC
81
VO/VC
140
–280
100M
3536 F12
Figure 12. Complete Loop Bode Plot for Boost Operation Mode
VO/VC
100
–90
100M
0
–30
80
3536 F11
–60
60
Figure 11. Compensated Error Amplifier Bode Plot
The Bode plot for the complete loop should be checked
overall operating conditions and for variations in component values to ensure that sufficient phase margin and
gain margin exists in all cases. The stability of the loop
should also be confirmed via time domain simulation and
by evaluating the transient response of the converter in
the actual circuit.
In this example the VIN varies from 1.8V to 5.5V. In buckboost operation (when 0.85 • VOUT < VIN < VOUT/0.85) the
Bode plot of the complete loop shows a phase margin of
–90
GAIN (dB)
40
GAIN
20
–120
0
–150
–20
–180
–40
–210
–60
–240
–80
1
10
100
1k
10k
100k
FREQUENCY (Hz)
1M
10M
PHASE (DEG)
The final step in the design process is to compute the Bode
plot for the entire loop using the designed compensation
network and confirm its phase margin and crossover
frequency. The complete loop Bode plot for this example
is shown in Figure 12. The loop crossover frequency is
37.8kHz which matches the design target and the phase
margin is approximately 60°.
PHASE
–270
100M
3536 F13
Figure 13. Complete Loop Bode Plot for
Buck-Boost Operation Mode
40° for VIN = VOUT = 3.3V. In fact in this mode of operation
the DC gain increase and often make this the most critical
region to compensate.
3536fa
21
LTC3536
Applications Information
In order to improve the stability also in buck-boost mode
of operation, the two compensation zeros could be move
to different frequency:
1
2πRFBCFB
= 5.4kHz
The new RFB value is:
1
RFB =
= 81.9kΩ ≈ 80.6kΩ
2π • (180pF ) • 10.8kHz
As consequence the fPOLE2 will move to higher frequency:
fPOLE2 =
2πCPOLERFB
= 532kHz
In buck mode there is no right-half plane zero and the
stability is normally achieved.
VO/VC
120
–60
100
–80
80
–100
PHASE
–140
GAIN
–160
0
–180
–20
–200
–40
–220
–60
–240
–80
–260
10
100
1k
10k
100k
FREQUENCY (Hz)
1M
10M
PHASE (DEG)
GAIN (dB)
40
1
100
–60
80
–80
60
–100
–120
PHASE
20
–140
GAIN
0
–160
–20
–180
–40
–200
–60
–220
–80
–240
–100
–260
1
10
100
1k
10k
100k
FREQUENCY (Hz)
1M
10M
–280
100M
3536 F15
Figure 15. Complete Loop Bode Plot for
Buck-Boost Operation Mode
Output Voltage Programming
The output voltage is set via the external resistor divider
comprised of resistors RTOP and RBOT. The resistor divider
values determine the output regulation voltage according to:
 R

VOUT = 0.6  1+ TOP  V
 RBOT 
–120
60
–100
–40
–120
1
As shown from Figures 14 and 15, the stability is now
improved for the buck-boost region (VIN = 3V) and remains
good for the boost region (VIN = 1.8V).
20
120
40
= 2 • fZERO2 = 10.8kHz
–20
PHASE (DEG)
fZERO1 =
2πR TOPCFF
GAIN (dB)
fZERO2 =
1
VO/VC
140
–280
100M
In addition to setting the output voltage, the value of RTOP
is instrumental in controlling the dynamics of the compensation network. When changing the value of this resistor,
care must be taken to understand the impact this will have
on the compensation network. As noted in the Input and
Peak Current Limit section, “for current limit feature to
be most effected, the Thevenin resistance (RTOP//RBOT)
from FB to ground should exceed 100k.”
VOUT
LTC3536
RTOP
FB
RBOT
3536 F14
3536 F16
Figure 14. Complete Loop Bode Plot for Boost Operation Mode
22
Figure 16. FB Resistor Network
3536fa
LTC3536
Applications Information
Switching Frequency Selection
Higher switching frequencies facilitate the use of smaller
inductors as well as smaller input and output filter capacitors which results in a smaller solution size and reduced
component height. However, higher switching frequencies
also generally reduce conversion efficiency due to the
increased switching losses. In addition, the maximum
voltage step-up ratio is reduced slightly at higher switching
frequencies as shown in the maximum duty cycle versus
switching frequency curve in the Typical Performance
Characteristics section of this data sheet.
PCB Layout Considerations
The LTC3536 buck-boost converter switches large currents
at high frequencies. Special attention should be paid to the
PC board layout to ensure a stable, noise-free and efficient
application circuit. A few key guidelines are provided:
1. The parasitic inductance and resistance of all circulating
high current paths should be minimized. This can be
accomplished by keeping the routes as short and as
wide as possible. Capacitor ground connections should
via down to the ground plane by way of the shortest
route possible. The bypass capacitors on PVIN and
VOUT should be placed as close to the IC as possible
and should have the shortest possible paths to ground.
2. The exposed pad is the electrical power ground connection for the LTC3536 in the DD package. Multiple vias
should connect the backpad directly to the ground plane.
In addition, maximization of the metallization connected
to the backpad will improve the thermal environment
and improve the power handling capabilities of the IC
in either package.
3. The components their connections with high current
should all be placed over a complete ground plane to
minimize loop cross-sectional areas. This minimizes
EMI and reduces inductive drops.
4. Connections to all of the components with high current
should be made as wide as possible to reduce the series
resistance. This will improve efficiency and maximize the
output current capability of the buck-boost converter
5. To prevent large circulating currents in the ground plane
from disrupting operation of the LTC3536, all smallsignal grounds should return directly to GND by way
of a dedicated Kelvin route. This includes the ground
connection for the RT pin resistor and the ground connection for the feedback network.
6. Keep the routes connecting to the high impedance,
noise sensitive inputs FB and RT as short as possible
to reduce noise pick-up. Example from MODE route in
case the chip is synchronized with external clock.
3536fa
23
LTC3536
Applications Information
Figure 17a. Fabrication Layer of Example PCB with 4 Layers
Figure 17b. Top Layer of Example PCB
24
Figure 17c. Bottom Layer of Example PCB
3536fa
LTC3536
Typical Applications
300kHz High Efficiency Li-Ion to 3.6V at 1A,
Pulsed with Manual Mode Control
Efficiency 3.6V, 300kHz vs Load Current
100
15µH*
10µF
SW1
VIN
SW2
VOUT
100pF
LTC3536
OFF ON
332k
845k
MODE/SYNC FB
SHDN
1nF
49.9k
RT
47µF
15k
VC
VOUT
3.6V
100µF 1A
80
EFFICIENCY (%)
VIN
3V TO 4.2V
90
70
60
50
40
30
20
GND
169k
VIN = 3V
VIN = 3.6V
VIN = 4.2V
10
0
0.001
3536 TA02
*COILCRAFT XAL4040
0.01
0.1
LOAD CURRENT (A)
1
3536 TA02b
USB to 5V Converter
Load Step
2.2µH*
USB POWER
4.3V TO 5.5V
47µF
SW1
VIN
SW2
VOUT
18pF
LTC3536
2MHz EXTERNAL CLOCK
OFF ON
49.9k
1100k
MODE/SYNC FB
SHDN
33k
RT
22.6k
VOUT
200mV/DIV
ILOAD
500mA/DIV
200pF
VC
GND
VOUT
5V
22µF 1A
150k
VIN = 4.3V
VOUT = 5V
100µs/DIV
3536 TA03b
3536 TA03
*COILCRAFT XFL4020
3536fa
25
LTC3536
Package Description
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
DD Package
10-Lead Plastic DFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1699 Rev C)
R = 0.125
TYP
0.40 ± 0.10
6
10
0.70 ±0.05
3.55 ±0.05
1.65 ±0.05
2.15 ±0.05 (2 SIDES)
3.00 ±0.10
(4 SIDES)
1.65 ± 0.10
(2 SIDES)
PIN 1 NOTCH
R = 0.20 OR
0.35 × 45°
CHAMFER
PIN 1
PACKAGE
TOP MARK
OUTLINE (SEE NOTE 6)
0.25 ± 0.05
0.75 ±0.05
0.200 REF
0.50
BSC
2.38 ±0.05
(2 SIDES)
5
0.00 – 0.05
1
(DD) DFN REV C 0310
0.25 ± 0.05
0.50 BSC
2.38 ±0.10
(2 SIDES)
BOTTOM VIEW—EXPOSED PAD
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
NOTE:
1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-2).
CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS OF VARIATION ASSIGNMENT
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE
TOP AND BOTTOM OF PACKAGE
MSE Package
12-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1666 Rev D)
BOTTOM VIEW OF
EXPOSED PAD OPTION
2.845 ± 0.102
(.112 ± .004)
5.23
(.206)
MIN
2.845 ± 0.102
(.112 ± .004)
0.889 ± 0.127
(.035 ± .005)
6
1
1.651 ± 0.102
(.065 ± .004)
1.651 ± 0.102 3.20 – 3.45
(.065 ± .004) (.126 – .136)
12
0.65
0.42 ± 0.038
(.0256)
(.0165 ± .0015)
BSC
TYP
RECOMMENDED SOLDER PAD LAYOUT
0.254
(.010)
0.35
REF
4.039 ± 0.102
(.159 ± .004)
(NOTE 3)
0.12 REF
DETAIL “B”
CORNER TAIL IS PART OF
DETAIL “B” THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
7
NO MEASUREMENT PURPOSE
0.406 ± 0.076
(.016 ± .003)
REF
12 11 10 9 8 7
DETAIL “A”
0° – 6° TYP
3.00 ± 0.102
(.118 ± .004)
(NOTE 4)
4.90 ± 0.152
(.193 ± .006)
GAUGE PLANE
0.53 ± 0.152
(.021 ± .006)
DETAIL “A”
1.10
(.043)
MAX
0.18
(.007)
SEATING
PLANE
0.22 – 0.38
(.009 – .015)
TYP
1 2 3 4 5 6
0.650
(.0256)
BSC
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
26
0.86
(.034)
REF
0.1016 ± 0.0508
(.004 ± .002)
MSOP (MSE12) 0910 REV D
3536fa
LTC3536
Revision History
REV
DATE
DESCRIPTION
A
11/11
Add new bullet Output Disconnect in Shutdown to Features bullet list.
PAGE NUMBER
1
In the Absolute Maximum Ratings section change (Notes 1, 2) to (Note 1) and (Note 2) to (Notes 2, 3).
2
In Electrical Characteristics table add conditions for Error Amplifier Sink Current and Error Amplifier Source Current.
3
In Pin Functions add Exposed Pad Pin 13 and remove last sentence to PGND pin description.
7
Change negative input of Peak Current Limit comparator to 3.4V and negative input of UVLO comparator to 1.75V.
8
Add new section Output Disconnect to Operations section.
12
3536fa
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
27
LTC3536
Typical Application
Backup Power Supply
4.7µH*
VSUPERCAP
1.8V TO 5.5V
SW1
VIN
10µF
SW2
VOUT
6.49k
MODE/SYNC
LTC3536
100k
300mA FOR VIN ≥ 1.8V
1A FOR VIN ≥ 3V
VSYS
3.3V
MAIN POWER
12V
DC/DC
22µF
845k
0.1µF
R2
47pF
SHDN
FB
RT
VC
49.9k
VH
VL
330pF
GND
182k
10pF
20k
20k
VCC
UV
OV
LTC2912-2
R1
866k
DIS
GND
6.04k
TMR
CRT
PWM BURST
3536 TA04
*COILCRAFT XFL4020
Related Parts
PART NUMBER DESCRIPTION
COMMENTS
LTC3440
600mA (IOUT) 2MHz Synchronous Buck-Boost DC/DC Converter
95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 2.5V, IQ = 25µA,
ISD ≤ 1µA, 10-Lead MSOP and DFN Packages
LTC3442
1.2A (IOUT) 2MHz Synchronous Buck-Boost DC/DC Converter with
Programmable Burst Mode Operation
VIN: 2.4V to 5.5V, VOUT: 2.4 to 5.25V, IQ = 35µA, ISD ≤ 1µA,
DFN Package
LTC3444
400mA (IOUT) 2MHz Synchronous Buck-Boost DC/DC Converter
VIN: 2.75V to 5.5V, VOUT: 0.5 to 5V, IQ = 35µA, ISD ≤ 1µA, DFN
Package
LTC3101
Wide VIN, Multi-Output DC/DC Converter and PowerPath™ Controller
VIN: 1.8V to 5.5V, VOUT: 1.5V to 5.25V, IQ = 38µA, ISD ≤ 15µA,
QFN Package
LTC3113
3A (IOUT), 2MHz Synchronous Buck-Boost DC/DC Converter
VIN: 1.8V to 5.5V, VOUT: 1.8V to 5.25V, IQ = 30µA, ISD < 1µA,
DFN and TSSOP Packages
LTC3533
2A (IOUT), 2MHz Synchronous Buck-Boost DC/DC Converter
VIN: 1.8V to 5.5V, VOUT: 1.8V to 5.25V, IQ = 40µA, ISD < 1QA,
DFN Package
LTC3534
7V, 500mA (IOUT), 1MHz Synchronous Buck-Boost DC/DC Converter
94% Efficiency, VIN: 2.4V to 7V, VOUT: 1.8V to 7V, IQ = 25µA,
ISD < 1µA, DFN and GN Packages
LTC3530
600mA (IOUT), 2MHz Synchronous Buck-Boost DC/DC Converter
VIN: 1.8V to 5.5V, VOUT: 1.8V to 5.25V, IQ = 12µA, ISD < 2µA,
QFN Package
LTC3112
2.5A (IOUT), 15V Synchronous Buck-Boost DC/DC Converter
VIN: 2.7V to 15V, VOUT: 2.5V to 14V, IQ = 40µA, ISD < 1µA,
DFN and TSSOP Packages
LTC3127
1A (IOUT), 1.2MHz Synchronous Buck-Boost DC/DC Converter with
Programmable Input Current Limit
96% Efficiency, VIN: 1.8V to 5.5V, VOUT: 1.8V to 5.25V, IQ =
30µA,
ISD < 4µA, DFN and MSOP Packages
LTC3780
High Efficiency, Synchronous, 4-Switch Buck-Boost Controller
98% Efficiency, VIN: 4V to 36V, VOUT: 0.8V < VOUT < 30V,
ISD = 55µA, SSOP and QFN Packages
LTC3785
10V, High Efficiency, Synchronous, No RSENSE™ Buck-Boost
Controller
96% Efficiency, VIN and VOUT: 2.7V to 10V, ISD = 15µA,
IQ = 86µA, QFN Package
LTC3789
High Efficiency, Synchronous, 4-Switch Buck-Boost Controller
98% Efficiency, VIN: 4V to 38V, VOUT: 0.8V < VOUT < 38V,
ISD = 40µA, SSOP and QFN Packages
28 Linear Technology Corporation
3536fa
LT 1111 REV A • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
 LINEAR TECHNOLOGY CORPORATION 2011