MIC2171 Micrel MIC2171 100kHz 2.5A Switching Regulator Preliminary Information General Description Features The MIC2171 is a complete 100kHz SMPS current-mode controller with an internal 65V 2.5A power switch. • • • • • • • • • • Although primarily intended for voltage step-up applications, the floating switch architecture of the MIC2171 makes it practical for step-down, inverting, and Cuk configurations as well as isolated topologies. Operating from 3V to 40V, the MIC2171 draws only 7mA of quiescent current, making it attractive for battery operated supplies. 2.5A, 65V internal switch rating 3V to 40V input voltage range Current-mode operation, 2.5A peak Internal cycle-by-cycle current limit Thermal shutdown Twice the frequency of the LM2577 Low external parts count Operates in most switching topologies 7mA quiescent current (operating) Fits LT1171/LM2577 TO-220 and TO-263 sockets Applications The MIC2171 is available in a 5-pin TO-220 or TO-263 for –40°C to +85°C operation. • • • • Laptop/palmtop computers Battery operated equipment Hand-held instruments Off-line converter up to 50W (requires external power switch) • Predriver for higher power capability 4 Typical Applications +5V (4.75V min.) IN VIN 4V to 6V C1* 47µF L1 15µH D1 VOUT +12V, 0.25A 1N5822 R1 10.7k 1% SW MIC2171 FB COMP R3 1k GND C3 1µF C2 470µF R2 1.24k 1% VOUT 5V, 0.5A T1 R4* C1 47µF D1* IN SW MIC2171 COMP R3 1k C3* D2 1N5818 C4 470µF 1:1.25 LPRI = 12µH FB GND C2 1µF * Locate near MIC2171 when supply leads > 2" * Optional voltage clipper (may be req’d if T1 leakage inductance too high) Figure 1. MIC2171 5V to 12V Boost Converter 1997 Figure 2. MIC2171 5V Flyback Converter 4-3 R1 3.74k 1% R2 1.24k 1% MIC2171 Micrel Ordering Information Part Number Temperature Range Package MIC2171BT –40°C to +85°C 5-lead TO-220 MIC2171BU –40°C to +85°C 5-lead TO-263 Pin Configuration 5 4 3 2 1 IN SW GND FB COMP 5 4 3 2 1 Tab GND IN SW GND FB COMP Tab GND 5-lead TO-220 (BT) 5-lead TO-263 (BU) Pin Description Pin Number Pin Name Pin Function 1 COMP 2 FB Feedback: Inverting input of error amplifier. Connect to external resistive divider to set power supply output voltage. 3 GND Ground: Connect directly to the input filter capacitor for proper operation (see applications info). 4 SW 5 IN Frequency Compensation: Output of transconductance-type error amplifier. Primary function is for loop stabilization. Can also be used for output voltage soft-start and current limit tailoring. Power Switch Collector: Collector of NPN switch. Connect to external inductor or input voltage depending on circuit topology. Supply Voltage: 3.0V to 40V 4-4 1997 MIC2171 Micrel Junction Temperature ................................ –55°C to 150°C Thermal Resistance θJA 5-lead TO-220, Note 1................................. 45°C/W θJA 5-lead TO-263, Note 2................................. 45°C/W Storage Temperature ............................... –65°C to +150°C Soldering (10 sec.) .................................................. +300°C Absolute Maximum Ratings Input Voltage (VIN) ........................................................ 40V Switch Voltage (VSW) .................................................... 65V Feedback Voltage (transient, 1ms) (VFB) ................... ±15V Operating Temperature Range ...................... –40 to +85°C Electrical Characteristics VIN = 5V; TA = 25°C, bold values indicate –40°C ≤ TA ≤ +85°C; unless noted. Parameter Conditions Min Typ Max Units 1.220 1.214 1.240 1.264 1.274 V V Reference Section Feedback Voltage (VFB) VCOMP = 1.24V Feedback Voltage Line Regulation 3V ≤ VIN ≤ 40V VCOMP = 1.24V .06 Feedback Bias Current (IFB) VFB = 1.24V 310 750 1100 nA nA %/V Error Amplifier Section Transconductance (gm) ∆ICOMP = ±25µA 3.0 2.4 3.9 6.0 7.0 µA/mV µA/mV Voltage Gain (AV) 0.9V ≤ VCOMP ≤ 1.4V 400 800 2000 V/V Output Current VCOMP = 1.5V 125 100 175 350 400 µA µA Output Swing High Clamp, VFB = 1V Low Clamp, VFB = 1.5V 1.8 0.25 2.1 0.35 2.3 0.52 V V Compensation Pin Threshold Duty Cycle = 0 0.8 0.6 0.9 1.08 1.25 V V 0.37 0.50 0.55 Ω Ω 5 5.5 5 A A A Output Switch Section ON Resistance ISW = 2A, VFB = 0.8V Current Limit Duty Cycle = 50%, TJ ≥ 25°C Duty Cycle = 50%, TJ < 25°C Duty Cycle = 80%, Note 3 2.5 2.5 2.0 3.6 4.0 3.0 Breakdown Voltage (BV) 3V ≤ VIN ≤ 40V ISW = 5mA 65 75 Frequency (fO) 88 85 100 112 115 kHz kHz Duty Cycle [δ(max)] 80 90 95 % 2.7 3.0 V V Oscillator Section Input Supply Voltage Section Minimum Operating Voltage Quiescent Current (IQ) 3V ≤ VIN ≤ 40V, VCOMP = 0.6V, ISW = 0 7 9 mA Supply Current Increase (∆IIN) ∆ISW = 2A, VCOMP = 1.5V, during swich on-time 9 20 mA General Note Devices are ESD sensitive. Handling precautions required. Note 1 Mounted vertically, no external heat sink, 1/4 inch leads soldered to PC board containing approximently 4 inch squared copper area surrounding leads. Note 2 All ground leads soldered to approximently 2 inches squared of horizontal PC board copper area. Note 3 For duty cycles (δ) between 50% and 95%, minimum guaranteed switch current is ICL = 1.66 (2-δ) Amp (Pk). 1997 4-5 4 MIC2171 Micrel Typical Performance Characteristics 2.9 2.7 2.6 Switch Current = 2A 2.5 2.4 -50 0 50 100 Temperature (°C) 700 600 500 400 300 200 100 0 -100 150 Supply Current ISW = 0 D.C. = 90% 11 10 D.C. = 50% 9 8 7 D.C. = 0% 0 10 20 30 VIN Operating Voltage (V) 40 Frequency (kHz) Switch ON Voltage (V) 0.8 0.6 TJ = 125°C 0 40 9 8 δ = 90% 30 20 δ = 50% 10 0 1 2 3 Switch Current (A) 40 Supply Current 10 VCOMP = 0.6V 7 6 5 4 3 2 1 0 -100 4 -50 0 50 100 Temperature (°C) 150 Current Limit 8 1 2 Switch Current (A) 90 80 60 -50 3 Error Amplifier Gain 0 50 100 Temperature (°C) –40°C 4 25°C 125°C 2 Error Amplifier Gain 3.0 2.5 2.0 1.5 1.0 0.5 150 0 20 40 60 80 Duty Cycle (%) 100 Error Amplifier Phase -30 6000 0 5000 30 Phase Shift (°) 4.0 3.5 -50 0 50 100 Temperature (°C) 6 0 150 7000 Transconductance (µS) Transconductance (µA/mV) 10 20 30 VIN Operating (V) 70 5.0 4.5 0.0 -100 0 150 100 0.2 0 TJ = -40°C -5 110 TJ = –40°C 0.4 -3 -4 Oscillator Frequency TJ = 25°C 1.0 TJ = 25°C -1 -2 120 1.4 1.2 TJ = 125°C 1 0 50 0 Switch On-Voltage 1.6 4 3 2 Switch Current (A) 13 12 5 Supply Current Average Supply Current (mA) Supply Current (mA) 15 14 -50 0 50 100 Temperature (°C) Supply Current (mA) 2.3 -100 Feedback Voltage Change (mV) 800 2.8 6 5 Feedback Voltage Line Regulation Feedback Bias Current Feedback Bias Current (nA) Minimum Operating Voltage (V) Minimum Operating Voltage 4000 3000 2000 60 90 120 150 1000 0 180 1 10 100 1000 Frequency (kHz) 4-6 10000 210 1 10 100 1000 Frequency (kHz) 10000 1997 MIC2171 Micrel Block Diagram MIC2171 IN Reg. D1 2.3V SW Anti-Sat. 100kHz Osc. Logic Q1 Driver Comparator FB 1.24V Ref. Current Amp. Error Amp. COMP GND response. Inherent cycle-by-cycle current limiting greatly improves the power switch reliability and provides automatic output current limiting. Finally, current-mode operation provides automatic input voltage feed forward which prevents instantaneous input voltage changes from disturbing the output voltage setting. Functional Description Refer to “Block Diagram MIC2171”. Internal Power The MIC2171 operates when VIN is ≥ 2.6V. An internal 2.3V regulator supplies biasing to all internal circuitry including a precision 1.24V band gap reference. Anti-Saturation The anti-saturation diode (D1) increases the usable duty cycle range of the MIC2171 by eliminating the base to collector stored charge which would delay Q1’s turnoff. PWM Operation The 100kHz oscillator generates a signal with a duty cycle of approximately 90%. The current-mode comparator output is used to reduce the duty cycle when the current amplifier output voltage exceeds the error amplifier output voltage. The resulting PWM signal controls a driver which supplies base current to output transistor Q1. Compensation Loop stability compensation of the MIC2171 can be accomplished by connecting an appropriate network from either COMP to circuit ground (see typical Applications) or COMP to FB. Current-Mode Advantages The error amplifier output (COMP) is also useful for soft start and current limiting. Because the error amplifier output is a transconductance type, the output impedance is relatively high which means the output voltage can be easily clamped or adjusted externally. The MIC2171 operates in current mode rather than voltage mode. There are three distinct advantages to this technique. Feedback loop compensation is greatly simplified because inductor current sensing removes a pole from the closed loop 1997 4-7 4 MIC2171 Micrel The device operating losses are the dc losses associated with biasing all of the internal functions plus the losses of the power switch driver circuitry. The dc losses are calculated from the supply voltage (VIN) and device supply current (IQ). The MIC2171 supply current is almost constant regardless of the supply voltage (see “Electrical Characteristics”). The driver section losses (not including the switch) are a function of supply voltage, power switch current, and duty cycle. Applications Information Soft Start A diode-coupled capacitor from COMP to circuit ground slows the output voltage rise at turn on (Figure 3). VIN IN ) ( ( P(bias+driver) = VIN IQ + VIN(min) × ISW × ∆IIN MIC2171 where: P(bias+driver) = device operating losses VIN(min) = supply voltage = VIN – VSW IQ = typical quiescent supply current ICL = power switch current limit ∆IIN = typical supply current increase As a practical example refer to Figure 1. COMP D1 D2 R1 C1 ) C2 Figure 3. Soft Start VIN = 5.0V IQ = 0.007A ICL = 2.21A δ = 66.2% (0.662) The additional time it takes for the error amplifier to charge the capacitor corresponds to the time it takes the output to reach regulation. Diode D1 discharges C1 when VIN is removed. Current Limit Then: VIN VIN(min) = 5 – (2.21 × 0.37) = 4.18V IN P(bias SW GND R1 VOUT COMP R3 Q1 C1 C2 = (5 × 0.007) + (4.18 × 2.21 × .009) P(bias+driver) = 0.1W Power switch dissipation calculations are greatly simplified by making two assumptions which are usually fairly accurate. First, the majority of losses in the power switch are due to on-losses. To find these losses, assign a resistance value to the collector/emitter terminals of the device using the saturation voltage versus collector current curves (see Typical Performance Characteristics). Power switch losses are calculated by modeling the switch as a resistor with the switch duty cycle modifying the average power dissipation. MIC2171 FB + driver) ICL ≈ 0.6V/R2 Note: Input and output returns not common. R2 PSW = (ISW)2 RSW δ where: δ = duty cycle Figure 4. Current Limit The maximum current limit of the MIC2171 can be reduced by adding a voltage clamp to the COMP output (Figure 4). This feature can be useful in applications requiring either a complete shutdown of Q1’s switching action or a form of current fold-back limiting. This use of the COMP output does not disable the oscillator, amplifiers or other circuitry, therefore the supply current is never less than approximately 5mA. δ= VOUT + VF – VIN(min) VOUT + VF VSW = ICL (RSW) VOUT = output voltage VF = D1 forward voltage drop at IOUT From the Typical performance Characteristics: Thermal Management Although the MIC2171 family contains thermal protection circuitry, for best reliability, avoid prolonged operation with junction temperatures near the rated maximum. RSW = 0.37Ω Then: The junction temperature is determined by first calculating the power dissipation of the device. For the MIC2171, the total power dissipation is the sum of the device operating losses and power switch losses. PSW = (2.21)2 × 0.37 × 0.662 PSW) = 1.2W P(total) = 1.2 + 0.1 P(total) = 1.3W 4-8 1997 MIC2171 Micrel The junction temperature for any semiconductor is calculated using the following: mode is preferred because the feedback control of the converter is simpler. TJ = TA + P(total) θJA When L1 discharges its current completely during the MIC2171 off-time, it is operating in discontinuous mode. Where: L1 is operating in continuous mode if it does not discharge completely before the MIC2171 power switch is turned on again. TJ = junction temperature TA = ambient temperature (maximum) P(total) = total power dissipation θJA = junction to ambient thermal resistance For the practical example: Discontinuous Mode Design Given the maximum output current, solve equation (1) to determine whether the device can operate in discontinuous mode without initiating the internal device current limit. TA = 70°C θJA = 45°C/W (TO-220) Then: TJ = 70 + (1.24 × 45) TJ = 126°C This junction temperature is below the rated maximum of 150°C. (1) IOUT (1a) δ= VOUT + VF – VIN(min) Grounding ICL = internal switch current limit ICL = 2.5A when δ < 50% ICL = 1.67 (2 – δ) when δ ≥ 50% (Refer to Electrical Characteristics.) IOUT = maximum output current VIN(min) = minimum input voltage = VIN – VSW δ = duty cycle VOUT = required output voltage VF = D1 forward voltage drop For the example in Figure 1. VIN IN SW MIC2171 FB COMP IOUT = 0.25A ICL = 1.67 (2–0.662) = 2.24A VIN(min) = 4.18V δ = 0.662 VOUT = 12.0V VF = 0.36V (@ .26A, 70°C) Single point ground Figure 5. Single Point Ground A single point ground is strongly recommended for proper operation. The signal ground, compensation network ground, and feedback network connections are sensitive to minor voltage variations. The input and output capacitor grounds and power ground conductors will exhibit voltage drop when carrying large currents. Keep the sensitive circuit ground traces separate from the power ground traces. Small voltage variations applied to the sensitive circuits can prevent the MIC2171 or any switching regulator from functioning properly. Then: IOUT 2.235 × 4.178 × 0.662 2 ≤ 12 IOUT ≤ 0.258A This value is greater than the 0.25A output current requirement, so we can proceed to find the minimum inductance value of L1 for discontinuous operation at POUT. Boost Conversion Refer to Figure 1 for a typical boost conversion application where a +5V logic supply is available but +12V at 0.25A is required. (2) L1 ≥ (VIN δ)2 2 POUT fSW Where: The first step in designing a boost converter is determining whether inductor L1 will cause the converter to operate in either continuous or discontinuous mode. Discontinuous 1997 VOUT + VF Where: Refer to Figure 5. Heavy lines indicate high current paths. GND ICL δ V 2 IN(min) ≤ VOUT POUT = 12 × 0.25 = 3W fSW = 1×105Hz (100kHz) 4-9 4 MIC2171 Micrel down (failure) of the MIC2171’s internal power switch. For our practical example: (3) (4.178 Discontinuous Mode Design × 0.662) 2 × 3.0 × 1× 105 L1 ≥ 12.4µH (use 15µH) Equation (3) solves for L1’s maximum current value. L1 ≥ IL1(peak) = 2 When designing a discontinuous flyback converter, first determine whether the device can safely handle the peak primary current demand placed on it by the output power. Equation (8) finds the maximum duty cycle required for a given input voltage and output power. If the duty cycle is greater than 0.8, discontinuous operation cannot be used. VIN T ON L1 Where: (8) TON = δ / fSW = 6.62×10-6 sec δ ≥ ( 2 POUT ICL VIN(min) – VSW ) For a practical example let: (see Figure 2) 4.178 × 6.62 × 10-6 15 × 10-6 IL1(peak) = 1.84A IL1(peak) = POUT = 5.0V × 0.5A = 2.5W VIN = 4.0V to 6.0V ICL = 2.5A when δ < 50% 1.67 (2 – δ) when δ ≥ 50% Use a 15µH inductor with a peak current rating of at least 2A. Flyback Conversion Then: Flyback converter topology may be used in low power applications where voltage isolation is required or whenever the input voltage can be less than or greater than the output voltage. As with the step-up converter the inductor (transformer primary) current can be continuous or discontinuous. Discontinuous operation is recommended. Figure 2 shows a practical flyback converter design using the MIC2171. Switch Operation During Q1’s on time (Q1 is the internal NPN transistor—see block diagrams), energy is stored in T1’s primary inductance. During Q1’s off time, stored energy is partially discharged into C4 (output filter capacitor). Careful selection of a low ESR capacitor for C4 may provide satisfactory output ripple voltage making additional filter stages unnecessary. C1 (input capacitor) may be reduced or eliminated if the MIC2171 is located near a low impedance voltage source. Output Diode The output diode allows T1 to store energy in its primary inductance (D2 nonconducting) and release energy into C4 (D2 conducting). The low forward voltage drop of a Schottky diode minimizes power loss in D2. Frequency Compensation ( VIN(min) = VIN – ICL × RSW VIN(min) = 4 – 0.78V VIN(min) = 3.22V δ ≥ 0.74 (74%), less than 0.8 so discontinous is permitted. A few iterations of equation (8) may be required if the duty cycle is found to be greater than 50%. Calculate the maximum transformer turns ratio a, or NPRI/NSEC, that will guarantee safe operation of the MIC2171 power switch. (9) Voltage Clipper Care must be taken to minimize T1’s leakage inductance, otherwise it may be necessary to incorporate the voltage clipper consisting of D1, R4, and C3 to avoid second break- a ≤ V CE FCE – VIN(max) V SEC Where: a = transformer maximum turns ratio VCE = power switch collector to emitter maximum voltage FCE = safety derating factor (0.8 for most commercial and industrial applications) VIN(max) = maximum input voltage VSEC = transformer secondary voltage (VOUT + VF) For the practical example: A simple frequency compensation network consisting of R3 and C2 prevents output oscillations. High impedance output stages (transconductance type) in the MIC2171 often permit simplified loop-stability solutions to be connected to circuit ground, although a more conventional technique of connecting the components from the error amplifier output to its inverting input is also possible. ) VCE = 65V max. for the MIC2171 FCE = 0.8 VSEC = 5.6V Then: a ≤ 65 × 0.8 – 6.0 5.6 a ≤ 8.2 (NPRI/NSEC) Next, calculate the maximum primary inductance required to store the needed output energy with a power switch duty cycle of 55%. 4-10 1997 MIC2171 (10) LPRI ≥ Micrel 0.5 fSW VIN(min)2 TON2 (12) POUT Where: 0.5 × 1× 105 × (3.22) 2 ( × 7.4 × 10-6 )2 2.5 (13) LPRI ≥ 11.4µH Use an 12µH primary inductance to overcome circuit inefficiencies. To complete the design the inductance value of the secondary is found which will guarantee that the energy stored in the transformer during the power switch on time will be completed discharged into the output during the off-time. This is necessary when operating in discontinuous-mode. L SEC ≤ (14) POUT IPEAK(pri) = VIN(min) T ON LPRI IPEAK(pri) = 3.22 × 7.6 × 10-6 12µH IPEAK(pri) = 2.1A Now find the minimum reverse voltage requirement for the output rectifier. This rectifier must have an average current rating greater than the maximum output current of 0.5A. VBR ≥ ( VIN(max) + V OUT a ) FBR a Where: LSEC = maximum secondary inductance TOFF = power switch off time VBR = output rectifier maximum peak reverse voltage rating a = transformer turns ratio (1.2) FBR = reverse voltage safety derating factor (0.8) Then: 0.5 × 1× 105 × (5.41) 2 ( × 2.6 × 10-6 )2 Then: 2.5 LSEC ≤ 7.9µH Finally, recalculate the transformer turns ratio to insure that it is less than the value earlier found in equation (9). 1997 11.4 = 1.20 7.9 So: 0.5 f SW V SEC 2 T OFF 2 Where: L SEC ≤ a ≤ This ratio is less than the ratio calculated in equation (9). When specifying the transformer it is necessary to know the primary peak current which must be withstood without saturating the transformer core. Then: (11) LPRI L SEC Then: LPRI = maximum primary inductance fSW = device switching frequency (100kHz) VIN(min) = minimum input voltage TON = power switch on time LPRI ≥ a ≤ VBR ≥ 6.0 + (5.0 × 1.2) 0.8 × 1.2 VBR ≥ 12.5V A 1N5817 will safely handle voltage and current requirements in this example. 4-11 4 MIC2171 Micrel Forward Converters Micrel’s MIC2171 can be used in several circuit configurations to generate an output voltage which is less than the input voltage (buck or step-down topology). Figure 7 shows the MIC2171 in a voltage step-down application. Because of the internal architecture of these devices, more external components are required to implement a step-down regulator than with other devices offered by Micrel (refer to the LM257x or MIC457x family of buck switchers). However, for step-down conversion requiring a transformer (forward), the MIC2171 is a good choice. A 12V to 5V step-down converter using transformer isolation (forward) is shown in Figure 7. Unlike the isolated flyback converter which stores energy in the primary inductance during the controller’s on-time and releases it to the load during the off-time, the forward converter transfers energy to the output during the on-time, using the off-time to reset the transformer core. In the application shown, the transformer core is reset by the tertiary winding discharging T1’s peak magnetizing current through D2. For most forward converters the duty cycle is limited to 50%, allowing the transformer flux to reset with only two times the input voltage appearing across the power switch. Although during normal operation this circuit’s duty cycle is well below 50%, the MIC2172 has a maximum duty cycle capability of 90%. If 90% was required during operation (start-up and high load currents), a complete reset of the transformer during the off-time would require the voltage across the power switch to be ten times the input voltage. This would limit the input voltage to 6V or less for forward converter applications. To prevent core saturation, the application given here uses a duty cycle limiter consisting of Q1, C4 and R3. Whenever the MIC2171 exceeds a duty cycle of 50%, T1’s reset winding current turns Q1 on. This action reduces the duty cycle of the MIC2171 until T1 is able to reset during each cycle. T1 1:1:1 D3 1N5819 VIN 12V R1* C2* L1 100µH D4 1N5819 VOUT 5V, 1A R4 C5 3.74k 470µF 1% D1* IN SW MIC2171 C1 22µF GND FB COMP R2 1k C3 1µF D2 1N5819 Q1† R3† R5 1.24k 1% C4† * Voltage clipper † Duty cycle limiter Figure 7. MIC2171 Forward Converter 4-12 1997