FAIRCHILD PQ

www.fairchildsemi.com
AN-9737
Design Guideline for Single-Stage Flyback AC-DC
Converter Using FL6961 for LED Lighting
Summary
Basic Operation: High Power Factor
Flyback Converter
This application note presents single-stage Power Factor
Correction (PFC) and focuses on how to select and design
the flyback transformer for 16.8W (24V/0.7A) solution for
universal input for LED lighting applications using FL6961.
The flyback converter using FL6961 operates in Critical
Conduction Mode (CRM) and has functions such as CC/CV
feedback circuit, soft-starting, and the cycle-by-cycle current
limit for LED lighting applications.
The basic idea of achieving high power factor (PF) flyback
converter is to use a Critical Conduction Mode (CRM) PFC
controller. The conventional PFC IC, such as FL6961, has
constant on-time and variable off-time control method,
which means the input average current always follows the
input voltage shape.
Figure 1 shows the typical application schematic of singlestage PFC topology. The main difference of normal CRM
boost converter is that single-stage PFC doesn’t use a large
electrolytic capacitor after the full rectification diode.
Normally, the single-stage PFC method uses a small
capacitor (C1 in Figure 1) to act as a noise filter to attenuate
high-frequency components and doesn’t use the INV pin for
output voltage regulation.
Introduction
These days, engineers use various types of LEDs for general
lighting systems because of their long life, excellent
efficacy, price, environmental benefits, and requirements
from end users. At the same time, high power factor (PF),
isolation for safety, and constant current control (CC) for
constant LED color are becoming requirements.
Conventional regulation is the minimum power factor
correction for input power base above 25W, but many want
to reduce power ratings and the new Energy-Star directive
for solid-state lighting requires a power factor greater than
0.9 for commercial applications. Expect PF regulations to
become more stringent.
T1
BR
D3
C4
R8
R5
D2
R1
D1
C5
U101
R2
VCC
INV
1
Fuse
C1
MOT
R3
C2
CS
EMI filter
C3
2
3
4
8
FL6961
COMP
7
6
OUT
GND
R6
Q1
R7
ZCD
5
R4
R8
Feedback
Figure 1. Simplified Schematic of High-Power Factor Flyback Converter with FLS6961
© 2011 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 4/13/11
www.fairchildsemi.com
AN-9737
APPLICATION NOTE
(NS) and naturally decreases to zero. The average current of
the secondary side is:
Figure 2 shows typical waveforms of the simplified circuit
of a flyback converter with CRM. When the MOSFET (Q1)
turns on, the primary current in primary side linearly
increases and is clamped at a certain internal level because
the FL6961 doesn’t have cycle-by-cycle current limit like a
conventional current mode control IC (such as FAN7527B).
Its peak level is determined by the primary magnetizing
inductance value and the fixed on-time. Instead of the cycleby-cycle primary current limit, the FL6961 has an overcurrent protection (OCP) function. If the current sensing
signal is larger than internal detection level, the FL6961
doesn’t get output signal for operating the MOSFET (Q1).
I AVG ( DIODE ) =
(3)
Since the diode forward-voltage drop decreases as current
decreases, the output voltage reflects the primary winding
and adds additional voltage due to overshoot made by
resonance between the leakage inductance on primary-side
winding and parasitic capacitance on the MOSFET (Q1). As
a result, a superimposed voltage occurs on the MOSFET
during off-time as:
VMOSFET ( off ) = V IN + V R + VOS
IDS (MOSFET Drain-to-Source Current))
IPK ( MOSFET
1 NP
I PK toff
2 NS
)
IAVG (MOSFET
where VR is the reflected voltage and VOS is the voltage
overshoot term.
)
The reflected voltage, VR, is affected by the turns ratio
between the primary and secondary side of the transformer
and the output voltage, calculated as:
time
ID (Diode Current)
(4)
IPK ( DIODE )
IAVG (DIODE
VR =
)
NP
VO
NS
(5)
Figure 3 shows the ideal waveforms of the primary-side
current at MOSFET (Q1) and the secondary-side current at
the diode. The input peak and average current on the
primary side follows input voltage instantaneously.
Normally, secondary-side current on the diode is larger than
the primary side because of the turns ratio.
time
VDS (MOSFET Voltage)
VOS
VR
VIN
time
tOFF
tON
tS
Figure 2. Key Waveforms of Flyback Converter on
CRM
The FL6961 has a constant on-time across the whole range.
The input average current always follows the peak input
current, as shown in the equation:
1
I AVG ( MOSFET ) = I PK tON
(1)
2
This is also proportional to the instantaneous input voltage.
This means the input current shape is always the same as the
input voltage shape. The reverse diode voltage is linearly
increased and is equal to:
VPK ( DIODE ) = VO + VIN
NS
NP
Figure 3. Ideal Waveforms
(2)
During the MOSFET off-time, which is also the diode ontime; the input current instantly drops to zero, the diode in
the secondary side conducts, and the diode current linearly
decreases. The peak current of the secondary side is the
same as the multiplication of the primary peak current and
turns ratio between the primary side (NP) and secondary side
© 2011 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 4/13/11
www.fairchildsemi.com
2
AN-9737
APPLICATION NOTE
As a result, designers should consider two conditions before
component selection: voltage and current capacity on
primary-side MOSFET(Q1) and secondary-side diode (D3)
to make a stable system with margin.
Figure 4 shows a guide to deciding two components on the
boundary condition of flyback converter topology.
Figure 4. Boundary Conditions of Flyback Converter
Topology (Refer to AN-8025)
Design Example
P = I o (Vo + Vd ) = 0.7(24 + 1) = 17.5 [W]
A. Transformer Design
A design guideline of 16.8W single-stage flyback AC-DC
converter using FL6961 is presented. The applied system
parameters are shown in Table 1.
Table 1.
Step 4. Calculate the maximum input current, Imax:
Po
I in (max) =
Vminη
System Parameters
Parameter
Main Input Voltage Range, VAC(main)
=
17.5
= 0.168 [A]
( 2 × 90)(0.82)
Step 5. Calculate the MOSFET voltage drop, Vvd:
Value
Vvd = I in (max) RMOS = 0.168 [V]
90V~265V
Output Voltage, VOUT
24V
Step 6. Calculate the primary voltage on transformer, Vp:
Output Current, IOUT
0.7A
VP = Vmin −V vd= 127 − 0.168 ≈ 127 [V]
Vp=126.83 use 127
Minimum Switching Frequency at VAC(min)_pk
50kHz
Diode Voltage Drop, Vd
1V
MOSFET On Resistance, RMOS
1Ω
Window Utilization
0.4
Target System Efficiency
0.82
Maximum Duty at Vac(min)_pk
0.35
Operating Maximum Flux Density
0.35
Regulation, α
0.5%
Step 7. Calculate the primary peak current, Ippk:
I ppk =
2TP
2( 20 × 10 −6 )(17.5)
=
= 0.96 [A]
ηV p t on (max) 0.82(127)(7 × 10 −6 )
Step 8. Calculate the primary rms current, Iprms:
I prms = I ppk
Note:
1. Regulation is strongly related with the copper loss and
0.5% regulation means 0.084W loss on transformer.
t on
(7 ×10 −6 )
= 0.96
= 0.32 [A]
3T
3(20 ×10 −6 )
Step 9. Calculate the required minimum inductance, L:
L=
There are many ways to decide core and coil size and turns,
such as using AL value and following common practices. In
this note, use the Kg value related with the core geometry to
find optimum core and coil information.
V p t on (max)
I ppk
=
127(7 ×10 −6 )
= 0.926 [mH]
0.96
L=0.926[mH] use 1[mH]
Step 10. Calculate the energy-handing capability in wattseconds, w-s:
Step 1. Calculate the total period, T:
1
= 20 [µs]
f
Step 2. Calculate the maximum on-time at MOSFET in
primary side.
ENG =
T=
2
LI ppk
2
=
(1× 10 −3 )(0.96 2 )
= 0.0004608 [w-s]
2
Step11. Calculate the electrical conditions, Ke:
K e = 0.145PBm2 × 10−4 = 0.145(17.5)(0.352 ) × 10−4 = 0.00003108
−6
t on = TDmax = (20 × 10 )(0.35) = 7 [µs]
Step 12. Calculate the core geometry, Kg:
Step 3. Calculate the output power:
© 2011 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 4/13/11
www.fairchildsemi.com
3
AN-9737
Kg =
APPLICATION NOTE
5
( ENG ) 2
(0.0004608) 2
=
= 0.0136 [cm ]
0.00003108(0.5)
K eα
AW ( B ) =
Step 23. Calculate the skin depth at expected operating
frequency at low input voltage. The skin depth is the radius
of the wire.
Step 13. See Table 2 for core size.
To prevent core saturation, select a little big core after
comparing two Kg values: calculate value at Step 12 vs. the
existing value in Table 2.
γ=
The PQ-42016 has a little bit big Kg value (0.01327) in
Table 2 with 2500 permeability (µi).
2 ( ENG ) × 10
B m AP K u
4
=
2
WireA = π (r 2 ) = 0.0027535 [cm ]
2
2 ( 0 . 0004608 ) × 10 4
= 265 [A/cm ]
0 . 35 ( 0 . 2484 )( 0 . 4 )
Step 25. Select a wire size with the required area from Table
4. If the area is not within 10% of the required area, then go
to the next smallest size.
Step 15. Calculate the required wire area. AW(B):
AW ( B ) =
I rms
0 . 32
=
= 0 . 001207
J
265
[cm2]
AWG=#23
AW(B)=0.00259[cm2]
Step 16. Calculate the number of turns, N:
N =
µΩ/cm=666
WaKu
0 . 4283 × 0 . 4
=
= 141 . 93 [T]
0 . 001207
Aw( B )
Step 26. Calculate the required number of primary strands,
Snp:
N=141.93; use 142 turns.
S np =
Step 17. Calculate the required gap, lg:
lg =
0.4π ( N)(∆I ) ×10−4 0.4π (142)(0.96) ×10−4
=
= 0.0489 [cm]
∆Bm
0.35
N=
MPL
µi
0.4π ( Ac )
=
Ns =
(V p Dmax )
N aux =
Step 19. Calculate the fringing flux, F:
Ac
ln
2G
0.0489 2(1.001)
) = (1 +
ln
) = 1.238
lg
0.0489
0.58
−8
(0.4π )( Ac ) F (10 )
=
I spk =
74(15 + 1)(1 − 0.35)
= 17.31
( 2 × 90)(0.35)
(1 − Dmax )
(1 − 0.35)
= 2.153
= 1.0021 [A]
3
3
Step 30. Calculate the secondary wire area, Asw(B):
I PK
0.96
) F (10 − 4 ) (0.4π )(74)(
)(1.238)(10 − 4 )
[T]
2
2
=
= 0.113
lg
0.0489
ASW ( B ) =
I rms 1.0021
2
=
= 0.003781 [cm ]
J
265
Step 31. Select a wire size with the required area from Table
4. If the area is not within 10% of the required area, go to
the next smallest size.
Step 22. Calculate the new wire size, AW(B) :
© 2011 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 4/13/11
=
2I o
2(0.7)
=
= 2.153 [A]
(1 − Dmax ) 1 − 0.35
I srms = I spk
Step 21. Calculate the AC flux density in Tesla, BAC:
(0.4π ) N (
(V p Dmax )
Step 29. Calculate the secondary rms current, Isrms:
0.0489 × 1 × 10 5
= 73.6 [T]
(0.4π )(0.58)(1.238)
Nnew=73.6; use 74.
B ac =
N p (Vo + Vd )(1 − Dmax )
Step 28. Calculate the secondary peak current, Ispk:
Step 20. Calculate the new turns, Nnew:
lg L
74(24 + 1)(1 − 0.35)
= 27.05
( 2 × 90)(0.35)
Naux=17.31; use 17.
where G is window height of selected core.
N=
=
Ns=27.05; use 27.
where µi is permeability of selected core material and
MPL is Magnetic Path Length of selected core.
lg
0.002315
= 0.8938
0.00259
N p (Vo + Vd )(1 − Dmax )
N=83.153; use 83[T].
F = (1 +
=
Step 27. Calculate the secondary and auxiliary turns, Ns
Naux:
3.74
)(10 8 )
[T]
2500
= 83.153
0.4π (0.58)
1 × 10 −3 (0.0489 +
)
Aw( B )
Wire A
This means that the selected wire from the Step 25, AWG23,
is enough or has enough margins for supplying the primaryside current on the flyback converter.
Step 18. Calculate the new turns using a gap from Step 15.
L(l g +
6.62
6.62
=
= 0.02960 [cm]
f
50 ×103
Step 24.Calculate the required wire area under considering
skin depth :
Step 14. Calculate the current density, J.:
J =
2
Wa K u 0.4283 × 0.4
=
= 0.002315 [A/cm ]
N new
74
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AN-9737
APPLICATION NOTE
AWG=#22
C. Sensing Resistor
AW(B) =0.003243[cm2]
The CS pin of FL6961 has over-current protection (OCP)
over the whole operating period and its internal clamping
level, VLIMIT, is 0.8V.
µΩ/cm=531.4
Step 32. Calculate the required number of primary strands,
Snp:
S np =
Asw( B )
Wire A
=
0.003243
= 1.2521
0.00259
This requires the AWG21 wire with two strands for
secondary-side winding on the flyback converter.
Adapted Core Size
Turns
PQ-42614
AWG
Primary
74
23
Secondary
27
22/ 2 Strands
Auxiliary
17
Estimated gap[mm]
0.489
Figure 5. Switching Current Limit
B. MOSFET and Diode Selection
Normally, it is reasonable to set the OCP level to 1.5 times
higher than the peak current at primary side.
Step 33. Calculate the maximum voltage of MOSFET drain
voltage at primary side:
VMOSFET(off ) = VIN + VR + VOS = VIN +
NP
VO + VOS = 490.54 [V]
NS
I LIMIT = 1.5I PPK =
where VOS is assumed ~50V and its peak can degrade
external snubber circuit performance. This means a 600V
MOSFET can be used with some margin. Minimum
requirements of the MOSFET are summarized below.
Current Rating [A]
+20% Margin
Calculation
+20% Margin
0.96
1.152
490.54
588.65
Rsen sin g ≤
0 .8
I LIMIT
= 0.55 [Ω ]
D. Voltage and Current Feedback for CC/CV
Function
The constant voltage and current output is adapted by
measuring the actual output voltage and current with
external passive components and an op amp in the
evaluation board. Because the output loads, the High
Bright LED (HB LED) and passive components are
effected by ambient temperature. Use the feedback path
for stable operation.
Step 34. Calculate the maximum voltage of diode at
secondary side:
VPK ( DIODE ) = VO + VIN
= 1.44
Calculate the sensing resistor as:
Voltage Rating [V]
Calculation
3TP
ηV p t on (max)
NS
27
= 24 + 265 2
= 160.74 [V]
NP
74
This means a 200V diode can be used with some margin.
The minimum requirement of the secondary diode as
summarized below.
Current rating [A]
Voltage rating [V]
Calculation
+20% Margin
Calculation
+20% Margin
2.153
2.584
160.74
192.88
Figure 6. Feedback Circuit for CC/CV Operation
© 2011 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 4/13/11
www.fairchildsemi.com
5
AN-9737
APPLICATION NOTE
E. Soft-Start / Overshoot Prevention Function
Normally, the CC block is dominate over the CV block in
steady state and the CV block acts as the Over-Voltage
Protection (OVP) at transient or abnormal mode, such as noload condition.
Normally, the High Bright (HB) LED has a forward-current
limitation to prevent the LED burn-out due to over-power
dissipation. Thererfore, the output overshoot function is
needed through the whole operating period. Though there
are CC/CV blocks for output regulation, those blocks do not
operate in transient modes, because they block have a long
response time and cannot act instantly. Figure 7 shows the
output voltage overshoot compression method using diode
and resistor. The current flows through resistor, R9, and
diode, D204, at startup, which is the period before activating
the CC/CV block, and then decrease at steady state. The
quantity of by-passing current goes into the feedback block
on the control IC, FL6961, and controls the output power
gradually.
The output signal of CC block is determined as:
VO _ cc = R4 (
Vsen sin g _ CC
R2
−
Vref
R3
)+
1 Vsen sin g _ CC Vref
(
−
)dt
C1 ∫
R2
R3
where the Vsensing_CC means the sensing voltage from the
sensing resistor (R1) and its values is as:
Vsen sin g _ CC = I o × R1
The output signal of CV block is determined as:
VO _ CV = (
R6
R
R6
)Vsen sin g _ CV + 8 [(
)
R5 + R6
R7 R5 + R6
Vsen sin g _ CV − Vref ] +
1 1
R6
(
Vsen sin g _ CV − Vref )dt
∫
C2 R7 R5 + R6
where the Vsensing_CV means the output voltage on this
circuit and this voltage is divided by two resistors, R5 and
R6, and connected to non-inverted pin at the op amp.
Normally, set this divided voltage, (
R6
)Vsen sin g _ CV , to
R5 + R6
Vref
or a little bit smaller value in steady state condition
because the main role of this block is over-voltage
protection. There are more high-voltage transfers to the
output stage at transient or an abnormal case such as overvoltage output condition than in the steady state.
Figure 7. Soft-Start / Overshoot Prevention Method
© 2011 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 4/13/11
www.fairchildsemi.com
6
AN-9737
APPLICATION NOTE
Table 2.
Various Core Types and Size
Part #
MLT
[cm]
MPL
[cm]
G[cm]
AC [cm]
Wa
2
[cm ]
Ap
4
[cm ]
Kg
5
[cm ]
Perm
AL
Manufacturer
RM-42316
4.17
3.80
1.074
0.640
0.454
0.2900
0.017820
2500
2200
Magnetics
PQ-42610
5.54
2.94
0.239
1.05
0.1177
0.1235
0.00937
2500
6310
Magnetics
PQ-42614
5.54
3.33
0.671
0.709
0.3304
0.2343
0.01200
2500
4585
Magnetics
PQ-42016
4.34
3.74
1.001
0.580
0.4283
0.2484
0.01327
2500
2930
Magnetics
EPC-25
4.930
5.92
1.800
0.4640
0.8235
0.3810
0.01438
2300
1560
Magnetics
EI-44008
7.77
5.19
0.356
0.9950
0.3613
0.3595
0.018416
2500
4103
Magnetics
EFD-25
4.78
5.69
1.86
0.5810
0.6789
0.3944
0.01917
1800
1800
Philips
Table 3.
PQ-42016 Core Dimensions
(Magnetics: http://www.mag-inc.com/home/Advanced-Search-Results?pn=42016
Table 4.
AWG
Wire Table
Bare Wire Area
Cm2
CIR-MIL
20
0.005188
1024.0
21
0.004116
812.30
22
0.003243
23
24
µΩ/cm
Heavy Insulation
Cm2
Turns/cm
Turns/cm2
332.3
0.006065
11.37
98.93
418.9
0.004837
12.75
124.0
640.10
531.4
0.003857
14.25
155.5
0.002588
510.80
666.0
0.003135
15.82
191.3
0.002047
404.0
842.1
0.002514
17.63
238.6
25
0.001623
320.40
1062.0
0.002002
19.8
299.7
26
0.001280
252.80
1345.0
0.001603
22.12
374.2
27
0.001021
201.60
1687.6
0.001313
24.44
456.9
28
0.008048
158.80
2142.7
0.0010515
27.32
570.6
29
0.0006470
127.70
2664.3
0.0008548
30.27
701.9
© 2011 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 4/13/11
www.fairchildsemi.com
7
AN-9737
APPLICATION NOTE
Schematic
FL6961
Figure 8. Schematic
© 2011 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 4/13/11
www.fairchildsemi.com
8
AN-9737
APPLICATION NOTE
Bill Of Materials
Item
Number
Part
Reference
Value
Quantity
Description (Manufacturer)
1
U101
FL6961
1
CRM PFC Controller (Fairchild Semiconductor)
2
U102
FOD817
1
Opto-Coupler (Fairchild Semiconductor)
3
U201
KA431
1
Shunt Regulator (Fairchild Semiconductor)
4
U202
KA358A(LM2904)
1
Dual Op Amp (Fairchild Semiconductor)
5
Q101
FQPF3N80C
1
800V/3A MOSFET (Fairchild Semiconductor)
6
D101
DF04
1
1.5A SMD Bridge-Diode (Fairchild Semiconductor)
7
D102
RS1M
1
1000V/1A Ultra-Fast Recovery Diode (Fairchild Semiconductor)
8
D103
RS1G
1
400V/1A Fast Recovery Diode (Fairchild Semiconductor)
9
D201,D204
EGP30D
2
200V/3A Ultra-Fast Recovery Diode (Fairchild Semiconductor)
10
D202,D203,
D205,D206
LL4148
3
General-Purpose Diode (Fairchild Semiconductor)
11
R101,R102,
R103
82KΩ
3
SMD Resistor1206
12
R104
120kΩ
1
SMD Resistor1206
13
R105
10KΩ
1
SMD Resistor1206
14
R106
20KΩ
1
SMD Resistor1206
15
R107
9.1kΩ
1
SMD Resistor1206
16
R108
47Ω
1
SMD Resistor 1206
17
R109
10Ω
1
SMD Resistor 1206
18
R110
220KΩ
1
2W
19
R111
30KΩ
1
SMD Resistor 1206
20
R112,R113
1Ω
2
SMD Resistor 1206
21
R201,R202,
R203
1Ω
3
SMD Resistor 1206
22
R204
2.2Ω
1
SMD Resistor 0806
23
R205
4.3KΩ
1
SMD Resistor 0806
24
R206
1.5KΩ
1
SMD Resistor 0806
25
R207
30KΩ
1
SMD Resistor 0806
26
R208
51KΩ
1
SMD Resistor 0806
27
R209
33KΩ
1
SMD Resistor 0806
28
R210
3.9KΩ
1
SMD Resistor 0806
29
R211
120KΩ
1
SMD Resistor 0806
30
R212
47KΩ
1
SMD Resistor 0806
31
R213
4.7KΩ
1
SMD Resistor 0806
32
R214
47KΩ
1
SMD Resistor 0806
© 2011 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 4/13/11
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9
AN-9737
APPLICATION NOTE
Bill Of Materials (Continued)
Item Number
Part Reference
Value
Quantity
Description (Manufacturer)
33
C101
100nF/250V
1
X – Capacitor
34
C102
47nF/250V
1
X – Capacitor
35
C103
100nF/630V
1
Film Capacitor
36
C104
33µF/35V
1
Electrolytic Capacitor
37
C105
2.2nF/1kV
1
Y-Capacitor
38
C106
2.2µF
1
SMD Capacitor 0805
39
C107
30pF
1
SMD Capacitor 0805
40
C108
100nF
1
SMD Capacitor 0805
41
C201,C202
470µF/35V
2
Electrolytic capacitor
42
C203
1µF
1
SMD Capacitor 0805
43
C204
470nF
1
SMD Capacitor 0805
44
C205
10µF/35V
1
Electrolytic Capacitor
45
LF101,LF102
80mH
2
Line Filter
46
L101
27µH
1
Line Filter
47
L102
6.8µH
1
Line Filter
48
L201
5µH
1
Output Inductor
49
F101
1A/250V
1
Fuse
50
T1
PQ-42016
1
1mH
© 2011 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 4/13/11
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10
AN-9737
APPLICATION NOTE
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WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF FAIRCHILD SEMICONDUCTOR CORPORATION.
As used herein:
1.
Life support devices or systems are devices or systems which,
(a) are intended for surgical implant into the body, or (b)
support or sustain life, or (c) whose failure to perform when
properly used in accordance with instructions for use provided
in the labeling, can be reasonably expected to result in
significant injury to the user.
© 2011 Fairchild Semiconductor Corporation
Rev. 1.0.0 • 4/13/11
2.
A critical component is any component of a life support device
or system whose failure to perform can be reasonably
expected to cause the failure of the life support device or
system, or to affect its safety or effectiveness.
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