MICREL MIC2589-1BM

MIC2589/MIC2595
Single-Channel, Negative High-Voltage Hot
Swap Power Controller/Sequencer
General Description
Features
The MIC2589 and MIC2595 are single-channel,
negative voltage hot swap controllers designed to
address the need for safe insertion and removal of
circuit boards into “live” system backplanes, while using
few external components. The MIC2589/MIC2589R and
the MIC2595/MIC2595R are each available in 14-pin
SOIC packaging and work in conjunction with an
external N-Channel MOSFET for which the gate drive is
controlled to provide inrush current limiting and output
voltage slew-rate control. Overcurrent fault protection is
also provided via a programmable overcurrent threshold
and filter. Very fast fault response is provided to ensure
that system power supplies maintain regulation even
during output short circuits. These controllers offer two
responses to a circuit breaker fault condition: the
MIC2589 and MIC2595 latch the circuit breaker’s output
off when the overcurrent threshold interval is exceeded
and the overcurrent filter times out while the MIC2589R
and MIC2595R automatically attempt to restart at a fixed
duty cycle after a current limit fault. A primary PowerGood signal and two secondary (delayed and
staggered) Power-Good signals are provided to indicate
that the output voltage is within its valid operating range.
These signals can be used to perform an all-at-once or
a sequenced enabling of one or more DC-DC power
modules.
All support documentation can be found on Micrel’s web
site at www.micrel.com.
• Provides safe insertion and removal from live –48V
(nominal) backplanes
• Operates from –19V to –80V
• Fast responding circuit breaker (<1µs) to short circuit
conditions
• User-programmable overcurrent detector response
time
• Electronic circuit breaker function:
Output latch OFF (MIC2589/MIC2595)
Output auto-retry (MIC2589R/MIC2595R)
• Active current regulation to control inrush currents
• Programmable undervoltage and overvoltage
lockouts (MIC2589/MIC2589R)
• Programmable UVLO hysteresis
(MIC2595/MIC2595R)
• Staggered ‘Power-Good’ output signals provide load
sequencing
Active-HIGH (-1)
Active-LOW (-2)
Applications
•
•
•
•
Central office switching
–48V power distribution
Distributed power systems
AdvancedTCA
_________________________________________________________________________________________________________
Ordering Information
Part Number
Standard
Pb-Free
PWRGD
Polarity
Lockout Functions
Circuit Breaker
Function
Package
MIC2589-1BM
MIC2589-1YM
Active-High
Programmable UVLO & OVLO
Latched Off
14-Pin SOIC
MIC2589-2BM
MIC2589-2YM
Active-Low
Programmable UVLO & OVLO
Latched Off
14-Pin SOIC
MIC2589R-1BM
MIC2589R-1YM
Active-High
Programmable UVLO & OVLO
Auto Retry
14-Pin SOIC
MIC2589R-2BM
MIC2589R-2YM
Active-Low
Programmable UVLO & OVLO
Auto Retry
14-Pin SOIC
MIC2595-1BM
MIC2595-1YM
Active-High
Programmable UVLO Hysteresis
Latched Off
14-Pin SOIC
MIC2595-2BM
MIC2595-2YM
Active-Low
Programmable UVLO Hysteresis
Latched Off
14-Pin SOIC
MIC2595R-1BM
MIC2595R-1YM
Active-High
Programmable UVLO Hysteresis
Auto Retry
14-Pin SOIC
MIC2595R-2BM
MIC2595R-2YM
Active-Low
Programmable UVLO Hysteresis
Auto Retry
14-Pin SOIC
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
December 2005
M9999-120505
(408) 955-1690
Micrel
MIC2589/MIC2595
Typical Applications
-48V RTN
(Long Pin)
-48V RTN
*D1
SMAT70A
1
R1
689k
1%
-48V RTN
(Short Pin)
R2
11.8k
1%
CTIMER
0.068µF
CFILTER
2.2µF
C1
0.47µF
R3
12.4k
1%
CNLD
0.068µF
MIC2589-2BM
VDD
/PWRGD1
14
2
PGTIMER
/PWRGD3
13
3
UV
/PWRGD2
12
4
OV
DRAIN
11
5
CFILTER
GATE
10
6
CNLD
7
VEE
SENSE
9
N/C
8
C4
100µF
IN+
C3
0.1µF
C6
100µF
C5
0.1µF
*C2
0.1µF
R4
10
C8
100µF
-48V RTN
(Long Pin)
RSENSE
0.01
5%
OUT+
+5VOUT
ON/OFF*
C7
0.1µF
M1
SUM110N10-09
IN–
OUT–
+5V RTN
IN+
OUT+
+2.5VOUT
ON/OFF*
IN–
OUT–
+2.5V RTN
IN+
OUT+
+1.8VOUT
ON/OFF*
IN–
OUT–
+1.8V RTN
Nominal Undervoltage and Overvoltage Thresholds:
VUV=36.5V
VOV=71.2V
Overcurrent TImer Delay
=30ms
tFLT~
*Optional components (See Funtional Description and Applications Information for more details)
#An external pull-up resistor for the power-good signal is necessary for DC-DC convertors (and all other load modules) not equipped with an
internal pull-up impedance
December 2005
2
M9999-120505
(408) 955-1690
Micrel
MIC2589/MIC2595
Pin Configuration
December 2005
14-Pin SOIC
MIC2589-1BM
MIC2589R-1BM
MIC2589-1YM
MIC2589R-1YM
14-Pin SOIC
MIC2589-2BM
MIC2589R-2BM
MIC2589-2YM
MIC2589R-2YM
14-Pin SOIC
MIC2595-1BM
MIC2595R-1BM
MIC2595-1YM
MIC2595R-1YM
14-Pin SOIC
MIC2595-2BM
MIC2595R-2BM
MIC2595-2YM
MIC2595R-2YM
3
M9999-120505
(408) 955-1690
Micrel
MIC2589/MIC2595
Pin Description
Pin Number
Pin Name
1
PWRGD1
(MIC25XX-1)
Active High
/PWRGD1
(MIC25XX-2)
Active Low
Pin Function
Power-Good Output 1: Asserted when the voltage on the DRAIN pin (VDRAIN) is within
VPGTH of VEE, indicating that the output voltage is within proper specifications. For
the MIC2589-1 and MIC2595-1, PWRGD1 will be high impedance when VDRAIN is
less than VPGTH, and will pull-down to VDRAIN when VDRAIN is greater than VPGTH. For
the MIC2589-2 and MIC2595-2, /PWRGD1 will pull-down to VDRAIN when VDRAIN is
less than VPGTH, and will be high-impedance when VDRAIN is greater than VPGTH.
2
PGTIMER
A capacitor connected from this pin to VEE sets the time interval between assertions
of PWRGD2 (or /PWRGD2) and PWRGD3 (or /PWRGD3) relative to PWRGD1 (or
/PWRGD1). See the “Functional Description” for further detail.
3
UV
Undervoltage Threshold Input: When the voltage at the UV pin is less than the VUVL
threshold, the GATE pin is immediately pulled low by an internal 100µA current pulldown. The UV pin is also used to cycle the device off and on to reset the circuit
breaker. Taken together, the OV and UV pins form a window comparator that defines
the limits of VEE to deliver power to the load.
(MIC2589
and
MIC2589R)
3
4
OFF
(MIC2595
and
MIC2595R)
Turn-Off Threshold: When the voltage at the OFF pin is less than the VOFFL
threshold, the GATE pin is immediately pulled low by an internal 100µA current pulldown. The OFF pin is also used to cycle the device off and on to reset the circuit
breaker. Taken together, the ON and OFF pins provide programmable hysteresis for
the MIC2595 to be enabled.
OV
Overvoltage Threshold Input: When the voltage at the OV pin is greater than the
VOVH threshold, the GATE pin is immediately pulled low by an internal 100µA current
pull-down.
(MIC2589
and
MIC2589R)
4
ON
(MIC2595
and
MIC2595R)
Turn-On Threshold: At initial system power-up or after the part has been shut off by
the OFF pin, the voltage on the ON pin must be above the VONH threshold in order for
the MIC2595 to be enabled.
5
CFILTER
Current Limit Response Timer: A capacitor connected between this pin and VEE
provides filtering against nuisance tripping of the circuit breaker by setting a time
delay, tFLT, for which an overcurrent event must last prior to signaling a fault condition
and latching the output off. The minimum time for tFLT will be the time it takes for the
output (capacitance) to charge to VEE during start-up. This pin is held to VEE with a
3µA current pull-down when no current limit condition exists. See the “Functional
Description” for further details.
6
CNLD
No-Load Detect Timer: The absence of a load for the MIC2589/MIC2589R is defined
for any current load that is less than 20% of the full-scale current limit (i.e., 0.20 ×
ILIM). A capacitor between CNLD and VEE sets the filter delay, tNLD, for a load current
that is 80% (or greater) below the full-scale current limit before the circuit breaker is
tripped.
7
VEE
Negative Supply Voltage Input: Connect the negative, or low side, terminal of the
input power supply.
8
NC
No Internal Connection
9
SENSE
10
GATE
Gate Drive Output: Connects to the Gate of an N-Channel MOSFET.
11
DRAIN
Drain Sense Input: Connects to the Drain of an N-Channel MOSFET.
December 2005
Circuit Breaker Sense Input: A resistor between this pin and VEE sets the current
limit trip point for the circuit. When the current limit threshold of IR = 50mV is
exceeded for tFLT, the circuit breaker is tripped and the GATE pin is immediately
pulled low by IGATEOFF. Toggling UV or OV will reset the circuit breaker. In order to
disable the circuit breaker (i.e., eliminate overcurrent VSENSE-VEE protection), connect
(short) the SENSE pin to VEE and also connect the CNLD pin to VEE to disable the
no-load detection feature.
4
M9999-120505
(408) 955-1690
Micrel
MIC2589/MIC2595
Pin Desription (cont.)
Pin Number
Pin Name
12
PWRGD2
(MIC25XX-1)
Power-Good Output 2: Asserted when the following is true: (PWRGD1 = Asserted)
AND (Time after Assertion of PWRGD1 = Time PWRGD2, as programmed by the
capacitor on PGTIMER). Once PWRGD1 is asserted, the PGTIMER pin begins to
charge and PWRGD2 will assert when PGTIMER crosses the PWRGD2 threshold
(VTHRESH(PG2) = 0.63V, typical). Also see PWRGD1 and PGTIMER pin descriptions
12
/PWRGD2
(MIC25XX-2)
/Power-Good Output 2: Asserted when the following is true: (/PWRGD1 = Asserted)
AND (Time after Assertion of /PWRGD1 = Time /PWRGD2, as programmed by the
capacitor on PGTIMER). Once /PWRGD1 is asserted, the PGTIMER pin begins to
charge and /PWRGD2 will assert when PGTIMER crosses the /PWRGD2 threshold
(VTHRESH(PG2) = 0.63V, typical). Also see /PWRGD1 and PGTIMER pin descriptions.
13
PWRGD3
(MIC25XX-1)
Power-Good Output 3: Asserted when the following is true: (PWRGD1 = Asserted)
AND (Time after Assertion of PWRGD1 = Time PWRGD3, as programmed by the
capacitor on PGTIMER). Once PWRGD1 is asserted, the PGTIMER pin begins to
charge and PWRGD3 will assert when PGTIMER crosses the PWRGD3 threshold
(VTHRESH(PG3) = 1.15V, typical). Also see PWRGD1 and PGTIMER pin descriptions.
13
/PWRGD3
(MIC25XX-2)
/Power-Good Output 3: Open Collector. Asserted when the following is true:
(/PWRGD1 = Asserted) AND (Time after Assertion of /PWRGD1 = Time /PWRGD3,
as programmed by the capacitor on PGTIMER). Once /PWRGD1 is asserted, the
PGTIMER pin begins to charge and /PWRGD3 will assert when PGTIMER crosses
the /PWRGD3 threshold (VTHRESH(PG3) = 1.15V, typical). Also see /PWRGD1 and
PGTIMER pin descriptions.
14
VDD
December 2005
Pin Function
Positive Supply Input: Connect to the positive, or high side, terminal of the input
power supply.
5
M9999-120505
(408) 955-1690
Micrel
MIC2589/MIC2595
Absolute Maximum Ratings(1)
Operating Ratings(2)
(All voltages are referred to VEE)
Supply Voltage (VDD – VEE) ........................... –0.3V to 100V
DRAIN, PWRGD pins.................................... –0.3V to 100V
GATE pin...................................................... –0.3V to 12.5V
SENSE, OV, UV, ON, OFF pins........................ –0.3V to 6V
Lead Temperature (soldering)
Standard Package (-xBM)
(IR Reflow, Peak Temperature ...........240°C +0°C/-5°C
Pb-Free Package (-xYM)
(IR Reflow, Peak Temperature ...........260°C +0°C/-5°C
ESD Ratings(3)
Human Body Model.................................................. 2kV
Machine Model ......................................................100V
Supply Voltage (VDD-VEE) ...............................+19V to +80V
Ambient Temperature Range (TA) .................–40°C to 85°C
Junction Temperature (TJ)..........................................125°C
Package Thermal Resistance
SOIC (θJA) ....................................................... 120°C/W
DC Electrical Characteristics(4)
VDD = 48V, VEE = 0V, TA = 25°C, unless otherwise noted. Bold indicates specifications apply over the full operating temperature
range of –40°C to 85°C.
Symbol
Parameter
VDD – VEE
Supply Voltage
IDD
Supply Current
VTRIP
Circuit Breaker Trip Voltage
INLDTH
No-Load Detect Threshold
(% of full-scale current limit)
Condition
Min
Typ
Max
Units
80
V
4
6
mA
50
60
mV
19
GATE Drive Voltage, (VGATE – VEE)
40
IOUT decreasing
IOUT increasing
20
22
%
%
2
%
INLDHYS
No-Load Detect Threshold
Hysteresis
VCNLD
No-Load Detect Timer High
Threshold Voltage
1.17
1.24
1.33
V
ICNLD
No-Load Detect Timer
(5)
Capacitor Charge Current
10
25
40
µA
VGATE
GATE Drive Voltage,
(VGATE – VEE)
15V ≤ (VDD – VEE) ≤ 80V
9
10
11
V
IGATEON
GATE Pin Pull-Up Current
VGATE = VEE to 8V
19V ≤ (VDD – VEE) ≤ 80V
30
45
60
µA
ISENSE
SENSE Pin Current
VSENSE = 50mV
IGATEOFF
GATE Pin Sink Current
(VSENSE – VEE) = 100mV
VGATE = 2V
ICFILTER
CFILTER Pin Charge Current
0.2
µA
100
240
mA
(VSENSE – VEE) > VTRIP
VCFILTER = 0.75V
VGATE = 3V
65
95
135
µA
CFILTER Discharge Current
(VSENSE – VEE) < VTRIP
VCFILTER = 0.75V
VGATE = 3V
2
4
6
µA
VCFILTER(TRIP)
High Threshold Voltage
Overcurrent Detect Timer
(VSENSE – VEE) > VTRIP
1.17
1.25
1.33
V
VCFILTER(RETRY)
Voltage on CFILTER
(decreasing) to Trigger AutoRetry
0.17
0.22
0.25
V
30
45
80
µA
(MIC2589R and MIC2595R)
IPGTIMER
December 2005
PGTIMER Charge Current
Voltage on PGTIMER = 0.75 V
6
M9999-120505
(408) 955-1690
Micrel
MIC2589/MIC2595
Notes:
1. Exceeding the absolute maximum rating may damage the device.
2. The device is not guaranteed to function outside its operating rating.
3. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5k in series with 100pF.
4. Specification for packaged product only.
5. Not 100% tested. Parameters are guaranteed by design.
DC Electrical Characteristics(6)
VDD = 48V, VEE = 0V, TA = 25°C, unless otherwise noted. Bold indicates specifications apply over the full operating temperature
range of –40°C to 85°C.
Symbol
Parameter
VTHRESH(PG2)
Min
Typ
Max
Units
PGTIMER Threshold Voltage for
PWRGD2 and /PWRGD2
0.5
0.63
0.8
V
VTHRESH(PG3)
PGTIMER Threshold Voltage for
PWRGD3 and /PWRGD3
1.00
1.15
1.30
V
RPGTIMER
PGTIMER Discharge Resistance
Voltage on PGTIMER = 0.5 V
250
500
750
Ω
VOVH
OV Pin High Threshold Voltage
(MIC2589 and MIC2589R)
Low-to-High transition
1.198
1.223
1.247
V
VOVL
OV Pin Low threshold Voltage
(MIC2589 and MIC2589R)
High-to-Low transition
1.165
1.203
1.232
V
VOVHYS
OV Pin Hysteresis
(MIC2589 and MIC2589R)
VUVL
UV Pin Low threshold Voltage
(MIC2589 and MIC2589R)
High-to-Low transition
1.198
1.223
1.247
V
VUVH
UV Pin High Threshold Voltage
(MIC2589 and MIC2589R)
Low-to-High transition
1.213
1.243
1.272
V
VUVHYS
UV Pin Hysteresis
(MIC2589 and MIC2589R)
VONH
ON Pin High Threshold Voltage
(MIC2595 and MIC2595R)
Low-to-High transition
1.198
1.223
1.247
V
VOFFL
OFF Pin Low Threshold Voltage
(MIC2595 and MIC2595R)
High-to-Low transition
1.198
1.223
1.247
V
ICNTRL
Input Current (OV, UV, ON, OFF
Pins)
VINPUT = 1.25V
0.5
µA
VPGTH
Power-Good Threshold
(VDRAIN – VEE)
High-to-Low Transition
1.40
V
VOLPG
PWRGD Output Voltage
(relative to voltage at the DRAIN
pin)
VOLPG – VDRAIN
0 ≤ IPG ≤ 1mA
PWRGD Output Leakage
Current
VPWRGD = VDD = 80 V
ILKG(PG)
Condition
20
mV
20
1.1
1.26
mV
MIC25XX-1
(VDRAIN – VEE) > VPGTH
-0.25
0.8
V
MIC25XX-2
(VDRAIN – VEE) < VPGTH
-0.25
0.8
V
1
µA
Note:
6. Specification for packaged product only.
December 2005
7
M9999-120505
(408) 955-1690
Micrel
MIC2589/MIC2595
AC Electrical Characteristics(7)
Symbol
Parameter
Condition
Min
Typ
Max
Units
tOCSENSE
Overcurrent Sense to GATE
Low Trip Time(8)
Figure 2
VSENSE – VEE = 100mV
3.5
µs
tOVPHL
OV High to GATE Low(8)
(MIC2589 and MIC2589R)
Figure 3
OV = 1.5V
1
µs
tOVPLH
OV Low to GATE High(8),
(MIC2589 and MIC2589R)
Figure 3
OV = 1.0V
1
µs
tUVPHL
UV Low to GATE Low(8),
(MIC2589 and MIC2589R)
Figure 4
UV = 1.0V
1
µs
tUVPLH
UV High to GATE High(8)
(MIC2589 and MIC2589R)
Figure 4
UV = 1.5V
1
µs
tOFFPHL
OFF Low to GATE Low(8),
Figure 5 (MIC2595 and
MIC2595R)
OFF = 1.0V
1
µs
tONPLH
ON High to GATE High(8)
(MIC2595 and MIC2595R)
Figure 5
ON = 1.5V
1
µs
tPGLH1
DRAIN Low to PWRGD1 Output
High(8)
CLOAD on PWRGDx = 50pF
RPULLUP = 100kΩ
3
µs
CLOAD on PWRGDx = 50pF
RPULLUP = 100kΩ
5
µs
CLOAD on /PWRGDx = 50pF
RPULLUP = 100kΩ
5
µs
CLOAD on /PWRGDx = 50pF
RPULLUP = 100kΩ
3
µs
(MIC25XX-1XX)
tPGHL1
DRAIN High to all PWRGDx
(8)
Outputs Low
(MIC25XX-1XX)
tPGHL2
DRAIN Low to /PWRGD1
Output Low(8)
(MIC25XX-2)
tPGLH2
DRAIN High to all /PWRGDx
Outputs High(8)
(MIC25XX-2)
Note:
7. Specification for packaged product only.
8. Not 100% production tested. Parameters are guaranteed by design.
December 2005
8
M9999-120505
(408) 955-1690
Micrel
MIC2589/MIC2595
Timing Diagrams
OVERCURRENT
EVENT
t < t FLT
t ≥ tFLT
ILIMIT
tNLD
ILOAD
INLDTH
0A
Output OFF
(at V DD)
Load current is regulated
at I LIMIT = 50mV/R SENSE
V DRAIN
(at V EE)
(at V EE)
(at V EE)
VUV (MIC2589)
VOFF , VON (MIC2595)
Reduction in V DRAIN to support
ILIMIT = 50mV/R SENSE
(V UV
VUVL
V EE)
VUVH
(V UV VEE)
(at V EE)
Figure 1. Overcurrent and Undercurrent (No Load) Response
Figure 2. SENSE to GATE Timing Response
Figure 3. MIC2589/MIC2595 Overvoltage Response
Figure 4. MIC2589/MIC2589R Undervoltage Response
December 2005
9
M9999-120505
(408) 955-1690
Micrel
MIC2589/MIC2595
Figure 5a. MIC2595/MIC2595R OFF to GATE Drive Response
1.223V
VON
tONLH
VGATE
1V
Figure 5b. MIC2595/MIC2595R ON to GATE Drive Response
Figure 6. DRAIN to Power-Good Response
December 2005
10
M9999-120505
(408) 955-1690
Micrel
MIC2589/MIC2595
Typical Characteristics
6
4
3
2
1
0
-40 -20 0 20 40 60 80 100
TEMPERATURE (°C)
10
4
8
3
2
25 35 45 55 65 75
SUPPLY VOLTAGE (V)
0
-40 -20 0 20 40 60 80 100
TEMPERATURE (°C)
85
GATE Pull-Up Current
vs. Temperature
40
250
4
20
200
150
100
85
UVH
1.25
1.24
VUVL
1.23
1.22
1.21
1.28
1.2
-40 -20 0 20 40 60 80 100
TEMPERATURE (°C)
1.25
1.24
VOVH
1.23
1.22
V
OVL
1.21
100
90
80
PG2TH
PG3TH
0
-40 -20 0 20 40 60 80 100
TEMPERATURE (°C)
December 2005
1.26
PGTimer Charge Current
vs. Temperature
IPGTIMER (µA)
0.8
0.6
0.4
0.2
1.27
1.2
-40 -20 0 20 40 60 80 100
TEMPERATURE (°C)
PGTimer Thresholds
vs. Temperature
1.6
1.4
1.2
1
Power-Good Threshold
vs. Temperature
70
60
50
I
PGTIMER
40
30
20
-40 -20 0 20 40 60 80 100
TEMPERATURE (°C)
11
POWER-GOOD THRESHOLD (V)
V
0
-40 -20 0 20 40 60 80 100
TEMPERATURE (°C)
OV Pin Threshold
vs. Temperature
UV Pin Threshold
vs. Temperature
1.26
50
0
-40 -20 0 20 40 60 80 100
TEMPERATURE (°C)
PGTIMER DISCHARGE CURRENT (mA)
25 35 45 55 65 75
SUPPLY VOLTAGE (V)
1.27
2
1.8
30
10
OV PIN THRESHOLD (V)
1.28
IGATEOFF (mA)
8
6
GATE Sink Current
vs. Temperature
350
300
0
15
UV PIN THRESHOLD (V)
4
50
2
PGTIMER THRESHOLDS (V)
6
10
IGATEON (µA)
VGATE (V)
60
vs. Temperature
2
1
GATE Drive (V GATE - VEE)
12
12
5
0
15
vs. Supply Voltage
GATE Drive (V GATE - VEE)
Supply Current
vs. Supply Voltage
VGATE (V)
5
SUPPLY CURRENT (mA)
SUPPLY CURRENT (mA)
6
Supply Current
vs. Temperature
2
1.8
1.6
1.4
1.2
PGTH+
1
0.8
0.6
PGTH-
0.4
0.2
0
-40 -20 0 20 40 60 80 100
TEMPERATURE (°C)
PGTimer Discharge Current
vs. Temperature
4
3.5
3
2.5
2
1.5
I
PGTIMEROFF
1
0.5
0
-40 -20 0 20 40 60 80 100
TEMPERATURE (°C)
M9999-120505
(408) 955-1690
Micrel
MIC2589/MIC2595
Typical Characteristics (cont.)
P
1000
0.5
60
0.45
58
800
0.4
56
0.35
0.3
54
52
R
VOLPG (V)
PGTIMER
500
400
0.25
0.2
300
0.15
200
0.1
40
40
35
35
I
NLDTH
20
30
25
I
CNLD
20
15
15
10
10
5
5
0
-40 -20 0 20 40 60 80 100
TEMPERATURE (°C)
0
-40 -20 0 20 40 60 80 100
TEMPERATURE (°C)
1.45
1.4
1.35
1.3
1.25
VCFILTER(trip)
1.2
1.15
1.1
46
1.4
1.2
1
0.8
0.6
0.4
0.2
0
-40 -20 0 20 40 60 80 100
TEMPERATURE (°C)
CFILTER (Auto-Retry) Threshold
vs. Temperature
0.5
VCFILTER(RETRY) (V)
1.5
CFILTER Threshold
vs. Temperature
48
No-Load Detect Timer Discharging
Current vs. Temperature
1.05
0.45
0.4
0.35
0.3
0.25
VCFILTER(retry)
0.2
0.15
0.1
0.05
1
-40 -20 0 20 40 60 80 100
TEMPERATURE (°C)
0
-40 -20 0 20 40 60 80 100
TEMPERATURE (°C)
CFILTER Discharge Current
vs. Temperature
CFILTER Charge Current
vs. Temperature
CFILTER CHARGE CURRENT (µA)
30
25
ICNLD (µA)
45
50
42
40
-40 -20 0 20 40 60 80 100
TEMPERATURE (°C)
50
45
ON Pin Threshold
vs. Temperature
140
120
100
80
60
40
20
0
-40 -20 0 20 40 60 80 100
TEMPERATURE (°C)
1.5
9
8
1.45
1.4
1.45
1.4
7
1.35
1.35
6
1.3
4
3
VONH (V)
10
5
VONH
1.25
1.2
1.15
1.5
1.3
1.25
1.1
1.1
1
1.05
1.05
December 2005
1
-40 -20 0 20 40 60 80 100
TEMPERATURE (°C)
12
OFF Pin Threshold
vs. Temperature
VOFFL
1.2
1.15
2
0
-40 -20 0 20 40 60 80 100
TEMPERATURE (°C)
VTRIP
44
No-Load Detect Timer Charging
Current vs. Temperature
50
INLDTH (%)
OLPG
0.05
0
-40 -20 0 20 40 60 80 100
TEMPERATURE (°C)
No-Load Detect Threshold
vs. Temperature
CFILTER DISCHARGE CURRENT (µA)
V
NO-LOAD DISCHARGE CURRENT (mA)
700
600
VTRIP (mV)
900
100
0
-40 -20 0 20 40 60 80 100
TEMPERATURE (°C)
VCFILTER(TRIP) (V)
Circuit Breaker Trip Voltage
vs. Temperature
Power-Good Low Voltage
vs. Temperature
VOFFL (V)
RPGTIMER (ohms)
Discharge Resistance
vs. Temperature
GTIMER
1
-40 -20 0 20 40 60 80 100
TEMPERATURE (°C)
M9999-120505
(408) 955-1690
Micrel
MIC2589/MIC2595
Test Circuit
-48V RTN
(Long Pin)
-48V RTN
R5
47k
*D1
SMAT70A
-48V RTN
(Short Pin)
1
R1
689k
1%
R2
11.8k
1%
CTIMER
CFILTER
2.2µF
C1
0.47µF
R3
12.4k
1%
*C2
0.22µF
CNLD
0.068µF
MIC2589-2BM
/PWRGD1
VDD
14
2
PGTIMER
/PWRGD3
13
3
UV
/PWRGD2
12
4
OV
DRAIN
11
5
CFILTER
GATE
10
6
CNLD
7
VEE
SENSE
9
N/C
8
R6
47k
R7
47k
CLOAD
C4
0.1µF
CGATE
R4
10
-48V RTN
(Long Pin)
RSENSE
0.01
5%
M1
SUM110N10-09
-48VOUT
Test Circuit
December 2005
13
M9999-120505
(408) 955-1690
Micrel
MIC2589/MIC2595
Functional Characteristics
December 2005
14
M9999-120505
(408) 955-1690
Micrel
MIC2589/MIC2595
Functional Diagram
Block Diagram
December 2005
15
M9999-120505
(408) 955-1690
Micrel
MIC2589/MIC2595
Functional Description
Hot Swap Insertion
When circuit boards are inserted into systems carrying
live supply voltages (“hot swapped”), high inrush
currents often result due to the charging of bulk
capacitance that resides across the circuit board’s
supply pins. These current spikes can cause the
system’s supply voltages to temporarily go out of
regulation causing data loss or system lock-up. In
more extreme cases, the transients occurring during a
hot swap event may cause permanent damage to
connectors or onboard components.
The MIC2589 and the MIC2595 are designed to
address these issues by limiting the maximum current
that is allowed to flow during hot swap events. This is
achieved by implementing a constant-current loop at
turn-on. In addition to inrush current control, the
MIC2589 and the MIC2595 incorporate input voltage
supervisory functions and user programmable
overcurrent protection, thereby providing robust
protection for both the system and the circuit board.
GATE Start-Up and Control
When the input voltage to the controller is between
the overvoltage and undervoltage threshold settings
(MIC2589) or is greater than the ON threshold setting
(MIC2595), a start cycle is initiated to deliver power to
the load. During the start-up cycle, the GATE pin of
the controller applies a constant charging current
(45µA, nominal) to the gate of the external MOSFET,
charging the MOSFET gate from 0V to 10V,
referenced to VEE. An external capacitor (C2) can be
used to adjust and control the slew rate of the GATE
output, while resistor R4 can be used to minimize the
potential for parasitic high-frequency oscillations
occurring on the gate of the external MOSFET (M1).
See Typical Application circuit.
The following
equation is used to approximate the expected inrush
current given the values of the capacitance at the gate
and the load (i.e., the gate of the external MOSFET
and the drain of the external MOSFET, respectively).
INRUSH =
CLOAD
× IGATE(ON)
C GATE
Active current limiting for the MIC2589/MIC2595 is
implemented by controlling the voltage on the GATE
pin via an internal feedback circuit.
The
MIC2589/MIC2595 is defined to be in current limit
when the GATE output voltage level is between 2.5V
and 5.5V. Once in current limit, the GATE output
voltage is regulated to limit the load current to the
programmed value (ILIMIT).
Additionally, the
December 2005
overcurrent delay and the no-load detection timers
must be set accordingly to allow the output load to
fully charge during the start-up cycle. See the “Circuit
Breaker Function” and “No-Load Detection” sections
for further details.
Resistor R4, in series with the power MOSFET’s gate,
may be required in some layouts to minimize the
potential for parasitic oscillations occurring in M1.
Note that resistance in this device of the circuit has a
slight
destabilizing
effect
upon
the
MIC2589/MIC2595’s current regulation loop. If
possible, use high-frequency PCB layout techniques
and use a dummy resistor (R4 = 0Ω) for the initial
evaluation. If during prototyping an R4 is required,
common values for R4 range between 4.7Ω to 20Ω for
various power MOSFETs.
Circuit Breaker Function
The MIC2589 and MIC2595 device family employs an
electronic circuit breaker that protects the external
power MOSFET and other system components
against large-scale faults, such as short circuits. The
current-limit threshold is set via an external resistor,
RSENSE, connected between the VEE and SENSE pins.
ILIMIT =
VTRIP
R SENSE
An overcurrent filter period is set via a capacitor from
the CFILTER pin to ground (CFILTER) that determines
the length of the time period (tFLT) for which the device
remains in current limit before the circuit breaker is
tripped. This programmable delay prevents tripping of
the circuit breaker due to the large inrush current
charging bulk and distributed capacitive loads.
Whenever the voltage across RSENSE exceeds 50mV,
two things happen:
1. A constant-current regulation loop is engaged
which is designed to hold the voltage across
RSENSE equal to 50mV. This protects both the
load and the MIC2589/MIC2595 circuits from
excessively high currents. This currentregulation loop will engage in less than 1µs
from the time at which the overcurrent trip
threshold on RSENSE is exceeded.
2. Capacitor CFILTER is charged up to an internal
VCFILTER(TRIP)
threshold
of
1.25V
by
ICFILTER(CHARGE) an internal 95µA current
source. If the voltage across CFILTER
crosses this threshold, the circuit breaker trips
and the GATE pin is immediately pulled low
by an internal current pull-down. This
operation turns off the MOSFET quickly and
disconnects the input from the load. The time
period that allows for the output to regulate in
16
M9999-120505
(408) 955-1690
Micrel
MIC2589/MIC2595
current limit is defined as the overcurrent fault
timer, tFLT, and is determined by the following
equation.
CFILTER × VCFILTER(trip)
tFLT =
ICFILTER(CHARGE)
The value of CFILTER should be selected to
allow the circuit’s minimum regulated value of
IOUT to equal ILIMIT for somewhat longer than
the time it takes to charge the total load
capacitance.
During startup, the CFILTER pin will begin to charge
once the GATE crosses 2.5V. In order to avoid falsetripping of the circuit breaker by allowing the
overcurrent filter to time out, the overcurrent delay
must be set to exceed the time it takes to ramp the
GATE output above 5.5V (i.e., charge the output load
capacitance).
An initial value for CFILTER is found by calculating the
time it will take for the MIC2589/MIC2595 to
completely charge up the output capacitive load.
Assuming the load is enabled by the PWRGDX (or
/PWRGDX) signal(s) of the controller, the turn-on
delay time is derived from I = C × (dv/dt):
C LOAD × (VDD − VEE )
ILIMIT
Using
parametric
values
specific
to
the
MIC2589/MIC2595, an expression relating CFILTER to
the circuit’s turn-on delay time is:
t TURN−ON =
CFILTER =
(t TURN−ON × ICFILTER )
VCFILTER
Substituting the variables above with the specification
limits of the MIC2589/MIC2595, an expression for the
worst-case value for CFILTER is given by:
⎛ 135µA ⎞
C FILTER(max ) = t TURN−ON × ⎜
⎟
⎝ 1.17V ⎠
µF ⎞
⎛
C FILTER(max ) = t TURN−ON × ⎜115 × 10 −6
⎟
sec ⎠
⎝
For example, in a system with a CLOAD = 1500µF, a
maximum (VDD – VEE) = 72V, and a maximum load
current on a nominal –48V buss of 2.5A, the nominal
circuit design equations steps are:
1. Choose ILIMIT = IHOT_SWAP(nom) = 3A (2.5A + 20%);
38.8mV
= 12.9mΩ (closest
3A
1% standard value is 13.0mΩ);
3. Using ICHARGE = ILIMIT = 3A, the application circuit
turn-on time is calculated:
2. Select an R SENSE =
December 2005
t TURN −ON =
(1500µF × 72V ) = 36ms (use 40ms)
3A
Allowing for capacitor tolerances and a nominal 40ms
turn-on time, an initial worst-case value for CFILTER is:
CFILTER(WORST-CASE) = 40ms×(115×10–6µF/sec) = 4.6µF
The closest standard ±10% tolerance capacitor value
is 4.7µF and would be a good initial starting value for
prototyping.
Whenever the hot swap controller is not in current
limit, CFILTER is discharged to VEE by an internal
4µA current source.
For the MIC2589R/MIC2595R devices, the circuit
breaker automatically resets after approximately 25
tFLT time constants (23.75 × tFLT_AUTO). If the fault
condition still exists, capacitor CFILTER will again
charge up to VCFILTER(TRIP), tripping the circuit breaker.
Capacitor CFILTER will then be discharged by an
internal 4µA current source until the voltage across
CFILTER goes below VCFILTER(RETRY), at which time
another start cycle is initiated. This will continue until
the fault condition is removed or input power is
removed/cycled. The duty cycle of the auto-restart
function is therefore fixed at 4.25% and the period of
the auto-restart cycle is given by:
t RETRY = t FLT + t FLT_AUTO
t RETRY = t FLT +
[CFILTER × (VCFILTER(TRIP) - VCFILTER(retry ) )]
ICFILTER(pu ll− down )
The auto-restart period for the example above where
the worst-case CFILTER was determined to be 4.7µF is:
tAUTO-RESTART = 1.27s
No-Load Detection
For applications in which a minimum load current will
always be present, the no-load detect capability of the
MIC2589 product family offers system designers the
ability to perform a shutdown operation on such fault
conditions, such as an unscheduled or unexpected
removal of PC boards from the system or on-board
fuse failure. As long as the minimum current drawn by
the load is at least 20% of the current limit (defined by
VTRIP
), the output of the hot swap controller will
R SENSE
remain enabled. If the output current falls below 20%
of the actual current limit, the controller’s no-load
detection loop is enabled. In this loop, an internal
current source, ICNLD, will charge an external capacitor
CNLD. An expression for the controller’s no-load timeout delay is given by:
17
M9999-120505
(408) 955-1690
Micrel
MIC2589/MIC2595
⎛C
t NLD = VCNLD × ⎜⎜ NLD
⎝ ICNLD
⎞
⎟
⎟
⎠
where VCNLD = 1.24V (typ); ICNLD = 25µA (typ); and
CNLD is an external capacitor connected from Pin 6 to
VEE. Once the voltage on CNLD reaches its no-load
threshold voltage, VCNLD, the loop times out and the
controller will shut down until it is reset manually
(MIC2589/MIC2595) or until it performs an auto-retry
operation (MIC2589R/MIC2595R). During start-up, the
no-load detection circuit begins to monitor the load
current and the CNLD pin starts ramping along with
the GATE output. In order to keep the output from
shutting down, tNLD must be long enough to ensure
that the output MOSFET switches on to deliver the
required minimum load-detect current to the output
load before the no-load timer times out.
The Power-Good Output Signals
For
the
MIC2589/MIC2595-1
and
MIC2589R/MIC2595R-1, power-good output signal
PWRGD1 will be high impedance when VDRAIN drops
below VPGTH, and will pull-down to the potential at the
DRAIN when VDRAIN is above VPGTH. For the
MIC2589/95-2 and the MIC2589R/95R-2, power-good
output signal /PWRGD1 will pull down to the potential
of the DRAIN pin when VDRAIN drops below VPGTH and
will be high impedance when VDRAIN is above VPGTH.
Hence, the -1 parts have an active-high PWRGDX
signal and the -2 parts have an active-low /PWRGDX
output. PWRGDX (or /PWRGDX) may be used as an
enable signal for one or more following DC/DC
converter modules or for other system uses as
desired. When used as an enable signal, the time
necessary for the PWRGD (or /PWRGD) signal to
pull-up (when in high impedance state) will depend
upon the load (RC) that is present on this output.
Power-good output signals PWRGD2 (/PWRGD2) and
PWRGD3 (/PWRGD3) follow the assertion of
PWRGD1 (/PWRGD1) with a sequencing delay set by
an external capacitor (CPG) from the controller’s
PGTIMER pin (Pin 2) to VEE. An expression for the
sequencing delay between PWRGD2 and PWRGD1
is given by:
t PGDLY2−1 =
VTHRESH(PG2) × C PG
(VTHRESH(PG3 ) − VTHRESH(PG2 ) ) × CPG
IPGTIMER
where VTHRESH(PG3) (1.15V, typical) is the PWRGD3
threshold voltage for PGTIMER. Therefore, powergood output signal PWRGD2 (/PWRGD2) will be
delayed after the assertion of PWRGD1 (/PWRGD1)
by:
tPGDLY2-1 (ms) ≅ 14 × CPG(µF)
Power-good output signal PWRGD3 (/PWRGD3)
follows the assertion of PWRGD2 by a delay:
tPGDLY3-2 (ms) ≅ 11.5 × CPG(µF)
For example, for a 10µF value for CPG, power-good
output signal PWRGD2 will be asserted 140ms after
PWRGD1. Power-good signal PWRGD3 will then be
asserted 115ms after PWRGD2 and 255ms after the
assertion of PWRGD1. The relationships between
VDRAIN, VPGTH, PWRGD1, PWRGD2, and PWRGD3
are shown in Figure 6.
Undervoltage/Overvoltage Detection (MIC2589 and
MIC2589R)
The MIC2589 and the MIC2589R have “UV” and “OV”
input pins that can be used to detect input supply rail
undervoltage
and
overvoltage
conditions.
Undervoltage lockout prevents the output from
switching on until the supply input is stable and within
tolerance. In a similar fashion, overvoltage shutdown
prevents damage to sensitive circuit components
should the input voltage exceed normal operating
limits. Each of these pins is internally connected to
analog comparators with 20mV of hysteresis. When
the UV pin falls below its VUVL threshold or the OV pin
is above its VOVH threshold, the GATE pin is
immediately pulled low. The GATE pin will be held low
until the UV pin is above its VUVH threshold and the OV
pin is below its VOVL threshold. The circuit’s UV and
OV threshold voltage levels are programmed using
the resistor divider R1, R2, and R3 as shown in the
“Typical Application” circuit and the equations to set
the trip points are shown below. The circuit’s UV
threshold is set to VUV = 37V and the OV threshold is
set at VOV = 72V, values commonly used in Central
Office power distribution applications.
(R1 + R2 + R3)
(R2 + R3)
(R1 + R2 + R3)
(typ) ×
VUV = VUVL (typ) ×
IPGTIMER
where VTHRESH(PG2) (= 0.63V, typically) is the
PWRGD2 threshold voltage for PGTIMER and IPGTIMER
(= 45µA, typically) is the internal PGTIMER charge
current. Similarly, an expression for the sequencing
delay between PWRGD3 and PWRGD2 is given by:
December 2005
tPGDLY3 − 2 =
VOV = VOVL
R3
Given VUV, VOV, and any one of the resistor values,
the remaining two resistor values can be determined.
A suggested value for R3 is selected to provide
approximately 100µA (or more) of current through the
voltage divider chain at VDD = VUV. This yields the
18
M9999-120505
(408) 955-1690
Micrel
MIC2589/MIC2595
following as a starting point:
R3 =
VOVH (typ) 1.223V
=
= 12.23kΩ
100µA
100µA
The closest standard 1% value for R3 = 12.4kΩ.
Solving for R2 and R1 yields:
⎡⎛ V
R2 = R3 × ⎢⎜⎜ OV
⎣⎢⎝ VUV
⎞ ⎤
⎟ − 1⎥
⎟
⎠ ⎦⎥
⎡⎛ 72V ⎞ ⎤
R2 = 12.4kΩ × ⎢⎜
⎟ − 1⎥ = 11.73kΩ
⎣⎝ 37V ⎠ ⎦
fault condition. Should either event occur, the GATE
pin is immediately pulled low and will remain low until
the ON pin voltage once again rises above its VONH
threshold. The circuit’s turn-on and turn-off voltage
levels are set using the resistor divider R1, R2, and
R3 similar to the “Typical Application” circuit and the
equations to set the trip points are shown below. For
the following example, the circuit’s ON threshold is set
to VON = 40V and the circuit’s OFF threshold is set to
VOFF = 35V.
VON = VONH (typ) ×
The closest standard 1% values for R2 = 11.8kΩ.
Lastly, the value for R1 is calculated:
− 1.223V ) ⎤
⎡ (V
R1 = R3 × ⎢ OV
⎥ − R2
1.223V
⎣
⎦
⎡ (72V − 1.223V ) ⎤
R1 = 12.4kΩ × ⎢
⎥ − 11.8kΩ
1.223V
⎣
⎦
R1 = 705.81kΩ
The closest standard 1% value for R1 = 698kΩ.
Using standard 1% resistor values, the circuit’s
nominal UV and OV thresholds are:
VUV = 36.5V
VOV = 71.2V
Good general engineering design practices must
consider the tolerances associated with these
parameters, including but not limited to, power supply
tolerance, undervoltage and overvoltage threshold
tolerances, and the tolerances of the external passive
components.
Programmable UVLO Hysteresis (MIC2595 and
MIC2595R)
The MIC2595 and the MIC2595R devices have userprogrammable hysteresis by means of the ON and
OFF pins (Pins 4 and 3, respectively). This allows
setting the MIC2595/MIC2595R to turn on at a voltage
V1, and not turn off until a second voltage V2, where
V2 < V1. This can significantly simplify dealing with
source impedances in the supply buss while at the
same time increasing the amount of available
operating time from a loosely regulated power rail (for
example, a battery supply). The MIC2595/MIC2595R
holds the output off until the voltage at the ON pin is
above its VONH threshold value given in the “Electrical
Characteristics” table. Once the output has been
enabled by the ON pin, it will remain on until the
voltage at the OFF pin falls below its respective VOFFL
threshold value, or the part turns off due to an external
December 2005
VOFF = VOFFL
(R1 + R2 + R3 )
R3
(R1 + R2 + R3 )
(typ) ×
R2 + R3
Given VOFF, VON, and any one of the resistor values,
the remaining two resistor values can be readily
determined. A suggested value for R3 is selected to
provide approximately 100µA (or more) of current
through the voltage divider chain at VDD = VOFF. This
yields the following as a starting point:
VOFFL (typ) 1.223V
=
= 12.23kΩ
100µA
100µA
The closest standard 1% value for R3 = 12.4kΩ.
Solving for R2 and R1 yields:
R3 =
⎡⎛ V
⎞⎤
R2 = R3 × ⎢⎜⎜ ON − 1⎟⎟⎥
⎠⎥⎦
⎣⎢⎝ VOFF
⎡⎛ 40V
⎞⎤
− 1⎟⎥ = 1.77kΩ
R2 = 12.4kΩ × ⎢⎜
35V
⎠⎦
⎣⎝
The closest standard 1% value for R2 = 1.78kΩ.
Lastly, the value for R1 is calculated:
R1 = R3 ×
(VON − 1.223V )
− R2
1.223V
40V − 1.223V
− 1.78kΩ
R1 = 12.4kΩ ×
1.223V
R1 = 391.38kΩ
The closest standard 1% value for R1 = 392kΩ.
Using standard 1% resistor values, the circuit’s
nominal ON and OFF thresholds are:
VON = 40.1V
VOFF = 35V
Good general engineering design practices must
consider the tolerances associated with these
parameters, including but not limited to, power supply
tolerance, undervoltage and overvoltage threshold
tolerances, and the tolerances of the external passive
components.
19
M9999-120505
(408) 955-1690
Micrel
MIC2589/MIC2595
Application Information
Optional External Circuits for Added
Protection/Performance
In many telecom applications, it is very common for
circuit boards to encounter large-scale supply-voltage
transients in backplane environments. Because
backplanes
present
a
complex
impedance
environment, these transients can be as high as 2.5
times steady-state levels, or 120V in worst-case
situations. In addition, a sudden load dump anywhere
on the circuit card can generate a very high voltage
spike at the drain of the output MOSFET that will
appear at the DRAIN pin of the MIC2589/MIC2595. In
both cases, it is good engineering practice to include
protective measures to avoid damaging sensitive ICs
or the hot swap controller from these large-scale
transients. Two typical scenarios in which large-scale
transients occur are described below:
1. An output current load dump with no bypass
(charge bucket or bulk) capacitance to VEE.
For example, if LLOAD = 5µH, VIN = 56V and
tOFF = 0.7µs, the resulting peak short-circuit
current prior to the MOSFET turning off
would reach:
(56V × 0.7 µs ) = 7.8 A
5 µH
If there is no other path for this current to
take when the MOSFET turns off, it will
avalanche the drain-source junction of the
MOSFET.
Since
the
total
energy
represented is small relative to the
sturdiness of modern power MOSFETs, it’s
unlikely that this will damage the transistor.
However, the actual avalanche voltage is
unknown; all that can be guaranteed is that
it
will
be
greater
than
the
VBD(D-S) of the MOSFET. The drain of the
transistor is connected to the DRAIN pin of
December 2005
the MIC2589/MIC2595, and the resulting
transient does have enough voltage and
energy to damage this, or any, high-voltage
hot swap controller.
2. If the load’s bypass capacitance (for
example, the input filter capacitors for DCDC converter module(s)) is on a board from
which the board with the MIC2589/MIC2595
and the MOSFET can be unplugged, the
same type of inductive transient damage
can occur to the MIC2589/MIC2595.
For many applications, the use of additional circuit
components can be implemented for optimum system
performance and/or protection. The circuit, shown in
Figure 7, includes several components to address
some the following system (dynamic) responses
and/or functions: 1) suppression of transient voltage
spikes, 2) elimination of false “tripping” of the circuit
breaker due to undervoltage and overcurrent glitches,
and 3) the implementation of an external reset circuit.
It is not mandatory that these techniques be utilized,
however, the application environment will dictate
suitability. For protection against sudden on-card load
dumps at the DRAIN pin of the MIC2589/MIC2595
controller, a 68V, 1W, 5% Zener diode clamp (D2)
connected from the DRAIN to the VEE of the
controller can be implemented as shown. To protect
the controller from large-scale transients at the card
input, a 100V clamp diode (D1, SMAT70A or
equivalent) can be used. In either case, very short
lead lengths and compact layout design is strongly
recommended to prevent unwanted transients in the
protection circuitry. Power buss inductance often
produces localized (plug-in card) high-voltage
transients during a turn-off event. Managing these
repeated voltage stresses with sufficient input bulk
capacitance and/or transient suppressing diode
clamps is highly recommended for maximizing the life
of the hot swap controller(s).
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MIC2589/MIC2595
-48V RTN
(Long Pin)
-48V RTN
R5
47k
*D1
SMAT70A
1
R1
689k
1%
-48V RTN
(Short Pin)
System
Reset
R2
11.8k
1%
C1
0.47µF
CFILTER
2.2µF
*C3
0.1µF
*M2
CTIMER
R3
12.4k
1%
CNLD
0.068µF
MIC2589-2BM
/PWRGD1
VDD
14
2
PGTIMER
/PWRGD3
13
3
UV
/PWRGD2
12
4
OV
DRAIN
11
GATE
10
5
CFILTER
6
CNLD
7
VEE
SENSE
9
N/C
8
C5
100µF
*C2
0.1µF
*D2
ZMM5266B
R4
10
RSENSE
0.01
5%
-48V RTN
(Long Pin)
C4
0.1µF
M1
SUM110N10-09
-48VOUT
*Optional components (See Functional Description and Applications Information for more details)
M2 is an SOT-323, B8S138W or equivalent
D2 is a 68V, 500mW Zener diode
Figure 7. Optional Components for Added Performance/Protection
The circuit in Figure 7 consisting of M2, R5, and a
digital control signal, can be used to reset the
controller after the GATE (and output) turns off. Once
the output has been latched off, applying a low-highlow pulse on the GATE of M2 via the System Enable
control can toggle the UV pin. System Enable is a
user-defined signal referenced to VEE.
value of RSENSE has been calculated, it is good
practice to check the maximum hot swap load current
(IHOT_SWAP(MAX)) that the circuit may let pass in the case
of tolerance build-up in the opposite direction. Here,
the worse case maximum is found using a VTRIP(MAX)
threshold of 60mV and a sense resistor 3% low in
value:
Sense Resistor Selection
The sense resistor is nominally valued at:
IHOT_SWAP(m ax) =
R SENSE(nom) =
VTRIP(typ)
IHOT_SWAP(n om)
where VTRIP(TYP) is the typical (or nominal) circuit
breaker threshold voltage (50mV) and IHOT_SWAP(NOM) is
the nominal load current level necessary to trip the
internal circuit breaker.
To accommodate worse-case tolerances in the sense
resistor (for a ±1% initial tolerance, allow ±3%
tolerance for variations over time and temperature)
and circuit breaker threshold voltages, a slightly more
detailed calculation must be used to determine the
minimum and maximum hot swap load currents.
As the MIC2589’s minimum current limit threshold
voltage is 40mV, the minimum hot swap load current
is determined where the sense resistor is 3% high:
IHOT_SWAP(m in) =
40mV
38.8mV
(1.03 × R SENSE(nom) ) = R SENSE(nom)
Keep in mind that the minimum hot swap load current
should be greater than the application circuit’s upper
steady-state load current boundary. Once the lower
December 2005
60mV
61.9mV
(0.97 × R SENSE(nom) ) = R SENSE(nom)
In this case, the application circuit must be sturdy
enough to operate up to approximately 1.5x the
steady-state hot swap load current. For example, if an
MIC2589 circuit must pass a minimum hot swap load
current of 4A without nuisance trips, RSENSE should be
set to:
R SENSE(nom) =
40mV
= 10mΩ
4A
where the nearest 1% standard value is 10.0mΩ. At
the other tolerance extremes, IHOT_SWAP(MAX) for the
circuit in question is then simply:
IHOT_SWAP(m ax) =
61.9mV
= 6.19A
10mΩ
With a knowledge of the application circuit’s maximum
hot swap load current, the power dissipation rating of
the sense resistor can be determined using P = I2 × R.
Here, the current is IHOT_SWAP(max) = 6.19A and the
resistance
RSENSE(max) = (1.03)(RSENSE(nom)) = 10.3mΩ.
Thus, the sense resistor’s maximum power dissipation
is:
PMAX = (6.19A)2 × (10.3mΩ) = 0.395W
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A 0.5W sense resistor is a good choice in this
application.
Power MOSFET Selection
Selecting the proper external MOSFET for use with
theMIC2589/MIC2595 involves three straightforward
tasks:
• Choice of a MOSFET that meets minimum voltage
requirements.
• Selection of a device to handle the maximum
continuous current (steady-state thermal issues).
• Verify the selected part’s ability to withstand any
peak currents (transient thermal issues).
Power MOSFET Operating Voltage Requirements
The first voltage requirement for the MOSFET is that
the drain-source breakdown voltage of the MOSFET
must be greater than VIN(MAX) = VDD – VEE(min).
The second breakdown voltage criterion that must be
met is the gate-source voltage. For the
MIC2589/MIC2595, the gate of the external MOSFET
is driven up to a maximum of 11V above VEE. This
means that the external MOSFET must be chosen to
have a gate-source breakdown voltage of 12V or
more; 20V is recommended. Most power MOSFETs
with a 20V gate-source voltage rating have a 30V
drain-source breakdown rating or higher. For many
48V telecom applications, transient voltage spikes can
approach, and sometimes exceed, 100V. The
absolute maximum input voltage rating of the
MIC2589/MIC2595 is 100V; therefore, a drain-source
breakdown voltage of 100V is suggested for the
external MOSFET. Additionally, an external input
voltage clamp is strongly recommended for
applications that do not utilize conditioned power
supplies.
MIC2589/MIC2595
voltage of 4.5V and 10V. For MIC2589/MIC2595
applications, choose the gate-source ON resistance at
10V and call this value RON. Since a heavily enhanced
MOSFET acts as an ohmic (resistive) device, almost
all that’s required to determine steady-state power
dissipation is to calculate I2R. The one addendum to
this is that MOSFETs have a slight increase in RON
with
increasing
die
temperature.
A
good
approximation for this value is 0.5% increase in RON
per °C rise in junction temperature above the point at
which RON was initially specified by the manufacturer.
For instance, if the selected MOSFET has a
calculated RON of 10mΩ at TJ = 25°C, and the actual
junction temperature ends up at 110°C, a good first
cut at the operating value for RON would be:
RON ≈ 10mΩ[1 + (110 – 25)(0.005)] ≈14.3mΩ
The final step is to make sure that the heat sinking
available to the MOSFET is capable of dissipating at
least as much power (rated in °C/W) as that with
which the MOSFET’s performance was specified by
the manufacturer. Here are a few practical tips:
1. The heat from a TO-263 power MOSFET
flows almost entirely out of the drain tab. If
the drain tab can be soldered down to one
square inch or more, the copper will act as
the heat sink for the part. This copper must
be on the same layer of the board as the
MOSFET drain.
2. Airflow works. Even a few LFM (linear feet
per minute) of air will cool a MOSFET down
substantially. If you can, position the
MOSFET(s) near the inlet of a power
supply’s fan, or the outlet of a processor’s
cooling fan.
3. The best test of a candidate MOSFET for
an application (assuming the above tips
show it to be a likely fit) is an empirical one.
Check the MOSFET’s temperature in the
actual layout of the expected final circuit, at
full operating current. The use of a
thermocouple on the drain leads, or infrared
pyrometer on the package, will then give a
reasonable idea of the device’s junction
temperature.
Power MOSFET Steady-State Thermal Issues
The selection of a MOSFET to meet the maximum
continuous current is a fairly straightforward exercise.
First, arm yourself with the following data:
• The value of ILOAD(CONT, MAX.) for the output in
question (see Sense Resistor Selection).
• The manufacturer’s datasheet for the candidate
MOSFET.
• The maximum ambient temperature in which the
device will be required to operate.
• Any knowledge you can get about the heat sinking
available to the device (e.g., can heat be dissipated
into the ground plane or power plane, if using a
surface-mount part? Is any airflow available?).
The datasheet will almost always give a value of ON
resistance for a given MOSFET at a gate-source
December 2005
Power MOSFET Transient Thermal Issues
If the prospective MOSFET has been shown to
withstand the environmental voltage stresses and the
worst-case steady-state power dissipation is
addressed, the remaining task is to verify if the
MOSFET is capable of handling extreme overcurrent
load faults, such as a short circuit, without
overheating. A power MOSFET can handle a much
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higher pulsed power without damage than its
continuous power dissipation ratings imply due to an
inherent trait, thermal inertia. With respect to the
specification and use of power MOSFETs, the
parameter of interest is the “Transient Thermal
Impedance”, or Zθ, which is a real number (variable
factor) used as a multiplier of the thermal resistance
(Rθ). The multiplier is determined using the given
“Transient Thermal Impedance Graph”, normalized to
Rθ, that displays curves for the thermal impedance
versus power pulse duration and duty cycle. The
single-pulse curve is appropriate for most hot swap
applications. Zθ is specified from junction-to-case for
power MOSFETs typically used in telecom
applications.
The following example provides a method for
estimating the peak junction temperature of a power
MOSFET in determining if the MOSFET is suitable for
a
particular
application.
VIN (VDD – VEE) = 48V, ILIM = 4.2A, tFLT is 20ms, and
the power MOSFET is the SUM110N10-09 (TO-263
package) from Vishay-Siliconix. This MOSFET has an
RON of 9.5mΩ (TJ = 25°C), the junction-to-case
thermal resistance (Rθ(J-C)) is 0.4°C/W, junction-toambient thermal resistance (Rθ(J-A)) is 40°C/W, and the
Transient Thermal Impedance Curve is shown in
Figure 8. Consider, say, the MOSFET is switched on
at time t1 and the steady-state load current passing
through the MOSFET is 3A. At some point in time
after t1, at time t2, there is an unexpected short-circuit
applied to the load, causing the MIC2589/MIC2595
controller to adjust the GATE output voltage and
regulate the load current for 20ms at the programmed
current limit value, 4.2A in this example. During this
short-circuit load condition, the dissipation in the
MOSFET is calculated by:
PD(short) = VDS × ILIM ; VDS = 0V – (-48V) = 48V
PD(short) = 48V × 4.2A = 201.6W for 20ms.
At first glance, it would appear that a very hefty
MOSFET is required to withstand this extreme
overload condition. Upon further examination, the
calculation to approximate the peak junction
temperature is not a difficult task. The first step is to
determine the maximum steady-state junction
temperature, then add the rise in temperature due to
the maximum power dissipated during a transient
overload caused by a short circuit condition. The
equation to estimate the maximum steady-state
junction temperature is given by:
TJ(steady-state) ≅ TC(max) + ∆TJ
(1)
TC(max) is the highest anticipated case temperature,
prior to an overcurrent condition, at which the
MOSFET will operate and is estimated from the
December 2005
MIC2589/MIC2595
following equation based on the highest ambient
temperature of the system environment.
(2)
TC(max) = TA(max) + PD × (Rθ(J-A) – Rθ(J-C))
Let’s assume a maximum ambient of 60°C. The
power dissipation of the MOSFET is determined by
the current through the MOSFET and the ON
resistance (I2RON), which we will estimate at 17mΩ
(specification given at TJ = 125°C).
Using our
example information and substituting into Equation 2,
TC(max)
= 60°C+[((3A)2×17mΩ)×(40–0.4)°C/W]
= 66.06°C
Substituting the variables into Equation 1, TJ is
determined by:
TJ(steady-state) ≅TC(max)+[RON+(TC(max)–TC)(0.005)
× (RON)][I2×(Rθ(J-A)–Rθ(J-C))]
≅ 66.06°C+[17mΩ+(66.06°C–25°C)(0.005/°C)
×(17mΩ)][(3A)2×(40–0.4)°C/W]
≅ 66.06°C + 7.30°C
≅ 73.36°C
Since this is not a closed-form equation, getting a
close approximation may take one or two iterations.
On the second iteration, start with TJ equal to the
value calculated above. Doing so in this example
yields;
TJ(steady-state) ≅66.06°C+[17mΩ+(73.36°C
-25°C)×(0.005/°C)
×(17mΩ)][(3A)2×(40–0.4)]°C/W
≅73.62°C
Another iteration shows that the result (73.63°C) is
converging quickly, so we’ll estimate the maximum
TJ(steady-state) at 74°C.
The use of the Transient Thermal Impedance Curves
is necessary to determine the increase in junction
temperature associated with a worst-case transient
condition.
From our previous calculation of the
maximum power dissipated during a short circuit
event for the MIC2589/MIC2595, we calculate the
transient junction temperature increase as:
TJ(transient) = PD(short) × Rθ(J-C) × Multiplier
(3)
Assume the MOSFET has been on for a long time –
several minutes or more – and delivering the steadystate load current of 3A to the load when the load is
short circuited. The controller will regulate the GATE
output voltage to limit the current to the programmed
value of 4.2A for 20ms before immediately shutting off
the output. For this situation and almost all hot swap
applications, this can be considered a single pulse
event as there is no significant duty cycle. From
Figure 8, find the point on the X-axis (“Square-Wave
Pulse Duration”) for 25ms, allowing for a 25% margin
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MIC2589/MIC2595
of the tFLT, and read up the Y-axis scale to find the
intersection of the Single Pulse curve. This point is
the normalized transient thermal impedance (Zθ(J-C)),
and the effective transient thermal impedance is the
product of Rθ(J-C) and the multiplier, 0.9 in this
example. Solving Equation 3,
TJ(transient) = (201.6W) × (0.4°C/W) × 0.9 = 72.6°C
Finally, add this result to the maximum steady state
junction temperature calculated previously to
determine the estimated maximum transient junction
temperature of the MOSFET: TJ(max.transient) =
74°C + 72.6°C = 146.6°C, which is safely under the
specified maximum junction temperature of 200°C for
the SUM110N10-09.
Figure 8. Transient Thermal Impedance – SUM110N10-09
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PCB Layout Considerations
4-Wire Kelvin Sensing
Because of the low value typically required for the
sense resistor, special care must be used to measure
accurately the voltage drop across it. Specifically, the
measurement technique across each RSENSE must
employ 4-wire Kelvin sensing. This is simply a means
of making sure that any voltage drops in the power
traces connecting to the resistors are not picked up by
the signal conductors measuring the voltages across
the sense resistors.
Figure 9 illustrates how to implement 4-wire Kelvin
sensing. As the figure shows, all the high current in
the circuit (from VEE through RSENSE, and then to the
source of the output MOSFET) flows directly through
the power PCB traces and RSENSE. The voltage drop
resulting across RSENSE is sampled in such a way that
the high currents through the power traces will not
introduce any parasitic voltage drops in the sense
leads. It is recommended to connect the hot swap
controller’s sense leads directly to the sense resistor’s
metalized contact pads.
Other Layout Considerations
Figure 10 is a suggested PCB layout diagram for the
MIC2589/MIC2595. Many hot swap applications will
require load currents of several amperes. Therefore,
December 2005
MIC2589/MIC2595
the power (VEE and Return) trace widths (W) need to
be wide enough to allow the current to flow while the
rise in temperature for a given copper plate (e.g., 1oz.
or 2oz.) is kept to a maximum of 10°C to 25°C. The
return (or power ground) trace should be the same
width as the positive voltage power traces (input/load)
and isolated from any ground and signal planes so
that the controller’s power is common mode. Also,
these traces should be as short as possible in order to
minimize the IR drops between the input and the load.
Finally, the use of plated-through vias will be
necessary to make circuit connections to the power,
ground and signal planes of multi-layer PCBs.
RSENSE metalized
contact pads
Power Trace
From VEE
RSENSE
PCB Track Width:
0.03" per Ampere
using 1oz Cu
Signal Trace
to MIC2589/MIC2595 VEE Pin
Power Trace
To MOSFET Source
Signal Trace
to MIC2589/MIC2595 SENSE Pin
Note: Each SENSE lead trace shall be
balanced for best performance with equal
length/equal aspect ratio.
Figure 9.
RSENSE
25
4-Wire Kelvin Sense Connections for
M9999-120505
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MIC2589/MIC2595
W
Current Flow
to the Load
Vias to bottom layer
Return/Ground plane
(VDD)
MIC2589-2BM
1 /PWRGD1
VDD 14
2 PGTIMER
/PWRGD3 13
3 UV
/PWRGD2 12
4 OV
DRAIN 11
**CTIMER
**R1
**R2
**R3
5
CFILTER
6
CNLD
GATE 10
**CFILTER
C1
0.47uF
SENSE 9
**CNLD
7 VEE
Via to bottom layer
Return/Ground plane
(VDD)
N/C 8
D1
SMAT70A
Via to bottom layer
Return/Ground plane
(VDD)
Current Flow
from the Load
**CGATE
**CLOAD1
**RGATE
CLOAD2
0.1uF
W
W
Via to the
ground plane
*SENSE RESISTOR
(WSR-2 or
WSL2512)
*POWER MOSFET
(TO-263)
- DRAWING IS NOT TO SCALE*See Table 1 for part numbers and vendors
**Component values application specific, determined by user
Trace width (W) guidelines and additional information given in "PCB Layout
Recommendations" section of the datasheet
Figure 10. Recommended PCB Layout for Sense Resistor, Power MOSFET, Overvoltage/Undervoltage
Resistive Divider Network, and Timer Capacitors
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MIC2589/MIC2595
caused by common mode transients. Such is the
case when an EMI filter is utilized to prevent DC-DC
converter switching noise from being injected back
onto the power supply. The circuit of Figure 11 shows
how to configure an optoisolator driven by the
/PWRGD signal of the MIC2589 controller.
Power-Good Signals Driving Optoisolators
The PWRGDx signals can be used to drive
optoisolators or LEDs. The use of an optoisolator is
sometimes needed to protect I/O signals (e.g.,
/PWRGD, RESET, ENABLE) of both the controller
and downstream DC-DC converter(s) from damage
R7
43k
R6
43k
R5
43k
1
OPTO#1
6
1
MOC207-M
2
OPTO#2
6
1
MOC207-M
5
2
OPTO#3
6
MOC207-M
5
2
5
-48V RTN
(Long Pin)
-48V RTN
*D1
SMAT70A
1
R1
689k
1%
-48V RTN
(Short Pin)
R2
11.8k
1%
CTIMER
0.068µF
CFILTER
2.2µF
C1
0.47µF
R3
12.4k
1%
CNLD
0.068µF
MIC2589-2BM
/PWRGD1
VDD
14
2
PGTIMER
/PWRGD3
13
3
UV
/PWRGD2
12
4
OV
DRAIN
11
5
CFILTER
GATE
10
6
CNLD
7
VEE
SENSE
9
N/C
8
IN+
C4
100µF
C3
0.1µF
C6
100µF
C5
0.1µF
*C2
0.1µF
R4
10
-48V RTN
(Long Pin)
C8
100µF
C7
0.1µF
+5VOUT
IN–
OUT–
+5V RTN
IN+
OUT+
+2.5VOUT
ON/OFF*
IN–
OUT–
+2.5V RTN
IN+
OUT+
+1.8VOUT
ON/OFF*
IN–
RSENSE
0.01
5%
OUT+
ON/OFF*
OUT–
+1.8V RTN
M1
SUM110N10-09
Nominal Undervoltage and Overvoltage Thresholds:
VUV = 36.5V
VOV = 71.2V
*Optional components (See Funtional Description and Applications Information for more details)
#An external pull-up resistor for the power-good signal is necessary for DC-DC convertors (and all other load modules) not equipped with an
internal pull-up impedance
Figure 11. Optoisolators Driven by /PWRGD Signals
December 2005
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MIC2589/MIC2595
MOSFET and Sense Resistor Vendors
Device types, part numbers, and manufacturer
contacts for power MOSFTETS and sense resistors
are provided in Table 1.
MOSFET Vendors
Breakdown Voltage (VDSS)
Contact Information
SUM75N06-09L (TO-263)
SUM70N06-11 (TO-263)
SUM50N06-16L (TO-263)
60V
60V
60V
www.siliconix.com
(203) 452-5664
SUP85N10-10 (TO-220AB)
SUB85N10-10 (TO-263)
SUM110N10-09 (TO-263)
SUM60N10-17 (TO-263)
100V
100V
100V
100V
www.siliconix.com
(203) 452-5664
International Rectifier
IRF530 (TO-220AB)
IRF540N (TO-220AB)
100V
100V
www.irf.com
(310) 322-3331
Renesas
2SK1298 (TO-3PFM)
2SK1302 (TO-220AB)
2SK1304 (TO-3P)
60V
100V
100V
www.renesas.com
(408) 433-1990
Vishay - Siliconix
Key MOSFET Type(s)
Resistor Vendors
Resistor Types
Contact Data
Vishay (Dale)
“WSL” Series
www.vishay.com/docs/wsl_30100.pdf
(203)452-5664
IRC
“OARS” Series
“LR” Series
www.irctt.com/pdf_files/OARS.pdf
www.irctt.com/pdf_files/LRC.pdf
(second source to “WSL”)
(828) 264-8861
Table 1. MOSFET and Sense Resistors
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MIC2589/MIC2595
Package Information
14-Pin SOIC (M)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http:/www.micrel.com
The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for
its use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a
product can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for
surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant
injury to the user. A Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk
and Purchaser agrees to fully indemnify Micrel for any damages resulting from such use or sale.
© 2004 Micrel, Incorporated.
December 2005
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