TPS5430 www.ti.com SLVS632 – JANUARY 2006 5.5-V to 36-V, 3-A STEP-DOWN SWIFT™ CONVERTER FEATURES APPLICATIONS • • • • • • • • • • • • • • • Wide Input Voltage Range: 5.5 V to 36 V Up to 3-A Continuous (4-A Peak) Output Current High Efficiency up to 95% Enabled by 110-mΩ Integrated MOSFET Switch Wide Output Voltage Range: Adjustable Down to 1.22 V with 1.5% Initial Accuracy Internal Compensation Minimizes External Parts Count Fixed 500 kHz Switching Frequency for Small Filter Size Improved Line Regulation and Transient Response by Input Voltage Feed Forward System Protected by Over Current Limiting and Thermal Shutdown –40°C to 125°C Operating Junction Temperature Range Available in Small Thermally Enhanced 8-Pin SOIC PowerPAD™ Package For SWIFT Documentation, Application Notes and Design Software, See the TI Website at www.ti.com/swift Consumer: Set-top Box, DVD, LCD Displays Industrial and Car Audio Power Supplies Battery Chargers, High Power LED Supply 12V/24V Distributed Power Systems DESCRIPTION As a member of the SWIFT™ family of DC/DC regulators, the TPS5430 is a high-output-current PWM converter that integrates a low resistance high side N-channel MOSFET. Included on the substrate with the listed features are a high performance voltage error amplifier that provides tight voltage regulation accuracy under transient conditions; an under-voltage-lockout circuit to prevent start-up until the input voltage reaches 5.5V; an internally set slow-start circuit to limit inrush currents; and a voltage feed-forward circuit to improve the transient response. Other features include an active high enable, over current protection and thermal shutdown. To reduce design complexity and external component count, the TPS5430 feedback loop is internally compensated. The TPS5430 device is available in a thermally enhanced, easy to use 8-pin SOIC PowerPAD™ package. TI provides evaluation modules and the SWIFT™ Designer software tool to aid in quickly achieving high-performance power supply designs to meet aggressive equipment development cycles. 100 SIMPLIFIED SCHEMATIC VIN 95 90 VOUT VIN PH NC BOOT NC Efficiency − % 85 TPS5430 80 75 70 65 ENA VSENSE GND VI = 12 V, V0 = 5 V, fs = 500 kHz 60 55 50 0 0.5 I O 1 1.5 2 2.5 − Output Current − A 3 3.5 Efficiency vs Output Current Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. SWIFT, PowerPAD are trademarks of Texas Instruments. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2006, Texas Instruments Incorporated TPS5430 www.ti.com SLVS632 – JANUARY 2006 These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. ORDERING INFORMATION (1) TJ OUTPUT VOLTAGE PACKAGE PART NUMBER –40°C to 125°C Adjustable to 1.22 V Thermally Enhanced SOIC (DDA) (1) TPS5430DDA The DDA package is also available taped and reeled. Add an R suffix to the device type (i.e., TPS5430DDAR). See applications section of data sheet for PowerPAD™ drawing and layout information. ABSOLUTE MAXIMUM RATINGS over operating free-air temperature range (unless otherwise noted) (1) (2) VALUE VI Input voltage range VIN –0.3 to 38 ENA –0.3 to 7 VSENSE –0.3 to 3 BOOT –0.3 to 48 BOOT-PH UNIT V 10 PH (steady-state) PH (transient < 10 ns) –0.6 to 38 –1.2 IO Source current PH Internally Limited IO Leakage current PH 10 µA TJ Operating virtual junction temperature range –40 to 125 °C TSTG Storage temperature –65 to 150 °C 300 °C Lead temperature 1,6 mm (1/16-inch) from case for 10 seconds (1) (2) Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings only and functional operation of the device at these or any other conditions beyond those indicated under recommended operating conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. All voltage values are with respect to network ground terminal. DISSIPATION RATINGS (1) (2) PACKAGE 8 Pin DDA (2-layer board with solder) (3) 33°C/W solder) (4) 26°C/W 8 Pin DDA (4-layer board with (1) (2) (3) (4) THERMAL IMPEDANCE JUNCTION-TO-AMBIENT Maximum power dissipation may be limited by overcurrent protection. Power rating at a specific ambient temperature TA should be determined with a junction temperature of 125°C. This is the point where distortion starts to substantially increase. Thermal management of the final PCB should strive to keep the junction temperature at or below 125°C for best performance and long-term reliability. See Thermal Calculations in applications section of this data sheet for more information. Test board conditions: a. 3 in x 3 in, 2 layers, thickness: 0.062 inch. b. 2 oz. copper traces located on the top and bottom of the PCB. c. 6 thermal vias in the PowerPAD area under the device package. Test board conditions: a. 3 in x 3 in, 4 layers, thickness: 0.062 inch. b. 2 oz. copper traces located on the top and bottom of the PCB. c. 2 oz. copper ground planes on the 2 internal layers. d. 6 thermal vias in the PowerPAD area under the device package. RECOMMENDED OPERATING CONDITIONS MIN NOM MAX UNIT VIN Input voltage range 5.5 36 V TJ Operating junction temperature –40 125 °C 2 TPS5430 www.ti.com SLVS632 – JANUARY 2006 ELECTRICAL CHARACTERISTICS TJ = –40°C to 125°C, VIN = 5.5 V to 36.0 V (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT SUPPLY VOLTAGE (VIN PIN) VIN Input voltage range IQ Quiescent current 5.5 36 V 3 4.4 mA 18 50 µA Start threshold voltage, UVLO 5.3 5.5 Hysteresis voltage, UVLO 330 VSENSE = 2 V, Not switching, PH pin open Shutdown, ENA = 0 V UNDER VOLTAGE LOCK OUT (UVLO) V mV VOLTAGE REFERENCE Voltage reference accuracy TJ = 25°C 1.202 1.221 1.239 Io = 0 A – 3 A 1.196 1.221 1.245 400 500 600 kHz 150 200 ns V OSCILLATOR Internally set free-running frequency Minimum controllable on time Maximum duty cycle 87 89 % ENABLE (ENA PIN) Start threshold voltage, ENA 1.3 Stop threshold voltage, ENA 0.5 Hysteresis voltage, ENA V 450 Internal slow-start time (0~100%) V mV 6.6 8 10 ms Current limit 4.0 5.0 6.0 A Current limit hiccup time 13 16 20 ms 135 162 °C 14 °C CURRENT LIMIT THERMAL SHUTDOWN Thermal shutdown trip point Thermal shutdown hysteresis OUTPUT MOSFET RDS-ON High side power MOSFET switch VIN = 5.5 V 150 VIN = 10 V – 36 V 110 230 mΩ 3 TPS5430 www.ti.com SLVS632 – JANUARY 2006 PIN ASSIGNMENTS DDA PACKAGE (TOP VIEW) 8 PH 7 VIN 3 6 GND 4 5 ENA BOOT 1 NC 2 NC VSENSE PowerPAD (Pin 9) TERMINAL FUNCTIONS TERMINAL NAME BOOT NC DESCRIPTION NO. 1 2, 3 Boost capacitor for the high-side FET gate driver. Connect 0.01 µF low ESR capacitor from BOOT pin to PH pin. Not connected internally. VSENSE 4 Feedback voltage for the regulator. Connect to output voltage divider. ENA 5 On/off control. Below 0.5 V, the device stops switching. Float the pin to enable. GND 6 Ground. Connect to PowerPAD. VIN 7 Input supply voltage. Bypass VIN pin to GND pin close to device package with a high quality, low ESR ceramic capacitor. PH 8 Source of the high side power MOSFET. Connected to external inductor and diode. PowerPAD 9 GND pin must be connected to the exposed pad for proper operation. 4 TPS5430 www.ti.com SLVS632 – JANUARY 2006 TYPICAL CHARACTERISTICS OSCILLATOR FREQUENCY vs JUNCTION TEMPERATURE NON-SWITCHING QUIESCENT CURRENT vs JUNCTION TEMPERATURE 530 3.5 I Q − Quiescent Current − mA f − Oscillator Frequency − kHz 520 510 500 490 480 3.25 3 2.75 470 460 −50 0 −25 25 50 75 100 2.5 −50 125 50 75 100 Figure 2. SHUTDOWN QUIESCENT CURRENT vs INPUT VOLTAGE VOLTAGE REFERENCE vs JUNCTION TEMPERATURE 20 T J = 125°C 15 T J = 27°C T J = –40°C 10 5 0 5 10 15 20 25 30 35 1.225 1.220 1.215 1.210 -50 40 -25 V I −Input V oltage −V 0 25 50 75 100 TJ - Junction Temperature - °C Figure 3. Figure 4. ON RESISTANCE vs JUNCTION TEMPERATURE INTERNAL SLOW START TIME vs JUNCTION TEMPERATURE 125 9 180 V I = 12 V TSS − Internal Slow Start Time − ms 170 125 1.230 ENA = 0 V VREF - Voltage Reference - V −µ A I SD −Shutdown Current 25 TJ − Junction Temperature − °C Figure 1. 25 160 150 140 130 120 110 r DS(on) −On Resistance −mΩ 0 −25 T − Junction Temperature − °C 100 8.5 8 7.5 90 80 −50 −25 0 25 50 75 100 T J −Junction Temperature − °C Figure 5. 125 7 −50 −25 0 25 50 75 100 TJ − Junction Temperature − °C 125 Figure 6. 5 TPS5430 www.ti.com SLVS632 – JANUARY 2006 TYPICAL CHARACTERISTICS (continued) MINIMUM CONTROLLABLE ON TIME vs JUNCTION TEMPERATURE MINIMUM CONTROLLABLE DUTY RATIO vs JUNCTION TEMPERATURE 8 170 7.75 Minimum Duty Ratio - % Minimum Controllable On Time − ns 180 160 150 140 7.25 130 120 −50 −25 0 25 50 75 100 TJ − Junction Temperature − °C Figure 7. 6 7.50 125 7 -50 -25 50 0 25 75 100 TJ - Junction Temperature - °C Figure 8. 125 TPS5430 www.ti.com SLVS632 – JANUARY 2006 APPLICATION INFORMATION FUNCTIONAL BLOCK DIAGRAM VIN VIN 1.221 V Bandgap Reference UVLO VREF SHDN Slow Start Boot Regulator BOOT HICCUP 5 µA ENABLE ENA SHDN SHDN VSENSE Z1 Thermal Protection SHDN NC VIN Ramp Generator SHDN SHDN Z2 Feed Forward Gain = 25 NC GND Error Amplifier SHDN PWM Comparator HICCUP Overcurrent Protection Oscillator SHDN Gate Drive Control POWERPAD Gate Driver SHDN TPS5430 BOOT PH VOUT DETAILED DESCRIPTION Oscillator Frequency The internal free running oscillator sets the PWM switching frequency at 500 kHz. The 500 kHz switching frequency allows less output inductance for the same output ripple requirement resulting in a smaller output inductor. Voltage Reference The voltage reference system produces a precision reference signal by scaling the output of a temperature stable bandgap circuit. The bandgap and scaling circuits are trimmed during production testing to an output of 1.221 V at room temperature. Enable (ENA) and Internal Slow Start The ENA pin provides electrical on/off control of the regulator. Once the ENA pin voltage exceeds the threshold voltage, the regulator starts operation and the internal slow start begins to ramp. If the ENA pin voltage is pulled below the threshold voltage, the regulator stops switching and the internal slow start resets. Connecting the pin to ground or to any voltage less than 0.5 V will disable the regulator and activate the shutdown mode. The quiescent current of the TPS5430 in shutdown mode is typically 18 µA. The ENA pin has an internal pullup current source, allowing the user to float the ENA pin. If an application requires controlling the ENA pin, use open drain or open collector output logic to interface with the pin. To limit the start-up inrush current, an internal slow start circuit is used to ramp up the reference voltage from 0 V to its final value linearly. The internal slow start time is 8ms typically. 7 TPS5430 www.ti.com SLVS632 – JANUARY 2006 APPLICATION INFORMATION (continued) Undervoltage Lockout (UVLO) The TPS5430 incorporates an under voltage lockout circuit to keep the device disabled when VIN (the input voltage) is below the UVLO start voltage threshold. During power up, internal circuits are held inactive until VIN exceeds the UVLO start threshold voltage. Once the UVLO start threshold voltage is reached, device start-up begins. The device operates until VIN falls below the UVLO stop threshold voltage. The typical hysteresis in the UVLO comparator is 330 mV. Boost Capacitor (BOOT) Connect a 0.01 µF low-ESR ceramic capacitor between the BOOT pin and PH pin. This capacitor provides the gate drive voltage for the high-side MOSFET. X7R or X5R grade dielectrics are recommended due to their stable values over temperature. Output Feedback (VSENSE) and Internal Compensation The output voltage of the regulator is set by feeding back the center point voltage of an external resistor divider network to the VSENSE pin. In steady-state operation, the VSENSE pin voltage should be equal to the voltage reference 1.221 V. The TPS5430 implements internal compensation to simplify the regulator design. Since the TPS5430 uses voltage mode control, a type 3 compensation network has been designed on chip to provide a high crossover frequency and a high phase margin for good stability. Refer to Internal Compensation Network in the applications section for more details. Voltage Feed Forward The internal voltage feed forward provides a constant DC power stage gain despite any variations with the input voltage. This greatly simplifies the stability analysis and improves the transient response. Voltage feed forward varies the peak ramp voltage inversely with the input voltage so that the modulator and power stage gain are constant at the feed forward gain, i.e. VIN Feed Forward Gain Ramp pkpk (1) The typical feed forward gain of TPS5430 is 25. Pulse-Width-Modulation (PWM) Control The regulator employs a fixed frequency pulse-width-modulator (PWM) control method. First, the feedback voltage (VSENSE pin voltage) is compared to the constant voltage reference by the high gain error amplifier and compensation network to produce a error voltage. Then, the error voltage is compared to the ramp voltage by the PWM comparator. In this way, the error voltage magnitude is converted to a pulse width which is the duty cycle. Finally, the PWM output is fed into the gate drive circuit to control the on-time of the high-side MOSFET. Overcurrent Protection Overcurrent protection is implemented by sensing the drain-to-source voltage across the high-side MOSFET. The drain to source voltage is then compared to a voltage level representing the overcurrent threshold limit. If the drain-to-source voltage exceeds the overcurrent threshold limit, the overcurrent indicator is set true. The system will ignore the overcurrent indicator for the leading edge blanking time at the beginning of each cycle to avoid any turn-on noise glitches. Once overcurrent indicator is set true, overcurrent protection is triggered. The high-side MOSFET is turned off for the rest of the cycle after a propagation delay. The overcurrent protection scheme is called cycle-by-cycle current limiting. If the sensed current continues to increase during cycle-by-cycle current limiting, the hiccup mode overcurrent protection will be triggered instead of cycle-by-cycle current limiting. During hiccup mode overcurrent protection, the voltage reference is grounded and the high-side MOSFET is turned off for the hiccup time. Once the hiccup time duration is complete, the regulator restarts under control of the slow start circuit. 8 TPS5430 www.ti.com SLVS632 – JANUARY 2006 APPLICATION INFORMATION (continued) Thermal Shutdown The TPS5430 protects itself from overheating with an internal thermal shutdown circuit. If the junction temperature exceeds the thermal shutdown trip point, the voltage reference is grounded and the high-side MOSFET is turned off. The part is restarted under control of the slow start circuit automatically when the junction temperature drops 14°C below the thermal shutdown trip point. PCB Layout Connect a low ESR ceramic bypass capacitor to the VIN pin. Care should be taken to minimize the loop area formed by the bypass capacitor connections, the VIN pin, and the TPS5430 ground pin. The best way to do this is to extend the top side ground area from under the device adjacent to the VIN trace, and place the bypass capacitor as close as possible to the VIN pin. The minimum recommended bypass capacitance is 10 uF ceramic with a X5R or X7R dielectric. There should be a ground area on the top layer directly underneath the IC, with an exposed area for connection to the PowerPAD. Use vias to connect this ground area to any internal ground planes. Use additional vias at the ground side of the input and output filter capacitors as well. The GND pin should be tied to the PCB ground by connecting it to the ground area under the device as shown below. The PH pin should be routed to the output inductor, catch diode and boot capacitor. Since the PH connection is the switching node, the inductor should be located very close to the PH pin and the area of the PCB conductor minimized to prevent excessive capacitive coupling. The catch diode should also be placed close to the device to minimize the output current loop area. Connect the boot capacitor between the phase node and the BOOT pin as shown. Keep the boot capacitor close to the IC and minimize the conductor trace lengths. The component placements and connections shown work well, but other connection routings may also be effective. Connect the output filter capacitor(s) as shown between the VOUT trace and GND. It is important to keep the loop formed by the PH pin, Lout, Cout and GND as small as is practical. Connect the VOUT trace to the VSENSE pin using the resistor divider network to set the output voltage. Do not route this trace too close to the PH trace. Due to the size of the IC package and the device pin-out, the trace may need to be routed under the output capacitor. Alternately, the routing may be done on an alternate layer if a trace under the output capacitor is not desired. If the grounding scheme shown is utilized, use a via connection to a different layer to route to the ENA pin. 9 TPS5430 www.ti.com SLVS632 – JANUARY 2006 APPLICATION INFORMATION (continued) PH BOOT CAPACITOR BOOT OUTPUT INDUCTOR RESISTOR DIVIDER VOUT PH NC VIN NC GND VSENSE ENA OUTPUT FILTER CAPACITOR Route feedback trace under output filter capacitor or on other layer CATCH DIODE INPUT INPUT BULK BYPASS CAPACITOR FILTER TOPSIDE GROUND AREA VIA to Ground Plane Signal VIA Figure 9. Design Layout 10 Vin TPS5430 www.ti.com SLVS632 – JANUARY 2006 APPLICATION INFORMATION (continued) 0.110 0.220 0.026 0.118 0.050 0.050 0.080 0.013 DIA 4 PL 0.040 0.098 All dimensions in inches Figure 10. TPS5430 Land Pattern Application Circuits Figure 11 shows the schematic for a typical TPS5430 application. The TPS5430 can provide up to 3-A output current at a nominal output voltage of 5.0 V. For proper thermal performance the exposed PowerPAD underneath the device must be soldered down to the printed circuit board. U1 TPS5430DDA 10.8 - 19.8 V VIN EN C1 10 mF 7 5 2 3 6 VIN C2 0.01 mF L1 15 mH 5V 1 VOUT BOOT ENA 8 NC PH NC 4 D1 B340A + C3 220 mF R1 10 kW VSNS GND PwPd 9 R2 3.24 kW Figure 11. Application Circuit, 12-V to 5.0-V Design Procedure The following design procedure can be used to select component values for the TPS5430. Alternately, the SWIFT Designer Software may be used to generate a complete design. The SWIFT Designer Software uses an iterative design procedure and accesses a comprehensive database of components when generating a design. This section presents a simplified discussion of the design process. To begin the design process a few parameters must be decided upon. The designer needs to know the following: 11 TPS5430 www.ti.com SLVS632 – JANUARY 2006 APPLICATION INFORMATION (continued) • • • • • • Input voltage range Output voltage Input ripple voltage Output ripple voltage Output current rating Operating frequency Design Parameters For this design example, use the following as the input parameters: (1) DESIGN PARAMETER (1) EXAMPLE VALUE Input voltage range 10.8 V to 19.8 V Output voltage 5.0 V Input ripple voltage 300 mV Output ripple voltage 30 mV Output current rating 3A Operating frequency 500 kHz As an additional constraint, the design is set up to be small size and low component height. Switching Frequency The switching frequency for the TPS5430 is internally set to 500 kHz. It is not possible to adjust the switching frequency. Input Capacitors The TPS5430 requires an input decoupling capacitor and, depending on the application, a bulk input capacitor. The recommended value for the decoupling capacitor, C1, is 10 µF. A high quality ceramic type X5R or X7R is required. For some applications a smaller value decoupling capacitor may be used, so long as the input voltage and current ripple ratings are not exceeded. The voltage rating must be greater than the maximum input voltage, including ripple. This input ripple voltage can be approximated by Equation 2 : VIN I OUT(MAX) 0.25 C BULK ƒ sw I OUT(MAX) ESR MAX (2) Where IOUT(MAX) is the maximum load current, fSW is the switching frequency, CIN is the input capacitor value and ESRMAX is the maximum series resistance of the input capacitor. The maximum RMS ripple current also needs to be checked. For worst case conditions, this can be approximated by Equation 3 : I OUT(MAX) I CIN 2 (3) In this case the input ripple voltage would be 156 mV and the RMS ripple current would be 1.5 A. The maximum voltage across the input capacitors would be VIN max plus delta VIN/2. The chosen input decoupling capacitor is rated for 25 V and the ripple current capacity is greater than 3 A, providing ample margin. It is very important that the maximum ratings for voltage and current are not exceeded under any circumstance. Additionally some bulk capacitance may be needed, especially if the TPS5430 circuit is not located within about 2 inches from the input voltage source. The value for this capacitor is not critical but it also should be rated to handle the maximum input voltage including ripple voltage and should filter the output so that input ripple voltage is acceptable. 12 TPS5430 www.ti.com SLVS632 – JANUARY 2006 Output Filter Componts Two components need to be selected for the output filter, L1 and C2. Since the TPS5430 is an internally compensated device, a limited range of filter component types and values can be supported. Inductor Selection To calculate the minimum value of the output inductor, use Equation 4: V V V OUT(MAX) IN(MAX) OUT L MIN V K I F IN(max) IND OUT SW (4) KIND is a coefficient that represents the amount of inductor ripple current relative to the maximum output current. Three things need to be considered when determining the amount of ripple current in the inductor: the peak to peak ripple current affects the output ripple voltage amplitude, the ripple current affects the peak switch current and the amount of ripple current determines at what point the circuit will become discontinuous. For designs using the TPS5430, KIND of 0.2 to 0.3 yields good results. Low output ripple voltages can be obtained when paired with the proper output capacitor, the peak switch current will be well below the current limit set point and relatively low load currents can be sourced before dicontinuous operation. For this design example use KIND = 0.2 and the minimum inductor value is calculated to be 12.5µH. The next highest standard value is 15 µH, which is used in this design. For the output filter inductor it is important that the RMS current and saturation current ratings not be exceeded. The RMS inductor current can be found from Equation 5: 2 I L(RMS) I2 1 OUT(MAX) 12 V OUT V IN(MAX) V OUT V L F 0.8 IN(MAX) OUT SW (5) and the peak inductor current can be determined with Equation 6: V I L(PK) I OUT(MAX) OUT V IN(MAX) V 1.6 V IN(MAX) L OUT OUT F SW (6) For this design, the RMS inductor current is 3.003 A, and the peak inductor current is 3.31 A. The chosen inductor is a Sumida CDRH104R-150 15µH. It has a saturation current rating of 3.4 A and a RMS current rating of 3.6 A, easily meeting these requirements. A lesser rated inductor could be used, however this device was chosen because of its low profile component height. In general, inductor values for use with the TPS5430 are in the range of 10 µH to 100 µH. Capacitor Selection The important design factors for the output capacitor are dc voltage rating, ripple current rating, and equivalent series resistance (ESR). The dc voltage and ripple current ratings cannot be exceeded. The ESR is important because along with the inductor ripple current it determines the amount of output ripple voltage. The actual value of the output capacitor is not critical, but some practical limits do exist. Consider the relationship between the desired closed loop crossover frequency of the design and LC corner frequency of the output filter. Due to the design of the internal compensation, it is desirable to keep the closed loop crossover frequency in the range 3 kHz to 30 kHz as this frequency range has adequate phase boost to allow for stable operation. For this design example, it is assumed that the intended closed loop crossover frequency will be between 2590 Hz and 24 kHz and also below the ESR zero of the output capacitor. Under these conditions the closed loop crossover frequency will be related to the LC corner frequency by: f CO f LC 2 85 VOUT (7) And the desired output capacitor value for the output filter to: 13 TPS5430 www.ti.com SLVS632 – JANUARY 2006 C OUT 1 3357 L OUT f CO V OUT (8) For a desired crossover of 18 kHz and a 15-µH inductor, the calculated value for the output capacitor is 220 µF. The capacitor type shold be chosen so that the ESR zero is above the loop crossover. The maximum ESR should be: (Add new equation) 1 ESR MAX 2 C OUT f CO (9) The maximum ESR of the output capacitor also determines the amount of output ripple as specified in the initial design parameters. The output ripple voltage is the inductor ripple current times the ESR of the output filter. Check that the maximum specified ESR as listed in the capacitor data sheet will result in an acceptable output ripple voltage: V p−p(MAX) ESRMAX VOUT VIN(MAX) V OUT NC VIN(MAX) LOUT F SW (10) Where: ∆ VP-P is the desired peak-to-peak output ripple. NC is the number of parallel output capacitors. FSW is the switching frequency. For this design example, a single 220-µF output capacitor is chosen for C3. The calculated RMS ripple current is 143 mA and the maximum ESR required is 40 mΩ. A capacitor that meets these requirements is a Sanyo Poscap 10TPB220M, rated at 10 V with a maximum ESR of 40 mΩ and a ripple current rating of 3.0 A. An additional small 0.1-µF ceramic bypass capacitor may also used, but is not included in this design. The minimum ESR of the output capacitor should also be considered. For good phase margin, the ESR zero when the ESR is at a minimum should not be too far above the internal compensation poles at 24 and 54 kHz. The selected output capacitor must also be rated for a voltage greater than the desired output voltage plus one half the ripple voltage. Any derating amount must also be included. The maximum RMS ripple current in the output capacitor is given by Equation 11: VOUT VIN(MAX) V OUT ICOUT(RMS) 1 12 V L F N IN(MAX) OUT SW C (11) Where: NC is the number of output capacitors in parallel. FSW is the switching frequency. Other capacitor types can be used with the TPS5430, depending on the needs of the application. Output Voltage Setpoint The output voltage of the TPS5430 is set by a resistor divider (R1 and R2) from the output to the VSENSE pin. Calculate the R2 resistor value for the output voltage of 5.0 V using Equation 12: R1 1.221 R2 V 1.221 OUT (12) For any TPS5430 design, start with an R1 value of 10 kΩ. R2 is then 3.24 kΩ. Boot Capacitor The boot capacitor should be 0.01 µF. 14 TPS5430 www.ti.com SLVS632 – JANUARY 2006 Catch Diode The TPS5430 is designed to operate using an external catch diode between PH and GND. The selected diode must meet the absolute maximum ratings for the application: Reverse voltage must be higher than the maximum voltage at the PH pin, which is VINMAX + 0.5 V. Peak current must be greater than IOUTMAX plus on half the peak to peak inductor current. Forward voltage drop should be small for higher efficiencies. It is important to note that the catch diode conduction time is typically longer than the high-side FET on time, so attention paid to diode parameters can make a marked improvement in overall efficiency. Additionally, check that the device chosen is capable of dissipating the power losses. For this design, a Diodes, Inc. B340A is chosen, with a reverse voltage of 40 V, forward current of 3 A, and a forward voltage drop of 0.5V. Additional Circuits Figure 12 shows an application circuit utilizing a wide input voltage range. The design parameters are simillar to those given for the design example, with a larger value output inductor and a lower closed loop crosover frequency. U1 TPS5430DDA VIN 12–35 V EN C1 10 mF 7 5 2 3 6 1 VIN BOOT ENA 8 PH NC 4 NC GND VSNS PwPd 9 L1 22 mH C2 0.01 mF 5V VOUT D1 + C3 220 mF R1 10 kW SBL845 R2 3.24 kW Figure 12. 12-35 V Input to 5 V Output Application Circuit ADVANCED INFORMATION Output Voltage Limitations Due to the internal design of the TPS5430, there are both upper and lower output voltage limits for any given input voltage. The upper limit of the output voltage set point is constrained by the maximum duty cycle of 87% and is given by: V OUTMAX 0.87 VINMIN I OMAX 0.230 VD I OMAX RL VD (13) Where VINMIN = minimum input voltage IOMAX = maximum load current VD = catch diode forward voltage. RL= output inductor series resistance. This equation assumes maximum on resistance for the internal high side FET. The lower limit is constrained by the minimum controllable on time which may be as high as 200nsec. The approximate minimum output voltage for a given input voltage and minimum load current is given by: V OUTMIN 0.12 VINMAX I OMIN 0.110 VD I OMIN RL VD (14) Where VINMAX = maximum input voltage IOMIN = minimum load current VD = catch diode forward voltage. RL= output inductor series resistance. 15 TPS5430 www.ti.com SLVS632 – JANUARY 2006 This equation assumes nominal on resistance for the high side FET and accounts for worst case variation of operating frequency set point. Any design operating near the operational limits of the device should be carefully checked to assure proper functionality. Internal Compensation Network The design equations given in the example circuit can be used to generate circuits using the TPS5430. These designs are based on certain assumptions and will tend to always select output capacitors within a limited range of ESR values. If a different capacitor type is desired, it may be posssible to to fit one to the internal compensation of the TPS5430. Equation 15 gives the nominal frequency response of the internal voltage-mode type III compensation network: s s 1 1 2Fz1 2Fz2 H(s) s s s s 1 1 1 2Fp0 2Fp1 2Fp2 2Fp3 (15) Where Fp0 = 2165 Hz, Fz1 = 2170 Hz, Fz2 = 2590 Hz Fp1 = 24 kHz, Fp2 = 54 kHz, Fp3 = 440 kHz Fp3 represents the non-ideal parasitics effect. Using this information along with the desired output voltage, feed forward gain and output filter characteristics,the closed loop transfer function can be derived. Thermal Calculations The following formulas show how to estimate the device power dissipation under continuous conduction mode operations. They should not be used if the device is working at light loads in the discontinuous conduction mode. Conduction Loss: Pcon=Io2× Rds,on × VOUT/VIN Switching Loss: Psw = VIN × IOUT × 0.01 Quiescent Current Loss: Pq = VIN × 0.01 Total Loss: Ptot = Pcon + Psw + Pq Given TA => Estimated Junction Temperature: TJ = TA + Rth × Ptot Given TJMAX = 125°C => Estimated Maximum Ambient Temperature: TAMAX = TJMAX– Rth x Ptot 16 TPS5430 www.ti.com SLVS632 – JANUARY 2006 PERFORMANCE GRAPHS The performance graphs in Figures 12 - 18 are applicable to the circuit in Figure 10. Ta = 25 °C. unless otherwise specified. 0.3 100 VI = 10.8 V 95 0.2 VI = 12 V Output Regulation - % Efficiency - % VI = 15 V 90 VI = 18 V 85 VI = 19.8 V 80 0.1 0 -0.1 -0.2 -0.3 75 0 0.5 1 1.5 2 2.5 IO - Output Current - A 3 3.5 Figure 13. Efficiency vs. Output Current 0 0.5 1 2 1.5 2.5 3 IO - Output Current - A Figure 14. Output Regulation % vs. Output Current 0.1 VIN = 100 mV/Div (AC Coupled) 0.08 0.06 IO = 3 A Input Regulation - % 0.04 IO = 1.5 A 0.02 0 PH = 5 V/Div -0.02 IO = 0 A -0.04 -0.06 -0.08 -0.1 10.8 13.8 16.8 VI - Input Voltage - V 19.8 Figure 15. Input Regulation % vs. Input Voltage t -Time - 500 ns/Div Figure 16. Input Voltage Ripple and PH Node, Io = 3 A. 17 TPS5430 www.ti.com SLVS632 – JANUARY 2006 VOUT = 50 mV/Div (AC Coupled) VOUT = 20 mV/Div (AC Coupled) PH = 5 V/Div IOUT = 1 A /Div t - Time = 200 ms/Div t - Time = 500 ns/Div Figure 17. Output Voltage Ripple and PH Node, Io = 3 A Figure 18. Transient Response, Io Step 0.75 to 2.25 A. VIN = 5 V/Div VOUT = 2 V/Div t - Time = 2 ms/Div Figure 19. Startup Waveform, Vin and Vout. 18 PACKAGE OPTION ADDENDUM www.ti.com 10-Feb-2006 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Drawing Pins Package Eco Plan (2) Qty TPS5430DDA ACTIVE SO Power PAD DDA 8 75 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM TPS5430DDAG4 ACTIVE SO Power PAD DDA 8 75 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM TPS5430DDAR ACTIVE SO Power PAD DDA 8 2500 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM TPS5430DDARG4 ACTIVE SO Power PAD DDA 8 2500 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM Lead/Ball Finish MSL Peak Temp (3) (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. 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