TI TPS54233DR

TPS54233
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2A, 28V INPUT, STEP DOWN SWIFT™ DC/DC CONVERTER WITH ECO-MODE™
FEATURES
1
• 3.5V to 28V Input Voltage Range
• Adjustable Output Voltage Down to 0.8V
• Integrated 80 mΩ High Side MOSFET Supports
up to 2A Continuous Output Current
• High Efficiency at Light Loads with a Pulse
Skipping Eco-mode™
• Fixed 300kHz Switching Frequency
• Typical 1µA Shutdown Quiescent Current
• Adjustable Slow Start Limits Inrush Currents
• Programmable UVLO Threshold
• Overvoltage Transient Protection
• Cycle by Cycle Current Limit, Frequency Fold
Back and Thermal Shutdown Protection
• Available in Easy-to-Use SOIC8 Package
• Supported by SwitcherPro™ Software Tool
(http://focus.ti.com/docs/toolsw/folders/print/s
witcherpro.html)
• For SWIFT™ Documentation, See the TI
Website at www.ti.com/swift
2
DESCRIPTION
The TPS54233 is a 28-V, 2-A non-synchronous buck
converter that integrates a low Rds(on) high side
MOSFET. To increase efficiency at light loads, a
pulse skipping Eco-mode™ feature is automatically
activated. Furthermore, the 1 µA shutdown supply
current allows the device to be used in battery
powered applications. Current mode control with
internal slope compensation simplifies the external
compensation calculations and reduces component
count while allowing the use of ceramic output
capacitors. A resistor divider programs the Hysteresis
of the input under-voltage lockout. An overvoltage
transient protection circuit limits voltage overshoots
during startup and transient conditions. A cycle by
cycle current limit scheme, frequency fold back and
thermal shutdown protect the device and the load in
the event of an overload condition. The TPS54233 is
available in an 8-pin SOIC package that has been
internally optimized to improve thermal performance.
APPLICATIONS
•
•
•
Consumer Applications such as Set-Top
Boxes, CPE Equipment, LCD Displays,
Peripherals, and Battery Chargers
Industrial and Car Audio Power Supplies
5V, 12V and 24V Distributed Power Systems
SIMPLIFIED SCHEMATIC
EFFICIENCY
100
Ren1
EN
VIN
Ren2
VO = 3.3 V
V = 12 V
VIN = 8 V IN
VIN = 15 V
95
VIN
CI
90
CBOOT
BOOT
LO
VOUT
PH
SS
COMP
D1
CO
RO1
Efficiency - %
TPS54233
85
VIN = 18 V
80
75
70
C1
CSS
C2
R3
65
VSENSE
GND
60
RO2
0
0.25 0.5 0.75 1 1.25 1.5 1.75
IO - Output Current - A
2
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
SWIFT, Eco-mode, SwitcherPro are trademarks of Texas Instruments.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2008, Texas Instruments Incorporated
TPS54233
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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
ORDERING INFORMATION (1)
(1)
(2)
TJ
PACKAGE
SWITCHING FREQUENCY
PART NUMBER (2)
–40°C to 150°C
8 pin SOIC
300 kHz
TPS54233D
For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI
web site at www.ti.com.
The D package is also available taped and reeled. Add an R suffix to the device type (i.e., TPS54233DR). See applications section of
data sheet for layout information.
ABSOLUTE MAXIMUM RATINGS (1)
over operating free-air temperature range (unless otherwise noted)
VALUE
Input Voltage
VIN
–0.3 to 30
EN
–0.3 to 5
BOOT
38
VSENSE
–0.3 to 3
COMP
–0.3 to 3
SS
–0.3 to 3
BOOT-PH
Output Voltage
Source Current
Sink Current
UNIT
V
8
PH
–0.6 to 30
V
PH (10 ns transient from ground to negative peak)
–5
EN
100
µA
BOOT
100
mA
VSENSE
10
µA
PH
6
A
VIN
6
A
COMP
100
SS
200
µA
Electrostatic Discharge (HBM)
2
Electrostatic Discharge (CDM)
500
V
Operating Junction Temperature
–40 to 150
°C
Storage Temperature
–65 to 150
°C
(1)
Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
PACKAGE DISSIPATION RATINGS (1)
(1)
(2)
(3)
2
kV
(2) (3)
PACKAGE
THERMAL IMPEDANCE JUNCTION TO
AMBIENT
PSEUDO THERMAL IMPEDANCE JUNCTION TO
TOP
SOIC8
100 °C/W
5 °C/W
Maximum power dissipation may be limited by overcurrent protection
Power rating at a specific ambient temperature TA should be determined with a junction temperature of 150°C. This is the point where
distortion starts to substantially increase. Thermal management of the PCB should strive to keep the junction temperature at or below
150°C for best performance and long-term reliability. See power dissipation estimate in application section of this data sheet for more
information.
Test board conditions:
a. 2 inches x 1.5 inches, 2 layers, thickness: 0.062 inch
b. 2-ounce copper traces located on the top and bottom of the PCB
c. 6 thermal vias located under the device package
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RECOMMENDED OPERATING CONDITIONS
over operating free-air temperature range (unless otherwise noted)
MIN
TYP
MAX
UNIT
Operating Input Voltage on (VIN pin)
3.5
28
V
Operating junction temperature, TJ
–40
150
°C
MAX
UNIT
ELECTRICAL CHARACTERISTICS
TJ = –40°C to 150°C, VIN = 3.5V to 28V (unless otherwise noted)
DESCRIPTION
TEST CONDITIONS
MIN
TYP
SUPPLY VOLTAGE (VIN PIN)
Internal undervoltage lockout threshold
Rising and Falling
Shutdown supply current
EN = 0V, VIN = 12V, –40°C to 85°C
3.5
V
1
4
µA
Operating – non switching supply current
VSENSE = 0.85 V
75
110
µA
Enable threshold
Rising and Falling
1.25
1.35
Input current
Enable threshold – 50 mV
-1
µA
Input current
Enable threshold + 50 mV
-4
µA
ENABLE AND UVLO (EN PIN)
V
VOLTAGE REFERENCE
Voltage reference
0.772
0.8
0.828
BOOT-PH = 3 V, VIN = 3.5 V
115
200
BOOT-PH = 6 V, VIN = 12 V
80
150
V
HIGH-SIDE MOSFET
On resistance
mΩ
ERROR AMPLIFIER
Error amplifier transconductance (gm)
–2 µA < ICOMP < 2 µA, V(COMP) = 1 V
Error amplifier DC gain (1)
VSENSE = 0.8 V
800
µmhos
V/V
Error amplifier unity gain bandwidth (1)
5 pF capacitance from COMP to GND pins
2.7
MHz
Error amplifier source/sink current
V(COMP) = 1.0 V, 100 mV overdrive
±7
µA
Switch current to COMP transconductance
VIN = 12 V
9
A/V
92
SWITCHING FREQUENCY
TPS54233 Switching Frequency
VIN = 12V
Minimum controllable on time
VIN = 12V, 25°C
Maximum controllable duty ratio (1)
BOOT-PH = 6 V
210
90
300
390
kHz
105
130
ns
93
%
100
mA
3.5
A
165
°C
PULSE SKIPPING ECO-MODE™
Pulse skipping Eco-mode™ switch current threshold
CURRENT LIMIT
Current limit threshold
VIN = 12 V
2.3
THERMAL SHUTDOWN
Thermal Shutdown
SLOW START (SS PIN)
Charge current
V(SS) = 0.4 V
2
µA
SS to VSENSE matching
V(SS) = 0.4 V
10
mV
(1)
Specified by design
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DEVICE INFORMATION
PIN ASSIGNMENTS
BOOT
1
8
PH
VIN
2
7
GND
EN
3
6
COMP
SS
4
5
VSENSE
TERMINAL FUNCTIONS
TERMINAL
DESCRIPTION
NAME
NO.
BOOT
1
A 0.1 µF bootstrap capacitor is required between BOOT and PH. If the voltage on this capacitor falls below the
minimum requirement, the high-side MOSFET is forced to switch off until the capacitor is refreshed.
VIN
2
Input supply voltage, 3.5 V to 28 V.
EN
3
Enable pin. Pull below 1.25V to disable. Float to enable. Programming the input undervoltage lockout with two
resistors is recommended.
SS
4
Slow start pin. An external capacitor connected to this pin sets the output rise time.
VSENSE
5
Inverting node of the gm error amplifier.
COMP
6
Error amplifier output, and input to the PWM comparator. Connect frequency compensation components to this pin.
GND
7
Ground.
PH
8
The source of the internal high-side power MOSFET.
4
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FUNCTIONAL BLOCK DIAGRAM
EN
VIN
165 C
Thermal
Shutdown
1 mA
3 mA
Shutdown
Shutdown
Logic
1.25 V
Enable
Threshold
Enable
Comparator
Boot
Charge
™
ECO-MODE
Minimum Clamp
Boot
UVLO
BOOT
2.1V
Error
Amplifier
VSENSE
2 mA
PWM
Comparator
Gate
Drive
Logic
gm = 92 mA/V
DC gain = 800 V/V
BW = 2.7 MHz
Voltage
Reference
SS
2 kW
0.8 V
S
Shutdown
PWM
Latch
9 A/V
Current
Sense
R
80 mW
Q
S
Slope
Compensation
PH
Discharge
Logic
VSENSE
Frequency
Shift
Oscillator
GND
COMP
Maximum
Clamp
TYPICAL CHARACTERISTICS
CHARACTERIZATION CURVES
ON RESISTANCE
vs
JUNCTION TEMPERATURE
SHUTDOWN QUIESCENT CURRENT
vs
INPUT VOLTAGE
4
110
105
SWITCHING FREQUENCY
vs
JUNCTION TEMPERATURE
310
VIN = 12 V
VIN = 12 V
TJ = 150°C
95
90
85
80
75
fsw - Oscillator Frequency - kHz
Isd - Shutdown Current - mA
Rdson - On Resistance - mW
100
3
TJ = -40°C
2
TJ = 25°C
1
70
305
300
295
65
60
-50
0
-25
0
25
50
75
100
TJ - Junction Temperature - °C
125
150
Figure 1.
3
8
13
18
VI - Input Voltage - V
23
Figure 2.
28
290
-50
-25
0
25
50
75
100
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150
Figure 3.
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125
TJ - Junction Temperature - °C
5
TPS54233
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TYPICAL CHARACTERISTICS (continued)
MINIMUM CONTROLLABLE ON
TIME
vs
JUNCTION TEMPERATURE
VOLTAGE REFERENCE
vs
JUNCTION TEMPERATURE
0.8120
0.8060
0.8000
0.7940
0.7880
0.7820
-25
0
25
50
75
100
TJ - Junction Temperature - °C
125
130
120
110
-25
0
25
50
75
100
TJ - Junction Temperature - °C
Figure 4.
3.50
3.25
125
-25
0
150
25
50
75
100
TJ - Junction Temperature - °C
Figure 5.
SS CHARGE CURRENT
vs
JUNCTION TEMPERATURE
125
150
Figure 6.
CURRENT LIMIT THRESHOLD
vs
INPUT VOLTAGE
4
Current Limit Threshold - A
2.10
ISS - Slow Start Charge Current - mA
3.75
3
-50
100
-50
150
VIN = 12 V
VIN = 12 V
Minimum Controllable Duty Ratio - %
Tonmin - Minimum Controllable On Time - ns
0.8180
Vref - Voltage Reference - V
4
140
0.8240
0.7760
-50
MINIMUM CONTROLLABLE DUTY
RATIO
vs
JUNCTION TEMPERATURE
2
TJ = 25°C
TJ = -40°C
3.5
TJ = 150°C
1.90
-50
3
-25
0
25
50
75
100
125
150
3
8
TJ - Junction Temperature - °C
13
18
VI - Input Voltage - V
Figure 7.
23
28
Figure 8.
SUPPLEMENTAL APPLICATION CURVES
TYPICAL MAXIMUM OUTPUT
VOLTAGE
vs
INPUT VOLTAGE
MAXIMUM POWER DISSIPATION
vs
JUNCTION TEMPERATURE
30
150
20
TJ - Junction Temperature - °C
VO - Output Voltage - V
25
IO = 1 A
15
IO = 2 A
10
125
100
75
50
5
0
3
8
13
18
VI - Input Voltage - V
23
28
25
0
0.2
Figure 9.
6
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0.4
0.6
0.8
1
1.2
PD - Power Dissipation - W
Figure 10.
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TYPICAL CHARACTERISTICS (continued)
OVERVIEW
The TPS54233 is a 28-V, 2-A, step-down (buck) converter with an integrated high-side n-channel MOSFET. To
improve performance during line and load transients, the device implements a constant frequency, current mode
control which reduces output capacitance and simplifies external frequency compensation design. The
TPS54233 has a pre-set switching frequency of 300kHz.
The TPS54233 needs a minimum input voltage of 3.5V to operate normally. The EN pin has an internal pull-up
current source that can be used to adjust the input voltage under-voltage lockout (UVLO) with two external
resistors. In addition, the pull-up current provides a default condition when the EN pin is floating for the device to
operate. The operating current is 75 µA typically when not switching and under no load. When the device is
disabled, the supply current is 1µA typically.
The integrated 80 mΩ high-side MOSFET allows for high efficiency power supply designs with continuous output
currents up to 2A.
The TPS54233 reduces the external component count by integrating the boot recharge diode. The bias voltage
for the integrated high-side MOSFET is supplied by an external capacitor on the BOOT to PH pin. The boot
capacitor voltage is monitored by an UVLO circuit and will turn the high-side MOSFET off when the voltage falls
below a preset threshold of 2.1V typically. The output voltage can be stepped down to as low as the reference
voltage.
By adding an external capacitor, the slow start time of the TPS54233 can be adjustable which enables flexible
output filter selection.
To improve the efficiency at light load conditions, the TPS54233 enters a special pulse skipping Eco-modeTM
when the peak inductor current drops below 100mA typically.
The frequency foldback reduces the switching frequency during startup and over current conditions to help
control the inductor current. The thermal shut down gives the additional protection under fault conditions.
DETAILED DESCRIPTION
FIXED FREQUENCY PWM CONTROL
The TPS54233 uses a fixed frequency, peak current mode control. The internal switching frequency of the
TPS54233 is fixed at 300kHz.
ECO-MODETM
The TPS54233 is designed to operate in pulse skipping Eco-modeTM at light load currents to boost light load
efficiency. When the peak inductor current is lower than 100 mA typically, the COMP pin voltage falls to 0.5V
typically and the device enters Eco-modeTM . When the device is in Eco-modeTM, the COMP pin voltage is
clamped at 0.5V internally which prevents the high side integrated MOSFET from switching. The peak inductor
current must rise above 100mA for the COMP pin voltage to rise above 0.5V and exit Eco-modeTM. Since the
integrated current comparator catches the peak inductor current only, the average load current entering
Eco-modeTM varies with the applications and external output filters.
VOLTAGE REFERENCE (Vref)
The voltage reference system produces a ±2% initial accuracy voltage reference (±3.5% over temperature) by
scaling the output of a temperature stable bandgap circuit. The typical voltage reference is designed at 0.8V.
BOOTSTRAP VOLTAGE (BOOT)
The TPS54233 has an integrated boot regulator and requires a 0.1 µF ceramic capacitor between the BOOT and
PH pin to provide the gate drive voltage for the high-side MOSFET. A ceramic capacitor with an X7R or X5R
grade dielectric is recommended because of the stable characteristics over temperature and voltage. To improve
drop out, the TPS54233 is designed to operate at 100% duty cycle as long as the BOOT to PH pin voltage is
greater than 2.1V typically.
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ENABLE AND ADJUSTABLE INPUT UNDER-VOLTAGE LOCKOUT (VIN UVLO)
The EN pin has an internal pull-up current source that provides the default condition of the TPS54233 operating
when the EN pin floats.
The TPS54233 is disabled when the VIN pin voltage falls below internal VIN UVLO threshold. It is recommended
to use an external VIN UVLO to add Hysteresis unless VIN is greater than (VOUT + 2V). To adjust the VIN
UVLO with Hysteresis, use the external circuitry connected to the EN pin as shown in Figure 11. Once the EN
pin voltage exceeds 1.25V, an additional 3µA of hysteresis is added. Use Equation 1 and Equation 2 to calculate
the resistor values needed for the desired VIN UVLO threshold voltages. The VSTART is the input start threshold
voltage, the VSTOP is the input stop threshold voltage and the VEN is the enable threshold voltage of 1.25V. The
VSTOP should always be greater than 3.5V.
VIN
Ren1
1 mA
3 mA
+
EN
Ren2
1.25 V
-
Figure 11. Adjustable Input Under-Voltage Lockout
Ren1 =
VSTART - VSTOP
3 mA
(1)
VEN
Ren2 =
VSTART - VEN
+ 1 mA
Ren1
(2)
PROGRAMMABLE SLOW START USING SS PIN
It is highly recommended to program the slow start time externally because no slow start time is implemented
internally. The TPS54233 effectively uses the lower voltage of the internal voltage reference or the SS pin
voltage as the power supply’s reference voltage fed into the error amplifier and will regulate the output
accordingly. A capacitor (Css) on the SS pin to ground implements a slow start time. The TPS54233 has an
internal pull-up current source of 2µA that charges the external slow start capacitor. The equation for the slow
start time (10% to 90%) is shown in Equation 3 . The Vref is 0.8V and the Iss current is 2µA.
CSS (nF ) ´ Vref (V )
TSS (ms ) =
ISS (mA )
(3)
The slow start time should be set between 1ms to 10ms to ensure good start-up behavior. The slow start
capacitor should be no more than 27 nF.
If during normal operation, the input voltage drops below the VIN UVLO threshold, or the EN pin is pulled below
1.25 V, or a thermal shutdown event occurs, the TPS54233 stops switching.
ERROR AMPLIFIER
The TPS54233 has a transconductance amplifier for the error amplifier. The error amplifier compares the
VSENSE voltage to the internal effective voltage reference presented at the input of the error amplifier. The
transconductance of the error amplifier is 92 µA/V during normal operation. Frequency compensation
components are connected between the COMP pin and ground.
SLOPE COMPENSATION
In order to prevent the sub-harmonic oscillations when operating the device at duty cycles greater than 50%, the
TPS54233 adds a built-in slope compensation which is a compensating ramp to the switch current signal.
8
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CURRENT MODE COMPENSATION DESIGN
To simplify design efforts using the TPS54233, the typical designs for common applications are listed in Table 1.
For designs using ceramic output capacitors, proper derating of ceramic output capacitance is recommended
when doing the stability analysis. This is because the actual ceramic capacitance drops considerably from the
nominal value when the applied voltage increases. Advanced users may refer to the Step by Step Design
Procedure in the Application Information section for the detailed guidelines or use SwitcherPro™ Software tool
(http://focus.ti.com/docs/toolsw/folders/print/switcherpro.html).
Table 1. Typical Designs (Referring to Simplified Schematic on page 1)
VIN
(V)
VOUT
(V)
Fsw
(kHz)
Lo
(µH)
Co
RO1
(kΩ)
RO2
(kΩ)
C2
(pF)
C1
(pF)
R3
(kΩ)
12
5
300
22
12
3.3
300
15
Ceramic 47 µF
10
1.91
68
1800
21
Ceramic 47µF
10.2
3.24
47
4700
12
1.8
300
10
21
Ceramic 100 µF x 2
10
8.06
100
4700
21
12
0.9
300
12
5
300
6.8
Ceramic 100 µFx2
10
80.6
100
4700
21
22
Aluminum 330 µF/160 mΩ
10
1.91
56
220
40.2
12
3.3
300
15
Aluminum 470 µF/160 mΩ
10.2
3.24
220
220
30.9
12
1.8
12
0.9
300
10
SP 220 µF/12 mΩ
10
8.06
100
4700
40.2
300
6.8
SP 220 µF/12 mΩ
10
80.6
100
1800
21
OVERCURRENT PROTECTION AND FREQUENCY SHIFT
The TPS54233 implements current mode control that uses the COMP pin voltage to turn off the high-side
MOSFET on a cycle by cycle basis. Every cycle the switch current and the COMP pin voltage are compared;
when the peak inductor current intersects the COMP pin voltage, the high-side switch is turned off. During
overcurrent conditions that pull the output voltage low, the error amplifier responds by driving the COMP pin high,
causing the switch current to increase. The COMP pin has a maximum clamp internally, which limit the output
current.
The TPS54233 provides robust protection during short circuits. There is potential for overcurrent runaway in the
output inductor during a short circuit at the output. The TPS54233 solves this issue by increasing the off time
during short circuit conditions by lowering the switching frequency. The switching frequency is divided by 8, 4, 2,
and 1 as the voltage ramps from 0V to 0.8V on VSENSE pin. The relationship between the switching frequency
and the VSENSE pin voltage is shown in Table 2.
Table 2. Switching Frequency Conditions
SWITCHING FREQUENCY
VSENSE PIN VOLTAGE
300 kHz
VSENSE ≥ 0.6 V
300 kHz / 2
0.6 V > VSENSE ≥ 0.4 V
300 kHz / 4
0.4 V > VSENSE ≥ 0.2 V
300 kHz / 8
0.2 V > VSENSE
OVERVOLTAGE TRANSIENT PROTECTION
The TPS54233 incorporates an overvoltage transient protection (OVTP) circuit to minimize output voltage
overshoot when recovering from output fault conditions or strong unload transients. The OVTP circuit includes an
overvoltage comparator to compare the VSENSE pin voltage and internal thresholds. When the VSENSE pin
voltage goes above 109% × Vref, the high-side MOSFET will be forced off. When the VSENSE pin voltage falls
below 107% × Vref, the high-side MOSFET will be enabled again.
THERMAL SHUTDOWN
The device implements an internal thermal shutdown to protect itself if the junction temperature exceeds 165°C.
The thermal shutdown forces the device to stop switching when the junction temperature exceeds the thermal
trip threshold. Once the die temperature decreases below 165°C, the device reinitiates the power up sequence.
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R6
3.24 kW
R5
30.9 kW
R2
68.1 kW
C1
10 mF
Vin 8 - 18 V
R1
332 kW
C4
0.1 mF
C5
0.015 mF
B230A
D1
C6
220 pF
L1
15 mH
+
C7
220 pF
C9
470 mF
R5
10.2 kW
Vout 3.3 V,
Iout Max 2 A
APPLICATION INFORMATION
Figure 12. Typical Application Schematic
10
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STEP BY STEP DESIGN PROCEDURE
The following design procedure can be used to select component values for the TPS54233. Alternately, the
SwitcherPro™Software may be used to generate a complete design. The SwitcherPro™ Software uses an
iterative design procedure and accesses a comprehensive database of components when generating a design.
This section presents a simplified discussion of the design process.
To begin the design process a few parameters must be decided upon. The designer needs to know the following:
• Input voltage range
• Output voltage
• Input ripple voltage
• Output ripple voltage
• Output current rating
• Operating frequency
For this design example, use the following as the input parameters
Table 3. Design Parameters
DESIGN PARAMETER
EXAMPLE VALUE
Input voltage range
8 V to 18 V
Output voltage
3.3 V
Input ripple voltage
300 mV
Output ripple voltage
30 mV
Output current rating
2A
Operating Frequency
300 kHz
SWITCHING FREQUENCY
The switching frequency for the TPS54233 is fixed at 300 kHz.
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OUTPUT VOLTAGE SET POINT
The output voltage of the TPS54233 is externally adjustable using a resistor divider network. In the application
circuit of Figure 12, this divider network is comprised of R5 and R6. The relationship of the output voltage to the
resistor divider is given by Equation 4 and Equation 5:
R5 ´ VREF
R6 =
VOUT - VREF
(4)
é R5 ù
VOUT = VREF ´ ê
+1ú
ë R6 û
(5)
Choose R5 to be approximately 10.0 kΩ. Slightly increasing or decreasing R5 can result in closer output voltage
matching when using standard value resistors. In this design, R4 = 10.2 kΩ and R = 3.24 kΩ, resulting in a 3.31
V output voltage. The zero ohm resistor R4 is provided as a convenient place to break the control loop for
stability testing.
INPUT CAPACITORS
The TPS54233 requires an input decoupling capacitor and depending on the application, a bulk input capacitor.
The typical recommended value for the decoupling capacitor is 10 µF. A high-quality ceramic type X5R or X7R is
recommended. The voltage rating should be greater than the maximum input voltage. A smaller value may be
used as long as all other requirements are met; however 10 µF has been shown to work well in a wide variety of
circuits. Additionally, some bulk capacitance may be needed, especially if the TPS54233 circuit is not located
within about 2 inches from the input voltage source. The value for this capacitor is not critical but should be rated
to handle the maximum input voltage including ripple voltage, and should filter the output so that input ripple
voltage is acceptable. For this design two 4.7 µF capacitors are used for the input decoupling capacitor. They are
X7R dielectric rated for 50 V. The equivalent series resistance (ESR) is approximately 2mΩ, and the current
rating is 3 A. Additionally, a small 0.01 µF capacitor is included for high frequency filtering.
This input ripple voltage can be approximated by Equation 6
IOUT(MAX) ´ 0.25
+ IOUT(MAX) ´ ESRMAX
DVIN =
CBULK ´ fSW
(
)
(6)
Where IOUT(MAX) is the maximum load current, fSW is the switching frequency, CBULK is the bulk capacitor value
and ESRMAX is the maximum series resistance of the bulk capacitor.
The maximum RMS ripple current also needs to be checked. For worst case conditions, this can be
approximated by Equation 7
IOUT(MAX)
ICIN =
2
(7)
In this case, the input ripple voltage would be 143 mV and the RMS ripple current would be 1.5 A. It is also
important to note that the actual input voltage ripple will be greatly affected by parasitics associated with the
layout and the output impedance of the voltage source. The actual input voltage ripple for this circuit is shown in
Design Parameters and is larger than the calculated value. This measured value is still below the specified input
limit of 300 mV. The maximum voltage across the input capacitors would be VIN max plus ΔVIN/2. The chosen
bulk and bypass capacitors are each rated for 50 V and the ripple current capacity is greater than 3 A, both
providing ample margin. It is very important that the maximum ratings for voltage and current are not exceeded
under any circumstance.
OUTPUT FILTER COMPONENTS
Two components need to be selected for the output filter, L1 and C2. Since the TPS54233 is an externally
compensated device, a wide range of filter component types and values can be supported.
Inductor Selection
To calculate the minimum value of the output inductor, use Equation 8
12
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LMIN =
(VIN(MAX) - VOUT )
VOUT(MAX) ´
VIN(MAX) ´ KIND ´ IOUT ´ FSW
(8)
KIND is a coefficient that represents the amount of inductor ripple current relative to the maximum output current.
In general, this value is at the discretion of the designer; however, the following guidelines may be used. For
designs using low ESR output capacitors such as ceramics, a value as high as KIND = 0.3 may be used. When
using higher ESR output capacitors, KIND = 0.2 yields better results.
For this design example, use KIND = 0.3 and the minimum inductor value is calculated to be 5.7µH. For this
design, a large value was chosen: 6.8 µH.
For the output filter inductor, it is important that the RMS current and saturation current ratings not be exceeded.
The RMS inductor current can be found from Equation 9
2
IL(RMS) = IOUT(MAX)
(
)
æ V
ö
OUT ´ VIN(MAX) - VOUT
1
÷
+
´ ç
ç VIN(MAX) ´ LOUT ´ FSW ´ 0.7 ÷
12
è
ø
2
(9)
and the peak inductor current can be determined with Equation 10
IL(PK) = IOUT(MAX) +
VOUT ´
(VIN(MAX)
- VOUT
)
1.4 ´ VIN(MAX) ´ LOUT ´ FSW
(10)
For this design, the RMS inductor current is 3.02 A and the peak inductor current is 3.54 A. The chosen inductor
is a Sumida CDRH103-6R8 6.8 µH. It has a saturation current rating of 3.84 A and an RMS current rating of 3.60
A, meeting these requirements. Smaller or larger inductor values can be used depending on the amount of ripple
current the designer wishes to allow so long as the other design requirements are met. Larger value inductors
will have lower ac current and result in lower output voltage ripple, while smaller inductor values will increase ac
current and output voltage ripple. In general, inductor values for use with the TPS54233 are in the range of
6.8 µH to 47µH.
Capacitor Selection
The important design factors for the output capacitor are dc voltage rating, ripple current rating, and equivalent
series resistance (ESR). The dc voltage and ripple current ratings cannot be exceeded. The ESR is important
because along with the inductor current it determines the amount of output ripple voltage. The actual value of the
output capacitor is not critical, but some practical limits do exist. Consider the relationship between the desired
closed loop crossover frequency of the design and LC corner frequency of the output filter. In general, it is
desirable to keep the closed loop crossover frequency at less than 1/5 of the switching frequency. With high
switching frequencies such as the 300-kHz frequency of this design, internal circuit limitations of the TPS54233
limit the practical maximum crossover frequency to about 25 kHz. In general, the closed loop crossover
frequency should be higher than the corner frequency determined by the load impedance and the output
capacitor. This limits the minimum capacitor value for the output filter to:
CO _ min = 1 /(2 ´ p ´ RO ´ FCO _ max )
(11)
Where RO is the output load impedance (VO/IO) and fCO is the desired crossover frequency. For a desired
maximum crossover of 25 kHz the minimum value for the output capacitor is around 3.8µF. This may not satisfy
the output ripple voltage requirement. The output ripple voltage consists of two components; the voltage change
due to the charge and discharge of the output filter capacitance and the voltage change due to the ripple current
times the ESR of the output filter capacitor. The output ripple voltage can be estimated by:
é ( D - 0 .5 )
ù
+ R ESR ú
V O PP = I LPP ê
ë 4 ´ F SW ´ C O
û
(12)
Where NC is the number of output capacitors in parallel.
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The maximum ESR of the output capacitor is determined by the amount of allowable output ripple as specified in
the initial design parameters; so the maximum specified ESR as listed in the capacitor data sheet is given by
Equation 13:
ESRmax =
VOPPMAX
ILPP
-
(D
- 0.5)
4 ´ FSW ´ CO
(13)
Where ΔVp-p is the desired peak-to-peak output ripple.
For this design example, a single 470-µF aluminum electrolytic output capacitor is chosen for C9. This is a
Panasonic, EEVFK1A471P rated at 10 V with a maximum ESR of 160mΩ and a ripple current rating of 600 mA.
The maximum RMS output ripple current can be calculated using Equation 14
æ VOUT × VIN(MAX) - VOUT ö
1
÷
ICOUT(RMS) =
× ç
ç VIN(MAX) × LOUT × FSW × NC ÷
12
è
ø
(
)
(14)
The calculated total RMS ripple current is 216 mA and the maximum total ESR required is 43 mΩ. These output
capacitors exceed the requirements by a wide margin and will result in a reliable, high-performance design. The
selected output capacitor must be rated for a voltage greater than the desired output voltage plus = the ripple
voltage. Any derating amount must also be included.
Other capacitor types work well with the TPS54233, depending on the needs of the application.
COMPENSATION COMPONENTS
The external compensation used with the TPS54233 allows for a wide range of output filter configurations. A
large range of capacitor values and types of dielectric are supported. The design example uses ceramic X5R
dielectric output capacitors, but other types are supported.
A Type II compensation scheme is recommended for the TPS54233. The compensation components are chosen
to set the desired closed loop cross over frequency and phase margin for output filter components. The type II
compensation has the following characteristics; a dc gain component, a low frequency pole, and a mid frequency
zero / pole pair.
The dc gain is determined by Equation 15:
Vggm ´ VREF
GDC =
VO
(15)
Where:
Vggm = 800
VREF = 0.8 V
The low-frequency pole is determined by Equation 16:
VPO = 1/ (2 ´ p ´ ROO ´ CZ )
(16)
The mid-frequency zero is determined by Equation 17:
FZ1 = 1/ (2 ´ p ´ R Z ´ CZ )
(17)
And, the mid-frequency pole is given by Equation 18:
FP1 = 1/ (2 ´ p ´ R Z ´ CP )
(18)
The first step is to choose the closed loop crossover frequency. In general, the closed-loop crossover frequency
should be less than 1/8 of the minimum operating frequency, but for the TPS54233it is recommended that the
maximum closed loop crossover frequency be not greater than 25 kHz. Next, the required gain and phase boost
of the crossover network needs to be calculated. By definition, the gain of the compensation network must be the
inverse of the gain of the modulator and output filter. For this design example, where the ESR zero is less than
the closed loop crossover frequency, the gain of the modulator and output filter can be approximated by
Equation 19:
14
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æ RO ö
æ RO ö
Gain = 20 log ç
÷ - 20 log ç
÷
è RSENSE ø
è RESR ø
(19)
Where:
RSENSE = 1Ω/9
RO = VO/IO
RESR = Equivalent series resistance of the output capacitor
The phase loss is given by Equation 20:
PL = a tan (2 ´ p ´ FCO ´ RESR ´ CO ) - a tan (2 ´ p ´ FCO ´ RO ´ CO )
(20)
Where:
RESR = Equivalent series resistance of the output capacitor
RO = VO/IO
Now that the phase loss is known the required amount of phase boost to meet the phase margin requirement
can be determined. The required phase boost is given by Equation 21:
PB = (PM - 90 deg ) - PL
(21)
Where PM = the desired phase margin.
A zero / pole pair of the compensation network will be placed symmetrically around the intended closed loop
frequency to provide maximum phase boost at the crossover point. The amount of separation can be determined
by Equation 22 and the resultant zero and pole frequencies are given by Equation 23 and Equation 24
ö
æ PB
k = tanç
+ 45 deg ÷
ø
è 2
FZ 1 =
(22)
FCO
k
(23)
FP1 = FCO ´ k
(24)
The low-frequency pole is set so that the gain at the crossover frequency is equal to the inverse of the gain of the
modulator and output filter. Due to the relationships established by the pole and zero relationships, the value of
RZ can be derived directly by Equation 25 :
RZ =
VO ´ ROA ´ 0.98
GMCOMP ´ Vggm ´ VREF ´ RESR
(25)
Where:
VO = Output voltage
ROA = 8.696 MΩ
GMCOMP = 9 A/V
Vggm = 800
VREF = 0.8 V
RESR = Equivalent series resistance of the output capacitor
With RZ known, CZ and CP can be calculated using Equation 26 and Equation 27:
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CZ =
CP =
1
2 ´ p ´ FZ 1 ´ Rz
(26)
1
2 ´ p ´ FP1 ´ Rz
(27)
For this design, a singe 470-µF output capacitor is used. The ESR is approximately .160 Ω. The desired closed
loop crossover frequency is 22000 Hz.
Using Equation 19 and Equation 20, the output stage gain and phase loss are equivalent as:
Gain = –3.114 dB
and
PL - –4.96 degrees
For 60 degrees of phase margin, Equation 21 requires no additional phase boost, so K can be set equal to 1.
Equation 22, Equation 23, and Equation 24 are used to find the zero and pole frequencies of:
FZ1 = 22000 Hz
And
FP1 = 22000 Hz
RZ, CZ, and CP are calculated using Equation 25, Equation 26, and Equation 27:
Rz =
Cz =
Cp =
2.5 ´ 8.696 ´ 106 ´ 0.98
= 30.5 kW
9 ´ 800 ´ 0.8 ´ 0.160
(28)
1
= 237 pF
2 ´ p ´ 22000 ´ 30500
(29)
1
= 237 pF
2 ´ p ´ 22000 ´ 30500
(30)
Using standard values for R3, C6, and C7 in the application schematic of Figure 12:
R3 = 30.9 kΩ
C6 = 220 pF
C7 = 220 pF
The measured overall loop response for the circuit is given in Figure 20. Note that the actual closed loop
crossover frequency is higher than intended at about 25 kHz. This is primarily due to variation in the actual
values of the output filter components and tolerance variation of the internal feed-forward gain circuitry. Overall
the design has greater than 60 degrees of phase margin and will be completely stable over all combinations of
line and load variability.
BOOTSTRAP CAPACITOR
Every TPS54233 design requires a bootstrap capacitor, C4. The bootstrap capacitor must be 0.1 µF. The
bootstrap capacitor is located between the PH pins and BOOT pin. The bootstrap capacitor should be a
high-quality ceramic type with X7R or X5R grade dielectric for temperature stability.
16
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CATCH DIODE
The TPS54233 is designed to operate using an external catch diode between PH and GND. The selected diode
must meet the absolute maximum ratings for the application: Reverse voltage must be higher than the maximum
voltage at the PH pin, which is VINMAX + 0.5 V. Peak current must be greater than IOUTMAX plus on half the
peak to peak inductor current. Forward voltage drop should be small for higher efficiencies. It is important to note
that the catch diode conduction time is typically longer than the high-side FET on time, so attention paid to diode
parameters can make a marked improvement in overall efficiency. Additionally, check that the device chosen is
capable of dissipating the power losses. For this design, a Diodes, Inc. B340A is chosen, with a reverse voltage
of 40 V, forward current of 3 A, and a forward voltage drop of 0.5 V.
OUTPUT VOLTAGE LIMITATIONS
Due to the internal design of the TPS54233, there are both upper and lower output voltage limits for any given
input voltage. The upper limit of the output voltage set point is constrained by the maximum duty cycle of 91%
and is given by Equation 31:
VOmax = 0.91 ×
((VIN min
- IO max × RDSon max ) + VD ) - (IO max × RL ) - VD
(31)
Where:
VIN min = Minimum input voltage
IO max = Maximum load current
VD = Catch diode forward voltage
RL = Output inductor series resistance
The equation assumes maximum on resistance for the internal high-side FET.
The lower limit is constrained by the minimum controllable on time which may be as high as 160 ns. The
approximate minimum output voltage for a given input voltage and minimum load current is given by Equation 32:
VOmin = 0.051 ´
((VIN max
- IOmin ´ Rin ) + VD ) - (IO min ´ RL
)
- VD
(32)
Where:
VIN max = Maximum input voltage
IO min = Minimum load current
VD = Catch diode forward voltage
RL = Output inductor series resistance
This equation assumes nominal on-resistance for the high-side FET and accounts for worst case variation of
operating frequency set point. Any design operating near the operational limits of the device should be carefully
checked to assure proper functionality.
POWER DISSIPATION ESTIMATE
The following formulas show how to estimate the device power dissipation under continuous conduction mode
operations. They should not be used if the device is working in the discontinuous conduction mode (DCM) or
pulse skipping Eco-modeTM.
The device power dissipation includes:
1) Conduction loss: Pcon = IOUT2 x Rds(on) x VOUT/VIN
2) Switching loss: Psw = 0.5 x 10-9 x VIN2 x IOUT x Fsw
3) Gate charge loss: Pgc = 22.8 x 10-9 x Fsw
4) Quiescent current loss: Pq = 0.075 x 10-3 x VIN
Where:
IOUT is the output current (A).
Rds(on) is the on-resistance of the high-side MOSFET (Ω).
VOUT is the output voltage (V).
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VIN is the input voltage (V).
Fsw is the switching frequency (Hz).
So
Ptot = Pcon + Psw + Pgc + Pq
For given TA , TJ = TA + Rth x Ptot.
For given TJMAX = 150°C, TAMAX = TJMAX– Rth x Ptot.
Where:
Ptot is the total device power dissipation (W).
TA is the ambient temperature (°C).
TJ is the junction temperature (°C) .
Rth is the thermal resistance of the package (°C/W).
TJMAX is maximum junction temperature (°C).
TAMAX is maximum ambient temperature (°C).
PCB LAYOUT
The VIN pin should be bypassed to ground with a low ESR ceramic bypass capacitor. Care should be taken to
minimize the loop area formed by the bypass capacitor connections, the VIN pin, and the anode of the catch
diode. The typical recommended bypass capacitance is 10-µF ceramic with a X5R or X7R dielectric and the
optimum placement is closest to the VIN pins and the source of the anode of the catch diode. See Figure 13 for
a PCB layout example. The GND D pin should be tied to the PCB ground plane at the pin of the IC. The source
of the low-side MOSFET should be connected directly to the top side PCB ground area used to tie together the
ground sides of the input and output capacitors as well as the anode of the catch diode. The PH pin should be
routed to the cathode of the catch diode and to the output inductor. Since the PH connection is the switching
node, the catch diode and output inductor should be located very close to the PH pins, and the area of the PCB
conductor minimized to prevent excessive capacitive coupling. For operation at full rated load, the top side
ground area must provide adequate heat dissipating area. The TPS54233 uses a fused lead frame so that the
GND pin acts as a conductive path for heat dissipation from the die. Many applications have larger areas of
internal or back side ground plane available, and the top side ground area can be connected to these areas
using multiple vias under or adjacent to the device to help dissipate heat. The additional external components
can be placed approximately as shown. It may be possible to obtain acceptable performance with alternate
layout schemes, however this layout has been shown to produce good results and is intended as a guideline.
18
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OUTPUT
FILTER
CAPACITOR
TOPSIDE
GROUND
AREA
Route BOOT CAPACITOR
trace on other layer to provide
wide path for topside ground
Vout
Feedback Trace
OUTPUT
INDUCTOR
CATCH
DIODE
PH
INPUT
BYPASS
CAPACITOR
BOOT
Vin
UVLO
RESISTOR
DIVIDER
VIN
GND
EN
COMP
SS
VSENSE
SLOW START
CAPACITOR
Thermal VIA
BOOT
CAPACITOR
PH
COMPENSATION
NETWORK
RESISTOR
DIVIDER
Signal VIA
Figure 13. TPS54233 Board Layout
Estimated Circuit Area
The estimated printed circuit board area for the components used in the design of Figure 12 is 0.72 in2. This area
does not include test points or connectors.
ELECTROMAGNETIC INTERFERENCE (EMI) CONSIDERATIONS
As EMI becomes a rising concern in more and more applications, the internal design of the TPS54233 takes
measures to reduce the EMI. The high-side MOSFET gate drive is designed to reduce the PH pin voltage
ringing. The internal IC rails are isolated to decrease the noise sensitivity. A package bond wire scheme is used
to lower the parasitics effects.
To achieve the best EMI performance, external component selection and board layout are equally important.
Follow the Step by Step Design Procedure above to prevent potential EMI issues.
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APPLICATION CURVES
100
100
VO = 3.3 V
95
VIN = 8 V
VIN = 12 V
95
VIN = 15 V
VIN = 18 V
90
VIN = 12 V
VIN = 15 V
90
Efficiency - %
Efficiency - %
85
85
VIN = 18 V
80
75
80
75
VIN = 18 V
70
65
70
60
65
55
50
60
0
0.25
0.5 0.75
1
1.25 1.5
IO - Output Current - A
1.75
2
0
20
Figure 14. TPS54233 Efficiency
0.05
0.025
0.04
0.020
IO = 1 A
0.015
0.02
VIN = 18 V
Output Regulation - %
Output Regulation - %
60 80 100 120 140 160 180 200
IO - Output Current - mA
Figure 15. TPS54233 Low Current Efficiency
0.03
VIN = 8 V
VIN = 12 V
0.01
0
-0.01
VIN = 15 V
-0.02
0.010
0.005
0
-0.005
-0.010
-0.015
-0.03
-0.020
-0.025
-0.04
0
0.2 0.4
0.6 0.8 1 1.2 1.4
IO - Output Current - A
1.6 1.8
2
8
Figure 16. TPS54233 Load Regulation
20
40
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9
10
11 12 13 14 15
VI - Input Voltage -V
16
17
18
Figure 17. TPS54233 Line Regulation
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180
60
Gain
Phase
VOUT
IOUT
-180
-60
10
1M
f - Frequency - Hz
Figure 18. TPS54233 Transient Response
Figure 19. TPS54233 Loop Response
VOUT
VIN
PH
PH
Figure 20. TPS54233 Input Ripple
Figure 21. TPS54233 Output Ripple
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VIN
ENA
VOUT
VOUT
Figure 22. TPS54233 Start Up
Figure 23. TPS54233 Start-up Relative to Enable
VOUT
PH
Figure 24. TPS54233 Eco-mode™ Operation
22
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PACKAGE MATERIALS INFORMATION
www.ti.com
6-Nov-2008
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
TPS54233DR
Package Package Pins
Type Drawing
SOIC
D
8
SPQ
Reel
Reel
Diameter Width
(mm) W1 (mm)
2500
330.0
12.4
Pack Materials-Page 1
A0 (mm)
B0 (mm)
K0 (mm)
P1
(mm)
6.4
5.2
2.1
8.0
W
Pin1
(mm) Quadrant
12.0
Q1
PACKAGE MATERIALS INFORMATION
www.ti.com
6-Nov-2008
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
TPS54233DR
SOIC
D
8
2500
346.0
346.0
29.0
Pack Materials-Page 2
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