TPS62290 www.ti.com SLVS764 – JUNE 2007 1-A Step Down Converter in 2 x 2 SON Package FEATURES DESCRIPTION High Efficiency Step Down Converter Output Current up to 1000 mA VIN Range From 2.3 V to 6 V 2.25 MHz Fixed Frequency Operation Power Save Mode at Light Load Currents Output Voltage Accuracy in PWM mode ±1.5% Typ. 15-µA Quiescent Current 100% Duty Cycle for Lowest Dropout Voltage Positioning at Light Loads Available in a 2 × 2 × 0,8 mm SON Package APPLICATIONS • • • • • • TPS62290DRV VIN CIN GND MODE VOUT 1.8 V, 1000 mA L1 2.2 mH R1 360 kW 10 mF The TPS62290 operates at 2.25-MHz fixed switching frequency and enters Power Save Mode operation at light load currents to maintain high efficiency over the entire load current range. The TPS62290 is available in a 2 mm × 2 mm 6 pin SON package. SW EN With an input voltage range of 2.3 V to 6 V, the device supports batteries with extended voltage range and are ideal to power portable applications like mobile phones and other portable equipment. The Power Save Mode is optimized for low output voltage ripple. For low noise applications, the device can be forced into fixed frequency PWM mode by pulling the MODE pin high. In the shutdown mode, the current consumption is reduced to less than 1 µA. TPS62290 allows the use of small inductors and capacitors to achieve a small solution size. Cell Phones, Smart-phones WLAN PDAs, Pocket PCs Low Power DSP Supply Portable Media Players POL VIN 2.7 V to 6 V The TPS62290 device is a high efficient synchronous step down dc-dc converter optimized for battery powered portable applications. It provides up to 1000-mA output current from a single Li-Ion cell. C1 22 pF 100 VIN = 4.2 V 90 VIN = 3.8 V COUT 10 mF 80 FB R2 180 kW Efficiency - % • • • • • • • • • • 70 VIN = 5 V VIN = 4.5 V 60 50 40 VOUT = 3.3 V, MODE = GND, L = 2.2 mH 30 0.00001 0.0001 0.001 0.01 0.1 IO - Output Current - A 1 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2007, Texas Instruments Incorporated TPS62290 www.ti.com SLVS764 – JUNE 2007 This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. ORDERING INFORMATION TA PART NUMBER (1) OUTPUT VOLTAGE (2) PACKAGE (3) PACKAGE DESIGNATOR ORDERING PACKAGE MARKING –40°C to 85°C TPS62290 adjustable SON 2 x 2 DRV TPS62290DRV BYN (1) (2) (3) The DRV package is available in tape on reel. Add R suffix to order quantities of 3000 parts per reel. Contact TI for other fixed output voltage options For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI website at www.ti.com. ABSOLUTE MAXIMUM RATINGS over operating free-air temperature range (unless otherwise noted) (1) VALUE VI Input voltage range (2) –0.3 to 7 –0.3 to VIN +0.3, ≤ 7 Voltage range at EN, MODE Voltage on SW ESD rating (3) Internally limited HBM Human body model 2 CDM Charge device model 1 Machine model A kV 200 V TJ Maximum operating junction temperature –40 to 125 Tstg Storage temperature range –65 to 150 (2) (3) V –0.3 to 7 Peak output current (1) UNIT °C Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings only and functional operation of the device at these or any other conditions beyond those indicated under recommended operating conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. All voltage values are with respect to network ground terminal. The human body model is a 100-pF capacitor discharged through a 1.5-kΩ resistor into each pin. The machine model is a 200-pF capacitor discharged directly into each pin. DISSIPATION RATINGS PACKAGE RθJA POWER RATING FOR TA≤ 25°C DERATING FACTOR ABOVE TA = 25°C DRV 76°C/W 1300 mW 13 mW/°C RECOMMENDED OPERATING CONDITIONS MIN VIN 2 Supply voltage NOM MAX 2 6 UNIT V Output voltage range for adjustable voltage 0.6 VIN V TA Operating ambient temperature –40 85 °C TJ Operating junction temperature –40 125 °C Submit Documentation Feedback TPS62290 www.ti.com SLVS764 – JUNE 2007 ELECTRICAL CHARACTERISTICS Over full operating ambient temperature range, typical values are at TA = 25°C. Unless otherwise noted, specifications apply for condition VIN = EN = 3.6V. External components CIN = 4.7µF 0603, COUT = 10µF 0603, L = 2.2µH, refer to parameter measurement information. PARAMETER TEST CONDITIONS MIN TYP MAX UNIT SUPPLY VI Input voltage range 2.3 6 VIN 2.7 V to 6 V IO Output current IQ Operating quiescent current ISD Shutdown current UVLO Undervoltage lockout threshold V 1000 VIN 2.5 V to 2.7 V 600 VIN 2.3 V to 2.5 V 300 mA IO = 0 mA, PFM mode enabled (MODE = GND) device not switching, See (1) 15 µA IO = 0 mA, switching with no load (MODE = VIN) PWM operation, VO = 1.8 V, VIN = 3V 3.8 mA EN = GND 0.1 Falling 1.85 Rising 1.95 1 µA V ENABLE, MODE VIH High level input voltage, EN, MODE 2.3 V ≤ VIN ≤ 6 V 1 VIN VIL Low level input voltage, EN, MODE 2.3 V ≤ VIN ≤ 6 V 0 0.4 II Input bias current, EN, MODE EN, MODE = GND or VIN 0.01 1 240 480 185 380 1.4 1.68 V V µA POWER SWITCH RDS(on) ILIMF TSD High side MOSFET on-resistance Low side MOSFET on-resistance VIN = VGS = 3.6 V, TA = 25°C Forward current limit MOSFET high-side and low side VIN = VGS = 3.6 V Thermal shutdown Increasing junction temperature 140 Thermal shutdown hysteresis Decreasing junction temperature 20 1.19 mΩ A °C OSCILLATOR fSW Oscillator frequency 2.3 V ≤ VIN ≤ 6 V 2.0 2.25 2.5 MHz VI V OUTPUT VO Adjustable output voltage range Vref Reference voltage 0.6 600 VFB(PWM) Feedback voltage MODE = VIN, PWM operation, 2.3 V ≤ VIN ≤ 6 V, See (2) VFB(PFM) Feedback voltage PFM mode MODE = GND, device in PFM mode, +1% voltage positioning active, See (1) Load regulation –1.5% 0 - 0.5 %/A 500 µs µs tStart Up Start-up time tRamp VO ramp-up time Time to ramp from 5% to 95% of VO 250 Leakage current into SW pin VI = 3.6 V, VI = VO = VSW, EN = GND, See (3) 0.1 (1) (2) (3) 1.5% 1% Time from active EN to reach 95% of VO Ilkg mV 1 µA In PFM mode, the internal reference voltage is set to typ. 1.01 × Vref . See the parameter measurement information. For VIN = VO + 1.0 V In fixed output voltage versions, the internal resistor divider network is disconnected from FB pin. Submit Documentation Feedback 3 TPS62290 www.ti.com SLVS764 – JUNE 2007 PIN ASSIGNMENTS DRV PACKAGE (TOP VIEW) SW MODE FB 1 2 3 D 6 PA 5 r we 4 Po GND VIN EN TERMINAL FUNCTIONS TERMINAL NAME I/O NO. DESCRIPTION VIN 5 PWR VIN power supply pin. GND 6 PWR GND supply pin EN 4 I SW 1 OUT This is the switch pin and is connected to the internal MOSFET switches. Connect the external inductor between this terminal and the output capacitor. FB 3 I Feedback Pin for the internal regulation loop. Connect the external resistor divider to this pin. In case of fixed output voltage option, connect this pin directly to the output capacitor MODE 2 I MODE pin = high forces the device to operate in fixed-frequency PWM mode. Mode pin = low enables the Power Save Mode with automatic transition from PFM mode to fixed-frequency PWM mode. This is the enable pin of the device. Pulling this pin to low forces the device into shutdown mode. Pulling this pin to high enables the device. This pin must be terminated. FUNCTIONAL BLOCK DIAGRAM VIN Current Limit Comparator VIN Undervoltage Lockout 1.8 V Thermal Shutdown Limit High Side EN Reference 0.6 V VREF FB PFM Comp . +1% Voltage positioning VREF + 1% Mode MODE Softstart VOUT RAMP CONTROL Error Amp Control Stage Gate Driver Anti Shoot-Through SW1 VREF Integrator FB FB Zero-Pole AMP. PWM Comp . Limit RI1 Low Side Int. Resistor Network RI3 RI..N Sawtooth Generator Current Limit Comparator 2.25 MHz Oscillator GND 4 Submit Documentation Feedback GND TPS62290 www.ti.com SLVS764 – JUNE 2007 PARAMETER MEASUREMENT INFORMATION TPS62290DRV VIN C IN 10 mF SW R1 EN GND L1 2 .2 mH V OUT C1 22 pF C OUT 10 mF FB MODE R2 L: LPS3015 2.2 mH, 110 mW CIN: GRM188R60J106M 10 mF Murata 0603 size COUT: GRM188R60J106M 10 mF Murata 0603 size TYPICAL CHARACTERISTICS Table 1. Table Of Graphs FIGURE Efficiency vs VO 1.8 V Power Save Mode Figure 1 Efficiency vs VO 1.8 V Forced Save Mode Figure 2 Efficiency vs VO 3.3 V Power Save Mode Figure 3 Efficiency vs VO 3.3 V Forced Save Mode Figure 4 VO ACCURACY Figure 5 PFM LOAD TRANSIENT Figure 6 PFM LINE TRANSIENT Figure 7 PWM LOAD TRANSIENT Figure 8 PWM LINE TRANSIENT Figure 9 TYPICAL OPERATION PFM MODE Figure 10 TYPICAL OPERATION PWM MODE Figure 11 Shutdown Current Into VIN vs Input Voltage, (TA = 85°C, TA = 25°C, TA = -40°C) Figure 12 Quiescent Current vs Input Voltage, (TA = 85°C, TA = 25°C, TA = -40°C) Figure 13 Static Drain-Source On-State Resistance vs Input Voltage, (TA = 85°C, TA = 25°C, TA = -40°C) Submit Documentation Feedback Figure 14 Figure 15 5 TPS62290 www.ti.com SLVS764 – JUNE 2007 EFFICIENCY (Power Save Mode) vs OUTPUT CURRENT EFFICIENCY (Forced PWM Mode) vs OUTPUT CURRENT 100 100 VIN = 2.7 V 90 L = 2.2 mH VIN = 3.3 V VIN = 3.3 V VIN = 4.5 V Efficiency - % Efficiency - % 80 VIN = 3.6 V 80 70 VIN = 5 V 60 50 30 0.01 VIN = 2.7 V VIN = 4.5 V 50 VIN = 3.6 V 20 0.1 100 10 1 IO - Output Current - mA 1000 100 10 IO - Output Current - mA 1 Figure 1. Figure 2. EFFICIENCY (Power Save Mode) vs OUTPUT CURRENT EFFICIENCY (Forced PWM Mode) vs OUTPUT CURRENT 1000 100 VIN = 4.2 V VIN = 3.8 V 80 VIN = 3.8 V 80 VIN = 5 V VIN = 5 V 70 Efficiency - % VIN = 4.5 V 70 60 60 VIN = 4.5 V 50 40 30 VOUT = 3.3 V, MODE = GND, L = 2.2 mH 40 30 0.01 VIN = 4.2 V 90 50 20 VOUT = 3.3 V, MODE = VIN, 10 L = 2.2 mH 0 0.1 100 10 1 IO - Output Current - mA 1000 1 Figure 3. 6 VIN = 5 V 60 30 100 Efficiency - % 70 40 VOUT = 1.8 V, MODE = GND, L = 2.2 mH 40 90 VOUT = 1.8 V, MODE = VIN, 90 100 10 IO - Output Current - mA Figure 4. Submit Documentation Feedback 1000 TPS62290 www.ti.com SLVS764 – JUNE 2007 OUTPUT VOLTAGE ACCURACY vs OUTPUT CURRENT PFM LOAD TRANSIENT 1.854 SW 2V/Div MODE = VIN, L = 2.2 mH 1.836 DC Output Voltage - V VIN = 2.7 V, TA = -40°C VOUT 50 mV/Div VIN = 3.6 V, TA = -40°C 1.818 VIN 3.6 V, VOUT 1.8 V, IOUT 50 mA to 250 mA, 250 mA MODE = GND VIN = 4.5 V, TA = -40°C 1.8 1.782 VIN = 2.7 V, TA = 25°C VIN = 4.5 V, TA = 85°C VIN = 3.6 V, TA = 25°C 1.764 1.746 0.01 VIN = 3.6 V, TA = 85°C VIN = 4.5 V, TA = 25°C 0.1 IOUT 200 mA/Div 50 mA VIN = 2.7 V, TA = 85°C 1 10 100 IO - Output Current - mA Icoil 500 mA/Div 1000 Time Base - 20 ms/Div Figure 5. Figure 6. PFM LINE TRANSIENT PWM LOAD TRANSIENT VIN 3.6 V, VOUT 1.8 V, IOUT 300 mA to 800 mA, MODE = GND VOUT 100 mV/Div IOUT 500 mA/Div VIN 3.6 V to 4.2 V 500 mV/Div 800 mA 300 mA VOUT = 1.8 V, 50 mV/Div, IOUT = 50 mA, MODE = GND Icoil 500 mA/Div Time Base - 100 ms/Div Time Base - 20 ms/Div Figure 7. Figure 8. Submit Documentation Feedback 7 TPS62290 www.ti.com SLVS764 – JUNE 2007 TYPICAL OPERATION vs PFM MODE PWM LINE TRANSIENT VIN 3.6 V to 4.2 V, 500 mV/Div VOUT 20 mV/Div VIN 3.6 V, VOUT 1.8 V, IOUT 10 mA, L 2.2 mH, COUT 10 mF 0603 SW 2 V/Div VOUT = 1.8 V, 50 mV/Div, IOUT = 250 mA, MODE = GND Icoil 200 mA/Div Time Base - 100 ms/Div Time Base - 10 ms/Div Figure 9. Figure 10. TYPICAL OPERATION vs PWM MODE SHUTDOWN CURRENT INTO VIN vs INPUT VOLTAGE 0.8 VIN 3.6 V, VOUT 1.8 V, IOUT 150 mA, L 2.2 mH, COUT 10 mF 0603 SW 2 V/Div Icoil 200 mA/Div ISD - Shutdown Current Into VIN − mA VOUT 10 mV/Div EN = GND 0.7 0.6 o TA = 85 C 0.5 0.4 0.3 0.2 o o TA = 25 C TA = -40 C 0.1 0 Time Base - 10 ms/Div 2 2.5 3 3.5 4 4.5 VIN − Input Voltage − V Figure 11. 8 Figure 12. Submit Documentation Feedback 5 5.5 6 TPS62290 www.ti.com SLVS764 – JUNE 2007 QUIESCENT CURRENT vs INPUT VOLTAGE STATIC DRAIN-SOURCE ON-STATE RESISTANCE vs INPUT VOLTAGE MODE = GND, EN = VIN, Devise Not Switching TA = 85oC 16 o TA = 25 C 14 12 TA = -40oC 10 8 2 2.5 3 3.5 4 4.5 5 5.5 6 VIN − Input Voltage − V High Side Switching 0.7 0.6 o TA = 85 C 0.5 o TA = 25 C 0.4 0.3 0.2 TA = -40oC 0.1 0 2 2.5 3 3.5 4 4.5 5 VIN − Input Voltage − V Figure 13. Figure 14. STATIC DRAIN-SOURCE ON-STATE RESISTANCE vs INPUT VOLTAGE RDS(on) - Static Drain-Source On-State Resistance − W IQ - Quiescent Current − mA 18 0.8 RDS(on) - Static Drain-Source On-State Resistance − W 20 0.4 Low Side Switching 0.35 0.3 o TA = 85 C 0.25 o TA = 25 C 0.2 0.15 0.1 TA = -40oC 0.05 0 2 2.5 3 3.5 4 4.5 5 VIN − Input Voltage − V Figure 15. Submit Documentation Feedback 9 TPS62290 www.ti.com SLVS764 – JUNE 2007 DETAILED DESCRIPTION OPERATION The TPS62290 step down converter operates with typically 2.25-MHz fixed frequency pulse width modulation (PWM) at moderate to heavy load currents. At light load currents, the converter can automatically enter Power Save Mode and operates then in PFM mode. During PWM operation, the converter use a unique fast response voltage mode controller scheme with input voltage feed-forward to achieve good line and load regulation allowing the use of small ceramic input and output capacitors. At the beginning of each clock cycle initiated by the clock signal, the High Side MOSFET switch is turned on. The current flows now from the input capacitor via the High Side MOSFET switch through the inductor to the output capacitor and load. During this phase, the current ramps up until the PWM comparator trips and the control logic turns off the switch. The current limit comparator also turns off the switch if the current limit of the High Side MOSFET switch is exceeded. After a dead time preventing shoot through current, the Low Side MOSFET rectifier is turned on and the inductor current ramps down. The current flows now from the inductor to the output capacitor and to the load. It returns to the inductor through the Low Side MOSFET rectifier. The next cycle is initiated by the clock signal again turning off the Low Side MOSFET rectifier and turning on the on the High Side MOSFET switch. POWER SAVE MODE The Power Save Mode is enabled with MODE Pin set to low level. If the load current decreases, the converter will enter Power Save Mode operation automatically. During Power Save Mode the converter skips switching and operates with reduced frequency in PFM mode with a minimum quiescent current to maintain high efficiency. The converter will position the output voltage typically +1% above the nominal output voltage. This voltage positioning feature minimizes voltage drops caused by a sudden load step. The transition from PWM mode to PFM mode occurs once the inductor current in the Low Side MOSFET switch becomes zero, which indicates discontinuous conduction mode. During the Power Save Mode the output voltage is monitored with a PFM comparator. As the output voltage falls below the PFM comparator threshold of VOUT nominal +1%, the device starts a PFM current pulse. For this the High Side MOSFET switch will turn on and the inductor current ramps up. After the On-time expires, the switch is turned off and the Low Side MOSFET switch is turned on until the inductor current becomes zero. The converter effectively delivers a current to the output capacitor and the load. If the load is below the delivered current, the output voltage will rise. If the output voltage is equal or higher than the PFM comparator threshold, the device stops switching and enters a sleep mode with typical 15µA current consumption. If the output voltage is still below the PFM comparator threshold, a sequence of further PFM current pulses are generated until the PFM comparator threshold is reached. The converter starts switching again once the output voltage drops below the PFM comparator threshold. With a fast single threshold comparator, the output voltage ripple during PFM mode operation can be kept small. The PFM Pulse is time controlled, which allows to modify the charge transferred to the output capacitor by the value of the inductor. The resulting PFM output voltage ripple and PFM frequency depend in first order on the size of the output capacitor and the inductor value. Increasing output capacitor values and inductor values will minimize the output ripple. The PFM frequency decreases with smaller inductor values and increases with larger values. The PFM mode is left and PWM mode entered in case the output current can not longer be supported in PFM mode. The Power Save Mode can be disabled through the MODE pin set to high. The converter will then operate in fixed frequency PWM mode. Dynamic Voltage Positioning This feature reduces the voltage under/overshoots at load steps from light to heavy load and vice versa. It is active in Power Save Mode and regulates the output voltage 1% higher than the nominal value. This provides more headroom for both the voltage drop at a load step, and the voltage increase at a load throw-off. 10 Submit Documentation Feedback TPS62290 www.ti.com SLVS764 – JUNE 2007 DETAILED DESCRIPTION (continued) Output voltage Voltage Positioning Vout +1% PFM Comparator threshold Light load PFM Mode Vout (PWM) moderate to heavy load PWM Mode Figure 16. Power Save Mode Operation 100% Duty Cycle Low Dropout Operation The device starts to enter 100% duty cycle Mode once the input voltage comes close the nominal output voltage. To maintain the output voltage, the High Side MOSFET switch is turned on 100% for one or more cycles. With further decreasing VIN the High Side MOSFET switch is turned on completely. In this case, the converter offers a low input-to-output voltage difference. This is particularly useful in battery-powered applications to achieve longest operation time by taking full advantage of the whole battery voltage range. The minimum input voltage to maintain regulation depends on the load current and output voltage, and can be calculated as: VINmin = VOmax + IOmax × (RDS(on)max + RL) With: IOmax = maximum output current plus inductor ripple current RDS(on)max = maximum P-channel switch RDS(on). RL = DC resistance of the inductor VOmax = nominal output voltage plus maximum output voltage tolerance Undervoltage Lockout The undervoltage lockout circuit prevents the device from malfunctioning at low input voltages and from excessive discharge of the battery and disables the output stage of the converter. The undervoltage lockout threshold is typically 1.85V with falling VIN. MODE SELECTION The MODE pin allows mode selection between forced PWM mode and Power Save Mode. Connecting this pin to GND enables the Power Save Mode with automatic transition between PWM and PFM mode. Pulling the MODE pin high forces the converter to operate in fixed frequency PWM mode even at light load currents. This allows simple filtering of the switching frequency for noise sensitive applications. In this mode, the efficiency is lower compared to the power save mode during light loads. The condition of the MODE pin can be changed during operation and allows efficient power management by adjusting the operation mode of the converter to the specific system requirements. Submit Documentation Feedback 11 TPS62290 www.ti.com SLVS764 – JUNE 2007 DETAILED DESCRIPTION (continued) ENABLE The device is enabled setting EN pin to high. During the start up time tStart Up the internal circuits are settled. Afterwards, the device activates the soft start circuit. The EN input can be used to control power sequencing in a system with various dc/dc converters. The EN pin can be connected to the output of another converter, to drive the EN pin high and getting a sequencing of supply rails. With EN = GND, the device enters shutdown mode. In this mode, all circuits are disabled. In fixed output voltage versions, the internal resistor divider network is disconnected from FB pin. SOFT START The TPS62290 has an internal soft start circuit that controls the ramp up of the output voltage. The output voltage ramps up from 5% to 95% of its nominal value within typical 250µs. This limits the inrush current in the converter during ramp up and prevents possible input voltage drops when a battery or high impedance power source is used. The soft start circuit is enabled within the start up time tStart Up. SHORT-CIRCUIT PROTECTION The High Side and Low Side MOSFET switches are short-circuit protected with maximum switch current = ILIMF. The current in the switches is monitored by current limit comparators. Once the current in the High Side MOSFET switch exceeds the threshold of it's current limit comparator, it turns off and the Low Side MOSFET switch is activated to ramp down the current in the inductor and High Side MOSFET switch. The High Side MOSFET switch can only turn on again, once the current in the Low Side MOSFET switch has decreased below the threshold of its current limit comparator. THERMAL SHUTDOWN As soon as the junction temperature, TJ, exceeds 140°C (typical) the device goes into thermal shutdown. In this mode, the High Side and Low Side MOSFETs are turned-off. The device continues its operation when the junction temperature falls below the thermal shutdown hysteresis. 12 Submit Documentation Feedback TPS62290 www.ti.com SLVS764 – JUNE 2007 APPLICATION INFORMATION TPS62290DRV VIN 2.7 V to 6 V VIN CIN 2.2 mH SW R1 EN 10 mF GND L1 360 kW COUT FB 10 mF R2 MODE VOUT 1.8 V, 1000 mA C1 22 pF 180 kW Figure 17. TPS62290DRV Adjustable 1.8 V TPS62290DRV VIN 3.7 V to 6 V VIN CIN GND MODE 2.2 mH SW R1 EN 10 mF L1 820 kW VOUT 3.3 V, 1000 mA C1 22 pF COUT FB R2 10 mF 182 kW Figure 18. TPS62290DRV Adjustable 3.3 V OUTPUT VOLTAGE SETTING The output voltage can be calculated to: R V OUT + VREF 1) 1 R2 with an internal reference voltage VREF typical 0.6V. ǒ Ǔ To minimize the current through the feedback divider network, R2 should be 180 kΩ or 360 kΩ. The sum of R1 and R2 should not exceed ~1MΩ, to keep the network robust against noise. An external feed forward capacitor C1 is required for optimum load transient response. The value of C1 should be in the range between 22pF and 33pF. Route the FB line away from noise sources, such as the inductor or the SW line. OUTPUT FILTER DESIGN (INDUCTOR AND OUTPUT CAPACITOR) The TPS62260 is designed to operate with inductors in the range of 1.5µH to 4.7µH and with output capacitors in the range of 4.7µF to 22µF. The part is optimized for operation with a 2.2µH inductor and 10µF output capacitor. Larger or smaller inductor values can be used to optimize the performance of the device for specific operation conditions. For stable operation, the L and C values of the output filter may not fall below 1µH effective inductance and 3.5µF effective capacitance. Inductor Selection The inductor value has a direct effect on the ripple current. The selected inductor has to be rated for its dc resistance and saturation current. The inductor ripple current (∆IL) decreases with higher inductance and increases with higher VI or VO. The inductor selection has also impact on the output voltage ripple in PFM mode. Higher inductor values will lead to lower output voltage ripple and higher PFM frequency, lower inductor values will lead to a higher output voltage ripple but lower PFM frequency. Submit Documentation Feedback 13 TPS62290 www.ti.com SLVS764 – JUNE 2007 APPLICATION INFORMATION (continued) Equation 1 calculates the maximum inductor current under static load conditions. The saturation current of the inductor should be rated higher than the maximum inductor current as calculated with Equation 2. This is recommended because during heavy load transient the inductor current will rise above the calculated value. DI L + Vout 1 * Vout Vin ƒ L I Lmax + I outmax ) (1) DI L 2 (2) With: f = Switching Frequency (2.25MHz typical) L = Inductor Value ∆IL = Peak to Peak inductor ripple current ILmax = Maximum Inductor current A more conservative approach is to select the inductor current rating just for the maximum switch current of the corresponding converter. Accepting larger values of ripple current allows the use of low inductance values, but results in higher output voltage ripple, greater core losses, and lower output current capability. The total losses of the coil have a strong impact on the efficiency of the dc/dc conversion and consist of both the losses in the dc resistance (R(DC)) and the following frequency-dependent components: • The losses in the core material (magnetic hysteresis loss, especially at high switching frequencies) • Additional losses in the conductor from the skin effect (current displacement at high frequencies) • Magnetic field losses of the neighboring windings (proximity effect) • Radiation losses Table 2. List of Inductors DIMENSIONS [mm3] INDUCTOR TYPE SUPPLIER 3 × 3 × 1.5 LPS3015 Coilcraft Output Capacitor Selection The advanced fast-response voltage mode control scheme of the TPS62290 allows the use of tiny ceramic capacitors. Ceramic capacitors with low ESR values have the lowest output voltage ripple and are recommended. The output capacitor requires either an X7R or X5R dielectric. Y5V and Z5U dielectric capacitors, aside from their wide variation in capacitance over temperature, become resistive at high frequencies. At nominal load current, the device operates in PWM mode and the RMS ripple current is calculated as: 1 * Vout 1 Vin I RMSCout + Vout ƒ L 2 Ǹ3 (3) At nominal load current, the device operates in PWM mode and the overall output voltage ripple is the sum of the voltage spike caused by the output capacitor ESR plus the voltage ripple caused by charging and discharging the output capacitor: DVout + Vout 1 * Vout Vin L ƒ ǒ8 1 Cout ƒ Ǔ ) ESR (4) At light load currents the converter operates in Power Save Mode and the output voltage ripple is dependent on the output capacitor and inductor value. Larger output capacitor and inductor values minimize the voltage ripple in PFM mode and tighten dc output accuracy in PFM mode. 14 Submit Documentation Feedback TPS62290 www.ti.com SLVS764 – JUNE 2007 Input Capacitor Selection The buck converter has a natural pulsating input current; therefore, a low ESR input capacitor is required for best input voltage filtering and minimizing the interference with other circuits caused by high input voltage spikes. For most applications, a 10-µF ceramic capacitor is recommended. The input capacitor can be increased without any limit for better input voltage filtering. Take care when using only small ceramic input capacitors. When a ceramic capacitor is used at the input and the power is being supplied through long wires, such as from a wall adapter, a load step at the output or VIN step on the input can induce ringing at the VIN pin. The ringing can couple to the output and be mistaken as loop instability or could even damage the part by exceeding the maximum ratings. Table 3. List of Capacitor CAPACITANCE TYPE SIZE SUPPLIER 10µF GRM188R60J106M69D 0603 1.6x0.8x0.8mm3 Murata LAYOUT CONSIDERATIONS As for all switching power supplies, the layout is an important step in the design. Proper function of the device demands careful attention to PCB layout. Care must be taken in board layout to get the specified performance. If the layout is not carefully done, the regulator could show poor line and/or load regulation, stability issues as well as EMI problems. It is critical to provide a low inductance, impedance ground path. Therefore, use wide and short traces for the main current paths. The input capacitor should be placed as close as possible to the IC pins as well as the inductor and output capacitor. Connect the GND Pin of the device to the Power Pad of the PCB and use this Pad as a star point. Use a common Power GND node and a different node for the Signal GND to minimize the effects of ground noise. Connect these ground nodes together to the Power Pad (star point) underneath the IC. Keep the common path to the GND PIN, which returns the small signal components and the high current of the output capacitors as short as possible to avoid ground noise. The FB line should be connected right to the output capacitor and routed away from noisy components and traces (e.g., SW line). Submit Documentation Feedback 15 TPS62290 www.ti.com SLVS764 – JUNE 2007 VOUT R2 GND C1 R1 COUT CIN VIN L G N D U Figure 19. Layout 16 Submit Documentation Feedback PACKAGE OPTION ADDENDUM www.ti.com 23-Jul-2007 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Drawing Pins Package Eco Plan (2) Qty TPS62290DRVR ACTIVE SON DRV 6 3000 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM TPS62290DRVRG4 ACTIVE SON DRV 6 3000 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM TPS62290DRVT ACTIVE SON DRV 6 250 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM TPS62290DRVTG4 ACTIVE SON DRV 6 250 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM Lead/Ball Finish MSL Peak Temp (3) (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. 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Addendum-Page 1 PACKAGE MATERIALS INFORMATION www.ti.com 27-Jun-2007 TAPE AND REEL INFORMATION Pack Materials-Page 1 PACKAGE MATERIALS INFORMATION www.ti.com Device 27-Jun-2007 Package Pins Site Reel Diameter (mm) Reel Width (mm) A0 (mm) B0 (mm) K0 (mm) P1 (mm) W Pin1 (mm) Quadrant TPS62290DRVR DRV 6 NSE 179 8 2.2 2.2 1.2 4 8 Q2 TPS62290DRVT DRV 6 NSE 179 8 2.2 2.2 1.2 4 8 Q2 TAPE AND REEL BOX INFORMATION Device Package Pins Site Length (mm) Width (mm) Height (mm) TPS62290DRVR DRV 6 NSE 195.0 200.0 45.0 TPS62290DRVT DRV 6 NSE 195.0 200.0 45.0 Pack Materials-Page 2 IMPORTANT NOTICE Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, modifications, enhancements, improvements, and other changes to its products and services at any time and to discontinue any product or service without notice. Customers should obtain the latest relevant information before placing orders and should verify that such information is current and complete. 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