ONSEMI NCP1002PG

NCP1000, NCP1001,
NCP1002
Integrated Off−Line
Switching Regulator
The NCP1000 through NCP1002 series of integrated switching
regulators, combine a fixed frequency PWM controller with an integrated
high voltage power switch circuit. This chip allows for simple design and
minimal parts count for very low cost applications which utilize an ac
input. This chip is designed to power a single ended topology, typically a
discontinuous mode flyback, with secondary side sensing.
The internal high voltage switch circuit and startup circuit can
function in continuous operation over a wide range of inputs, from
85 Vac to 265 Vac, and thus can be used in any existing power system in
the world. Though inexpensive, these devices include a number of
features such as undervoltage lockout, over−temperature protection,
bandgap reference and leading edge blanking that make them an
excellent value.
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Pin: 1.
2.
3, 6−8
4.
5.
Features
•
•
•
•
•
•
•
•
•
Highly Integrated Solution
Operates Over Universal Input Voltage Range (85 Vac to 265 Vac)
On−board 700 V Power Switch Circuit
Minimal External Parts Required
Input Undervoltage Lockout with Hysteresis
Very Low Standby Current
No Minimum Load Requirement
Opto Fail−Safe Shutdown Circuit
Pb−Free Packages are Available*
Typical Applications
• Cell Phone Chargers
• Wall Adapters
• On−board AC−DC Converters
VCC
INTERNAL BIAS
UVLO &
OPTO FAIL−SAFE
STARTUP
PWM
COMPARATOR
GND
VCC
Feedback Input
Ground
Startup
Power Switch Circuit
MARKING DIAGRAM
8
NCP100xP
AWL
YYWWG
1
x
A
WL
YY
WW
G
= Device Number 0, 1, or 2
= Assembly Location
= Wafer Lot
= Year
= Work Week
= Pb−Free Package
ORDERING INFORMATION*
STARTUP
CIRCUIT
FEEDBACK
INPUT
+
−
PDIP−8
P SUFFIX
CASE 626
8
PWM CONTROL
& POWER
SWITCH CIRCUIT
POWER
SWITCH
CIRCUIT
Device
Package
Shipping
NCP1000P
PDIP−8
50 Units/Rail
0.5
18
PDIP−8 50 Units/Rail
(Pb−Free)
0.5
18
50 Units/Rail
1.0
9
PDIP−8 50 Units/Rail
(Pb−Free)
1.0
9
50 Units/Rail
1.5
6
PDIP−8 50 Units/Rail
(Pb−Free)
1.5
6
NCP1000PG
NCP1001P
NCP1001PG
NCP1002P
OSCILLATOR
NCP1002PG
Figure 1. Simplified Block Diagram
Ipk Ron
Typ Max
(A) (W)
PDIP−8
PDIP−8
*Consult factory for additional optocoupler fail−safe
latching, frequency, and current limit options.
*For additional information on our Pb−Free strategy and soldering details, please
download the ON Semiconductor Soldering and Mounting Techniques
Reference Manual, SOLDERRM/D.
© Semiconductor Components Industries, LLC, 2006
February, 2006 − Rev. 8
1
Publication Order Number:
NCP1000/D
NCP1000, NCP1001, NCP1002
FUNCTIONAL PIN DESCRIPTION
Pin
Function
Description
1
VCC
Positive input supply voltage. This pin is connected to an external capacitor for energy storage.
The startup circuit sources current out of this pin to initially charge the capacitor. When the
voltage reaches the upper threshold limit of the undervoltage lockout circuit, the startup circuit
will turn off, and the power supply will begin operation. Power is then supplied to the chip via
this pin, by virtue of the auxiliary winding.
2
Feedback Input
The error signal from the optocoupler is fed into this input. It is loaded with a 2.7 kW resistor
which converts the opto current into a voltage. There is a 7.0 kHz, single pole, low pass filter
between this pin and the error amp input. A 10 volt clamp is also connected to this pin to protect
the device from ESD damage or overvoltage conditions.
3, 6, 7, 8
Ground
Ground reference pin for the circuit. These pins are part of the integrated circuit leadframe and
are an integral part of the heat flow path on the PDIP−8 package.
4
Startup
This pin is connected to the bulk DC input voltage supply. It feeds an internal current source that
initially charges up the VCC capacitor on power up.
5
Power Switch Circuit
The internal power switch circuit is connected between this pin and ground. This pin connects
directly to one end of the transformer primary winding.
MAXIMUM RATINGS (Notes 1 and 2)
Rating
Symbol
Value
Unit
VDS
IDS(pk)
*0.3 to 700
2.0 Ilim Max
V
A
Power Supply Voltage Range (Pin 1)
Vclp
*0.3 to 10
V
Feedback Input (Pin 2)
Voltage Range
Current
VI(fb)
lfb
*0.3 to 10
100
V
mA
Power Switch Circuit (Pin 5)
Drain Voltage Range
Drain Current Peak During Transformer Saturation
°C/W
Thermal Resistance
P Suffix, Plastic Package Case 626
Junction−to−Lead
Junction−to−Air, 2.0 Oz. Printed Circuit Copper Clad
0.36 Sq. Inch
1.0 Sq. Inch
RqJL
RqJA
5.0
45
35
Operating Junction Temperature
TJ
*40 to 125
°C
Storage Temperature
Tstg
*65 to +150
°C
1. This device series contains ESD protection and exceeds the following tests:
Pins 1−3: Human Body Model 2000 V per MIL−STD−883, Method 3015.
Machine Model Method 200 V.
Pins 4 and 5 are the HV startup and the drain of the LDMOS device, rated only to the max rating of the part , or 700 V.
2. This device contains Latchup protection and exceeds $200 mA per JEDEC Standard JESD78.
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NCP1000, NCP1001, NCP1002
ELECTRICAL CHARACTERISTICS (VCC = 8.6 volts, pin 2 grounded, TJ = 25°C for typical values. For min/max values, TJ is the
operating junction temperature that applies.)
Characteristics
Symbol
Min
Typ
Max
Unit
90
85
75
100
−
−
110
115
115
Av
DAv
−110
−
*136
0.2
−170
−
%/mA
(%/mA)/°C
D(max)
Ifb
68
1.8
72
−
74
−
%
mA
Vrpk
Vrvly
−
−
4.1
2.7
−
−
Vclp
8.3
8.2
7.2
−
8.55
8.5
7.5
1.0
8.9
8.8
8.0
−
2.0
1.5
3.4
2.6
4.2
4.2
OSCILLATOR
Frequency (lfb = 1.1 mA) (Note 4) (Figure 7)
TJ = 25°C
TJ = 0°C to 125°C
TJ = −40°C to 125°C
fOSC
kHz
PWM COMPARATOR
Feedback Input PWM Gain (TJ = 25°C) (lfb = 1.20 mA to 1.30 mA) (Figure 2)
Gain Temperature Coefficient (TJ = −40°C to TJ = 125°C) (Note 4)
PWM Duty Cycle (Pin 2)
Maximum (lfb = 0.8 mA)
Zero Duty Cycle Current
PWM Ramp
Peak
Valley
V
STARTUP CONTROL AND VCC LIMITER
V
Undervoltage Lockout (Figure 8)
VCC Clamp Voltage (ICC = 4.0 mA)
Startup Threshold (Vclp Increasing)
Minimum Operating Voltage After Turn−On
Hysteresis
Vclp(on)
Vclp(min)
VH
Startup Circuit, Pin 1 Output Current (Pin 4 = 50 V)
VCC = 0 V
VCC = 8.0 V
Istart
Minimum Startup Voltage (VCC = Vclp(on) −0.2 V, Istart = 0.5 mA)
Vstart
−
14.7
20
V
Drst
frst
4.0
−
5.0
1.2
6.0
−
%
Hz
VBR(st)
700
−
−
V
−
−
20
30
40
75
Auto Restart (CPin 1 = 47 mF, Pin 4 = 50 V) (Note 5)
Duty Cycle
Frequency
Startup Circuit Breakdown Voltage (I = 25 mA) (Note 5)
Startup Circuit Leakage Current (Pin 4 = 700 VDC)
TJ = 25°C
TJ = −40°C to 125°C
mA
mA
Ileak
3. Maximum package power dissipation limits must be observed.
4. Tested junction temperature range for this device series: Tlow = −40°C, Thigh = +125°C
5. Guaranteed by design only.
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NCP1000, NCP1001, NCP1002
ELECTRICAL CHARACTERISTICS
Characteristics
Symbol
Min
Typ
Max
Unit
POWER SWITCH CIRCUIT
Power Switch Circuit On−State Resistance
NCP1000 (ID = 50 mA)
TJ = 25°C
TJ = 125°C (Note 8)
NCP1001 (ID = 100 mA)
TJ = 25°C
TJ = 125°C (Note 8)
NCP1002 (ID = 150 mA)
TJ = 25°C
TJ = 125°C (Note 8)
R(on)
Power Switch Circuit Breakdown Voltage
(ID(off) = 100 mA, TJ = 25°C)
V(BR)
Power Switch Circuit Off−State Leakage Current (VDS = 650 V)
TJ = 25°C
TJ = −40°C to 125°C
W
−
−
13
24
18
36
−
−
7.0
14
9.0
18
−
−
4.0
8.0
6.0
12
700
−
−
−
−
0.25
−
1.0
50
−
−
50
50
−
−
0.42
0.84
1.26
0.48
0.96
1.43
0.54
1.08
1.6
−
−
−
0.500
1.000
1.500
−
−
−
−
10
18
−
25
35
mA
I(off)
Switching Characteristics (VDS = 50 V, RL set for ID = 0.7 IIim)
Turn−on Time (90% to 10%)
Turn−off Time (10% to 90%)
V
ns
ton
toff
CURRENT LIMIT AND THERMAL PROTECTION
Current Limit Threshold (TJ = 25°C) (Note 9)
NCP1000
NCP1001
NCP1002
Ilim
Current Limit, Peak Switch Current
NCP1000 (di/dt = 100 mA/ms)
NCP1001 (di/dt = 200 mA/ms)
NCP1002 (di/dt = 300 mA/ms)
Ipk
A
A
Opto Fail−safe Protection (Figure 12)
TJ = 25°C
TJ = 0°C to 125°C
IOfail
mA
Propagation Delay, Current Limit Threshold to Power Switch Circuit Output
(Leading Edge Blanking plus Current Limit Delay)
tPLH
−
220
−
tsd
tH
125
−
140
30
−
−
ns
°C
Thermal Protection (Note 6, 8)
Shutdown (Junction Temperature Increasing)
Hysteresis (Junction Temperature Decreasing)
TOTAL DEVICE (Pin 1)
Power Supply Current After UVLO Turn−On
Power Switch Circuit Enabled
NCP1000
NCP1001
NCP1002
Power Switch Circuit Disabled
mA
ICC1
ICC2
−
−
−
0.6
6. Maximum package power dissipation limits must be observed.
7. Tested junction temperature range for this device series:
Tlow = −40°C
Thigh = +125°C
8. Guaranteed by design only.
9. Actual peak switch current is increased due to the propagation delay time and the di/dt (see Figure 16).
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1.2
1.4
1.6
1.0
1.6
1.8
2.0
1.25
NCP1000, NCP1001, NCP1002
2
0.800
0.700
1.5
0.600
ICC (mA)
DC (%)
0.500
0.400
0.300
1
0.5
0.200
0.100
0.000
0.8
1
1.2
1.4
1.6
0
1.8
0
1
2
3
4
5
6
7
lfb (mA)
VCC (V)
Figure 2. Duty Cycle vs. FB Input Current
Figure 3. ICC vs. VCC
1000
8
9
4.00
Startup Current (mA)
VCC Charge Time (ms)
3.50
100
10
3.00
2.50
2.00
1.50
1.00
VCC = 4.0 V
0.50
1
0.00
1
10
100
1
10
100
1000
Capacitance (mF)
Startup Voltage (V) (Pin 4)
Figure 4. Charge Time vs. VCC Capacitance
Figure 5. Startup Current vs. Startup Voltage
100
104.0
102.0
NCP1002
98.0
fOSC, (kHz)
Capacitance (pF)
100.0
NCP1001
10
NCP1000
96.0
94.0
92.0
90.0
88.0
1.0
10
100
86.0
−40
1000
−25
0
25
50
75
Switch Voltage (V)
Temperature (°C)
Figure 6. Power Switch Circuit
Capacitance vs. Voltage
Figure 7. Oscillator Frequency
vs. Temperature
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100
125
NCP1000, NCP1001, NCP1002
1.80
VCC/UVLO Threshold (V)
8.40
Peak Current Limit Threshold (A)
8.60
VCC Turn On
8.20
8.00
7.80
7.60
VCC Turn Off
7.40
7.20
−40 −25
0
25
50
75
100
1.40
1.20
NCP1001
1.00
0.80
NCP1000
0.60
0.40
0.20
0.00
−40
125
−25
0
25
50
75
100
Temperature (°C)
Temperature (°C)
Figure 8. VCC/UVLO Threshold vs.
Temperature
Figure 9. Peak Current Limit Threshold
vs. Temperature
25.0
125
250
200
Leakage Current (mA)
20.0
On Resistance (W)
NCP1002
1.60
NCP1000
15.0
NCP1001
10.0
125°C
150
100
25°C
−40°C
50
5.0
NCP1002
0.0
−40
−25
0
25
50
75
100
0
125
0
400
600
800
1000
Temperature (°C)
Voltage (V)
Figure 10. Power Switch Circuit On Resistance
vs. Temperature
Figure 11. Power Switch Circuit Leakage Current
vs. Pin 5 Voltage
24.0
1.75
22.0
ZDC Feedback Current (mA)
Fail−safe Trigger Current (mA)
200
20.0
18.0
16.0
14.0
12.0
10.0
8.0
−40
−25
0
25
50
75
100
1.70
1.65
1.60
1.55
1.50
1.45
−40
125
−25
0
25
50
75
100
125
Temperature (°C)
Temperature (°C)
Figure 12. Opto Fail−safe Trigger Current
vs. Temperature
Figure 13. Zero Duty Cycle Feedback Current
vs. Temperature
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NCP1000, NCP1001, NCP1002
1.100
14.0
12.0
On Resistance (W)
1.060
1.040
1.020
1.000
0.960
−40
10.0
NCP1001
8.0
6.0
NCP1002
4.0
0.980
−25
0
25
50
75
100
125
2.0
0
0.5
1
Temperature (°C)
Power Switch Circuit Current (A)
Figure 14. Maximum Duty Cycle Feedback
Current vs. Temperature
Figure 15. On Resistance vs. Current
1600
1400
1200
di/dt (mA/ms)
MDC Feedback Current (mA)
NCP1000
1.080
1000
800
600
400
200
0
0
50
100
150
200
250
300
DIpk (mA)
Figure 16. Power Switch Circuit di/dt
vs. DIpk
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350
1.5
NCP1000, NCP1001, NCP1002
OPERATING DESCRIPTION
Introduction
consumption while in the standby operation mode. For
operation at no load, the output may skip cycles. This is a
common occurrence for this type of control circuit. The
converter will switch for several cycles, and due to delays in
the output filter and feedback loop, the duty cycle will not
be reduced until the output has exceeded it’s regulation limit.
The unit will then shut down for several cycles until the
voltage is below the regulation limit, and then it will switch
again. During the time that switching cycles are not present
the output voltage will decay according to it’s RC time
constant, which is based on the output capacitance and
internal loading from the regulation circuitry. During this
interval, the voltage on the VCC supply will also decay. If it
decays below the lower hysteretic turn off threshold, the unit
will shut down and recycle. This mode of operation is not
normally desirable. In order to avoid it, the time constant
for the VCC cap and load should be equal to, or greater
than the time constant of the output. If no load operation
is not required, a relatively small value (t10 mF) for the
VCC capacitor is acceptable.
The NCP1000 thru NCP1002 represent a new higher level
of integration by providing on a single monolithic chip all of
the active power, control, logic, and protection circuitry
required to implement a high voltage flyback or forward
converter. This device series is designed for direct operation
from a rectified 240 Vac line source and requires minimal
external components for a complete cost sensitive converter
solution. Potential markets include office automation,
industrial, residential, personal computer, and consumer. A
description of each of the functional blocks is given below,
and the representative block diagram is shown in Figure 17.
Oscillator
The Oscillator block consists of two comparators that
alternately gate on and off a trimmed current source and
current sink which are used to respectively charge and
discharge an on−chip timing capacitor between two voltage
levels. This configuration generates a precise linear
sawtooth ramp signal that is used to pulse width modulate
the MOSFET of the Power Switch circuit. During the charge
of the timing capacitor, the Oscillator duty cycle output
holds one input of the Driver low. This action keeps the
MOSFET of the Power Switch Circuit off, thus limiting the
maximum duty cycle. The Oscillator frequency is internally
programmed for 100 kHz operation with a controlled charge
to discharge current ratio that yields a maximum Power
Switch Circuit duty cycle of 72%. The Oscillator
temperature characteristics are shown in Figure 7.
Feedback Input
The feedback input, pin 2, accepts the DC error signal that
feeds the non−inverting input to the PWM. Pin 2 has a
nominal 2.7 kW internal resistor to ground, which converts
the optocoupler current into a voltage. Its’ signal is filtered
by a 7.0 kHz low pass filter which reduces high frequency
noise to the input of the PWM comparator.
Typically, the photo transistor of the optocoupler is
connected between VCC (pin 1) and the Feedback input
(pin 2). The photo transistor is effectively a current source
which is driven by the LED, which is connected to the output
regulation circuit of the power supply. An external capacitor
may be connected from pin 2 to ground for additional noise
filtering if necessary.
When the feedback input is below the lower threshold of
the ramp signal, the output of the power converter will be
operating at full duty cycle. The input current vs. duty cycle
transfer function is shown in Figure 2. As the voltage
increases, the duty cycle will vary linearly with the change
in voltage at the feedback input, between the upper and
lower extremes of the ramp waveform 2.7 V to 4.1 V. Above
the upper extreme point of the ramp, the duty cycle will be
zero and no power will be transmitted to the output.
The circuit should be designed such that when the output
is low, the optocoupler will be off, leaving the voltage at
pin 2 at ground (full duty cycle). As the output voltage
increases, the optocoupler will begin to conduct, such that
the voltage at pin 2 increases until the proper duty cycle is
reached to maintain regulation.
Pin 2 is protected from ESD transients by a 10 V Zener
diode to ground.
PWM Comparator and Latch
The pulse width modulator (PWM) consists of a
comparator with the Oscillator ramp output applied to the
inverting input. The Oscillator clock output applies a set
pulse to the PWM Latch when the timing capacitor reaches
its peak voltage, initiating Power Switch Circuit conduction.
As the timing capacitor discharges, the ramp voltage
decreases to a level that is less than the Error Amplifier
output, causing the PWM Comparator to reset the latch and
terminate Power Switch Circuit conduction for the duration
of the ramp−down period. This method of having the
Oscillator set and the PWM Comparator reset the Latch
prevents the possibility of multiple output pulses during a
given Oscillator clock cycle. This circuit configuration is
commonly referred to as double pulse suppression logic. A
timing diagram is shown in Figure 18 that illustrates the
behavior of the pulse width modulator.
No load operation. The pulse width modulator is designed
to operate between 73% and 0% duty cycle. The ability to
operate down to zero duty cycle allows for no load operation
without the burden of preloads. This feature is consistent
with the Blue Angle requirements, as it minimizes power
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NCP1000, NCP1001, NCP1002
Current Limit Comparator and Power Switch Circuit
High Voltage Startup
The NCP1000 series uses cycle−by−cycle current limiting
as a means of protecting the output switch transistor from
overstress. Current limiting is implemented by monitoring
the instantaneous output switch current during conduction,
and upon sensing an overcurrent condition, immediately
turning off the switch for the duration of the Oscillator
ramp−down period.
The Power Switch Circuit is constructed using a
SENSEFETt allowing a virtually lossless method of
monitoring the drain current. A small number of the power
MOSFET cells are used for current sensing by connecting
their individual sources to a single ground referenced sense
resistor, Rpk. The current limit comparator detects if the
voltage across Rpk exceeds the reference level that is present
at the noninverting input. If exceeded, the comparator
quickly resets the PWM Latch, thus protecting the Power
Switch Circuit. Figure 9 shows that this detection method
yields a relatively constant current limit threshold over
temperature. The high voltage Power Switch Circuit is
integrated with the control logic circuitry and is designed to
directly drive the converter transformer. The Power Switch
Circuit is capable of switching 700 V with an associated
drain current that ranges from 0.5 A to 1.5 A. Proper drain
voltage snubbing during converter startup and overload is
mandatory for reliable device operation.
A Leading Edge Blanking circuit was placed in the current
sensing signal path to prevent a premature reset of the PWM
Latch. A potential premature reset signal is generated each
time the Power Switch Circuit is driven into conduction and
appears as a narrow voltage spike across current sense
resistor Rpk. The spike is due to the MOSFET gate to source
capacitance, transformer interwinding capacitance, and
output rectifier recovery time. The Leading Edge Blanking
circuit has a dynamic behavior that masks the current signal
until the Power Switch Circuit turn−on transition is
completed.
The current limit propagation delay time is typically
220 ns. This time is measured from when an overcurrent
appears at the Power Switch Circuit drain, to the beginning
of turn−off. Care must be taken during transformer
saturation so that the maximum device current limit rating
is not exceeded. To determine the peak Power Switch Circuit
current at turn off, the effect of the propagation delay must
be taken into account. To do this, use the appropriate Current
Limit Threshold value from the electrical tables, and then
add the DIpk based on the di/dt from Figure 16. The di/dt of
the circuit can be calculated by the following formula:
The NCP1000−1002 contain an internal startup circuit
that eliminates the need for external startup components. In
addition, this circuit increases the efficiency of the supply as
it uses no power when in the normal mode of operation, but
instead uses the power supplied by the auxiliary winding.
Rectified, filtered ac line voltage is connected to pin 4. An
internal JFET allows current to flow from the startup pin, to
the VCC pin at a current of approximately 3.0 mA. Figure 5
shows the startup current out of pin 1 which charges the
capacitor(s) connected to this pin.
The start circuit will be enhanced (conducting) when the
voltage at Pin 1 (VCC) is less than 7.5 V. It will remain
enhanced until the VCC voltage reaches 8.5 V. At this point
the Power Switch Circuit will be disabled, and the unit will
generate voltage via the auxiliary winding to maintain
proper operation of the device. Figure 4 shows the charge
time for turn−on vs. VCC capacitance when the unit is
initially energized.
If the VCC voltage drops below 7.5 V (e.g. current limit
mode), the start circuit will again begin conducting, and will
charge up the VCC cap until the 8.5 V limit is reached.
VCC Limiter and Undervoltage Lockout
The undervoltage lockout (UVLO) is designed to
guarantee that the integrated circuit has sufficient voltage to
be fully functional before the output stage is enabled. It
inhibits operation of the major functions of the device by
disabling the Internal Bias circuitry, and assures that the
Power Switch Circuit remains in its “off’’ state as the bias
voltage is initially brought up from zero volts. When the
NCP100x is in the “off’’ state, the High Voltage Startup
circuit is operational. The UVLO is a hysteretic switch and
will hold the device in its “off’’ state any time that the VCC
voltage is less than 7.5 V. As the VCC increases past 7.5 V,
the NCP100x will remain off until the upper threshold of 8.6
V is reached. At this time the power converter is enabled and
will commence operation. The UVLO will allow the unit to
continue to operate as long as the VCC voltage exceeds 7.5 V.
The temperature characteristics of the UVLO circuit are
shown in Figure 8.
If the converter output is overloaded or shorted, the device
will enter the auto restart mode. This happens when the
auxiliary winding of the power transformer does not have
sufficient voltage to support the VCC requirements of the
chip. Once the chip is operational, if the VCC voltage falls
below 7.5 V the unit will shut down, and the High Voltage
Startup circuit will be enabled. This will charge the VCC cap
up to 8.5 V, which will clock the divide by eight counter. The
divide by eight counter holds the Power Switch Circuit off.
This causes the VCC cap to discharge. It will continue to
discharge and recharge for eight consecutive cycles. After
the eighth cycle, the unit will turn on again. If the fault
remains, the unit will again cycle through the auto restart
mode; if the fault has cleared the unit will begin normal
operation. The auto restart mode greatly reduces the power
dissipation of the power devices in the circuit and improves
dińdt (Ańms) + VńL
where:
V is the rectified, filtered input voltage (volts)
L is the primary inductance of the flyback transformer
(Henries)
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NCP1000, NCP1001, NCP1002
reliability in overload conditions. Figure 20 shows the
timing waveforms in auto restart mode.
The VCC pin receives its startup power from the high
voltage startup circuit. Once the undervoltage lockout trip
point is exceeded, the high voltage startup circuit turns off,
and the VCC pin receives its power from the auxiliary
winding of the power transformer. Once the converter is
enabled, the VCC voltage will be clamped by the 8.6 V
limiter. Since the voltage limiter will regulate the VCC
voltage at 8.6 V, it must shunt all excess current based on the
input impedance to this pin. A resistor is required between
the auxiliary winding filter capacitor and the VCC pin to
limit the current.
For minimum bias current:
Rs max +
(V
* 8.8 volts)
AUX min
I
CC1max
where:
VAUXmin is the minimum expected DC voltage from the
auxiliary winding.
Typically, this voltage will vary between "5% to "10%
from it’s nominal value.
ICC1max is the maximum rated bias current for the device
used. This value can found in the tables on the data sheet.
For the best optocoupler fail−safe response:
Optocoupler Fail−safe Circuit
Rs
The NCP100x has the ability to sense an open optocoupler
and protect the load in the event of a failure. This circuit
operates by sensing the current in the VCC limiter, and
detecting a high current which is an indication of an open
optocoupler.
The VCC pin receives the output of a current source which
is created by the voltage drop between the auxiliary winding
and the VCC limiter across the shunt resistor. The Vcc limiter
will clamp the VCC voltage to approximately 8.6 V. Any
current that is available at this pin, that is not needed for
either the chip bias current, or the opto current is shunted
through this limiter.
The opto fail−safe circuit operates on the premise that
under an open opto condition, the opto current will all be
shunted through the VCC limiter, and the output voltage (and
therefore the auxiliary winding voltage) will increase. The
increase in auxiliary winding voltage will cause an
amplified increase in the current into the VCC pin. To detect
an open opto condition, the current in the limiter is measured
and if it exceeds 10 milliamps, the chip will shut down and
go into burst mode operation. After a shutdown signal, the
optocoupler fail−safe circuit will enable the
divide−by−eight counter and attempt to restart the unit after
every eight VCC cycles.
For this circuit to operate properly, the shunt resistor must
be chosen prudently. There is a range of values for RS that
will determine the operation of this circuit. On one extreme,
a large value of RS will minimize the bias current, which will
have the effect of maximizing efficiency, while reducing the
response to an open optocoupler. The other extreme is the
minimum value of RS, which will maximize the bias current
into the chip and minimize the voltage overshoot in the event
of an open optocoupler.
min
+
(V
* 7.2 volts)
AUX max
1.0mA ) I
trip
where:
VAUXmax is the maximum expected DC voltage from the
auxiliary winding.
Itrip is the minimum trip current for the optocoupler
fail−safe. This information can be found in the tables under
Current Limit and Thermal Protection, as well as in
Figure 12.
The value of RS that is used in the circuit must be between
the two extreme values calculated. Setting it closer to RSmin
will optimize the optocoupler fail−safe feature, while setting
it closer to the RSmax value will minimize the bias current
Thermal Shutdown and Package
The internal Thermal Shutdown block protects the device
in the event that the maximum junction temperature is
exceeded. When activated, typically at 140°C, one input of
the Driver is held low to disable the Power Switch Circuit.
Thermal shutdown activation is non−latching and the Power
Switch Circuit is allowed resume operation when the
junction temperature falls below 110°C. The thermal
shutdown feature is provided to prevent catastrophic device
failures from accidental overheating. It is not intended to be
used as a substitute for proper heatsinking.
The die in the 8−pin dual−in−line package is mounted on
a special heat tab copper alloy lead frame. The tab consists
of pins 3, 6, 7, 8 is specifically designed to improve the
thermal conduction from the die to the printed circuit board.
This permits the use of standard layout and mounting
practices while having the ability to halve the junction to air
thermal resistance.
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10
NCP1000, NCP1001, NCP1002
Applications
offset by the board layout. The Power Switch Circuit pin can
be offset if additional layout creepage distance is required.
Due to the potentially high rate of change in switch current,
a capacitor (if used), at pin 2, should have traces as short as
possible, from pin 2 and ground. This will significantly
reduce the level of switching noise that can be imposed upon
the feedback control signal.
Do not attempt to construct a converter circuit on a
wire−wrap or plug−in prototype board. In order to ensure
proper device operation and stability, it is important to
minimize the lead length and the associated inductance of
the ground pin. This pin must connect as directly as possible
to the printed circuit ground plane and should not be bent or
+
AC Line
Input
+
Snubber
+
Auxiliary Output
Bypass
+
+
Startup
Auto Restart Timing
Power Switch
Circuit
5
4
VCC
1
Startup
10 V
VCC Limiter 8.6 V
Feedback
Input
Internal
Bias
Opto Fail−safe
10 V
2
+
10 mA
+
State Control
Power up−reset
Undervoltage
8.5 V/ Lockout
7.5 V
Ck R
Divide
by 8
Q
Thermal
Shutdown
Loop Compensation
Oscillator
Duty Cycle
PWM Latch
S
Q
R
Clock
Ramp
7.0 kHz
Filter
Driver
PWM
Comparator
Leading Edge
Blanking
2.7 k
+
Current Limit
Comparator
Ground
Rpk
3
Figure 17. Representative Block Diagram
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11
Converter
DC Output
−
NCP1000, NCP1001, NCP1002
Feedback Input
(Rfb Voltage)
Oscillator Ramp
Oscillator
Duty Cycle
Oscillator Clock
PWM
Comparator
Output
PWM Latch
Q Output
Current Limit
Propagation
Delay
Power Switch
Circuit Gate Drive
Current
Limit
Threshold
Leading Edge
Blanking Input
(Power Switch
Circuit Current)
Zero Duty Cycle
Normal PWM Operating Range
Output Overload
Figure 18. Pulse Width Modulation Timing Diagram
8.6 V
7.5 V
Regulation Threshold
VCC Limiter
(Pin 1)
0V
Flyback Voltage
Power Switch
Circuit Voltage
(Pin 5)
Rectified Line Voltage
Switching Disabled
Switching
Startup
Normal Operation
Figure 19. Startup and Normal Operation Timing Diagram
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12
0V
NCP1000, NCP1001, NCP1002
1
2
7
8
1
2
7
8
1
8.6 V
7.5 V
Hysteretic Regulation
Startup Circuit
On
Istart
Off
ICC1
On
Istart
Off
ICC2
On
Istart
Off On
ICC2 Istart
Off
ICC1
35% 65%
Startup MOSFET
Duty Cycle
During Auto Restart
VCC Limiter
(Pin 1)
Eight Cycles
0V
Power Switch Circuit Duty Cycle During Auto Restart
5%
95%
Flyback Voltage
Rectified Line Voltage
Power Switch
Circuit Voltage
(Pin 5)
0V
Switching Disabled
Switching
Switching
Switching Disabled
Startup
Switching Disabled
Switching
Auto Restart Operation with Overloaded or Shorted Output
Figure 20. Auto Restart Operation Timing Diagram
APPLICATIONS INFORMATION
Converter Test Data
APPLICATION #1: Offline Converter Provides
5.0 V, 1.0 A Output for Small Electronic Equipment
Parameter
ON Semiconductor’s NCP1000 series of offline converters
offers a low cost, high efficiency power source for low power,
electronic equipment. It serves the same function as small, line
frequency transformers, but with the added benefits of line and
load regulation, transient suppression, reduction in weight, and
operation across the universal input voltage range.
This kit provides a 5.0 V, 1.0 A output, which is derived from
an input source of 85 to 265 Vac, and 50 Hz to 60 Hz. This range
of input voltages will allow this circuit to function virtually
anywhere in the world without modification. The output is
regulated and current limited, and EMI filters are included on
both the input and output.
RS
F1
85 v v Vin v 265 v
DVo = 6.0 mV
Load Regulation
0 A v lo v 1.0 A
DVo = 8.0 mV
Combined Line/
Load Regulation
85 v v Vin v 265 v
.09 A v lo v 1.0 A
DVo = 10 mV
Output Ripple
lo = 1.0 A
100 mVpp
Input Power
Vin = 115 v, lo = 1.0 A
Vin = 220 v, lo = 1.0 A
7.75 watts
7.88 watts
Power Factor
Vin = 115 v, lo = 1.0 A
Vin = 220 v, lo = 1.0 A
−.57
−.49
Efficiency
Vin = 115 v, lo = 1.0 A
Vin = 220 v, lo = 1.0 A
h = 66%
h = 64%
L2
C1
D5
5 V, 1 A
C6
R2
+
V in
Data
Line Regulation
D1 − D4
6.8
Conditions
C7
+
R3
C3
R4
p/o U2
C2a C2b
R7
6.8
p/o U2
C10
D6
R1
U3
C8
1
R5
NCP
1000
C5
RETURN
C9
C4
Figure 21. Wall Adapter Schematic
For additional information on this application, please order application note AND8019/D from the Literature Distribution Center or download
from our website at http://onsemi.com.
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13
NCP1000, NCP1001, NCP1002
PACKAGE DIMENSIONS
PDIP−8
P SUFFIX
CASE 626−05
ISSUE L
8
NOTES:
1. DIMENSION L TO CENTER OF LEAD WHEN
FORMED PARALLEL.
2. PACKAGE CONTOUR OPTIONAL (ROUND OR
SQUARE CORNERS).
3. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
5
−B−
1
4
F
−A−
NOTE 2
L
C
J
−T−
MILLIMETERS
MIN
MAX
9.40
10.16
6.10
6.60
3.94
4.45
0.38
0.51
1.02
1.78
2.54 BSC
0.76
1.27
0.20
0.30
2.92
3.43
7.62 BSC
−−−
10 _
0.76
1.01
INCHES
MIN
MAX
0.370
0.400
0.240
0.260
0.155
0.175
0.015
0.020
0.040
0.070
0.100 BSC
0.030
0.050
0.008
0.012
0.115
0.135
0.300 BSC
−−−
10_
0.030
0.040
N
SEATING
PLANE
D
H
DIM
A
B
C
D
F
G
H
J
K
L
M
N
M
K
G
0.13 (0.005)
M
T A
M
B
M
The products described herein (NCP1000, 1001, 1002), may be covered by one or more of the following U.S. patents: 4,553,084; 5,418,410; 5,477,175;
5,760,639; 5,859,768. There may be other patents pending.
ON Semiconductor and
are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice
to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability
arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages.
“Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All
operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. SCILLC does not convey any license under its patent rights
nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications
intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where personal injury or death may occur. Should
Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold SCILLC and its officers, employees, subsidiaries, affiliates,
and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death
associated with such unintended or unauthorized use, even if such claim alleges that SCILLC was negligent regarding the design or manufacture of the part. SCILLC is an Equal
Opportunity/Affirmative Action Employer. This literature is subject to all applicable copyright laws and is not for resale in any manner.
PUBLICATION ORDERING INFORMATION
LITERATURE FULFILLMENT:
N. American Technical Support: 800−282−9855 Toll Free
Literature Distribution Center for ON Semiconductor
USA/Canada
P.O. Box 61312, Phoenix, Arizona 85082−1312 USA
Phone: 480−829−7710 or 800−344−3860 Toll Free USA/Canada Japan: ON Semiconductor, Japan Customer Focus Center
2−9−1 Kamimeguro, Meguro−ku, Tokyo, Japan 153−0051
Fax: 480−829−7709 or 800−344−3867 Toll Free USA/Canada
Phone: 81−3−5773−3850
Email: [email protected]
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ON Semiconductor Website: http://onsemi.com
Order Literature: http://www.onsemi.com/litorder
For additional information, please contact your
local Sales Representative.
NCP1000/D