SLUS562A − JUNE 2003 − REVISED SEPTEMBER 2003
FEATURES
D Wide Input Supply Range: −36 V to −80 V
D Transient Rating to −100 V
D Improved Transient Response
D Enable Input (EN)
D Programmable Current Limit
D Programmable Current Slew Rate
D Fault Timer to Eliminate Nuisance Trips
D Open-Drain Power Good Output (PG)
D 8-Pin MSOP Package
DESCRIPTION
The TPS2398 and TPS2399 integrated circuits
are hot swap power managers optimized for use
in nominal −48-V systems. For redundant-supply
systems, they incorporate an improved circuit
breaker response that provides rapid protection
from short circuits, while still enabling plug-ins to
tolerate large transients that can be generated by
the sudden switchover to a higher voltage supply.
They are designed for supply voltage ranges up to
−80 V, and are rated to withstand spikes to −100 V.
In conjunction with an external N-channel FET
and sense resistor, they can be used to enable live
insertion of plug-in cards and modules in powered
systems. Both devices provide load current slew
rate and peak magnitude limiting, easily
programmed by sense resistor value and a singleexternal capacitor.
⋅APPLICATIONS
D −48-V Distributed Power Systems
D Redundant Negative Voltage Supplies
D Central Office Switching
APPLICATION DIAGRAM
−48V_RTN
VIN+
R3
30 kΩ
0.5 W
EN
VDD
VOUT+
10 µA
+
R2
100 kΩ
VIN−
D2
−48V_INB
D1
C1
0.047 µF
1 PG
RTN 8
2 EN
GATE 7
+
COUT
C3
100 µF
100 V
TPS2398/TPS2399
VOUT+
VOUT−
VOUT−
DC/DC CONVERTER
Q1
IRF530
3 FLTTIME ISENS 6
4 IRAMP
−VIN 5
C2
3900 pF
R1
0.02 Ω
1%
UDG−03069
−48V_INA
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments
semiconductor products and disclaimers thereto appears at the end of this data sheet.
!"# $ %&'# "$ (&)*%"# +"#',
+&%#$ %! # $('%%"#$ (' #-' #'!$ '."$ $#&!'#$
$#"+"+ /""#0, +&%# (%'$$1 +'$ # '%'$$"*0 %*&+'
#'$#1 "** (""!'#'$,
Copyright 2002, Texas Instruments Incorporated
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1
SLUS562A − JUNE 2003 − REVISED SEPTEMBER 2003
DESCRIPTION (continued)
They also provide single-line fault reporting, electrical isolation of faulty cards, and protection against nuisance
overcurrent trips. The TPS2398 latches off in response to current faults, while the TPS2399 periodically retries
the load in the event of a fault.
ABSOLUTE MAXIMUM RATINGS (See Note 1)
Input voltage range, all pins except RTN, EN, PG(2)
Input voltage range, RTN(2)
Input voltage range, EN(2)(3)
Output voltage range, PG(2)(4)
TPS2398/1
UNIT
−0.3 V to 15
V
−0.3 V to 100
V
−0.3 V to 100
V
−0.3 V to 100
V
10
mA
Continuous output current, PG
Continuous total power dissipation
see Dissipation Rating Table
Operating junction temperature range, TJ
−55_C to 125_C
_C
Storage temperature range, Tstg
−65_C to 150_C
_C
260_C
_C
Lead temperature soldering 1,6 mm (1/16 inch) from case for 10 seconds
NOTES 1: Stresses beyond those listed under ”absolute maximum ratings” may cause permanent damage to the device. These are stress ratings
only and functional operation of the device at these or any other conditions beyond those indicated under ”recommended operating
conditions” is not implied. Exposure to absolute−maximum−rated conditions for extended periods may affect device reliability.
2: All voltages are with respect to −VIN (unless otherwise noted).
3: With 100-kΩ minimum input series resistance, −0.3 V to 15 V with low impedance.
4: With 10-kΩ minimum series resistance, −0.3 V to 80 V with low impedance.
ELECTROSTATIC DISCHARGE (ESD) PROTECTION
MIN
UNIT
Human Body Model (HBM)
1.5
kV
Charged Device Model (CDM)
1.5
kV
RECOMMENDED OPERATING CONDITIONS†
MIN
Nominal input supply, −VIN to RTN
Operating junction temperature range
† All voltages are with respect to −VIN (unless otherwise noted)
MAX
UNIT
−80
NOM
−36
V
−40
85
_C
DISSIPATION RATING TABLE
PACKAGE
TA < 25_C
POWER RATING
DERATING FACTOR
ABOVE TA = 25_C
TA = 85_C
POWER RATING
MSOP-8
420 mW
4.3 mW/_C
160 mW
AVAILABLE OPTIONS
OPERATING
TA
−40_C to 85_C
2
FAULT
OPERATION
PACKAGED DEVICES
MSOP (DGK)
Latch off
TPS2398DGK
Periodically retry
TPS2399DGK
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SLUS562A − JUNE 2003 − REVISED SEPTEMBER 2003
ELECTRICAL CHARACTERISTICS
VI(−VIN) = −48 V with respect to RTN, VI(EN) = 2.8 V, VI(ISENS) = 0, all outputs unloaded, TA = −40_C to 85_C
(unless otherwise noted)(1)(2)
input supply
PARAMETER
TEST CONDITIONS
ICC1
ICC2
Supply current, RTN
Supply current, RTN
VI(RTN) = 48 V
VI(RTN) = 80 V
VUVLO_L
VHYS
UVLO threshold, input voltage rising
To GATE pull-up, referenced to RTN
UVLO hysteresis
MIN
TYP
MAX
UNIT
700
1000
µA
1000
1500
µA
−36
−30
−25
V
1.8
2.3
3.0
V
enable input (EN)
PARAMETER
TEST CONDITIONS
VTH
VHYS_EN
Threshold voltage, input voltage rising
IIH
High-level input current
To GATE pull-up
EN hysteresis
VI(EN) = 5 V
MIN
TYP
MAX
UNIT
1.3
1.4
1.5
V
22
60
90
mV
−2
1
2
µA
linear current amplifier (LCA)
PARAMETER
VOH
ISINK
High-level output, GATE
II
VREF_K
VIO
Input offset voltage
TEST CONDITIONS
MIN
TYP
11
14
Output sink current
VI(ISENS) = 0 V
VI(ISENS) = 80 mV, VO(GATE) = 5V, Fault mode
50
100
Input current, ISENS
0 V < VI(ISENS) < 0.2 V
−1
Reference clamp voltage
VO(IRAMP) = open
VO(IRAMP) = 2 V
33
40
−7
MAX
17
UNIT
V
mA
1
µA
46
mV
6
mV
ramp generator
PARAMETER
TEST CONDITIONS
ISRC1
ISRC2
IRAMP source current, slow turn-on rate
VOL
AV
Low-level output voltage
IRAMP source current, normal rate
VO(IRAMP) = 0.25 V
VO(IRAMP) = 1 V, 3 V
VI(EN) = 0 V
VO(IRAMP) = 1 V, 3 V
Voltage gain, relative to ISENS
MIN
TYP
MAX
UNIT
−850
−600
−400
nA
−11
−10
−9
µA
5
mV
9.5
10.0
10.5
mV/V
MAX
UNIT
overload comparator
PARAMETER
VTH_OL
tDLY
TEST CONDITIONS
Current overload threshold, ISENS
Glitch filter delay time
VI(ISENS) = 200 mV
MIN
TYP
80
100
120
mV
2
4
7
µs
fault timer
PARAMETER
TEST CONDITIONS
VOL
ICHG
Low-level output voltage
VFLT
IDSG
Fault threshold voltage
Discharge current, retry mode
TPS2399
D
Output duty cycle
TPS2399
Charging current, current limit mode
VI(EN) = 0 V
VI(ISENS) = 80 mV, VO(FLTTIME) = 2 V
VI(ISENS) = 80 mV, VO(FLTTIME) = 2 V
IRST
Discharge current, timer reset mode
VO(FLTTIME) = 2 V,VI(ISENS) = 0 V
NOTES 1: All voltages are with respect to the −VIN terminal unless otherwise stated.
2: Currents are positive into and negative out of the specified terminal.
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MIN
TYP
MAX
UNIT
5
mV
−55
−50
−45
µA
3.75
4.00
4.25
V
0.38
0.75
µA
1
1.5
%
1
mA
3
SLUS562A − JUNE 2003 − REVISED SEPTEMBER 2003
ELECTRICAL CHARACTERISTICS (continued)
VI(−VIN) = −48 V with respect to RTN, VI(EN) = 2.8 V, VI(ISENS) = 0, all outputs unloaded, TA = −40_C to 85_C
(unless otherwise noted)(1)(2)
PG output
PARAMETER
IOH
RDS(ON)
TEST CONDITIONS
High-level output (leakage) current
MIN
VI(EN) = 0 V,
VO(PG) = 65 V
VI(ISENS) = 80 mV, VO(FLTTIME) = 5V,
IO(PG) = 1 mA
Driver ON resistance
NOTES 1: All voltages are with respect to the −VIN terminal unless otherwise stated.
2: Currents are positive into and negative out of the specified terminal.
TERMINAL FUNCTIONS
TERMINAL
NAME
I/O
DESCRIPTION
NO.
EN
2
I
Enable input to turn on/off power to the load.
PG
FLTTIME
1
O
Open-drain, active-low indication of a load power good condition.
3
I/O
Connection for user-programming of the fault timeout period.
GATE
IRAMP
7
O
Gate drive for external N-channel FET.
4
I/O
Programming input for setting the inrush current slew rate.
ISENS
6
I
Current sense input.
RTN
8
I
Positive supply input for the TPS2398 and TPS2399.
−VIN
5
I
Negative supply input and reference pin for the TPS2398 and TPS2399.
DGK PACKAGE
(TOP VIEW)
4
PG
1
8
RTN
EN
2
7
GATE
FLTTIME
3
6
ISENS
IRAMP
4
5
−VIN
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TYP
35
MAX
UNIT
10
µA
80
Ω
SLUS562A − JUNE 2003 − REVISED SEPTEMBER 2003
DETAILED PIN DESCRIPTIONS
EN: Enable input to turn on/off power to the load. The EN pin is referenced to the −VIN potential of the circuit.
When this input is pulled high (above the nominal 1.4-V threshold) the device enables the GATE output, and
begins the ramp of current to the load. When this input is low, the linear current amplifier (LCA) is disabled, and
a large pull-down device is applied to the FET gate, disabling power to the load.
FLTTIME: Connection for user-programming of the fault timeout period. An external capacitor connected from
FLTTIME to −VIN establishes the timeout period to declare a fault condition. This timeout protects against
indefinite current sourcing into a faulted load, and also provides a filter against nuisance trips from momentary
current spikes or surges. The TPS2398 and TPS2399 define a fault condition as voltage at the ISENS pin at
or greater than the 40-mV fault threshold. When a fault condition exists, the timer is active. The devices manage
fault timing by charging the external capacitor to the 4-V fault threshold, then subsequently discharging it to reset
the timer (TPS2398), or discharging it at approximately 1% the charge rate to establish the duty cycle for retrying
the load (TPS2399). Whenever the internal fault latch is set (timer expired), the pass FET is rapidly turned off,
and the PG output is deasserted.
GATE: Gate drive for external N-channel FET. When enabled, and the input supply is above the UVLO
threshold, the gate drive is enabled and the device begins charging an external capacitor connected to the
IRAMP pin. This pin voltage is used to develop the reference voltage at the non-inverting input of the internal
LCA. The inverting input is connected to the current sense node, ISENS. The LCA acts to slew the pass FET
gate to force the ISENS voltage to track the reference. The reference is internally clamped at 40 mV, so the
maximum current that can be sourced to the load is determined by the sense resistor value as IMAX ≤ 40
mV/RSENSE. Once the load voltage has ramped up to the input dc potential, and current demand drops off, the
LCA drives the GATE output to about 14 V to fully enhance the pass FET, completing the low-impedance supply
return path for the load.
IRAMP: Programming input for setting the inrush current slew rate. An external capacitor connected between
this pin and −VIN establishes the load current slew rate whenever power to the load is enabled. The device
charges the external capacitor to establish the reference input to the LCA. The closed-loop control of the LCA
and pass FET acts to maintain the current sense voltage at ISENS at the reference potential. Since the sense
voltage is developed as the drop across a resistor, the charging current ramp rate is set by the voltage ramp
rate at the IRAMP pin. When the output is disabled via the EN input or due to a load fault, the capacitor is
discharged and held low to initialize for the next turn-on.
ISENS: Current sense input. An external low value resistor connected between this pin and −VIN is used to feed
back current magnitude information to the TPS2398/99. There are two internal device thresholds associated
with the voltage at the ISENS pin. During charging of the load’s input capacitance, or during other periods of
excessive demand, the HSPM acts to limit this voltage to 40 mV. Whenever the LCA is in current regulation
mode, the capacitor at FLTTIME is charged to activate the timer. If, when the LCA is driving to its supply rail,
a fast-acting fault such as a short-circuit, causes the ISENS voltage to exceed 100 mV (the overload threshold),
the GATE pin is pulled low rapidly, bypassing the fault timer.
PG: Open-drain, active-low indication of load power good. A power good status is declared when the output
is enabled, the GATE pin voltage has ramped to at least 7 V, and the voltage on the IRAMP pin exceeds
approximately 5 V. This last condition assures that full programmed sourcing current is available prior to
declaring power good, even with very slow current ramp rates. This additional protection prevents potential
discharging of the module input bulk capacitance during load turn-on.
RTN: Positive supply input for the TPS2398/99. For negative voltage systems, the supply pin connects directly
to the return node of the input power bus. Internal regulators step down the input voltage to generate the various
supply levels used by the TPS2398 and TPS2399.
−VIN: Negative supply input and reference pin for the TPS2398/99. This pin connects directly to the input supply
negative rail. The input and output pins and all internal circuitry are referenced to this pin, so it is essentially the
GND or VSS pin of the device.
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5
SLUS562A − JUNE 2003 − REVISED SEPTEMBER 2003
TYPICAL CHARACTERISTICS
LIVE INSERTION EVENT
VIN = −48 V
LIVE INSERTION EVENT
VIN = −80 V
EN (20 V/div.)
EN (20 V/div.)
VDRAIN (50 V/div.)
VDRAIN (20 V/div.)
Power Applied
Power Applied
CLOAD = 100 µF
CIRAMP = 3900 pF
CFLT = 0.1 µF
CLOAD = 50 µF
ILOAD
(500 mA/div.)
ILOAD (500 mA/div.)
t − TIme − 1 ms/div
t − TIme − 1 ms/div
Figure 1
Figure 2
LOAD CURRENT RAMP PROFILES
START-UP FROM ENABLE ASSERTION
IRAMP (2 V/div.)
EN (5 V/div.)
CIRAMP = .022 µF
VDRAIN 50 V/div.
CIRAMP =
3900 pF
CIRAMP = .047 µF
IRAMP (5 V/div.)
CLOAD = 100 µF
EN driven from logiclevel signal, ref to −VIN
ILOAD (1 A/div.)
ILOAD
t − TIme − 10 ms/div
t − TIme − 1 ms/div
Figure 4
Figure 3
6
CFLT = 0.33 µF
CLOAD = 600 µF
(500 mA/div.)
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SLUS562A − JUNE 2003 − REVISED SEPTEMBER 2003
TYPICAL CHARACTERISTICS
TURN-ON INTO SHORTED LOAD
(TPS2399)
TURN-ON INTO SHORTED LOAD
(TPS2398)
PG (50 V/div.)
PG (50 V/div.)
VIRAMP (5 V/div.)
VIRAMP (5 V/div.)
FLTTIME (2 V/div.)
FLTTIME (2 V/div.)
ILOAD
(1 A/div.)
CIRAMP = 3900 pF
CFLT = 0.047 µF
RPG = 100 kΩ
CIRAMP = 3900 pF
CFLT = 0.047 µF
RPG = 100 kΩ
t − TIme − 1 ms/div
t − TIme − 1 ms/div
Figure 5
FAULT RETRY OPERATION
(TPS2399)
ILOAD
(1 A/div.)
Figure 6
RECOVERY FROM A FAULT − LARGE SCALE VIEW
(TPS2399)
PG (50 V/div.)
PG (50 V/div.)
FLTTIME (2 V/div.)
FLTTIME (2 V/div.)
VDRAIN (20 V/div.)
VDRAIN (20 V/div.)
CIRAMP = 3900 pF
CFLT = 0.047 µF
CLOAD = 100 µF
CIRAMP = 3900 pF
CFLT = 0.047 µF
CLOAD = 100 µF
RLOAD = 12.5 Ω
ILOAD (1 A/div.)
ILOAD (1 A/div.)
t − TIme − 50 ms/div
t − TIme − 50 ms/div
Figure 7
Figure 8
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7
SLUS562A − JUNE 2003 − REVISED SEPTEMBER 2003
TYPICAL CHARACTERISTICS
PG OUTPUT TIMING,
VOLTAGE QUALIFIED
RECOVERY FROM A FAULT − EXPANDED VIEW
(TPS2399)
PG (50 V/div)
CIRAMP = 3900 pF
CFLT = 0.1 µF
CLOAD = 150 µF
FLTTIME (2 V/div.)
VIRAMP (2 V/div)
VDRAIN (20 V/div)
VDRAIN (20 V/div)
CIRAMP = 3900 pF
CFLT = 0.047 µF
CLOAD = 100 µF
ILOAD (1 A/div)
PG (50 V/div)
t − TIme − 1 ms/div
t − TIme − 1 ms/div
Figure 10
Figure 9
SUPPLY CURRENT
vs
AMBIENT TEMPERATURE
PG OUTPUT TIMING,
CURRENT QUALIFIED
1200
VTH_PG
1000
ICC − Supply Current − µA
CIRAMP = 0.01 µF
CFLT = 0.1 µF
CLOAD = 50 µF
VIRAMP (2 V/div)
VDRAIN (20 V/div)
VRTN = 80 V
800
600
VRTN = 48 V
400
VRTN = 36 V
200
PG (50 V/div)
0
−40
t − TIme − 1 ms/div
−15
10
35
60
TA − Ambient Temperature − °C
Figure 11
8
Figure 12
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85
SLUS562A − JUNE 2003 − REVISED SEPTEMBER 2003
TYPICAL CHARACTERISTICS
IRAMP OUTPUT CURRENT
vs
AMBIENT TEMPERATURE, SLOW TURN-ON
GATE HIGH-LEVEL OUTPUT
vs
AMBIENT TEMPERATURE
−0.50
17.0
ISRC1 − IRAMP Output Current − µA
VI(ISENS) = 0 V
VOH − Output Voltage − V
16.5
VRTN = 80 V
16.0
VRTN = 48 V
15.5
15.0
−0.54
VRTN = 80 V
−0.58
VRTN = 48 V
VRTN = 36 V
−0.62
14.5
VO(IRAMP) = 0.25 V
VRTN = 36 V
−066
14.0
−40
−15
10
35
60
−40
85
−15
35
85
Figure 14
Figure 13
IRAMP OUTPUT CURRENT
vs
AMBIENT TEMPERATURE, NORMAL RATE
TIMER CHARGING CURRENT
vs
AMBIENT TEMPERATURE
−9.5
−45
VI(ISENS) = 80 mV
VO(FLTTIME) =2V
Average for VO(IRAMP) = 1 V, 3 V
VRTN = 36 V to 80 V
−9.7
ICHG − Charging Current − µA
−47
−9.9
−10.1
−10.3
−49
VRTN = 80 V
−51
−53
VRTN = 36 V
VRTN = 48 V
−10.5
−40
60
TA − Ambient Temperature − °C
TA − Ambient Temperature − °C
ISRC2 − IRAMP Output Current − µA
10
−55
−15
10
35
60
85
TA − Ambient Temperature − °C
−40
−15
10
35
60
85
TA − Ambient Temperature − °C
Figure 15
Figure 16
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9
SLUS562A − JUNE 2003 − REVISED SEPTEMBER 2003
TYPICAL CHARACTERISTICS
FLTTIME DISCHARGE CURRENT
vs
AMBIENT TEMPERATURE (TPS2399)
FAULT LATCH THRESHOLD
vs
AMBIENT TEMPERATURE
4.25
0.50
VFLT − Threshold Voltage − V
IDSG − Discharge Current − µA
0.47
VRTN = 48 V
VI(ISENS) = 80 mV
VO(FLTTIME) = 2 V
VRTN = 36 V to 80 V
0.44
0.41
0.38
0.35
0.32
4.13
4.00
3.88
0.29
0.26
−40
−15
10
35
60
85
−15
10
35
60
TA − Ambient Temperature − °C
TA − Ambient Temperature − °C
Figure 17
10
3.75
−40
Figure 18
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85
SLUS562A − JUNE 2003 − REVISED SEPTEMBER 2003
APPLICATION INFORMATION
When a plug-in module or printed circuit card is inserted into a live chassis slot, discharged supply bulk
capacitance on the board can draw huge transient currents from the system supplies. Without some form of
inrush limiting, these currents can reach peak magnitudes ranging up to several hundred amps, particularly in
high-voltage systems. Such large transients can damage connector pins, PCB etch, and plug-in and supply
components. In addition, current spikes can cause voltage droops on the power distribution bus, causing other
boards in the system to reset.
The TPS2398 and TPS2399 are hot swap power managers designed to limit these peaks to preset levels, as
well as control the slew rate (di/dt) at which charging current ramps to the user-programmed limit. These devices
use an external N-Channel pass FET and sense element to provide closed-loop control of current sourced to
the load. Input supply undervoltage lockout (UVLO) protection allows hot swap circuits to turn on automatically
with the application of power, or to be controlled with a system command via the EN input. External capacitors
control both the current ramp rate, and the time−out period for load voltage ramping. In addition, an internal
overload comparator provides circuit breaker protection against shorts occurring during steady-state
(post-turn-on) operation of the card.
The TPS2398 and TPS2399 operate directly from the input supply (nominal −48 VDC rail). The −VIN pin
connects to the negative voltage rail, and the RTN pin connects to the supply return. Internal regulators convert
input power to the supply levels required by the device circuitry. An input UVLO circuit holds the GATE output
low until the supply voltage reaches a nominal 30-V level. A second comparator monitors the EN input; this pin
must be pulled above the 1.4-V enable threshold to turn on power to the load.
Once enabled, and when the input supply is above the UVLO threshold, the GATE pull-down is removed, the
linear control amplifier (LCA) is enabled, and a large discharge device in the RAMP CONTROL block is turned
off. Subsequently, a small current source is now able to charge an external capacitor connected to the IRAMP
pin. This results in a linear voltage ramp at IRAMP. The voltage ramp on the capacitor actually has two discrete
slopes. As shown in Figure 17, charging current is supplied from either of two sources. Initially at turn-on, the
600-nA source is selected, to provide a slow turn-on rate. This slow turn-on helps ensure that the LCA is pulled
out of saturation, and is slewing to the voltage at its non-inverting input before normal rate load charging is
allowed. This mechanism helps reduce current steps at turn-on. Once the voltage at the IRAMP pin reaches
approximately 0.5 V, an internal comparator deasserts the SLOW signal, and the 10-µA source is selected for
the remainder of the ramp period.
The voltage at IRAMP is divided down by a factor of 100, and applied to the non-inverting input of the LCA. Load
current magnitude information at the ISENS pin is applied to the inverting input. This voltage is developed by
connecting the current sense resistor between ISENS and −VIN. The LCA slews the gate of the external pass
FET to force the ISENS voltage to track the divided down IRAMP voltage. Consequently, the load current slew
rate tracks the linear voltage ramp at the IRAMP pin, producing a linear di/dt of the load current. The IRAMP
capacitor is charged to about 6.5 V; however, the LCA input is clamped at 40 mV. Therefore, the current sourced
to the load during turn-on is limited to a value given by IMAX ≤ 40 mV/RSENSE, where RSENSE is the value of
the sense resistor.
The resultant load current, regulated by the controller, charges the module’s input bulk capacitance in a safe
fashion. Under normal conditions, this capacitance eventually charges up to the dc input potential. At this point,
the load demand drops off, and the voltage at ISENS decreases. The LCA now drives the GATE output to its
supply rail.
The device detects this condition as the GATE voltage rises through 7 V or 8 V, latches this status and asserts
the PG output. If the full sourced current limit is not yet available to the load, as evidenced by the IRAMP voltage
being less than 5 V, then the PG assertion is delayed until that condition is also met.
The peak, steady-state GATE pin output, typically 14 V, ensures sufficient overdrive to fully enhance the external
FET, while not exceeding the typical 20-V VGS rating of common N-channel power FETs.
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11
SLUS562A − JUNE 2003 − REVISED SEPTEMBER 2003
APPLICATION INFORMATION
GATEHI
Ramp Control
600 nA
10 µA
1
RAMPHI
S
PG
Q
FLT
SLOW
R Q
ENA
VDD
IRAMP 4
99 R
R
+
LCA
40 mV
OVERLOAD
COMPARATOR
ENA
EN 2
+
EN_AMP
1.4 V
7 GATE
OC
6 ISENS
+
OL
100 mV
50 µA
RTN 8
OC
4V
ON
+
+
30 V
FLT
VDD
Q
S
LATCH/
LOGIC RTRY
RST
3 FLTTIME
0.4 µA
+
0.5 V
14 V
−VIN
5
DCHG
TPS2399 ONLY
TIMER BLOCK
ENA
UDG−03068
Figure 19. Block Diagram
Fault timing is accomplished by connecting a capacitor between the FLTTIME and −VIN pins, allowing
user-programming of the timeout period. Whenever the hot swap controller is in current control mode as
described above, the LCA asserts an overcurrent indication (OC in the Figure 17 diagram). Overcurrent fault
timing is inhibited during the slow turn-on portion of the IRAMP waveform. However, once the device transitions
to the normal rate current ramp (VO(IRAMP) ≥ 0.5 V), the external capacitor is charged by a 50-µA source,
generating a voltage ramp at the FLTTIME pin. If the load voltage ramps successfully, the fault capacitor is
discharged (DCHG signal), and load initialization can begin. However, if the timing capacitor voltage attains the
4-V fault threshold, the LCA is disabled, the pass FET is rapidly turned off, and the fault is latched. Fault capacitor
charging ceases, and the capacitor is then discharged. In addition, latching of a fault condition causes rapid
discharge of the IRAMP capacitor. In this manner, the soft-start function is then reset and ready for the next
output enable, if and when conditions permit.
12
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SLUS562A − JUNE 2003 − REVISED SEPTEMBER 2003
APPLICATION INFORMATION
Subsequent to a plug-in’s start-up, and during the module’s steady-state operation, load faults that force current
limit operation also initiate fault timing cycles as described above. In this case, a fault timeout also clears the
previously latched power good status.
The TPS2398 latches off in response to faults; once a fault timeout occurs, the DCHG signal turns on a large
NMOS device to rapidly discharge the external capacitor, resetting the timer for any subsequent device reset.
The TPS2398 can only be reset by cycling power to the device, or by cycling the EN input.
In response to a latched fault condition, the TPS2399 enters a fault retry mode, wherein it periodically retries
the load to test for continued existence of the fault. In this mode, the FLTTIME capacitor is discharged slowly
by a about a 0.4-µA constant-current sink. When the voltage at the FLTTIME pin decays below 0.5 V, the LCA
and RAMP CONTROL circuits are re-enabled, and a normal turn-on current ramp ensues. Again, during the
load charging, the OC signal causes charging of the FLTTIME capacitor until the next delay period elapses. The
sequential charging and discharging of the FLTTIME capacitor results in a typical 1% retry duty cycle. If the fault
subsides, the timing capacitor is rapidly discharged, duty-cycle operation stops, and the PG output is asserted.
Note that because of the timing inhibit during the initial slow ramp period, the duty cycle in practice is slightly
greater than the nominal 1% value. However, sourced current during this period peaks at only about one-eighth
the maximum limit. The duty cycle of the normal ramp and constant-current periods is approximately 1%.
The FAULT LOGIC within the TIMER BLOCK automatically manages capacitor charge and discharge actions,
and the enabling of the GATE output (DCHG and ON signals).
supply transient response
The TPS2398 and TPS2399 also feature a fast-acting overload comparator which acts to clamp large transients
from catastrophic faults occurring once the pass FET is fully enhanced, such as short circuits. This function
provides a back-up protection to the LCA by providing a hard gate discharge action when the LCA is saturated.
If sense voltage excursions above 100 mV are detected, this comparator rapidly pulls down the GATE output,
bypassing the fault timer, and terminating the short−circuit condition. Once the spike has been brought down
below the overload threshold, the GATE output is released, allowing the circuit to turn on again in either
current-ramp or current-limit mode. A 4−µs deglitch filter is applied to the OL signal to help reduce the
occurrence of nuisance trips.
In redundant-supply systems, the sudden switchover to a supply of higher voltage potential is one more source
of large current spikes. Due to the low impedance of filter capacitance under such high-frequency transients,
these spikes are generally indistinguishable from true short-circuit faults to a hot swap controller. However, the
TPS2398and TPS2399 transient response addresses this issue by providing rapid circuit-breaker protection
for load faults along with minimal interruption of power flow during supply switching events. The scope plots in
Figure 20 illustrate how.
Figure 20 is a scope capture of the TPS2398/99 response in a diode-OR configuration to such an input transient
event. (All waveforms are referenced to the −VIN pin.) In this example, the module is initially operating from
a nominal −44-V supply (relative to the backplane supply return node). At the second major time division,
another power supply, with an output of −48 V, is suddenly hot swapped into a secondary, or INB, input. This
sudden voltage step is reflected in the −48V_RTN trace. On this board, the 4−V potential difference caused
a greater than 6-A spike, as shown by the IINB trace. The GATE pin is rapidly pulled low, which quickly terminates
the overload spike. However, it is quickly released, and seen to drive back to the pass FET ON-threshold, in
this case, about 4.5 V. The resultant current-limit operation of the circuit is evidenced by the 2-A load on the B
supply. Once supply current is flowing again, the filter capacitance is charged up to the new input supply level,
seen here on the VDRAIN trace. Once the capacitance is fully charged, the load demand rolls off to the operating
1-A level. As an added benefit, this event is transparent to the PG signal, which remains asserted throughout
the disturbance.
www.ti.com
13
SLUS562A − JUNE 2003 − REVISED SEPTEMBER 2003
APPLICATION INFORMATION
INPUT TRANSIENT RESPONSE
−48V_RTN (5 V/div)
(Offset 44 V)
CLOAD = 100 µF
RLOAD = 50 Ω
RSNS = 20 mΩ
CIRAMP = 3900 pF
IINB= (2 A/div)
GATE (5 V/div)
VDRAIN (5 V/div)
PG (50 V/div)
t − TIme − 100 µs/div
Figure 20
In order for downstream loads (bricks, etc.) to operate through the distribution bus transient, it is important to
properly size the filtering capacitance to supply the needed energy during the OFF-time of the pass FET. In this
example, once the RTN node stabilizes at about 3.5 V higher than the original potential, about 4.5 V develops
across the FET, indicating approximately a 1-V droop across the brick input. Therefore, due to the fast response
of the TPS2398/99 devices, the 100-µF capacitor achieves excellent hold-up of the brick input voltage. Actual
requirements depend heavily on the individual application. Whether the device turns back on in either
current-ramp or current-limit mode depends in part on the size of the ramp capacitor (CIRAMP) and the input
capacitance of the pass FET. But in any case, the circuit turns back on in a controlled-current manner after
rapidly clamping the potentially damaging spike.
14
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SLUS562A − JUNE 2003 − REVISED SEPTEMBER 2003
APPLICATION INFORMATION
setting the sense resistor value
Due to the current-limiting action of the internal LCA, the maximum allowable load current for an implementation
is easily programmed by selecting the appropriate sense resistor value. The LCA acts to limit the sense voltage
VI(ISENS) to its internal reference. Once the voltage at the IRAMP pin exceeds approximately 4 V, this limit is
the clamp voltage, VREF_K. Therefore, a maximum sense resistor value can be determined from equation (1).
R SENSE v 33 mV
IMAX
(1)
where:
•
•
RSENSE is the resistor value, and
IMAX is the desired current limit.
When setting the sense resistor value, it is important to consider two factors, the minimum current that may be
imposed by the TPS2398 or TPS2399, and the maximum load under normal operation of the module. For the
first factor, the specification minimum clamp value is used, as seen in equation (1). This method accounts for
the tolerance in the sourced current limit below the typical level expected (40 mV/RSENSE). (The clamp
measurement includes LCA input offset voltage; therefore, this offset does not have to be factored into the
current limit again.) Second, if the load current varies over a range of values under normal operating conditions,
then the maximum load level must be allowed for by the value of RSENSE. One example of this is when the load
is a switching converter, or brick, which draws higher input current, for a given power output, when the
distribution bus is at the low end of its operating range, with decreasing draw at higher supply voltages. To avoid
current-limit operation under normal loading, some margin should be designed in between this maximum
anticipated load and the minimum current limit level, or IMAX > ILOAD(max), for equation (1).
For example, using a 20-mΩ sense resistor for a nominal 1-A load application provides a minimum of 650 mA
of overhead for load variance/margin. Typical bulk capacitor charging current during turn-on is 2 A
(40 mV/20 mΩ).
setting the inrush slew rate
The TPS2398 and TPS2399 devices enable user-programming of the maximum current slew rate during load
start-up events. A capacitor tied to the IRAMP pin (C2 in the typical application diagram) controls the di/dt rate.
Once the sense resistor value has been established, a value for ramp capacitor CIRAMP, in microfarads, can be
determined from equation (2).
C IRAMP +
11
100
R SENSE
ǒdtdiǓ
(2)
MAX
where:
•
•
RSENSE is in ohms, and
(di/dt)MAX is the desired maximum slew rate, in amperes/second.
For example, if the desired slew rate for the typical application shown is 1500 mA/ms, the calculated value for
CIRAMP is about 3700 pF. Selecting the next larger standard value of 3900 pF (as shown in the diagram) provides
some margin for capacitor and sense resistor tolerances.
www.ti.com
15
SLUS562A − JUNE 2003 − REVISED SEPTEMBER 2003
APPLICATION INFORMATION
As described earlier in this section, the TPS2398 and TPS2399 initiate ramp capacitor charging, and
consequently, load current di/dt at a reduced rate. This reduced rate applies until the voltage on the IRAMP pin
is about 0.5 V. The maximum di/dt rate, as set by equation (2), is effective once the device has switched to the
10-µA charging source.
setting the fault timing capacitor
The fault timeout period is established by the value of the capacitor connected to the FLTTIME pin, CFLT. The
timeout period permits riding out spurious current glitches and surges that may occur during operation of the
system, and prevents indefinite sourcing into faulted loads swapped into a live system. However, to ensure
smooth voltage ramping under all conditions of load capacitance and input supply potential, the minimum
timeout should be set to accommodate these system variables. To do this, a rough estimate of the maximum
voltage ramp time for a completely discharged plug-in card provides a good basis for setting the minimum timer
delay.
Due to the three-phase nature of the load current at turn-on, the load voltage ramp potentially has three distinct
phases ( compare Figures 1 and 2). This profile depends on the relative values of load capacitance, input dc
potential, maximum current limit and other factors. The first two phases are characterized by the two different
slopes of the current ramp; the third phase, if required for bulk capacitance charging, is the constant-current
charging at IMAX. Considering the two current ramp phases to be one period at an average di/dt simplifies
calculation of the required timing capacitor.
For the TPS2398 and TPS2399, the typical duration of the soft-start ramp period, tSS, is given by equation (3).
t SS + 1183
C IRAMP
(3)
where:
•
•
tSS is the soft-start period in ms, and
CIRAMP is given in µF
During this current ramp period, the load voltage magnitude which is attained is estimated by equation (4).
V LSS +
i AVG
2
C LOAD
C IRAMP
100
R SENSE
ǒtSSǓ
2
(4)
where:
•
•
•
•
VLSS is the load voltage reached during soft-start,
iAVG is 3.38 µA for the TPS2398 and TPS2399,
CLOAD is the amount of the load capacitance, and
tSS is the soft-start period, in seconds
The quantity iAVG in equation (4) is a weighted average of the two charge currents applied to CIRAMP during
turn-on, considering the typical output values.
16
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SLUS562A − JUNE 2003 − REVISED SEPTEMBER 2003
APPLICATION INFORMATION
If the result of equation (4) is larger than the maximum input supply value, then the load can be expected to
charge completely during the inrush slewing portion of the insertion event. However, if this voltage is less than
the maximum supply input, VIN(max), the HSPM transitions to the constant-current charging of the load. The
remaining amount of time required at IMAX is determined from equation (5).
t CC +
C LOAD
ǒ
ǒVIN (max) * VLSSǓ
V
REF_K (min)
R
SENSE
Ǔ
(5)
where:
•
•
tCC is the constant-current voltage ramp time, in seconds, and
VREF_K(min) is the minimum clamp voltage, 33 mV.
With this information, the minimum recommended value timing capacitor CFLT can be determined. The delay
time needed will be either a time tSS2 or the sum of tSS2 and tCC, according to the estimated time to charge the
load. The quantity tSS2 is the duration of the normal rate current ramp period, and is given by equation (6).
t
SS2
+ 0.35
C
RAMP
(6)
where:
•
CRAMP is given in microfarads
Since fault timing is generated by the constant−current charging of CFLT, the capacitor value is determined from
either equation (7) or (8), as appropriate.
C
C
FLT(min)
+
FLT(min)
+
55
55
t
SS2
3.75
(7)
ǒtSS2 ) tCCǓ
3.75
(8)
where:
•
•
•
CFLT(min) is the recommended capacitor value, in microfarads,
tSS2 is the result of equation (6), in seconds, and
tCC is the result of equation (5), in seconds.
Continuing the typical application example, using a 100-µF input capacitor (CLOAD), equations (3) and (4)
estimate the load voltage ramping to approximately −46 V during the soft-start period. If the module should
operate down to −72 V input supply, approximately another 1.58-ms of constant-current charging may be
required. Therefore, equations (6) and (8) are used to determine CFLT(min), and the result of 0.043-µF suggests
the 0.047-µF standard value.
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17
PACKAGE OPTION ADDENDUM
www.ti.com
30-Mar-2005
PACKAGING INFORMATION
Orderable Device
Status (1)
Package
Type
Package
Drawing
Pins Package Eco Plan (2)
Qty
TPS2398DGK
ACTIVE
MSOP
DGK
8
TPS2398DGKR
ACTIVE
MSOP
DGK
TPS2398DGKRG4
ACTIVE
MSOP
TPS2399DGK
ACTIVE
TPS2399DGKR
ACTIVE
80
Lead/Ball Finish
MSL Peak Temp (3)
Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-1-260C-UNLIM
8
2500 Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-1-260C-UNLIM
DGK
8
2500 Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-1-260C-UNLIM
MSOP
DGK
8
80
TBD
CU NIPDAU
Level-1-220C-UNLIM
MSOP
DGK
8
2500
TBD
CU NIPDAU
Level-1-220C-UNLIM
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS) or Green (RoHS & no Sb/Br) - please check
http://www.ti.com/productcontent for the latest availability information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements
for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered
at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame
retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder
temperature.
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Addendum-Page 1
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