TI TPS54218

TPS54218
www.ti.com ......................................................................................................................................................................................... SLVS974 – SEPTEMBER 2009
2.95 V to 6 V Input, 2 A Output, 2MHz, Synchronous Step Down
Switcher With Integrated FETs ( SWIFT™)
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FEATURES
DESCRIPTION
•
The TPS54218 device is a full featured 6 V, 2 A,
synchronous step down current mode converter with
two integrated MOSFETs.
1
•
•
•
•
•
•
•
•
•
•
Two 30 mΩ (typical) MOSFETs for High
Efficiency at 2 A loads
200kHz to 2MHz Switching Frequency
0.8 V ± 1% Voltage Reference Over
Temperature
Synchronizes to External Clock
Adjustable Slow Start/Sequencing
UV and OV Power Good Output
Low Operating and Shutdown Quiescent
Current
Safe Start-up into Pre-Biased Output
Cycle by Cycle Current Limit, Thermal and
Frequency Fold Back Protection
–40°C to 150°C Operating Junction
Temperature Range
Thermally Enhanced 3mm × 3mm 16-pin QFN
APPLICATIONS
•
•
•
Low-Voltage, High-Density Power Systems
Point of Load Regulation for High Performance
DSPs, FPGAs, ASICs and Microprocessors
Broadband, Networking and Optical
Communications Infrastructure
Efficiency is maximized through the integrated 30mΩ
MOSFETs and 350μA typical supply current. Using
the enable pin, shutdown supply current is reduced to
2 μA by entering a shutdown mode.
Undervoltage lockout is internally set at 2.6 V, but
can be increased by programming the threshold with
a resistor network on the enable pin. The output
voltage startup ramp is controlled by the slow start
pin. An open drain power good signal indicates the
output is within 93% to 107% of its nominal voltage.
Frequency fold back and thermal shutdown protects
the device during an overcurrent condition.
For more SWIFTTM documentation, see the TI
website at www.ti.com/swift.
CBOOT
VIN
The TPS54218 provides accurate regulation for a
variety of loads with an accurate ±1% Voltage
Reference (Vref) over temperature.
The TPS54218 is supported in the SwitcherProTM
Software Tool at www.ti.com/switcherpro.
SIMPLIFIED SCHEMATIC
VIN
The TPS54218 enables small designs by integrating
the MOSFETs, implementing current mode control to
reduce external component count, reducing inductor
size by enabling up to 2 MHz switching frequency,
and minimizing the IC footprint with a small 3mm x
3mm thermally enhanced QFN package.
BOOT
R4
CI
TPS54218
EN
LO
VOUT
95
CO
R5
VSENSE
SS
RT /CLK
COMP
GND
AGND
POWERPAD
C ss
90
R1
PWRGD
RT
100
PH
R3
C1
R2
Efficiency - %
2
85
80
75
70
65
VI = 5 V,
VO = 1.8 V,
FS = 1000 kHz
60
55
50
0
0.25
0.5
0.75
1
1.25 1.5
IO - Output Current - A
1.75
2
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
SWIFT is a trademark of Texas Instruments.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2009, Texas Instruments Incorporated
TPS54218
SLVS974 – SEPTEMBER 2009 ......................................................................................................................................................................................... www.ti.com
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
ORDERING INFORMATION
(1)
TJ
PACKAGE
–40°C to 150°C
3 × 3 mm QFN
PART NUMBER
(1)
TPS54218RTET
TPS54218RTER
The RTE package is only available taped and reeled. To order, add the suffix “R” to the end of the part
number for a reel of 3000, or add the suffix “T” to the end of the part number for a reel of 250 (e.g.,
TPS54218RTER).
ABSOLUTE MAXIMUM RATINGS
VALUE
UNIT
VIN
–0.3 to 7
V
EN
–0.3 to 7
BOOT
Input voltage
PH + 8
VSENSE
–0.3 to 3
COMP
–0.3 to 3
PWRGD
–0.3 to 7
SS
–0.3 to 3
RT/CLK
–0.3 to 6
BOOT-PH
Output voltage
8
PH
PH 10 ns Transient
Source current
Sink current
V
–0.6 to 7
–2 to 7
EN
100
μA
RT/CLK
100
μA
COMP
100
μA
PWRGD
10
mA
SS
100
μA
Electrostatic discharge (HBM)
2
kV
Electrostatic discharge (CDM)
500
V
Operating Junction temperature, Tj
–40 to 150
°C
Storage temperature, Tstg
–65 to 150
°C
PACKAGE DISSIPATION RATINGS (1)
(2) (3)
over operating free-air temperature range (unless otherwise noted)
(1)
(2)
(3)
2
PACKAGE
THERMAL IMPEDANCE
JUNCTION TO AMBIENT
φJT THERMAL CHARACTERISTIC
JUNCTION TO TOP
RTE
37°C/W
1°C/W
Maximum power dissipation may be limited by overcurrent protection
Power rating at a specific ambient temperature TA should be determined with a junction temperature of 150°C. This is the point where
distortion starts to substantially increase. Thermal management of the PCB should strive to keep the junction temperature at or below
150°C for best performance and long-term reliability. See power dissipation estimate in application section of this data sheet for more
information.
Test boards conditions:
(a) 2 inches x 2 inches, 4 layers, thickness: 0.062 inch
(b) 2 oz. copper traces located on the top of the PCB
(c) 2 oz. copper ground planes on the 2 internal layers and bottom layer
(d) 4 thermal vias (10mil) located under the device package
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ELECTRICAL CHARACTERISTICS
TJ = –40°C to 150°C, VIN = 2.95 to 6 V (unless otherwise noted)
DESCRIPTION
CONDITIONS
MIN
TYP
MAX
UNIT
2.6
2.8
V
2
5
μA
350
500
μA
1.25
1.37
V
SUPPLY VOLTAGE (VIN PIN)
Operating input voltage
2.95
Internal under voltage lockout threshold
No voltage hysteresis, rising and falling
Shutdown supply current
EN = 0 V, 25°C, 2.95 V ≤ VIN ≤ 6 V
Quiescent Current - Iq
VSENSE = 0.9 V, VIN = 5 V, 25°C, RT = 400 kΩ
6
V
ENABLE AND UVLO (EN PIN)
Enable threshold
Input current
Rising
1.16
Falling
1.18
Enable rising threshold + 50 mV
-3.2
Enable falling threshold – 50 mV
-0.65
μA
VOLTAGE REFERENCE (VSENSE PIN)
Voltage Reference
2.95 V ≤ VIN ≤ 6 V, –40°C <TJ < 150°C
0.795
0.803
0.811
BOOT-PH = 5 V
30
60
BOOT-PH = 2.95 V
44
70
VIN = 5 V
30
60
VIN = 2.95 V
44
70
V
MOSFET
High side switch resistance
Low side switch resistance
mΩ
mΩ
ERROR AMPLIFIER
Input current
7
nA
Error amplifier transconductance (gm)
–2 μA < I(COMP) < 2 μA, V(COMP) = 1 V
225
μmhos
Error amplifier transconductance (gm) during
slow start
–2 μA < I(COMP) < 2 μA, V(COMP) = 1 V,
VSENSE = 0.4 V
70
μmhos
Error amplifier source/sink
V(COMP) = 1 V, 100 mV overdrive
COMP to Iswitch gm
±20
μA
13
A/V
3.6
A
175
°C
15
°C
CURRENT LIMIT
Current limit threshold
2.9
THERMAL SHUTDOWN
Thermal Shutdown
Hysteresis
TIMING RESISTOR AND EXTERNAL CLOCK (RT/CLK PIN)
Switching frequency range using RT mode
Switching frequency
200
RT = 400 kΩ
Switching frequency range using CLK mode
500
300
Minimum CLK pulse width
RT/CLK voltage
400
2000
kHz
600
kHz
2000
kHz
75
R(RT/CLK) = 400kΩ
ns
0.5
RT/CLK high threshold
1.6
RT/CLK low threshold
0.4
V
2.2
V
0.6
V
RT/CLK falling edge to PH rising edge delay
Measure at 500 kHz with RT resistor in series
90
ns
PLL lock in time
Measure at 500 kHz
14
μs
Measured at 50% points on PH, IOUT = 2 A
60
ns
Measured at 50% points on PH, VIN = 5 V,
IOUT = 0 A
110
PH (PH PIN)
Minimum On time
Minimum Off time
Prior to skipping off pulses, BOOT-PH = 2.95 V,
IOUT = 2 A
60
ns
Rise/Fall Time
VIN = 5 V
1.5
V/ns
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ELECTRICAL CHARACTERISTICS (continued)
TJ = –40°C to 150°C, VIN = 2.95 to 6 V (unless otherwise noted)
DESCRIPTION
CONDITIONS
MIN
TYP
MAX
UNIT
BOOT (BOOT PIN)
BOOT Charge Resistance
VIN = 5 V
16
Ω
BOOT-PH UVLO
VIN = 2.95 V
2.1
V
Charge Current
V(SS) = 0.4 V
1.8
μA
SS to reference crossover
98% nominal
0.9
V
SS discharge voltage (overload)
VSENSE = 0 V
20
μA
SS discharge current (UVLO, EN, Thermal
Fault)
VIN = 5 V, V(SS) = 0.5 V
1.25
mA
SLOW START (SS PIN)
POWER GOOD (PWRGD PIN)
VSENSE threshold
VSENSE falling (Fault)
91
% Vref
VSENSE rising (Good)
93
% Vref
VSENSE rising (Fault)
107
% Vref
VSENSE falling (Good)
105
% Vref
2
% Vref
2
nA
100
Ω
Hysteresis
VSENSE falling
Output high leakage
VSENSE = VREF, V(PWRGD) = 5.5 V
On resistance
Output low
I(PWRGD) = 3.5 mA
0.3
Minimum VIN for valid output
V(PWRGD) < 0.5 V at 100 μA
1.2
4
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V
1.6
V
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PIN CONFIGURATION
2
GND
3
GND
4
EN
PWRGD
BOOT
14
13
Thermal
Pad
(17)
AGND
5
6
7
8
RT/CLK
VIN
15
COMP
1
16
VSENSE
VIN
VIN
QFN16
RTE PACKAGE
(TOP VIEW)
12
PH
11
PH
10
PH
9
SS
PIN FUNCTIONS
PIN
DESCRIPTION
NAME
NO.
AGND
5
Analog Ground should be electrically connected to GND close to the device.
BOOT
13
A bootstrap capacitor is required between BOOT and PH. If the voltage on this capacitor is below the minimum
required by the BOOT UVLO, the output is forced to switch off until the capacitor is refreshed.
COMP
7
Error amplifier output, and input to the output switch current comparator. Connect frequency compensation
components to this pin.
EN
15
Enable pin, internal pull-up current source. Pull below 1.2 V to disable. Float to enable. Can be used to set the
on/off threshold (adjust UVLO) with two additional resistors.
GND
3, 4
Power Ground. This pin should be electrically connected directly to the power pad under the IC.
PH
10, 11,
12
The source of the internal high side power MOSFET, and drain of the internal low side (synchronous) rectifier
MOSFET.
PowerPAD
17
GND pin should be connected to the exposed power pad for proper operation. This power pad should be
connected to any internal PCB ground plane using multiple vias for good thermal performance.
PWRGD
14
An open drain output, asserts low if output voltage is low due to thermal shutdown, overcurrent,
over/under-voltage or EN shut down.
RT/CLK
8
Resistor Timing or External Clock input pin.
SS
9
Slow-start. An external capacitor connected to this pin sets the output voltage rise time.
VIN
1, 2, 16
VSENSE
6
Input supply voltage, 2.95 V to 6 V.
Inverting node of the transconductance (gm) error amplifier.
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FUNCTIONAL BLOCK DIAGRAM
PWRGD
EN
VIN
i1
Shutdown
93%
ihys
Thermal
Shutdown
Enable
Comparator
Logic
UVLO
Shutdown
Shutdown
Logic
107%
Enable
Threshold
Boot
Charge
Voltage
Reference
Boot
UVLO
Minimum
COMP Clamp
ERROR
AMPLIFIER
Current
Sense
PWM
Comparator
VSENSE
SS
BOOT
Logic and PWM
Latch
Shutdown
Logic
S
COMP
Slope
Compensation
PH
Frequency
Shift
Overload
Recovery
Maximum
Clamp
Oscillator
with PLL
GND
TPS54218
AGND
6
POWERPAD
RT/CLK
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TYPICAL CHARACTERISTICS CURVES
FREQUENCY vs TEMPERATURE
550
0.055
High Side Rdson
VI = 3.3 V
Low Side Rdson
VI = 3.3 V
0.05
0.045
High Side Rdson
VI = 5 V
0.04
0.035
Low Side Rdson
VI = 5 V
0.03
RT = 400 kW,
VI = 3.3 V
540
fs - Switching Frequency - kHz
RDSON - Static Drain-Source On-State Resistance - W
HIGH SIDE AND LOW SIDE RDS (on) vs TEMPERATURE
530
520
510
500
490
480
470
0.025
460
0.02
-50
-25
0
25
50
75
100
TJ - Junction Temperature - °C
125
450
-50
150
-25
0
25
50
75
100
TJ - Junction Temperature - °C
Figure 1.
HIGH SIDE CURRENT LIMIT vs TEMPERATURE
VOLTAGE REFERENCE vs TEMPERATURE
0.808
3.9
0.806
3.8
Vref - Voltage Reference - V
High Side Switching Current - A
150
Figure 2.
4
3.7
VI = 3 V
3.6
3.5
3.4
3.3
0.804
VI = 3.3 V
0.802
0.8
0.798
0.796
3.2
0.794
3.1
3
-50
-25
0
25
50
75
100
TJ - Junction Temperature - °C
125
0.792
-50
150
-25
0
25
50
75
100
TJ - Junction Temperature - °C
125
Figure 3.
Figure 4.
SWITCHING FREQUENCY vs
RT RESISTANCE LOW FREQUENCY RANGE
SWITCHING FREQUENCY vs
RT RESISTANCE HIGH FREQUENCY RANGE
1900
fs - Switching Frequency - kHz
900
800
700
600
500
400
300
200
100
150
2000
1000
fs - Switching Frequncy - kHz
125
1800
1700
1600
1500
1400
1300
1200
1100
200
300
400
500
600
700
RT - Resistance - kW
800
900
1000
1000
80
Figure 5.
100
120
140
160
RT - Resistance kW
180
200
Figure 6.
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TYPICAL CHARACTERISTICS CURVES (continued)
SWITCHING FREQUENCY vs VSENSE
TRANSCONDUCTANCE vs TEMPERATURE
100
280
VIN = 3 .3 V
260
EA - Transconductance - mA/V
Normal Switching Frequency - %
Vsense Falling
75
Vsense Rising
50
25
240
220
200
180
160
0
0
0.1
0.2
0.3
0.4
0.5
Vsense - V
0.6
0.7
140
-50
0.8
-25
0
25
50
75
100
TJ - Junction Temperature - °C
Figure 7.
150
Figure 8.
TRANSCONDUCTANCE (SLOW START) vs
JUNCTION TEMPERATURE
ENABLE PIN VOLTAGE vs TEMPERATURE
1.3
90
85
125
VIN = 3 .3 V
1.29
1.28
1.26
75
EN - Threshold - V
EA - Transconductance - mA/V
VIN = 3.3 V, rising
1.27
80
70
65
60
55
1.25
1.24
1.23
1.22
1.21
1.2
VIN = 3.3 V, falling
1.19
1.18
50
1.17
45
40
-50
-25
0
25
50
75
100
TJ - Junction Temperature - °C
125
1.16
1.15
-50
150
-25
0
25
50
75
100
TJ - Junction Temperature - °C
Figure 9.
PIN CURRENT vs TEMPERATURE
-0.35
-0.45
-3.05
-0.55
EN - Pin Current - mA
EN Pin Current - mA
VIN = 5 V,
VEN = Rising Threshold +50 mV
-3.15
-3.25
-3.35
-3.45
150
-0.65
-0.75
-0.85
-0.95
-3.55
-1.05
-3.65
-1.15
-25
0
25
50
75
100
TJ - Junction Temperature - °C
125
150
VIN = 5 V,
VEN = Falling Threshold -50 mV
-1.25
-50
-25
Figure 11.
8
125
PIN CURRENT vs TEMPERATURE
-0.25
-2.95
-3.75
-50
150
Figure 10.
-2.75
-2.85
125
0
25
50
75
100
TJ - Junction Temperature - °C
Figure 12.
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TYPICAL CHARACTERISTICS CURVES (continued)
CHARGE CURRENT vs TEMPERATURE
DISCHARGE CURRENT vs TEMPERATURE
-1
105
-1.2
103
SS/TR - Discharge Current - mA
SS/TR - Charge Current - mA
-1.4
VIN = 5 V
-1.6
-1.8
-2
-2.2
-2.4
-2.6
-2.8
-3
-50
101
97
95
93
91
89
87
-25
0
25
50
75
100
TJ - Junction Temperature - °C
125
85
-50
150
-25
0
25
50
75
100
TJ - Junction Temperature - °C
125
150
Figure 13.
Figure 14.
INPUT VOLTAGE vs TEMPERATURE
SHUTDOWN SUPPLY CURRENT vs TEMPERATURE
3
3
VIN = 3.3 V
2.9
2.5
2.7
Shutdown Supply Current - mA
2.8
VIN - Input Voltage - V
VIN = 5 V
99
UVLO Start Switching
2.6
2.5
UVLO Stop Switching
2.4
2.3
2.2
2
1.5
1
0.5
2.1
2
-50
-25
0
25
50
75
100
TJ - Junction Temperature - °C
125
0
-50
150
-25
0
25
50
75
100
TJ - Junction Temperature - °C
Figure 15.
125
150
125
150
Figure 16.
SHUTDOWN SUPPLY CURRENT vs INPUT VOLTAGE
SUPPLY CURRENT vs TEMPERATURE
3
400
TJ = 25°C
390
VIN = 3.3 V
380
ICC - Supply Current - mA
Shutdown Supply Current mA
2.5
2
1.5
1
370
360
350
340
330
320
0.5
310
0
3
3.5
4
4.5
5
VIN - Input Voltage - V
5.5
6
300
-50
-25
Figure 17.
0
25
50
75
100
TJ - Junction Temperature - °C
Figure 18.
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TYPICAL CHARACTERISTICS CURVES (continued)
SUPPLY CURRENT vs INPUT VOLTAGE
PWRGD THRESHOLD vs TEMPERATURE
110
400
108
380
106
PWRGD - Threshold - % Vref
ICC - Supply Current - mA
TJ = 25°C
390
370
360
350
340
330
320
Vsense Rising, VIN = 3.3 V
Vsense Falling
104
102
100
98
96
94
Vsense Rising
92
310
90
300
88
-50
3
3.5
4
4.5
5
VIN - Input Voltage - V
5.5
6
Vsense Falling
-25
0
Figure 19.
25
50
75
100
TJ - Junction Temperature - °C
125
150
Figure 20.
RDSON - Static Drain-Sourec On State Resistance - W
PWRGD ON RESISTANCE vs TEMPERATURE
200
180
VIN = 3.3 V
160
140
120
100
80
60
40
20
0
-50
-25
0
25
50
75
100
TJ - Junction Temperature - °C
125
150
Figure 21.
OVERVIEW
The TPS54218 is a 6-V, 2 A, synchronous step-down (buck) converter with two integrated n-channel MOSFETs.
To improve performance during line and load transients the device implements a constant frequency, peak
current mode control which reduces output capacitance and simplifies external frequency compensation design.
The wide switching frequency of 200 kHz to 2000 kHz allows for efficiency and size optimization when selecting
the output filter components. The switching frequency is adjusted using a resistor to ground on the RT/CLK pin.
The device has an internal phase lock loop (PLL) on the RT/CLK pin that is used to synchronize the power
switch turn on to a falling edge of an external system clock.
The TPS54218 has a typical default start up voltage of 2.6 V. The EN pin has an internal pull-up current source
that can be used to adjust the input voltage under voltage lockout (UVLO) with two external resistors. In addition,
the pull up current provides a default condition when the EN pin is floating for the device to operate. The total
operating current for the TPS54218 is 350 μA when not switching and under no load. When the device is
disabled, the supply current is less than 5 μA.
The integrated 30 mΩ MOSFETs allow for high efficiency power supply designs with continuous output currents
up to 2 amperes.
10
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The TPS54218 reduces the external component count by integrating the boot recharge diode. The bias voltage
for the integrated high side MOSFET is supplied by a capacitor on the BOOT to PH pin. The boot capacitor
voltage is monitored by an UVLO circuit and turns off the high side MOSFET when the voltage falls below a
preset threshold. This BOOT circuit allows the TPS54218 to operate approaching 100%. The output voltage can
be stepped down to as low as the 0.8 V reference.
The TPS54218 has a power good comparator (PWRGD) with 2% hysteresis.
The TPS54218 minimizes excessive output overvoltage transients by taking advantage of the overvoltage power
good comparator. When the regulated output voltage is greater than 109% of the nominal voltage, the
overvoltage comparator is activated, and the high side MOSFET is turned off and masked from turning on until
the output voltage is lower than 105%.
The SS (slow start) pin is used to minimize inrush currents or provide power supply sequencing during power up.
A small value capacitor should be coupled to the pin for slow start. The SS pin is discharged before the output
power up to ensure a repeatable restart after an over-temperature fault, UVLO fault or disabled condition.
The use of a frequency foldback circuit reduces the switching frequency during startup and over current fault
conditions to help limit the inductor current.
DETAILED DESCRIPTION
FIXED FREQUENCY PWM CONTROL
The TPS54218 uses an adjustable fixed frequency, peak current mode control. The output voltage is compared
through external resistors on the VSENSE pin to an internal voltage reference by an error amplifier which drives
the COMP pin. An internal oscillator initiates the turn on of the high side power switch. The error amplifier output
is compared to the high side power switch current. When the power switch current reaches the COMP voltage
level the high side power switch is turned off and the low side power switch is turned on. The COMP pin voltage
increases and decreases as the output current increases and decreases. The device implements a current limit
by clamping the COMP pin voltage to a maximum level and also implements a minimum clamp for improved
transient response performance.
SLOPE COMPENSATION AND OUTPUT CURRENT
The TPS54218 adds a compensating ramp to the switch current signal. This slope compensation prevents
sub-harmonic oscillations as duty cycle increases. The available peak inductor current remains constant over the
full duty cycle range.
BOOTSTRAP VOLTAGE (BOOT) AND LOW DROPOUT OPERATION
The TPS54218 has an integrated boot regulator and requires a small ceramic capacitor between the BOOT and
PH pin to provide the gate drive voltage for the high side MOSFET. The value of the ceramic capacitor should be
0.1 μF. A ceramic capacitor with an X7R or X5R grade dielectric with a voltage rating of 10 V or higher is
recommended because of the stable characteristics overtemperature and voltage.
To improve drop out, the TPS54218 is designed to operate at 100% duty cycle as long as the BOOT to PH pin
voltage is greater than 2.5 V. The high side MOSFET is turned off using an UVLO circuit, allowing for the low
side MOSFET to conduct when the voltage from BOOT to PH drops below 2.5 V. Since the supply current
sourced from the BOOT pin is low, the high side MOSFET can remain on for more switching cycles than are
required to refresh the capacitor, thus the effective duty cycle of the switching regulator is high.
ERROR AMPLIFIER
The TPS54218 has a transconductance amplifier. The error amplifier compares the VSENSE voltage to the lower
of the SS pin voltage or the internal 0.8 V voltage reference. The transconductance of the error amplifier is
225 μA/V during normal operation. When the voltage of VSENSE pin is below 0.8 V and the device is regulating
using the SS voltage, the gm is 70 μA/V. The frequency compensation components are placed between the
COMP pin and ground.
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VOLTAGE REFERENCE
The voltage reference system produces a precise ±1% voltage reference overtemperature by scaling the output
of a temperature stable bandgap circuit. The bandgap and scaling circuits produce 0.8 V at the non-inverting
input of the error amplifier.
ADJUSTING THE OUTPUT VOLTAGE
The output voltage is set with a resistor divider from the output node to the VSENSE pin. It is recommended to
use divider resistors with 1% tolerance or better. Start with a 100 kΩ for the R1 resistor and use the Equation 1
to calculate R2. To improve efficiency at very light loads consider using larger value resistors. If the values are
too high the regulator is more susceptible to noise and voltage errors from the VSENSE input current are
noticeable.
æ
ö
0.8 V
R2 = R1 ´ ç
÷
è VO - 0.8 V ø
(1)
TPS54218
VO
R1
VSENSE
R2
0.8 V
+
Figure 22. Voltage Divider Circuit
ENABLE AND ADJUSTING UNDER-VOLTAGE LOCKOUT
The TPS54218 is disabled when the VIN pin voltage falls below 2.6 V. If an application requires a higher
under-voltage lockout (UVLO), use the EN pin as shown in Figure 23 to adjust the input voltage UVLO by using
two external resistors. It is recommended to use the enable resistors to set the UVLO falling threshold (VSTOP)
above 2.7 V. The rising threshold (VSTART) should be set to provide enough hysteresis to allow for any input
supply variations. The EN pin has an internal pull-up current source that provides the default condition of the
TPS54218 operating when the EN pin floats. Once the EN pin voltage exceeds 1.25 V, an additional 2.55 μA of
hysteresis is added. When the EN pin is pulled below 1.18 V, the 2.55 μA is removed. This additional current
facilitates input voltage hysteresis.
TPS54218
i hys
VIN
2.55 mA
i1
R1
0.6 mA
R2
EN
+
-
Figure 23. Adjustable Under Voltage Lock Out
12
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R1 =
R2 =
0.944 × VSTART - VSTOP
2.59 ´ 10-6
(2)
1.18 × R1
VSTOP - 1.18 + R1 × 3.2 ´ 10 - 6
(3)
SLOW START PIN
The TPS54218 regulates to the lower of the SS pin and the internal reference voltage. A capacitor on the SS pin
to ground implements a slow start time. The TPS54218 has an internal pull-up current source of 1.8μA which
charges the external slow start capacitor. Equation 4 calculates the required slow start capacitor value where Tss
is the desired slow start time in ms, ISS is the internal slow start charging current of 1.8 μA, and Vref is the
internal voltage reference of 0.8 V.
T (mS) ´ ISS (mA)
CSS (nF) = SS
Vref (V)
(4)
If during normal operation, the VIN goes below the UVLO, EN pin pulled below 1.2 V, or a thermal shutdown
event occurs, the TPS54218 stops switching and the SS is discharged to 0 volts before reinitiating a powering up
sequence.
SEQUENCING
Many of the common power supply sequencing methods can be implemented using the SS, EN and PWRGD
pins. The sequential method can be implemented using an open drain or collector output of a power on reset pin
of another device. Figure 24 shows the sequential method. The power good is coupled to the EN pin on the
TPS54218 which enables the second power supply once the primary supply reaches regulation.
Ratiometric start up can be accomplished by connecting the SS pins together. The regulator outputs ramp up
and reach regulation at the same time. When calculating the slow start time the pull up current source must be
doubled in Equation 4. The ratiometric method is illustrated in Figure 26.
TPS54218
PWRGD
EN1 = 2 V / div
EN
EN
SS
SS
PWRGD1 = 2 V / div
Vout1 = 1 / div
PWRGD
Vout2 = 1 V / div
Time = 5 msec / div
Figure 24. Sequential Start-Up Sequence
Figure 25. Sequential Startup using EN and
PWRGD
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TPS54218
EN1 = 2 V / div
EN
SS
Vout1 = 1 V / div
PWRGD
Vout2 = 1 V / div
TPS54218
EN
Time = 5 msec / div
SS
PWRGD
Figure 26. Schematic for Ratiometric Start-Up
Sequence
Figure 27. Ratiometric Start-Up using Coupled SS
Pins
CONSTANT SWITCHING FREQUENCY and TIMING RESISTOR (RT/CLK Pin)
The switching frequency of the TPS54218 is adjustable over a wide range from 200 kHz to 2000 kHz by placing
a maximum of 1000kΩ and minimum of 85 kΩ, respectively, on the RT/CLK pin. An internal amplifier holds this
pin at a fixed voltage when using an external resistor to ground to set the switching frequency. The RT/CLK is
typically 0.5 V. To determine the timing resistance for a given switching frequency, use the curve in Figures 5
and 6, or Equation 5.
311890
RT (kW) =
Fsw(kHz)1.0793
(5)
Fsw(kHz) =
133870
RT(kW)0.9393
(6)
To reduce the solution size one would typically set the switching frequency as high as possible, but tradeoffs of
the efficiency, maximum input voltage and minimum controllable on time should be considered.
The minimum controllable on time is typically 60 ns at full current load and 110 ns at no load, and limits the
maximum operating input voltage or output voltage.
OVERCURRENT PROTECTION
The TPS54218 implements a cycle by cycle current limit. During each switching cycle the high side switch
current is compared to the voltage on the COMP pin. When the instantaneous switch current intersects the
COMP voltage, the high side switch is turned off. During overcurrent conditions that pull the output voltage low,
the error amplifier responds by driving the COMP pin high, increasing the switch current. The error amplifier
output is clamped internally. This clamp functions as a switch current limit.
FREQUENCY SHIFT
To operate at high switching frequencies and provide protection during overcurrent conditions, the TPS54218
implements a frequency shift. If frequency shift was not implemented, during an overcurrent condition the low
side MOSFET may not be turned off long enough to reduce the current in the inductor, causing a current
runaway. With frequency shift, during an overcurrent condition the switching frequency is reduced from 100%,
then 75%, then 50%, then 25% as the voltage decreases from 0.8 to 0 volts on VSENSE pin to allow the low
side MOSFET to be off long enough to decrease the current in the inductor. During start-up, the switching
frequency increases as the voltage on VSENSE increases from 0 to 0.8 volts. See Figure 7 for details.
14
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REVERSE OVERCURRENT PROTECTION
The TPS54218 implements low side current protection by detecting the voltage across the low side MOSFET.
When the converter sinks current through its low side FET, the control circuit turns off the low side MOSFET if
the reverse current is more than 1.3 A. By implementing this additional protection scheme, the converter is able
to protect itself from excessive current during power cycling and start-up into pre-biased outputs.
SYNCHRONIZE USING THE RT/CLK PIN
The RT/CLK pin is used to synchronize the converter to an external system clock. See Figure 28. To implement
the synchronization feature in a system, connect a square wave to the RT/CLK pin with an on time of at least
75ns. If the pin is pulled above the PLL upper threshold, a mode change occurs and the pin becomes a
synchronization input. The internal amplifier is disabled and the pin is a high impedance clock input to the
internal PLL. If clocking edges stop, the internal amplifier is re-enabled and the mode returns to the frequency set
by the resistor. The square wave amplitude at this pin must transition lower than 0.6 V and higher than 1.6 V
typically. The synchronization frequency range is 300 kHz to 2000 kHz. The rising edge of the PH is
synchronized to the falling edge of RT/CLK pin.
TPS54218
SYNC Clock = 2 V / div
PLL
PH = 2 V / div
RT/CLK
Clock
Source
RT
Time = 500 nsec / div
Figure 28. Synchronizing to a System Clock
Figure 29. Plot of Synchronizing to System Clock
POWER GOOD (PWRGD PIN)
The PWRGD pin output is an open drain MOSFET. The output is pulled low when the VSENSE voltage enters
the fault condition by falling below 91% or rising above 107% of the nominal internal reference voltage. There is
a 2% hysteresis on the threshold voltage, so when the VSENSE voltage rises to the good condition above 93%
or falls below 105% of the internal voltage reference the PWRGD output MOSFET is turned off. It is
recommended to use a pull-up resistor between the values of 1kΩ and 100kΩ to a voltage source that is 6 V or
less. The PWRGD is in a valid state once the VIN input voltage is greater than 1.2 V.
OVERVOLTAGE TRANSIENT PROTECTION
The TPS54218 incorporates an overvoltage transient protection (OVTP) circuit to minimize voltage overshoot
when recovering from output fault conditions or strong unload transients. The OVTP feature minimizes the output
overshoot by implementing a circuit to compare the VSENSE pin voltage to the OVTP threshold which is 109%
of the internal voltage reference. If the VSENSE pin voltage is greater than the OVTP threshold, the high side
MOSFET is disabled preventing current from flowing to the output and minimizing output overshoot. When the
VSENSE voltage drops lower than the OVTP threshold the high side MOSFET is allowed to turn on the next
clock cycle.
THERMAL SHUTDOWN
The device implements an internal thermal shutdown to protect itself if the junction temperature exceeds 175°C.
The thermal shutdown forces the device to stop switching when the junction temperature exceeds the thermal
trip threshold. Once the die temperature decreases below 160°C, the device reinitiates the power up sequence
by discharging the SS pin to 0 volts. The thermal shutdown hysteresis is 15°C.
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SMALL SIGNAL MODEL FOR LOOP RESPONSE
Figure 30 shows an equivalent model for the TPS54218 control loop which can be modeled in a circuit simulation
program to check frequency response and dynamic load response. The error amplifier is a transconductance
amplifier with a gm of 225 μA/V. The error amplifier can be modeled using an ideal voltage controlled current
source. The resistor Ro and capacitor Co model the open loop gain and frequency response of the amplifier. The
1-mV AC voltage source between the nodes a and b effectively breaks the control loop for the frequency
response measurements. Plotting a/c shows the small signal response of the frequency compensation. Plotting
a/b shows the small signal response of the overall loop. The dynamic loop response can be checked by
replacing the RL with a current source with the appropriate load step amplitude and step rate in a time domain
analysis.
PH
VO
Power Stage
13.0 A/V
a
b
R1
c
RL
COMP
R3
C2
RESR
C1
CO RO
0.8 V
VSENSE
gm
225 uA/V
COUT
R2
Figure 30. Small Signal Model for Loop Response
SIMPLE SMALL SIGNAL MODEL FOR PEAK CURRENT MODE CONTROL
Figure 30 is a simple small signal model that can be used to understand how to design the frequency
compensation. The TPS54218 power stage can be approximated to a voltage controlled current source (duty
cycle modulator) supplying current to the output capacitor and load resistor. The control to output transfer
function is shown in Equation 7 and consists of a dc gain, one dominant pole and one ESR zero. The quotient of
the change in switch current and the change in COMP pin voltage (node c in Figure 30) is the power stage
transconductance. The gm for the TPS54218 is 13 A/V. The low frequency gain of the power stage frequency
response is the product of the transconductance and the load resistance as shown in Equation 8. As the load
current increases and decreases, the low frequency gain decreases and increases, respectively. This variation
with load may seem problematic at first glance, but the dominant pole moves with load current [see Equation 9].
The combined effect is highlighted by the dashed line in the right half of Figure 31. As the load current
decreases, the gain increases and the pole frequency lowers, keeping the 0-dB crossover frequency the same
for the varying load conditions which makes it easier to design the frequency compensation.
16
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VO
Adc
VC
RESR
fp
RL
gmps
COUT
fz
Figure 31. Simple Small Signal Model and Frequency Response for Peak Current Mode Control
æ
ç 1+
vo
è 2p
= Adc ´
vc
æ
ç 1+
è 2p
ö
s
÷
× ¦z ø
ö
s
÷
× ¦p ø
(7)
Adc = gmps ´ RL
¦p =
¦z =
C OUT
(8)
1
´ RL ´ 2p
(9)
1
COUT ´ RESR ´ 2p
(10)
SMALL SIGNAL MODEL FOR FREQUENCY COMPENSATION
The TPS54218 uses a transconductance amplifier for the error amplifier and readily supports two of the
commonly used frequency compensation circuits. The compensation circuits are shown in Figure 32. The Type 2
circuits are most likely implemented in high bandwidth power supply designs using low ESR output capacitors. In
Type 2A, one additional high frequency pole is added to attenuate high frequency noise.
VO
R1
VSENSE
COMP
gmea
R2
Vref
RO
CO
5pF
Type 2A
R3
C2
Type 2B
R3
C1
C1
Figure 32. Types of Frequency Compensation
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The design guidelines for TPS54218 loop compensation are as follows:
1. The modulator pole, fpmod, and the esr zero, fz1 must be calculated using Equation 11 and Equation 12.
Derating the output capacitor (COUT) may be needed if the output voltage is a high percentage of the
capacitor rating. Use the capacitor manufacturer information to derate the capacitor value. Use Equation 13
and Equation 14 to estimate a starting point for the crossover frequency, fc. Equation 13 is the geometric
mean of the modulator pole and the esr zero and Equation 14 is the mean of modulator pole and the
switching frequency. Use the lower value of Equation 13 or Equation 14 as the maximum crossover
frequency.
IOUT max
¦p mod =
2p ´ VOUT ´ COUT
(11)
¦ z mod =
1
2p ´ RESR ´ COUT
¦C =
¦p mod ´ ¦ z mod
¦C =
¦p mod ´
(12)
(13)
¦ sw
2
(14)
2. R3 can be determined by
2p × ¦ c ´ VOUT ´ COUT
R3 =
gmea ´ Vref ´ gmps
(15)
Where is the gmea amplifier gain (225 μA/V), gmps is the power stage gain (13 A/V).
¦p =
1
C OUT ´ R L ´ 2 p . C1 can be determined by
3. Place a compensation zero at the dominant pole
R ´ COUT
C1 = L
R3
4. C2 is optional. It can be used to cancel the zero from te COUT ESR.
´ COUT
R
C2 = ESR
R3
18
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APPLICATION INFORMATION
DESIGN GUIDE – STEP-BY-STEP DESIGN PROCEDURE
This example details the design of a high frequency switching regulator design using ceramic output capacitors.
This design is available as the HPA511 evaluation module (EVM). A few parameters must be known in order to
start the design process. These parameters are typically determined on the system level. For this example, we
start with the following known parameters:
Output Voltage
1.8 V
Transient Response 1 to 2A load step
ΔVOUT = 3%
Maximum Output Current
2A
Input Voltage
3.3 V nom. 3 V to 6 V
Output Voltage Ripple
< 30 mVPP
Start Input Voltage (rising VIN)
3.1 V
Stop Input Voltage (falling VIN)
2.8 V
Switching Frequency (Fsw)
1000 kHz
SELECTING THE SWITCHING FREQUENCY
The first step is to decide on a switching frequency for the regulator. Typically, you want to choose the highest
switching frequency possible since this produces the smallest solution size. The high switching frequency allows
for lower valued inductors and smaller output capacitors compared to a power supply that switches at a lower
frequency. However, the highest switching frequency causes extra switching losses, which hurt the converter’s
performance. The converter is capable of running from 200 kHz to 2 MHz. Unless a small solution size is an
ultimate goal, a moderate switching frequency of 1MHz is selected to achieve both a small solution size and a
high efficiency operation. Using Equation 5, R4 is calculated to be 180 kΩ. A standard 1% 182 kΩ value was
chosen in the design.
VIN = 3 - 6 V
Figure 33. High Frequency, 1.8 V Output Power Supply Design with Adjusted UVLO
OUTPUT INDUCTOR SELECTION
The inductor selected works for the entire TPS54218 input voltage range. To calculate the value of the output
inductor, use Equation 18. KIND is a coefficient that represents the amount of inductor ripple current relative to the
maximum output current. The inductor ripple current is filtered by the output capacitor. Therefore, choosing high
inductor ripple currents impacts the selection of the output capacitor since the output capacitor must have a
ripple current rating equal to or greater than the inductor ripple current. In general, the inductor ripple value is at
the discretion of the designer; however, KIND is normally from 0.1 to 0.3 for the majority of applications.
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For this design example, use KIND = 0.3 and the inductor value is calculated to be 2.10 μH. For this design, a
nearest standard value was chosen: 2.2 μH. For the output filter inductor, it is important that the RMS current
and saturation current ratings not be exceeded. The RMS and peak inductor current can be found from
Equation 20 and Equation 21.
For this design, the RMS inductor current is 2 A and the peak inductor current is 2.3 A. The chosen inductor is a
Coilcraft XPL7030-222ML_. It has a RMS current rating of 9.7 A and a saturation current rating of 16 A. The
current ratings for this exceed the requirement, but the inductor was chosen for small physical size and low
series resistance for high efficiency.
The current flowing through the inductor is the inductor ripple current plus the output current. During power up,
faults or transient load conditions, the inductor current can increase above the calculated peak inductor current
level calculated above. In transient conditions, the inductor current can increase up to the switch current limit of
the device. For this reason, the most conservative approach is to specify an inductor with a saturation current
rating equal to or greater than the switch current limit rather than the peak inductor current.
V max - VOUT
VOUT
L1 = IN
´
IOUT ´ K ind
VINmax ´ ¦ sw
(18)
Iripple =
VINmax - VOUT
VOUT
´
L1
VINmax ´ ¦ sw
ILrms = IOUT 2 +
ILpeak = IOUT +
æV
´ (VINmax - VOUT ) ö
1
´ ç OUT
÷
12
è VINmax ´ L1 ´ ¦ sw ø
(19)
2
(20)
Iripple
2
(21)
OUTPUT CAPACITOR
There are three primary considerations for selecting the value of the output capacitor. The output capacitor
determines the modulator pole, the output voltage ripple, and how the regulator responds to a large change in
load current. The output capacitance needs to be selected based on the more stringent of these three criteria.
The desired response to a large change in the load current is the first criteria. The output capacitor needs to
supply the load with current when the regulator can not. This situation would occur if there are desired hold-up
times for the regulator where the output capacitor must hold the output voltage above a certain level for a
specified amount of time after the input power is removed. The regulator is temporarily not able to supply
sufficient output current if there is a large, fast increase in the current needs of the load such as transitioning
from no load to a full load. The regulator usually needs two or more clock cycles for the control loop to see the
change in load current and output voltage and adjust the duty cycle to react to the change. The output capacitor
must be sized to supply the extra current to the load until the control loop responds to the load change. The
output capacitance must be large enough to supply the difference in current for 2 clock cycles while only allowing
a tolerable amount of droop in the output voltage. Equation 22 shows the necessary minimum output
capacitance.
For this example, the transient load response is specified as a 3% change in VOUT for a load step from 1A (50%
load) to 2A (100%). For this example, ΔIOUT = 2-1 = 1 A and ΔVOUT = 0.03 × 1.8 = 0.054 V. Using these numbers
gives a minimum capacitance of 37 μF. This value does not take the ESR of the output capacitor into account in
the output voltage change. For ceramic capacitors, the ESR is usually small enough to ignore in this calculation.
Equation 23 calculates the minimum output capacitance needed to meet the output voltage ripple specification.
Where fsw is the switching frequency, Vripple is the maximum allowable output voltage ripple, and Iripple is the
inductor ripple current. In this case, the maximum output voltage ripple is 30 mV. Under this requirement,
Equation 23 yields 2.18 nF.
20
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COUT >
COUT
2 ´ DIOUT
¦ sw ´ DVOUT
(22)
1
1
´
>
V
8 ´ ¦ sw
Oripple
Iripple
(23)
Where ΔIOUT is the change in output current, Fsw is the regulators switching frequency and ΔVOUT is the
allowable change in the output voltage.
Equation 24 calculates the maximum ESR an output capacitor can have to meet the output voltage ripple
specification. Equation 24 indicates the ESR should be less than 57 mΩ. In this case, the ESR of the ceramic
capacitor is much less than 57 mΩ.
Additional capacitance de-ratings for aging, temperature and DC bias should be factored in which increases this
minimum value. For this example, two 22 μF 10 V X5R ceramic capacitors with 3 mΩ of ESR are used.
Capacitors generally have limits to the amount of ripple current they can handle without failing or producing
excess heat. An output capacitor that can support the inductor ripple current must be specified. Some capacitor
data sheets specify the RMS (Root Mean Square) value of the maximum ripple current. Equation 25 can be used
to calculate the RMS ripple current the output capacitor needs to support. For this application, Equation 25 yields
151mA.
VOripple
RESR <
Iripple
(24)
Icorms =
VOUT ´ (VINmax - VOUT )
12 ´ VINmax ´ L1 ´ ¦ sw
(25)
INPUT CAPACITOR
The TPS54218 requires a high quality ceramic, type X5R or X7R, input decoupling capacitor of at least 4.7 μF of
effective capacitance and in some applications a bulk capacitance. The effective capacitance includes any DC
bias effects. The voltage rating of the input capacitor must be greater than the maximum input voltage. The
capacitor must also have a ripple current rating greater than the maximum input current ripple of the TPS54218.
The input ripple current can be calculated using Equation 26.
The value of a ceramic capacitor varies significantly over temperature and the amount of DC bias applied to the
capacitor. The capacitance variations due to temperature can be minimized by selecting a dielectric material that
is stable over temperature. X5R and X7R ceramic dielectrics are usually selected for power regulator capacitors
because they have a high capacitance to volume ratio and are fairly stable over temperature. The output
capacitor must also be selected with the dc bias taken into account. The capacitance value of a capacitor
decreases as the dc bias across a capacitor increases.
For this example design, a ceramic capacitor with at least a 10 V voltage rating is required to support the
maximum input voltage. For this example, one 10 μF and one 0.1 μF 10 V capacitors in parallel have been
selected. The input capacitance value determines the input ripple voltage of the regulator. The input voltage
ripple can be calculated using Equation 27. Using the design example values, IOUTmax = 2 A, CIN = 10 μF,
Fsw = 1 MHz, yields an input voltage ripple of 34 mV and a rms input ripple current of 0.98 A.
Icirms = IOUT ´
DVIN
VOUT
´
VINmin
(VINmin
- VOUT )
VINmin
(26)
I
max ´ 0.25
= OUT
CIN ´ ¦ sw
(27)
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SLOW START CAPACITOR
The slow start capacitor determines the minimum amount of time it takes for the output voltage to reach its
nominal programmed value during power up. This is useful if a load requires a controlled voltage slew rate. This
is also used if the output capacitance is very large and would require large amounts of current to quickly charge
the capacitor to the output voltage level. The large currents necessary to charge the capacitor may make the
TPS54218 reach the current limit or excessive current draw from the input power supply may cause the input
voltage rail to sag. Limiting the output voltage slew rate solves both of these problems.
The slow start capacitor value can be calculated using Equation 28. For the example circuit, the slow start time is
not too critical since the output capacitor value is 44 μF which does not require much current to charge to 1.8 V.
The example circuit has the slow start time set to an arbitrary value of 4ms which requires a 10 nF capacitor. In
TPS54218, Iss is 2 μA and Vref is 0.8 V.
T (mS) ´ ISS (mA)
CSS (nF) = SS
Vref (V)
(28)
BOOTSTRAP CAPACITOR SELECTION
A 0.1 μF ceramic capacitor must be connected between the BOOT to PH pin for proper operation. It is
recommended to use a ceramic capacitor with X5R or better grade dielectric. The capacitor should have 10 V or
higher voltage rating.
UNDERVOLTAGE LOCK OUT SET POINT
The Undervoltage Lock Out (UVLO) can be adjusted using an external voltage divider on the EN pin of the
TPS54218. The UVLO has two thresholds, one for power up when the input voltage is rising and one for power
down or brown outs when the input voltage is falling. For the example design, the supply should turn on and start
switching once the input voltage increases above 3.1 V (VSTART). After the regulator starts switching, it should
continue to do so until the input voltage falls below 2.8 V (VSTOP).
The programmable UVLO and enable voltages are set using a resistor divider between Vin and ground to the EN
pin. Equation 29 and Equation 30 can be used to calculate the resistance values necessary. From Equation 29
and Equation 30, a 48.7 kΩ between Vin and EN and a 32.4 kΩ between EN and ground are required to produce
the 3.1 and 2.8 volt start and stop voltages.
0.944 × VSTART - VSTOP
R1 =
2.59 ´ 10-6
(29)
R2 =
1.18 × R1
VSTOP - 1.18 + R1 × 3.2 ´ 10 - 6
(30)
OUTPUT VOLTAGE AND FEEDBACK RESISTORS SELECTION
For the example design, 100 kΩ was selected for R6. Using Equation 31, R7 is calculated as 80 kΩ. The nearest
standard 1% resistor is 80.5 kΩ.
Vref
R7 =
R6
VOUT - Vref
(31)
Due to the internal design of the TPS54218, there is a minimum output voltage limit for any given input voltage.
The output voltage can never be lower than the internal voltage reference of 0.8 V. Above 0.8 V, the output
voltage may be limited by the minimum controllable on time. The minimum output voltage in this case is given by
Equation 32
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(
(
)))- (IOUTmin ´ (RL + RDS ))
(
VOUT min = Ontimemin ´ FSmax ´ VINmax - IOUT min ´ (2 ´ RDS )
(32)
Where:
VOUTmin = minimum achievable output voltage
Ontimemin = minimum controllable on-time (60 ns typical. 110 nsec no load)
FSmax = maximum switching frequency including tolerance
VINmax = maximum input voltage
IOUTmin = minimum load current
RDS = minimum high side MOSFET on resistance (30 - 44 mΩ)
RL = series resistance of output inductor
There is also a maximum achievable output voltage which is limited by the minimum off time. The maximum
output voltage is given by Equation 33
)(
(
(
))
VOUT max = 1 - (Offtimemax ´ FSmax ) ´ VINmin - IOUT max ´ (2 ´ RDS ) - IOUT max ´ (RL + RDS )
(33)
Where:
VOUTmax = maximum achievable output voltage
Offtimeman = maximum off time (60 nsec typical)
FSmax = maximum switching frequency including tolerance
VINmin = minimum input voltage
IOUTmax = maximum load current
RDS = maximum high side MOSFET on resistance (60 - 70 mΩ)
RL = series resistance of output inductor
COMPENSATION
There are several industry techniques used to compensate DC/DC regulators. The method presented here is
easy to calculate and yields high phase margins. For most conditions, the regulator has a phase margin between
60 and 90 degrees. The method presented here ignores the effects of the slope compensation that is internal to
the TPS54218. Since the slope compensation is ignored, the actual cross over frequency is usually lower than
the cross over frequency used in the calculations. Use SwitcherPro software for a more accurate design.
To get started, the modulator pole, fpmod, and the esr zero, fz1 must be calculated using Equation 34 and
Equation 12. For COUT, derating the capacitor is not needed as the 1.8 V output is a small percentage of the 10 V
capacitor rating. If the output is a high percentage of the capacitor rating, use the capacitor manufacturer
information to derate the capacitor value. Use Equation 36 and Equation 37 to estimate a starting point for the
crossover frequency, fc. For the example design, fpmod is 4.02 kHz and fpmod is 1206 kHz. Equation 36 is the
geometric mean of the modulator pole and the esr zero and Equation 37 is the mean of modulator pole and the
switching frequency. Equation 36 yields 69.6 kHz and Equation 37 gives 44.8 kHz. Use the lower value of
Equation 36 or Equation 37 as the maximum crossover frequency. For this example, fc is 45 kHz. Next, the
compensation components are calculated. A resistor in series with a capacitor is used to create a compensating
zero. A capacitor in parallel to these two components forms the compensating pole (if needed).
IOUT max
¦p mod =
2p ´ VOUT ´ COUT
(34)
¦ z mod =
1
2p ´ RESR ´ COUT
¦C =
¦p mod ´ ¦ z mod
¦C =
¦p mod ´
(35)
(36)
¦ sw
2
(37)
The compensation design takes the following steps:
1. Set up the anticipated cross-over frequency. Use Equation 38 to calculate the compensation network’s
resistor value. In this example, the anticipated cross-over frequency (fc) is 45 kHz. The power stage gain
(gmps) is 13 A/V and the error amplifier gain (gmea) is 225 A/V.
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R3 =
2p × ¦ c ´ VOUT ´ COUT
gm ´ Vref ´ VIgm
(38)
2. Place compensation zero at the pole formed by the load resistor and the output capacitor. The compensation
network’s capacitor can be calculated from Equation 39.
´ COUT
R
C3 = OUT
R3
(39)
3. An additional pole can be added to attenuate high frequency noise. In this application, it is not necessary to
add it.
From the procedures above, the compensation network includes a 9.53 kΩ resistor and a 3900 pF capacitor.
APPLICATION CURVES
EFFICIENCY
vs
LOAD CURRENT
EFFICIENCY
vs
LOAD CURRENT
100
100
95
90
80
90
VI = 3.3 V
85
70
VI = 5 V
80
Efficiency - %
Efficiency - %
VI = 3.3 V
75
70
60
40
65
30
60
20
55
10
50
0
0.25
0.5
0.75
1
1.25
IO - Output Current - A
1.5
1.75
2
Vin = 5 V
50
0
0.001
0.01
0.1
IO - Output Current - A
1
Figure 34.
Figure 35.
TRANSIENT RESPONSE, 1 A STEP
TRANSIENT RESPONSE, 2 A STEP
10
VOUT = 50 mV/div (ac coupled)
VOUT = 50 mV/div (ac coupled)
IOUT = 1 A/div,
0.5 to 1.5 A step
IOUT = 1 A/div,
0 to 2 A step
Time = 2 ms/div
Time = 2 ms/div
Figure 36.
24
Figure 37.
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POWER UP VOUT, VIN
POWER DOWN VOUT, VIN
VI = 2 V/div
VI = 2 V/div
EN = 1 V/div
EN = 1 V/div
SS = 1 V/div
SS = 1 V/div
VO = 1 V/div
VO = 1 V/div
Time = 5 ms/div
Time = 500 ms/div
Figure 38.
Figure 39.
POWER UP VOUT, EN
POWER DOWN VOUT, EN
VI = 2 V/div
VI = 2 V/div
EN = 1 V/div
EN = 1 V/div
SS = 1 V/div
SS = 1 V/div
VO = 1 V/div
VO = 1 V/div
Time = 500 ms/div
Time = 5 ms/div
Figure 40.
Figure 41.
OUTPUT RIPPLE, 0 A
OUTPUT RIPPLE, 2 A
VOUT = 10 mV/div (ac coupled)
VOUT = 10 mV/div (ac coupled)
PH = 2 V/div
PH = 2 V/div
Time = 500 ns/div
Time = 500 ns/div
Figure 42.
Figure 43.
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INPUT RIPPLE, 0 A
INPUT RIPPLE, 2 A
VI = 50 mV/div (ac coupled)
VI = 10 mV/div (ac coupled)
PH = 2 V/div
PH = 2 V/div
Time = 500 ns/div
Time = 500 ns/div
Figure 44.
Figure 45.
CLOSED LOOP RESPONSE, VIN (3.3 V), 2 A
LOAD REGULATION
vs
LOAD CURRENT
60
0.25
180
0.2
Fhase
Output Voltage Change - %
0.15
Gain
Phase
Gain
0.1
0.05
VI = 3.3 V
0
-0.05
-0.1
-0.15
-0.2
-60
10
-180
1M
f - Frequency - Hz
-0.25
0
LOAD REGULATION
vs
LOAD CURRENT
REGULATION
vs
INPUT VOLTAGE
1.81
0.2
1.808
1.5
1.75
2
1.806
VI = 5 V
0.1
VO - Output Voltage - V
Output Voltage Change - %
0.75
1
1.25
IO - Output Current - A
Figure 47.
0.15
0.05
0
-0.05
-0.1
1.804
1.802
1.798
1.796
-0.15
1.794
-0.2
1.792
0.25
0.5
0.75
1
1.25
IO - Output Current - A
1.5
1.75
2
IO = 1
1.8
1.79
3
Figure 48.
26
0.5
Figure 46.
0.25
-0.25
0
0.25
3.5
4
4.5
5
VI - Input Voltage - V
5.5
6
Figure 49.
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POWER DISSIPATION ESTIMATE
The following formulas show how to estimate the IC power dissipation under continuous conduction mode (CCM)
operation. The power dissipation of the IC (Ptot) includes conduction loss (Pcon), dead time loss (Pd), switching
loss (Psw), gate drive loss (Pgd) and supply current loss (Pq).
Pcon = IOUT 2 × RDS(on)
Pd = ƒsw × IOUT × 0.7 × 60 × 10-9
Psw = 2 × VIN 2 × ƒsw × IOUT × 0.25 × 10-9
Pgd = 2 × VIN × 3 × 10-9 × ƒsw
Pq = 350 × 10-6 × VIN
Where:
IOUT is the output current (A).
RDS(on) is the on-resistance of the high-side MOSFET (Ω).
VOUT is the output voltage (V).
VIN is the input voltage (V).
ƒsw is the switching frequency (Hz).
So
Ptot = Pcon + Pd + Psw + Pgd + Pq
For given TA,
TJ = TA + Rth × Ptot
For given TJ max = 150°C
TAmax = TJ max – Rth × Ptot
Where:
Ptot is the total device power dissipation (W).
TA is the ambient temperature (°C).
TJ is the junction temperature (°C).
Rth is the thermal resistance of the package (°C/W).
TJmax is maximum junction temperature (°C).
TAmax is maximum ambient temperature (°C).
There are additional power losses in the regulator circuit due to the inductor ac and dc losses and trace
resistance that impact the overall efficiency of the regulator. As an example, the maximum ambient temperature
versus power dissipation for the EVM is shown in Figure 51.
150
140
TA = room temperature, no air flow
TJ - Junction Temperature - °C
130
120
110
100
90
80
70
60
50
40
30
20
0
0.5
1
1.5
2
2.5
PD - Power Dissipated - W
3
3.5
Figure 50. Junction Temperature vs IC Power Dissipation
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TAmax - Maximum Ambient Temperature - °C
150
140
130
120
TJmax = 150°C, no air flow
110
100
90
80
70
60
50
40
30
20
0
0.5
1
1.5
2
2.5
PD - Power Dissipated - W
3
3.5
Figure 51. Maximum Ambient Temperature vs IC power Dissipation
LAYOUT
Layout is a critical portion of good power supply design. There are several signal paths that conduct fast
changing currents or voltages that can interact with stray inductance or parasitic capacitance to generate noise
or degrade the power supplies performance. Care should be taken to minimize the loop area formed by the
bypass capacitor connections and the VIN pins. See Figure 52 for a PCB layout example. The GND pins and
AGND pin should be tied directly to the power pad under the IC. The power pad should be connected to any
internal PCB ground planes using multiple vias directly under the IC. Additional vias can be used to connect the
top side ground area to the internal planes near the input and output capacitors. For operation at full rated load,
the top side ground area along with any additional internal ground planes must provide adequate heat dissipating
area.
Locate the input bypass capacitor as close to the IC as possible. The PH pin should be routed to the output
inductor. Since the PH connection is the switching node, the output inductor should be located close to the PH
pins, and the area of the PCB conductor minimized to prevent excessive capacitive coupling. The boot capacitor
must also be located close to the device. The sensitive analog ground connections for the feedback voltage
divider, compensation components, slow start capacitor and frequency set resistor should be connected to a
separate analog ground trace as shown. The RT/CLK pin is particularly sensitive to noise so the RT resistor
should be located as close as possible to the IC and routed with minimal lengths of trace. The additional external
components can be placed approximately as shown. It is possible to obtain acceptable performance with
alternate PCB layouts; however, this layout has been shown to produce good results and is meant as a
guideline.
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VIA to
Ground
Plane
UVLO SET
RESISTRORS
VIN
INPUT
BYPASS
CAPACITOR
BOOT
PWRGD
EN
VIN
VIN
BOOT
CAPACITOR
VIN
OUTPUT
INDUCTOR
PH
VIN
PH
EXPOSED
POWERPAD
AREA
GND
PH
GND
VOUT
OUTPUT
FILTER
CAPACITOR
PH
SLOW START
CAPACITOR
RT/CLK
COMP
VSENSE
AGND
SS
FEEDBACK
RESISTORS
ANALOG
GROUND
TRACE
FREQUENCY
SET
RESISTOR
TOPSIDE
GROUND
AREA
COMPENSATION
NETWORK
VIA to Ground Plane
Figure 52. PCB Layout Example
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PACKAGE OPTION ADDENDUM
www.ti.com
23-Sep-2009
PACKAGING INFORMATION
Orderable Device
Status (1)
Package
Type
Package
Drawing
Pins Package Eco Plan (2)
Qty
TPS54218RTER
ACTIVE
QFN
RTE
16
3000 Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
TPS54218RTET
ACTIVE
QFN
RTE
16
250
CU NIPDAU
Level-2-260C-1 YEAR
Green (RoHS &
no Sb/Br)
Lead/Ball Finish
MSL Peak Temp (3)
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check
http://www.ti.com/productcontent for the latest availability information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements
for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered
at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and
package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS
compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame
retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder
temperature.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is
provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the
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