TPA3117D2 www.ti.com SLOS672 – OCTOBER 2010 15-W Filter-Free Stereo Class-D Audio Power Amplifier with SpeakerGuard™ Check for Samples: TPA3117D2 FEATURES APPLICATIONS • • • • 1 2 • • • • • • • • • • • 15-W/ch into 8Ω Loads at 10% THD+N From a 16V Supply 10-W/ch into 8Ω Loads at 10% THD+N From a 13V Supply 90% Efficient Class-D Operation Eliminates Need for Heat Sinks Wide Supply Voltage Range Allows Operation from 8V to 26V Filter-Free Operation SpeakerGuard™ Speaker Protection Includes Adjustable Power Limiter Excellent THD+N / Pop-Free Performance Four Selectable, Fixed Gain Settings Differential Inputs Selectable Switching Frequency (290kHz or 390kHz) Allows Multiple Devices to be Used in One System Integrated 5V Regulator With up to 30 mA of Output Current for Powering External Data Converters 5mm x 5mm QFN Packaging Televisions Consumer Audio Equipment All-in-One Computers DESCRIPTION The TPA3117D2 is a 15W (per channel) efficient, Class-D audio power amplifier for driving bridged-tied stereo speakers. Advanced EMI Suppression Technology enables the use of inexpensive ferrite bead filters at the outputs while meeting EMC requirements. SpeakerGuard™ speaker protection circuitry includes an adjustable power limiter. The adjustable power limiter allows the user to set a "virtual" voltage rail lower than the chip supply to limit the amount of current through the speaker. The TPA3117D2 can drive stereo speakers as low as 4Ω. The high efficiency of the TPA3117D2, 90%, eliminates the need for an external heat sink when playing music. The outputs are also fully protected against shorts to GND, VCC, and output-to-output. The short-circuit protection and thermal protection includes an auto-recovery feature. SPACER 1mF Audio Source OUTL+ LINP OUTL- LINN OUTR+ RINP OUTR- TPA3117D2 RINN VDD GAIN0 GAIN1 REG_OUT 2.2mF 10W OUTPL OUTNL OUTPR OUTNR FERRITE BEAD FILTER 15W 8W FERRITE BEAD FILTER 15W 8W PLIMIT PBTL FSEL SD PVCC 8 to 26V Figure 1. TPA3117D2 Simplified Application Schematic 1 2 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. SpeakerGuard is a trademark of Texas Instruments. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2010, Texas Instruments Incorporated TPA3117D2 SLOS672 – OCTOBER 2010 www.ti.com These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. ABSOLUTE MAXIMUM RATINGS over operating free-air temperature range (unless otherwise noted) (1) UNIT VCC Supply voltage AVCC, PVCC –0.3V to 30V SD, GAIN0, GAIN1, PBTL, FSEL (2) VI Interface pin voltage –0.3V to VCC + 0.3V PLIMIT –0.3V to REG_OUT + 0.3V RINN, RINP, LINN, LINP –0.3V to 6.3V Continuous total power dissipation See Dissipation Rating Table TA Operating free-air temperature range –40°C to 85°C TJ Operating junction temperature range (3) –40°C to 150°C Tstg Storage temperature range –65°C to 150°C RL Minimum Load Resistance BTL: PVCC > 15V 4.8 BTL: PVCC ≤ 15V 3.2 PBTL ESD (1) (2) (3) (4) (5) Electrostatic discharge 3.2 Human body model (4) Charged-device model (all pins) (5) ±2kV (all pins) ±500V Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings only, and functional operations of the device at these or any other conditions beyond those indicated under recommended operating conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. For input voltage >6V, a series current limiting resistor of at least 100kΩ is recommended. The TPA3117D2 incorporates an exposed thermal pad on the underside of the chip. This acts as a heatsink, and it must be connected to a thermally dissipating plane for proper power dissipation. Failure to do so may result in the device going into thermal protection shutdown. In accordance with JEDEC Standard 22, Test Method A114-B. In accordance with JEDEC Standard 22, Test Method C101-A RECOMMENDED OPERATING CONDITIONS over operating free-air temperature range (unless otherwise noted) MIN MAX VCC Supply voltage PARAMETER PVCC, AVCC TEST CONDITIONS 8 26 VIH High-level input voltage SD, GAIN0, GAIN1, PBTL, FSEL 2 VIL Low-level input voltage SD, GAIN0, GAIN1, PBTL, FSEL 0.8 V IIH High-level input current SD, GAIN0, GAIN1, PBTL, FSEL, VI = 2V, VCC = 18V 50 µA IIL Low-level input current SD, GAIN0, GAIN1, PBTL, FSEL, VI = 0.8V, VCC = 18V 5 µA TA Operating free-air temperature 85 °C –40 UNIT V V THERMAL INFORMATION THERMAL METRIC (1) TPA3117D2 RHB (32 PINS) qJA Junction-to-ambient thermal resistance 33.7 qJC(top) Junction-to-case(top) thermal resistance 36.3 qJB Junction-to-board thermal resistance 9.8 yJT Junction-to-top characterization parameter 0.6 yJB Junction-to-board characterization parameter 9.5 qJC(bottom) Junction-to-case(bottom) thermal resistance 3.2 (1) 2 UNITS °C/W For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953. Submit Documentation Feedback Copyright © 2010, Texas Instruments Incorporated Product Folder Link(s) :TPA3117D2 TPA3117D2 www.ti.com SLOS672 – OCTOBER 2010 DC CHARACTERISTICS TA = 25°C, VCC = 24 V, RL = 8 Ω (unless otherwise noted) PARAMETER TEST CONDITIONS | VOS | Class-D output offset voltage (measured differentially) VI = 0V, Gain = 18.2dB ICC Quiescent supply current ICC(SD) Quiescent supply current in shutdown mode rDS(on) UNIT 15 mV SD = 2V, no load, VCC = 26V 25 50 mA SD = 0.8V, no load, VCC = 24V 2.5 5 mA Gain GAIN1 = 2V ton Turn-on time SD = 2V tOFF Turn-off time SD = 0.8V High Side 240 Low side 240 GAIN0 = 0.8 V GAIN1 = 0.8V G TYP MAX 1.5 VCC = 12V, IO = 500mA, TJ = 25°C Drain-source on-state resistance MIN mΩ 8 9 10 GAIN0 = 2 V 11.1 12.1 13.1 GAIN0 = 0.8 V 14.2 15.2 16.2 GAIN0 = 2 V 17.2 18.2 19.2 dB dB 14 ms 2 ms DC CHARACTERISTICS TA = 25°C, VCC = 12 V, RL = 8 Ω (unless otherwise noted) PARAMETER TEST CONDITIONS | VOS | Class-D output offset voltage (measured differentially) VI = 0V, Gain = 18.2dB ICC Quiescent supply current ICC(SD) Quiescent supply current in shutdown mode rDS(on) UNIT 15 mV SD = 2V, no load, VCC = 12V 15 35 mA SD = 0.8V, no load, VCC = 12V 2.5 5 mA Gain GAIN1 = 2V tON Turn-on time SD = 2V tOFF Turn-off time SD = 0.8V High Side 240 Low side 240 GAIN0 = 0.8 V GAIN1 = 0.8V G TYP MAX 1.5 VCC = 12V, IO = 500mA, TJ = 25°C Drain-source on-state resistance MIN mΩ 8 9 10 GAIN0 = 2 V 11.1 12.1 13.1 GAIN0 = 0.8 V 14.2 15.2 16.2 GAIN0 = 2 V 17.2 18.2 19.2 dB dB 14 ms 2 ms LDO CHARACTERISTICS TA = 25°C, VCC = 12 V (unless otherwise noted) PARAMETER TEST CONDITIONS TYP MAX 8 12 26 UNIT V 30 mA Input voltage IO Continuous output current VO Output voltage 0 < IO < 30mA, 10.8V < VIN < 13.2 5 V Line regulation IL = 5mA, 10.8V < VIN < 13.2V 6 µV Load regulation IL = 0 - 30mA, VIN = 12V, Measurement taken with an external 10Ω series resistor Power supply ripple rejection VCC = 12V, IL = 20mA PSRR VCC MIN VI 10.2 f = 100Hz 92 f = 1kHz 72 mV / mA Submit Documentation Feedback Copyright © 2010, Texas Instruments Incorporated Product Folder Link(s) :TPA3117D2 dB 3 TPA3117D2 SLOS672 – OCTOBER 2010 www.ti.com AC CHARACTERISTICS TA = 25°C, VCC = 24 V, RL = 8 Ω (unless otherwise noted) PARAMETER TEST CONDITIONS MIN KSVR Power Supply ripple rejection 200mVPP ripple at 1kHz, Gain = 18.2dB, Inputs ac-coupled to AGND PO Continuous output power THD+N Total harmonic distortion + noise Vn TYP MAX UNIT –70 dB THD+N = 10%, f = 1kHz, VCC = 16V 15 W VCC = 16V, f = 1kHz, PO = 7.5W (half-power) 0.1 % 55 µV Output integrated noise 20Hz to 22kHz, A-weighted filter, Gain = 9dB Crosstalk VO = 1Vrms, Gain = 9dB, f = 1kHz SNR Signal-to-noise ratio fOSC Oscillator frequency –85 dBV –100 dB Maximum output at THD+N < 1%, f = 1kHz, Gain = 9dB, A-weighted 102 dB FSEL = 0.8V 290 FSEL = 2V 390 Thermal trip point Thermal hysteresis kHz 150 °C 15 °C AC CHARACTERISTICS TA = 25°C, VCC = 12 V, RL = 8 Ω (unless otherwise noted) PARAMETER TEST CONDITIONS MIN KSVR Supply ripple rejection 200mVPP ripple from 20Hz – 1kHz, Gain = 18.2dB, Inputs ac-coupled to AGND PO Continuous output power THD+N = 10%, f = 1kHz; VCC = 13V THD+N Total harmonic distortion + noise RL = 8Ω, f = 1kHz, PO = 5W (half-power) Vn Output integrated noise 20Hz to 22kHz, A-weighted filter, Gain = 9dB Crosstalk Po = 1W, Gain = 9dB, f = 1kHz SNR Signal-to-noise ratio fOSC Oscillator frequency TYP MAX UNIT –70 dB 10 W 0.06 % 48 µV –86 dBV –100 dB Maximum output at THD+N < 1%, f = 1kHz, Gain = 9dB, A-weighted 102 dB FSEL = 0.8V 290 FSEL = 2V 390 Thermal trip point Thermal hysteresis kHz 150 °C 15 °C 28 BSPL 29 PVCCL NC 30 PVCCL SD 31 NC FSEL 32 27 26 25 LINN 1 24 OUTPL GAIN0 2 23 PGND 3 22 OUTNL AVCC 4 21 BSNL AGND 5 20 BSNR 6 19 OUTNR PLIMIT 7 18 PGND RINN 8 17 OUTPR 9 10 11 12 13 14 15 16 NC PBTL NC NC PVCCR PVCCR BSPR REG_OUT Thermal Pad RINP GAIN1 4 LINP RHB (QFN) PACKAGE (TOP VIEW) Submit Documentation Feedback Copyright © 2010, Texas Instruments Incorporated Product Folder Link(s) :TPA3117D2 TPA3117D2 www.ti.com SLOS672 – OCTOBER 2010 PIN FUNCTIONS PIN NAME NUMBER I/O/P DESCRIPTION LINN 1 I Negative audio input for left channel. Biased at 3V. GAIN0 2 I Gain select least significant bit. TTL logic levels with compliance to AVCC. GAIN1 3 I Gain select most significant bit. TTL logic levels with compliance to AVCC. AVCC 4 P Analog supply AGND 5 REG_OUT 6 O 5V regulated output. Connect 2.2µF to AGND after the series 10 Ω resistor. PLIMIT 7 I Power limit level adjust. Connect a resistor divider from REG_OUT to AGND to set power limit. Connect directly to REG_OUT for no power limit. RINN 8 I Negative audio input for right channel. Biased at 3V. 9 I Positive audio input for right channel. Biased at 3V. RINP NC PBTL Analog signal ground. Connect to the thermal pad. 10, 12, 13, 28, 29 Not connected 11 I Parallel BTL mode switch (low = BTL mode, high = PBTL mode) 14, 15 P Power supply for right channel H-bridge. Right channel and left channel power supply inputs are connected internally. PVCCR and PVCCL must be connected together on the PCB. BSPR 16 I Bootstrap I/O for right channel, positive high-side FET. OUTPR 17 O Class-D H-bridge positive output for right channel. PGND 18 OUTNR 19 O Class-D H-bridge negative output for right channel. BSNR 20 I Bootstrap I/O for right channel, negative high-side FET. BSNL 21 I Bootstrap I/O for left channel, negative high-side FET. OUTNL 22 O Class-D H-bridge negative output for left channel. PGND 23 OUTPL 24 O Class-D H-bridge positive output for left channel. BSPL 25 I Bootstrap I/O for left channel, positive high-side FET. 26, 27 P Power supply for left channel H-bridge. Right channel and left channel power supply inputs are connected internally. PVCCR and PVCCL must be connected together on the PCB. SD 30 I Shutdown logic input for audio amp (LOW = outputs Hi-Z, HIGH = outputs enabled). TTL logic levels with compliance to AVCC. FSEL 31 I Frequency select input pin (low = 300kHz, high = 400kHz) LINP 32 I Positive audio input for left channel. Biased at 3V. PVCCR PVCCL Power ground for the H-bridges. Power ground for the H-bridges. Submit Documentation Feedback Copyright © 2010, Texas Instruments Incorporated Product Folder Link(s) :TPA3117D2 5 TPA3117D2 SLOS672 – OCTOBER 2010 www.ti.com FUNCTIONAL BLOCK DIAGRAM REG_OUT PVCCL BSPL PVCCL PBTL Select OUTPL FB Gate Drive OUTPL OUTPL FB LINN Gain Control PGND PWM Logic PLIMIT REG_OUT LINP PVCCL BSNL PVCCL OUTNL FB OUTNL FB Gate Drive OUTNL SD FSEL GAIN0 TTL Buffer SC Detect PGND Gain Control GAIN1 FSEL Ramp Generator Biases and References Startup Protection Logic PLIMIT Reference PLIMIT Thermal Detect UVLO/OVLO REG_OUT AVDD AVCC PVCCR BSNR PVCCR LDO Regulator REG_OUT Gate Drive REG_OUT OUTNR OUTNR_FB OUTNR FB RINN Gain Control PLIMIT PWM Logic RINP PGND REG_OUT PVCCR BSPR PVCCR OUTPR_FB Gate Drive PBTL TTL Buffer PBTL Select OUTPR PBTL Select OUTPR FB AGND PGND 6 Submit Documentation Feedback Copyright © 2010, Texas Instruments Incorporated Product Folder Link(s) :TPA3117D2 TPA3117D2 www.ti.com SLOS672 – OCTOBER 2010 TYPICAL CHARACTERISTICS (All Measurements taken at 1 kHz, unless otherwise noted. Measurements were made using the TPA3117D2 EVM which is available at ti.com.) TOTAL HARMONIC DISTORTION vs FREQUENCY (BTL) TOTAL HARMONIC DISTORTION vs FREQUENCY (BTL) 10 Gain = 18.2 dB VCC = 12 V ZL = 8 Ω + 66 µH THD − Total Harmonic Distortion − % THD − Total Harmonic Distortion − % 10 1 0.1 PO = 5 W PO = 0.5 W 0.01 Gain = 18.2 dB VCC = 18 V ZL = 8 Ω + 66 µH 1 0.1 PO = 10 W PO = 1 W 0.01 PO = 5 W PO = 2.5 W 0.001 20 100 1k 10k 0.001 20 20k 100 f − Frequency − Hz 1k 10k G001 G002 Figure 2. Figure 3. TOTAL HARMONIC DISTORTION vs FREQUENCY (BTL) TOTAL HARMONIC DISTORTION vs FREQUENCY (BTL) 10 Gain = 18.2 dB VCC = 24 V ZL = 8 Ω + 66 µH THD − Total Harmonic Distortion − % THD − Total Harmonic Distortion − % 10 1 0.1 20k f − Frequency − Hz PO = 10 W PO = 1 W 0.01 Gain = 18.2 dB VCC = 12 V ZL = 6 Ω + 47 µH 1 0.1 PO = 5 W PO = 0.5 W 0.01 PO = 2.5 W PO = 5 W 0.001 20 100 1k 10k 20k 0.001 20 f − Frequency − Hz 100 1k 10k 20k f − Frequency − Hz G003 Figure 4. G004 Figure 5. Submit Documentation Feedback Copyright © 2010, Texas Instruments Incorporated Product Folder Link(s) :TPA3117D2 7 TPA3117D2 SLOS672 – OCTOBER 2010 www.ti.com TYPICAL CHARACTERISTICS (continued) (All Measurements taken at 1 kHz, unless otherwise noted. Measurements were made using the TPA3117D2 EVM which is available at ti.com.) TOTAL HARMONIC DISTORTION vs FREQUENCY (BTL) TOTAL HARMONIC DISTORTION vs FREQUENCY (BTL) 10 Gain = 18.2 dB VCC = 18 V ZL = 6 Ω + 47 µH THD − Total Harmonic Distortion − % THD − Total Harmonic Distortion − % 10 1 PO = 10 W 0.1 0.01 PO = 1 W Gain = 18.2 dB VCC = 12 V ZL = 4 Ω + 33 µH 1 0.1 PO = 10 W PO = 1 W 0.01 PO = 5 W PO = 5 W 0.001 20 100 1k 10k 0.001 20 20k 100 1k f − Frequency − Hz 10k G005 G006 Figure 6. Figure 7. TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT POWER (BTL) TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT POWER (BTL) 10 Gain = 18.2 dB VCC = 12 V ZL = 8 Ω + 66 µH THD+N − Total Harmonic Distortion + Noise − % THD+N − Total Harmonic Distortion + Noise − % 10 1 f = 20 Hz 0.1 f = 1 kHz 0.01 f = 10 kHz 0.001 0.01 0.1 1 PO − Output Power − W 10 50 Gain = 18.2 dB VCC = 18 V ZL = 8 Ω + 66 µH 1 f = 1 kHz f = 20 Hz 0.1 0.01 f = 10 kHz 0.001 0.01 G007 Figure 8. 8 20k f − Frequency − Hz 0.1 1 PO − Output Power − W 10 50 G008 Figure 9. Submit Documentation Feedback Copyright © 2010, Texas Instruments Incorporated Product Folder Link(s) :TPA3117D2 TPA3117D2 www.ti.com SLOS672 – OCTOBER 2010 TYPICAL CHARACTERISTICS (continued) (All Measurements taken at 1 kHz, unless otherwise noted. Measurements were made using the TPA3117D2 EVM which is available at ti.com.) TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT POWER (BTL) TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT POWER (BTL) 10 Gain = 18.2 dB VCC = 12 V ZL = 6 Ω + 47 µH THD+N − Total Harmonic Distortion + Noise − % THD+N − Total Harmonic Distortion + Noise − % 10 1 f = 1 kHz 0.1 f = 20 Hz 0.01 f = 10 kHz 0.001 0.01 0.1 1 10 PO − Output Power − W 1 f = 1 kHz f = 20 Hz 0.1 0.01 f = 10 kHz 0.001 0.01 50 0.1 1 10 PO − Output Power − W G010 Figure 10. Figure 11. TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT POWER (BTL) MAXIMUM OUTPUT POWER vs PLIMIT VOLTAGE (BTL) 10 50 G011 16 Gain = 18.2 dB VCC = 12 V ZL = 4 Ω + 33 µH PO(Max) − Maximum Output Power − W THD+N − Total Harmonic Distortion + Noise − % Gain = 18.2 dB VCC = 18 V ZL = 6 Ω + 47 µH 1 f = 1 kHz 0.1 0.01 f = 20 Hz 14 Gain = 18.2 dB VCC = 24 V ZL = 8 Ω + 66 µH 12 10 8 6 4 2 f = 10 kHz 0.001 0.01 0.1 1 PO − Output Power − W 10 50 0 0.0 G012 Figure 12. 0.5 1.0 1.5 2.0 2.5 VPLIMIT − PLIMIT Voltage − V 3.0 G013 Figure 13. Submit Documentation Feedback Copyright © 2010, Texas Instruments Incorporated Product Folder Link(s) :TPA3117D2 9 TPA3117D2 SLOS672 – OCTOBER 2010 www.ti.com TYPICAL CHARACTERISTICS (continued) (All Measurements taken at 1 kHz, unless otherwise noted. Measurements were made using the TPA3117D2 EVM which is available at ti.com.) OUTPUT POWER vs PLIMIT VOLTAGE (BTL) OUTPUT POWER vs SUPPLY VOLTAGE (BTL) 30 35 Gain = 18.2 dB VCC = 12 V ZL = 4 Ω + 33 µH 25 25 PO − Output Power − W PO − Output Power − W 30 Gain = 18.2 dB ZL = 8 Ω + 66 µH 20 15 10 20 THD = 10% 15 THD = 1% 10 5 5 0 0 0 1 2 3 4 5 VPLIMIT − PLIMIT Voltage − V 6 6 8 10 G014 Note: Dashed Lines represent thermally limited regions. Figure 14. 14 16 20 22 24 26 G016 EFFICIENCY vs OUTPUT POWER (BTL) 25 100 Gain = 18.2 dB ZL = 4 Ω + 33 µH VCC = 12 V 90 20 VCC = 18 V VCC = 24 V 80 70 η − Efficiency − % THD = 10% 15 THD = 1% 10 60 50 40 30 5 20 Gain = 18.2 dB ZL = 8 Ω + 66 µH 10 0 0 6 8 10 12 14 16 VCC − Supply Voltage − V Figure 16. 18 0 5 10 15 20 25 30 35 PO − Output Power − W G017 Note: Dashed Lines represent thermally limited regions. 10 18 Note: Dashed Lines represent thermally limited regions. Figure 15. OUTPUT POWER vs SUPPLY VOLTAGE (BTL) PO − Output Power − W 12 VCC − Supply Voltage − V 40 G018 Note: Dashed Lines represent thermally limited regions. Figure 17. Submit Documentation Feedback Copyright © 2010, Texas Instruments Incorporated Product Folder Link(s) :TPA3117D2 TPA3117D2 www.ti.com SLOS672 – OCTOBER 2010 TYPICAL CHARACTERISTICS (continued) (All Measurements taken at 1 kHz, unless otherwise noted. Measurements were made using the TPA3117D2 EVM which is available at ti.com.) EFFICIENCY vs OUTPUT POWER (BTL with LC FILTER) EFFICIENCY vs OUTPUT POWER (BTL) 100 100 VCC = 12 V 90 VCC = 18 V 90 80 VCC = 24 V η − Efficiency − % 70 60 50 40 30 60 50 40 30 20 20 Gain = 18.2 dB LC Filter = 22 µH + 0.68 µF RL = 8 Ω 10 Gain = 18.2 dB ZL = 6 Ω + 47 µH 10 0 0 0 5 10 15 20 PO − Output Power − W 25 0 5 10 15 20 PO − Output Power − W G032 25 G019 Note: Dashed Lines represent thermally limited regions. Figure 19. Figure 18. EFFICIENCY vs OUTPUT POWER (BTL with LC FILTER) EFFICIENCY vs OUTPUT POWER (BTL) 100 100 90 90 VCC = 12 V VCC = 12 V 80 80 VCC = 18 V 70 η − Efficiency − % 70 η − Efficiency − % VCC = 18 V 80 70 η − Efficiency − % VCC = 12 V 60 50 40 30 60 50 40 30 20 20 Gain = 18.2 dB LC Filter = 22 µH + 0.68 µF RL = 6 Ω 10 Gain = 18.2 dB ZL = 4 Ω + 33 µH 10 0 0 0 5 10 15 PO − Output Power − W 20 25 0 G033 Figure 20. 3 6 9 12 15 PO − Output Power − W 18 G020 Figure 21. Submit Documentation Feedback Copyright © 2010, Texas Instruments Incorporated Product Folder Link(s) :TPA3117D2 11 TPA3117D2 SLOS672 – OCTOBER 2010 www.ti.com TYPICAL CHARACTERISTICS (continued) (All Measurements taken at 1 kHz, unless otherwise noted. Measurements were made using the TPA3117D2 EVM which is available at ti.com.) EFFICIENCY vs OUTPUT POWER (BTL with LC FILTER) SUPPLY CURRENT vs TOTAL OUTPUT POWER (BTL) 2.6 100 2.4 90 ICC − Supply Current − A 2.0 70 η − Efficiency − % VCC = 18 V 2.2 VCC = 12 V 80 60 50 40 30 1.8 1.6 VCC = 12 V 1.4 1.2 VCC = 24 V 1.0 0.8 0.6 20 Gain = 18.2 dB LC Filter = 22 µH + 0.68 µF RL = 4 Ω 10 0.4 Gain = 18.2 dB ZL = 8 Ω + 66 µH 0.2 0.0 0 0 5 10 15 20 0 25 5 10 20 25 35 40 G021 Note: Dashed Lines represent thermally limited regions. Figure 23. SUPPLY CURRENT vs TOTAL OUTPUT POWER (BTL) CROSSTALK vs FREQUENCY (BTL) 3.2 −20 Gain = 18.2 dB ZL = 4 Ω + 33 µH 2.8 −30 −40 2.4 Gain = 18.2 dB VCC = 12 V VO = 1 Vrms ZL = 8 Ω + 66 µH −50 Crosstalk − dB 2.0 VCC = 12 V 1.6 1.2 −60 −70 −80 Right to Left −90 −100 0.8 Left to Right −110 0.4 −120 0.0 0 5 10 15 20 25 PO(Tot) − Total Output Power − W 30 −130 20 G022 Note: Dashed Lines represent thermally limited regions. Figure 24. 12 30 G034 Figure 22. ICC − Supply Current − A 15 PO(Tot) − Total Output Power − W PO − Output Power − W Submit Documentation Feedback 100 1k 10k 20k f − Frequency − Hz G023 Figure 25. Copyright © 2010, Texas Instruments Incorporated Product Folder Link(s) :TPA3117D2 TPA3117D2 www.ti.com SLOS672 – OCTOBER 2010 TYPICAL CHARACTERISTICS (continued) (All Measurements taken at 1 kHz, unless otherwise noted. Measurements were made using the TPA3117D2 EVM which is available at ti.com.) SUPPLY RIPPLE REJECTION RATIO vs FREQUENCY (BTL) TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY (PBTL) −20 10 Gain = 18.2 dB Vripple = 200 mVpp ZL = 8 Ω + 66 µH THD+N − Total Harmonic Distortion + Noise − % KSVR − Supply Ripple Rejection Ratio − dB 0 −40 −60 VCC = 12 V −80 −100 −120 20 100 1k 10k Gain = 18.2 dB VCC = 24 V ZL = 4 Ω + 33 µH 1 PO = 5 W 0.1 PO = 0.5 W 0.01 PO = 2.5 W 0.001 20 20k 100 1k 10k 20k f − Frequency − Hz f − Frequency − Hz G025 G024 Figure 26. Figure 27. TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT POWER (PBTL) OUTPUT POWER vs SUPPLY VOLTAGE (PBTL) 40 Gain = 18.2 dB VCC = 24 V ZL = 4 Ω + 33 µH Gain = 18.2 dB ZL = 4 Ω + 33 µH 35 30 1 PO − Output Power − W THD+N − Total Harmonic Distortion + Noise − % 10 f = 1 kHz 0.1 25 THD = 10% 20 THD = 1% 15 10 0.01 f = 20 Hz 5 f = 10 kHz 0.001 0.01 0 0.1 1 PO − Output Power − W Figure 28. 10 50 6 8 10 12 14 16 18 VCC − Supply Voltage − V G026 20 G028 Note: Dashed Lines represent thermally limited regions. Figure 29. Submit Documentation Feedback Copyright © 2010, Texas Instruments Incorporated Product Folder Link(s) :TPA3117D2 13 TPA3117D2 SLOS672 – OCTOBER 2010 www.ti.com TYPICAL CHARACTERISTICS (continued) (All Measurements taken at 1 kHz, unless otherwise noted. Measurements were made using the TPA3117D2 EVM which is available at ti.com.) EFFICIENCY vs OUTPUT POWER (PBTL) SUPPLY CURRENT vs OUTPUT POWER (PBTL) 2.8 100 2.4 80 2.2 η − Efficiency − % 70 ICC − Supply Current − A VCC = 18 V VCC = 12 V 60 50 40 30 2.0 1.8 VCC = 12 V 1.6 1.4 VCC = 18 V 1.2 1.0 0.8 0.6 20 0.4 Gain = 18.2 dB ZL = 4 Ω + 33 µH 10 0.2 0.0 0 0 5 10 15 20 25 30 35 40 0 45 15 20 25 30 35 40 G029 Figure 31. SUPPLY RIPPLE REJECTION RATIO vs FREQUENCY (PBTL) SUPPLY RIPPLE REJECTION RATIO vs FREQUENCY (REG_OUT) 45 G030 0 PSRR – Power Supply Rejection Ratio – dB Gain = 18.2 dB Vripple = 200 mVpp ZL = 8 Ω + 66 µH −40 −60 VCC = 12 V −80 −100 −120 20 10 Figure 30. 0 −20 5 PO − Output Power − W PO − Output Power − W KSVR − Supply Ripple Rejection Ratio − dB Gain = 18.2 dB ZL = 4 Ω + 33 µH 2.6 90 100 1k 10k 20k –20 VCC = 12 V, VO = 4.9 V, IL = 20 mA –40 –60 –80 –100 –120 20 f − Frequency − Hz 100 1k f – Frequency – Hz 10k 20k G031 Figure 32. 14 Figure 33. Submit Documentation Feedback Copyright © 2010, Texas Instruments Incorporated Product Folder Link(s) :TPA3117D2 TPA3117D2 www.ti.com SLOS672 – OCTOBER 2010 DEVICE INFORMATION Gain setting via GAIN0 and GAIN1 inputs The gain of the TPA3117D2 is set by two input terminals, GAIN0 and GAIN1. The gains listed in Table 1 are realized by changing the taps on the input resistors inside the amplifier. This causes the input impedance (ZI) to be dependent on the gain setting. The actual gain settings are controlled by ratios of resistors, so the gain variation from part-to-part is small. However, the input impedance from part-to-part at the same gain may shift by ±20% due to shifts in the actual resistance of the input resistors. For design purposes, the input network (discussed in the next section) should be designed assuming an input impedance of 60 kΩ, which is the absolute minimum input impedance of the TPA3117D2. At the lower gain settings, the input impedance could increase as high as 256 kΩ Table 1. Gain Setting GAIN1 GAIN0 AMPLIFIER GAIN (dB) INPUT IMPEDANCE (kΩ) TYP TYP 0 0 9 213 0 1 12.1 149 1 0 15.2 104 1 1 18.2 74 SD OPERATION The TPA3117D2 employs a shutdown mode of operation designed to reduce supply current (ICC) to the absolute minimum level during periods of nonuse for power conservation. The SD input terminal should be held high (see specification table for trip point) during normal operation when the amplifier is in use. Pulling SD low causes the outputs to mute and the amplifier to enter a low-current state. Never leave SD unconnected, because amplifier operation would be unpredictable. For the best power-off pop performance, place the amplifier in the shutdown mode prior to removing the power supply voltage. 5V regulator (REG_OUT) is active in the shutdown state. PLIMIT The voltage at pin 7 can used to limit the power to levels below that which is possible based on the supply rail. Add a resistor divider from REG_OUT to ground to set the voltage at the PLIMIT pin. An external reference may also be used if tighter tolerance is required. Also add a 1mF capacitor from pin 7 to ground. Vinput PLIMIT = REG_OUT Pout = 11.8W PLIMIT = 3V Pout = 10W PLIMIT = 1.8V Pout = 5W TPA3117D2 Power Limit Function Vin=1.13VPP Freq=1kHz RLoad=8W Figure 34. PLIMIT Circuit Operation Submit Documentation Feedback Copyright © 2010, Texas Instruments Incorporated Product Folder Link(s) :TPA3117D2 15 TPA3117D2 SLOS672 – OCTOBER 2010 www.ti.com The PLIMIT circuit sets a limit on the output peak-to-peak voltage. The limiting is done by limiting the duty cycle to fixed maximum value. This limit can be thought of as a "virtual" voltage rail which is lower than the supply connected to PVCC. This "virtual" rail is 4 times the voltage at the PLIMIT pin. This output voltage can be used to calculate the maximum output power for a given maximum input voltage and speaker impedance. POUT ææ ö ö RL çç ç ÷ x VP ÷÷ è RL + 2 x RS ø ø = è 2 x RL 2 for unclipped power (1) Where: RS is the total series resistance including RDS(on), and any resistance in the output filter. RL is the load resistance. VP is the peak amplitude of the output possible within the supply rail. VP = 4 × PLIMIT voltage if PLIMIT < 4 × VP POUT (10% THD) = 1.25 × POUT (unclipped) REG_OUT Regulator The TPA3117D2 has an integrated 5V regulator for driving external circuitry. Maximum output current is 30mA. The regulator is always active when power is applied to the device. The SD pin does not disable operation. Connect a series 10Ω resister followed by a 2.2µF capacitor to AGND before routing to the external circuitry. When not used for powering external devices, a series 10Ω resistor with 2.2 µF of decoupling is still required. PBTL Select TPA3117D2 offers the feature of parallel BTL operation with two outputs of each channel connected directly. If the PBTL pin (pin 11) is tied high, the positive and negative outputs of each channel (left and right) are synchronized and in phase. To operate in this PBTL (mono) mode, apply the input signal to the RIGHT input and place the speaker between the LEFT and RIGHT outputs. Connect the positive and negative output together for best efficiency. For normal BTL operation, connect the PBTL pin to local ground. SHORT-CIRCUIT PROTECTION TPA3117D2 has protection from overcurrent conditions caused by a short circuit on the output stage. Amplifier outputs are switched to a Hi-Z state when the short circuit protection latch is engaged. After a typical delay of 250ms, the outputs will resume normal operation until another short occurs. It is not necessary to cycle pin SD to restart the device operation after a short circuit event. THERMAL PROTECTION Thermal protection on the TPA3117D2 prevents damage to the device when the internal die temperature exceeds 150°C. There is a ±15°C tolerance on this trip point from device to device. Once the die temperature exceeds the thermal set point, the device enters into the shutdown state and the outputs are disabled. This is not a latched fault. The thermal fault is cleared once the temperature of the die is reduced by 15°C. The device begins normal operation at this point with no external system interaction. It is not necessary to cycle SD terminal to restart device operation after a short circuit event. FSEL FUNCTIONALITY This terminal is used to select the switching frequency of the amplifier. In applications where more than one device is needed, configure one device with FSEL = LOW (290kHz switching) and the other device with FSEL = HIGH (390kHz switching). 16 Submit Documentation Feedback Copyright © 2010, Texas Instruments Incorporated Product Folder Link(s) :TPA3117D2 TPA3117D2 www.ti.com SLOS672 – OCTOBER 2010 APPLICATION INFORMATION PVCC 100 μF 0.1 μF 1000 pF 100 kΩ Control System 30 31 1 mF 32 1 mF 1 2 3 PVCC 4 1 mF 5 1 mF 2.2 mF 10 W 6 10 kW 7 SD PVCCL FSEL PVCCL LINP BSPL LINN OUTPL GAIN0 PGND GAIN1 OUTNL BSNL AVCC TPA3117D2 BSNR AGND REG_OUT OUTNR PGND PLIMIT 27 26 25 0.22 μF FB 24 1000 pF 23 22 1000 pF 21 0.22 μF 20 0.22 μF FB FB 19 1000 pF 18 10 kΩ 1 mF Audio Source 8 9 RINN OUTPR RINP BSPR 1 mF PVCCR 11 PBTL GND PVCCR 17 1000 pF 16 FB 0.22 μF 15 100 μF 14 0.1 μF 1000 pF Thermal Pad PVCC Note: Pins 10, 12, 13, 28 and 29 are NC (not internally connected) Figure 35. Stereo Class-D Amplifier with BTL Output and Single-Ended Inputs Submit Documentation Feedback Copyright © 2010, Texas Instruments Incorporated Product Folder Link(s) :TPA3117D2 17 TPA3117D2 SLOS672 – OCTOBER 2010 www.ti.com TPA3117D2 Modulation Scheme The TPA3117D2 uses a modulation scheme that allows operation without the classic LC reconstruction filter when the amp is driving an inductive load. Each output is switching from 0 volts to the supply voltage. The OUTP and OUTN are in phase with each other with no input so that there is little or no current in the speaker. The duty cycle of OUTP is greater than 50% and OUTN is less than 50% for positive output voltages. The duty cycle of OUTP is less than 50% and OUTN is greater than 50% for negative output voltages. The voltage across the load sits at 0V throughout most of the switching period, reducing the switching current, which reduces any I2R losses in the load. OUTP OUTN OUTP OUTP-OUTN No Output 0V Speaker Current OUTP OUTN Positive Output PVCC OUTP-OUTN 0V Speaker Current 0A OUTP Negative Output OUTN OUTP-OUTN 0V -PVCC Speaker 0A Current Figure 36. The TPA3117D2 Output Voltage and Current Waveforms Into an Inductive Load Ferrite Bead Filter Considerations Using the Advanced Emissions Suppression Technology in the TPA3117D2 amplifier it is possible to design a high efficiency Class-D audio amplifier while minimizing interference to surrounding circuits. It is also possible to accomplish this with only a low-cost ferrite bead filter. In this case it is necessary to carefully select the ferrite bead used in the filter. One important aspect of the ferrite bead selection is the type of material used in the ferrite bead. Not all ferrite material is alike, so it is important to select a material that is effective in the 10 to 100 MHz range which is key to the operation of the Class D amplifier. Many of the specifications regulating consumer electronics have emissions limits as low as 30 MHz. It is important to use the ferrite bead filter to block radiation in the 30 MHz and above range from appearing on the speaker wires and the power supply lines which are good antennas for these signals. The impedance of the ferrite bead can be used along with a small capacitor with a value in the range of 1000 pF to reduce the frequency spectrum of the signal to an acceptable level. For best performance, the resonant frequency of the ferrite bead/ capacitor filter should be less than 10 MHz. 18 Submit Documentation Feedback Copyright © 2010, Texas Instruments Incorporated Product Folder Link(s) :TPA3117D2 TPA3117D2 www.ti.com SLOS672 – OCTOBER 2010 Also, it is important that the ferrite bead is large enough to maintain its impedance at the peak currents expected for the amplifier. Some ferrite bead manufacturers specify the bead impedance at a variety of current levels. In this case it is possible to make sure the ferrite bead maintains an adequate amount of impedance at the peak current the amplifier will see. If these specifications are not available, it is also possible to estimate the bead current handling capability by measuring the resonant frequency of the filter output at low power and at maximum power. A change of resonant frequency of less than fifty percent under this condition is desirable. Examples of ferrite beads which have been tested and work well with the TPA3117D2 include 28L0138-80R-10 and HI1812V101R-10 from Steward and the 742792510 from Wurth Electronics. A high quality ceramic capacitor (x5R or better) is also needed for the ferrite bead filter. A low ESR capacitor with good temperature and voltage characteristics will work best. Additional EMC improvements may be obtained by adding snubber networks from each of the class D outputs to ground. Suggested values for a simple RC series snubber network would be 10 Ω in series with a 330 pF capacitor although design of the snubber network is specific to every application and must be designed taking into account the parasitic reactance of the printed circuit board as well as the audio amp. Take care to evaluate the stress on the component in the snubber network especially if the amplifer is running at high PVCC. Also, make sure the layout of the snubber network is tight and returns directly to the PGND or the thermal pad beneath the chip. Efficiency: LC Filter Required With the Traditional Class-D Modulation Scheme The main reason that the traditional class-D amplifier needs an output filter is that the switching waveform results in maximum current flow. This causes more loss in the load, which causes lower efficiency. The ripple current is large for the traditional modulation scheme, because the ripple current is proportional to voltage multiplied by the time at that voltage. The differential voltage swing is 2 × VCC, and the time at each voltage is half the period for the traditional modulation scheme. An ideal LC filter is needed to store the ripple current from each half cycle for the next half cycle, while any resistance causes power dissipation. The speaker is both resistive and reactive, whereas an LC filter is almost purely reactive. The TPA3117D2 modulation scheme has little loss in the load without a filter because the pulses are short and the change in voltage is VCC instead of 2 × VCC. As the output power increases, the pulses widen, making the ripple current larger. Ripple current could be filtered with an LC filter for increased efficiency, but for most applications the filter is not needed. An LC filter with a cutoff frequency less than the class-D switching frequency allows the switching current to flow through the filter instead of the load. The filter has less resistance but higher impedance at the switching frequency than the speaker, which results in less power dissipation, therefore increasing efficiency. When to Use an Output Filter for EMI Suppression The TPA3117D2 has been tested with a simple ferrite bead filter for a variety of applications including long speaker wires up to 125 cm and high power. The TPA3117D2 EVM passes FCC Class B specifications under these conditions using twisted speaker wires. The size and type of ferrite bead can be selected to meet application requirements. Also, the filter capacitor can be increased if necessary with some impact on efficiency. There may be a few circuit instances where it is necessary to add a complete LC reconstruction filter. These circumstances might occur if there are nearby circuits which are sensitive to noise. In these cases a classic second order Butterworth filter similar to those shown in the figures below can be used. Some systems have little power supply decoupling from the AC line but are also subject to line conducted interference (LCI) regulations. These include systems powered by "wall warts" and "power bricks." In these cases, an LC reconstruction filter can be the lowest cost means to pass LCI tests. Common mode chokes using low frequency ferrite material can also be effective at preventing line conducted interference. Submit Documentation Feedback Copyright © 2010, Texas Instruments Incorporated Product Folder Link(s) :TPA3117D2 19 TPA3117D2 SLOS672 – OCTOBER 2010 www.ti.com 33 mH OUTP L1 C2 1 mF 33 mH OUTN L2 C3 1 mF Figure 37. Typical LC Output Filter, Cutoff Frequency of 27 kHz, Speaker Impedance = 8 Ω 15 mH OUTP L1 C2 2.2 mF 15 mH OUTN L2 C3 2.2 mF Figure 38. Typical LC Output Filter, Cutoff Frequency of 27 kHz, Speaker Impedance = 4 Ω Ferrite Chip Bead OUTP 1 nF Ferrite Chip Bead OUTN 1 nF Figure 39. Typical Ferrite Chip Bead Filter (Chip Bead Example: ) 20 Submit Documentation Feedback Copyright © 2010, Texas Instruments Incorporated Product Folder Link(s) :TPA3117D2 TPA3117D2 www.ti.com SLOS672 – OCTOBER 2010 INPUT RESISTANCE Changing the gain setting can vary the input resistance of the amplifier from its smallest value, ±20%, to the largest value, ±20%. As a result, if a single capacitor is used in the input high-pass filter, the -3 dB or cutoff frequency may change when changing gain steps. Zf Ci IN Input Signal Zi The -3-dB frequency can be calculated using Equation 2. Use the ZI values given in Table 1. f = 1 2p Zi Ci (2) INPUT CAPACITOR, CI In the typical application, an input capacitor (CI) is required to allow the amplifier to bias the input signal to the proper dc level for optimum operation. In this case, CI and the input impedance of the amplifier (ZI) form a high-pass filter with the corner frequency determined in Equation 3. -3 dB fc = 1 2p Zi Ci fc (3) The value of CI is important, as it directly affects the bass (low-frequency) performance of the circuit. Consider the example where ZI is 60 kΩ and the specification calls for a flat bass response down to 20 Hz. Equation 3 is reconfigured as Equation 4. Ci = 1 2p Zi fc (4) In this example, CI is 0.13 µF; so, one would likely choose a value of 0.15 mF as this value is commonly used. If the gain is known and is constant, use ZI from Table 1 to calculate CI. A further consideration for this capacitor is the leakage path from the input source through the input network (CI) and the feedback network to the load. This leakage current creates a dc offset voltage at the input to the amplifier that reduces useful headroom, especially in high gain applications. For this reason, a low-leakage tantalum or ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor should face the amplifier input in most applications as the dc level there is held at 3 V, which is likely higher than the source dc level. Note that it is important to confirm the capacitor polarity in the application. Additionally, lead-free solder can create dc offset voltages and it is important to ensure that boards are cleaned properly. Submit Documentation Feedback Copyright © 2010, Texas Instruments Incorporated Product Folder Link(s) :TPA3117D2 21 TPA3117D2 SLOS672 – OCTOBER 2010 www.ti.com POWER SUPPLY DECOUPLING, CS The TPA3117D2 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to ensure that the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also prevents oscillations for long lead lengths between the amplifier and the speaker. Optimum decoupling is achieved by using a network of capacitors of different types that target specific types of noise on the power supply leads. For higher frequency transients due to parasitic circuit elements such as bond wire and copper trace inductances as well as lead frame capacitance, a good quality low equivalent-series-resistance (ESR) ceramic capacitor of value between 220 pF and 1000 pF works well. This capacitor should be placed as close to the device PVCC pins and system ground (either PGND pins or PowerPad) as possible. For mid-frequency noise due to filter resonances or PWM switching transients as well as digital hash on the line, another good quality capacitor typically 0.1 mF to 1 µF placed as close as possible to the device PVCC leads works best For filtering lower frequency noise signals, a larger aluminum electrolytic capacitor of 220 mF or greater placed near the audio power amplifier is recommended. The 220 mF capacitor also serves as a local storage capacitor for supplying current during large signal transients on the amplifier outputs. The PVCC terminals provide the power to the output transistors, so a 220 µF or larger capacitor should be placed on each PVCC terminal. A 10 µF capacitor on the AVCC terminal is adequate. Also, a small decoupling resistor between AVCC and PVCC can be used to keep high frequency class D noise from entering the linear input amplifiers. BSN and BSP CAPACITORS The full H-bridge output stages use only NMOS transistors. Therefore, they require bootstrap capacitors for the high side of each output to turn on correctly. A 0.22 mF ceramic capacitor, rated for at least 25 V, must be connected from each output to its corresponding bootstrap input. Specifically, one 0.22 mF capacitor must be connected from OUTPx to BSPx, and one 0.22 mF capacitor must be connected from OUTNx to BSNx. (See the application circuit diagram in Figure 1.) The bootstrap capacitors connected between the BSxx pins and corresponding output function as a floating power supply for the high-side N-channel power MOSFET gate drive circuitry. During each high-side switching cycle, the bootstrap capacitors hold the gate-to-source voltage high enough to keep the high-side MOSFETs turned on. DIFFERENTIAL INPUTS The differential input stage of the amplifier cancels any noise that appears on both input lines of the channel. To use the TPA3117D2 with a differential source, connect the positive lead of the audio source to the INP input and the negative lead from the audio source to the INN input. To use the TPA3117D2 with a single-ended source, ac ground the INP or INN input through a capacitor equal in value to the input capacitor on INN or INP and apply the audio source to either input. In a single-ended input application, the unused input should be ac grounded at the audio source instead of at the device input for best noise performance. For good transient performance, the impedance seen at each of the two differential inputs should be the same. The impedance seen at the inputs should be limited to an RC time constant of 1 ms or less if possible. This is to allow the input dc blocking capacitors to become completely charged during the 14 ms power-up time. If the input capacitors are not allowed to completely charge, there will be some additional sensitivity to component matching which can result in pop if the input components are not well matched. USING LOW-ESR CAPACITORS Low-ESR capacitors are recommended throughout this application section. A real (as opposed to ideal) capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this resistance, the more the real capacitor behaves like an ideal capacitor. 22 Submit Documentation Feedback Copyright © 2010, Texas Instruments Incorporated Product Folder Link(s) :TPA3117D2 TPA3117D2 www.ti.com SLOS672 – OCTOBER 2010 PRINTED-CIRCUIT BOARD (PCB) LAYOUT The TPA3117D2 can be used with a small, inexpensive ferrite bead output filter for most applications. However, since the Class-D switching edges are fast, it is necessary to take care when planning the layout of the printed circuit board. The following suggestions will help to meet EMC requirements. • Decoupling capacitors—The high-frequency decoupling capacitors should be placed as close to the PVCC and AVCC terminals as possible. Large (220 µF or greater) bulk power supply decoupling capacitors should be placed near the TPA3117D2 on the PVCCL and PVCCR supplies. Local, high-frequency bypass capacitors should be placed as close to the PVCC pins as possible. These caps can be connected to the thermal pad directly for an excellent ground connection. Consider adding a small, good quality low ESR ceramic capacitor between 220 pF and 1000 pF and a larger mid-frequency cap of value between 0.1mF and 1mF also of good quality to the PVCC connections at each end of the chip. • Keep the current loop from each of the outputs through the ferrite bead and the small filter cap and back to PGND as small and tight as possible. The size of this current loop determines its effectiveness as an antenna. • Grounding—The AVCC (pin 4) decoupling capacitor should be grounded to analog ground (AGND). The PVCC decoupling capacitors should connect to PGND. Analog ground and power ground should be connected at the thermal pad, which should be used as a central ground connection or star ground for the TPA3117D2. • Output filter—The ferrite EMI filter (Figure 39) should be placed as close to the output terminals as possible for the best EMI performance. The LC filter (Figure 37 and Figure 38) should be placed close to the outputs. The capacitors used in both the ferrite and LC filters should be grounded to power ground. • Thermal Pad—The thermal pad must be soldered to the PCB for proper thermal performance and optimal reliability. For recommended PCB footprints, see figures at the end of this data sheet. For an example layout, see the TPA3117D2 Evaluation Module (TPA3117D2EVM) User Manual. Both the EVM user manual and the thermal pad application report are available on the TI Web site at http://www.ti.com. Submit Documentation Feedback Copyright © 2010, Texas Instruments Incorporated Product Folder Link(s) :TPA3117D2 23 PACKAGE OPTION ADDENDUM www.ti.com 30-Oct-2010 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Drawing Pins Package Qty Eco Plan (2) Lead/ Ball Finish MSL Peak Temp (3) TPA3117D2RHBR ACTIVE QFN RHB 32 3000 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR TPA3117D2RHBT ACTIVE QFN RHB 32 250 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR Samples (Requires Login) (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. 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Addendum-Page 1 PACKAGE MATERIALS INFORMATION www.ti.com 28-Oct-2010 TAPE AND REEL INFORMATION *All dimensions are nominal Device Package Package Pins Type Drawing SPQ Reel Reel A0 Diameter Width (mm) (mm) W1 (mm) B0 (mm) K0 (mm) P1 (mm) W Pin1 (mm) Quadrant TPA3117D2RHBR QFN RHB 32 3000 330.0 12.4 5.3 5.3 1.5 8.0 12.0 Q2 TPA3117D2RHBT QFN RHB 32 250 180.0 12.4 5.3 5.3 1.5 8.0 12.0 Q2 Pack Materials-Page 1 PACKAGE MATERIALS INFORMATION www.ti.com 28-Oct-2010 *All dimensions are nominal Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm) TPA3117D2RHBR QFN RHB 32 3000 346.0 346.0 29.0 TPA3117D2RHBT QFN RHB 32 250 190.5 212.7 31.8 Pack Materials-Page 2 IMPORTANT NOTICE Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, modifications, enhancements, improvements, and other changes to its products and services at any time and to discontinue any product or service without notice. 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