IR3638 High Frequency Synchronous Step Down PWM Controller for Tracking Applications The IR3638 controller IC is designed to provide a simple synchronous Buck regulator for on−board DC to DC applications in a 14−pin SOIC. The IR3638 is designed specifically for tracking applications by providing the track input. The IR3638 operates at a fixed internal 400 kHz switching frequency allowing the use of small external components. The device features a programmable soft start set by an external capacitor, under−voltage lockout and output under−voltage detection that latches off the device when an output short is detected. Features • • • • • Power up Sequencing / Tracking Enable Input Internal 400 kHz Oscillator Programmable Soft−Start Fixed Frequency Voltage Mode 14 1 A WL Y WW G MARKING DIAGRAM SOIC−14 D SUFFIX CASE 751A = Assembly Location = Wafer Lot = Year = Work Week = Pb−Free Package IR3638G AWLYWW 1 PIN CONNECTIONS FB VP NC 1 SS COMP NC Applications • • • • • http://onsemi.com Tracking Applications Game Consoles Computing Peripheral Voltage Regulators Graphics Cards General DC to DC Converters VCC NC NC VC LDRV HDRV GND PGND (Top View) ORDERING INFORMATION See detailed ordering and shipping information in the package dimensions section on page 12 of this data sheet. Figure 1. Typical Application Circuit © Semiconductor Components Industries, LLC, 2009 June, 2009 − Rev. 0 1 Publication Order Number: IR3638/D http://onsemi.com 2 Figure 2. Simplified Block Diagram COMP FB VP/EN SS GND VCC 22 mA VBIAS VCC UVLO 25 k 25 k POR POR Error Amp 0.4 V Error Comparator CT Oscillator 64 mA Max 0.65 V VP/EN S R Q POR S R Q Reset Dom POR POR PWM FAULT SS 2V FAULT Delay Delay VCC PGND LDRV HDRV VC IR3638 CIrcuit Description: Block Diagram IR3638 Table 1. PIN FUNCTION DESCRIPTION Pin Name Description 1 FB Inverting input to the error amplifier. This pin is connected to the output of the regulator via resistor divider to set the output voltage and provide feedback to the error amplifier. 2 VP/EN 3 NC 4 VCC 5 NC 6 LDRV Output driver for low side MOSFET. 7 GND IC ground for internal control circuitry. 8 PGND Power Ground. This pin serves as a separate ground for the MOSFET drivers and should be connected to the system’s power ground plane. 9 HDRV Output driver for high side MOSFET. The negative voltage at this pin may cause instability for the gate drive circuit. To prevent this, a low forward voltage drop diode (e.g. BAT54 or 1N4148) is required between this pin and Power Ground. 10 VC This pin powers the high side driver. 11 NC No Connect 12 COMP 13 SS Soft start. This pin provides user programmable soft−start function. Connect an external capacitor from this pin to ground to set the start up time of the output voltage. 14 NC No Connect Dual function pin. Non inverting input to the error amplifier. Enable input. No Connect This pin provides power for the internal blocks of the IC as well as powers the low side driver. A minimum of 0.1 mF, high frequency capacitor must be connected from this pin to power ground. No Connect Output of error amplifier. An external resistor and capacitor network is typically connected from this pin to ground to provide loop compensation. Table 2. ABSOLUTE MAXIMUM RATINGS Rating Main Supply Voltage Input Main Supply Voltage Input 200 ns wide spikes, 400 kHz Supply Voltage for the High side driver Supply Voltage for the High side driver 200 ns wide spikes, 400 kHz VP/EN pin Voltage FB pin Voltage Rating Symbol min max Unit VCC −0.3 20 V VCC_SPK −0.3 22 V VC −0.3 20 V VC_SPK −0.3 22 V VP/EN −0.3 10 or VCC (Note 1) V VFB −0.3 10 or VCC (Note 1) V Symbol Value Unit Thermal Resistance, Junction−to−Ambient (Note 2) Rthja 90 K/W Storage Temperature Range Tstg −65 to 150 °C Junction Operating Temperature TJ 0 to 150 °C ESD Withstand Voltage (Note 3) Human Body Model Machine Model VESD 2.0 200 kV V Moisture Sensitivity Level MSL JEDEC Level 1 @ 260°C Stresses exceeding Maximum Ratings may damage the device. Maximum Ratings are stress ratings only. Functional operation above the Recommended Operating Conditions is not implied. Extended exposure to stresses above the Recommended Operating Conditions may affect device reliability. NOTE: All voltages are referenced to GND pin unless otherwise stated. 1. Maximum = 10 V or VCC, whichever is lower. 2. JEDEC High−K model 3. This device series contains ESD protection and exceeds the following tests: Human Body Model (HBM) ±2.0 kV per JEDEC standard: JESD22−A114 Machine Model (MM) ±200 V per JEDEC standard: JESD22−A115 http://onsemi.com 3 IR3638 Table 3. RECOMMENDED OPERATING CONDITIONS Symbol Definition Min Max Units VCC Supply Voltage 7 20 V VC Supply Voltage Converter Voltage + 5 V, (Note 4) 20 V TJ Junction Temperature 0 125 °C NOTE: All voltages are referenced to GND pin. 4. Depend on high side MOSFET VGS Table 4. ELECTRICAL SPECIFICATIONS Unless otherwise specified, VCC = VC = 12 V, 0°C < TJ < 125°C Parameter Symbol Test Condition ICC(Static) Min Typ Max Units VP/EN = 0 V, No Switching 1.5 5 mA ICC(Dynamic) fSW = 400 kHz, CL = 1.5 nF 10 15 mA IC(Static) VP/EN = 0 V, No Switching 0.1 1 mA IC(Dynamic) fSW = 400 kHz, CL = 1.5 nF 9 15 mA VCC−Start−Threshold VCC UVLO (R) Supply voltage Rising 6.3 6.6 7.0 V VCC−Stop−Threshold VCC UVLO (F) Supply voltage Falling 6.0 6.3 6.6 V VCC (Hyst) Supply ramping up and down 0.2 0.3 0.4 V Enable−Start−Threshold VP/EN UVLO (R) Supply voltage Rising 0.6 0.65 0.7 V Enable−Stop−Threshold VP/EN UVLO (F) Supply voltage Falling 0.56 0.6 0.66 V VP/EN (Hyst) Supply ramping up and down 25 42.5 60 mV VFB UVLO FB ramping down 0.3 0.4 0.5 V 360 400 440 kHz SUPPLY CURRENT VCC Supply Current (Static) VCC Supply Current (Dynamic) VC Supply Current (Static) VC Supply Current (Dynamic) UNDER VOLTAGE LOCKOUT VCC−Hysteresis Enable−Hysteresis FB UVLO OSCILLATOR Frequency Ramp Amplitude fSW VRAMP (Note 5) Min Duty Cycle DMIN VFB =1V, VP/EN = 0.8 V Max Duty Cycle DMAX fSW = 400 kHz, VFB = 0.6 V, VP/EN = 0.8 V FB Input Bias Current IFB1 FB Input Bias current 1.25 V 0 % 85 95 % VSS = 3 V −0.1 −0.5 mA IFB2 VSS = 0 V 64 IVP/EN VSS = 3 V −0.1 81 ERROR AMPLIFIER VP/EN Input Bias Current Transconductance gm Input Offset Voltage VOS VP/EN = 0.8 V, VCOMP = 2.0 V −6 VCOMN (Note 5) 0.6 VP/EN Common Mode Range 440 0 mA −0.5 mA 1300 mmho +6 mV 1.5 V ERROR AMPLIFIER DESIGN SPECIFICATIONS OTA output current IOTA (SINK) VFB = 1.2 V, VP/EN = 1.0 V, VCOMP = 2.0 V, (Note 5) 100 mA OTA output current IOTA (SOURCE) VFB = 0.8 V, VP/EN = 1.0 V, VCOMP = 2.0 V, (Note 5) 100 mA 5. Guaranteed by Design but not tested in production. http://onsemi.com 4 IR3638 Table 5. ELECTRICAL SPECIFICATIONS Unless otherwise specified, VCC = VC = 12 V, 0°C < TJ < 125°C Symbol Test Condition Min Typ Max Units Soft Start Current ISS VSS = 0 V 12 22 32 mA Soft Start Turn On SS (on) 1.8 2 2.2 V Parameter SOFT START OUTPUT DRIVERS LO Drive Rise Time tr(Lo) CL = 1.5 nF (See Figure 3) 30 60 ns HI Drive Rise Time tr(Hi) CL = 1.5 nF (See Figure 3) 30 60 ns LO Drive Fall Time tf(Lo) CL = 1.5 nF (See Figure 3) 30 60 ns HI Drive Fall Time tf(Hi) CL = 1.5 nF (See Figure 3) 30 60 ns Dead Band Time tDEAD (See Figure 3) 90 150 ns tr 35 tf 9V High Side Driver (HDRV) tDEAD tDEAD 2V tr tf 9V Low Side Driver (LDRV) 2V Deadband H to L Deadband L to H Figure 3. Definition of Rise/Fall Time and Deadband Time TYPICAL CHARACTERISTICS 7.00 6.90 6.80 VCC (V) 6.70 Rising 6.60 6.50 6.40 6.30 Falling 6.20 6.10 6.00 0 20 40 60 80 TEMPERATURE (°C) Figure 4. VCC UVLO http://onsemi.com 5 100 120 IR3638 TYPICAL CHARACTERISTICS 0.50 0.70 0.48 0.68 0.46 Rising 0.44 0.64 0.62 VFB (V) VP/EN (V) 0.66 Falling 0 20 40 60 80 100 0.32 0.30 120 40 60 80 Figure 5. VP/EN UVLO Figure 6. FB UVLO 440 93 430 100 120 100 120 420 fSW (kHz) 89 87 85 410 400 390 380 83 370 0 20 40 60 80 100 360 120 0 20 40 60 80 TEMPERATURE (°C) TEMPERATURE (°C) Figure 7. Maximum Duty Cycle Figure 8. Switching Frequency 150 1300 1200 130 1100 1000 110 t (ns) 900 800 700 High to Low 90 Low to High 70 600 50 500 400 20 TEMPERATURE (°C) 95 81 0 TEMPERATURE (°C) 91 DMAX (%) 0.38 0.34 0.58 gm (mmho) 0.40 0.36 0.60 0.56 0.42 0 20 40 60 80 100 30 120 0 20 40 60 80 TEMPERATURE (°C) TEMPERATURE (°C) Figure 9. Error Amplifier Transconductance Figure 10. Deadtime http://onsemi.com 6 100 120 IR3638 Detailed Description Introduction The value of the output capacitor should be calculated using the following equation: The IR3638 is voltage mode PWM synchronous controller designated to drive two external N-channel MOSFETs. Switching frequency is fixed at 400 kHz. Output voltage is determined by feedback resistor divider and external reference voltage. Reference voltage input can be used to enabling and disabling operation and for tracking function. C OUT w The undervoltage lockout circuit ensures that the IC does not start and work until VCC and VP/EN are over set thresholds. If these conditions are not fulfilled output drivers are in the off state. Input Capacitor Selection The input capacitor is used to supply current pulses while the high side MOSFET is on. When the MOSFET is off, the input capacitor is being charged. The value of this capacitor can be selected with the Equation (4): Disable Function The output voltage can be disabled by pulling the VP/EN pin below 0.6 V. At this time are output drivers in the off state. I OUT @ C IN w Output Voltage Ǔ (eq. 1) Ǔǒ 1* V OUT V INmax Ǔ Ǔ VOUT VIN (eq. 4) Ǹ ǒ Ǔ VOUT V IN (eq. 5) V IN Power MOSFET Selection The IR3638 uses two N-channel MOSFETs. They can be primarily selected according to RDS(ON), maximum drain to source voltage, and gate charge. RDS(ON) impacts conductive losses and gate charge impacts switching losses. The low side MOSFET is selected primarily for conduction losses, and the high side MOSFET is selected to reduce switching losses especially when the output voltage is less than 30% of the input voltage. The drain to source breakdown voltage must be higher than the maximum input voltage. Conductive power losses can be calculated using the following Equations (6) and (7): The inductor selection is based on the output power, frequency, input and output voltages, and efficiency requirements. High inductor values cause low current ripple, slower transient response, higher efficiency and increased size. Inductor design can be reduced to desired maximum current ripple in the inductor. It is good to have current ripple (ΔILmax) between 20% and 50% of the output current. For a buck converter, the inductor should be chosen according to Equation (2). V OUT f SW @ DI Lmax ǒ @ 1* f SW @ DV IN I RMS + I OUT @ Inductor Selection ǒ VIN V OUT @ 1 * where VP/EN is the external reference voltage at VP/EN pin that is connected to noninverting input of error amplifier. R1 and R2 resistors create voltage divider from output to FB pin that is connected to inverting input of error amplifier. Absolute values of resistors R1 and R2 depend on the compensation network type. See discussion of compensation description for details. L+ VOUT where ΔVIN is the input voltage ripple and the recommended value is about 2–5% of VIN. The input capacitor must be able to handle the input ripple current. Its value should be calculated using Equation (5): Output voltage can be set by an external resistor divider and external reference voltage at VP/EN pin according to Equation (1): ǒ (eq. 3) For a higher switching frequency, it is suitable to use a multilayer ceramic capacitor (MLCC) with very low ESR. The advantages are small size, low output voltage ripple and fast transient response. The disadvantage of the MLCC type is the requirement to use a Type III compensation network. Under-Voltage Lockout V OUT + V PńEN @ 1 ) R1 R2 DI L 8 @ f SW @ (DV OUT * DI L @ ESR) P COND−HIGHFET + I OUT 2 @ R DS(ON) @ ǒ P COND−LOWFET + I OUT 2 @ R DS(ON) @ 1 * (eq. 2) V OUT (eq. 6) V IN Ǔ V OUT V IN (eq. 7) Switching losses are dependent on the drain to source voltage at turn-off state, output current, and switch-on and switch-off times, as is shown by Equation (8). Output Capacitor Selection The output voltage ripple and transient requirements determine the output capacitor type and value. The important parameter for the selection of the output capacitor is equivalent serial resistance (ESR). If the capacitor has low ESR, it often has sufficient capacity for filtering as well as an adequate RMS current rating. P SW + V DS(OFF) 2 @ (t ON ) t OFF) @ f SW @ I OUT (eq. 8) tON and tOFF times are dependent on the transistor gate charge. http://onsemi.com 7 IR3638 The MOSFET output capacitance loss is caused by the charging and discharging during the switching process and can be computed using Equation (9). P COSS + C OSS @ V IN 2 @ f SW 2 QRR is the diode recovery charge as given in the manufacturer’s datasheet. For some types of MOSFETs, this dissipation may be dominant at high input voltages. It is necessary to take care when selecting a MOSFET. An external Schottky diode across the low side MOSFET can be used to eliminate the reverse recovery charge power loss. The Schottky diode’s forward voltage should be lower than that of the body diode, and reverse recovery time (trr) should be lower then that of the body diode. The Schottky diode’s capacitance loss can be calculated as shown in Equation (11). (eq. 9) where COSS = CDS + CGS. Some power dissipation is caused by the reverse recovery charge in the low side MOSFET body diode, which conducts at dead time. This charge is needed to close the diode. The current from the input power supply flows through the high side MOSFET to the low side MOSFET body diode. This power dissipation can be calculated using the following Equation (10): P QRR + Q RR @ V IN @ f SW P C(schottky) + C schottky @ V IN 2 @ f SW 2 (eq. 11) (eq. 10) tdead tdead High Side Logic Signal Low Side Logic Signal td(on) tf RDSmax High Side MOSFET RDS RDS(ON)min tr td(off) tr tf RDSmax Low Side MOSFET RDS(ON)min td(on) td(off) Figure 11. MOSFETs Timing Diagram Soft Start The MOSFET delay, turn-on and turn-off times must be short enough to prevent cross conduction. If not, there will be cross conduction from the input through both MOSFETs to ground. Due to this fact, the following conditions must be true: t d(on) t d(on) high low ) t dead u t d(off) ) t dead u t d(off) low high ) tf low ) tf high The soft start time is set by a capacitor connected between the SS pin and ground. This function is used for controlling the output voltage slope and limiting start-up currents. The start-up sequence initiates when the Power On Ready (POR) internal signal rises to logic level high. That means the supply voltage and VP/EN voltage are over the set thresholds. The soft start capacitor is charged by a 22 mA current source. If POR is low, the SS pin is internally pulled to GND, which means that the IR3638 is in a shutdown state. The SS pin voltage (0 V to 2 V) controls the internal current source (eq. 12) Where tdead is the controller dead band time, td(on), tr, td(off) and tf are the MOSFET parameters. These parameters can be found in the datasheet for specific conditions. http://onsemi.com 8 IR3638 (64 mA to 0 mA) with a negative linear characteristic. This current source injects current into the resistor (25 kW) connected between the FB pin and the negative input of the error amplifier and into the external feedback resistor network. Voltage drop on these resistors is over 1.6 V, which is enough to force the error amplifier into a negative saturation state and to block switching. When the soft start pin reaches around 1.2 V (exact value depends on feedback and compensation network and on the soft start capacitor; a larger soft start capacitor and a lower compensation capacity decrease this level), the IC starts switching. The impact of the controlled current source decreases and the output voltage starts to rise. When the soft start capacitor voltage reaches 2 V, the output voltage is at nominal value. The soft start time must be at least 10 times longer than the time needed to charge the compensation network from the output of the error amplifier. If the soft start time is not long enough, the soft start sequence would be faster than the charging compensation network and the IC would start without slowly increasing the output voltage. The soft start capacitance can be calculated using Equation (13): C SS + 22 @ 10 −6 @ T SS (eq. 13) VCC = VC VIN VP_EN POR 3V 2V VSS 1V 0V VOUT 64 mA Internal IFB Vneg_error_amp >1.6 V VP/EN VP/EN VFB 0V Figure 12. Start-up Sequence Start to Pre-biased Output The IR3638 is able to start up into a pre-biased output capacitor. The low side MOSFET does not turn on before the output voltage is at set value. During this time, the energy is not discharged by the low side MOSFET (current flows through low side MOSFET body diode) until the soft start sequence ends. http://onsemi.com 9 IR3638 Vout 3V VSS 2V 1V VLDRV VHDRV Figure 13. Start-up to Pre-biased Output Short Circuit Protection comparator that compares FB voltage to 0.4 V. If FB voltage is below 0.4 V then IC goes to latch state and switch output drivers to off state. Latch state can be released by decrease VCC or VP/EN voltage below threshold. The output of convertor with IR3638 is protected against short circuit conditions. This protection is sensing output voltage through feedback divider on FB pin. On this pin is VCC or VP/EN Threshold Vout Output shorted VSS VLDRV VHDRV Figure 14. Short Circuit Protection (Start Up, Short, Latch, Latch Release and New Start-up) http://onsemi.com 10 IR3638 Compensation Circuit One zero of this LC filter is given by the output capacitance and output capacitor ESR. Its value can be calculated using the following equation: The IR3638 is a voltage mode buck converter with a transconductance error amplifier compensated by an external compensation network. Compensation is needed to achieve accurate output voltage regulation and fast transient response. The goal of the compensation circuit is to provide a loop gain function with the highest crossing frequency and adequate phase margin (minimally 45°). The transfer function of the power stage (the output LC filter) is a double pole system. The resonance frequency of this filter is expressed as follows: f PO + 1 2 @ p @ ǸL @ C OUT f Z0 + 1 2 @ p @ C OUT @ ESR (eq. 15) The next parameter that must be chosen is the zero crossover frequency f0. It can be chosen to be 1/10–1/5 of the switching frequency. These three parameters show the necessary type of compensation that can be selected from Table 6. (eq. 14) Table 6. COMPENSATION TYPES Zero Crossover Frequency Condition Compensation Type Typical Output Capacitor Type fP0 < fZ0< f0 < fSW/2 Type II (PI) Electrolytic, Tantalum fP0 < f0< fZ0 < fSW/2 Type III (PID) Method I Tantalum, Ceramic fP0 < f0 < fSW/2 < fZ0 Type III (PID) Method II Ceramic Compensation Type II (PI) Compensation Type III (PID) This compensation is suitable for low-cost electrolytic capacitors. The zero created by the capacitor’s ESR is a few kHz, and the zero crossover frequency is chosen to be 1/10 of the switching frequency. Components of the PI compensation (Figure 15) network can be specified by the following equations: Tantalum and ceramic capacitors have lower ESR than electrolytic capacitors, so the zero of the output LC filter goes to a higher frequency above the zero crossover frequency. This situation needs to be compensated by the PID compensation network that is shown in Figure 16. VOUT VOUT CC2 RFB1 R1 − + CFB1 RC1 CC1 CC2* R2 1 0.75 @ 2 @ p @ f P0 @ R C1 C C2 + 1 p @ R C1 @ f SW V OUT * V PńEN V PńEN − OTA + There are two methods to select the zeros and poles of the compensation network. The first one (method I) is usable for tantalum output capacitors, which have a higher ESR than ceramics, and its zeros and poles can be calculated as shown below: 2 @ p @ f 0 @ L @ V RAMP @ V OUT R C1 + ESR @ V IN @ V PńEN @ gm C C1 + VP/EN CC1 Figure 16. PID Compensation (Type III) Figure 15. PI compensation (Type II) R1 + RC1 OTA VP/EN R2 R1 *Optional (eq. 16) f Z1 + 0.75 @ f P0 f Z2 + f P0 @ R2 f P2 + f Z0 f P3 + VRAMP is the peak-to-peak voltage of the oscillator ramp, and gm is the transconductance error amplifier gain. Capacitor CC2 is optional. http://onsemi.com 11 f SW 2 (eq. 17) IR3638 The second one (method II) is for ceramic capacitors: f Z2 + f 0 @ f P2 + f 0 @ Ǹ Ǹ The remaining calculations are the same for both methods. 1 * sin q max 1 ) sin q max 1 ) sin q max 1 * sin q max 2 R C1 uu gm (eq. 18) f Z1 + 0.5 @ f Z2 f P3 + 0.5 @ f SW C C1 + 1 2 @ p @ f Z1 @ R C1 C C2 + 1 2 @ p @ f P3 @ R C1 (eq. 19) C FB1 + 2 @ p @ f 0 @ L @ V RAMP @ C OUT V IN @ R C1 R FB1 + 1 2 @ p @ C FB1 @ f P2 1 * R FB1 2 @ p @ C FB1 @ f Z2 V PńEN R2 + @ R1 V OUT * V PńEN R1 + To check the design of this compensation network, the following equation must be true: R1 @ R2 @ R FB1 1 (eq. 20) u gm R1 @ R FB1 ) R2 @ R FB1 ) R1 @ R2 If it is not true, then a higher value of RC1 must be selected. ORDERING INFORMATION Device IR3638DR2G Package Shipping† SOIC−14 (Pb−Free) 2500 / Tape & Reel †For information on tape and reel specifications, including part orientation and tape sizes, please refer to our Tape and Reel Packaging Specifications Brochure, BRD8011/D. http://onsemi.com 12 IR3638 PACKAGE DIMENSIONS SOIC−14 CASE 751A−03 ISSUE J NOTES: 1. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. 2. CONTROLLING DIMENSION: MILLIMETER. 3. DIMENSIONS A AND B DO NOT INCLUDE MOLD PROTRUSION. 4. MAXIMUM MOLD PROTRUSION 0.15 (0.006) PER SIDE. 5. DIMENSION D DOES NOT INCLUDE DAMBAR PROTRUSION. ALLOWABLE DAMBAR PROTRUSION SHALL BE 0.127 (0.005) TOTAL IN EXCESS OF THE D DIMENSION AT MAXIMUM MATERIAL CONDITION. −A− 14 8 −B− P 7 PL 0.25 (0.010) B M 7 1 G −T− 0.25 (0.010) M T B S A DIM A B C D F G J K M P R J M K D 14 PL F R X 45 _ C SEATING PLANE M S MILLIMETERS MIN MAX 8.55 8.75 3.80 4.00 1.35 1.75 0.35 0.49 0.40 1.25 1.27 BSC 0.19 0.25 0.10 0.25 0_ 7_ 5.80 6.20 0.25 0.50 INCHES MIN MAX 0.337 0.344 0.150 0.157 0.054 0.068 0.014 0.019 0.016 0.049 0.050 BSC 0.008 0.009 0.004 0.009 0_ 7_ 0.228 0.244 0.010 0.019 SOLDERING FOOTPRINT 7X 7.04 14X 1.52 1 14X 0.58 1.27 PITCH DIMENSIONS: MILLIMETERS ON Semiconductor and are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice to any products herein. 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