TPS5420 www.ti.com SLVS642A – APRIL 2006 – REVISED AUGUST 2006 2-A, WIDE INPUT RANGE, STEP-DOWN SWIFT™ CONVERTER FEATURES APPLICATIONS • • • • • • • • • • • • • • • Wide Input Voltage Range: 5.5 V to 36 V Up to 2-A Continuous (3-A Peak) Output Current High Efficiency up to 95% Enabled by 110-mΩ Integrated MOSFET Switch Wide Output Voltage Range: Adjustable Down to 1.22 V with 1.5% Initial Accuracy Internal Compensation Minimizes External Parts Count Fixed 500-kHz Switching Frequency for Small Filter Size Improved Line Regulation and Transient Response by Input Voltage Feed Forward System Protected by Overcurrent Limiting and Thermal Shutdown –40°C to 125°C Operating Junction Temperature Range Available in Small 8-Pin SOIC Package For SWIFT Documentation, Application Notes and Design Software, See the TI Website at www.ti.com/swift Consumer: Set-top Box, DVD, LCD Displays Industrial and Car Audio Power Supplies Battery Chargers, High Power LED Supply 12-V/24-V Distributed Power Systems DESCRIPTION As a member of the SWIFT™ family of DC/DC regulators, the TPS5420 is a high-output-current PWM converter that integrates a low resistance high side N-channel MOSFET. Included on the substrate with the listed features is a high performance voltage error amplifier that provides tight voltage regulation accuracy under transient conditions; an undervoltage-lockout circuit to prevent start-up until the input voltage reaches 5.5 V; an internally set slow-start circuit to limit inrush currents; and a voltage feed-forward circuit to improve the transient response. Using the ENA pin, shutdown supply current is reduced to 18 µA typically. Other features include an active high enable, overcurrent protection and thermal shutdown. To reduce design complexity and external component count, the TPS5420 feedback loop is internally compensated. The TPS5420 device is available in an easy to use 8-pin SOIC package. TI provides evaluation modules and the SWIFT™ Designer software tool to aid in quickly achieving high-performance power supply designs to meet aggressive equipment development cycles. SIMPLIFIED SCHEMATIC Efficiency vs Output Current VIN PH VIN 100 VOUT 95 TPS5420 90 BOOT NC ENA VSENSE GND 85 Efficiency − % NC VI = 12 V 80 75 70 65 60 55 50 0 0.5 1 1.5 2 I − Output Current− A O 2.5 3 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. SWIFT is a trademark of Texas Instruments. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2006, Texas Instruments Incorporated TPS5420 www.ti.com SLVS642A – APRIL 2006 – REVISED AUGUST 2006 These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. ORDERING INFORMATION TJ INPUT VOLTAGE –40°C to 125°C (1) (2) OUTPUT VOLTAGE 5.5 V to 36 V Adjustable to 1.22 V PACKAGE (1) SOIC PART NUMBER (D) (2) TPS5420D For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI web site at www.ti.com. The D package is available taped and reeled. Add an R suffix to the device type (i.e., TPS5420DR). ABSOLUTE MAXIMUM RATINGS over operating free-air temperature range (unless otherwise noted) (1) (2) VALUE VIN –0.3 to BOOT –0.3 to 50 –0.6 to 40 (3) PH (steady-state) VI Input voltage range UNIT 40 (3) EN –0.3 to 7 VSENSE –0.3 to 3 BOOT-PH V 10 PH (transient < 10 ns) –1.2 IO Source current PH Ilkg Leakage current PH 10 µA TJ Operating virtual junction temperature range –40 to 150 °C Tstg Storage temperature –65 to 150 °C (1) (2) (3) Internally Limited Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings only and functional operation of the device at these or any other conditions beyond those indicated under recommended operating conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. All voltage values are with respect to network ground terminal. Approaching the absolute maximum rating for the VIN pin may cause the voltage on the PH pin to exceed the absolute maximum rating. DISSIPATION RATINGS (1) (2) THERMAL IMPEDANCE JUNCTION-TO-AMBIENT PACKAGE 8 Pin D (1) (2) (3) (3) 75°C/W Maximum power dissipation may be limited by overcurrent protection. Power rating at a specific ambient temperature TA should be determined with a junction temperature of 125°C. This is the point where distortion starts to substantially increase. Thermal management of the final PCB should strive to keep the junction temperature at or below 125°C for best performance and long-term reliability. See Thermal Calculations in applications section of this data sheet for more information. Test board conditions: a. 3 in x 3 in, 2 layers, thickness: 0.062 inch. b. 2 oz. copper traces located on the top and bottom of the PCB. RECOMMENDED OPERATING CONDITIONS MIN 2 NOM MAX UNIT VI Input voltage range, VIN 5.5 36 V TJ Operating junction temperature –40 125 °C Submit Documentation Feedback TPS5420 www.ti.com SLVS642A – APRIL 2006 – REVISED AUGUST 2006 ELECTRICAL CHARACTERISTICS TJ = –40°C to 125°C, VIN = 5.5 V to 36 V (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT 3 4.4 mA 18 50 µA Start threshold voltage, UVLO 5.3 5.5 Hysteresis voltage, UVLO 330 SUPPLY VOLTAGE (VIN PIN) IQ Quiescent current VSENSE = 2 V, Not switching, PH pin open Shutdown, ENA = 0 V UNDERVOLTAGE LOCK OUT (UVLO) V mV VOLTAGE REFERENCE Voltage reference accuracy TJ = 25°C 1.202 1.221 1.239 IO = 0 A – 2 A 1.196 1.221 1.245 400 500 600 kHz 150 200 ns 87% 89% 1.3 V V OSCILLATOR Internally set free-running frequency Minimum controllable on time Maximum duty cycle ENABLE (ENA PIN) Start threshold voltage, ENA Stop threshold voltage, ENA 0.5 Hysteresis voltage, ENA V 450 Internal slow-start time (0 ~ 100%) 6.6 8 mV 10 ms CURRENT LIMIT Current limit Current limit hiccup time 3 4 5 A 13 16 20 ms 135 162 °C 14 °C THERMAL SHUTDOWN Thermal shutdown trip point Thermal shutdown hysteresis OUTPUT MOSFET rDS(on) High side power MOSFET switch VIN = 5.5 V 150 VIN = 10 V - 36 V 110 Submit Documentation Feedback 230 mΩ 3 TPS5420 www.ti.com SLVS642A – APRIL 2006 – REVISED AUGUST 2006 PIN ASSIGNMENTS D PACKAGE (TOP VIEW) BOOT 1 8 PH NC 2 7 VIN NC 3 6 GND VSENSE 4 5 ENA TERMINAL FUNCTIONS TERMINAL NAME NO. BOOT 1 NC 4 2, 3 DESCRIPTION Boost capacitor for the high-side FET gate driver. Connect 0.01 µF low ESR capacitor from BOOT pin to PH pin. Not connected internally. VSENSE 4 Feedback voltage for the regulator. Connect to output voltage divider. ENA 5 On/off control. Below 0.5 V, the device stops switching. Float the pin to enable. GND 6 Ground. VIN 7 Input supply voltage. Bypass VIN pin to GND pin close to device package with a high quality, low ESR ceramic capacitor. PH 8 Source of the high side power MOSFET. Connected to external inductor and diode. Submit Documentation Feedback TPS5420 www.ti.com SLVS642A – APRIL 2006 – REVISED AUGUST 2006 TYPICAL CHARACTERISTICS OSCILLATOR FREQUENCY vs JUNCTION TEMPERATURE 3.5 f − Oscillator Frequency − kHz 520 510 500 490 480 470 460 -50 180 VI = 12 V 170 Minimum Controllable On Time − ns IQ − Operating Quiescent Current − mA 530 3.25 3 2.75 2.5 -25 0 50 25 75 100 -50 125 -25 0 50 25 75 100 160 150 140 130 120 -50 125 -25 o TJ − Junction Temperature − oC TJ − Junction Temperature − C 0 25 50 75 100 Figure 2. Figure 3. VOLTAGE REFERENCE vs JUNCTION TEMPERATURE ON STATE RESISTANCE vs JUNCTION TEMPERATURE INTERNAL SLOW START TIME vs JUNCTION TEMPERATURE 180 9 VI = 12 V 1.22 1.215 tSS − Internal Slow Start Time − ms rDS(on) − On-State Resistance − mW 170 1.225 125 o TJ − Junction Temperature − C Figure 1. 1.23 160 150 140 130 120 110 100 8.5 8 7.5 90 1.21 80 -25 0 50 25 75 100 125 -50 -25 0 50 25 100 7 125 -25 TJ − Junction Temperature − oC Figure 4. 0 25 50 75 100 125 o TJ − Junction Temperature − C Figure 5. Figure 6. MINIMUM CONTROLLABLE DUTY RATIO vs JUNCTION TEMPERATURE SHUTDOWN QUIESCENT CURRENT vs INPUT VOLTAGE 25 75 -50 o TJ − Junction Temperature − C 8 ENA = 0 V o TJ = 125 C 20 Minimum Duty Ratio − % -50 ISD − Shutdown Current − mA Vref − Voltage Reference − V MINIMUM CONTROLLABLE ON TIME vs JUNCTION TEMPERATURE OPERATING QUIESCENT CURRENT vs JUNCTION TEMPERATURE TJ = 27oC 15 10 7.5 7.5 7.25 TJ = -40oC 5 7 0 5 10 15 20 25 VI − Input Voltage − V 30 35 40 -50 -25 0 25 50 75 100 125 TJ − Junction Temperature − oC Figure 7. Submit Documentation Feedback Figure 8. 5 TPS5420 www.ti.com SLVS642A – APRIL 2006 – REVISED AUGUST 2006 APPLICATION INFORMATION FUNCTIONAL BLOCK DIAGRAM VIN VIN 1.221 V Bandgap Reference UVLO VREF SHDN Slow Start Boot Regulator BOOT HICCUP 5 µA ENA ENABLE SHDN SHDN VSENSE Z1 Thermal Protection NC SHDN VIN Ramp Generator NC GND Z2 Feed Forward Gain = 25 SHDN SHDN Error Amplifier SHDN PWM Comparator HICCUP Overcurrent Protection Oscillator SHDN Gate Drive Control Gate Driver SHDN BOOT PH VOUT DETAILED DESCRIPTION Oscillator Frequency The internal free running oscillator sets the PWM switching frequency at 500 kHz. The 500 kHz switching frequency allows less output inductance for the same output ripple requirement resulting in a smaller output inductor. Voltage Reference The voltage reference system produces a precision reference signal by scaling the output of a temperature stable bandgap circuit. The bandgap and scaling circuits are trimmed during production testing to an output of 1.221 V at room temperature. Enable (ENA) and Internal Slow Start The ENA pin provides electrical on/off control of the regulator. Once the ENA pin voltage exceeds the threshold voltage, the regulator starts operation and the internal slow start begins to ramp. If the ENA pin voltage is pulled below the threshold voltage, the regulator stops switching and the internal slow start resets. Connecting the pin to ground or to any voltage less than 0.5 V disables the regulator and activate the shutdown mode. The quiescent current of the TPS5420 in shutdown mode is typically 18 µA. The ENA pin has an internal pullup current source, allowing the user to float the ENA pin. If an application requires controlling the ENA pin, use open drain or open collector output logic to interface with the pin. To limit the start-up inrush current, an internal slow start circuit is used to ramp up the reference voltage from 0 V to its final value linearly. The internal slow start time is 8 ms typically. 6 Submit Documentation Feedback TPS5420 www.ti.com SLVS642A – APRIL 2006 – REVISED AUGUST 2006 APPLICATION INFORMATION (continued) Undervoltage Lockout (UVLO) The TPS5420 incorporates an undervoltage lockout circuit to keep the device disabled when VIN (the input voltage) is below the UVLO start voltage threshold. During power up, internal circuits are held inactive until VIN exceeds the UVLO start threshold voltage. Once the UVLO start threshold voltage is reached, device start-up begins. The device operates until VIN falls below the UVLO stop threshold voltage. The typical hysteresis in the UVLO comparator is 330 mV. Boost Capacitor (BOOT) Connect a 0.01 µF low-ESR ceramic capacitor between the BOOT pin and PH pin. This capacitor provides the gate drive voltage for the high-side MOSFET. X7R or X5R grade dielectrics are recommended due to their stable values over temperature. Output Feedback (VSENSE) The output voltage of the regulator is set by feeding back the center point voltage of an external resistor divider network to the VSENSE pin. In steady-state operation, the VSENSE pin voltage should be equal to the voltage reference 1.221 V. Internal Compensation The TPS5420 implements internal compensation to simplify the regulator design. Since the TPS5420 uses voltage mode control, a type 3 compensation network has been designed on chip to provide a high crossover frequency and a high phase margin for good stability. See the Internal Compensation Network in the applications section for more details. Voltage Feed Forward The internal voltage feed forward provides a constant DC power stage gain despite any variations with the input voltage. This greatly simplifies the stability analysis and improves the transient response. Voltage feed forward varies the peak ramp voltage inversely with the input voltage so that the modulator and power stage gain are constant at the feed forward gain, i.e. VIN Feed Forward Gain = Ramppk-pk (1) The typical feed forward gain of TPS5420 is 25. Pulse-Width-Modulation (PWM) Control The regulator employs a fixed frequency pulse-width-modulator (PWM) control method. First, the feedback voltage (VSENSE pin voltage) is compared to the constant voltage reference by the high gain error amplifier and compensation network to produce a error voltage. Then, the error voltage is compared to the ramp voltage by the PWM comparator. In this way, the error voltage magnitude is converted to a pulse width which is the duty cycle. Finally, the PWM output is fed into the gate drive circuit to control the on-time of the high-side MOSFET. Overcurrent Protection Overcurrent protection is implemented by sensing the drain-to-source voltage across the high-side MOSFET. The drain to source voltage is then compared to a voltage level representing the overcurrent threshold limit. If the drain-to-source voltage exceeds the overcurrent threshold limit, the overcurrent indicator is set true. The system ignores the overcurrent indicator for the leading edge blanking time at the beginning of each cycle to avoid any turn-on noise glitches. Once overcurrent indicator is set true, overcurrent protection is triggered. The high-side MOSFET is turned off for the rest of the cycle after a propagation delay. The overcurrent protection scheme is called cycle-by-cycle current limiting. Submit Documentation Feedback 7 TPS5420 www.ti.com SLVS642A – APRIL 2006 – REVISED AUGUST 2006 APPLICATION INFORMATION (continued) If the sensed current continues increasing even with the cycle-by-cycle current limiting which may happen during short-circuit or under other circumstances, the hiccup mode overcurrent protection is triggered instead of the cycle-by-cycle current limiting. During the hiccup mode overcurrent protection, the voltage reference is grounded and the high-side MOSFET is turned off for the hiccup time. Once the hiccup time is complete, the regulator restarts. Thermal Shutdown The TPS5420 protects itself from overheating with an internal thermal shutdown circuit. If the junction temperature exceeds the thermal shutdown trip point, the voltage reference is grounded and the high-side MOSFET is turned off. The part is restarted under control of the slow start circuit automatically when the junction temperature drops 14°C below the thermal shutdown trip point. PCB Layout Connect a low ESR ceramic bypass capacitor to the VIN pin. Care should be taken to minimize the loop area formed by the bypass capacitor connections, the VIN pin, and the TPS5420 ground pin. The best way to do this is to extend the top side ground area from under the device adjacent to the VIN trace, and place the bypass capacitor as close as possible to the VIN pin. The minimum recommended bypass capacitance is 10-µF ceramic with a X5R or X7R dielectric. There should be a ground area on the top layer directly underneath the IC to connect the GND pin of the device and the anode of the catch diode. The GND pin should be tied to the PCB ground by connecting it to the ground area under the device as shown in Figure 9. The PH pin should be routed to the output inductor, catch diode and boot capacitor. Since the PH connection is the switching node, the inductor should be located close to the PH pin, and the area of the PCB conductor minimized to prevent excessive capacitive coupling. The catch diode should also be placed close to the device to minimize the output current loop area. Connect the boot capacitor between the phase node and the BOOT pin as shown. Keep the boot capacitor close to the IC and minimize the conductor trace lengths. The component placements and connections shown work well, but other connection routings may also be effective. Connect the output filter capacitor(s) as shown between the VOUT trace and GND. It is important to keep the loop formed by the PH pin, Lout, Cout and GND as small as is practical. Connect the VOUT trace to the VSENSE pin using the resistor divider network to set the output voltage. Do not route this trace too close to the PH trace. Due to the size of the IC package and the device pinout, the trace may need to be routed under the output capacitor. The routing may be done on an alternate layer if a trace under the output capacitor is not desired. If the grounding scheme shown is used via a connection to a different layer to route to the ENA pin. 8 Submit Documentation Feedback TPS5420 www.ti.com SLVS642A – APRIL 2006 – REVISED AUGUST 2006 APPLICATION INFORMATION (continued) PH BOOT CAPACITOR OUTPUT INDUCTOR RESISTOR DIVIDER VOUT BOOT PH NC VIN NC GND VSENSE ENA OUTPUT FILTER CAPACITOR Route feedback trace under the output filter capacitor or on the other layer. CATCH DIODE INPUT INPUT BULK BYPASS CAPACITOR FILTER Vin TOPSIDE GROUND AREA VIA to Ground Plane Signal VIA Figure 9. Design Layout Submit Documentation Feedback 9 TPS5420 www.ti.com SLVS642A – APRIL 2006 – REVISED AUGUST 2006 0.050 APPLICATION INFORMATION (continued) 0.026 0.220 0.080 All dimensions in inches Figure 10. TPS5420 Land Pattern Application Circuits Figure 11 shows the schematic for a typical TPS5420 application. The TPS5420 can provide up to 2-A output current at a nominal output voltage of 5 V. U1 TPS5420D 10 V - 35 V 7 VIN ENA C1 4.7 mF C4 4.7 mF VIN BOOT 5 ENA 2 NC 1 C2 0.01 mF TP5 PH 8 3 NC 6 GND VSNS 4 L1 33 mH 5V VOUT D1 B340A + C3 100 mF (See Note A) R1 10 kW R2 3.24 kW A. C3 = Tantalum AVX TPSD107M010R0080 Figure 11. Application Circuit, 10-V — 35 V to 5-V 10 Submit Documentation Feedback TPS5420 www.ti.com SLVS642A – APRIL 2006 – REVISED AUGUST 2006 APPLICATION INFORMATION (continued) Design Procedure The following design procedure can be used to select component values for the TPS5420. Alternately, the SWIFT™ Designer Software may be used to generate a complete design. The SWIFT™ Designer Software uses an iterative design procedure and accesses a comprehensive database of components when generating a design. This section presents a simplified discussion of the design process. To • • • • • • begin the design process, a few parameters must be determined. The designer must know the following: Input voltage range Output voltage Input ripple voltage Output ripple voltage Output current rating Operating frequency Design Parameters For this design example, use the following as the input parameters: (1) DESIGN PARAMETER (1) EXAMPLE VALUE Input voltage range 10 V to 36 V Output voltage 5V Input ripple voltage 300 mV Output ripple voltage 30 mV Output current rating 2A Operating frequency 500 kHz As an additional constraint, the design is set up to be small size and low component height. Switching Frequency The switching frequency for the TPS5420 is internally set to 500 kHz. It is not possible to adjust the switching frequency. Input Capacitors The TPS5420 requires an input decoupling capacitor and, depending on the application, a bulk input capacitor. The recommended value for the decoupling capacitor is 10 µF. A high quality ceramic type X5R or X7R is required. For some applications, a smaller value decoupling capacitor may be used, if the input voltage and current ripple ratings are not exceeded. The voltage rating must be greater than the maximum input voltage, including ripple. For this design, two 4.7 µF capacitors, C1 and C4 are used to allow for smaller 1812 case size to be used while maintaining a 50 V rating. This input ripple voltage can be approximated by Equation 2 : IOUT(MAX) x 0.25 + IOUT(MAX) x ESRMAX DVIN = CBULK x ƒSW ( ) (2) Where IOUT(MAX) is the maximum load current, fSW is the switching frequency, CI is the input capacitor value and ESRMAX is the maximum series resistance of the input capacitor. The maximum RMS ripple current also needs to be checked. For worst case conditions, this is approximated by Equation 3: Submit Documentation Feedback 11 TPS5420 www.ti.com SLVS642A – APRIL 2006 – REVISED AUGUST 2006 I I CIN + OUT(MAX) 2 (3) In this example, the calculated input ripple voltage is 118 mV, and the RMS maximum voltage across the input capacitors would be VIN max plus delta VIN/2. capacitors are rated for 50 V, and the ripple current capacity for each is 3 A at 500 The actual measured input ripple voltage may be larger than the calculated value of the input voltage source and parasitics associated with the layout. ripple current is 1.0 A. The The chosen input decoupling kHz, providing ample margin. due to the output impedance CAUTION: The maximum ratings for voltage and current are not to be exceeded under any circumstance. Additionally, some bulk capacitance may be needed, especially if the TPS5420 circuit is not located within approximately 2 inches from the input voltage source. The value for this capacitor is not critical but it should be rated to handle the maximum input voltage including ripple voltage and should filter the output so that input ripple voltage is acceptable. Output Filter Components Two components need to be selected for the output filter, L1 and C2. Since the TPS5420 is an internally compensated device, a limited range of filter component types and values can be supported. Inductor Selection To calculate the minimum value of the output inductor, use Equation 4: LMIN = ( VOUT x VIN(MAX) - VOUT ) VIN(max) x KIND x IOUT x FSW x 0.8 (4) KIND is a coefficient that represents the amount of inductor ripple current relative to the maximum output current. Three things need to be considered when determining the amount of ripple current in the inductor: the peak to peak ripple current affects the output ripple voltage amplitude, the ripple current affects the peak switch current, and the amount of ripple current determines at what point the circuit becomes discontinuous. For designs using the TPS5420, KIND of 0.2 to 0.3 yields good results. Low output ripple voltages is obtained when paired with the proper output capacitor, the peak switch current is below the current limit set point, and low load currents can be sourced before discontinuous operation. For this design example, use KIND = 0.2, and the minimum inductor value is 31 µH. The next highest standard value used in this design is 33 µH. For the output filter inductor, it is important that the RMS current and saturation current ratings not be exceeded. The RMS inductor current can be found from Equation 5: I L(RMS) + Ǹ I2 1 ) OUT(MAX) 12 ǒ V V OUT ǒVIN(MAX) * VOUTǓ L IN(MAX) OUT F SW 0.8 Ǔ 2 (5) and the peak inductor current can be determined using Equation 6: ( ) VOUT x VIN(MAX) - VOUT IL(PK) = IOUT(MAX) + 1.6 x VIN(MAX) x LOUT x FSW (6) For this design, the RMS inductor current is 2.002 A, and the peak inductor current is 2.16 A. The chosen inductor is a Coilcraft MSS1260-333 type. The nominal inductance is 33 µH. It has a saturation current rating of 2.2 A and a RMS current rating of 2.7 A, which meets the requirements. Inductor values for use with the TPS5420 are in the range of 10 µH to 100 µH. 12 Submit Documentation Feedback TPS5420 www.ti.com SLVS642A – APRIL 2006 – REVISED AUGUST 2006 Capacitor Selection The important design factors for the output capacitor are dc voltage rating, ripple current rating, and equivalent series resistance (ESR). The dc voltage and ripple current ratings cannot be exceeded. The ESR is important because along with the inductor ripple current it determines the amount of output ripple voltage. The actual value of the output capacitor is not critical, but some practical limits do exist. Consider the relationship between the desired closed loop crossover frequency of the design and LC corner frequency of the output filter. Due to the design of the internal compensation, it is recommended to keep the closed loop crossover frequency in the range 3 kHz to 30 kHz as this frequency range has adequate phase boost to allow for stable operation. For this design example, the intended closed loop crossover frequency is between 2590 Hz and 24 kHz, and below the ESR zero of the output capacitor. Under these conditions, the closed loop crossover frequency is related to the LC corner frequency as: f CO + f LC 2 85 VOUT (7) and the desired output capacitor value for the output filter to: 1 C OUT + 3357 L OUT f CO V OUT (8) For a desired crossover of 18 kHz and a 33-µH inductor, the calculated value for the output capacitor is 100 µF. The capacitor type should be chosen so that the ESR zero is above the loop crossover. The maximum ESR is: 1 ESR MAX + 2p C OUT f CO (9) The maximum ESR of the output capacitor also determines the amount of output ripple as specified in the initial design parameters. The output ripple voltage is the inductor ripple current times the ESR of the output filter. Check that the maximum specified ESR as listed in the capacitor data sheet results in an acceptable output ripple voltage: VPP(MAX) = ( ) ESRMAX x VOUT x VIN(MAX) - VOUT NC x VIN(MAX) x LOUT x FSW x 0.8 (10) Where: ∆ VPP is the desired peak-to-peak output ripple. NC is the number of parallel output capacitors. FSW is the switching frequency. The minimum ESR of the output capacitor should also be considered. For a good phase margin, if the ESR is zero when the ESR is at its minimum, it should not be above the internal compensation poles at 24 kHz and 54 kHz. The selected output capacitor must also be rated for a voltage greater than the desired output voltage plus one half the ripple voltage. Any derating amount must also be included. The maximum RMS ripple current in the output capacitor is given by Equation 11: ICOUT(RMS) 1 = Ö12 x [ ( VOUT x VIN(MAX) - VOUT ) VIN(MAX) x LOUT - FSW x 0.8 x NC ] (11) Where: NC is the number of output capacitors in parallel. FSW is the switching frequency. Submit Documentation Feedback 13 TPS5420 www.ti.com SLVS642A – APRIL 2006 – REVISED AUGUST 2006 For this design example, a single 100-µF output capacitor is chosen for C3. The calculated RMS ripple current is 143 mA and the maximum ESR required is 88 mΩ. A capacitor that meets these requirements is a AVX TPSD107M010R0080, rated at 10 V with a maximum ESR of 80 mΩ and a ripple current rating of 1.369 A. This capacitor results in a peak-to-peak output ripple of 26 mV using equation 10. An additional small 0.1-µF ceramic bypass capacitor may also used, but is not included in this design. Other capacitor types can be used with the TPS5420, depending on the needs of the application. Output Voltage Setpoint The output voltage of the TPS5420 is set by a resistor divider (R1 and R2) from the output to the VSENSE pin. Calculate the R2 resistor value for the output voltage of 5 V using Equation 12: R1 1.221 R2 + V * 1.221 OUT (12) For any TPS5420 design, start with an R1 value of 10 kΩ. R2 is then 3.24 kΩ. Boot Capacitor The boot capacitor should be 0.01 µF. Catch Diode The TPS5420 is designed to operate using an external catch diode between PH and GND. The selected diode must meet the absolute maximum ratings for the application: Reverse voltage must be higher than the maximum voltage at the PH pin, which is VINMAX + 0.5 V. Peak current must be greater than IOUTMAX plus on half the peak-to-peak inductor current. Forward voltage drop should be small for higher efficiencies. It is important to note that the catch diode conduction time is typically longer than the high-side FET on time; therefore, the diode parameters improve the overall efficiency. Additionally, check that the device chosen is capable of dissipating the power losses. For this design, a Diodes, Inc. B340A is chosen, with a reverse voltage of 40 V, forward current of 3 A, and a forward voltage drop of 0.5 V. Additional Circuits Figure 12 shows an application circuit using a wide input voltage range. The design parameters are similar to those given for the design example, with a larger value output inductor and a lower closed loop crossover frequency. 10 V - 21 V VIN ENA C1 10 mF U1 TPS5420D 7 5 2 3 6 VIN BOOT C2 0.01 mF TP5 L1 27 mH 5V 1 VOUT ENA PH NC NC VSNS 8 4 D1 B340A + C3 100 mF (See Note A) R1 10 kW GND R2 3.24 kW A. C3 = Tantalum AVX TPSD107M010R0080 Figure 12. 10-V — 21-V Input to 5-V Output Application Circuit 14 Submit Documentation Feedback TPS5420 www.ti.com SLVS642A – APRIL 2006 – REVISED AUGUST 2006 ADVANCED INFORMATION Output Voltage Limitations Due to the internal design of the TPS5420, there are both upper and lower output voltage limits for any given input voltage. The upper limit of the output voltage set point is constrained by the maximum duty cycle of 87% and is given by: V OUTMAX + 0.87 ǒǒVINMIN * I OMAX Ǔ Ǔ ǒ 0.230 ) VD * I OMAX Ǔ RL * VD (13) Where: VINMIN = minimum input voltage IOMAX = maximum load current VD = catch diode forward voltage. RL= output inductor series resistance. This equation assumes maximum on resistance for the internal high side FET. The lower limit is constrained by the minimum controllable on time which may be as high as 200 ns. The approximate minimum output voltage for a given input voltage and minimum load current is given by: V OUTMIN + 0.12 ǒǒVINMAX * I OMIN Ǔ Ǔ ǒ 0.110 ) VD * I OMIN Ǔ RL * VD (14) Where: VINMAX = maximum input voltage IOMIN = minimum load current VD = catch diode forward voltage. RL= output inductor series resistance. This equation assumes nominal on resistance for the high side FET and accounts for worst case variation of operating frequency set point. Any design operating near the operational limits of the device should be checked to assure proper functionality. Internal Compensation Network The design equations given in the example circuit can be used to generate circuits using the TPS5420. These designs are based on certain assumptions, and always select output capacitors within a limited range of ESR values. If a different capacitor type is desired, it may be possible to fit one to the internal compensation of the TPS5420. Equation 15 gives the nominal frequency response of the internal voltage-mode type III compensation network: s s 1) 1) 2p Fz1 2p Fz2 H(s) + s s s s 1) 1) 1) 2p Fp0 2p Fp1 2p Fp2 2p Fp3 (15) ǒ ǒ Ǔ ǒ Ǔ ǒ Ǔ ǒ Ǔ Ǔ ǒ Ǔ Where Fp0 = 2165 Hz, Fz1 = 2170 Hz, Fz2 = 2590 Hz Fp1 = 24 kHz, Fp2 = 54 kHz, Fp3 = 440 kHz Fp3 represents the non-ideal parasitics effect. Using this information along with the desired output voltage, feed forward gain and output filter characteristics, the closed loop transfer function can be derived. Submit Documentation Feedback 15 TPS5420 www.ti.com SLVS642A – APRIL 2006 – REVISED AUGUST 2006 Thermal Calculations The following formulas show how to estimate the device power dissipation under continuous conduction mode operations. They should not be used if the device is working at light loads in the discontinuous conduction mode. Conduction Loss: Pcon = IOUT2 x rDS(on) x VOUT / VIN Switching Loss: Psw = VIN x IOUT x 0.01 Quiescent Current Loss: Pq = VIN x 0.01 Total Loss: Ptot = Pcon + Psw + Pq Given TA => Estimated Junction Temperature: TJ = TA + Rth x Ptot Given TJMAX = 125°C => Estimated Maximum Ambient Temperature: TAMAX = TJMAX – Rth x Ptot PERFORMANCE GRAPHS The performance graphs in Figure 13 - Figure 19 are applicable to the circuit in Figure 11. TA = 25°C. unless otherwise specified. 100 VI = 10.8 V 0.2 0.2 Output Regulation - % 90 85 VI = 18 V VI = 19.8 V 80 Output Regulation - % IO = 2 A VI = 15 V Efficiency - % 0.3 VI = 12 V 95 0.1 0 -0.1 -0.2 0 0.5 1 1.5 2 IO - Output Current - A 2.5 Figure 13. Efficiency vs. Output Current VIN = 100 mV/Div (AC Coupled) PH = 5 V/Div t - Time - 1 ms / Div Figure 16. Input Voltage Ripple and PH Node, IO = 3 A 3 0.1 IO = 0 A 0 -0.1 IO = 1 A -0.2 -0.3 75 16 0.3 -0.3 0 0.5 1 1.5 2 2.5 3 IO - Output Current - A Figure 14. Output Regulation % vs. Output Current VOUT = 20 mV/Div (AC Coupled) PH = 5 V/Div t - Time - 1 ms / Div Figure 17. Output Voltage Ripple and PH Node, IO = 3 A Submit Documentation Feedback 0 0.5 1 1.5 2 2.5 3 VI - Input Voltage - V Figure 15. Input Regulation % vs. Input Voltage VOUT = 50 mV/Div (AC Coupled) IOUT = 500 mA/Div t - Time = 200 μs/Div Figure 18. Transient Response, Io Step 0.5 to 1.5 A TPS5420 www.ti.com SLVS642A – APRIL 2006 – REVISED AUGUST 2006 VIN = 10 V/Div ENA = 2 V/Div VOUT = 2 V/Div VOUT = 2 V/Div t - Time = 5 ms/Div Figure 19. Startup Waveform, VIN and VOUT t - Time = 5 ms/Div Figure 20. Startup Waveform, ENA and VOUT Submit Documentation Feedback 17 PACKAGE OPTION ADDENDUM www.ti.com 12-Sep-2006 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Drawing Pins Package Eco Plan (2) Qty TPS5420D ACTIVE SOIC D 8 75 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR TPS5420DG4 ACTIVE SOIC D 8 75 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR TPS5420DR ACTIVE SOIC D 8 2500 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR TPS5420DRG4 ACTIVE SOIC D 8 2500 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR Lead/Ball Finish MSL Peak Temp (3) (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. 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