HR1000 - Monolithic Power Systems

AN054
Resonant Converter Controller HR1000
The Future of Analog IC Technology
Application Note
for an LLC Resonant Converter Using
Resonant Controller HR1000
Prepared by Jeff Jin
Sept. 01, 2011
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AN054―RESONANT CONVERTER CONTROLLER HR1000
ABSTRACT
This application note presents design guidelines for an LLC resonant converter using resonant controller
HR1000 in applications such as the one shown in Figure 1. The first part introduces the HR1000’s
features. This section is followed by an introduction to the LLC resonant converter and a time-domain and
frequency-domain analysis of its operating principle. The third section discusses a step-by-step design
methodology for an LLC resonant converter using the HR1000. The last section discusses a methodology
to verify a design, using a 90W adapter prototype as an example.
Input
85-265VAC
BO
Output
CS
SS
TIMER
CT
Fset
Burst
CS
BO
LATCH
1
2
16
15
BST
HG
SW
3
4
5
14
13
N.C.
HR1000 12
VCC
6
7
11
10
GND
8
9
LG
PFC
Interface to PFC
Figure 1: LLC Resonant Converter Using Resonant Controller—HR1000
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AN054―RESONANT CONVERTER CONTROLLER HR1000
INDEX
1. AN INTRODUCTION TO THE HR1000 ......................................................................................................... 5
2. AN INTRODUCTION TO THE LLC RESONANT CONVERTER .................................................................. 5
2.1 LLC Resonant Converter Introduction ....................................................................................... 5
2.2 Key Operating Principle............................................................................................................. 7
A. Operating Waveform at fm<fs<fr ....................................................................................................... 8
B. Operating Principle at fs=fr............................................................................................................... 8
C. Operating Waveform at fs>fr ............................................................................................................ 9
2.3 Frequency Domain Analysis.....................................................................................................11
2.4 Performance Analysis of LLC resonant converter.................................................................... 16
A. Loss analysis on the operation at resonant frequency ................................................................. 16
B. Performance Analysis during Holdup............................................................................................ 19
3. DESIGN PROCEDURE ............................................................................................................................... 20
3.1 Predetermined Input and Output Specifications....................................................................... 22
3.2 Determining the Transformer Turns Ratio ............................................................................... 22
3.3 Design of Primary-Side Inductor, Lm ........................................................................................ 22
3.4 Determining Lr, Cr.................................................................................................................... 23
3.5 Transformer Design ................................................................................................................ 25
A. Transformer Core Selection .......................................................................................................... 25
B. Primary and Secondary Winding Turns ........................................................................................ 25
C. Wire Size ....................................................................................................................................... 25
D. Air Gap .......................................................................................................................................... 26
3.6 Inductor Design ....................................................................................................................... 27
A. Inductor Core Selection................................................................................................................. 27
3.7 Parameter Design ................................................................................................................... 28
A. FSET, CT, SS................................................................................................................................ 28
B. Burst Mode ................................................................................................................................. 30
C. Current Sensing Methods ............................................................................................................. 31
D. Input Voltage Sensing................................................................................................................... 33
F. Low-Side Gate Driver .................................................................................................................... 34
4. EXAMPLE DESIGN...................................................................................................................................... 34
4.1 Specification............................................................................................................................ 35
4.2 Schematic ............................................................................................................................... 35
4.3 Design Spreadsheet Using MPS’s Design Toolⅲ ...................................................................... 36
A. Input and Output Specifications ................................................................................................. 36
B. Transformer Turns Ratio ............................................................................................................... 36
C. Transformer Primary Inductance .................................................................................................. 36
D. h, Lr, and Cr .................................................................................................................................. 36
4.4 Transformer Core and Winding Turns...................................................................................... 37
4.5 Resonant Inductor Core and Winding Turns............................................................................ 38
4.6 Control Circuit Design ............................................................................................................. 38
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AN054―RESONANT CONVERTER CONTROLLER HR1000
4.7 Transformer and Inductor Design ........................................................................................... 41
4.8 Evaluation Board for 90W Slim Adapter.................................................................................. 42
5. EXPERIMENTAL VERIFICATION ............................................................................................................... 42
5.1 Efficiency................................................................................................................................. 42
5.2 Startup Operation .................................................................................................................... 44
5.3 Steady-State Operation ........................................................................................................... 46
5.4 No-Load Operation.................................................................................................................. 47
5.5 Transient ................................................................................................................................. 48
5.6 Short-Circuit Protection ........................................................................................................... 48
5.7 Over-Load Protection .............................................................................................................. 49
6. REFERENCES............................................................................................................................................. 49
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AN054―RESONANT CONVERTER CONTROLLER HR1000
1. AN INTRODUCTION TO THE HR1000
The HR1000 is a controller designed specifically for the resonant half-bridge topology. It has two output
channels of complementary driving signals that run at 50% duty cycle. An integrated bootstrap diode
simplifies the external driving circuit for the high-side switch. A fixed dead-time inserted between the two
complementary gate drivers guarantees soft-switching during the transient and enables high-frequency
operation. Modulating the operating frequency regulates the output voltage. A programmable oscillator
sets both the maximum and minimum switching frequency.
The IC initially works at the programmed maximum switching frequency that gradually falls until the control
loop takes over in order to prevent the excessive inrush current. The IC can be forced to enter a controlled
burst-mode operation at light-load to minimize the power consumption and tighten output regulation.
Protections features—such as latched shutdown or auto-recovery for over-current or over-voltage
conditions or brown-outs—contribute to a safer converter design without increasing circuitry complexity.
This paper provides practical design guidelines for an LLC resonant converter using the HR1000, and
includes step-by-step design guidelines that include a resonant parameters selection, transformer
design, resonant inductor design and control parameters design.
2. AN INTRODUCTION TO THE LLC RESONANT CONVERTER
2.1 LLC Resonant Converter Introduction
Conventional PWM converters regulate the output voltage by adjusting the switching-cycle pulse width.
The maximum duty cycle and main component parameter must be designed for the minimum input
voltage condition so that the duty cycle gradually decreases as the input voltage increases. However,
this causes the convert efficiency to drop substantially at normal input and at high line. The problem
becomes serious when an application requires optimal converter efficiency at a high input voltage with a
wide voltage range. By comparison, the LLC resonant converter can realize a wide input voltage range
without sacrificing efficiency.
Another of the LLC resonant converter’s merits is the capacity to achieve zero-voltage switching (ZVS)
for primary side switches and zero-current switching (ZCS) for the secondary side rectifier. LLC
converters do not suffer from reverse recovery issues and severe switching noise when compared
against conventional converters, which generally have serious reverse recovery issues induced by hard
switching. Soft-switching greatly reduces switching losses, allowing for LLC use in high-frequency
applications. Operating at higher frequencies reduces the size of passive component, such as
transformer and inductor, considerably and permits higher power densities.
Figure 1 shows the half-bridge LLC topology. The circuit can be divided into the following function blocks:
the square-wave generator, the series resonant tank, the transformer, the output rectifier circuit, and the
output filter. S1 and S2 implement the square wave generator, which commutates at a 50% duty cycle.
The series resonant tank is composed of a series resonant inductor, Lr, a series resonant capacitor, Cr,
and the Lm formed by the magnetizing inductance of transformer T1. The series resonant inductor can
be an external component or the leakage inductance of T1. The rectifier circuit—which includes D1 and
D2—converts the resonant current into a unidirectional current. The output filter, Cf, modulates the
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AN054―RESONANT CONVERTER CONTROLLER HR1000
high-frequency ripple current.
A conventional series resonant converter (SRC)—which features an infinite magnetizing inductor, Lm and
can only work above the resonant frequency to achieve the ZVS condition: The SRC DC gain is always
<1. However an LLC converter that substitutes Lm, with a shunt inductor can not only work above the
Lr·Cr resonant frequency (fs), but also below fs and above the Cr·( Lr+Lm) resonant frequency (fm).
The resonant frequency, fs, is defined as:
1
2π Lr ⋅ Cr
(1)
1
2π ( Lr + Lm ) ⋅ Cr
(2)
fs =
The resonant frequency, fm, is defined as:
fm =
To reiterate, the LLC-SRC can operate not only in the range of f>fs, but also in the range of fm<f<fs.
S1
Lr
Cr
T
Vin
D1
S2
Co
R
Lm
D2
Figure 2: Half-Bridge LLC Converter
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AN054―RESONANT CONVERTER CONTROLLER HR1000
2.2 Key Operating Principle
S1
Lr
iD1
Cr
T
Vin
S2
D1
im
ir
Vo
Lm
D2
(a) Stage 1 [t0, t1]
S1
Lr
iD1
Cr
T
Vin
ir
S2
D1
im
Vo
Lm
D2
(b) Stage 2 [t1, t2]
S1
Lr
iD1
Cr
T
Vin
S2
ir
im
D1
Vo
Lm
D2
(c) Stage 3 [t2, t3]
Figure 3: Equivalent main circuit at fm<f<fs
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AN054―RESONANT CONVERTER CONTROLLER HR1000
S1
S1
S2
S2
ir
ir
im
im
Vds1
tdead
i D1
irec
iD2
Vc
t0
t1
t2 t3 t4
t5 t6
Figure 4: Operating waveform at fm<fs<fr
A. Operating Waveform at fm<fs<fr
The equivalent circuit and operating waveform are illustrated in Figure 3 and Figure 4, respectively. The
switch turns off at time t0 and the resonant current firstly discharges the parasitic capacitor and then
flows through the body diode of S1. The output rectifier diode, D1, continues to deliver energy to the load.
Cr and Lr resonate because the voltage across Lm is clamped at the reflected output voltage. The
magnetizing current, im, increases linearly during this period.
At t1, S1 turns on due to the ZVS condition. The current drops to 0A and before reversing and flowing
through the switch, S1. The resonant current waveform increases in amplitude from being clamped by
the voltage difference between Vin and the reflected output voltage. The load current is proportional to
the difference between the resonant current, ir, and im. At t2, the load current drops to zero due to the
resonant current, ir, equaling im, and causes D1 to turn off. Since the switching period is longer than the
Lr·Cr resonant period, S1 continues to conduct until t3. In the fm<fs<fr operating range, im implements the
primary-side ZVS condition—which is independent of the load current and input voltage, thus extending
the ZVS range beyond that of most of soft-switching topologies. Also, the secondary-side diode turns off
after the current drops to 0A, thus eliminating the reverse-recovery problem, and diodes operate under
the ZCS condition.
B. Operating Principle at fs=fr
When the switching frequency equals the Lr·Cr resonant frequency, the resonant current waveform
shape resembles a sinusoid beginning at t3 and ending at t6. The diode currents on the secondary-side
are in boundary mode. Under this condition, the conduction loss is at its lowest and the conversion
efficiency is at its best.
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AN054―RESONANT CONVERTER CONTROLLER HR1000
C. Operating Waveform at fs>fr
Figure 5 and Figure 6 show the equivalent circuit and operation waveform, respectively.
From t0 to t1, the S1 and D1 switch on and Cr and Lr resonate because the voltage across Lm is clamped
at the reflected output voltage. im increases linearly from –im to +im.
From t1 to t2, the S1 turns off before the resonance current equals the magnetizing current as the
switching period drops to the resonant period. The resonance current is still larger than the magnetizing
current, thus the current difference ir-im is still fed to the load. The resonance current starts to flow
through the body diode D2 after discharging the parasitic capacitor. The VO reflection voltage blocks ir
and it decreases rapidly.
At stage to t2 to t3, the switch S2 turns on at the ZVS condition, the resonant current decreases to im and
D1 turns off.
For fs>fr, the primary-side ZVS condition remains, while the ZCS condition is lost because VO forces the
diode current to 0A.
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AN054―RESONANT CONVERTER CONTROLLER HR1000
S1
D1
Lr
iD1
Cr
T
Vin
ir
S2
D1
im
Vo
Lm
D2
D2
(a) Stage 1 [t0, t1]
S1
Lr
iD1
Cr
T
Vin
ir
S2
D1
im
Vo
Lm
D2
(b) Stage 2 [t1, t2]
S1
Lr
iD1
Cr
T
Vin
S2
ir
im
D1
Vo
Lm
D2
(c) Stage 3 [t2, t3]
Figure 5: Equivalent main circuit at fs>fr
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AN054―RESONANT CONVERTER CONTROLLER HR1000
S1
S1
S2
S2
ir
ir
im
im
Vds1
iD1
irec
i D1
iD2
iD2
Vc
t0
t1 t2 t3
Figure 6: Operating waveform at fs>fr
2.3 Frequency Domain Analysis
Lr
Cr
N:1
ir
Vab (1)
Lr
im
irec
Lm
Ro.eq
Cr
nVo
ir
Vab (1)
im
Lm
Req
Figure 7: Simplified AC circuit of LLC-SRC
Exact analysis of LLC-SRC converter requires the time-domain method, which leads to complex models
without a useful design methodology. Instead, this paper uses first-harmonic approximation (FHA) to
simplify calculations by assuming that the input-output power transfer is due to first order harmonics of
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AN054―RESONANT CONVERTER CONTROLLER HR1000
the fundamental Fourier series of the currents and voltages.i
This approach neglects the harmonic current and assumes that the resonant current is purely sinusoidal.
This is accurate when the switching frequency is equal to or greater than the resonant frequency in
continuous mode. It is still valid, though less accurate, when the switching frequency drops below the
resonant frequency when the current is discontinuous.
Figure 7 shows the LLC-SRC circuit simplified via FHA. During operation, the two half-bridge MOSFETs
turn on and off symmetrically at a 50% duty cycle. Thus the tank input voltage Vab is a square waveform
at the amplitude VDC with a DC component of VDC/2. Thus, Cr acts as not only the resonant tank
capacitor but also the DC-blocking capacitor.
The circuit on the left in Figure 7 can be further simplified as the circuit on the right, where Req is,
Req = N 2
8
π2
RL
(3)
and RL is the load impedance.
The fundamental of the Fourier component analysis of the input voltage can be expressed as,
Vab (1) =
2
π
Vdc sin(2π f swt )
(4)
Also the fundamental of the Fourier component analysis of the output voltage can be expressed as,
Vo (1) =
4
π
Vo sin(2π f swt − φ )
(5)
Based on the simplified AC circuit illustrated in Figure 7, the voltage gain of the output and input can be
reduced to:
M ( h, Q , f n ) =
jω Lm // Req
NV0
1
=
=
1
Vdc / 2
1
1
1
jω Lm +
+ jω Lm // Req
(1 + − 2 ) 2 + Q 2 ( f n − ) 2
jωCr
h hf n
fn
(6)
Where the parameters defined as follows:
The inductor ratio:
h=
Lm
Lr
(7)
Normalized frequency:
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AN054―RESONANT CONVERTER CONTROLLER HR1000
fn =
fs
fr
(8)
Characteristic impedance:
Z 0 = Lr / Cr
(9)
Quality factor:
Q=
Lr / Cr
Z0
=
N 2 Req
N 2 Req
(10)
Figure 8 shows a family of plots of voltage gain versus normalized frequency. For different Q values at
the inductance value, h=10, the LLC-SRC has a load-independent point at the resonant frequency (fn=1)
where all curves are tangential to its unity gain. This load-independent point is located at an inductive
zone which means the current lags behind the voltage.
In Figure 9 we can see that as h decreases, the gain curve shrinks towards to fn=1, meaning that the
minimum gain at no-load decreases.
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AN054―RESONANT CONVERTER CONTROLLER HR1000
4
Q=0.1
Q=0.3
Q=0.5
Q=0.8
Q=1
Q=2
Q=5
M
3.5
3
2.5
2
ZVS
1.5
1
ZCS
ZVS
0.5
0
0
0.5
fn
1
1.5
Figure 8: Gain characteristics of LLC-SRC
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AN054―RESONANT CONVERTER CONTROLLER HR1000
2
M
M
h=10
h=5
1.5
1
0.5
Q
Q
0
0
0.5
fn
2
1.5
2
2
h=2
M
M
h=1.1
1.5
1.5
1
1
0.5
0.5
Q
Q
0
1
fn
0
0.5
1
1.5
2
0
0
0.5
1
1.5
2
fn
fn
Figure 9: Shrinking Effect as h Increases
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AN054―RESONANT CONVERTER CONTROLLER HR1000
2.4 Performance Analysis of LLC resonant converter
A. Loss analysis on the operation at resonant frequency
At the resonance frequency, the resonant current is purely sinusoidal after ignoring the dead-time, and
the magnetizing current is a triangle waveform, as shown in Figure 10.
Figure 10: Waveforms of Resonant Current and Magnetizing Current
The resonant current can be expressed by:
ir (t ) = 2I rms _ pri sin(ωt − φ )
ω = 2π f s =
2π
T
(11)
Where Irms_pri is the rms current of the resonant current, ω is the angular representation of the resonant
frequency, and T is the switching period predetermined by the switching converter.
Since the output voltage clamps the magnetizing inductor in the first half of a PWM cycle and negative
output voltage in the second half, it can be reduced to:
NV0
⎧
if
⎪− I m + L t
⎪
m
im (t ) = ⎨
⎪ I − NV0 (t − T )
⎪⎩ m Lm
2
0<t <
if
T
2
T
<t <T
2
(12)
At time t, im(t) is equal to the peak magnetizing current, Im. Therefore, Im can be represented as:
Im =
NV0T
4 Lm
(13)
At time T, the resonant current equals the magnetizing current, therefore:
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AN054―RESONANT CONVERTER CONTROLLER HR1000
2 I rms _ pri sin(−φ ) = −
NV0T
4 Lm
(14)
The current fed to the load is the difference between ir and im,
1
T /2
T /2
∫[
2 I rms _ pri sin(ωt − φ ) +
0
NV0T NV0
V
−
t ]dt = 0
Lm
N ⋅ RL
4 Lm
(15)
Where RL is the load resistance.
From this equation, the rms of the tank current can be solved as:
T2
4π + N RL
Lm 2
2
V0
I rms _ pri =
4
2
4 2 ⋅ N ⋅ RL
(16)
The turn ratio N, the load resistance RL, the output voltage VO, and switching period T are predetermined
for a specific converter, so the rms current is only related to Lm. The lower rms value of the primary side
current translates to lower conduction loss generated by the MOSFET RDS(ON) and the inductor RDC.
Based on the turn ratio and quality factor, Q, equation (16) can be rewritten as below:
I rms _ pri =
V0
π6
4π 2 +
2
4 2 NRL
16 ( h ⋅ Q )
1
(17)
Normalizing the equation with the load current reflected on the primary side produces:
I rms _ pri _ norm =
1
4 2
4π 2 +
π6
16 ( h ⋅ Q )
2
(18)
The relationship between the h and Q and the primary rms current is shown Figure 11. The primary-side
RMS current decreases as h×Q increases. However, the effectiveness of increasing h×Q is limited when
it exceeds 6.
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AN054―RESONANT CONVERTER CONTROLLER HR1000
1.5
1.4
Irms_pri_norm ( hQ) 1.3
1.2
1.119
1.1
0
2
4
0
6
8
hQ
10
10
Figure 11: Irms_pri_norm vs. h×Q
Secondary-side conduction loss is still a concern, especially in low-voltage high-output–current
applications. Although the conduction loss is related to the forward voltage drop of diode and output
current, minimize the secondary-size RMS current when accounting for the diode’s equivalent resistance
or using SR. The output current is thus the difference between the resonant current and magnetizing
current, and its RMS is:
V0 12π 4 +
I rms _ sec = 3
5π 2 − 48 4 2 2
N RL T
Lm 2
24π RL
(19)
Based on h and Q, the equation (19) can be rewritten as:
12π 2 +
I rms _ sec = 3
V0
RL
5π 2 − 48
16 ( h ⋅ Q )
2
π4
24
(20)
It can further be normalized with the load current:
12π 2 +
I rms _ sec_ norm = 3
5π 2 − 48
16 ( h ⋅ Q )
2
π4
24
(21)
The relationship between h and Q and the secondary rms current is shown in Figure 12. The
secondary-side RMS current decreases with the increasing hQ. However, the effectiveness of increasing
of hQ is limited when it is larger than 1.
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AN054―RESONANT CONVERTER CONTROLLER HR1000
Figure 12: Irms_sec_norm vs. h×Q
As previously discussed, both the primary and secondary rms currents are determined by the
magnetizing inductor. An hQ value higher than 6 limits its effectiveness on the rms current. Increasing
the inductance of the magnetizing inductor increases hQ, which reduces the rms current and the
conduction loss but needs to choose a larger core. Decreasing the inductance of the magnetizing
inductor decreases hQ, which increases the rms current and the conduction loss but a smaller core
could be used. So it is a trade-off for hQ between the core size and the rms current. Also, the hQ value
will affect the peak gain of the converter and affect the holdup time. There is a trade-off between them.
In addition to conduction loss, the switching loss—which is composed of the turn-on and turn-off
losses—contributes substantially to the circuit efficiency. The primary-side MOSFET has zero turn-on
loss due to the ZVS condition, however hard-switching turn-offs at the peak magnetizing current
generates substantial losses. Selecting a suitable magnetizing inductance can reduce both ZVS turn-on
and turn-off loss.
B. Performance Analysis during Holdup
During holdup, the LLC boosts the output voltage by reducing the switching frequency. The minimum
input voltage that can be regulated to the normal output voltage depends on the peak voltage gain.
Efficiency is not a concern during holdup as it only last 20ms. For example, if the minimum required input
voltage is 200V and the normal input voltage is 400V, then the minimum peak gain required here is 2
since the switching frequency is designed at the resonant frequency, which has unity voltage gain.
As shown in the family of curves of voltage gain versus h and Q values in Figure 9, the peak gain equals
to one when the converter runs at the resonant frequency regardless of the h and Q values. However,
the achievable peak gain changes with the h and Q values. This estimate is based on the FHA method
for simplicity, though with an increase in error because the tank current is not precisely sinusoidal when
converter runs below the resonant frequency.
Figure 13 shows the achievable peak gain with different h-Q combinations with a 3D plot. For each h-Q
combination, there is one corresponding peak gain. This peak gain increases when h×Q drops. The map
helps to narrow down the range of valid h-Q values that meet the peak gain. For instance, if the
converter requires a peak gain that exceeds 2, then use a plane with gain equal to 2 to intersect with the
peak gain surface: The h and Q values above the plane are valid design choices.
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AN054―RESONANT CONVERTER CONTROLLER HR1000
Given the high number of h-Q combinations that meet the gain requirement, narrowing h and Q values
requires examining trade-offs between the efficiency, size, and active-component stress.
M max , M2
Figure 13: Peak Voltage Gain for Different h-Q Combinations
3. DESIGN PROCEDURE
The design goal for an LLC converter is to minimize power loss and to achieve a suitable peak gain that
ensures a wider input voltage range. As previously discussed, the conduction and switching losses
relate only to the magnetizing inductance, and discusses the relationship between the achievable peak
gain and h-Q combinations. The following methodology for LLC converter design uses these analyses,
and Figure 14 shows the flowchart of the LLC design procedure.
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AN054―RESONANT CONVERTER CONTROLLER HR1000
S ta rt
E n ter th e V in _d c _m in ,
V in _d c _ n o m , V in _ d c _m ax sp ec fo r
th e in p u t
E n ter th e V o , Io sp ec fo r th e o u tp u t , th e
estim ate efficien cy o f th e circu it
G e t th e in p u t an d to ta l o u tp u t p o w e r
G et th e tra n sfo rm er tu rn s ra tio ,N
M O S F E T ,C o ss
S w itch in g freq u en cy ,fs
G et th e M axim u m m ag n etiz in g in d u ctan ce
Lm
G e t th e p ro d u ct , h *Q
E n ter in d u ctan ce ratio , h
G e t th e q u ality facto r , Q
M ak e the Lm an d h
low er , get the high
peak gain
M eet p eak g ain req u irem en t
D es ig n th e tran sfo rm er an d res o n an t in d u cto r
Y
N
D esig n b y yo u rself ?
E n ter th e flu x d en sity ,B m
E n ter th e flu x d en s ity ,B m
E n ter th e tran sfo rm er co re sh ap e
G et th e tu rn s fo r each w in d in g
G et th e tran sfo rm er co re A e a n d A w
G et th e tu rn s fo r each w in d in g
E n ter th e each co il n u m b er
G et th e p arallel w in d in g s
E n ter th e each co il n u m b er
G et th e p arallel w in d in g s
G et th e w in d in g lo s s a n d c o re lo ss
N
N
G et th e w in d in g lo ss an d co re lo ss
P o w er L o ss o p tim iz ed
P o w er L o ss o p tim iz ed
E stim ate th e fill facto r
F ill facto < 0 .3 ?
E stim ate th e fill facto r
F ill facto < 0.3?
N
N
Y
G e t th e fin a l p a ra m e te rs c h e c k lis t
Figure 14: LLC Design Procedure
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AN054―RESONANT CONVERTER CONTROLLER HR1000
3.1 Predetermined Input and Output Specifications
The following specifications a predetermined in each LLC converter design:
- DC input voltage range: Vin_dc_min, Vin_dc_nom, Vin_dc_max, for example Vin_dc_min =320VDC, Vin_dc_nom=390VDC,
Vin_dc_max=400VDC.
- Output: VO, IO, POUT
- Switching frequency: fs
- Estimated power conversion efficiency: η
Then the maximum input power can be given as:
Pin =
POUT
η
(22)
Generally, LLC-SRC switching frequency is designed for the load-independent resonant frequency, fr, at
the normal input voltage for optimizing the efficiency. This leaves the resonant tank’s step-up capability
to handle the minimum input voltage during voltage dips.
The series resonant frequency of an LLC-SRC can be set as:
fr = fs
(23)
3.2 Determining the Transformer Turns Ratio
Selecting the transformer turns ratio provides control over the design of LLC-SRC switching frequency at
the load-independent point in normal input conditions where the voltage gain is unity. To ensure that the
LLC-SRC operates at fr, the turns ratio should meet the equation below:
N=
Vin _ dc _ nom / 2
VO
or
N=
Vin _ dc _ nom
VO
(24)
Where VO is the output voltage and Vin_dc_nom is the normal DC input voltage: the left equation is for
half-bridge applications, and the right equation is for full-bridge applications.
3.3 Design of Primary-Side Inductor, Lm
The previous section discusses the relationship between the conduction loss and switching loss; that the
conduction loss is determined by the magnetizing inductance, and that the larger inductance of
magnetizing inductor leads to lower conduction loss. Besides the conduction loss, the turn-off loss
depends on the switch-off current, which is equal to Im. Also larger inductors result in lower turn-off
losses. As discusses earlier in this document, the LLC converter has the advantage of easily achieving
the ZVS turn-on condition regardless of the load current. Discharging the MOSFET junction capacitor
during dead time ensures the ZVS condition: The discharge current equals the peak magnetizing current,
which is inversely proportional to the inductance of Lm: a larger magnetizing inductance results in a
smaller magnetizing current.
Figure 15 shows the equivalent circuit during dead time. Discharge the voltage on the MOSFET VDS to
zero during dead time to ensure the ZVS such that,
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AN054―RESONANT CONVERTER CONTROLLER HR1000
Imtdead = 2CeqVin
(25)
where Im is the peak magnetizing current during dead time, tdead is the dead time, Ceq is MOSFET
equivalent output capacitance, and Vin is the LLC bus voltage.
Given Im as expressed in Equation (13), the magnetizing inductance should satisfy the following term for
a half-bridge topology:
Lm <
T ⋅ tdead
16Ceq
(26)
Where T is the switching period (which equals the resonant period), tdead is the dead time, and Ceq is
equivalent output capacitor of MOSFET.
For full-bridge applications, the magnetizing inductance should meet the following term:
Lm <
T ⋅ tdead
8C j
(27)
So the conduction loss is determined by the magnetizing inductance, and the larger inductance of the
magnetizing inductor leads to lower conduction loss. Given that soft-switching maximizes the
magnetizing inductor, the optimizing the magnetizing inductor design is a matter of meeting the
soft-switching requirement.
Ceq
Lr
Cr
T
Vin
D1
Ceq
Im
Vo
Lm
D2
Figure 15: Equivalent Circuit during Dead Time
3.4 Determining Lr, Cr.
During holdup, the LLC boosts the output voltage by reducing the switching frequency, and regulating
the minimum input voltage to the normal output voltage relies on the peak voltage gain. Figure 13 gives
the achievable peak gain for different combinations of h and Q.
There are apparent choices to meet the minimum peak gain. To further narrow the design parameters,
we examine the magnetizing inductor. The soft-switching and conduction loss requirement determines
the magnetizing inductor. However, choosing the magnetizing inductor fixes the relationship between h
and Q in place. From the definition of h and Q, we get:
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AN054―RESONANT CONVERTER CONTROLLER HR1000
h ⋅Q =
Lm
Lr
Lr / Cr
Req
Lm Lr / Cr
π 2 2π fr Lm
=
L r N2 8 R
8 N 2RL
L
2
=
(28)
π
Where fr is resonant frequency, which equals the switching frequency, RL is the load resistor. Using the
STP11NK60 as an example, Lm can be calculated based on the ZVS requirement. Then the product of h
and Q is:
h ⋅ Q = 2.38
(29)
Narrowing down the number valid h and Q further requires a trade-off between peak voltage gain and
the efficiency, size and stress the active components. Figure 16 shows a family of gain curves with the
same product of h and Q. If h decreases, then Lr increases due to Lm; thus the Lr loss and size increases,
and Q increases accordingly. Cr decreases and leads to high voltage stress on the resonant capacitor.
This leads the peak gain and the peak current will to decrease following the expression:
I pk =
Vin / 2
Lr / Cr
sin(π
fr
fstart
)
(30)
Where the Vin is the input voltage, fr is the resonant frequency and fstart is the startup frequency.
For optimized design, select h between 4 and 10.
3
3
2.5
(
) 2
M ( 5 , 0.3 , Ω.1 )
M ( 6 , 0.25 , Ω.1 ) 1.5
M ( 10 , 0.15 , Ω.1 )
M 3 , 0.5 , Ω.1
1
0.5
0
0
0
0
0.5
1
Ω.1
1.5
1.5
Figure 16: Family of Gain Curves with the Same h×Q
As long as the inductance ratio h is define, then Lr can be calculated according equation (7), and the Cr
derived from equation (1).
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AN054―RESONANT CONVERTER CONTROLLER HR1000
3.5 Transformer Design
A. Transformer Core Selection
Select an appropriate core for the specific output power at the operating frequency; typically ferrite for
most applications. The core area product (AP) — the core magnetic cross-section area multiplied by
window area available for winding—provides an initial estimate of core size for a given application. A
rough indication of the required AEAW (cm4) is given by following equation 31ii:
AE ⋅ A W
⎛ Lm ⋅ Ip ⋅ Irms × 10 4
=⎜
⎜ B ⋅K ⋅K
max
u
j
⎝
4
⎞3
4
⎟⎟ cm
⎠
(31)
Where Ku is winding factor (typically 0.1 to 0.25 for an off-line transformer), Kj is the current-density
coefficient (typically 400 to 450 for a ferrite core), Bmax is the maximum allowable flux density at normal
operation (usually preset to be the saturation flux density of the core material; 0.1T to 0.3T), and Ipeak is
the primary-side peak current from equation (16):
Ip = 2 ⋅ Irms _ pri
Where Irms is the transformer’s total RMS current that includes the current through the primary side and
the current reflected from the secondary side. The RMS current can be derived as.
Irms = Irms _ pri +
Irms _ sec
N
(32)
B. Primary and Secondary Winding Turns
With a defined core size, the turns of secondary side can be easily deduced since the output voltage
clamps the winding:
Ns =
Vo
4fr Bmax A e
(33)
Where:
VO is the output voltage,
Bmax is the allowable flux density (generally selected according to the core loss), and
Ae is the effective area cross sectional core,
Secondary winding (Ns) is a function of N and Np, as shown in equation (34).
Np = Ns ⋅ N
(34)
C. Wire Size
Once all the winding turns are determined, select the wire size to minimize the winding conduction loss.
The winding loss depends on the RMS current value, the length and the cross section of the wire, and
the transformer structure.
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AN054―RESONANT CONVERTER CONTROLLER HR1000
Determine the wire size through the winding RMS current. For an LLC converter, the RMS current on
primary side and the secondary side are represented by equation (16) and equation (19), respectively.
The required wire size for the primary and secondary side is (respectively):
Spri =
Ssec =
Irms _ pri
J
(35)
Irms _ sec
J
Where J is the current density of the wire, which is typically 450A/cm2.
Due to the skin effect and proximity effect of the conductor, the diameter of the wire should be less than
2*Δd (where Δd is the skin-effect depth):
Δd =
1
* 10 3 (mm)
π ⋅ fs ⋅ μ ⋅ σ
(36)
Where μ is the magnetic permeability of the conductor, which usually equals to the permeability of
vacuum for most conductor, i.e. 4π×10-7H/m, and σ is the conductivity of the wire (for copper, σ is
typically 6×107S/m at 0° that increases with the temperature, which means Δd decreases).
If the required winding size is larger than Δd, use multiple strands of thinner wire or Litz wire to minimize
the AC resistance. The effective cross section area of multiple wire strands or Litz wire must meet the
requirement set by the current density.
After determining the wire size, determine whether the window area with the selected core can
accommodate the windings. Calculate the window area required by each winding and include the area
for inter-winding insulation, bobbin and spaces existing between the turns. Select a fill factor (the
winding area to the whole window area of the core) well below 1 because of the inter-winding insulation
and spaces between turns: For best results, select a fill factor no greater than about 30%. Use smaller fill
factors for transformers with multiple outputs.
Compare the total window area required to the available window area of a selected core based on these
considerations. If the required window area exceeds the selected one, either reduce the wire size select
a larger core. However, reducing the wire size increases the copper loss of the transformer.
D. Air Gap
With the selected core and winding turns, the air gap of the core is given as equation (38):
la =
μ0 * Np 2 * A e
Lm
−
lc
μr
(38)
Where:
Ae is the cross sectional area of the selected core,
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AN054―RESONANT CONVERTER CONTROLLER HR1000
μ0 is the permeability of vacuum 4π×10-7H/m,
Lm and Np are the primary winding inductance and turns, respectively,
IC is the magnetic path core length and
µr is the relative magnetic permeability of the core material.
For a ferrite core, µr is very large, so Ia can be approximated as:
la =
μ0 * Np 2 * A e
Lm
(39)
3.6 Inductor Design
A. Inductor Core Selection
To design the transformer, choose an appropriate core based on the AP value. The AP value is the
product of effective are of core (AE) and the winding window (Aw). The following equation estimates
AEAW (cm4)[1]:
AP = A E ⋅ A W
⎛ Lr ⋅ Ip ⋅ Irms × 10 4
=⎜
⎜ B ⋅K ⋅K
⎝ max u j
4
⎞3
4
⎟⎟ cm
⎠
(40)
Where:
Ku is winding factor (typically 0.2 to 0.3),
Kj is the current-density coefficient (typically 400 to 450 A/cm2 for a ferrite core),
Bmax is the maximum allowable flux density in normal operation, which is usually preset to the saturation
flux density of the core material (0.3T to 0.4T)
Ipeak is the primary-side peak current (the maximum peak current occurs at startup, so use equation
(30)).
Based on the design notes in transformer design section, the wire size is:
SL =
Irms _ pri
J
(41)
Where J is the current density of the wire (typically 450A/cm2).
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AN054―RESONANT CONVERTER CONTROLLER HR1000
3.7 Parameter Design
A. Fset, CT, SS
Oscillator internal
block diagram
Ifmin
2V
Fset
RFmax
RFmin
Rss
SS
Css
CT
VCO
Figure 17: Oscillator Internal Block Diagram
3.9V
CT
0.9V
LG
UG
SW
Figure 18: Operating Oscillator Waveform
The LLC-SRC regulates the output voltage by adjusting the operating frequency. The voltage-controlled
oscillator (VCO) shown in Figure 17 changes the frequency as programmed by the external capacitor on
CT pin. This capacitor alternately charges and discharges from a current value determined by an
external network on the Fset pin. Larger current sources lead to higher oscillator frequency. The pin
provides a 2V reference voltage with about a 2mA source-current capacity. The network on the Fset pin
is as follows:
1) RFmin. Resistor that determines the minimum frequency.
2) RFmax. Resistor connected between the Fset pin and the collector of optocoupler. The optocoupler
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AN054―RESONANT CONVERTER CONTROLLER HR1000
transfers the feedback signal from the secondary side to the primary side by modulating the collector
current—and therefore the frequency—to regulate the voltage. RFmax defines the maximum frequency
where the optocoupler is fully saturated.
3) An RC-series network. This connects between the pin and ground to form the soft-start circuit, which
sets a frequency shift during startup. Note that the contribution of this branch is zero during steady state.
Figure 18 shows the timing diagram between the oscillator waveform, the gate driver, and the swing
node of the half bridge. Note that the low-side driver is on while the triangle waveform is ramping up, and
the high-side driver is on while the triangle waveform is ramping down. This procedure ensures that the
low-side MOSFET turns on first to charge the bootstrap capacitor at startup or when the IC resumes
operation during burst mode, and guarantees the bootstrap capacitor is charged and ready to supply the
high-side driver. The triangle waveform swings between 0.9V and 3.9V as defined by internal two
comparators. Thus the minimum frequency (fmin) and maximum frequency (fmax) are:
1
3 ⋅ CF ⋅ RFmin
(42)
1
3 ⋅ CF ⋅ ( RFmin // RFmax )
(43)
f min =
f max =
After CF is fixed at hundreds of PF or nF, depending on the maximum source current capability and the
device power consumption. Select RFmin and RFmax so that the selected oscillator frequency can cover
the regulatory range; from the minimum frequency (minimum input and maximum load), to the maximum
frequency (maximum input and minimum load).
RFmin =
1
3 ⋅ CF ⋅ f min
(44)
RFmin
f max
−1
f min
(45)
RFmax =
Here RFmax determines the maximum frequency where the controller will enter burst mode operation at
the minimum load. However, if the controller enters the burst mode operation under some load, POUT, the
RFmax can be determined as:
RFmax =
3 RFmin
8 f max − 1
f min
(46)
POUT is such that the transformer peak current is low enough not to cause audible noise.
The soft-start circuit progressively increases the converter power to avoid large inrush current. The
soft-start circuit can be implemented by RSS and CSS. Since the voltage gain is inversely proportion to the
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AN054―RESONANT CONVERTER CONTROLLER HR1000
switching frequency, the soft-start functions by sweeping from the maximum frequency until the control
loop takes over.
Initially, CSS is fully discharged and RSS is effectively in parallel with RFmin so that the initial frequency is:
f start =
1
3 ⋅ CF ⋅ ( RFmin // Rss )
(47)
RFmin
f start
−1
f min
(48)
3 ⋅10 −3
Rss
(49)
Then determine RSS and CSS as：
Rss =
Css =
Where fstart is less 3 times of the resonant frequency, and CSS selection is a compromise between the
soft-start function and the OCP function.
B.
Burst Mode
Fset
RFmax
RFmin
4
HR1000
Burst
5
Figure 19: Functional Diagram of Burst Mode
When the converter runs at light load or no load, the switching frequency approaches the maximum
frequency. The magnetizing current must be high enough to continue soft-switching. This results in large
switch-off and conduction losses that keep the no-load power loss relatively high. To overcome this
issue, design the burst mode function to allow the converter to operate intermittently at no-load or at
light-load. It operates with only a few switching cycles spaced out by a long idle period where the two
MOSFETs are OFF. The result is a substantially reduced equivalent switching frequency, which reduces
the associated power loss. This facilitates converter compliance with the energy-saving no-load
requirement.
To implement burst mode, connect a resistor between the optocoupler collector and the Burst pin. If the
Burst pin voltage is lower than 1.25V, the HR1000 enters burst mode where not only the two MOSFETs
are OFF, but the oscillator stops and the output voltage continues to drop. Then the voltage on the
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AN054―RESONANT CONVERTER CONTROLLER HR1000
optocoupler collector ramps up until it exceeds 1.25V and then the IC resumes operations.
C. Current Sensing Methods
Vbus
S1
S2
6
CS
HR1000
τ≈
10
f min
Cr
Rs
Figure 20: Current Sensing with a Sense Resistor
Vbus
S1
S2
6
CS
CA RA
HR1000
CB RB
Cr
Figure 21: Current Sensing with Lossless Network
The LLC resonant converter is essentially a voltage-mode converter. Unlike conventional PWM
converters where the duty cycle controls the power, the LLC resonant converter duty cycle is fixed and
its switching frequency controls its output power. In addition, when the current exceeds a preset value,
the converter increases the switching frequency to limit the current. Frequency changes take at least
until the next cycle, making cycle-by-cycle limitation impossible.
Figure 20 shows current sensing for over-current protection using a sense resistor, and Figure 21 shows
current sensing with a lossless network
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AN054―RESONANT CONVERTER CONTROLLER HR1000
The HR1000 integrates a sophisticated over-current protection system using the CS pin. The CS pin
connects to two comparators: the first comparator has a 0.8V threshold, and the second comparator has
a 1.5V threshold. When the voltage on CS pin exceeds the 0.8V threshold, the first comparator trips and
discharges CSS. The switching frequency increases quickly to decrease the power delivered. The
discharge continues until the voltage on the CS pin drops by 50mV. Under the output-short condition,
the peak current is nearly constant by this frequency change.
If the voltage on the CS pin exceeds 1.5V, the second comparator triggers the IC to shutdown and latch
off. Restarting the IC requires that VCC drop below the UVLO threshold before rising again.
Using sense resistor for current sensing requires assuming that the RC filter’s time constant is ten times
the minimum frequency, such that the sense resistor value is:
Rs ≈
4
I crpk
(50)
Where the Icrpkx is the desired peak current through the primary switch or the resonant capacitor.
Using a lossless network requires two conditions.
1) If RA in series with CA is small (> several hundred Ωs), CA operates like a current divider. Use the
following equations to design the lossless sensing circuit.
Cr
100
(51)
0.8π
C
(1 + r )
I Crpk
CA
(52)
CA <
RB =
(2) If the resistor RA is not small (~10kΩ), then the sensing network works like a divider for the ripple
voltage on Cr. Use the following equations:
CA <
0.8π
RB =
I Crpk
Cr
100
RA 2 + X C A 2
X Cr
(53)
(54)
Where the reactance calculations of CA and Cr are based on the frequency where the maximum peak
resonant current occurs. Empirically, the RB and CB time constant is in range of 10/fmin.
With either circuit, Consider the calculated value a cut value that needs adjustment based on
experimental results to meet the design goals.
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AN054―RESONANT CONVERTER CONTROLLER HR1000
D. Input Voltage Sensing
Figure 22: Input-Voltage Sensing Block
The HR1000 provides a brown-out function when the voltage on the BO pin goes below 1.25V. The
controller then remains OFF under this condition until the soft-start capacitor discharges, the PFC-STOP
pin is open, and the IC is disabled. As the voltage on the BO pin rises and exceeds 1.25V, the IC restarts.
The internal comparator provides a current hysteresis of 15µA; this hysteresis is off, which occurs when
the BO voltage rises above the internal 1.25V reference, and on when the BO voltage drops below the
1.25V reference. This ensures the LLC resonant controller works within the defined input voltage range
to prevent over-current and voltage stress. Connect the BO pin to the tap of a resistor divider connected
to either the AC rectifier voltage or the DC bus voltage.
Based on Figure 22, the RH and RL resistors can be expressed as:
RH =
Vinon − Vinoff
RL = RH
15 ⋅10−6
(51)
1.25
Vinoff − 1.25
(52)
Where the Vinon and Vinoff are the ON/OFF threshold of the input voltage.
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AN054―RESONANT CONVERTER CONTROLLER HR1000
E. Boot-Strap Capacitor
Vcc
BST
CBST
HIGH SIDE
DRIVER
HG
SW
LEVEL
SHIFTER
Figure 23: High-Side Gate Driver
The external BST capacitor powers the high-side gate driver. VCC charges this capacitor through an
integrated bootstrap diode, which simplifies the external driving circuit for the high-side switch by
allowing to the BST capacitor to be charged when the low-side MOSFET is ON. To provide sufficient
energy and without a long charge time, select a BST capacitor value in the range of from 470nF to 1µF.
F. Low-Side Gate Driver
The LG pin provides the gate-drive signal for the low-side MOSFET. As the maximum absolute rating
table shows, the maximum voltage on the LG pin is 16V. During severe conditions—such as a short
circuit—hard-switching is unavoidable and will generate high voltage spikes on the LG pin due to the
oscillations from the long gate-drive wire and the MOSFET’s parasitic capacitance and small gate drive
resistor. This high voltage spike poses a threat on the LG pin, so add a 15V Zener diode placed close to
the LG and GND pins.
SW
Vs
LOW SIDE
DRIVER
LG
Rg Cgd
Cds
Cgs
15 V
GND
Figure 24: Low-Side Gate Driver
4. EXAMPLE DESIGN
This application note describes a 90W adapter as a reference design for the LLC resonant converter as
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AN054―RESONANT CONVERTER CONTROLLER HR1000
shown in Figure 25. The circuit consists of two stages: a front-end PFC using the MP44010, and a
resonant DC/DC converter using the HR1000. The PFC stage delivers a stable 400VDC and reduces the
mains harmonic to meet European standard EN61000-3-2. The second stage is a resonant converter
with a half-bridge topology that works in ZVS. The HR1000 controller incorporates the necessary
functions to properly drive the half-bridge with a 50% fixed-duty cycle with dead-time, and works using a
variable frequency.
This note only introduces the LLC design and the spreadsheet design tool. For PFC design, please refer
to AN045.
4.1 Specification
Table 1: Specifications for a 90W Adapter
Parameter
Symbol Value Unit
Value
Unit
Input Voltage
VAC
90 to 265
VAC
Line Frequency
fline
47 to 63
Hz
Output Voltage
VO
19.2
V
Output Current
IO
4.7
A
4.2 Schematic
PGND
PGND
VG2
VG1
VDD
EN
LL
MP6922 RCP
NC
NC
VD1
VD2
VS1
VS2
Bridge
PFC
Front
Stage
Cin
AC
input
S1
RH
Cr1
Lr
BO
Cbulk
S3
S2
Lm
RL
Output
Cr2
S4
SS
Rss
TIMER
CT
Css
CT
Fset
Rfmax Burst
Rfmin
CS
BO
LATCH
1
16
2
15
14
3
4
5
HR1000 13
12
6
11
7
10
8
9
BST
Cbst
HG
SW
TL431
N.C.
VCC
LG
GND
PFC
CB
RB
CA RA
Figure 25: Schematic for a 90W Adapter
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AN054―RESONANT CONVERTER CONTROLLER HR1000
4.3 Design Spreadsheet Using MPS’s Design Toolⅲ
A. Input and Output Specifications
The underlined red data is user input. This tool can calculate the values shown in cyan.
B. Transformer Turns Ratio
The LLC-SRC switching frequency can be designed at a load-independent point for normal input
conditions where the voltage gain is at unity. The turns ratio only depends on the input and output
voltage.
C. Transformer Primary Inductance
The peak magnetizing current must fully discharge the MOSFET’s capacitance during dead time to
satisfy the ZVS condition. The primary inductance must be lower than the value calculated in equations
26 and 27.
D. h, Lr, and Cr
Selecting the magnetizing inductor cements the relationship between h and Q. Valid h and Q values
must trade off between the peak voltage gain, efficiency, size and stress on active components. To
optimize the design, select an inductance ratio between 4 and 10 and then check whether the
achievable peak gain satisfies the minimum input voltage requirement. The required peak gain is:
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AN054―RESONANT CONVERTER CONTROLLER HR1000
Gain max =
Vin _ dc _ nom
= 1.25
Vin _ dc _ min
(53)
As shown in plot of gain vs. fn, the achievable gain is 1.41 at the minimum frequency, 60kHz, thus it
exceeds the 1.25 gain requirement. If the peak gain exceeds the gain requirement, increase h to
decrease Lr.. The Lr loss and size decreases accordingly, and increase Cr to lower the voltage stress on
the resonant capacitor.
As long as the inductance ratio h is defined, then calculate Lr using equation (7) and Cr using equation
(1).
Gain vs fn
3.0
fs
Gain(fn, Q)
2.5
2.0
1.5
1.0
0.5
0.0
0.0
0.5
1.0
1.5
2.0
fn
Gain
Gain(max)
fsmin
fs
fsmax
Figure 26: Gain vs. fn with Fixed h and Q
4.4 Transformer Core and Winding Turns
The tool provides a transformer auto-design feature: The user selects the core shape, and the tool selects
a suitable core for the specifications and calculates the number of windings. The user then determines the
diameter of the windings, and the tool calculates the fill factor. The tool also calculates the winding loss
and core loss.
If the winding loss far exceeds the core loss, using a larger winding area reduces the winding loss. To
keep the same fill factor, the user must increase the flux density (Bm) to reduce the number of turns. Adjust
Bm to balance the winding loss against the core loss to optimize transformer design.
The tool also allows for manual design. The user determines which core to use and then input Ae and Aw.
This tool calculates the number of turns, and the user determines the diameter of the windings. The tool
calculates the fill factor and the winding and core losses. Here show the transformer design based on
manual design.
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AN054―RESONANT CONVERTER CONTROLLER HR1000
4.5 Resonant Inductor Core and Winding Turns
The tool can automatically design a resonant inductor. The user selects the core shape, and the tool
auto-selects a suitable core for the specifications and calculate the number of windings. The user needs
to determine the diameter of the windings. The tool calculates the fill factor; if the fill factor exceeds 0.3,
the user must choose a larger core to accommodate the windings. The tool also calculates the winding
loss and core loss. But it is not shown here.
This tool also provides a manual design tool for inductor, just shown as follows.
4.6 Control Circuit Design
The tool can calculate the control parameters, given the following user inputs:
•
fstart: start frequency,
•
fs_min: minimum switching frequency,
•
fs_max: maximum frequency.
Select a minimum frequency range that not only guarantees inductive operation mode but also meets
the gain requirement. To guarantee inductive operation mode, select an fs_min larger than fr—the
resonant frequency between the Cr and Lr+Lm in series—and below fmin as defined as:
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AN054―RESONANT CONVERTER CONTROLLER HR1000
1
fmin = fr
1
)
Mmax
(54)
NVO
Vin _ dc _ min/ 2
(55)
1 + h(1 −
Where Mmax is:
Mmax =
At no-load condition, the converter will regulate to the maximum frequency,
fmax = fr
AN054 Rev. 1.0
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1
1 + h(1 −
1
)
Mmin
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(56)
39
AN054―RESONANT CONVERTER CONTROLLER HR1000
Generate the parameters check list
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AN054―RESONANT CONVERTER CONTROLLER HR1000
4.7 Transformer and Inductor Design
The transformer used in this design has a turns ratio of 31:3:3:3 (N1:N2:N3:N4) with 850uH primary
inductance. The core selected is EC26B.
PRI.
1
SEC.
7
N1
N3
3
2
6
N4
N2
4
5
Winding Start
TEFLON TUBE
Figure 27: Transformer Connection Diagram
Pri. Side
Sec. Side
1Ts shielding to GND
1Ts
N4
1Ts
N3
1Ts shielding to GND
N2
1Ts
Figure 28: Transformer Winding Diagram
N1
1T
Figure 28: Winding Diagram
Table 2: Transformer Winding Order
Tape(T)
Terminal
Wire size
Turns
(start-end)
(φ)
(T)
N1
1Æ3
0.23mm*2
31
N2
2Æ4
0.3mm*1
3
Winding
1
1
1
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AN054―RESONANT CONVERTER CONTROLLER HR1000
Shielding to GND
1
N3
5Æ6
N4
6Æ7
0.23mm*7
3 layers insulated wire
0.23mm*7
3 layers insulated wire
3
3
Shielding to GND
1
The resonant inductor used in this design has a turn of 50 with 100uH inductance. The core selected is EPC13.
Table 3: Inductor Winding Order
Tape(T)
Winding
1
N1
Terminal
Wire size
Turns
(start-end)
(φ)
(T)
5Æ6
0.25mm*2
50
4.8 Evaluation Board for 90W Slim Adapter
Figure 29: EV44010-S+HR1000-S-01B: 90W Slim Adapter
Based on the above design, an evaluation board for 90W slim adapter is made as shown in Figure 29.For
more detailed information of this evaluation board, please refer to the EV44010-S+HR1000-S-01B
Datasheetⅳ.
5. EXPERIMENTAL VERIFICATION
5.1 Efficiency
Table 4 and Table 5 show measured AC input power and output voltage at nominal mains with different
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AN054―RESONANT CONVERTER CONTROLLER HR1000
load conditions. Then the efficiency is calculated.
Table 4: Efficiency Measurement vs. Load at 115VAC
Vout(V)
Iout(A)
Po(W)
Pin(W)
Efficiency(%)
Full load
18.93
4.7513
89.94
99.55
90.34
3/4 load
18.95
3.965
75.14
82.95
90.57
1/2 load
18.99
2.3844
45.28
50.17
90.24
1/4 load
19.01
1.2013
22.84
26.27
86.92
Table 5: Efficiency measurement VS load at 230Vac
Vout(V)
Iout(A)
Po(W)
Pin(W)
Efficiency(%)
Full load
19.03
4.7513
90.42
97.37
92.86
3/4 load
18.97
3.9650
75.22
81.52
92.26
1/2 load
19.00
2.3838
45.29
49.31
91.84
1/4 load
19.03
1.2000
22.84
25.95
88.01
94%
93%
92%
91%
90%
89%
88%
115VAC
87%
230VAC
86%
85%
1
2
3
4
5
Figure 30: Efficiency Curve vs. Load
The measured efficiency at low AC input is shown in Table 6, which is also quite good.
Table 6: Efficiency Measurement at Low Line
Vac
Vout(V)
Iout(A)
Po(W)
Pin(W)
Efficiency(%)
90Vac
18.91
4.7075
89.02
100.4
88.64
100Vac
18.91
4.7075
89.02
99.61
89.36
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AN054―RESONANT CONVERTER CONTROLLER HR1000
5.2 Startup Operation
Figure 31: Start-Up Current Waveform
Ch1: Low-Side Driver
Ch3: SW
Ch4: Primary-Side Resonant Current
Figure 32: VCC, PFC Output Voltage Waveform at Start-Up
Ch1: VCC
Ch2: PFC Output Voltage
Ch3: SW
Ch4: Primary-Side Resonant Current
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AN054―RESONANT CONVERTER CONTROLLER HR1000
Figure 33: SS (Pin 1), Output Voltage Waveform at Start-Up
Ch1: SS (Pin 1)
Ch2: 19V Output Voltage
Ch3: SW
Ch4: Primary-Side Resonant Current
Figure 34: Start-Up Frequency, fstart, Waveform
Ch1: Low-Side Driver
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AN054―RESONANT CONVERTER CONTROLLER HR1000
Figure35: Minimum Frequency, fmin, Waveform
Ch1: Low-Side Driver
Ch3: High-Side Driver
5.3 Steady-State Operation
Figure 36: Steady-State Driver and Current Waveform
Ch1: Low-Side Driver
Ch3: SW
Ch4: Primary-Side Resonant Current
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AN054―RESONANT CONVERTER CONTROLLER HR1000
Figure 37: Maximum Frequency, fmax, at No-Load Condition
Ch1: Low-Side Driver
Ch3: SW
Ch4: Primary-Side Resonant Current
5.4 No-Load Operation
Test condition: Vac=115VAC, Vo=19.2V, Po=0W
Figure 38: Output Voltage Ripple at No-Load Condition
Ch1: Low-Side Driver
Ch2: Output Voltage Ripple
Ch3: SW
Ch4: Primary-Side Resonant Current
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AN054―RESONANT CONVERTER CONTROLLER HR1000
5.5 Transient
Test condition: Vac=115VAC, Vo=19.2V, Io=0A-4.7A
Figure 39: Transient from No-Load to Full-Load
Ch2: Output Voltage Ripple
Ch4: Output Current
5.6 Short-Circuit Protection
Figure 40: Resonant Current at Short-Circuit
Ch1: Low-Side Driver
Ch3: SW
Ch4: Primary-Side Resonant Current
NOTICE: The information in this document is subject to change without notice. Users should warrant and guarantee that third
party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not
assume any legal responsibility for any said applications.
AN054 Rev. 1.0
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AN054―RESONANT CONVERTER CONTROLLER HR1000
5.7 Over-Load Protection
Figure 41: Timer (Pin 2), CS (Pin 6) Waveform at Overload
Ch1: Timer (Pin 2)
Ch2: CS (Pin 6)
Ch3: SW
Ch4: Primary-Side Resonant Current
6. REFERENCES
i
R. L. Steigerwald, "A comparison of half-bridge resonant converter topologies," Power Electronics,
IEEE Transactions on, vol. 3, pp. 174-182, 1988.
ii
Dixon, Lloyd H. 1990. Magnetics Design for Switching Power Supplies. Unitrode Magnetics Design
Handbook. (publisher location, publisher name)
ⅲ
HR1000 Design Assistant
ⅳ
EV44010-S+HR1000-S-01B Datasheet
NOTICE: The information in this document is subject to change without notice. Users should warrant and guarantee that third
party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not
assume any legal responsibility for any said applications.
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49