Generic - Monolithic Power Systems

AN052
Reduction of No-load Power
Consumption
The Future of Analog IC Technology
Application Note for Reduction of
No-load Power Consumption
Prepared by Hommy Ding
Oct 09, 2011
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1
AN052
Reduction of No-load Power
Consumption
The Future of Analog IC Technology
ABSTRACT
This note presents a method of reducing no-load power consumption for flyback converters. Under
normal operation, power loss of a flyback converter includes conduction loss, switching loss and control
circuit loss. In no-load condition, the current in the circuit is very small which makes the conduction loss
almost negligible. Switching loss and control circuit loss are major sources of power loss, and must be
minimized to reduce no-load power consumption.
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AN052 – REDUCTION OF NO-LOAD POWER CONSUMPTION
INDEX
ABSTRACT ............................................................................................................................... 2
NO-LOAD POWER LOSS ANALYSIS...................................................................................... 4
Xcap ........................................................................................................................................ 4
Input Capacitor ....................................................................................................................... 4
RCD Snubber .......................................................................................................................... 5
Switching Components.......................................................................................................... 6
(1) MOSFET ............................................................................................................................ 6
(2) Diode.................................................................................................................................. 8
5. Transformer ............................................................................................................................ 9
6. Control Circuit ...................................................................................................................... 11
(1) IC Controller ..................................................................................................................... 11
(2) Feedback Circuit .............................................................................................................. 11
1.
2.
3.
4.
EXAMPLE ............................................................................................................................... 13
APPLICATION SUGGESTIONS ............................................................................................. 15
Discharging Resistors ................................................................................................................ 15
Electrical Capacitor .................................................................................................................... 15
Switch Component ..................................................................................................................... 15
Transformer ................................................................................................................................ 15
Control Circuit ............................................................................................................................ 15
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AN052 – REDUCTION OF NO-LOAD POWER CONSUMPTION
NO-LOAD POWER LOSS ANALYSIS
The analysis presented in this note is based on a flyback converter with an MPS current mode
controller. For a flyback converter working under no-load condition, power losses can be divided into six
parts: (1) Xcap discharge loss, (2) Input capacitor loss, (3) RCD snubber loss, (4) Loss of switching
components, (5) Transformer power loss, (6) Loss of control circuit.
1. Xcap
An Xcap is a kind of safety capacitor connected between L and N. It acts as a filter on the differential
mode interference of the power supply. If the capacitance exceeds 0.1μF, when input is disconnected,
the circuit automatically discharges the capacitor to avoid any potential electrical shock, the discharging
time must not exceed 1s for pluggable power supplies. The relevant time constant is the product of the
effective capacitance and the discharging resistance in the circuit. However, because determining the
effective capacitance and resistance values precisely is difficult and the Xcap is usually the dominant
capacitance, then we can estimate time constant as a function of the Xcap and the discharging
resistors. So that the discharging time constantτ of a RC network is then:
τ = R x ⋅ Cx
(1)
Where Cx is the Xcap, Rx is the discharging resistance. As the time constant τ can not exceed 1s, the
discharging resistors must be smaller than 1/Cx. The discharging resistors continuously dissipate power
throughout operation. The power dissipated by the discharging resistors Pdischarge can be calculated as:
Pdisch arg e
VAC 2
=
Rx
(2)
Where VAC is the rms value of the AC input voltage.
The power loss through discharging resistor contributes significantly to the no-load power loss
especially in the high-input condition. To decrease the no-load power consumption, increase the
discharging resistance, through in some instances the Xcap must decrease in order to increase the
discharging resistance, which may deteriorate EMI performance. As a compromise, choose the
appropriate Xcap and discharging resistors according to each application.
2. Input Capacitor
Power loss of electrical capacitor induced by the leakage current IR can not be ignored when the
capacitor voltage is very high. To decrease the no-load power consumption, lower the leakage current
of the input capacitor as much as possible. The leakage current IR can be calculated as:
IR = K ⋅ Cin ⋅ Vin
(3)
Where K is the coefficient of leakage current, Cin is the capacitance, and Vin is the DC input voltage.
We can obtain the loss induced by the input capacitor PCapacitor as:
PCapacitor = K ⋅ Cin ⋅ Vin2
(4)
For a flyback converter, Cin is defined by the power of the converter, and Vin is defined by the rms value
of the AC input voltage. Therefore, the coefficient K dominates the loss induced by the capacitor. It is
correlative with material purity of capacitor and use condition. At typical temperature, K is 0.01 for a
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AN052 – REDUCTION OF NO-LOAD POWER CONSUMPTION
general specific product, and it is 0.0001 for a premium product. The loss induced by the capacitor can
be several to tens of mW. Choose a capacitor with a low leakage current to minimize the standby
power loss.
3. RCD Snubber
In operation, the energy stored in the leakage inductance can not transfer to the output side of a flyback
converter. This energy may result in a high voltage spike across the MOSFET and the rectifier diode,
which can cause severe EMI noise and device failure.
The RCD snubber shown in Figure 1 suppresses the voltage spike to protect the component.
+
Vclamp
Rsn
Csn
Dsn
Lleakage
Rsn Csn
Lleakage
D
Figure 1: RCD Snubber on Primary and Secondary Side
The RCD snubber dissipates the energy of the leakage inductance and limits the voltage spikes.
Accurate analysis of the RCD snubber power loss is affected by the leakage inductance, snubber diode
and parasitic capacitance, but can be roughly estimated by assuming the energy stored in the leakage
inductance is completely dissipated by the RCD snubber circuit in steady state.
The energy stored in the leakage inductance can be expressed as:
Pleakage =
1
⋅ Lleakage ⋅ Ip r i _ peak 2 ⋅ f
2
(5)
Where Lleakage is the leakage inductance, Ipri_peak is the primary peak current, and f is the switching
frequency.
When the converter works in no-load condition, the current sense voltage threshold Vpeak can be
obtained from the datasheet of the IC controller. So the primary peak current Ipri_peak is determined by
the sense resistor Rsense. The peak current Ipri_peak under no-load condition is given as:
Ip r i _ peak =
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Vpeak
Rsense
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AN052 – REDUCTION OF NO-LOAD POWER CONSUMPTION
Meanwhile, the converter enters burst mode when working in no-load condition. An equivalent
switching frequency fs can be used to substitute the switching frequency f and be calculated by (5).
fs =
Nsw
Nsw ⋅ t sw + t burst
(7)
As shown in Figure 2, Nsw is the number of switchings in one burst time, tsw is the switching period and
tburst is the burst time.
Vgs
NSW
0
tsw
t
tburst
Figure 2: Vgs in Burst mode
By substituting (4) (5) into (3), we can obtain the energy stored in the leakage inductance as:
Pleakage =
V
Nsw
1
⋅ Lleakage ⋅ ( peak )2 ⋅
2
Rsense
Nsw ⋅ t sw + t burst
(8)
4. Switching Components
Generally, switching components include the MOSFET and diode in a flyback converter.
(1) MOSFET
Power loss of MOSFET can be divided into conduction loss, switching loss and gate driving loss. As
mentioned above, the primary peak current in no-load condition can be calculated with the current
sense voltage threshold Vpeak to derive the conduction loss of MOSFET. Usually, the gate drive stage is
integrated in the controller so the gate driving loss of MOSFET can be also included in the loss of the
control circuit.
Figure 3 shows the flyback transformer magnetizing current at no-load condition; generally it is in DCM
mode.
Ip
0
ton
toff tsw
t
Figure 3: Magnetizing Current in Transformer
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AN052 – REDUCTION OF NO-LOAD POWER CONSUMPTION
The primary side MOSFET on time ton and secondary side diode on time toff can be calculated as:
Lm ⋅ Ipri _ peak
t on =
t off =
Vin
Lm ⋅ Ipri _ peak
N ⋅ (Vout + VF )
(9)
(10)
Where Lm is the transformer magnetizing inductance, N is the transformer turn ratio (primary side to
secondary side), Vout is the output voltage, and VF is the forward voltage drop of secondary diode.
The primary side current can be calculated as:
Ipri (t) =
Vin
⋅ t,0 ≤ t ≤ t on
Lm
(11)
From (7) (9), the primary side rms and average current can be obtained as:
Ipri _ rms =
1
t SW
Ipri _ avg =
ton
∫I
pri
(t)2 dt
(12)
(t)dt
(13)
0
1
t SW
t on
∫I
pri
0
The conduction loss of MOSFET PMOSFET_conduction in no-load condition is given as:
PMOSFET _ conduction = Ipri _ rms 2 ⋅ Rds(on)
(14)
Where Rds(on) is the on state resistance of MOSFET.
The converter works in DCM mode in no-load condition. That means that the switch turns on in zerocurrent condition. So the turn on power loss of MOSFET is dominated by the equivalent primary-side
parasitic capacitance Coss which includes the MOSFET junction capacitance, transformer parasitic
capacitance, diode junction capacitance etc. In no load condition, the converter usually operates in
deep DCM mode, and the MOSFET drain-source voltage is Vds. The power loss during MOSFET turnon is:
PMOSFET _ Coss =
AN052 Rev. 1.0
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1
⋅ Coss ⋅ Vds 2 ⋅ fs
2
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AN052 – REDUCTION OF NO-LOAD POWER CONSUMPTION
Figure 4 shows the voltage and current waveforms of MOSFET during turn-off.
VDS
Vds
IDS
Ip
Td(off)
Tf
Figure 4: Voltage and Current Waveforms of MOSFET during Turning off
The switching off loss of MOSFET PMOSFET_switching is:
PMOSFET _ switching =
1
⋅ Vds ⋅ Ip ⋅ (t d(off ) + t f ) ⋅ fs
2
(16)
We can find turn-off delay time td(off) and fall time tf in the datasheet of MOSFET with a given gate
resistance. fs is the equivalent switching frequency given in (7).
(2) Diode
For low-output-voltage application, use a Schottky diode to reduce the conduction loss and avoid
potential diode reverse-recovery problem. The diode works in zero-current condition. The diode switchoff loss can be ignored and the snubber capacitance Cdiode dominates the switch-on loss. So the power
loss during diode turn-on is:
2
Pdiode _ sw int ch =
1
⎛V
⎞
⋅ Cdiode ⋅ ⎜ in + Vout ⎟ ⋅ fs
2
⎝ N
⎠
(17)
The conduction loss of diode is induced by the forward voltage drop VF of diode. VF always changes
with the forward current. For simplicity, use a constant value to calculate the power loss. In no-load
condition, VF is the voltage of diode under the secondary peak current Isec_peak.
With the primary peak current Ipri_peak, the secondary peak current Isec_peak can be calculated as:
Isec _ peak = N ⋅ Ipri _ peak
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AN052 – REDUCTION OF NO-LOAD POWER CONSUMPTION
Figure 5 shows the diode current waveform.
V GS
0
t on
t sw
Diode
Current(A)
I secondary
0
t on t on+toff
Figure 5: Current Waveform of Diode
The average value of secondary current can be written as follow:
Isec _ avg
1
⋅ I sec _ peak ⋅t off
=2
ts
(19)
The conduction loss of diode in no-load condition can be obtained as:
PDiode _ conduction = VF ⋅ Isec_ avg
(20)
5. Transformer
Transformer power loss can be divided into copper loss Pcopper and core loss Pcore. The copper loss is
caused by the winding resistance. The resistance contains DC impedance and AC impedance. So the
copper loss also contains DC loss and AC loss. The DC loss can be obtained as:
Pcopper _ DC = Ipri _ rms 2 ⋅ Rpri _ winding + Isec_ rms 2 ⋅ Rsec_ winding
(21)
Where Rpri_winding is the resistance of primary winding, Rsec_winding is the resistance of secondary winding,
which are given as:
Rp r i _ winding = ρcopper ⋅
Rsec_ winding = ρcopper ⋅
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Np r i ⋅ Lbobbin
Npri _ strand ⋅ Spri _ copper
Nsec ⋅ Lbobbin
Nsec_ sec tion ⋅ Ssec_ copper
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(23)
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AN052 – REDUCTION OF NO-LOAD POWER CONSUMPTION
Where
z
ρcopper is the conductivity of copper,
z
Npri is the number of primary turns,
z
Nsec is the number of secondary turns,
z
Lbobbin is the mean length of the turn,
z
Npri_strand is the strands of primary winding wire,
z
Nsec_strand is the strands of secondary winding wire,
z
Spri_copper is the cross section area of single primary winding wire,
z
Ssec_copper is the cross section area of single secondary winding wire.
The analysis of AC loss is difficult to calculate because of the difficulty in calculating the AC impedance
and AC current accurately. Based on the AC transformer winding resistance calculation model of
Dowell, the AC power loss can be calculated as:
Pcopper _ AC = Ip r i _ ACrms 2 ⋅ αR p r i _ winding +Isec_ ACrms 2 ⋅ αRsec_ winding
(24)
Where α is the empirical factor to estimate the AC resistance due to the calculation model. It is about
1.5 to 2 in this note.
The rms AC current value of primary and secondary side can be can be estimated by:
Ip r i _ ACrms = Ipri _ rms 2 − Ip r i _ avg2
(25)
Isec_ ACrms = Isec_ rms 2 − Isec_ avg2
(26)
The core loss can be calculated with an empirical formula as below:
PCore = Cm ⋅ (fs )x ⋅ (Bmax )y ⋅ (Ct0 − Ct1 ⋅ TCore + Ct 2 ⋅ TCore 2 ) ⋅ Ve
(27)
Where Cm, x, y, Ct0, Ct1, and Ct2 are coefficients related to the material of core, Bmax is the maximum
magnetic flux, Tcore is the temperature of core.
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AN052 – REDUCTION OF NO-LOAD POWER CONSUMPTION
6. Control Circuit
(1) IC Controller
The power loss of IC controller contains internal IC consumption and the loss induced by the start up
circuit.
For internal IC consumption, the power loss can be calculated as:
PIC = VCC ⋅ ICC
(28)
Where Vcc is the supply voltage of the IC controller, Icc is the operation current in no-load condition. To
decrease the no-load power consumption, select the lowest-possible voltage. This voltage must be
higher than the lowest operating voltage of the IC controller. Select the voltage VCC using the turn-ratio
between the secondary and auxiliary winding for a given output voltage.
The IC controller needs a circuit to start up. Some ICs have an HV pin, some have startup resistors.
Both the circuits will consume the power.
For the ICs with an HV pin, we can find the leakage current Ileakage from HV pin in the datasheet. The
loss induced by the leakage current can be estimated as:
PLeakage _ current = Vin ⋅ ILeakage
(29)
Where Vin is the DC input voltage. Minimize the leakage current on the HV pin to decrease this power
loss. However, it is mainly determined by the chip process.
For the IC with startup resistors, estimate the loss induced by the startup resistors as:
Pstartup
( V − VCC )
= in
2
(30)
Rstartup
To decrease this power loss, use large startup resistors. However, large startup resistors slow the
startup speed and may even result in startup failure.
(2) Feedback Circuit
For a flyback converter with isolated output, typically adopt an optocoupler and three-terminal
programmable shunt regulator like the TL431 to achieve output voltage feedback. For example, the
HFC0300 feedback circuit shown in Figure 6 consumes some power for normal operation. To minimize
the no-load power consumption, minimize the power loss of the feedback circuit. For an isolated flyback
converter, the output voltage usually powers the optocoupler and the regulator. If the converter has
multiple outputs, choose the lower voltage as the supply voltage.
Alternatively, choose a regulator with a lower operating current and an optocoupler with a high current
transfer ratio (CTR).
Vout
VCC
R1
R2
Vcomp
Ccomp
V ref
Rcomp
PGND
SGND
Figure 6: Feedback Circuit
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AN052 – REDUCTION OF NO-LOAD POWER CONSUMPTION
For the HFC0300, Vcomp is about 3.1V when the converter is working in burst mode condition. The
primary and secondary optocoupler current can be estimated as:
Isec _ photocoupler =
Ip r i _ photocoupler =
Vcomp
Rcomp
Isec _ photocoupler
CTR
(31)
(32)
Where CTR is the current transfer ratio of the optocoupler.
The power loss of feedback circuit contains two parts as shown below:
Pfeedback = VCC ⋅ Isec _ photocoupler + Vout ⋅ Ip r i _ photocoupler
(33)
Using a regulator with a low operating current and high CTR optocoupler, the primary and secondary
side current of the feedback circuit drops and reduces the power loss. If the voltage supply for the
feedback circuit is very high (typically range 12V- 24V), we can save 10mW-20mW power loss by
choosing a regulator with a lower operating current and an optocoupler with a higher CTR.
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AN052 – REDUCTION OF NO-LOAD POWER CONSUMPTION
EXAMPLE
In order to show the validity of no-load consumption analysis, a flyback converter controlled by the
HFC0300 was built and tested. The AC input is 90 Vrms to 264Vrms; the outputs are 5V/3A and 24V/1.5A
respectively. The circuit of the converter is shown in Figure 7.
L
N
1
1
1
F1
2
RT1
5.1
CX1
0.22uF
1M
R1A
1 2
1M
R2A
LX1
32mH
3
4
2
BD1
3
3
R5
1
R4
1
Q1
R3
1
P1065ATF
1
4
1
2
R7
10
C1
100uF
2
C2
1
R6
20K
R8
JR3
JR2
3.48K
3.48K
RF2
R9
2.2nF
C4
D1
1
1
2
HV
FR107
N/C
D2
DRV
VCC
FR107
CS
10K
GND
FSET
U3
COMP
200K
1
2
3
33pF
4
2
1K
1
3
5
6
R20
0
R10
1
8
7
6
C7
5
1
EER28
12
11
8
7
10
9
R27
4
2
3
51
R12
R11
0
10K
C19
C3
0.1uF
CY4 4.7nF
T1
C20
470pF 47uF
10nF
3
1
D3 1nF
R16
2.2
D4
C8
MBRF10150CT
3
1
SP1060
2
1
2
2
1
1
C12
1000uF
L1
3.3uH
2
C15
2
1
L2
3.3uH
220uF
1
C14
2
1000uF
2
C16
2
3
220uF
1
2
2K
R15
R17
10K
1
2.2nF
C10
C11
1000uF
U1
C9
10nF
PC817A
1
2
1K
R14
1
R13
20K
U2
TL431
2
1uF
C17
1uF
C18
R23
274K
12.4K
R24
1
2
1
2
CN2
CN3
Figure 7: Circuit of HFC0300 Controlled Flyback Converter
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AN052 – REDUCTION OF NO-LOAD POWER CONSUMPTION
The measured no-load power consumption was 91.5mW at 264Vrms AC input. To decrease the standby
power loss, some measure is given as: 1) Increase the discharging resistors to 4MΩ. 2) Optimize the
transformer design. 3) Choose 5V output voltage as the supply voltage of feedback circuit and use a
regulator of 1.24V reference. The measured no-load power consumption decreased to 51.7mW, and
the waveform of burst mode at no-load condition is as shown in Figure 8.
CH1:Vds
CH1:Vds
CH2:Vcc
CH2:Vcc
CH3:Vcomp
CH3:Vcomp
Figure 8: Waveform of Burst Mode at No-load
We can obtain the equivalent switch frequency fs from the waveform. As per the analysis of no-load
power consumption previously discussed, we can calculate the no-load power loss of each part and
summarized them in Table 1.
Table 1: No-Load Power Loss Breakdown
Input Voltage
264VAC
fs
107Hz
No-load power loss breakdown
Discharging
Input capacitor
17.42mW
resistor
RCD snubber
MOSFET
0.15mW
Diode
Transformer
1.76mW
IC(HFC0300)
Feedback circuit
15.34mw
7.27mW
3.46mW
0.12mW
1.07mW
Thus, we can find the loss through the discharging resistor and IC are the major part of total no-load
power loss. However, with the increased equivalent frequency, the power loss of MOSFET, diode and
transformer increase significantly and dominates.
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AN052 – REDUCTION OF NO-LOAD POWER CONSUMPTION
APPLICATION SUGGESTIONS
Follow the suggestions below to decrease the no-load power consumption by decreasing the loss on
each component.
Discharging Resistors
Small discharging resistors will cause higher power loss. Choose a suitable Xcap to optimize the noload loss and EMI.
Electrical Capacitor
Choose an appropriate capacitor with relatively low leakage current, balanced against increased cost.
Switch Component
z
Choose a MOSFET with low Rds(on), high switching speed, and low output capacitance.
z
Use an application-appropriate gate drive resistor for the MOSFET to balance efficiency against
EMI.
z
Use a Schottky diode with a low forward voltage drop
Transformer
z
Choose an appropriate winding size and use multiple strands of wire
z
Use a transformer with a sandwich winding structure to decrease the leakage inductance
z
Choose a low loss core material
Control Circuit
z
Optimize the IC losses by decreasing the loss on the startup circuit and the operation current in noload condition.
z
Use a regulator with a low operating current and an optocoupler with a high-CTR
z
Design an appropriate feedback circuit to decrease the equivalent frequency as much as possible.
NOTICE: The information in this document is subject to change without notice. Users should warrant and guarantee that third
party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not
assume any legal responsibility for any said applications.
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