Design Guidelines for Quasi-Resonant Flyback Converters Using

AN027
Quasi-Resonant Flyback Converters
The Future of Analog IC Technology
Design Guidelines for Quasi-Resonant
Flyback Converters Using HFC0100
Application Note
Prepared by Lei Miao
Jan. 11, 2010
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AN027
Quasi-Resonant Flyback Converters
The Future of Analog IC Technology
ABSTRACT
This paper presents design guidelines for quasi-resonant flyback converters with current mode
controller-HFC0100 of MPS. Design of a quasi-resonant flyback converter with HFC0100 is made easier
through use of this step-by-step design procedure from this paper. Experimental results based on the
design example are presented in the last part.
T1
*
Input
85-265VAC
Output
+
+
*
RTN
*
HV
4
5
Drive
N/C
3
6
CS
VCC
2
7
GND
VSD
1
8
FB
HFC0100
Basic Quasi―Resonant Flyback Converter Using HFC0100
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AN027 –QUASI-RESONANT FLYBACK CONVERTERS
INDEX
1.
HFC0100 INTRODUCTION ......................................................................................................... 4
2.
QUASI-RESONANT OPERATION INTRODUCTION ................................................... 4
3.
DESIGN PROCEDURE ............................................................................................................................ 6
A. Predetermine the input and output specifications. .......................................................................... 6
B. Determine the Startup Circuitry ......................................................................................................... 7
C. Valley Switching Detector and OVP Circuitry .................................................................................. 8
D. Turns ratio-N, Primary MOSFET and Secondary Rectifier Diode Selection ................................. 9
E. Primary inductance Lm, peak current IP and OLP function.......................................................... 11
F. Transformer Design ......................................................................................................................... 13
F-1. Transformer core selection ...................................................................................................... 13
F-2. Primary and secondary winding turns ..................................................................................... 14
F-3. Wire size ......................................................................................................................... 14
F-4. Air gap........................................................................................................................................ 15
G. Design the RCD snubber ................................................................................................................. 16
H. Design the output filters ................................................................................................................... 17
4.
DESIGN SUMMARY ..................................................................................................................... 17
5.
EXPERIMENTAL VERIFICATION ......................................................................................... 18
6.
REFERENCES ................................................................................................................................. 21
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AN027 –QUASI-RESONANT FLYBACK CONVERTERS
1. HFC0100 INTRODUCTION
HFC0100 is a peak current mode controller that integrated with a high voltage current source; its internal
valley detector ensures minimum Drain-Source voltage switching (Quasi-Resonant operation). When the
output power falls below a given level, the controller enters the burst mode for lower power loss at
no/light load condition. HFC0100 can be adopted in off-line, telecom and non-isolated applications.
Internal Vcc Undervoltage Lockout (UVLO), Over Load Protection (OLP), Over Voltage Protection (OVP)
are all provided to minimize the external component count. This paper presents practical design
guidelines for a quasi-resonant flyback converter employing HFC0100. Step-by-step design procedure
for quasi-resonant flyback converter using HFC0100 can be applied to various offline applications,
mainly including transformer design, output filter design and component selection.
2. QUASI-RESONANT OPERATION INTRODUCTION
Quasi-resonant conversion works in quite a different way than the well-known resonant converter to cut
losses. Figure 1 shows the drain-source voltage waveform of primary switch in a current-mode flyback
converter operating in the discontinuous conduction mode (DCM). During the first time interval, the drain
current ramps up until the desired current level is reached. The power MOSFET then turns off. The
leakage inductance in the flyback transformer rings with the MOSFET parasitic capacitance and causes
a high voltage spike, which is limited by a clamp circuit. After the inductive spike has damped, the drain
voltage equals to the input voltage plus the reflected output voltage. The drain voltage would
immediately drop to the bus voltage when the current in the output diode drops to zero if the parasitic
ring of the primary inductance and the parasitic capacitance is ignored. However, the drain voltage rings
down to this level as shown in Fig 1 due to the parasitic resonance by the primary inductance and
parasitic capacitance.
For example the inductance is 1mH and the parasitic capacitance is 100pF, then the resonant
frequency is 500 kHz. The resonant circuit is lightly damped and the resonant frequency given below is
independent of the input voltage and load currents:
fresonant =
1
2π ⋅ Lm ⋅ Ceqp
where Lm is the primary inductance; Ceqp is the equivalent primary side parasitic capacitance which
including parasitic capacitance of the primary winding, the parasitic capacitance of the MOSFET and the
parasitic capacitance of the secondary side (including the secondary winding and output rectifier diode)
reflect to primary side.
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AN027 –QUASI-RESONANT FLYBACK CONVERTERS
Figure 1 — Single-Pulsed Flyback Converter
In a conventional fix frequency flyback converter at DCM operation the primary switch (MOSFET) is
turned on at a fixed frequency and turned off when the current reaches the desired level. The device's
turn-on time may occur at any point during this parasitic resonance. In some cases the device may turn
on when the drain voltage is lower than the bus voltage (means low switching losses and high efficiency),
and in some cases the switch will turn on when the drain voltage is higher above the bus voltage (means
high switching loss). This characteristic is often observed on the efficiency curves of a discontinuous
flyback converters with a constant load, the efficiency fluctuated with the input voltage as the turn-on
switching loss changes due to the variation of the drain voltage at the turn on point.
In quasi-resonant (QR) operation, the switch does not have a fixed switching frequency. Instead, the
switch will always turn on by the controller when the drain voltage reaches its minimum value (valley
point, refer to Fig.2). The time period Tw is half the resonant period determined by the transformer
magnetizing inductance and parasitic capacitance. The switch on time (T1 in Fig 2) is determined by the
output feedback loop as conventional peak current mode control. The energy stored in the magnetizing
inductor is fully transferred to the output. For light load condition, the time T1 is small as less energy is
required by the load, resulting in small peak current, and also a shorter output diode conduction time.
Therefore, the switching frequency increases in the QR flyback as load decreasing, which may
deteriorate the light load efficiency and challenges the EMI design. In order to eliminate these problems,
a frequency-clamp function is usually added into the controller to limit the maximum switching frequency
when load decreasing, which further improves the light load efficiency and simplifies the converter
design.
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AN027 –QUASI-RESONANT FLYBACK CONVERTERS
Figure 2 — Quasi-Resonant Flyback Converter
Compared to the traditional flyback under CCM and DCM operation, the Quasi-resonant operation can
minimize the turn on switching loss of the switch by switching at the valley point, thus increasing
efficiency and lowering device temperature rise. Its main disadvantage that of higher switching
frequency at light load is eliminated by the frequency-clamp function in controllers. Also, by switching at
low voltages and currents, the EMI generated by quasi-resonant operation is relatively low.
MPS provides HFC0100 for the quasi-resonant flyback solution, the internal valley detector ensures
valley switching for high efficiency and low EMI. The minimum 8us off time limits the maximum switching
frequency when load decreases. When the load is fairly light, HFC0100 will force the system into burst
mode operation to further reduce the equivalent switching frequency.
3. DESIGN PROCEDURE
A. Predetermine the Input and Output Specifications.
- Input AC voltage range: Vac(min), Vac(max), for example 90Vac~265Vac RMS
- DC bus voltage range: Vin(max), Vin(min).
- Output: Vout , Iout(min), Iout(max), Pout
- Estimated efficiency: η, It is used to estimate the power conversion efficiency to calculate
the maximum input power. Generally, η is set to be 0.8~0.9 according to different output
applications.
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AN027 –QUASI-RESONANT FLYBACK CONVERTERS
Then the maximum input power can be given as:
Pin =
Pout
η
(1)
Fig.3 shows the typical DC bus voltage waveform. The DC input capacitor Cin is usually set as 2uF/W of
input power Pin for the universal input condition. For 230V single range application, the capacitance can
be half the value.
Figure 3 — Input Voltage Waveform
From the waveform above, the AC input Voltage VAC and DC input Voltage VDC can be got as:
VAC (Vac ,t) = 2 ⋅ Vac ⋅ cos(2 ⋅ π ⋅ f ⋅ t)
VDC (Vac ,t) = 2 × Vac 2 −
2 × Pin
×t
Cin
(2)
(3)
By setting VAC=VDC, T1 where DC input voltage had reached to its minimum VDC(min) can be calculated.
VDC(min) = VDC (Vac(min) ,T1)
(4)
Then, the minimum average DC input voltage Vin(min) can be got as:
Vin(min) =
2 ⋅ Vac(min) + VDC(min)
2
(5)
The maximum average DC input voltage Vin(max) can be got as:
Vin(max) = 2 ⋅ Vac(max)
(6)
B. Determine the Startup Circuitry
Fig.4 shows the startup circuit, when power is on, the internal 2mA current charges C1 through R1
connected on HV pin of HFC0100. Once VCC voltage reaches 12V, the internal high voltage current
source (2mA) turned off and IC start switching, then the auxiliary winding take over the power supply. If
VCC dropped below 8V before the auxiliary winding take over the power supply, the switching is stopped
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AN027 –QUASI-RESONANT FLYBACK CONVERTERS
and the internal high voltage current source turned on again, re-charge the VCC external capacitor C1
(see Fig.5).
Figure 4 — The Startup Circuit with HFC0100
Figure 5 — The Startup and VCC UVLO of HFC0100
C. Valley Switching Detector and OVP Circuitry
Fig.6 shows the VSD pin circuitry. The VSD pin is connected to the auxiliary winding by RVSD. The VSD
pin is used for two functions:
z
Detects the valley voltage of the switching waveform to achieve the valley voltage switching,
which ensures QR operation to minimize the switching losses and reduces EMI.
z
An internal voltage comparator and 6V reference voltage provide an output OVP
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AN027 –QUASI-RESONANT FLYBACK CONVERTERS
Figure 6 — The VSD Pin Circuitry of HFC0100 to Ensure Valley Switching
The internal valley switching signal occurs when the VSD pin voltage reaches:
(VDS − Vin ) ⋅
Naux
24kΩ
⋅
< 55mV
Npri 24kΩ + R VSD
where VDS is the Drain-Source Voltage of the primary FET; Naux is the auxiliary winding Turns of the
transformer and Npri is primary winding turns of the transformer.
The output OVP is achieved by detecting the positive plateau of auxiliary winding voltage which is
proportional to the output voltage (see Fig.7). The output OVP setting point can be calculated as:
Vout −p ⋅
Naux
24kΩ
⋅
> 6V
Nsec 24kΩ + R VSD
where Vout-p is the output OVP setting voltage; Naux is the auxiliary winding Turns of the transformer and
Nsec is secondary winding turns of the transformer.
To avoid the mis-trigger due to the oscillation of the leakage inductance and the parasitic capacitance
(see Fig.7) after primary switch turns off, the OVP sampling has a TOVPS blanking period, typical 3.5us.
Figure 7— The OVP Sampling and TOVPS Blanking
D. Turns Ratio-N, Primary MOSFET and Secondary Rectifier Diode Selection
Fig.8 shows the typical Drain-Source voltage waveform of the primary MOSFET and secondary rectifier
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AN027 –QUASI-RESONANT FLYBACK CONVERTERS
diode in a quasi-resonant flyback converter. From the waveform, the primary MOSFET Drain-Source
voltage rating VP-MOS can be got as:
VP −MOS =
Vin(max) + VRO + 60V
k
(7)
where k is the derating factor which is typically selected as 0.9. VRO is the reflected output
voltage: VRO = N ⋅ VOUT , 60V spike voltage is assumed here.
The secondary rectifier diode voltage rating VDIODE can be got as:
VDIODE =
Vin(max) / N + VOUT
k
(8)
Figure 8 — Drain-Source Voltage of Primary MOSFET and Secondary Rectifier Diode
From (7) and (8), the voltage rating of primary MOSFET and secondary rectifier diode versus turns-ratio
N can is shown in Fig 9. Then the turns-ratio N can be determined for the required MOSFET and
Rectifier diode voltage rating. For example, in 24V output adapter application, 650V MOSFET and 100V
rectifier diode is preferred for better performance. From Fig.9, N=6 is selected for the required voltage
rating. Sometimes N can be selected within a range, then smaller N means larger duty on secondary
and lower voltage stress on primary (more power loss on secondary and less power loss on primary);
while larger N means smaller duty on secondary and higher voltage stress on primary (more power loss
on primary and less power loss on secondary).
Figure 9—Voltage Rating of Primary MOSFET and Secondary Rectifier Diode vs. Turn Ratio-N
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AN027 –QUASI-RESONANT FLYBACK CONVERTERS
E. Primary Inductance Lm, Peak Current IP and OLP Function
Primary inductance Lm and peak current IP determines the power can be transferred to the output side:
Pin =
1
⋅ Lm ⋅ IP 2 ⋅ fs
2
(9)
Fig.10 shows the primary side MOSFET Drain-Source voltage and current waveform in a quasi-resonant
flyback converter. It is preferred that the primary switch turns on at the first valley at heavy load condition
to minimize the peak current. Each time period can be calculated as:
Ton =
Lm ⋅ IP
Vin
TOFF =
Lm ⋅ IP
N ⋅ VOUT
Tw = π ⋅ Lm ⋅ Ceqp
(10)
(11)
(12)
where Ceqp is the equivalent parasitic capacitance of the primary side, including parasitic capacitance of
the primary winding and secondary winding, the parasitic capacitance of the MOSFET and the output
rectifier.
Figure 10 — Drain-Source voltage and current of Primary MOSFET in Quasi-Resonant Flyback
Converter
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AN027 –QUASI-RESONANT FLYBACK CONVERTERS
From (10) (11) and (12), the switching frequency is given as:
fs =
1
TON + TOFF + Tw
(13)
Take (1) (10) (11) (12) and (13) into equation (9), assume the power loss mainly is dominated by the
secondary side:
POUT 1
1
= ⋅ Lm ⋅ IP 2 ⋅
Lm ⋅ IP
L ⋅I
2
η
+ m P + π ⋅ Lm ⋅ Ceqp
Vin
N ⋅ VOUT
(14)
Usually the Tw only has a small amount of the total switching period, which can be neglected for
simplicity. Thus, (14) can be simplified as:
POUT 1
1
= ⋅ Lm ⋅ IP 2 ⋅
Lm ⋅ IP
L ⋅I
2
η
+ m P
Vin
N ⋅ VOUT
(15)
From (15), the primary peak current IP can be got as:
IP (Vin ) =
2 ⋅ POUT ⎡ 1
1 ⎤
⋅⎢
+
⎥
η
⎣ Vin N ⋅ VOUT ⎦
(16)
Then, the maximum peak current IP at the minimum input line is IP(Vin(min)). So that the primary inductance
Lm can be got from (9) as:
Lm =
2 ⋅ POUT
1
⋅
η
IP( Vin(min))2 ⋅ fs min
(17)
Where fsmin is the expected minimum switching frequency at low line with full load condition, for example
assume the fsmin as 60kHz, then IP and Lm can be calculated according to (16) and (17).
As introduced in the previous section, in order to avoid the very high switching frequency at light load
condition for quasi-resonant operation, the controller usually contains a frequency-clamp function to
limit the maximum switching frequency. For HFC0100, it has 8us minimum-off time to limit the switching
frequency.
Be sure that the off time at low line and full load does not fall below the allowed minimum-off time of
HFC0100. The calculated inductance use (17) should be above the minimum inductance, which is
given as:
Lm ≥
N ⋅ VOUT ⋅ (Toff _ min − TW )
Ip( Vin(min))
(18)
Where Toff_min is the minimum off time, i.e. 8us; and Tw is given in (12).
If the calculated inductance at low line and full load condition with (17) is above the minimum value, the
design is OK. Otherwise, we need to adjust the transformer turns-ratio N or decrease fsmin to do the
re-design
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AN027 –QUASI-RESONANT FLYBACK CONVERTERS
HFC0100 employs an external sense resistor RSENSE to detect the load for primary current limiting. If the
primary peak current IP is high enough which causes the voltage on RSENSE exceeds the internal current
limiting reference voltage (Vref=1V), the controller will shut off the switching cycle. To simply explain the
design of RSENSE, it is assumed that, the current limiting function is expected to be triggered when output
current has reached 105% Iout(max). Then the output power will be α·Pout (α=105%), (15) changes to:
α ⋅ POUT 1
1
= ⋅ Lm ⋅ IP −L 2 ⋅
Lm ⋅ IP −L Lm ⋅ IP −L
η
2
+
Vin(min) N ⋅ VOUT
(19)
where IP-L is the primary peak current limiting value, from (19) it can be got that:
IP−L =
2 ⋅ α ⋅ POUT
η
⎡ 1
1 ⎤
⋅⎢
+
⎥
⎢⎣ Vin(min) N ⋅ VOUT ⎥⎦
(20)
then the sense resistor RSENSE can be calculated as:
RSENSE =
Vref
IP−L
(21)
where Vref is the internal current limiting reference voltage, which is about 1V.
For HFC0100, it has internal OLP (Over Load Protection) function for safe operation. Because the
maximum output power is limited by the maximum switching frequency and maximum primary peak
current. If the output consumes more than the maximum output power, the output voltage will drop below
the set point, the feedback loop will reduce the current through the optocoupler LED to try to keep the
output voltage, and FB pin voltage will increase.
By continuously monitoring the FB pin voltage, when the feedback voltage exceeds the threshold
VOLP—3.4V, the gate drive signal is prohibted. The HFC0100 enters a safe low power operation that
prevents from any thermal runaway or electrical over stress. As soon as the fault disappears, the power
supply resumes operation.
During the start up or load transient, the FB voltage will be high enough temporarily to mis-trigger the
OLP, to prevent this undesired protection, OLP circuit is designed to be triggered after Vcc is decreased
below 8.5V.
F. Transformer Design
F-1. Transformer Core Selection
A core appropriate for certain output power at the operating frequency needs to be selected. Ferrite is
widely adopted in flyback transformer. The core area product (AeAW) which is the core magnetic
cross-section area multiplied by window area available for winding, is widely used for an initial estimate
of core size for a given application. A rough indication of the required area product is given by
following[1]:
AE ⋅ A W
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⎛ Lm ⋅ IP ⋅ Irms × 104 ⎞
=⎜
⎟⎟
⎜ B ⋅K ⋅K ⋅ f
⎝ max u j smin ⎠
4/3
cm4
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AN027 –QUASI-RESONANT FLYBACK CONVERTERS
where Ku is winding factor which is usually 0.25~0.3 for an off-line transformer. Kj is the current-density
coefficient (typically 400~450 for ferrite core). IP and Irms is the
maximum peak current and RMS current of the primary inductance. Bmax is the allowed maximum flux
density in normal operation which is usually preset to be the saturation flux density of the core material
(0.3T~0.4T). fsmin is the minimum switching frequency at low line with full load condition.
Irms
Lm ⋅ IP3
≈
⋅ fs
3 ⋅ Vin(min)
(23)
Please refer to the manufacture’s datasheet to select the proper core. For discontinuous mode operation,
the winding area window is chosen as wide as possible to minimize AC winding losses. EC, ETD, EFD,
LP cores are all E-E core shapes with large wide windows. Applications requiring low profile can benefit
from using EFD cores.
F-2. Primary and Secondary Winding Turns
With a given core size, there is a minimum number of turns for the transformer primary side winding to
avoid saturation. The normal saturation specification is E-T or volt-second rating. The E-T rating is the
maximum voltage, E, which can be applied over a time of T seconds. (The E-T rating is identical to the
product of inductance L and peak current) Equation (24) defines a minimum value of NP for the
transformer primary winding to avoid the core saturation:
NP =
Lm ⋅ IP
× 106
B ⋅ AE
(24)
Where:
Lm = the primary inductance of the transformer
B= the maximum allowable flux density
AE= the effective cross sectional core
IP= the peak current in the primary side of the transformer, which is given in (16).
The maximum allowable flux density B should be smaller than the saturation flux density Bsat. Since Bsat
decreases as the temperature goes high, the high temperature characteristics should be considered.
Secondary turn count is a function of turn ratio n and primary turn count NP:
NS = NP / N
(25)
F-3. Wire Size
Once all the winding turns have been determined, wire size must be properly chosen to minimize the
winding conduction loss and leakage inductance. The winding loss depends on the RMS current value,
the length and the cross section of wire.
The wire size could be determined by the RMS current of the winding. For a flyback converter, the RMS
current on secondary side is:
Isec −rms ≈
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Lm ⋅ IP3
⋅ fs ⋅ N
3 ⋅ VOUT
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AN027 –QUASI-RESONANT FLYBACK CONVERTERS
Then, the wire size required on secondary side is:
S=
Isec −rms
(mm2 )
J
(27)
Here J is the current density of the wire which is 5A/mm2 typically.
Due to the skin effect and proximity effect of the conductor, the diameter of the wire selected is usually
less than 2*Δd (Δd: skin effect depth):
Δd =
1
π ⋅ fs ⋅ μ ⋅ σ
(28)
where μ is the magnetic permeability of the conductor, which is usually equals to the permeability of
vacuum for most conductor, i.e. 4π × 10 −7 H/m, σ is the conductivity of the wire (for copper, σ is typically
6 × 10 7 S/m at 0 deg, σ will be larger as temperature increases, which means the Δd will get smaller).
Therefore, multiple strands of thinner wire or Litz wire is usually adopted to minimize the AC resistance,
the effective cross section area of multi-strands wire or Litz wire should large enough to meet the
requirement set by the current density. Equation (23) and (27) can be used for primary wire size
selection.
After the wire sizes have been determined, it is necessary to check whether the window area with
selected core can accommodate the windings calculated in the previous steps. The window area
required by each winding should be calculated respectively and added together, the area for
interwinding insulation and spaces existing between the turns should also be taken into consideration.
The fill factor, means the winding area to the whole window area of the core, should be well below 1 due
to these interwinding insulation and spaces between turns. It is recommended that a fill factor no greater
than about 30% be used. For transformers with multiple outputs this factor may need to be reduced
further.
Based on these considerations, the total required window area is then compared to the available window
area of a selected core. If the required window area is larger than the selected one, either wire size must
be reduced, or a larger core must be chosen. Of course, a reduction in wire size increase the copper
loss of the transformer.
F-4. Air Gap
With the selected core and winding turns, the air gap of the core is given as:
G = μ0 ⋅ A E ⋅
NP2 lc
−
Lm μr
(29)
where AE is the cross sectional area of the selected core, μ0 is the permeability of vacuum which equals
4π × 10 −7 H/m. Lm and NP is the primary winding inductance and turns respectively, l c is the core
magnetic path length and μ r is the relative magnetic permeability of the core material.
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AN027 –QUASI-RESONANT FLYBACK CONVERTERS
G. Design the RCD Snubber
In application, a small amount of energy is stored in the leakage inductance, which cannot be transferred
to the output side in flyback converter. This amount of energy may result in a high voltage spike on the
drain of the main switch, which should be well damped to protect the MOSFET.
The RCD snubber is usually adopted to clamp the drain voltage as shown in Fig 11. The value of the
capacitor, Csn, and resistor, Rsn, depend on the energy stored in the parasitic inductance, as the energy
must be dissipated by the RC network during each cycle. The voltage across the capacitor and resistor
sets the clamp voltage, VCLAMP. Fig.12 shows the voltage of the primary MOSFET during turn-off phase
with respect to VCLAMP.
Figure 11 — RCD Snubber on Primary Side
V CLAMP
VRO
VIN
Figure 12 — MOSFET Drain Voltage and Snubber Capacitor Voltage
The energy stored in the leakage inductance can be obtained as:
Psn =
1
⋅ Lleakage ⋅ IP 2 ⋅ fs
2
(30)
where IP is the peak current in primary side.
For small leakage inductance application, the leakage energy given in (30) is partly dissipated in this
RCD clamp circuit. For large leakage inductance application, the energy absorbed by the RCD clamp
circuit is much larger than that given in (30) due to the long commutation time between the primary side
and secondary side, thus part of the magnetizing energy is fed to the RCD clamp circuit instead of the
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AN027 –QUASI-RESONANT FLYBACK CONVERTERS
output side. The theoretical analysis is quite complex and will not be elaborated on here. Generally, we
simply assume the energy stored in the leakage inductance is completely dissipated in the RCD clamp
circuit. Typically, we can assume an acceptable clamp voltage VCLAMP which is usually 50~100% higher
than the reflected output voltage VRO. The resistance can be calculated based on (31). In practice, this
resistance can be adjusted based on the power loss and the acceptable clamp voltage, which usually is
slightly higher than that calculated by (31).
VCLAMP = Psn ⋅ R sn
(31)
The snubber capacitor Csn should be selected considering the voltage ripple of VCLAMP given in (32).
Generally, a 5~10% ripple voltage is reasonable.
ΔVsn =
VCLAMP
Csn ⋅ Rsn ⋅ fs
(32)
H. Design the Output Filters
The RMS current of the output capacitor can be obtained as:
Icap−out = Irms−sec 2 − Iout 2
(33)
where Iout is the output current and Irms-sec is the secondary RMS current in (26).
The RMS current should be smaller than the RMS current specification of the capacitor.
The voltage ripple on the output can be estimated by:
ΔVout =
Iout ⋅ (TOFF + Tw )
+ (Isec −P − Iout ) ⋅ RESR
Cout
(34)
where Isec-P is the secondary side peak current; RESR is the ESR of output capacitor; TOFF and Tw can be got
from (11) (12).
Sometimes it is impossible to meet the ripple specification with a single electrolytic cap due to the high ESR.
Then, additional LC filter or extra ceramic capacitor with low ESR parallelled with the electrolytic capacitor
can be used.
4. DESIGN SUMMARY
z
z
z
z
A detailed reference design of quasi-resonant flyback converter is shown in Fig 13. The input
voltage is 90VAC to 265VAC with 24V/1.5A output capability. HFC0100 from MPS is adopted as the
controller, which has valley switching to minimize the switching loss with improved efficiency and
EMI.
OCP function is achieved by the primary sense resistor connected with CS pin of HFC0100, when
the primary current is high enough which caused the CS pin voltage exceeds 1V, the controller will
turn off the switch to limit the current.
By sensing the voltage on auxiliary winding into VSD pin of HFC0100, OVP function is achieved.
When output voltage is too high which make the voltage on auxiliary winding trigger the OVP
threshold of VSD pin, HFC0100 will latch off the switch until VCC voltage restarted.
The transformer used in this design has a turn ratio of 84:14:8 (Ns: Np2:Np1) turn ratio with 820uH
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AN027 –QUASI-RESONANT FLYBACK CONVERTERS
primary inductance. The auxiliary winding is used for powering the controller and detecting the VDS to
ensure DCM and valley switching.
1
3
2
4
AP2764I-A
3
4
2
1
Figure 13 — Schematic of Quasi-Resonant Flyback Converter with HFC0100
5. EXPERIMENTAL VERIFICATION
In order to show the validity of the design procedure presented by this paper, the quasi-resonant flyback
converter of the design example has been built and tested (Input: 90VAC~265VAC; Output: 24V/1.5A).
All the components in the circuit are used as design example in Fig.13.
Fig.14 and Fig.15 shows the valley switching on primary side. At low line input in Fig.10, the switch is
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AN027 –QUASI-RESONANT FLYBACK CONVERTERS
turned on at the first valley point of Drain-Source voltage to minimize the turn-on loss; At high line input,
because the time period between the switch turned off and the 1st valley point on drain-source voltage is
smaller than the minimum-off-time (8us) of the controller, the switch is not turned off until the 2nd valley
point on drain-source voltage is reached.
Fig.16 shows the Burst Mode function of the controller at light load. To minimize the power dissipation in
no load or light load, HFC0100 enters the burst mode operation. As the load decreases, the FB voltage
decreases, the HFC0100 stops the switching cycle when the FB voltage drops below the threshold
VBRUL—0.5V. And the output voltage starts to drop at a rate dependent on the load. This causes the FB
voltage to rise again. Once the FB voltage exceeds the threshold VBRUH—0.7V, switching resumes. The
FB voltage then falls and rises periodically. The burst mode operation alternately enables and disables
switching cycle of the MOSFET thereby reducing switching loss in the no load or light load conditions.
Fig.17 and Fig.18 shows the OCP and OVP function, when the output current or the output voltage
exceeds the setting value, HFC0100 will latch off the switch to protect the whole system.
Fig.19 shows the measured efficiency versus output current. Benefit from valley switching, the efficiency
exceeds 90% at full load condition. Also the power consumption at no load is very low (130mW at low
line; 160mW at high line) because of the burst mode operation by HFC0100.
Figure 14 — Waveform of Valley Switching of Primary at Low Line Input
(CH1: VGS of Q1; CH2: VDS of Q1)
Figure 15 — Waveform of Valley Switching of Primary at High Line Input (CH1: VGS of Q1; CH2:
VDS of Q1)
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AN027 –QUASI-RESONANT FLYBACK CONVERTERS
Figure 16 — The Burst Mode Function of HFC0100 (CH1: VFB; CH2: VDS)
Figure 17 — The OCP Function of HFC0100 (CH1: VOUT; CH4: IOUT)
Figure 18 — The OVP Function of HFC0100 (CH1: VOUT)
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AN027 –QUASI-RESONANT FLYBACK CONVERTERS
Efficiency vs Load
92
Efficiency(%)
90
88
86
84
82
80
0.2
0.4
0.6
0.8
1
1.2
1.4
1.6
Load(A)
Figure 19 — The Efficiency of the System (220VAC Input)
6. REFERENCES
[1]. Lloyd H. Dixon, “Magnetics Design for Switching Power Supplies,” in Unitrode Magnetics Design
Handbook, 1990.
NOTICE: The information in this document is subject to change without notice. Users should warrant and guarantee that third
party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not
assume any legal responsibility for any said applications.
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21