Power Factor Correction Controllers ISL6730A, ISL6730B, ISL6730C, ISL6730D The ISL6730A, ISL6730B, ISL6730C, ISL6730D are active Features power factor correction (PFC) controller ICs that use a boost topology. (ISL6730B, ISL6730C, ISL6730D are Coming Soon.) The controllers are suitable for AC/DC power systems, up to 2kW and over the universal line input. The ISL6730A, ISL6730B, ISL6730C, ISL6730D are operated in continuous current mode. Accurate input current shaping is achieved with a current error amplifier. A patent pending breakthrough negative capacitance technology minimizes zero crossing distortion and reduces the magnetic components size. The small external components result in a low cost design without sacrificing performance. The internally clamped 12.5V gate driver delivers 1.5A peak current to the external power MOSFET. The ISL6730A, ISL6730B, ISL6730C, ISL6730D provide a highly reliable system that is fully protected. Protection features include cycle-by-cycle overcurrent, over power limit, over-temperature, input brownout, output overvoltage and undervoltage protection. • Reduce component size requirements - Enables smaller, thinner AC/DC adapters - Choke and cap size can be reduced by 66% - Lower cost of materials • Excellent power factor over line and load regulation - Internal current compensation - CCM Mode with Patent pending IP for smaller EMI filter • Better light load efficiency - Automatic pulse skipping - Programmable or automatic shutdown • High reliable design - Cycle-by-cycle current limit - Input average power limit - OVP and OTP protection - Input brownout protection The ISL6730A, ISL6730B provide excellent power efficiency and transitions into a power saving skip mode during light load conditions, thus improving efficiency automatically. The ISL6730A, ISL6730B, ISL6730C, ISL6730D can be shut down by pulling the FB pin below 0.5V or grounding the BO pin. The ISL6730C, ISL6730D have no skip mode. • Small 10 Ld MSOP package Two switching frequency options are provided. The ISL6730B, ISL6730D switch at 62kHz, and the ISL6730A, ISL6730C switch at 124kHz. • TV AC/DC power supply • Desktop computer AC/DC adaptor • Laptop computer AC/DC adaptor • AC/DC brick converters 100 VI VLINE Applications + VOUT 95 EFFICIENCY (%) 90 VCC ISEN GATE ICOMP GND ISL6730 VIN FB ISL6730A, SKIP 80 ISL6730C 75 70 COMP BO 85 65 VREG 60 0 20 FIGURE 1. TYPICAL APPLICATION 40 60 OUTPUT POWER (W) 80 100 FIGURE 2. PFC EFFICIENCY TABLE 1. KEY DIFFERENCES IN FAMILY OF ISL6730 February 26, 2013 FN8258.0 VERSION ISL6730A ISL6730B ISL6730C ISL6730D Switching Frequency 124kHz 62kHz 124kHz 62kHz Skip Mode Yes-Fixed Yes-Fixed No No 1 CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 1-888-468-3774 | Copyright Intersil Americas LLC 2013. All Rights Reserved Intersil (and design) is a trademark owned by Intersil Corporation or one of its subsidiaries. All other trademarks mentioned are the property of their respective owners. ISL6730A, ISL6730B, ISL6730C, ISL6730D Pin Configuration ISL6730A, ISL6730B, ISL6730C, ISL6730D (10 LD MSOP) TOP VIEW GND 1 10 GATE ISEN 2 9 VCC ICOMP 3 8 VREG VIN 4 7 FB BO 5 6 COMP Pin Descriptions PIN # I/O SYMBOL DESCRIPTION 1 - GND Ground pin. All voltage levels refer to this pin. 2 I ISEN Current sense pin. The current through this pin is proportional to the inductor current. 3 I/O ICOMP 4 I VIN Input voltage sense. This pin provides the reference voltage to shape inductor current. Connect this pin to a resistor divider from the rectified input voltage. The resistor divider ratio is used to adjust the phase lag between input voltage and the input current. The phase lag is required to compensate the phase lead generated by the EMI filter. 5 I/O BO This pin should be decoupled to GND with a minimum 0.1µF ceramic capacitor. The BO pin is a voltage follower, which will follow the DC voltage of the VIN pin. The BO pin is internally tied to GND through a resistor RIS. The decoupling capacitor provides ripple filtering. When the voltage at the BO pin (VBO) drops below brownout voltage threshold, the controller enters shutdown mode and the gate drive is disabled. The BO pin will be disabled when the FB pin drops below the enabling threshold. 6 I/O COMP Output of the error amplifier. The voltage of the COMP pin sets the input power. During start-up, a small charge current will slowly ramp up the voltage of the COMP pin. 7 I FB Voltage feed back pin. Connect this pin to a resistor divider from the output. The resistor divider sets the output voltage. When the FB pin voltage exceeds 104% of the reference voltage, overvoltage-protection is triggered and gate drive is disabled. When the FB pin is below 10%, the device is put into shutdown mode. There is an internal pull-down current source for open loop protection. 8 - VREG Output of internal regulator. The voltage having a ±2% tolerance over line, load and operating temperature. Bypass to GND with a 47nF low ESR capacitor. VREG can source up to 10mA. This pin is not recommended for usage other than bypass. 9 I VCC Power supply pin. The VCC pin should be decoupled to GND with a minimum 0.1µF ceramic capacitor. 10 O GATE Push-pull gate drive for the external MOSFET. Output voltage is clamped at 12.5V. This pin provides typically 2A sink and 1.5A source capability. Current error amplifier output pin. Ordering Information PART NUMBER (Notes 1, 2, 3) PART MARKING TEMP. RANGE (°C) PACKAGE (Pb-Free) PKG. DWG. # ISL6730AFUZ 6730A -40 to +125 10 Ld MSOP M10.118 ISL6730BFUZ (Coming Soon) 6730B -40 to +125 10 Ld MSOP M10.118 ISL6730CFUZ (Coming Soon) 6730C -40 to +125 10 Ld MSOP M10.118 ISL6730DFUZ (Coming Soon) 6730D -40 to +125 10 Ld MSOP M10.118 NOTES: 1. Add “-T*” suffix for tape and reel. Please refer to TB347 for details on reel specifications. 2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020. 3. For Moisture Sensitivity Level (MSL), please see device information page for ISL6730A. For more information on MSL please see techbrief TB363. 2 FN8258.0 February 26, 2013 Block Diagram VI EMI CHOKE D VOUT L VLINE CF3 CF2 COUT Q1 CF1 Lm RCS 3 CREG VCC CURRENT MIRROR 2:1 RSEN ISEN VREG LINEAR REGULATOR UVLO VCC GATE I OC I > -------------CS 2 CONTROL LOGIC I CS GND OTP PWM COMP SKIP CEQ GEN. ICOMP V IREF Gmi CS R ×I IS ISEN = ----------------------------------2 RIS = 16kΩ OSCILLATOR 0.25 × VIN -----------------------------COMPB 2 BO RIN2 SOFT-START ENABLE 2.5V VIN COMPB SKIP CLAMP COMP-1V FB OVER POWER LIMIT RFB2 Gmv 20µA RIN1 IFB SKIP COMP BO FN8258.0 February 26, 2013 CBO RFB1 ISL6730A, ISL6730B, ISL6730C, ISL6730D DF1 DF2 Application Schematics Typical 300W Application Schematic D1 2 3 D8 S1M-13-F VCC R8 4.3M 3.16k R9 R11 4.3M TP6 2 4 TP3 R13 65k 3 CR12 47n U1 GND VIN ISL6730B TP4 C14 470n FB R18 TP2 82.5k C18 1.5u 1 P7 P6 GATE ISEN C13 220p C9 1u VCC ICOMP 5 100p C10 18n TP5 C12 C11 1.2n R14 4.02k COMP D7 S1M-13-F GND TP10 2 220n 3.3V DZ1 C8 1 6 470p 470p P3 51k R5 0.068 C6 P2 270u C1 R10 3.3M R4 9 PE R2 2.2 VCC C5 P5 D5 R3 2M C3 470n BO AC2 3 2 P4 D6 TP9 3 C2 IPA60R600C6 Q1 1 C4 470n DC+ C3D0406 0E R6 3.3M 2 S3KB-TP S3KB-TP 100n D2 D4 8 L3 R1 2M REG 85~265Vac S3KB-TP D3 F1 4 AC1 L2 0.6m 390V 1 2 L1 100u S3KB-TP 1 4 P1 C15 100n 10 1 7 TP1 R17 1.5k R19 40k C16 1n ISL6730A, ISL6730B, ISL6730C, ISL6730D S3KB-TP FN8258.0 February 26, 2013 Application Schematics (Continued) Typical 85W Application Schematic D1 2 S3KB-TP 3 AC2 C5 P5 PE D5 R3 2M C3 220n D6 R2 2.2 470p 470p D8 D7 S1M-13-F 220n 470n C8 1 S1M-13-F 2 4 9 VIN C13 220p 5 R13 65k GND ISEN ISL6730A FB R18 TP4 R15 DNP 3 C14 470n 1 PFC_EN R20 10k P9 C17 1n Q2 2N7002 2 P8 P7 P6 GATE ICOMP BO TP3 100p 3 VCC COMP 68n C9 1u 10 1 7 TP1 R17 1.5k 6 TP6 TP5 C12 8 C10 VCC R11 4.3M CR12 47n U1 REG 2.1k R9 4.3M C11 470p R14 6.2k 1 GND TP10 2 3.3V DZ1 VCC R8 P3 51k R5 0.22 C7 P2 56u C1 R10 3.3M R4 C6 TP9 R6 3.3M IPA60R600C6 Q1 1 C4 330n DC+ C3D0406 0E 2 7.5m 3 2 P4 D2 D4 C2 100n L2 2.2m 2 R1 2M L1 128u 3 L3 S3KB-TP S3KB-TP S3KB-TP D3 S3KB-TP 68k C18 2.2u TP2 C15 100n R16 DNP R19 40k C16 1n ISL6730A, ISL6730B, ISL6730C, ISL6730D 5 85~265Vac F1 1 AC1 4 P1 390V 1 FN8258.0 February 26, 2013 ISL6730A, ISL6730B, ISL6730C, ISL6730D Absolute Maximum Ratings Thermal Information VCC to GND. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +28V GATE to GND. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +18V VIN, BO, ISEN, FB and COMP to GND. . . . . . . . . . . . . . . . . . . . . . . .-0.3V to +6.3V ESD Rating Human Body Model (Tested per JESD22-A114) . . . . . . . . . . . . . . .2.5kV Machine Model (Tested per JESD22-A115). . . . . . . . . . . . . . . . . . . . 200V Charged Device Model (Tested per JESD-C101E. . . . . . . . . . . . . . . . . 1kV Latch Up (Tested per JESD-78B; Class 2, Level A) . . . . . . . . . . . . . . 100mA Thermal Resistance (Typical) θJA (°C/W) θJC (°C/W) MSOP Package (Notes 4, 5) . . . . . . . . . . . . 137 39 Maximum Junction Temperature (Plastic Package) . . . . . . . . . . . .+150°C Maximum Storage Temperature Range . . . . . . . . . . . . . .-65°C to +150°C Ambient Temperature Range . . . . . . . . . . . . . . . . . . . . . . .-40°C to +125°C Junction Temperature Range . . . . . . . . . . . . . . . . . . . . . . .-40°C to +125°C Pb-Free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . see link below http://www.intersil.com/pbfree/Pb-FreeReflow.asp Recommended Operating Conditions VCC to GND. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15V to + 20V Ambient Temperature Range . . . . . . . . . . . . . . . . . . . . . . .-40°C to +125°C CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and result in failures not covered by warranty. NOTES: 4. θJA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details. 5. For θJC, the "case temp" location is taken at the package top center. Electrical Specifications -40°C to +125°C. PARAMETER Operating Conditions: VCC = 15V, TA = +25°C. Boldface limits apply over the operating temperature range, SYMBOL TEST CONDITIONS MIN (Note 8) TYP MAX (Note 8) UNITS VCC SUPPLY CURRENT Start Up Current ISTART VFB = 1V, VCC < VCC(ON) 73 106 139 µA Standby Current ISTDN VFB = GND, VCC > VCC(ON) 179 237 295 µA VFB = 2.5V, COMP = SKIP*0.25 +1V 580 690 800 µA GATE is floating 3.2 3.7 4.2 µA Skip Mode Current ICCSKIP Operating Current (Note 6) ICC VCC UVLO UVLO Rising Threshold VCC(ON) 9 10 11 V UVLO Falling Threshold VCC(OFF) 6.7 7.5 8.3 V UVLO Threshold Hysteresis VCC(HYS) - 2.5 - V 5.16 5.4 5.6 V 30 50 70 mA REGULATOR VOLTAGE VREG Overall Accuracy IREF = 0 to -10mA, VCC = 15V, Load Capacitor = 47nF Current Limit PWM CONVERTERS Minimum Duty Cycle, ISL6730A fSW = 124kHz - 2.57 4.66 % Maximum Duty Cycle, ISL6730A fSW = 124kHz 94.8 96.5 - % OSCILLATOR Free Running Frequency, ISL6730A TA = -40°C to +125°C, VIN = 0.6V 98 107 116 kHz Free Running Frequency, ISL6730A TA = -40°C to +125°C, VIN = 2.5V 114 125 136 kHz 1.33 1.46 1.59 V - 2.33 4.46 Ω 0.15 0.3 0.45 V - 1.5 - A PWM Ramp Amplitude Vm GATE DRIVER Gate Drive Pull-Down Resistance VCC = 15V, IGATE = 15mA Gate Drive Pull-Up Voltage Drop VCC = 9V, IGATE = 15mA Gate Drive Max. Sourcing/Sinking Current 6 FN8258.0 February 26, 2013 ISL6730A, ISL6730B, ISL6730C, ISL6730D Electrical Specifications -40°C to +125°C. (Continued) PARAMETER Operating Conditions: VCC = 15V, TA = +25°C. Boldface limits apply over the operating temperature range, SYMBOL TEST CONDITIONS MIN (Note 8) TYP MAX (Note 8) UNITS Rise Time CO = 2.2nF, VCC = 15V, Gate Voltage Rise Time from 10% to 90% of VGC - 34 62 ns Fall Time CO = 2.2nF, VCC = 15V, Gate Voltage Fall Time from 10% to 90% of VGC - 34 57 ns VGC 11 12 13 V VREF 2.48 2.5 2.52 V IFB - 65 - nA Rising Threshold to Enable Converter FB_EN 280 300 320 mV Falling Threshold to Disable Converter FB_DIS 190 202 214 mV Enable Hysteresis FB_Hys - 100 - mV - 92 - dB 50 77 104 µA/V - 13 - µA Gate Clamp Voltage VOLTAGE REFERENCE Reference Voltage Feedback Pin Pull-Down Current VOLTAGE ERROR AMPLIFIER DC Gain Error Amp Transconductance gmv ISource/Sink COMP Offset Voltage VCOMP_OFF 0.95 1.01 1.07 V COMP Soft-Start Enable Voltage VCOMP_EN 0.58 0.64 0.75 V - 9 - nA 0.196 0.25 0.296 V/V INPUT VOLTAGE SENSING VIN Leakage Current MULTIPLIER GAIN GMUL COMP = 2.5V, VIN = 1.0V, BO = 1.0V, ISEN = 50µA CURRENT ERROR AMPLIFIER Current DC Gain AiDC ΔIICOMP/ΔIISEN 1.6 1.9 2.2 A/A Error Amp Transconductance gmi IICOMP = ±20µA 205 268 331 µA/V ICOMP Source/Sink Current (Note 7) - 60 - µA Current Sensing Input Offset -3 2 7 mV VSCMT 1.32 1.36 1.4 V COMP Upper Limit VCUL 3.53 3.85 4.17 V COMP Valid Range VCUL-1V 2.5 2.83 3.16 V FB Exit Threshold Voltage VFB_EXIT IISEN = 0µA 87 88 89 % ISEN Exit Threshold Current ISEN_EXIT VFB = 2.5V -38 -29 -20 µA LIGHT LOAD EFFICIENCY ENHANCEMENT AND OVERPOWER PROTECTION Skip Mode COMP Threshold BROWNOUT DETECTION Brownout Rising Threshold VBO_R 478 494 510 mV Brownout Falling Threshold VBO_F 387 401 415 V 102.9 104.1 105.3 V -197 -177 -159 µA OVERVOLTAGE PROTECTION Overvoltage Protection VOVP Fraction of the set point; ~1µs noise filter OVERCURRENT PROTECTION Overcurrent Threshold IOC 7 FN8258.0 February 26, 2013 ISL6730A, ISL6730B, ISL6730C, ISL6730D Electrical Specifications -40°C to +125°C. (Continued) Operating Conditions: VCC = 15V, TA = +25°C. Boldface limits apply over the operating temperature range, MIN (Note 8) TYP MAX (Note 8) UNITS Shutdown Temperature (Note 7) - 160 - °C Thermal Shutdown Hysteresis (Note 7) - 25 - °C PARAMETER SYMBOL TEST CONDITIONS THERMAL SHUTDOWN NOTES: 6. This is the VCC current consumed when the device is active but not switching. Does not include gate drive current. 7. Limits should be considered typical and are not production tested. 8. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. 100.50 101.0 100.25 100.5 FSW NORMALIZED (%) VFB NORMALIZED (%) Typical Performance Curves 100.00 99.75 VIN = 2.5V 100.0 99.5 VIN = 0.6V 99.50 -40 -20 0 20 40 60 80 TEMPERATURE (°C) 100 120 99.0 -40 140 FIGURE 3. FEEDBACK ACCURACY -20 0 20 40 60 80 TEMPERATURE (°C) 100 120 140 FIGURE 4. FSW vs TEMPERATURE, VCC = 15V 105 100 FSW NORMALIZED (%) AIDC NORMALIZED (%) 101 100 99 98 95 90 85 80 97 75 -40 -20 0 20 40 60 80 TEMPERATURE (°C) FIGURE 5. AIDC vs TEMPERATURE 8 100 120 140 0 0.5 1.0 1.5 2.0 2.5 3.0 VIN (V) FIGURE 6. FSW vs VIN, TA = +25°C FN8258.0 February 26, 2013 ISL6730A, ISL6730B, ISL6730C, ISL6730D Typical Performance Curves (Continued) VCC CURRENT NORMALIZED (%) 101 HYSTERSIS 100 UP THRESHOLD 99 DOWN THRESHOLD 98 -40 -20 0 20 40 60 80 100 120 102 101 100 ICC (GATE FLOATING) 99 98 -40 140 ISTART -20 0 20 TEMPERATURE (°C) 40 60 80 100 120 140 TEMPERATURE (°C) FIGURE 7. UVLO THRESHOLDS vs TEMPERATURE FIGURE 8. VCC SUPPLY CURRENT vs TEMPERATURE 112 DRIVER TIME NORMALIZED (%) UVLO THRESHOLD NORMALIZED (%) 102 110 108 FALL TIME 106 104 102 RISE TIME 100 98 96 -40 -20 0 20 40 60 80 100 120 140 TEMPERATURE (°C) FIGURE 9. GATE DRIVER ABILITY vs TEMPERATURE (LOAD = 2.2nF) 9 FN8258.0 February 26, 2013 ISL6730A, ISL6730B, ISL6730C, ISL6730D Functional Description EMI CHOKE VCC Undervoltage Lockout (UVLO) VLINE The ISL6730A, ISL6730B, ISL6730C, ISL6730D start automatically once the voltage at VCC exceeds the UVLO threshold. CF3 CF2 Lm DF1 Shutdown DF2 When the VFB pin is below 0.2V, the controller is disabled and the PWM output driver is tri-stated. When disabled, the IC power will be reduced. During shutdown, the COMP pin is discharged to GND and the controller is disabled. The Over-Temperature Protection (OTP) is still alive to prevent the controller from starting up in a high temperature ambient condition. RIN2 VIN BO RIN1 CBO In the event that the FB pin is disconnected from the feedback resistors, the FB pin is pulled to ground by an internal current source IFB. When the FB pin voltage drops below 0.2V, the gate driver is disabled. The ISL6730A, ISL6730B, ISL6730C, ISL6730D enters shutdown mode. The BO pin also utilizes the VIN resistor divider for voltage sensing. Set the resistor divider ratio to satisfy the brownout requirement. Soft-Start First, calculate the resistor divider ratio, KBO. The COMP pin is released once the soft-start operation begins. A 13µA current sources out to the RC network connected from the COMP pin until the FB pin voltage reaches 90% of the reference voltage. V BORMAX K BO = ------------------------------------------V RMSmin – 2V F Switching is inhibited when the COMP pin voltage is below 1V. When the COMP pin reaches 1V, the current error amplifier and the gate driver are activated and the converter starts switching. During UVLO, Brownout and Shutdown, the COMP is pulled to the ground. Input Voltage Sensing The VIN pin is needed to sense the rectified input voltage. The sensed semi-sinusoidal waveform is needed to shape inductor current, which helps achieves unity power factor. At the same time, the voltage on the VIN pin is used to generate the negative capacitive element at the input. This will cancel the input filter capacitor, CF. Canceling the effect of CF will increase the displacement power factor and alleviate the zero crossing distortion, which is related to the distortion power factor. FIGURE 10. INPUT VOLTAGE SENSING SCHEMATIC (EQ. 1) Where VF is the forward voltage drop of the bridge rectifier and the voltage drop of DF1; DF2. Then, select the RIN2 based on the highest reasonable resistance value. Then select the RIN1 based upon the desirable minimum RMS value of the line voltage for the PFC operation. K BO R IN1 = --------------------- ⋅ R IN2 1 – K BO (EQ. 2) Inductor Current Sensing The current sensing of the converter has two purposes. One is to force the inductor current to track the input semi-sinusoidal waveform. The other purpose is for overcurrent protection. Refer to Figure 11 for the current sensing scheme. The sensed current ICS is in proportion to the inductor current, IL as described in Equation 3. 1 R CS I CS = --- ⋅ ---------------- ⋅ I L 2 R SEN (EQ. 3) where: RCS is the current sensing resistor with low value in the return path to the bridge rectifier. RSEN is the current scaling resistor connected between ISEN to the RCS. 10 FN8258.0 February 26, 2013 ISL6730A, ISL6730B, ISL6730C, ISL6730D VI EMI CHOKE VOUT L Q1 COUT VLINE BRIDGE RECFIFIER LF CF3 CF1 CF1 CF2 Lm RCS FIGURE 12. TYPICAL PFC INPUT FILTER CIRCUIT CURRENT MIRROR 2:1 RSEN I BRIDGE RECFIFIER EMI CHOKE LF CS VLINE CF3 ISEN CF1 CF2 Lm I CS > 0.5 I OC FIGURE 11. INDUCTOR CURRENT SENSING SCHEME A high value RCS renders more accurate current sensing. It is recommended to use the RCS to render 120mV peak voltage at the maximum line voltage during full load condition. 120mV ⋅ V RMSMAX ⋅ η R CS > ------------------------------------------------------------2 ⋅ P Omax FIGURE 13. LOW COST PFC INPUT FILTER CIRCUIT For applications where the output power is above 500W, the negative capacitance helps to improve the power factor dramatically. Please refer to Table 2 for the recommended filtering capacitor to be placed after the bridge rectifier, CF1. TABLE 2. (EQ. 4) Where η is the efficiency of the converter at the maximum line input with full load. Since the RCS sees the average input current, high value RCS generates high power dissipation on the RCS. Use a reasonable RCS according to the resistor power rating. The worst-case power dissipation occurs at the input low line when input current is at its maximum. Power dissipation by the resistor is: (EQ. 5) P RCS = ( I RMSMAX ) 2 ⋅ R CS where: IRMSMAX is the maximum input RMS current at the minimum input line voltage, VRMSmin. Select the RSEN according to the peak current limit requirement. The resistor is sized for an overload current 25% more than the peak inductor peak current. Negative Input Capacitor Generation (Patent Pending) The patent pending negative capacitor generation capability of the ISL6730A, ISL6730B, ISL6730C, ISL6730D allows the capacitor CF2 to be moved from before the bridge rectifier (Figure 12) to after the bridge rectifier (Figure 13). Thus, a smaller lower cost CF2 can be used. The change in topology reduces the size of the EMI filter. Furthermore, CF1 can be increased thus decreasing the size of LF (Figure 13). CF1 Po < 100W Typical C(µF)/100W 0.68 100W < Po < 500W Po > 500W 0.33 0.22 Additional CF1 may be used to accommodate the use of small boost inductor or to eliminate the differential mode filter inductor as long as the equipment meets the power factor or goal. The equivalent negative capacitor is a function of the input voltage divider ratio, KBO, the current sensing gain and current compensation error integration gain. Adjusting the negative Ceq can be achieved by adjusting the current compensation network. Frequency Modulation The ISL6730A, ISL6730B, ISL6730C, ISL6730D can further reduce EMI filter size by lowering the differential noise power density. The reduction is achieved by switching frequency modulation. The frequency varies with the VIN pin. The switching frequency reaches the peak value when the VIN pin voltage is 2V as shown in Figure 6. The peak value of ISL6730A/C is 124kHz, and the ISL6730B/D is 62kHz. Output Voltage Regulation The output voltage is sensed through a resistor divider. The middle point of the resistor divider is fed to the FB pin. The resistor divider ratio sets the output voltage. The transconductance error amplifier generates a current in proportion to the difference between the FB pin and the 2.5V internal reference. The PFC is stabilized by the compensation network that is connected from the COMP pin to the ground. The voltage of the COMP sets the input average power by determining the amplitude of the current reference. To keep the 11 FN8258.0 February 26, 2013 ISL6730A, ISL6730B, ISL6730C, ISL6730D harmonic distortion minimum, it is desirable to set the control bandwidth much lower than twice of the line frequency. The recommended voltage loop bandwidth is 10Hz. During start-up, the compensation capacitors and the charging current from the error amplifier sets the input power increase rate. Thus, soft-start is achieved. The COMP is discharged during shutdown and fault conditions. Light Load Efficiency Enhancement For PC, adaptor and TV applications, it is desirable to achieve high efficiency at light load conditions and low standby current. The ISL6730A, ISL6730B, ISL6730C, ISL6730D can enter light load efficiency mode automatically. The voltage error amplifier output, COMP, is an indicator of the average input power level. The controller compares the V(COMP) and V(SKIP). If V(COMP)-1V is less than V(SKIP)*0.25, the PFC controller stops gate switching and the COMP pin voltage is clamped to V(SKIP)+0.6V. ISL6730A/C use a fixed V(SKIP), which is 1.4V; for ISL6730B/D, the SKIP function are disabled. The controller exits skip mode when VFB drops to 88% (typical) of the reference voltage or when the sensed returned current exceeds 29µA. Protection Circuits Input Brownout, BO Protection Brownout occurs when there is a drop in the line voltage. The BO pin is a dual function pin. The BO pin detects the brownout condition and shuts down the gate driver and controller. During normal operation, the BO pin is used to compensate the effect of the input line voltage change on the voltage loop. To keep the harmonic distortion low, the corner frequency formed by the RBO and CBO should be lower than 6Hz. The BO pin is the output of the average voltage of the rectified voltage. The PFC controller is turned off when the BO pin drops below 0.4V. This protects the PFC power stage to enable operation at or below brownout condition for long periods of time. The controller resumes operation when the BO pin returns to 0.5V. The BO pin is usually connected to GND through a capacitor, CBO. To avoid distortion on the VIN pin, select CBO so that: Overvoltage Protection If the voltage on the FB pin exceeds the reference voltage by about 4%, the gate driver is turned off. The controller resumes normal operation after the FB pin drops below reference voltage. Over-Temperature Protection The ISL6730A, ISL6730B, ISL6730C, ISL6730D is protected against over-temperature conditions. When the junction temperature exceeds +160°C, the PWM shuts down. Normal operation is resumed when the junction temperature decreases below +135°C. Application Guidelines Layout Considerations As in any high frequency switching converter, layout is very important. Switching current from one power device to another can generate voltage transients across the impedances of the interconnecting bond wires and circuit traces. These interconnecting impedances should be minimized by using wide, short printed circuit traces. The critical components should be located as close together as possible using ground plane construction or single point grounding. Figure 14 shows the critical power components; Q1, D and COUT. To minimize the voltage overshoot, the interconnecting wires indicated by heavy lines should be part of the ground or the power plane in a printed circuit board. The components shown in Figure 14 should be located as close together as possible. Please note that the capacitors CVCC and CO each represent numerous physical capacitors. Locate the ISL6730A, ISL6730B, ISL6730C, ISL6730D within 2 inches of the MOSFET, Q1. The circuit traces for the MOSFETs’ gate and source connections from the ISL6730A, ISL6730B, ISL6730C, ISL6730D must be sized to handle up to 1.5A peak current. D L Q1 COUT GATE (EQ. 6) C BO » 0.22μF VCC Overcurrent Protection The peak current limiting function prevents the inductor from saturation. The gate driver turns off when the current goes above the current limit. Overpower Protection The overpower protection is implemented by limiting the COMP pin voltage higher than 3.85V (typical). 12 CVCC FIGURE 14. CRITICAL CURRENT POWER COMPONENTS Component Selection Guidelines A 300W, universal input, PFC converter design is provided for demonstration. The design method is for a continuous current mode power factor correction boost converter with the ISL6730B/D. The switching frequency is 62kHz. FN8258.0 February 26, 2013 ISL6730A, ISL6730B, ISL6730C, ISL6730D Table 3 shows the design parameters. MIN TYP MAX UNIT Select the bridge diode using Equation 15 and sufficient reverse breakdown voltage. Assuming the forward voltage, VF,BR, is 1V across each rectifier diode. The power loss of the rectifier bridge can be calculated: VLINE 85 115 265 VAC P BR = 2 • V F, BR • I INAVE ( MAX ) (EQ. 15) FLINE 47 63 Hz 300 W P BR = 2 • 1V • 3.5A = 7W (EQ. 16) ms INPUT CAPACITOR SELECTION % Refer to Table 2 for the recommended input filter capacitor value. TABLE 3. CONVERTER DESIGN PARAMETERS PARAMETER CONDITIONS POMAX Maximum Output Power THOLD Hold Up Time Efficiency VLINE = 115VAC 20 92 0.33 C IN = 300W • ----------- = 0.99μF 100 BOOST INDUCTOR SELECTION First, calculate the maximum input RMS current, IINMAX. P OMAX I INMAX = ----------------------------------η • V RMSmin (EQ. 7 Where η is the converter efficiency at VRMSmin. PF is the power factor at VRMSmin. 300W I INMAX = ---------------------------- = 3.84A 0.92 • 85V (EQ. 8) Assuming the current is sinusoidal and the peak to peak ripple at line is 40%. The boost inductor, LBST, is given by the following equation: 2 • V RMSmin⎞ 2V RMSmin ⎛ L BST ≥ ---------------------------------------------------------------- • ⎜ 1 – ---------------------------------------⎟ V OUT 0.4 • F sw • 2 • I INMAX ⎝ ⎠ 85V 2 • 85V L BST ≥ ------------------------------------------------------ • ⎛ 1 – ------------------------⎞ = 617μH 0.4 • 62kHz • 3.88 A ⎝ 390V ⎠ (EQ. 9) (EQ. 17) This is the recommended capacitor used after the diode bridge. For better power factor, less capacitance can be used. To lower the input filter inductor size, more capacitance can be used. Two 0.47µF capacitors in parallel are used for CIN. BOOST DIODE SELECTION The boost diode loss is determined by the diode forward voltage drop, VF and the output average current. The maximum output current is: P OMAX I OUT ( max ) = -------------------V OUT (EQ. 18) 300W I OUT ( max ) = ---------------- = 0.77A 390V (EQ. 19) The forward power loss on the diode is: P FD = I OUT ( max ) • V F (EQ. 20) P FD = 0.77A • 1.85V = 1.42W (EQ. 21) (EQ. 10) The peak current of the inductor is the sum of the average peak inductor current and half of the peak to peak ripple current. Select and design the boost inductor as given by Equation 11. The ISL6730A, ISL6730B, ISL6730C, ISL6730D provides peak current limit function that can prevent the boost inductor saturation. Assuming 25% margin is given to the OCP threshold, select and design the boost inductor with saturation current given by Equation 11 with 25% more. I LPeak = 0.4 2 • I INMAX • ⎛ 1 + --------⎞ ⎝ 2 ⎠ (EQ. 11) I LPeak = 0.4 2 • 3.88A • ⎛ 1 + --------⎞ = 6.5A ⎝ 2 ⎠ (EQ. 12) The IDD03E60 part is selected. The reverse recovery loss on the diode can be calculated. The QRR is found from the diode datasheet. QRR = 220nC when IF = 3.5A. The reverse recover loss on the diode can be estimated: 1 P RRD = --- • Q • V OUT • F 4 RR sw (EQ. 22) 1 P RRD = --- • 220nC • 390V • 62kHz = 1.33W 4 (EQ. 23) The total power loss on the diode is: (EQ. 24) P D = P FD + P RRD = ( 1.42 + 1.35 )W = 2.75W INPUT RECTIFIER The maximum average input current is calculated: MOSFET POWER DISSIPATION 2 • 2 • I INMAX I INAVE ( max ) = -----------------------------------------π (EQ. 13) 2 • 2 • 3.88A I INAVE ( max ) = -------------------------------------- = 3.5A π (EQ. 14) The power dissipation on the MOSFET is from two different types of losses; the condition loss and the switching loss. For the MOSFET, the worst case is at minimum line input voltage. First, the drain to source RMS current is calculated: 8 2 V RMSmin I DS ( max ) = I INMAX 1 – ----------- • -------------------------V 3π (EQ. 25) 8 2 85V I DS ( max ) = 3.88A 1 – ----------- • -------------- = 3.3A 3π 390V (EQ. 26) OUT 13 FN8258.0 February 26, 2013 ISL6730A, ISL6730B, ISL6730C, ISL6730D The MOSFET, SPP20N60C3 is selected. (EQ. 27) 2 P COND = I DS ( max ) • R DS ( on ) 2 P COND = 3.3A • 0.3Ω = 3.27W When RG = 3.6Ω, ID = 6A, EON = 0.015mJ, EOFF = 0.007mJ. The switching loss due to transition is calculated: P SW = ( E ON + E OFF ) • F sw (EQ. 29) P SW = ( 0.015mJ + 0.007mJ ) • 62kHz = 1.36W (EQ. 30) The diode reverse recovery incurs additional power loss on the MOSFET. This loss can be estimated as: (EQ. 31) This loss is also related the di/dt during the MOSFET turn-on. The di/dt can be found out from the MOSFET datasheet. At RG = 3.6Ω, the turn-on di/dt is 4000A/µs. From the Typical Reverse Recovery Charge curve at TJ = +125°C, the QRR = 220nC when IF = 3.5A. (EQ. 32) RR = 3.27W + 1.36W + 5.32W = 9.95W CURRENT SENSING RESISTORS Please refer to Equation 4 for calculation of the current sensing resistor RCS. 120mV ⋅ 265V ⋅ 0.92 R CS ≥ ------------------------------------------------------- = 0.069Ω 2 ⋅ 300W (EQ. 40) While a large RCS renders better current sensing accuracy, larger RCS also incurs higher power dissipation. Select RCS from available standard value resistors to determine the sense resistor. (EQ. 41) R CS = 0.068Ω The maximum power dissipation on the RCS occurs at low line and full load condition. The maximum power dissipation is calculated: P RCSMAX = 3.88A • 0.068Ω = 1.023W (EQ. 42) (EQ. 43) The resistor, RSEN sets the overcurrent protection limit. From Equation 3, RSEN should be greater than: R CS • I LPeak • ( 1 + 0.25 ) R SEN ≥ -------------------------------------------------------------------2 • 0.5 I OC (EQ. 44) Where |x| stands for the ABS(x) function. (EQ. 34) 0.068Ω • 6.6A • 1.25 R SEN ≥ -------------------------------------------------------- = 3.117kΩ 2 • 0.18mA (EQ. 35) Calculate the ripple RMS current through the capacitor: (EQ. 45) Select RSEN from available standard value resistors, the selected RSEN is 3.16kΩ. CURRENT LOOP COMPENSATION V OUT 8 2 I CORMS ( max ) = I OUT ( max ) ----------- • -------------------------- – 1 3π V RMSmin (EQ. 36) 8 2 390V I CORMS ( max ) = 0.77A ----------- • -------------- – 1 = 1.633A 3π 85V (EQ. 37) Select the proper capacitor according to the hold time and ripple RMS current requirement. The actual capacitance is 270µF. It is important to make sure the output peak-to-peak ripple is less than the minimum OVP threshold as specified in the “Electrical Specifications” table on page 6. The ESR at 2 times of the line frequency of the capacitor is found in the capacitor datasheet. The ESR of the output capacitor is 770mΩ at 100Hz. 14 The minimum OVP threshold is 103% of the nominal output value. The maximum output peak to peak ripple should be less than 6% of the nominal value, which is 23.4VP-P. 2 (EQ. 33) The output capacitor, COUT, is required to hold the output above 300V during one line cycle. For capacitors with 20% tolerance, the tolerance should be taken into consideration. Thus, the output capacitance should be greater than: 2 ⋅ 20ms ⋅ 300W C OUT ≥ ---------------------------------------------- ⋅ ( 1 + 0.2 ) = 232μF 2 ( 390 ) – ( 300V ) 2 (EQ. 39) 2 OUTPUT CAPACITOR SELECTION 2 ⋅ T HOLD ⋅ P OMAX C OUT ≥ ---------------------------------------------------- ⋅ ( 1 + 0.2 ) 2 2 V OUT – V HOLD ( 2π ⋅ 100Hz ⋅ 270μF ⋅ 0.77Ω ) + 1 V Opp = 0.77A ⋅ ----------------------------------------------------------------------------------------------- = 6.6V ( 2π ⋅ 100Hz ) ⋅ 270μF ⋅ 0.8 P RCSMAX = I INMAX • R CS THE TOTAL LOSS ON THE MOSFET P COND + P SW + P (EQ. 38) OUT 2 From the MOSFET datasheet, the typical switching losses curves are provided. P RR = 220nC • 390V • 62 kHz = 5.32W line (EQ. 28) The switching loss of the MOSFET consists of three parts: the turn-on loss, the turn-off loss and the diode reverse recovery loss. P RR = Q RR • V OUT • F sw 2 ( 2πf line ⋅ C OUT ⋅ ESR ) + 1 V Opp = I OUT ( max ) ⋅ ------------------------------------------------------------------------------( 2πf )⋅C ⋅ 0.8 The input current shaping is achieved by comparing the sensed current signal to the sensed input voltage signal. The current error amplifier (Gmi), together with the current compensation network, adjusts the duty cycle so that the inductor current traces the sensed rectified voltage. Thus, unity power factor is achieved. The compensation network consists of the Trans-Conductance error amplifier (Gmi) and the impedance network (ZICOMP). The goal of the compensation network is to provide a closed loop transfer function with the sufficient 0dB crossing frequency (f0dB) and adequate phase margin. Phase margin is the difference between the open loop phase at f0dB and 180°. The following equations relate the compensation network’s poles, zeros and gain to the components (Ric, Cic and Cip) in Figure 15. FN8258.0 February 26, 2013 ISL6730A, ISL6730B, ISL6730C, ISL6730D VI The cross over frequency of the current loop should be set between 2kHz to 100kHz. At cross over frequency, the transfer function from duty cycle to inductor current is well approximated by Equation 48: VOUT L Q1 COUT CF1 V OUT G id ( s ) = ---------------------L BST ⋅ s RCS CURRENT MIRROR I 2:1 RSEN CS ICOMP IREF RIS Gmi Ric 2 ⎛⎛ A iDC R CS ⎞ V OUT 1 + ( fc ⁄ fz ) ⎞ C ip + C ic = ⎜ ⎜ --------------------------------------- ⋅ ------------- ⋅ ----------------⎟ ⋅ ------------------------------⎟ ⎜⎝ 2 V 2⎟ R SEN⎠ + ( ⁄ f ) 1 f m L ⋅ ( 2πf ) c p ⎠ ⎝ BST c FROM DUTY TO INDUCTOR CURRENT GAIN (dB) COMPENSATION GAIN FP FZ -20 -80 -100 C ip + C ic = ( 19.8 )nF (EQ. 52) fz C ip = ( C ip + C iC ) ---f (EQ. 53) The value of the noise filtering capacitor is: 2.12kHz C ip = 14.9nF ⋅ ----------------------- = 1.35nF 31kHz CURRENT GAIN MODULATOR GAIN 100 1k 10k (EQ. 54) The value of Cic is: 100k (EQ. 55) C ic = 19.8nF – 1.35nF = 18.4nF FREQUENCY (Hz) FIGURE 16. ASYMPTOTIC BODE PLOT OF CURRENT LOOP GAIN 1 F Z = -----------------------------------2π • R ic • C (EQ. 51) p OPEN LOOP GAIN -40 (EQ. 50) The total compensation capacitance is calculated: 40 -60 Where FC = FS/6 = 10.3kHz, ΦM is the phase margin, which is 60°. FP = FS/2 = 31kHz. ( 62KHz ) ⁄ 6 F Z = ------------------------------------------------------------- = 2.12kHz 2 ⎛ tan atan ⎛ ---⎞ + 60deg⎞ ⎝ ⎝ 6⎠ ⎠ 80 0 (EQ. 49) Thus, the current loop compensation zero is: Cip FIGURE 15. INDUCTOR CURRENT SENSING SCHEME 20 It is recommended to set the cross over frequency from 1/10 to 1/6 of the switching frequency with phase margin of 60°. A high frequency pole is set at 1/2 of the switching frequency for ripple filtering. In this example, we set the cross over, FC at 1/6 of the switching frequency. FC F Z = -------------------------------------------------------⎛ ⎞ ⎛ F C⎞ tan ⎜ atan ⎜ -------⎟ + Φ M⎟ ⎝ ⎠ ⎝ F P⎠ ISEN Cic (EQ. 48) (EQ. 46) ic The value of Ric is: 1 R ic = ----------------------------------------------------------- = 4.11kΩ 2π ⋅ 2.12kHz ⋅ 18.4nF (EQ. 56) Select the RC value from the standard value, we have: 1 F P = --------------------------------------------------C ip • C ic 2π • R ic • -----------------------C ip + C ic (EQ. 47) Ric = 4.02kΩ, Cic = 18nF, Cip = 1.2nF. Figure 17 shows the actual bode plot of current loop gain. Use the following guidelines for locating the poles and zeros of the compensation network. 15 FN8258.0 February 26, 2013 ISL6730A, ISL6730B, ISL6730C, ISL6730D 80 For example, CF1 = 0.68µF, CIN = 0.94µF, using the low cost EMI filter shown in Figure 13. When VLINE = 230VAC, fLINE = 50Hz, PO = 60W. 10.5kHz GAIN (dB) 60 40 Assuming 95% efficiency under the above test condition, the resistive component, which is in phase to voltage: 20 Po I a = --------------------------------- = 0.275A V LINE ⋅ 0.95 0 -20 The reactive current through the input capacitors: PHASE (°) 180 10.5kHz (EQ. 63) I c = V LINE • ( 2π ⋅ f LINE ) • ( C F1 + C IN ) = 0.117A 135 Thus, the displacement power factor is: 90 Ia PF DIS = ----------------------------------- = 0.92 2 2 ( Ia ) + ( Ic ) 60 45 45 0 10 (EQ. 62) 1x103 100 1x103 1x103 FREQUENCY (Hz) FIGURE 17. BODE PLOT OF THE ACTUAL CURRENT LOOP GAIN (EQ. 64) The reactive current generated by the equivalent negative capacitor is: (EQ. 65) I cneg = V LINE • ( 2π ⋅ f LINE ) • ( C NEG ) = 0.045A INPUT VOLTAGE SETTING First, set the BO resistor divider gain, KBO according to Equation 1 and Equation 2. With the equivalent negative capacitor, the total reactive current reduces to: Assuming the converter starts at VLINE = 80VRMS, then the BO resistor divider gain, KBO should be: I c – I cneg = 0.072A (EQ. 57) 0.5V K BO = ------------------------ = 0.00641 80V – 2V In this design, two 3.3MΩ resistors in series are used for RIN2. So, RIN1 is calculated: 0.00641 R IN1 = ------------------------------- ⋅ ( 6.6MΩ ) = 42.6kΩ 1 – 0.00641 (EQ. 58) (EQ. 66) The displacement power factor increases to: Ia PF DIS = -------------------------------------------------------- = 0.967 2 2 ( I a ) + ( I c – I cneg ) (EQ. 67) VOLTAGE LOOP COMPENSATION The average diode forward current can be approximated by: Using resistor from the standard value, RIN1 = 43kΩ, the actual KBO is calculated: P in I D ( ave ) = ---------------V OUT R IN1 K BO = --------------------------------- = 0.00647 R IN1 + R IN2 Assuming the input current traces the input voltage perfectly. The input power is in proportion to (VCOMP - 1V). (EQ. 59) NEGATIVE INPUT CAPACITOR GENERATION The ISL6730A, ISL6730B, ISL6730C, ISL6730D generates an equivalent negative capacitance at the input to cancel the input filter capacitance. Thus, more input capacitors can be used without reducing the power factor. The input equivalent negative capacitance is a function of the current sensing gain, BO resistor divider gain and the compensation components. V m ⎞ R SEN ⎛ C NEG = ⎜ K BO ⋅ 0.8 – ----------------⎟ -------------------------- ( C ic + C ip ) V OUT⎠ R CS A iDC ⎝ (EQ. 68) ⎛ ⎞ R SEN 1 0.25 I D ( ave ) = --------------------------------------- • ---------------- • ⎜ ------------------------------------------------⎟ • Δ COMP ⎜ ⎟ 2 R CS ⋅ 0.5 ⋅ R IS V OUT ⎝ ( ( 2 2 ) ⁄ π ) ⋅ K BO⎠ (EQ. 69) Where ΔCOMP is the VCOMP - 1V. 1V is the offset voltage. RIS is the internal current scaling resistor. RIS = 16kΩ. A I D ( ave ) = 0.598 ---- • Δ COMP V (EQ. 70) (EQ. 60) 1.5 3.16k C NEG = ⎛ 0.00647 ⋅ 0.8 – ----------⎞ --------------------------- ( 18nF + 1.2nF ) = 0.62μF ⎝ 390⎠ 0.068 ⋅ 1.9 (EQ. 61) This equivalent negative capacitor cancels the input filter capacitor required for EMI filtering. Therefore, the displacement power factor significantly improves. 16 FN8258.0 February 26, 2013 ISL6730A, ISL6730B, ISL6730C, ISL6730D VOUT 2.5V RFB2 FB The zero, FZv is calculated: F CV F Zv = -----------------------------------------------------------------------------tan ( Φ m + atan ( F CV ⁄ ( F Pv ) ) ) (EQ. 76) 8Hz F Zv = ------------------------------------------------------------------------------------------------ = 1.15Hz tan ( 60deg + atan ( ( 8Hz ) ⁄ ( 20Hz ) ) ) (EQ. 77) Gmv RFB1 IFB Then the total capacitance used for compensation is calculated: 2 G PS ( i • ( 2πF CV ) ) • G DIV • Gmv ( F CV ⁄ F ZV ) + 1 C vc + C vp = ------------------------------------------------------------------------------------------- • ------------------------------------------2 ( 2πF CV ) ( F CV ⁄ F PV ) + 1 (EQ. 78) COMP Rvc Thus, the total compensation capacitance is: Cvp Cvc FIGURE 18. OUTPUT VOLTAGE SENSING AND COMPENSATION Thus, the transfer function from VCOMP to VOUT is: I D ( ave ) V OUT ( s ) 1 G PS ( s ) = ------------------------ = ---------------- ⋅ -------------------Δ COMP C O ⋅ s Δ COMP (EQ. 71) ⎛ I D ( ave ) 1 ⎞ 0.598 G PS ( s ) = ⎜ ------------------- ⋅ --------------------⎟ = ---------------CO ⋅ s ⎝ C O ⋅ s Δ COMP⎠ (EQ. 72) As shown in Figure 18, the voltage loop gain is: F ZV C vp = 1829nF • ----------- = 105nF F ZP (EQ. 81) 1 R vc = ------------------------------------------- = 81.2kΩ 2 ⋅ π ⋅ F ZV ⋅ C VC (EQ. 82) Choose components from the standard values. We have CVP = 100nF, CVC = 1500nF, RVC = 82.5kΩ. The actual bode plot is shown in Figure 20. 60 V REF G DIV = ---------------V OUT (EQ. 74) GAIN (dB) 40 The output feedback resistor divider gain, GDIV is: R vc • C vc • s + 1 1 Z COMP ( s ) = --------------------------------------- • ------------------------------------------------------------( C vc + C vp ) ⋅ s R vc • C vc • C vp ------------------------------------------ • s + 1 C vc + C vp 20 0 0 -20 The compensation gain uses external impedance networks as shown in Figure 18, ZCOMP(s) is given by: -40 90 (EQ. 75) PHASE (deg) 75 The targeted cross over frequency, FCV is 8Hz. The high frequency pole, FPv is required in order to reject the 2 time line frequency component. FPv = 20Hz. The targeted phase margin is 60°. 60 45 30 15 0 FZv 1 10 100 1x103 FREQUENCY (Hz) 80 FCV FPv FIGURE 20. BODE PLOT OF THE ACTUAL VOLTAGE LOOP GAIN 60 GAIN (dB) (EQ. 80) C vc = 1829nF – 105nF = 1724nF (EQ. 73) G VLOOP ( s ) = G PS ( s ) • G DIV • gmv • Z COMP ( s ) 100 (EQ. 79) C vc + C vp = 1829nF 40 ZCOMP(s) GPS(s) 20 0 GVLOOP(s) -20 -40 -60 gmv*GDIV 1 10 100 1k FREQUENCY (Hz) FIGURE 19. ASYMPTOTIC BODE PLOT OF CURRENT LOOP GAIN 17 FN8258.0 February 26, 2013 ISL6730A, ISL6730B, ISL6730C, ISL6730D Revision History The revision history provided is for informational purposes only and is believed to be accurate, but not warranted. Please go to the web to make sure that you have the latest revision. DATE REVISION February 26, 2013 FN8258.0 CHANGE Initial Release About Intersil Intersil Corporation is a leader in the design and manufacture of high-performance analog, mixed-signal and power management semiconductors. The company's products address some of the fastest growing markets within the industrial and infrastructure, personal computing and high-end consumer markets. For more information about Intersil or to find out how to become a member of our winning team, visit our website and career page at www.intersil.com. For a complete listing of Applications, Related Documentation and Related Parts, please see the respective product information page. Also, please check the product information page to ensure that you have the most updated datasheet: ISL6730A. To report errors or suggestions for this datasheet, please go to: www.intersil.com/askourstaff Reliability reports are available from our website at: http://rel.intersil.com/reports/search.php For additional products, see www.intersil.com/en/products.html Intersil products are manufactured, assembled and tested utilizing ISO9000 quality systems as noted in the quality certifications found at www.intersil.com/en/support/qualandreliability.html Intersil products are sold by description only. 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For information regarding Intersil Corporation and its products, see www.intersil.com 18 FN8258.0 February 26, 2013 ISL6730A, ISL6730B, ISL6730C, ISL6730D Package Outline Drawing M10.118 10 LEAD MINI SMALL OUTLINE PLASTIC PACKAGE Rev 1, 4/12 5 3.0±0.05 A DETAIL "X" D 10 1.10 MAX SIDE VIEW 2 0.09 - 0.20 4.9±0.15 3.0±0.05 5 0.95 REF PIN# 1 ID 1 2 0.50 BSC B GAUGE PLANE TOP VIEW 0.55 ± 0.15 0.25 3°±3° 0.85±010 H DETAIL "X" C SEATING PLANE 0.18 - 0.27 0.08 M C A-B D 0.10 ± 0.05 0.10 C SIDE VIEW 1 (5.80) NOTES: (4.40) (3.00) 1. Dimensions are in millimeters. 2. Dimensioning and tolerancing conform to JEDEC MO-187-BA and AMSEY14.5m-1994. 3. Plastic or metal protrusions of 0.15mm max per side are not included. 4. Plastic interlead protrusions of 0.15mm max per side are not included. (0.50) (0.29) (1.40) 5. Dimensions are measured at Datum Plane "H". 6. Dimensions in ( ) are for reference only. TYPICAL RECOMMENDED LAND PATTERN 19 FN8258.0 February 26, 2013 Mouser Electronics Authorized Distributor Click to View Pricing, Inventory, Delivery & Lifecycle Information: Intersil: ISL6730AFUZ ISL6730AFUZ-T