Multiphase PWM Regulator for IMVP-6.5™ Mobile CPUs ISL62883, ISL62883B Features The ISL62883 is a multiphase PWM buck regulator for miroprocessor core power supply. The multiphase buck converter uses interleaved phase to reduce the total output voltage ripple with each phase carrying a portion of the total load current, providing better system performance, superior thermal management, lower component cost, reduced power dissipation, and smaller implementation area. The ISL62883 uses two integrated gate drivers and an external gate driver to provide a complete solution. The PWM modulator is based on Intersil's Robust Ripple Regulator (R3) technology™. Compared with traditional modulators, the R3™ modulator commands variable switching frequency during load transients, achieving faster transient response. With the same modulator, the switching frequency is reduced at light load, increasing the regulator efficiency. • Precision Multiphase Core Voltage Regulation - 0.5% System Accuracy Over-Temperature - Enhanced Load Line Accuracy The ISL62883 is fully compliant with IMVP-6.5™ specifications. It responds to PSI# and DPRSLPVR signals by adding or dropping PWM3 and Phase-2 respectively, adjusting overcurrent protection threshold accordingly, and entering/exiting diode emulation mode. It reports the regulator output current through the IMON pin. It senses the current by using either a discrete resistor or inductor DCR whose variation over temperature can be thermally compensated by a single NTC thermistor. It uses differential remote voltage sensing to accurately regulate the processor die voltage. The adaptive body diode conduction time reduction function minimizes the body diode conduction loss in diode emulation mode. User-selectable overshoot reduction function offers an option to aggressively reduce the output capacitors as well as the option to disable it for users concerned about increased system thermal stress. In 2-Phase configuration, the ISL62883 offers the FB2 function to optimize 1-Phase performance. • Current Monitor and Thermal Monitor The ISL62883B has the same functions as the ISL62883, but comes in a different package. • Notebook Computers June 21, 2011 FN6891.4 1 • Microprocessor Voltage Identification Input - 7-Bit VID Input, 0.300V to 1.500V in 12.5mV Steps - Supports VID Changes On-The-Fly • Supports Multiple Current Sensing Methods - Lossless Inductor DCR Current Sensing - Precision Resistor Current Sensing • Supports PSI# and DPRSLPVR modes • Superior Noise Immunity and Transient Response • Differential Remote Voltage Sensing • High Efficiency Across Entire Load Range • Programmable 1-, 2- or 3-Phase Operation • Two Integrated Gate Drivers • Excellent Dynamic Current Balance Between Phases • FB2 Function in 2-Phase Configuration to Optimize 1-Phase Performance • Adaptive Body Diode Conduction Time Reduction • User-selectable Overshoot Reduction Function • Small Footprint 40 Ld 5x5 or 48 Ld 6x6 TQFN Package • Pb-Free (RoHS Compliant) Applications CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 1-888-468-3774 | Copyright Intersil Americas Inc. 2009-2011. All Rights Reserved Intersil (and design) is a trademark owned by Intersil Corporation or one of its subsidiaries. All other trademarks mentioned are the property of their respective owners. ISL62883, ISL62883B Ordering Information PART NUMBER (Notes 1, 2, 3) PART MARKING TEMP. RANGE (°C) PACKAGE (Pb-Free) PKG. DWG. # ISL62883HRTZ 62883 HRTZ -10 to +100 40 Ld 5x5 TQFN L40.5x5 ISL62883IRTZ 62883 IRTZ -40 to +100 40 Ld 5x5 TQFN L40.5x5 ISL62883BHRTZ 62883 BHRTZ -10 to +100 48 Ld 6x6 TQFN L48.6x6 NOTES: 1. Add “-T*” suffix for tape and reel. Please refer to TB347 for details on reel specifications. 2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil Pbfree products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020. 3. For Moisture Sensitivity Level (MSL), please see device information page for ISL62883, ISL62883B. For more information on MSL please see techbrief TB363. Pin Configurations 24 PWM3 8 23 LGATE1 9 22 VSSP1 21 PHASE1 ISEN2 10 2 BOOT1 UGATE1 VIN IMON VDD ISUM- ISUM+ RTN VSEN ISEN1 11 12 13 14 15 16 17 18 19 20 VW NC VID0 VID1 VID3 VID4 VID5 VR_ON CLK_EN# VID2 31 NC (BOTTOM) 30 VCCP 8 29 PWM3 COMP 9 28 LGATE1 FB 10 27 VSSP1 ISEN3/FB2 11 26 PHASE1 NC 12 25 UGATE1 13 14 15 16 17 18 19 20 21 22 23 24 NC SEN3/FB2 GND 7 BOOT1 FB 25 VCCP 32 LGATE2 IMON COMP 7 NTC 6 NC 6 26 LGATE2 VIN VW GND PAD (BOTTOM) 33 VSSP2 VR_TT# 5 VDD NTC 5 34 PHASE2 ISUM+ 27 VSSP2 PSI# 3 RBIAS 4 RTN 28 PHASE2 ISUM- RBIAS 3 PGOOD 2 36 BOOT2 35 UGATE2 VSEN 29 UGATE2 VR_TT# 4 DPRSLPVR NC PSI# 2 NC 1 ISEN1 30 BOOT2 ISEN2 VID1 VID2 VID4 VID3 VID6 VID5 VR_ON DPRSLPVR CLK_EN# VID0 48 47 46 45 44 43 42 41 40 39 38 37 40 39 38 37 36 35 34 33 32 31 PGOOD 1 VID6 ISL62883B (48 LD TQFN) TOP VIEW ISL62883 (40 LD TQFN) TOP VIEW FN6891.4 June 21, 2011 ISL62883, ISL62883B Pin Function Descriptions RTN GND Remote voltage sensing return. Connect to ground at microprocessor die. Signal common of the IC. Unless otherwise stated, signals are referenced to the GND pin. ISUM- and ISUM+ PGOOD Droop current sense input. Power-Good open-drain output indicating when the regulator is able to supply regulated voltage. Pull-up externally with a 680Ω resistor to VCCP or 1.9kΩ to 3.3V. VDD 5V bias power. VIN PSI# Low load current indicator input. When asserted low, indicates a reduced load-current condition. For ISL62883, when PSI# is asserted low, PWM3 will be disabled. RBIAS Battery supply voltage, used for feed-forward. IMON An analog output. IMON outputs a current proportional to the regulator output current. 147k resistor to GND sets internal current reference. BOOT1 VR_TT# Connect an MLCC capacitor across the BOOT1 and the PHASE1 pins. The boot capacitor is charged through an internal boot diode connected from the VCCP pin to the BOOT1 pin, each time the PHASE1 pin drops below VCCP minus the voltage dropped across the internal boot diode. Thermal overload output indicator. NTC Thermistor input to VR_TT# circuit. VW A resistor from this pin to COMP programs the switching frequency (8kΩ gives approximately 300kHz). COMP This pin is the output of the error amplifier. Also, a resistor across this pin and GND adjusts the overcurrent threshold. FB This pin is the inverting input of the error amplifier. ISEN3/FB2 When the ISL62883 is configured in 3-phase mode, this pin is ISEN3. ISEN3 is the individual current sensing for phase 3. When the ISL62883 is configured in 2-phase mode, this pin is FB2. There is a switch between the FB2 pin and the FB pin. The switch is on in 2-phase mode and is off in 1-phase mode. The components connecting to FB2 are used to adjust the compensation in 1-phase mode to achieve optimum performance. ISEN2 Individual current sensing for Phase-2. When ISEN2 is pulled to 5V VDD, the controller will disable Phase-2 and allow other phases to operate. UGATE1 Output of the Phase-1 high-side MOSFET gate driver. Connect the UGATE1 pin to the gate of the Phase-1 high-side MOSFET. PHASE1 Current return path for the Phase-1 high-side MOSFET gate driver. Connect the PHASE1 pin to the node consisting of the high-side MOSFET source, the low-side MOSFET drain, and the output inductor of Phase-1. VSSP1 Current return path for the Phase-1 low-side MOSFET gate driver. Connect the VSSP1 pin to the source of the Phase-1 low-side MOSFET through a low impedance path, preferably in parallel with the trace connecting the LGATE1 pin to the gate of the Phase-1 low-side MOSFET. LGATE1 Output of the Phase-1 low-side MOSFET gate driver. Connect the LGATE1 pin to the gate of the Phase-1 low-side MOSFET. PWM3 PWM output for Channel 3. When PWM3 is pulled to 5V VDD, the controller will disable Phase-3 and allow other phases to operate. VCCP Individual current sensing for Phase-1. Input voltage bias for the internal gate drivers. Connect +5V to the VCCP pin. Decouple with at least 1µF of an MLCC capacitor to VSSP1 and VSSP2 pins respectively. VSEN LGATE2 Remote core voltage sense input. Connect to microprocessor die. Output of the Phase-2 low-side MOSFET gate driver. Connect the LGATE2 pin to the gate of the Phase-2 low-side MOSFET. ISEN1 3 FN6891.4 June 21, 2011 ISL62883, ISL62883B VSSP2 VID0, VID1, VID2, VID3, VID4, VID5, VID6 Current return path for the Phase-2 converter low-side MOSFET gate driver. Connect the VSSP2 pin to the source of the Phase-2 low-side MOSFET through a low impedance path, preferably in parallel with the trace connecting the LGATE2 pin to the gate of the Phase-2 low-side MOSFET. VID input with VID0 = LSB and VID6 = MSB. PHASE2 DPRSLPVR Current return path for the Phase-2 high-side MOSFET gate driver. Connect the PHASE2 pin to the node consisting of the high-side MOSFET source, the low-side MOSFET drain, and the output inductor of Phase-2. Deeper sleep enable signal. A high level logic signal on this pin indicates that the microprocessor is in deeper sleep mode. UGATE2 Open drain output to enable system PLL clock. It goes low 13 switching cycles after Vcore is within 10% of Vboot. Output of the Phase-2 high-side MOSFET gate driver. Connect the UGATE2 pin to the gate of the Phase-2 high-side MOSFET. VR_ON Voltage regulator enable input. A high level logic signal on this pin enables the regulator. CLK_EN# NC BOOT2 No Connect. Connect an MLCC capacitor across the BOOT2 and the PHASE2 pins. The boot capacitor is charged through an internal boot diode connected from the VCCP pin to the BOOT2 pin, each time the PHASE2 pin drops below VCCP minus the voltage dropped across the internal boot diode. BOTTOM (on ISL62883B) 4 The bottom pad of ISL62883B is electrically connected to the GND pin inside the IC. FN6891.4 June 21, 2011 ISL62883, ISL62883B Block Diagram VDD VIN VSEN ISEN1 ISEN3 ISEN2 PGOOD CLK_EN# 6µA 54µA 1.20V VR_ON PGOOD & CLK_EN# LOGIC CURRENT BALANCE DPRSLPVR NTC IBAL PROTECTION BOOT2 FLT VID0 IBAL VIN VDAC VID1 VID3 WOC OC VIN DAC AND SOFT START MODULATOR CLOCK COMP VDAC COMP VID4 VW COMP VID5 IBAL VIN VDAC VID6 MODULATOR Σ RTN COMP MODULATOR IDROOP WOC IMON 2.5X ISUM+ CURRENT SENSE ISUM- COMP CURRENT COMPARATORS NUMBER OF OC PHASES GAIN SELECT 5 UGATE2 PHASE2 SHOOT THROUGH PROTECTION LGATE2 DRIVER VSSP2 PWM3 BOOT1 IBAL VIN VDAC IMON DRIVER COMP E/A FB VW PWM CONTROL LOGIC RBIAS VID2 VR_TT# 1.24V 60UA PWM CONTROL LOGIC PSI# MODE CONTROL DRIVER UGATE1 PHASE1 SHOOT THROUGH PROTECTION VCCP DRIVER LGATE1 VSSP1 Σ ADJ. OCP THRESHOLD COMP GND FN6891.4 June 21, 2011 ISL62883, ISL62883B Absolute Maximum Ratings Thermal Information Supply Voltage, VDD. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +7V Battery Voltage, VIN . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +28V Boot Voltage (BOOT) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +33V Boot to Phase Voltage (BOOT-PHASE) . . . . . . . . . . . . . . . . -0.3V to +7V(DC) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +9V(<10ns) Phase Voltage (PHASE) . . . . . . . . . . . . . . . . -7V (<20ns Pulse Width, 10µJ) UGATE Voltage (UGATE) . . . . . . . . . . . . . . . . . . . PHASE - 0.3V (DC) to BOOT . . . . . . . . . . . . . . . . . . . . . . . PHASE-5V (<20ns Pulse Width, 10µJ) to BOOT LGATE Voltage (LGATE). . . . . . . . . . . . . . . . . . . . . . -0.3V (DC) to VDD + 0.3V . . . . . . . . . . . . . . . . . . . . . . . -2.5V (<20ns Pulse Width, 5µJ) to VDD + 0.3V All Other Pins . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to (VDD +0.3V) Open Drain Outputs, PGOOD, VR_TT#, CLK_EN# . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +7V Thermal Resistance (Typical) θJA (°C/W) θJC (°C/W) 40 Ld TQFN Package (Notes 4, 5) . . . . . . . 32 3 48 Ld TQFN Package (Notes 4, 5) . . . . . . . 29 2 Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . .+150°C Maximum Storage Temperature Range . . . . . . . . . . . . . .-65°C to +150°C Pb-Free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . see link below http://www.intersil.com/pbfree/Pb-FreeReflow.asp Recommended Operating Conditions Supply Voltage, VDD. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +5V ±5% Battery Voltage, VIN . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +4.5V to 25V Ambient Temperature ISL62883HRTZ, ISL62883BHRTZ . . . . . . . . . . . . . . . . .-10°C to +100°C ISL62883IRTZ. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-40°C to +100°C Junction Temperature ISL62883HRTZ, ISL62883BHRTZ . . . . . . . . . . . . . . . . .-10°C to +125°C ISL62883IRTZ. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-40°C to +125°C CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and result in failures not covered by warranty. NOTES: 4. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See Tech Brief TB379. 5. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside. Electrical Specifications Operating Conditions: VDD = 5V, TA = -40°C to +100°C, fSW = 300kHz, unless otherwise noted. Boldface limits apply over the operating temperature range, -40°C to +100°C. PARAMETER SYMBOL TEST CONDITIONS MIN (Note 6) TYP MAX (Note 6) UNITS INPUT POWER SUPPLY +5V Supply Current IVDD VR_ON = 1V 4 4.6 mA VR_ON = 0V 1 µA 1 µA Battery Supply Current IVIN VR_ON = 0V VIN Input Resistance RVIN VR_ON = 1V 900 Power-On-Reset Threshold PORr VDD rising 4.35 PORf VDD falling 4.00 kΩ 4.5 4.15 V V SYSTEM AND REFERENCES System Accuracy HRTZ No load; closed loop, active mode range %Error (VCC_CORE) VID = 0.75V to 1.50V -0.5 +0.5 % VID = 0.5V to 0.7375V -8 +8 mV VID = 0.3V to 0.4875V -15 +15 mV -0.8 +0.8 % -10 +10 mV -18 +18 mV IRTZ No load; closed loop, active mode range %Error (VCC_CORE) VID = 0.75V to 1.50V VID = 0.5V to 0.7375V VID = 0.3V to 0.4875V VBOOT 1.0945 1.100 1.1055 V Maximum Output Voltage VCC_CORE(max) VID = [0000000] 1.500 V Minimum Output Voltage VCC_CORE(min) VID = [1100000] 0.300 V RBIAS Voltage RBIAS = 147kΩ 6 1.45 1.47 1.49 V FN6891.4 June 21, 2011 ISL62883, ISL62883B Electrical Specifications Operating Conditions: VDD = 5V, TA = -40°C to +100°C, fSW = 300kHz, unless otherwise noted. Boldface limits apply over the operating temperature range, -40°C to +100°C. (Continued) PARAMETER SYMBOL TEST CONDITIONS MIN (Note 6) TYP 285 300 MAX (Note 6) UNITS CHANNEL FREQUENCY Nominal Channel Frequency fSW(nom) Rfset = 7kΩ, 3 channel operation, VCOMP = 1V Adjustment Range 315 kHz 200 500 kHz -0.15 +0.15 mV AMPLIFIERS Current-Sense Amplifier Input Offset Error Amp DC Gain IFB = 0A Av0 Error Amp Gain-Bandwidth Product GBW CL= 20pF 90 dB 18 MHz ISEN Imbalance Voltage Maximum of ISENs - Minimum of ISENs 1 Input Bias Current 20 mV nA POWER GOOD AND PROTECTION MONITORS PGOOD Low Voltage VOL IPGOOD = 4mA PGOOD Leakage Current IOH PGOOD = 3.3V -1 PGOOD Delay tpgd CLK_EN# LOW to PGOOD HIGH 6.3 0.26 0.4 V 1 µA 7.6 8.9 ms 1.5 Ω GATE DRIVER UGATE Pull-Up Resistance RUGPU 200mA Source Current 1.0 UGATE Source Current IUGSRC UGATE - PHASE = 2.5V 2.0 UGATE Sink Resistance RUGPD 250mA Sink Current 1.0 UGATE Sink Current IUGSNK UGATE - PHASE = 2.5V 2.0 LGATE Pull-Up Resistance RLGPU 250mA Source Current 1.0 LGATE Source Current ILGSRC LGATE - VSSP = 2.5V 2.0 A 1.5 Ω A 1.5 Ω A Ω LGATE Sink Resistance RLGPD 250mA Sink Current 0.5 LGATE Sink Current ILGSNK LGATE - VSSP = 2.5V 4.0 A UGATE to LGATE Deadtime tUGFLGR UGATE falling to LGATE rising, no load 23 ns LGATE to UGATE Deadtime tLGFUGR LGATE falling to UGATE rising, no load 28 ns 0.9 BOOTSTRAP DIODE Forward Voltage VF PVCC = 5V, IF = 2mA 0.58 V Reverse Leakage IR VR = 25V 0.2 µA PROTECTION Overvoltage Threshold OVH VSEN rising above setpoint for >1ms Severe Overvoltage Threshold OVHS VSEN rising for >2µs OC Threshold Offset at Rcomp = Open Circuit Current Imbalance Threshold 150 195 240 mV 1.525 1.55 1.575 V 3-phase configuration, ISUM- pin current 28.4 30.3 32.2 µA 2-phase configuration, ISUM- pin current 18.3 20.2 22.1 µA 1-phase configuration, ISUM- pin current 8.2 10.1 12.0 µA One ISEN above another ISEN for >1.2ms Undervoltage Threshold UVf VSEN falling below setpoint for >1.2ms 9 -355 -295 mV -235 mV 0.3 V LOGIC THRESHOLDS VR_ON Input Low VIL(1.0V) 7 FN6891.4 June 21, 2011 ISL62883, ISL62883B Electrical Specifications Operating Conditions: VDD = 5V, TA = -40°C to +100°C, fSW = 300kHz, unless otherwise noted. Boldface limits apply over the operating temperature range, -40°C to +100°C. (Continued) PARAMETER SYMBOL VR_ON Input High TEST CONDITIONS MIN (Note 6) TYP MAX (Note 6) UNITS HRTZ VIH(1.0V) 0.7 V IRTZ VIH(1.0V) 0.75 V VID0-VID6, PSI#, and DPRSLPVR Input Low VIL(1.0V) VID0-VID6, PSI#, and DPRSLPVR Input High VIH(1.0V) 0.3 0.7 V V PWM PWM3 Output Low VOL(5.0V) Sinking 5mA PWM3 Output High VOH(5.0V) Sourcing 5mA PWM Tri-State Leakage 1.0 3.5 PWM = 2.5V V V 2 µA THERMAL MONITOR NTC Source Current NTC = 1.3V Over-Temperature Threshold V (NTC) falling 53 60 67 µA 1.18 1.2 1.22 V RTT I = 20mA 6.5 9 Ω CLK_EN# Low Output Voltage VOL I = 4mA 0.26 0.4 V CLK_EN# Leakage Current IOH CLK_EN# = 3.3V 1 µA VR_TT# Low Output Resistance CLK_EN# OUTPUT LEVELS -1 CURRENT MONITOR IMON Output Current IIMON IMON Clamp Voltage ISUM- pin current = 20μA 108 120 132 µA ISUM- pin current = 10μA 51 60 69 µA ISUM- pin current = 5μA 22 30 37.5 µA 1.1 1.15 VIMONCLAMP Current Sinking Capability V 275 µA 0 µA INPUTS VR_ON Leakage Current IVR_ON VR_ON = 0V -1 VR_ON = 1V VIDx Leakage Current IVIDx VIDx = 0V 0 -1 VIDx = 1V PSI# Leakage Current IPSI# PSI# = 0V PSI# = 1V DPRSLPVR Leakage Current IDPRSLPVR DPRSLPVR = 0V 0 0.45 -1 DPRSLPVR = 1V 1 µA µA 1 0 0.45 µA µA 0 0.45 -1 1 µA µA 1 µA 6.5 mV/µs SLEW RATE Slew Rate (For VID Change) SR 5 NOTE: 6. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization and are not production tested. 8 FN6891.4 June 21, 2011 ISL62883, ISL62883B Gate Driver Timing Diagram PWM tLGFUGR tFU tRU 1V UGATE 1V LGATE tRL tFL tUGFLGR 9 FN6891.4 June 21, 2011 ISL62883, ISL62883B Simplified Application Circuits V+5 V+5 Vin V+5 Rbias RBIAS Rntc C L3 ISL6208 BOOT PWM LGATE GND PWM3 NTC o Vin VCC UGATE FCCM PHASE VDD VCCP VIN Rs3 PGOOD VR_TT# CLK_EN# VID<0:6> PSI# DPRSLPVR VR_ON PGOOD VR_TT# CLK_EN# VIDs PSI# DPRSLPVR VR_ON VW ISEN3 BOOT2 Cs3 UGATE2 L2 PHASE2 LGATE2 VSSP2 Rs2 ISEN2 ISL62883 BOOT1 Rfset Vo Cs2 UGATE1 L1 PHASE1 COMP LGATE1 VSSP1 FB Rs1 ISEN1 Rdroop VSEN Cs1 Rsum3 ISUM+ Rsum2 VCCSENSE VSSSENSE Rn Ris RTN Cn Rimon o C Rsum1 Cis Ri IMON IMON (Bottom Pad) VSS ISUM- FIGURE 1. TYPICAL APPLICATION CIRCUIT USING DCR SENSING V+5 V+5 Vin V+5 Rbias RBIAS Rsen3 L2 Rsen2 L1 Rsen1 BOOT PWM LGATE GND PWM3 NTC C L3 ISL6208 Rntc o Vin VCC UGATE FCCM PHASE VDD VCCP VIN Rs3 PGOOD VR_TT# CLK_EN# VIDs PSI# DPRSLPVR VR_ON VW PGOOD VR_TT# CLK_EN# VID<0:6> PSI# DPRSLPVR VR_ON ISEN3 BOOT2 Cs3 UGATE2 PHASE2 LGATE2 VSSP2 Rs2 ISEN2 ISL62883 BOOT1 Rfset Vo Cs2 UGATE1 PHASE1 COMP FB LGATE1 VSSP1 Rs1 ISEN1 Rdroop Cs2 VSEN Rsum3 ISUM+ Rsum2 VCCSENSE VSSSENSE Ris RTN Cn Rimon Rsum1 Cis Ri IMON IMON (Bottom Pad) VSS ISUM- FIGURE 2. TYPICAL APPLICATION CIRCUIT USING RESISTOR SENSING 10 FN6891.4 June 21, 2011 ISL62883, ISL62883B Theory of Operation VW Multiphase R3™ Modulator MASTER CLOCK CIRCUIT MASTER CLOCK COMP Phase Vcrm Sequencer VW MASTER CLOCK gmVo COMP Clock1 Clock2 Clock3 Vcrm Crm Master Clock SLAVE CIRCUIT 1 VW Clock1 S R Q PWM1 Phase1 L1 IL1 Vcrs1 Clock1 Vo PWM1 Co Clock2 PWM2 gm Crs1 Clock3 SLAVE CIRCUIT 2 VW Clock2 S R Q PWM2 Phase2 L2 PWM3 VW IL2 Vcrs2 gm Crs2 SLAVE CIRCUIT 3 VW Clock3 S R Q PWM3 Phase3 L3 IL3 Vcrs3 Vcrs1 Vcrs3 Vcrs2 FIGURE 5. R3™ MODULATOROPERATION PRINCIPLES IN LOAD INSERTION RESPONSE gm Crs3 FIGURE 3. R3™ MODULATORCIRCUIT VW Hysteretic W indow Vcrm The ISL62883 is a multiphase regulator, which implements Intel™ IMVP-6.5™ protocol. It can be programmed for 1-, 2- or 3-phase operation for microprocessor core applications. It uses Intersil patented R3™ (Robust Ripple Regulator™) modulator. The R3™ modulator combines the best features of fixed frequency PWM and hysteretic PWM while eliminating many of their shortcomings. Figure 3 conceptually shows the ISL62883 multiphase R3™ modulator circuit, and Figure 4 shows the operation principles. A current source flows from the VW pin to the COMP pin, creating a voltage window set by the resistor between the two pins. This voltage window is called VW window in the following discussion. COMP Master Clock Inside the IC, the modulator uses the master clock circuit to generate the clocks for the slave circuits. The modulator discharges the ripple capacitor Crm with a current source equal to gmVo, where gm is a gain factor. Crm voltage Vcrm is a sawtooth waveform traversing between the VW and COMP voltages. It resets to VW when it hits COMP, and generates a oneshot master clock signal. A phase sequencer distributes the master clock signal to the slave circuits. If the ISL62883 is in 3-phase mode, the master clock signal will be distributed to the three phases, and the Clock1~3 signals will be 120° out-ofphase. If the ISL62883 is in 2-phase mode, the master clock signal will be distributed to Phases 1 and 2, and the Clock1 and Clock2 signals will be 180° out-of-phase. If the ISL62883 is in 1-phase mode, the master clock signal will be distributed to Phases 1 only and be the Clock1 signal. Clock1 PW M1 Clock2 PW M2 Clock3 PW M3 VW Vcrs2 Vcrs3 Vcrs1 FIGURE 4. R3™ MODULATOR OPERATION PRINCIPLES IN STEADY STATE 11 Each slave circuit has its own ripple capacitor Crs, whose voltage mimics the inductor ripple current. A gm amplifier converts the inductor voltage into a current source to charge and discharge Crs. The slave circuit turns on its PWM pulse upon receiving the clock signal, and the current source charges Crs. When Crs FN6891.4 June 21, 2011 ISL62883, ISL62883B voltage VCrs hits VW, the slave circuit turns off the PWM pulse, and the current source discharges Crs. CCM/DCM BOUNDARY VW Since the ISL62883 works with Vcrs, which are large-amplitude and noise-free synthesized signals, the ISL62883 achieves lower phase jitter than conventional hysteretic mode and fixed PWM mode controllers. Unlike conventional hysteretic mode converters, the ISL62883 has an error amplifier that allows the controller to maintain a 0.5% output voltage accuracy. Vcrs Figure 5 shows the operation principles during load insertion response. The COMP voltage rises during load insertion, generating the master clock signal more quickly, so the PWM pulses turn on earlier, increasing the effective switching frequency, which allows for higher control loop bandwidth than conventional fixed frequency PWM controllers. The VW voltage rises as the COMP voltage rises, making the PWM pulses wider. During load release response, the COMP voltage falls. It takes the master clock circuit longer to generate the next master clock signal so the PWM pulse is held off until needed. The VW voltage falls as the VW voltage falls, reducing the current PWM pulse width. This kind of behavior gives the ISL62883 excellent response speed. Vcrs The fact that all the phases share the same VW window voltage also ensures excellent dynamic current balance among phases. Diode Emulation and Period Stretching Phase UG ATE LG ATE iL VW LIGHT DCM iL VW DEEP DCM Vcrs iL FIGURE 7. PERIOD STRETCHING Figure 7 shows the operation principle in diode emulation mode at light load. The load gets incrementally lighter in the three cases from top to bottom. The PWM on-time is determined by the VW window size, therefore is the same, making the inductor current triangle the same in the three cases. The ISL62883 clamps the ripple capacitor voltage Vcrs in DE mode to make it mimic the inductor current. It takes the COMP voltage longer to hit Vcrs, naturally stretching the switching period. The inductor current triangles move further apart from each other such that the inductor current average value is equal to the load current. The reduced switching frequency helps increase light load efficiency. Start-up Timing IL FIGURE 6. DIODE EMULATION ISL62883 can operate in diode emulation (DE) mode to improve light load efficiency. In DE mode, the low-side MOSFET conducts when the current is flowing from source to drain and doesn’t not allow reverse current, emulating a diode. As Figure 6 shows, when LGATE is on, the low-side MOSFET carries current, creating negative voltage on the phase node due to the voltage drop across the ON-resistance. The ISL62883 monitors the current through monitoring the phase node voltage. It turns off LGATE when the phase node voltage reaches zero to prevent the inductor current from reversing the direction and creating unnecessary power loss. If the load current is light enough, as Figure 6 shows, the inductor current will reach and stay at zero before the next phase node pulse, and the regulator is in discontinuous conduction mode (DCM). If the load current is heavy enough, the inductor current will never reach 0A, and the regulator is in CCM although the controller is in DE mode. With the controller's VDD voltage above the POR threshold, the start-up sequence begins when VR_ON exceeds the 3.3V logic high threshold. The ISL62883 uses digital soft start to ramp up DAC to the boot voltage of 1.1V at about 2.5mV/µs. Once the output voltage is within 10% of the boot voltage for 13 PWM cycles (43µs for frequency = 300kHz), CLK_EN# is pulled low and DAC slews at 5mV/µs to the voltage set by the VID pins. PGOOD is asserted high in approximately 7ms. Figure 8 shows the typical start-up timing. Similar results occur if VR_ON is tied to VDD, with the soft-start sequence starting 120µs after VDD crosses the POR threshold. VDD 5mV/µs VR_ON 2.5mV/µs 90% VBOOT 800µs DAC VID COMMAND VOLTAGE 13 SWITCHING CYCLES CLK_EN# ~7ms PGOOD FIGURE 8. SOFT-START WAVEFORMS 12 FN6891.4 June 21, 2011 ISL62883, ISL62883B Voltage Regulation and Load Line Implementation TABLE 1. VID TABLE (Continued) VID6 VID5 VID4 VID3 VID2 VID1 VID0 VO (V) After the start sequence, the ISL62883 regulates the output voltage to the value set by the VID inputs per Table 1. The ISL62883 will control the no-load output voltage to an accuracy of ±0.5% over the range of 0.75V to 1.5V. A differential amplifier allows voltage sensing for precise voltage regulation at the microprocessor die. 0 1 0 0 1 0 0 1.0500 0 1 0 0 1 0 1 1.0375 0 1 0 0 1 1 0 1.0250 0 1 0 0 1 1 1 1.0125 0 1 0 1 0 0 0 1.0000 TABLE 1. VID TABLE 0 1 0 1 0 0 1 0.9875 VID6 VID5 VID4 VID3 VID2 VID1 VID0 VO (V) 0 1 0 1 0 1 0 0.9750 0 0 0 0 0 0 0 1.5000 0 1 0 1 0 1 1 0.9625 0 0 0 0 0 0 1 1.4875 0 1 0 1 1 0 0 0.9500 0 0 0 0 0 1 0 1.4750 0 1 0 1 1 0 1 0.9375 0 0 0 0 0 1 1 1.4625 0 1 0 1 1 1 0 0.9250 0 0 0 0 1 0 0 1.4500 0 1 0 1 1 1 1 0.9125 0 0 0 0 1 0 1 1.4375 0 1 1 0 0 0 0 0.9000 0 0 0 0 1 1 0 1.4250 0 1 1 0 0 0 1 0.8875 0 0 0 0 1 1 1 1.4125 0 1 1 0 0 1 0 0.8750 0 0 0 1 0 0 0 1.4000 0 1 1 0 0 1 1 0.8625 0 0 0 1 0 0 1 1.3875 0 1 1 0 1 0 0 0.8500 0 0 0 1 0 1 0 1.3750 0 1 1 0 1 0 1 0.8375 0 0 0 1 0 1 1 1.3625 0 1 1 0 1 1 0 0.8250 0 0 0 1 1 0 0 1.3500 0 1 1 0 1 1 1 0.8125 0 0 0 1 1 0 1 1.3375 0 1 1 1 0 0 0 0.8000 0 0 0 1 1 1 0 1.3250 0 1 1 1 0 0 1 0.7875 0 0 0 1 1 1 1 1.3125 0 1 1 1 0 1 0 0.7750 0 0 1 0 0 0 0 1.3000 0 1 1 1 0 1 1 0.7625 0 0 1 0 0 0 1 1.2875 0 1 1 1 1 0 0 0.7500 0 0 1 0 0 1 0 1.2750 0 1 1 1 1 0 1 0.7375 0 0 1 0 0 1 1 1.2625 0 1 1 1 1 1 0 0.7250 0 0 1 0 1 0 0 1.2500 0 1 1 1 1 1 1 0.7125 0 0 1 0 1 0 1 1.2375 1 0 0 0 0 0 0 0.7000 0 0 1 0 1 1 0 1.2250 1 0 0 0 0 0 1 0.6875 0 0 1 0 1 1 1 1.2125 1 0 0 0 0 1 0 0.6750 0 0 1 1 0 0 0 1.2000 1 0 0 0 0 1 1 0.6625 0 0 1 1 0 0 1 1.1875 1 0 0 0 1 0 0 0.6500 0 0 1 1 0 1 0 1.1750 1 0 0 0 1 0 1 0.6375 0 0 1 1 0 1 1 1.1625 1 0 0 0 1 1 0 0.6250 0 0 1 1 1 0 0 1.1500 1 0 0 0 1 1 1 0.6125 0 0 1 1 1 0 1 1.1375 1 0 0 1 0 0 0 0.6000 0 0 1 1 1 1 0 1.1250 1 0 0 1 0 0 1 0.5875 0 0 1 1 1 1 1 1.1125 1 0 0 1 0 1 0 0.5750 0 1 0 0 0 0 0 1.1000 1 0 0 1 0 1 1 0.5625 0 1 0 0 0 0 1 1.0875 1 0 0 1 1 0 0 0.5500 0 1 0 0 0 1 0 1.0750 1 0 0 1 1 0 1 0.5375 0 1 0 0 0 1 1 1.0625 1 0 0 1 1 1 0 0.5250 13 FN6891.4 June 21, 2011 ISL62883, ISL62883B TABLE 1. VID TABLE (Continued) TABLE 1. VID TABLE (Continued) VID6 VID5 VID4 VID3 VID2 VID1 VID0 VO (V) VID6 VID5 VID4 VID3 VID2 VID1 VID0 VO (V) 1 0 0 1 1 1 1 0.5125 1 1 1 1 0 1 0 0.0000 1 0 1 0 0 0 0 0.5000 1 1 1 1 0 1 1 0.0000 1 0 1 0 0 0 1 0.4875 1 1 1 1 1 0 0 0.0000 1 0 1 0 0 1 0 0.4750 1 1 1 1 1 0 1 0.0000 1 0 1 0 0 1 1 0.4625 1 1 1 1 1 1 0 0.0000 1 0 1 0 1 0 0 0.4500 1 1 1 1 1 1 1 0.0000 1 0 1 0 1 0 1 0.4375 1 0 1 0 1 1 0 0.4250 1 0 1 0 1 1 1 0.4125 1 0 1 1 0 0 0 0.4000 1 0 1 1 0 0 1 0.3875 1 0 1 1 0 1 0 0.3750 1 0 1 1 0 1 1 0.3625 1 0 1 1 1 0 0 0.3500 1 0 1 1 1 0 1 0.3375 1 0 1 1 1 1 0 0.3250 1 0 1 1 1 1 1 0.3125 1 1 0 0 0 0 0 0.3000 1 1 0 0 0 0 1 0.2875 1 1 0 0 0 1 0 0.2750 1 1 0 0 0 1 1 0.2625 1 1 0 0 1 0 0 0.2500 1 1 0 0 1 0 1 0.2375 1 1 0 0 1 1 0 0.2250 1 1 0 0 1 1 1 0.2125 1 1 0 1 0 0 0 0.2000 1 1 0 1 0 0 1 0.1875 1 1 0 1 0 1 0 0.1750 1 1 0 1 0 1 1 0.1625 1 1 0 1 1 0 0 0.1500 1 1 0 1 1 0 1 0.1375 1 1 0 1 1 1 0 0.1250 1 1 0 1 1 1 1 0.1125 1 1 1 0 0 0 0 0.1000 1 1 1 0 0 0 1 0.0875 1 1 1 0 0 1 0 0.0750 1 1 1 0 0 1 1 0.0625 1 1 1 0 1 0 0 0.0500 1 1 1 0 1 0 1 0.0375 1 1 1 0 1 1 0 0.0250 1 1 1 0 1 1 1 0.0125 1 1 1 1 0 0 0 0.0000 1 1 1 1 0 0 1 0.0000 14 RDROOP VCC SENSE VDROOP FB VR LOCAL VO “CATCH” RESISTOR IDROOP E/A COMP Σ VIDS VDAC VID<0:6> DAC RTN INTERNAL TO IC X1 VSSSENSE VSS “CATCH” RESISTOR FIGURE 9. DIFFERENTIAL SENSING AND LOAD LINE IMPLEMENTATION As the load current increases from zero, the output voltage will droop from the VID table value by an amount proportional to the load current to achieve the load line. The ISL62883 can sense the inductor current through the intrinsic DC Resistance (DCR) resistance of the inductors Figure 1 shows on page 10 or through resistors in series with the inductors as Figure 2 shows also on page 10. In both methods, capacitor Cn voltage represents the inductor total currents. A droop amplifier converts Cn voltage into an internal current source with the gain set by resistor Ri. The current source is used for load line implementation, current monitor and overcurrent protection. Figure 9 shows the load line implementation. The ISL62883 drives a current source Idroop out of the FB pin, described by Equation 1. 2xV Cn I droop = ---------------Ri (EQ. 1) When using inductor DCR current sensing, a single NTC element is used to compensate the positive temperature coefficient of the copper winding thus sustaining the load line accuracy with reduced cost. Idroop flows through resistor Rdroop and creates a voltage drop of: V droop = R droop × I droop (EQ. 2) Vdroop is the droop voltage required to implement load line. Changing Rdroop or scaling Idroop can both change the load line slope. Since Idroop also sets the overcurrent protection level, it is recommended to first scale Idroop based on OCP requirement, then select an appropriate Rdroop value to obtain the desired load line slope. FN6891.4 June 21, 2011 ISL62883, ISL62883B Differential Sensing Figure 9 also shows the differential voltage sensing scheme. VCCSENSE and VSSSENSE are the remote voltage sensing signals from the processor die. A unity gain differential amplifier senses the VSSSENSE voltage and add it to the DAC output. The error amplifier regulates the inverting and the non-inverting input voltages to be equal as shown in Equation 3: VCC SENSE + V droop = V DAC + VSS SENSE (EQ. 3) The ISL62883 will adjust the phase pulse-width relative to the other phases to make VISEN1 = VISEN2 = VISEN3, thus to achieve IL1 = IL2 = IL3, when there are Rdcr1 = Rdcr2 = Rdcr3 and Rpcb1 = Rpcb2 = Rpcb3. Using same components for L1, L2 and L3 will provide a good match of Rdcr1, Rdcr2 and Rdcr3. Board layout will determine Rpcb1, Rpcb2 and Rpcb3. It is recommended to have symmetrical layout for the power delivery path between each inductor and the output voltage rail, such that Rpcb1 = Rpcb2 = Rpcb3. Rewriting Equation 3 and substitution of Equation 2 gives: VCC SENSE – VSS SENSE = V DAC – R droop × I droop (EQ. 4) ISEN3 Equation 4 is the exact equation required for load line implementation. Cs The VCCSENSE and VSSSENSE signals come from the processor die. The feedback will be open circuit in the absence of the processor. As shown in Figure 9, it is recommended to add a “catch” resistor to feed the VR local output voltage back to the compensator, and add another “catch” resistor to connect the VR local output ground to the RTN pin. These resistors, typically 10Ω~100Ω, will provide voltage feedback if the system is powered up without a processor installed. INTERNAL TO IC ISEN2 Cs Rdcr3 L3 INTERNAL TO IC ISEN2 L2 Phase2 Rs Rdcr2 Rdcr1 L1 Phase1 Rs Rpcb2 Vo Rpcb1 IL1 Cs FIGURE 10. CURRENT BALANCING CIRCUIT The ISL62883 monitors individual phase average current by monitoring the ISEN1, ISEN2, and ISEN3 voltages. Figure 10 shows the current balancing circuit recommended for ISL62883. Each phase node voltage is averaged by a low-pass filter consisting of Rs and Cs, and presented to the corresponding ISEN pin. Rs should be routed to inductor phase-node pad in order to eliminate the effect of phase node parasitic PCB DCR. Equations 5 thru 7 give the ISEN pin voltages: V ISEN1 = ( R dcr1 + R pcb1 ) × I L1 (EQ. 5) V ISEN2 = ( R dcr2 + R pcb2 ) × I L2 (EQ. 6) V ISEN3 = ( R dcr3 + R pcb3 ) × I L3 (EQ. 7) V3n Rs V2p L2 Rdcr2 Rpcb2 IL2 Rs Vo V2n Rs ISEN1 V1p Phase1 Rs Cs L1 Rs Rdcr1 IL1 Rpcb1 V1n Sometimes, it is difficult to implement symmetrical layout. For the circuit shown in Figure 10, asymmetric layout causes different Rpcb1, Rpcb2 and Rpcb3 thus current imbalance. Figure 11 shows a differential-sensing current balancing circuit recommended for ISL62883. The current sensing traces should be routed to the inductor pads so they only pick up the inductor DCR voltage. Each ISEN pin sees the average voltage of three sources: its own phase inductor phase-node pad, and the other two phases inductor output side pads. Equations 8 thru 10 give the ISEN pin voltages: V ISEN1 = V 1p + V 2n + V 3n (EQ. 8) V ISEN2 = V 1n + V 2p + V 3n (EQ. 9) V ISEN3 = V 1n + V 2n + V 3p (EQ. 10) The ISL62883 will make VISEN1 = VISEN2 = VISEN3 as in: V 1p + V 2n + V 3n = V 1n + V 2p + V 3n (EQ. 11) V 1n + V 2p + V 3n = V 1n + V 2n + V 3p (EQ. 12) Rewriting Equation 11 gives: V 1p – V 1n = V 2p – V 2n (EQ. 13) and rewriting Equation 12 gives: where Rdcr1, Rdcr2 and Rdcr3 are inductor DCR; Rpcb1, Rpcb2 and Rpcb3 are parasitic PCB DCR between the inductor output side pad and the output voltage rail; and IL1, IL2 and IL3 are inductor average currents. 15 IL3 Rpcb3 FIGURE 11. DIFFERENTIAL-SENSING CURRENT BALANCING CIRCUIT IL2 Cs ISEN1 Rpcb3 IL3 Cs Rdcr3 Rs Phase3 ISEN3 L3 Rs Phase2 Rs Phase Current Balancing Rs V3p Phase3 Rs V 2p – V 2n = V 3p – V 3n (EQ. 14) Combining Equations 13 and 14 gives: V 1p – V 1n = V 2p – V 2n = V 3p – V 3n (EQ. 15) Therefore: R dcr1 × I L1 = R dcr2 × I L2 = R dcr3 × I L3 (EQ. 16) FN6891.4 June 21, 2011 ISL62883, ISL62883B Current balancing (IL1 = IL2 = IL3) will be achieved when there is Rdcr1 = Rdcr2 = Rdcr3. Rpcb1, Rpcb2 and Rpcb3 will not have any effect. REP RATE = 10kHz Since the slave ripple capacitor voltages mimic the inductor currents, R3™ modulator can naturally achieve excellent current balancing during steady state and dynamic operations. Figure 12 shows current balancing performance of the ISL62883 evaluation board with load transient of 12A/51A at different rep rates. The inductor currents follow the load current dynamic change with the output capacitors supplying the difference. The inductor currents can track the load current well at low rep rate, but cannot keep up when the rep rate gets into the hundred-kHz range, where it’s out of the control loop bandwidth. The controller achieves excellent current balancing in all cases. CCM Switching Frequency REP RATE = 25kHz REP RATE = 50kHz The Rfset resistor between the COMP and the VW pins sets the sets the VW windows size, therefore sets the switching frequency. When the ISL62883 is in continuous conduction mode (CCM), the switching frequency is not absolutely constant due to the nature of the R3™ modulator. As explained in the Multiphase R3™ Modulator section, the effective switching frequency will increase during load insertion and will decrease during load release to achieve fast response. On the other hand, the switching frequency is relatively constant at steady state. Variation is expected when the power stage condition, such as input voltage, output voltage, load, etc. changes. The variation is usually less than 15% and doesn’t have any significant effect on output voltage ripple magnitude. Equation 17 gives an estimate of the frequency-setting resistor Rfset value. 8kΩ Rfset gives approximately 300kHz switching frequency. Lower resistance gives higher switching frequency. R fset ( kΩ ) = ( Period ( μs ) – 0.29 ) × 2.65 REP RATE = 100kHz (EQ. 17) Phase Count Configurations The ISL62883 can be configured for 3-, 2- or 1-phase operation. For 2-phase configuration, tie the PWM3 pin to 5V. Phase-1 and Phase-2 PWM pulses are 180° out-of-phase. In this configuration, the ISEN3/FB2 pin (pin 9) serves the FB2 function. For 1-phase configuration, tie the PWM3 and ISEN2 pins to 5V. In this configuration, only Phase-1 is active. The ISEN3/FB2, ISEN2, and ISEN1 pins are not used because there is no need for either current balancing or FB2 function. REP RATE = 200kHz FIGURE 12. ISL62883 EVALUATION BOARD CURRENT BALANCING DURING DYNAMIC OPERATION. CH1: IL1, CH2: ILOAD, CH3: IL2, CH4: IL3 16 FN6891.4 June 21, 2011 ISL62883, ISL62883B Modes of Operation TABLE 2. ISL62883 MODES OF OPERATION CONFIGURATION OPERATIONAL MODE When the unfiltered version is 20mV below the filtered version, the controller knows there is a fast voltage dip due to load insertion, hence issues an additional master clock signal to deliver a PWM pulse immediately. PSI# DPRSLPVR 0 0 2-phase CCM 0 1 1-phase DE 1 0 3-phase CCM Protections 1 1 1-phase DE 0 0 1-phase CCM The ISL62883 provides overcurrent, current-balance, undervoltage, overvoltage, and over-temperature protections. 0 1 1-phase DE 1 0 2-phase CCM 1 1 1-phase DE 0 0 1-phase CCM 0 1 1-phase DE 1 0 1-phase CCM 1 1 1-phase DE 3-phase Configuration 2-phase Configuration 1-phase Configuration Table 2 shows the ISL62883 operational modes, programmed by the logic status of the PSI# and the DPRSLPVR pins. In 3-phase configuration, the ISL62883 enters 2-phase CCM for (PSI# = 0 and DPRSLPVR = 0) by dropping PWM3 and operating phases 1 and 2 180° out-of-phase. It also reduces the overcurrent and the way-overcurrent protection levels to 2/3 of the initial values. The ISL62883 enters 1-phase DE mode when DPRSLPVR = 1. It drops phases 2 and 3, and reduces the overcurrent and the wayovercurrent protection levels to 1/3 of the initial values. In 2-phase configuration, the ISL62883 enters 1-phase CCM for (PSI# = 0 and DPRSLPVR = 0). It drops Phase-2 and reduces the overcurrent and the way-overcurrent protection levels to 1/2 of the initial values. The ISL62883 enters 1-phase DE mode when DPRSLPVR = 1 by dropping phase 2. In 1-phase configuration, the ISL62883 does not change the operational mode when the PSI# signal changes status. It enters 1-phase DE mode when DPRSLPVR = 1. Dynamic Operation The ISL62883 responds to VID changes by slewing to the new voltage at 5mV/µs slew rate. As the output approaches the VID command voltage, the dv/dt moderates to prevent overshoot. Geyserville-III transitions commands one LSB VID step (12.5mV) every 2.5µs, controlling the effective dv/dt at 5mv/µs. The ISL62883 is capable of 5mV/µs slew rate. When the ISL62883 is in DE mode, it will actively drive the output voltage up when the VID changes to a higher value. It’ll resume DE mode operation after reaching the new voltage level. If the load is light enough to warrant DCM, it will enter DCM after the inductor current has crossed zero for four consecutive cycles. The ISL62883 will remain in DE mode when the VID changes to a lower value. The output voltage will decay to the new value and the load will determine the slew rate. During load insertion response, the Fast Clock function increases the PWM pulse response speed. The ISL62883 monitors the VSEN pin voltage and compares it to 100ns - filtered version. 17 The R3™ modulator intrinsically has voltage feed forward. The output voltage is insensitive to a fast slew rate input voltage change. The ISL62883 determines overcurrent protection (OCP) by comparing the average value of the droop current Idroop with an internal current source threshold. It declares OCP when Idroop is above the threshold for 120µs. A resistor Rcomp from the COMP pin to GND programs the OCP current source threshold, as Table 3 shows. It is recommended to use the nominal Rcomp value. The ISL62883 detects the Rcomp value at the beginning of start up, and sets the internal OCP threshold accordingly. It remembers the Rcomp value until the VR_ON signal drops below the POR threshold. TABLE 3. ISL62883 OCP THRESHOLD Rcomp NOMINAL MIN. (kΩ) (kΩ) OCP THRESHOLD (µA) MAX. (kΩ) 1-PHASE MODE 2-PHASE MODE 3-PHASE MODE none none 20 40 60 320 400 480 22.7 45.3 68 210 235 260 20.7 41.3 62 155 165 175 18 36 54 104 120 136 20 37.33 56 78 85 92 22.7 38.7 58 62 66 70 20.7 42.7 64 45 50 55 18 44 66 The default OCP threshold is the value when Rcomp is not populated. It is recommended to scale the droop current Idroop such that the default OCP threshold gives approximately the desired OCP level, then use Rcomp to fine tune the OCP level if necessary. For overcurrent conditions above 2.5 times the OCP level, the PWM outputs will immediately shut off and PGOOD will go low to maximize protection. This protection is also referred to as way-overcurrent protection or fast-overcurrent protection, for short-circuit protections. The ISL62883 monitors the ISEN pin voltages to determine current-balance protection. If the ISEN pin voltage difference is greater than 9mV for 1ms, the controller will declare a fault and latch off. The ISL62883 will declare undervoltage (UV) fault and latch off if the output voltage is less than the VID set value by 300mV or more for 1ms. It’ll turn off the PWM outputs and dessert PGOOD. The ISL62883 has two levels of overvoltage protections. The first level of overvoltage protection is referred to as PGOOD overvoltage protection. If the output voltage exceeds the VID set FN6891.4 June 21, 2011 ISL62883, ISL62883B value by +200mV for 1ms, the ISL62883 will declare a fault and dessert PGOOD. The ISL62883 takes the same actions for all of the above fault protections: desertion of PGOOD and turn-off of the high-side and low-side power MOSFETs. Any residual inductor current will decay through the MOSFET body diodes. These fault conditions can be reset by bringing VR_ON low or by bringing VDD below the POR threshold. When VR_ON and VDD return to their high operating levels, a soft-start will occur. The second level of overvoltage protection is different. If the output voltage exceeds 1.55V, the ISL62883 will immediately declare an OV fault, dessert PGOOD, and turn on the low-side power MOSFETs. The low-side power MOSFETs remain on until the output voltage is pulled down below 0.85V when all power MOSFETs are turned off. If the output voltage rises above 1.55V again, the protection process is repeated. This behavior provides the maximum amount of protection against shorted high-side power MOSFETs while preventing output ringing below ground. Resetting VR_ON cannot clear the 1.55V OVP. Only resetting VDD will clear it. The 1.55V OVP is active all the time when the controller is enabled, even if one of the other faults have been declared. This ensures that the processor is protected against high-side power MOSFET leakage while the MOSFETs are commanded off. The ISL62883 has a thermal throttling feature. If the voltage on the NTC pin goes below the 1.18V OT threshold, the VR_TT# pin is pulled low indicating the need for thermal throttling to the system. No other action is taken within the ISL62883 in response to NTC pin voltage. Table 4 summarizes the fault protections. . TABLE 4. FAULT PROTECTION SUMMARY FAULT TYPE FAULT DURATION BEFORE PROTECTION Overcurrent 120µs Way-Overcurrent (2.5xOC) <2µs Overvoltage +200mV 1ms PROTECTION ACTION FAULT RESET PWM tri-state, PGOOD latched low VR_ON toggle or VDD toggle Phase Current Unbalance Immediately Over-Temperature Low-side MOSFET VDD toggle on until Vcore <0.85V, then PWM tri-state, PGOOD latched low. 1ms N/A Current Monitor The ISL62883 provides the current monitor function. The IMON pin outputs a high-speed analog current source that is 3 times of the droop current flowing out of the FB pin. Thus Equation 18: I IMON = 3 × I droop (EQ. 18) 18 The IMON pin voltage range is 0V to 1.1V. A clamp circuit prevents the IMON pin voltage from going above 1.1V. FB2 Function The FB2 function is only available when the ISL62883 is in 2-phase configuration, when pin 9 serves the FB2 function instead of the ISEN3 function. C1 R2 CONTROLLER IN 2-PHASE MODE C2 R3 VSEN C3.1 FB2 FB Vref C2 R3 C3.2 R1 E/A C1 R2 CONTROLLER IN 1-PHASE MODE VSEN COMP C3.1 FB2 C3.2 R1 FB E/A COMP Vref FIGURE 13. FB2 FUNCTION IN 2-PHASE MODE Figure 13 shows the FB2 function. A switch (called FB2 switch) turns on to short the FB and the FB2 pins when the controller is in 2-phase mode. Capacitors C3.1 and C3.2 are in parallel, serving as part of the compensator. When the controller enters 1-phase mode, the FB2 switch turns off, removing C3.2 and leaving only C3.1 in the compensator. The compensator gain will increase with the removal of C3.2. By properly sizing C3.1 and C3.2, the compensator cab be optimal for both 2-phase mode and 1-phase mode. When the FB2 switch is off, C3.2 is disconnected from the FB pin. However, the controller still actively drives the FB2 pin voltage to follow the FB pin voltage such that C3.2 voltage always follows C3.1 voltage. When the controller turns on the FB2 switch, C3.2 will be reconnected to the compensator smoothly. The FB2 function ensures excellent transient response in both 2-phase mode and 1-phase mode. If one decides not to use the FB2 function, simply populate C3.1 only. Adaptive Body Diode Conduction Time Reduction Undervoltage -300mV Overvoltage 1.55V As Figures 1 and 2 show, a resistor Rimon is connected to the IMON pin to convert the IMON pin current to voltage. A capacitor can be paralleled with Rimon to filter the voltage information. The IMVP-6.5™ specification requires that the IMON voltage information be referenced to VSSSENSE. In DCM, the controller turns off the low-side MOSFET when the inductor current approaches zero. During on-time of the low-side MOSFET, phase voltage is negative and the amount is the MOSFET Rdson voltage drop, which is proportional to the inductor current. A phase comparator inside the controller monitors the phase voltage during on-time of the low-side MOSFET and compares it with a threshold to determine the zero-crossing point of the inductor current. If the inductor current has not reached zero when the low-side MOSFET turns off, it’ll flow through the low-side MOSFET body diode, causing the phase node to have a larger voltage drop until it decays to zero. If the inductor current has crossed zero and reversed the direction when the low-side MOSFET turns off, it’ll flow through the high-side MOSFET body diode, causing the phase node to have a spike until it decays to zero. The controller continues monitoring the phase voltage after turning off the low-side MOSFET and adjusts the phase FN6891.4 June 21, 2011 ISL62883, ISL62883B comparator threshold voltage accordingly in iterative steps such that the low-side MOSFET body diode conducts for approximately 40ns to minimize the body diode-related loss. droop amplifier. The preferred values are Ris = 82.5Ω and Cis = 0.01µF. Slight deviations from the recommended values are acceptable. Large deviations may result in instability. Overshoot Reduction Function Inductor DCR Current-Sensing Network The ISL62883 has an optional overshoot reduction function. Using RBIAS = 47kΩ enables this function and using RBIAS = 147kΩ disables this function. When a load release occurs, the energy stored in the inductors will dump to the output capacitor, causing output voltage overshoot. The inductor current freewheels through the low-side MOSFET during this period of time. The overshoot reduction function turns off the low-side MOSFET during the output voltage overshoot, forcing the inductor current to freewheel through the low-side MOSFET body diode. Since the body diode voltage drop is much higher than MOSFET Rdson voltage drop, more energy is dissipated on the low-side MOSFET therefore the output voltage overshoot is lower. If the overshoot reduction function is enabled, the ISL62883 monitors the COMP pin voltage to determine the output voltage overshoot condition. The COMP voltage will fall and hit the clamp voltage when the output voltage overshoots. The ISL62883 will turn off LGATE1 and LGATE2, and tri-state PWM3 when COMP is being clamped. All the low-side MOSFETs in the power stage will be turned off. When the output voltage has reached its peak and starts to come down, the COMP voltage starts to rise and is no longer clamped. The ISL62883 will resume normal PWM operation. When PSI# is low, indicating a low power state of the CPU, the controller will disable the overshoot reduction function as large magnitude transient event is not expected and overshoot is not a concern. While the overshoot reduction function reduces the output voltage overshoot, energy is dissipated on the low-side MOSFET, causing additional power loss. The more frequent transient event, the more power loss dissipated on the low-side MOSFET. The MOSFET may face severe thermal stress when transient events happen at a high repetitive rate. User discretion is advised when this function is enabled. Key Component Selection RBIAS The ISL62883 uses a resistor (1% or better tolerance is recommended) from the RBIAS pin to GND to establish highly accurate reference current sources inside the IC. Using RBIAS = 47kΩ enables the overshoot reduction function and using RBIAS = 147kΩ disables this function. Do not connect any other components to this pin. Do not connect any capacitor to the RBIAS pin as it will create instability. Care should be taken in layout that the resistor is placed very close to the RBIAS pin and that a good quality signal ground is connected to the opposite side of the RBIAS resistor. Ris and Cis As Figures 1 and 2, show, the ISL62883 needs the Ris - Cis network across the ISUM+ and the ISUM- pins to stabilize the 19 Phase1 Phase2 Phase3 Rsum Rsum ISUM+ Rsum L L L Rntcs Rp DCR DCR DCR Cn Vcn Rntc Ro ISUM- Ri Ro Ro Io FIGURE 14. DCR CURRENT-SENSING NETWORK Figure 14 shows the inductor DCR current-sensing network for a 3-phase solution. An inductor current flows through the DCR and creates a voltage drop. Each inductor has two resistors in Rsum and Ro connected to the pads to accurately sense the inductor current by sensing the DCR voltage drop. The Rsum and Ro resistors are connected in a summing network as shown, and feed the total current information to the NTC network (consisting of Rntcs, Rntc and Rp) and capacitor Cn. Rntc is a negative temperature coefficient (NTC) thermistor, used to temperature-compensate the inductor DCR change. The inductor output side pads are electrically shorted in the schematic, but have some parasitic impedance in actual board layout, which is why one cannot simply short them together for the current-sensing summing network. It is recommended to use 1Ω~10Ω Ro to create quality signals. Since Ro value is much smaller than the rest of the current sensing circuit, the following analysis will ignore it for simplicity. The summed inductor current information is presented to the capacitor Cn. Equations 19 thru 23 describe the frequency-domain relationship between inductor total current Io(s) and Cn voltage VCn(s): ⎛ ⎞ R ntcnet ⎜ DCR⎟ V Cn ( s ) = ⎜ ----------------------------------------- × ------------⎟ × I o ( s ) × A cs ( s ) N ⎟ R sum ⎜ ⎝ R ntcnet + ------------⎠ N ( R ntcs + R ntc ) × R p R ntcnet = --------------------------------------------------R ntcs + R ntc + R p s 1 + -----ωL A cs ( s ) = ---------------------s 1 + -----------ω sns (EQ. 19) (EQ. 20) (EQ. 21) where N is the number of phases. FN6891.4 June 21, 2011 ISL62883, ISL62883B DCR ω L = -----------L io (EQ. 22) 1 ω sns = -----------------------------------------------------R sum R ntcnet × -------------N ----------------------------------------- × C n R sum R ntcnet + -------------N (EQ. 23) Vo Transfer function Acs(s) always has unity gain at DC. The inductor DCR value increases as the winding temperature increases, giving higher reading of the inductor DC current. The NTC Rntc values decreases as its temperature decreases. Proper selections of Rsum, Rntcs, Rp and Rntc parameters ensure that VCn represent the inductor total DC current over the temperature range of interest. FIGURE 16. LOAD TRANSIENT RESPONSE WHEN Cn IS TOO SMALL io There are many sets of parameters that can properly temperaturecompensate the DCR change. Since the NTC network and the Rsum resistors form a voltage divider, Vcn is always a fraction of the inductor DCR voltage. It is recommended to have a higher ratio of Vcn to the inductor DCR voltage, so the droop circuit has higher signal level to work with. A typical set of parameters that provide good temperature compensation are: Rsum = 3.65kΩ, Rp = 11kΩ, Rntcs = 2.61kΩ and Rntc = 10kΩ (ERT-J1VR103J). The NTC network parameters may need to be fine tuned on actual boards. One can apply full load DC current and record the output voltage reading immediately; then record the output voltage reading again when the board has reached the thermal steady state. A good NTC network can limit the output voltage drift to within 2mV. It is recommended to follow the Intersil evaluation board layout and current-sensing network parameters to minimize engineering time. VCn(s) also needs to represent real-time Io(s) for the controller to achieve good transient response. Transfer function Acs(s) has a pole ωsns and a zero ωL. One needs to match ωL and ωsns so Acs(s) is unity gain at all frequencies. By forcing ωL equal to ωsns and solving for the solution, Equation 24 solves for the value of Cn. L C n = -----------------------------------------------------------R sum R ntcnet × -------------N ----------------------------------------- × DCR R sum R ntcnet + -------------N Vo FIGURE 17. LOAD TRANSIENT RESPONSE WHEN Cn IS TOO LARGE For example, given N = 3, Rsum = 3.65kΩ, Rp = 11kΩ, Rntcs = 2.61kΩ, Rntc = 10kΩ, DCR = 0.88mΩ and L = 0.36µH, Equation 24 gives Cn = 0.406µF. Assuming the compensator design is correct, Figure 15 shows the expected load transient response waveforms if Cn is correctly selected. When the load current Icore has a square change, the output voltage Vcore also has a square response. If Cn value is too large or too small, VCn(s) will not accurately represent real-time Io(s) and will worsen the transient response. Figure 16 shows the load transient response when Cn is too small. Vcore will sag excessively upon load insertion and may create a system failure. Figure 17 shows the transient response when Cn is too large. Vcore is sluggish in drooping to its final value. There will be excessive overshoot if load insertion occurs during this time, which may potentially hurt the CPU reliability. (EQ. 24) io io Vo iL Vo RING BACK FIGURE 18. OUTPUT VOLTAGE RING BACK PROBLEM FIGURE 15. DESIRED LOAD TRANSIENT RESPONSE WAVEFORMS 20 FN6891.4 June 21, 2011 ISL62883, ISL62883B ISUM+ Rntcs Cn.1 Phase1 Phase2 Phase3 L L L DCR DCR DCR Cn.2 Vcn Rp Rntc Resistor Current-Sensing Network Rn OPTIONAL Rsum Ri Rsum ISUM- ISUM+ Rsum Rip Cip Rsen Rsen Rsen Vcn Ro Ro FIGURE 19. OPTIONAL CIRCUITS FOR RING BACK REDUCTION Figure 19 shows two optional circuits for reduction of the ring back. Rip and Cip form an R-C branch in parallel with Ri, providing a lower impedance path than Ri at the beginning of io change. Rip and Cip do not have any effect at steady state. Through proper selection of Rip and Cip values, idroop can resemble io rather than iL, and Vo will not ring back. The recommended value for Rip is100Ω. Cip should be determined through tuning the load transient response waveforms on an actual board. The recommended range for Cip is 100pF~2000pF. Cn is the capacitor used to match the inductor time constant. It usually takes the parallel of two (or more) capacitors to get the desired value. Figure 19 shows that two capacitors Cn.1 and Cn.2 are in parallel. Resistor Rn is an optional component to reduce the Vo ring back. At steady state, Cn.1+Cn.2 provides the desired Cn capacitance. At the beginning of io change, the effective capacitance is less because Rn increases the impedance of the Cn.1 branch. As explained in Figure 16, Vo tends to dip when Cn is too small, and this effect will reduce the Vo ring back. This effect is more pronounced when Cn.1 is much larger than Cn.2. It is also more pronounced when Rn is bigger. However, the presence of Rn increases the ripple of the Vn signal if Cn.2 is too small. It is recommended to keep Cn.2 greater than 2200pF. Rn value usually is a few ohms. Cn.1, Cn.2 and Rn values should be determined through tuning the load transient response waveforms on an actual board. 21 ISUM- Ro OPTIONAL Figure 18 shows the output voltage ring back problem during load transient response. The load current io has a fast step change, but the inductor current iL cannot accurately follow. Instead, iL responds in first order system fashion due to the nature of current loop. The ESR and ESL effect of the output capacitors makes the output voltage Vo dip quickly upon load current change. However, the controller regulates Vo according to the droop current idroop, which is a real-time representation of iL; therefore it pulls Vo back to the level dictated by iL, causing the ring back problem. This phenomenon is not observed when the output capacitor have very low ESR and ESL, such as all ceramic capacitors. Cn Ri Io FIGURE 20. RESISTOR CURRENT-SENSING NETWORK Figure 20 shows the resistor current-sensing network for a 3-phase solution. Each inductor has a series current-sensing resistor Rsen. Rsum and Ro are connected to the Rsen pads to accurately capture the inductor current information. The Rsum and Ro resistors are connected to capacitor Cn. Rsum and Cn form a a filter for noise attenuation. Equations 25 thru 27 give VCn(s) expression: R sen V Cn ( s ) = ------------ × I o ( s ) × A Rsen ( s ) N 1 A Rsen ( s ) = ---------------------s 1 + -----------ω sns 1 ω Rsen = --------------------------R sum -------------- × C n N (EQ. 25) (EQ. 26) (EQ. 27) Transfer function ARsen(s) always has unity gain at DC. Current-sensing resistor Rsen value will not have significant variation over temperature, so there is no need for the NTC network. The recommended values are Rsum = 1kΩ and Cn = 5600pF. Overcurrent Protection Refer to Equation 1 and Figures 9, 14 and 20; resistor Ri sets the droop current Idroop. Table 3 shows the internal OCP threshold. It is recommended to design Idroop without using the Rcomp resistor. For example, the OCP threshold is 60µA for 3-phase solution. We will design Idroop to be 38.8µA at full load, so the OCP trip level is 1.55 times of the full load current. For inductor DCR sensing, Equation 28 gives the DC relationship of Vcn(s) and Io(s). ⎛ ⎞ R ntcnet ⎜ DCR⎟ V Cn = ⎜ ----------------------------------------- × ------------⎟ × I o R sum N ⎟ ⎜ ⎝ R ntcnet + ------------⎠ N (EQ. 28) FN6891.4 June 21, 2011 ISL62883, ISL62883B Substitution of Equation 28 into Equation 1 gives Equation 29: R ntcnet DCR 2 I droop = ----- × ----------------------------------------- × ------------ × I o N R sum Ri R ntcnet + -------------N (EQ. 29) Therefore: 2R ntcnet × DCR × I o R i = -------------------------------------------------------------------------------R sum N × ⎛ R ntcnet + --------------⎞ × I droop ⎝ N ⎠ (EQ. 30) Substitution of Equation 20 and application of the OCP condition in Equation 30 gives Equation 31: ( R ntcs + R ntc ) × R p 2 × --------------------------------------------------- × DCR × I omax R ntcs + R ntc + R p R i = ------------------------------------------------------------------------------------------------------------------------⎛ ( R ntcs + R ntc ) × R p R sum⎞ N × ⎜ --------------------------------------------------- + --------------⎟ × I droopmax N ⎠ ⎝ R ntcs + R ntc + R p (EQ. 31) where Iomax is the full load current, Idroopmax is the corresponding droop current. For example, given N = 3, Rsum = 3.65kΩ, Rp = 11kΩ, Rntcs = 2.61kΩ, Rntc = 10kΩ, DCR = 0.88mΩ, Iomax = 51A and Idroopmax = 40.9µA, Equation 31 gives Ri = 606Ω. For resistor sensing, Equation 32 gives the DC relationship of Vcn(s) and Io(s). R sen V Cn = ------------ × I o N (EQ. 32) Substitution of Equation 32 into Equation 1 gives Equation 33: 2 R sen I droop = ----- × ------------ × I o N Ri (EQ. 33) For resistor sensing, substitution of Equation 33 into Equation 2 gives the load line slope expression: 2R sen × R droop V droop LL = ------------------ = ----------------------------------------N × Ri Io Substitution of Equation 30 and rewriting Equation 36, or substitution of Equation 34 and rewriting Equation 37 gives the same result in Equation 38: Io R droop = ---------------- × LL I droop 2R sen × I o R i = --------------------------N × I droop (EQ. 34) Substitution of Equation 34 and application of the OCP condition in Equation 30 gives: 2R sen × I omax R i = -------------------------------------N × I droopmax (EQ. 35) where Iomax is the full load current, Idroopmax is the corresponding droop current. For example, given N = 3, Rsen = 1mΩ, Iomax = 51A and Idroopmax = 40.9µA, Equation 35 gives Ri = 831Ω. A resistor from COMP to GND can adjust the internal OCP threshold, providing another dimension of fine-tune flexibility. Table 3 shows the detail. It is recommended to scale Idroop such that the default OCP threshold gives approximately the desired OCP level, then use Rcomp to fine tune the OCP level if necessary. (EQ. 38) One can use the full load condition to calculate Rdroop. For example, given Iomax = 51A, Idroopmax = 40.9µA and LL = 1.9mΩ, Equation 38 gives Rdroop = 2.37kΩ. It is recommended to start with the Rdroop value calculated by Equation 38, and fine tune it on the actual board to get accurate load line slope. One should record the output voltage readings at no load and at full load for load line slope calculation. Reading the output voltage at lighter load instead of full load will increase the measurement error. Current Monitor Refer to Equation 18 for the IMON pin current expression. Refer to Figures 1 and 2, the IMON pin current flows through Rimon. The voltage across Rimon is expressed in Equation 39: V Rimon = 3 × I droop × R imon (EQ. 39) Rewriting Equation 38 gives Equation 40: Io I droop = ------------------ × LL R droop Therefore: (EQ. 37) (EQ. 40) Substitution of Equation 40 into Equation 39 gives Equation 41: 3I o × LL V Rimon = --------------------- × R imon R droop (EQ. 41) Rewriting Equation 41 and application of full load condition gives Equation 42: V Rimon × R droop R imon = -------------------------------------------3I o × LL (EQ. 42) For example, given LL = 1.9mΩ, Rdroop = 2.37kΩ, VRimon = 963mV at Iomax = 51A, Equation 42 gives Rimon = 7.85kΩ. A capacitor Cimon can be paralleled with Rimon to filter the IMON pin voltage. The RimonCimon time constant is the user’s choice. It is recommended to have a time constant long enough such that switching frequency ripples are removed. Compensator Load Line Slope Refer to Figure 9. For inductor DCR sensing, substitution of Equation 29 into Equation 2 gives the load line slope expression: 2R droop R ntcnet V droop DCR LL = ------------------ = ---------------------- × ----------------------------------------- × -----------Io Ri R sum N R ntcnet + -------------N 22 (EQ. 36) Figure 15 shows the desired load transient response waveforms. Figure 21 shows the equivalent circuit of a voltage regulator (VR) with the droop function. A VR is equivalent to a voltage source (= VID) and output impedance Zout(s). If Zout(s) is equal to the load line slope LL, i.e. constant output impedance, in the entire frequency range, Vo will have square response when Io has a square change. FN6891.4 June 21, 2011 ISL62883, ISL62883B L Zout(s)=LL VID VR io Vo Q1 Vin Load Vo GATE Q2 DRIVER io Cout LOAD LINE SLOPE 20 Ω FIGURE 21. VOLTAGE REGULATOR EQUIVALENT CIRCUIT EA Mod. Comp Intersil provides a Microsoft Excel-based spreadsheet to help design the compensator and the current sensing network, so the VR achieves constant output impedance as a stable system. Please contact Intersil Application support at www.intersil.com/design/. Figure 24 shows a screenshot of the spreadsheet. A VR with active droop function is a dual-loop system consisting of a voltage loop and a droop loop which is a current loop. However, neither loop alone is sufficient to describe the entire system. The spreadsheet shows two loop gain transfer functions, T1(s) and T2(s), that describe the entire system. Figure 22 conceptually shows T1(s) measurement set-up and Figure 23 conceptually shows T2(s) measurement set-up. The VR senses the inductor current, multiplies it by a gain of the load line slope, then adds it on top of the sensed output voltage and feeds it to the compensator. T(1) is measured after the summing node, and T2(s) is measured in the voltage loop before the summing node. The spreadsheet gives both T1(s) and T2(s) plots. However, only T2(s) can be actually measured on an ISL62883 regulator. T1(s) is the total loop gain of the voltage loop and the droop loop. It always has a higher crossover frequency than T2(s) and has more meaning of system stability. T2(s) is the voltage loop gain with closed droop loop. It has more meaning of output voltage response. Design the compensator to get stable T1(s) and T2(s) with sufficient phase margin, and output impedance equal or smaller than the load line slope. 23 VID ISOLATION TRANSFORMER CHANNEL B LOOP GAIN = CHANNEL A CHANNEL A NETWORK ANALYZER CHANNEL B EXCITATION OUTPUT FIGURE 22. LOOP GAIN T1(s) MEASUREMENT SET-UP L Vo Q1 Vin GATE Q2 DRIVER Cout io LOAD LINE SLOPE 20 EA Mod. Comp Ω VID ISOLATION TRANSFORMER CHANNEL B LOOP GAIN = CHANNEL A CHANNEL A NETWORK ANALYZER CHANNEL B EXCITATION OUTPUT FIGURE 23. LOOP GAIN T2(s) MEASUREMENT SET-UP FN6891.4 June 21, 2011 Compensation & Current Sensing Network Design for Intersil Multiphase R^3 Regulators for IMVP-6.5 24 Changing the settings in red requires deep understanding of control loop design Place the 2nd compensator pole fp2 at: 2.2 xfs (Switching Frequency) Tune Ki to get the desired loop gain bandwidth Tune the compensator gain factor Ki: (Recommended Ki range is 0.8~2) 1.3 Loop Gain, Gain Curve 0DJQLWXGHPRKP 7V 7V ( 3KDVHGHJUHH ( ( )UHTXHQF\+] ( ( ( Loop Gain, Phase Curve 7V 7V ( FN6891.4 June 21, 2011 ( ( )UHTXHQF\+] ( ( ( ( ( )UHTXHQF\+] ( ( ( ( Output Impedance, Phase Curve ( ( ( ( )UHTXHQF\+] ( Operation Parameters Inductor DCR 0.88 m : Rsum 3.65 k : Rntc 10 k : Rntcs 2.61 k : Rp 11 k : Recommended Value Cn 0.406 uF Ri 606.036 : 3KDVHGHJUHH *DLQG% Output Impedance, Gain Curve Current Sensing Network Parameters ( FIGURE 24. SCREENSHOT OF THE COMPENSATOR DESIGN SPREADSHEET User Selected Value Cn 0.406 uF Ri 604 : ISL62883, ISL62883B Jia Wei, [email protected], 919-405-3605 Attention: 1. "Analysis ToolPak" Add-in is required. To turn on, go to Tools--Add-Ins, and check "Analysis ToolPak" 2. Green cells require user input Compensator Parameters Operation Parameters Controller Part Number: ISL6288x § s · § s · ¸ ¸ ¨1 KZi Zi ¨¨1 Phase Number: 3 2Sf z1 ¸¹ ¨© 2Sf z 2 ¸¹ © AV ( s ) Vin: 12 volts § s ·¸ §¨ s ·¸ Vo: 1.15 volts 1 s ¨1 ¨ 2Sf p1 ¸¹ ¨© 2Sf p 2 ¸¹ © Full Load Current: 51 Amps Estimated Full-Load Efficiency: 87 % Number of Output Bulk Capacitors: 4 Recommended Value User-Selected Value Capacitance of Each Output Bulk Capacitor: 270 uF R1 2.369 k : R1 2.37 k : ESR of Each Output Bulk Capacitor: 4.5 m : ESL of Each Output Bulk Capacitor: 0.6 nH R2 338.213 k : R2 324 k : Number of Output Ceramic Capacitors: 24 R3 0.530 k : R3 0.536 k : Capacitance of Each Output Ceramic Capacitor: 10 uF C1 148.140 pF C1 150 pF C2 455.369 pF C2 390 pF ESR of Each Output Ceramic Capacitor: 3 m: ESL of Each Output Ceramic Capacitor: 3 nH C3 40.069 pF C3 39 pF Switching Frequency: 300 kHz Use User-Selected Value (Y/N)? N Inductance Per Phase: 0.36 uH CPU Socket Resistance: 0.9 m : Desired Load-Line Slope: 1.9 m : Performance and Stability Desired ISUM- Pin Current at Full Load: 40.9 uA T1 Bandwidth: 212kHz T2 Bandwidth: 66kHz (This sets the over-current protection level) T1 Phase Margin: 58.9° T2 Phase Margin: 89.3° ISL62883, ISL62883B Optional Slew Rate Compensation Circuit For 1-Tick VID Transition Rdroop Vcore OPTIONAL Ivid COMP VIDs Σ VDACDAC RTN X1 INTERNAL TO IC (EQ. 44) dV fb C out × LL dV core C vid × ------------ = ------------------------ × -----------------dt dt R droop (EQ. 45) and: Idroop_vid E/A –t ------------------------------⎞ dV fb ⎛ R ×C I vid ( t ) = C vid × ------------ × ⎜ 1 – e vid vid⎟ ⎜ ⎟ dt ⎝ ⎠ It is desired to let Ivid(t) cancel Idroop_vid(t). So there are: Rvid Cvid FB In the mean time, the Rvid-Cvid branch current Ivid time domain expression is shown in Equation 44: R vid × C vid = C out × LL VID<0:6> VSSSENSE VSS (EQ. 46) The result is expressed in Equation 47: R vid = R droop (EQ. 47) and: dV core C out × LL ----------------dt C vid = ------------------------ × -----------------R droop dV fb -----------dt VID<0:6> (EQ. 48) For example: given LL = 1.9mΩ, Rdroop = 2.37kΩ, Cout = 1320µF, dVcore/dt = 5mV/us and dVfb/dt = 15mV/µs, Equation 47 gives Rvid = 2.37kΩ and Equation 48 gives Cvid = 350pF. Vfb Ivid It’s recommended to select the calculated Rvid value and start with the calculated Cvid value and tweak it on the actual board to get the best performance. Vcore During normal transient response, the FB pin voltage is held constant, therefore is virtual ground in small signal sense. The Rvid-Cvid network is between the virtual ground and the real ground, and hence has no effect on transient response. Idroop_vid FIGURE 25. OPTIONAL SLEW RATE COMPENSATION CIRCUIT FOR1-TICK VID TRANSITION Voltage Regulator Thermal Throttling During a large VID transition, the DAC steps through the VIDs at a controlled slew rate of 2.5µs per tick (12.5mV), controlling output voltage Vcore slew rate at 5mV/µs. Figure 25 shows the waveforms of 1-tick VID transition. During 1-tick VID transition, the DAC output changes at approximately 15mV/µs slew rate, but the DAC cannot step through multiple VIDs to control the slew rate. Instead, the control loop response speed determines Vcore slew rate. Ideally, Vcore will follow the FB pin voltage slew rate. However, the controller senses the inductor current increase during the up transition, as the Idroop_vid waveform shows, and will droop the output voltage Vcore accordingly, making Vcore slew rate slow. Similar behavior occurs during the down transition. To control Vcore slew rate during 1-tick VID transition, one can add the Rvid-Cvid branch, whose current Ivid cancels Idroop_vid. When Vcore increases, the time domain expression of the induced Idroop change is expressed in Equation 43: 54uA 64uA VR_TT# SW1 NTC + VNTC - + RNTC Rs 1.24V SW2 1.20V INTERNAL TO ISL62882 FIGURE 26. CIRCUITRY ASSOCIATED WITH THE THERMAL THROTTLING FEATURE OF THE ISL62882 –t -------------------------⎞ C out × LL dV core ⎛ C × LL⎟ I droop ( t ) = ------------------------ × ------------------ × ⎜ 1 – e out ⎜ ⎟ dt R droop ⎝ ⎠ (EQ. 43) where Cout is the total output capacitance. 25 FN6891.4 June 21, 2011 ISL62883, ISL62883B Figure 26 shows the thermal throttling feature with hysteresis. An NTC network is connected between the NTC pin and GND. At low temperature, SW1 is on and SW2 connects to the 1.20V side. The total current flowing out of the NTC pin is 60µA. The voltage on NTC pin is higher than threshold voltage of 1.20V and the comparator output is low. VR_TT# is pulled up by the external resistor. When temperature increases, the NTC thermistor resistance decreases so the NTC pin voltage drops. When the NTC pin voltage drops below 1.20V, the comparator changes polarity and turns SW1 off and throws SW2 to 1.24V. This pulls VR_TT# low and sends the signal to start thermal throttle. There is a 6µA current reduction on NTC pin and 40mV voltage increase on threshold voltage of the comparator in this state. The VR_TT# signal will be used to change the CPU operation and decrease the power consumption. When the temperature drops down, the NTC thermistor voltage will go up. If NTC voltage increases to above 1.24V, the comparator will flip back. The external resistance difference in these two conditions is expressed in Equation 49: 1.24V 1.20V --------------- – --------------- = 2.96k 54μA 60μA (EQ. 49) One needs to properly select the NTC thermistor value such that the required temperature hysteresis correlates to 2.96kΩ resistance change. A regular resistor may need to be in series with the NTC thermistor to meet the threshold voltage values. For example, given Panasonic NTC thermistor with B = 4700, the resistance will drop to 0.03322 of its nominal at +105°C, and drop to 0.03956 of its nominal at +100°C. If the required temperature hysteresis is +105°C to +100°C, the required resistance of NTC will be: 2.96kΩ ------------------------------------------------------= 467kΩ ( 0.03956 – 0.03322 ) represent the DC current flowing through the inductors. Recommended values are Rs = 10kΩ and Cs = 0.22µF. Layout Guidelines Table 5 shows the layout considerations. The designators refer to the reference design shown in Figure 27. TABLE 5. LAYOUT CONSIDERATION PIN NAME LAYOUT CONSIDERATION EP GND Create analog ground plane underneath the controller and the analog signal processing components. Don’t let the power ground plane overlap with the analog ground plane. Avoid noisy planes/traces (e.g.: phase node) from crossing over/overlapping with the analog plane. 1 PGOOD No special consideration 2 PSI# No special consideration 3 RBIAS Place the Rbias resistor (R16) in general proximity of the controller. Low impedance connection to the analog ground plane. 4 VR_TT# No special consideration 5 NTC The NTC thermistor (R9) needs to be placed close to the thermal source that is monitor to determine thermal throttling. Usually it’s placed close to Phase-1 high-side MOSFET. 6 VW Place the capacitor (C4) across VW and COMP in close proximity of the controller 7 COMP 8 FB Place the compensator components (C3, C6 R7, R11, R10 and C11) in general proximity of the controller. 9 (EQ. 50) Therefore a larger value thermistor, such as 470k NTC should be used. At +105°C, 470kΩ NTC resistance becomes (0.03322×470kΩ) = 15.6kΩ. With 60µA on the NTC pin, the voltage is only (15.6kΩ×60µA) = 0.937V. This value is much lower than the threshold voltage of 1.20V. Therefore, a regular resistor needs to be in series with the NTC. The required resistance can be calculated by Equation 51: 1.20V --------------- – 15.6kΩ = 4.4kΩ 60μA ISEN3/FB2 A capacitor (C7) decouples it to VSUM-. Place it in general proximity of the controller. An optional capacitor is placed between this pin and COMP. (It’s only used when the controller is configured 2-phase). Place it in general proximity of the controller. 10 ISEN2 A capacitor (C9) decouples it to VSUM-. Place it in general proximity of the controller. 11 ISEN1 A capacitor (C10) decouples it to VSUM-. Place it in general proximity of the controller. 12 VSEN 13 RTN Place the VSEN/RTN filter (C12, C13) in close proximity of the controller for good decoupling. (EQ. 51) 4.42k is a standard resistor value. Therefore, the NTC branch should have a 470k NTC and 4.42k resistor in series. The part number for the NTC thermistor is ERTJ0EV474J. It is a 0402 package. The NTC thermistor will be placed in the hot spot of the board. Current Balancing Refer to Figures 1 and 2. The ISL62883 achieves current balancing through matching the ISEN pin voltages. Rs and Cs form filters to remove the switching ripple of the phase node voltages. It is recommended to use rather long RsCs time constant such that the ISEN voltages have minimal ripple and 26 FN6891.4 June 21, 2011 ISL62883, ISL62883B TABLE 5. LAYOUT CONSIDERATION (Continued) TABLE 5. LAYOUT CONSIDERATION (Continued) PIN NAME LAYOUT CONSIDERATION PIN NAME LAYOUT CONSIDERATION 14 ISUM- 25 VCCP 15 ISUM+ Place the current sensing circuit in general proximity of the controller. Place C82 very close to the controller. Place NTC thermistors R42 next to Phase-1 inductor (L1) so it senses the inductor temperature correctly. Each phase of the power stage sends a pair of VSUM+ and VSUM- signals to the controller. Run these two signals traces in parallel fashion with decent width (>20mil). IMPORTANT: Sense the inductor current by routing the sensing circuit to the inductor pads. Route R63 and R71 to the Phase-1 side pad of inductor L1. Route R88 to the output side pad of inductor L1. Route R65 and R72 to the Phase-2 side pad of inductor L2. Route R90 to the output side pad of inductor L2. Route R67 and R73 to the Phase-3 side pad of inductor L3. Route R92 to the output side pad of inductor L3. If possible. Route the traces on a different layer from the inductor pad layer and use vias to connect the traces to the center of the pads. If no via is allowed on the pad, consider routing the traces into the pads from the inside of the inductor. The following drawings show the two preferred ways of routing current sensing traces. A capacitor (C22) decouples it to GND. Place it in close proximity of the controller. 26 LGATE2 27 VSSP2 Run these two traces in parallel fashion with decent width (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close. Recommend routing VSSP2 to the Phase-2 lowside MOSFET (Q5 and Q1) source pins instead of general power ground plane for better performance. 28 PHASE2 29 UGATE2 30 BOOT2 Use decent wide trace (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close. 31~37 VID0~6 No special consideration. VR_ON No special consideration. Inductor Inductor 38 39 40 Current-Sensing Traces DPRSLPVR No special consideration. CLK_EN# No special consideration. Other Phase Node Minimize phase node copper area. Don’t let the phase node copper overlap with/getting close to other sensitive traces. Cut the power ground plane to avoid overlapping with phase node copper. Other Vias Run these two traces in parallel fashion with decent width (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close. Recommend routing PHASE2 trace to the Phase-2 high-side MOSFET (Q4 and Q10) source pins instead of general Phase-2 node copper. Minimize the loop consisting of input capacitor, high-side MOSFETs and low-side MOSFETs (e.g.: C27, C33, Q2, Q8, Q3 and Q9). Current-Sensing Traces 16 VDD A capacitor (C16) decouples it to GND. Place it in close proximity of the controller. 17 VIN A capacitor (C17) decouples it to GND. Place it in close proximity of the controller. 18 IMON Place the filter capacitor (C21) close to the CPU. 19 BOOT1 Use decent wide trace (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close. 20 UGATE1 21 PHASE1 Run these two traces in parallel fashion with decent width (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close. Recommend routing PHASE1 trace to the Phase-1 high-side MOSFET (Q2 and Q8) source pins instead of general Phase-1 node copper. 22 VSSP1 23 LGATE1 24 PWM3 Run these two traces in parallel fashion with decent width (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close. Recommend routing VSSP1 to the Phase-1 lowside MOSFET (Q3 and Q9) source pins instead of general power ground plane for better performance. No special consideration. 27 FN6891.4 June 21, 2011 5 4 VIN EP Q6 7 6 10UF 10UF 10UF 10UF 10UF 10UF 10UF 10UF C44 270UF 10UF 10UF 10UF 10UF C57 270UF 10UF 10UF 10UF 10UF C52 270UF 10UF 10UF 10UF 10UF C39 270UF C40 10UF C49 10UF C60 10UF VSUM+ U3 IN VSUM- PHASE C48 C47 C56 C43 C55 C42 C54 C41 C50 C66 C65 C64 C63 C61 C74 C73 C72 C71 SAME RULE APPLIES TO OTHER PHASES FCCM PWM VCC GND LGATE ISL6208 4 FIGURE 27. 3-PHASE REFERENCE DESIGN IN 1UF BOOT C26 UGATE VSUM- IN ROUTE LGATE1 TRACE IN PARALLEL WITH THE VSSP1 TRACE GOING TO THE SOURCE OF Q3 AND Q9 ISEN3 R67 VSSSENSE 1 IRF7832 Q13 R73 IRF7832 NOTE: ROUTE UGATE1 TRACE IN PARALLEL WITH THE PHASE1 TRACE GOING TO THE SOURCE OF Q2 AND Q8 VSUM+ IN C68 1 10K R88 0.22UF LAYOUT 10K R92 L3 0.36UH Q7 PLACE NEAR L1 5 C32 3.65K 0 0.22UF C21 10K 2.61K NTC R50 7.87K R41 R38 11K -----> R42 C20 0.1UF C18 ---- 820PF 100 ------------OPTIONAL 0.47UF R26 0.039UF C16 1UF C17 C82 R30 604 ------------C81 R109 ---- 10 C15 C13 IN R18 0.01UF 82.5 ----C12 1000PF 330PF ----- ---- R58 IMON B Q12 OUT VSSSENSE 10 IN +5V VIN C DNP OUT IN VCCSENSE IN 0 C67 VSUM- 1 ISEN2 10K R90 VSUM+ R72 3.65K R65 IRF7821 OUT VCORE OPTIONAL ---- IN R40 1 0.22UF C10 0.22UF C9 0.22UF R20 R17 OUT R37 C59 10UF 10UF C27 21 R71 Q9 11 12 13 14 15 16 17 18 19 20 41 Q3 22 C29 2.37K 10UF PHASE1 0.22UF 0.36UH IRF7832 VSUM- VSSP1 ISEN2 0 IRF7832 ISEN1 ISEN3 +5V C30 23 LGATE1 FB IN 24 PWM3 ISL62883HRZ R56 R63 COMP 0 8 25 VCCP VW L1 26 LGATE2 R11 150PF 324K ----ISEN3 IN ISEN2 IN ISEN1 IN A 27 VSSP2 U6 NTC 3.65K VR_TT# Q8 VSUM+ PHASE2 DNP IRF7821 Q2 VCORE 28 10UF 390PF RBIAS 29 10UF C35 536 9 10 UGATE2 C22 R7 C11 30 BOOT2 PSI# 1UF 39PF R10 PGOOD 10UF C33 CLK_EN# DPRSLPVR VR_ON VID6 VID5 VID4 VID3 VID2 VID1 VID0 R19 R12 499 7 0.36UH IRF7832 Q11 OUT C6 C7 6 8 0.22UF C4 1000PF 8.66K DNP ------R6 ------R4 5 IRF7832 Q5 OUT 560PF 2.37K ------------- NTC TBD 4 0.22UF OUT ------------C83 R110 TBD 147K R9 C31 0 OUT +5V TITLE: ISL62883 DATE: ENGINEER: PAGE: REFERENCE DESIGN 3-PHASE, DCR SENSING 3 JIA WEI 2 JULY 2009 1 OF 1 1 A ISL62883, ISL62883B R8 3 R57 OUT 1 2 R16 VR_TT# OUT ---- OPTIONAL B D OUT IN C C3 DNP Q10 OUT IN ------- OPTIONAL IRF7821 Q4 40 39 38 37 36 35 34 33 32 31 1.91K PGOOD OUT PSI# +1.1V C28 R23 IN ISEN1 VSEN RTN ISUMISUM+ VDD VIN IMON BOOT1 UGATE1 28 +3.3V 1 L2 1.91K D 2 IN C24 VID0 IN VID1 IN VID2 IN VID3 IN VID4 IN VID5 IN VID6 IN VR_ON IN DPRSLPVR IN CLK_EN# OUT 3 10UF C34 6 56UF 7 56UF C25 8 FN6891.4 June 21, 2011 ISL62883, ISL62883B Reference Design Bill of Materials QTY REFERENCE VALUE DESCRIPTION 1 C11 390pF Multilayer Cap, 16V, 10% GENERIC H1045-00391-16V10 SM0603 1 C12 330pF Multilayer Cap, 16V, 10% GENERIC H1045-00331-16V10 SM0603 1 C13 1000pF Multilayer Cap, 16V, 10% GENERIC H1045-00102-16V10 SM0603 1 C15 0.01µF Multilayer Cap, 16V, 10% GENERIC H1045-00103-16V10 SM0603 3 C16, C22, C26 1µF Multilayer Cap, 16V, 20% GENERIC H1045-00105-16V20 SM0603 1 C18 0.47µF Multilayer Cap, 16V, 10% GENERIC H1045-00474-16V10 SM0603 1 C20 0.1µF Multilayer Cap, 16V, 10% GENERIC H1045-00104-16V10 SM0603 8 C21, C7, C9, C10, C17, C30, C31, C32 0.22µF Multilayer Cap, 16V, 10% GENERIC H1045-00224-16V10 SM0603 2 C24, C25 56µF Radial SP Series Cap, 25V, 20% 25SP56M CASE-CC 6 C27, C28, C29, C33, C34, C35 10µF Multilayer Cap, 25V, 20% GENERIC H1065-00106-25V20 SM1206 1 C3 150pF Multilayer Cap, 16V, 10% GENERIC H1045-00151-16V10 SM0603 4 C39, C44, C52, C57 270µF SPCAP, 2V, 4.5MOHM POLYMER CAP, 2.5V, 4.5mΩ 1 C4 24 C40-C43, C47-C50, C53-C56, C59-C69, C78 10µF Multilayer Cap, 6.3V, 20% MURATA PANASONIC TDK 1 C6 39pF Multilayer Cap, 16V, 10% 1 C81 820pF 1 C82 1 C83 3 1000pF Multilayer Cap, 16V, 10% MANUFACTURER SANYO PANASONIC KEMET PACKAGE EEFSX0D471E4 T520V477M2R5A(1)E4R5 H1045-00102-16V10 SM0603 GRM21BR61C106KE15L ECJ2FB0J106K C2012X5R0J106K SM0805 GENERIC H1045-00390-16V10 SM0603 Multilayer Cap, 16V, 10% GENERIC H1045-00821-16V10 SM0603 0.039µF Multilayer Cap, 16V, 10% GENERIC H1045-00393-16V10 SM0603 Multilayer Cap, 16V, 10% GENERIC H1045-00561-16V10 SM0603 L1, L2, L3 0.36µH Inductor, Inductance 20%, DCR 5% NEC-TOKIN PANASONIC MPCH1040LR36 ETQP4LR36AFC 10mmx10mm 3 Q2, Q4, Q6 N-Channel Power MOSFET IR IRF7821 PWRPAKSO8 6 Q3, Q5, Q7, Q9, Q11, Q13 N-Channel Power MOSFET IR IRF7832 PWRPAKSO8 3 Q8, Q10, Q12 DNP 1 R10 536 Thick Film Chip Resistor, 1% GENERIC H2511-05360-1/16W1 SM0603 1 R109 100 Thick Film Chip Resistor, 1% GENERIC H2511-01000-1/16W1 SM0603 1 R11 2.37k Thick Film Chip Resistor, 1% GENERIC H2511-02371-1/16W1 SM0603 1 R110 2.37k Thick Film Chip Resistor, 1% GENERIC H2511-02371-1/16W1 SM0603 1 R12 499 Thick Film Chip Resistor, 1% GENERIC H2511-04990-1/16W1 SM0603 1 R16 147k Thick Film Chip Resistor, 1% GENERIC H2511-01473-1/16W1 SM0603 2 R17, R18 10 Thick Film Chip Resistor, 1% GENERIC H2511-00100-1/16W1 SM0603 4 R19, R71, R72, R73 10k Thick Film Chip Resistor, 1% GENERIC H2511-01002-1/16W1 SM0603 1 R23 1.91k Thick Film Chip Resistor, 1% GENERIC H2511-01911-1/16W1 SM0603 1 R26 82.5 Thick Film Chip Resistor, 1% GENERIC H2511-082R5-1/16W1 SM0603 5 R20, R40, R56, R57, R58 0 Thick Film Chip Resistor, 1% GENERIC H2511-00R00-1/16W1 SM0603 1 R30 604 Thick Film Chip Resistor, 1% GENERIC H2511-06040-1/16W1 SM0603 560pF 29 GENERIC PART NUMBER FN6891.4 June 21, 2011 ISL62883, ISL62883B Reference Design Bill of Materials (Continued) QTY REFERENCE VALUE DESCRIPTION 4 R37, R88, R90, R92 1 Thick Film Chip Resistor, 1% GENERIC H2511-01R00-1/16W1 SM0603 1 R38 11k Thick Film Chip Resistor, 1% GENERIC H2511-01102-1/16W1 SM0603 1 R4 DNP 1 R41 2.61k Thick Film Chip Resistor, 1% GENERIC H2511-02611-1/16W1 SM0603 1 R42 ERT-J1VR103J SM0603 1 R50 7.87k Thick Film Chip Resistor, 1% GENERIC H2511-07871-1/16W1 SM0603 1 R6 8.66k Thick Film Chip Resistor, 1% GENERIC H2511-08662-1/16W1 SM0603 3 R63, R65, R67 3.65k Thick Film Chip Resistor, 1% GENERIC H2511-03651-1/16W1 SM0805 2 R8, R9 DNP 1 R7 324k Thick Film Chip Resistor, 1% GENERIC H2511-03243-1/16W1 SM0603 1 U3 Synchronous Rectified MOSFET Driver INTERSIL ISL6208CBZ SOIC8_150_50 1 U6 IMVP-6.5 PWM Controller INTERSIL ISL62883HRTZ QFN-40 10k NTC Thermistor, 10k NTC 30 MANUFACTURER PANASONIC PART NUMBER PACKAGE FN6891.4 June 21, 2011 ISL62883, ISL62883B Typical Performance 92 1.10 90 1.08 1.06 86 VIN = 8V 84 1.04 VIN = 12V 82 VOUT (V) EFFICIENCY(%) 88 VIN = 19V 80 78 1.02 1.00 0.98 76 0.96 74 0.94 72 70 0 5 10 15 20 25 30 35 40 45 50 55 60 0.92 65 0 5 10 15 20 25 IOUT (A) 30 35 40 IOUT (A) 45 50 55 60 65 FIGURE 29. 3-PHASE CCM LOAD LINE, VID = 1.075V, VIN1 = 8V, VIN2 = 12.6V AND VIN3 = 19V FIGURE 28. 3-PHASE CCM EFFICIENCY, VID = 1.075V, VIN1 = 8V, VIN2 = 12.6V AND VIN3 = 19V 95 0.875 85 VOUT (V) EFFICIENCY (%) 0.885 VIN = 8V 90 80 VIN = 12V 75 VIN = 19V 0.865 0.855 70 0.845 65 0.835 60 0 1 2 3 4 5 6 7 8 9 IOUT(A) 10 11 12 13 14 15 0.825 0 1 2 3 4 5 6 10 11 12 13 14 15 7 8 9 IOUT (A) FIGURE 31. 2-PHASE CCM LOAD LINE, VID = 0.875V, VIN1 = 8V, VIN2 = 12.6V AND VIN3 = 19V FIGURE 30. 2-PHASE CCM EFFICIENCY, VID = 0.875V, VIN1 = 8V, VIN2 = 12.6V AND VIN3 = 19V 95 0.885 85 0.875 VIN = 8V VOUT (V) EFFICIENCY (%) 90 80 75 VIN = 12V 70 65 60 0.1 0.865 0.855 0.845 0.835 VIN = 19V 1 10 IOUT (A) FIGURE 32. 1-PHASE DEM EFFICIENCY, VID = 0.875V, VIN1 = 8V, VIN2 = 12.6V AND VIN3 = 19V 31 100 0.825 0 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 IOUT (A) FIGURE 33. 1-PHASE DEM LOAD LINE, VID = 0.875V, VIN1 = 8V, VIN2 = 12.6V AND VIN3 = 19V FN6891.4 June 21, 2011 ISL62883, ISL62883B Typical Performance (Continued) FIGURE 34. SOFT-START, VIN = 19V, IO = 0A, VID = 0.95V, Ch1: PHASE1, Ch2: VO, Ch3: PHASE2, Ch4: PHASE3 FIGURE 35. SHUT DOWN, VIN = 19V, IO = 1A, VID = 0.95V, Ch1: PHASE1, Ch2: VO, Ch3: PHASE2, Ch4: PHASE3 FIGURE 36. CLK_EN# DELAY, VIN = 19V, IO = 2A, VID = 1.5V, Ch1: PHASE1, Ch2: VO, Ch3: IMON, Ch4: CLK_EN# FIGURE 37. PRE-CHARGED START UP, VIN = 19V, VID = 0.95V, Ch1: PHASE1, Ch2: VO, Ch3: IMON, Ch4: VR_ON IMON-VSSSENSE (mV) 1000 900 800 700 600 500 VIN = 19V 400 300 SPEC VIN = 12V 200 100 0 0 FIGURE 38. STEADY STATE, VIN = 19V, IO = 51A, VID = 0.95V, Ch1: PHASE1, Ch2: VO, Ch3: PHASE2, Ch4: PHASE3 32 VIN = 8V 5 10 15 20 25 30 IOUT (A) 35 40 45 50 FIGURE 39. IMON, VID = 1.075V FN6891.4 June 21, 2011 ISL62883, ISL62883B Typical Performance (Continued) FIGURE 40. LOAD TRANSIENT RESPONSE WITH OVERSHOOT REDUCTION FUNCTION DISABLED, VIN = 12V, SV CLARKSFIELD CPU TEST CONDITION: VID = 0.95V, IO = 12A/51A, di/dt = “FASTEST”, LL = 1.9mΩ FIGURE 41. LOAD TRANSIENT RESPONSE WITH OVERSHOOT REDUCTION FUNCTION DISABLED, VIN = 12V, SV CLARKSFIELD CPU TEST CONDITION: VID = 0.95V, IO = 12A/51A, di/dt = “FASTEST”, LL = 1.9mΩ FIGURE 42. LOAD TRANSIENT RESPONSE WITH OVERSHOOT REDUCTION FUNCTION DISABLED, VIN = 12V, SV CLARKSFIELD CPU TEST CONDITION: VID = 0.95V, IO = 12A/51A, di/dt = “FASTEST”, LL = 1.9mΩ FIGURE 43. LOAD TRANSIENT RESPONSE WITH OVERSHOOT REDUCTION FUNCTION DISABLED, VIN = 12V, SV CLARKSFIELD CPU TEST CONDITION: VID = 0.95V, IO = 12A/51A, di/dt = “FASTEST”, LL = 1.9mΩ FIGURE 44. 2-PHASE MODE LOAD INSERTION RESPONSE WITH OVERSHOOT REDUCTION FUNCTION DISABLED, 3-PHASE CONFIGURATION, PSI# = 0, DPRSLPVR = 0, VIN = 12V, VID = 0.875V, IO = 4A/17A, di/d = “FASTEST FIGURE 45. 2-PHASE MODE LOAD INSERTION RESPONSE WITH OVERSHOOT REDUCTION FUNCTION DISABLED, 3-PHASE CONFIGURATION, PSI# = 0, DPRSLPVR=0, VIN = 12V, VID = 0.875V, IO = 4A/17A, di/dt = “FASTEST” 33 FN6891.4 June 21, 2011 ISL62883, ISL62883B Typical Performance (Continued) FIGURE 46. PHASE ADDING/DROPPING (PSI# TOGGLE), IO = 15A, VID = 1.075V, Ch1: PHASE1, Ch2: VO, Ch3: PHASE2, Ch4: PHASE3 FIGURE 47. DEEPER SLEEP MODE ENTRY/EXIT, IO = 1.5A, HFM VID = 1.075V, LFM VID = 0.875V, DEEPER SLEEP VID = 0.875V, Ch1: PHASE1, Ch2: VO, Ch3: PHASE2, Ch4: PHASE3 FIGURE 48. VID ON THE FLY, 1.075V/0.875V, 3-PHASE CONFIGURATION, PSI#=1, DPRSLPVR=0, Ch1: PHASE1, Ch2: VO, Ch3: PHASE2, Ch4: PHASE3 FIGURE 49. VID ON THE FLY, 1.075V/0.875V, 3-PHASE CONFIGURATION, PSI#=0, DPRSLPVR=0, Ch1: PHASE1, Ch2: VO, Ch3: PHASE2, Ch4: PHASE3 FIGURE 50. VID ON THE FLY, 1.075V/0.875V, 3-PHASE CONFIGURATION, PSI# = 0, DPRSLPVR = 1, Ch1: PHASE1, Ch2: VO, Ch3: PHASE2, Ch4: PHASE3 FIGURE 51. VID ON THE FLY, 1.075V/0.875V, 3-PHASE CONFIGURATION, PSI# = 1, DPRSLPVR = 1, Ch1: PHASE1, Ch2: VO, Ch3: PHASE2, Ch4: PHASE3 34 FN6891.4 June 21, 2011 ISL62883, ISL62883B Typical Performance (Continued) Phase Margin Gain FIGURE 53. REFERENCE DESIGN LOOP GAIN T2(s) MEASUREMENT RESULT 1000 5.0 900 4.5 800 4.0 700 3.5 600 3.0 500 VIN = 19V 400 300 SPEC 2.0 1.0 VIN = 8V 100 5 10 15 20 25 30 IOUT (A) PSI# = 0, DPRSLPVR = 0, 2-PHASE CCM 2.5 1.5 VIN = 12V 200 0 0 Z(f) (mΩ) IMON-VSSSENSE (mV) FIGURE 52. LOAD TRANSIENT RESPONSE WITH OVERSHOOT REDUCTION FUNCTION ENABLED, VIN = 12V, SV CLARKSFIELD CPU TEST CONDITION: VID = 0.95V, IO = 12A/51A, di/dt = “FASTEST”, LL = 1.9mΩ, Ch1: LGATE1, Ch2: VO, Ch3: LGATE2, Ch4: ISL6208 LGATE PSI# = 1, DPRSLPVR = 0, 3-PHASE CCM 0.5 35 FIGURE 54. IMON, VID = 1.075V 40 45 50 0.0 1k 1M 100k 10k FREQUENCY (Hz) FIGURE 55. REFERENCE DESIGN FDIM RESULT For additional products, see www.intersil.com/product_tree Intersil products are manufactured, assembled and tested utilizing ISO9000 quality systems as noted in the quality certifications found at www.intersil.com/design/quality Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries. For information regarding Intersil Corporation and its products, see www.intersil.com 35 FN6891.4 June 21, 2011 ISL62883, ISL62883B Package Outline Drawing L40.5x5 40 LEAD THIN QUAD FLAT NO-LEAD PLASTIC PACKAGE Rev 1, 9/10 5.00 4X 3.60 A B 36X 0.40 6 PIN #1 INDEX AREA (4X) 3.50 5.00 6 PIN 1 INDEX AREA 0.15 40X 0.4± 0 .1 BOTTOM VIEW TOP VIEW 0.20 b 4 0.10 M C A B PACKAGE OUTLINE 0.40 0.750 SEE DETAIL “X” SIDE VIEW 3.50 5.00 0.050 // 0.10 C C BASE PLANE SEATING PLANE 0.08 C (36X 0.40 0.2 REF (40X 0.20) C (40X 0.60) 5 0.00 MIN 0.05 MAX TYPICAL RECOMMENDED LAND PATTERN DETAIL "X" NOTES: 1. Dimensions are in millimeters. Dimensions in ( ) for Reference Only. 2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994. 3. Unless otherwise specified, tolerance : Decimal ± 0.05 4. Dimension b applies to the metallized terminal and is measured between 0.15mm and 0.27mm from the terminal tip. 5. Tiebar shown (if present) is a non-functional feature. 6. The configuration of the pin #1 identifier is optional, but must be located within the zone indicated. The pin #1 identifier may be 7. JEDEC reference drawing: MO-220WHHE-1 either a mold or mark feature. 36 FN6891.4 June 21, 2011 ISL62883, ISL62883B Package Outline Drawing L48.6x6 48 LEAD THIN QUAD FLAT NO-LEAD PLASTIC PACKAGE Rev 1, 4/07 4X 4.4 6.00 44X 0.40 A B 6 PIN 1 INDEX AREA 6 PIN #1 INDEX AREA 48 37 1 6.00 36 4 .40 ± 0.15 25 12 0.15 (4X) 13 24 0.10 M C A B 0.05 M C TOP VIEW 48X 0.45 ± 0.10 4 48X 0.20 BOTTOM VIEW SEE DETAIL "X" 0.10 C BASE PLANE MAX 0.80 ( SEATING PLANE 0.08 C ( 44 X 0 . 40 ) ( 5. 75 TYP ) C SIDE VIEW 4. 40 ) C 0 . 2 REF 5 ( 48X 0 . 20 ) 0 . 00 MIN. 0 . 05 MAX. ( 48X 0 . 65 ) DETAIL "X" TYPICAL RECOMMENDED LAND PATTERN NOTES: 1. Dimensions are in millimeters. Dimensions in ( ) for Reference Only. 2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994. 3. Unless otherwise specified, tolerance : Decimal ± 0.05 4. Dimension b applies to the metallized terminal and is measured between 0.15mm and 0.30mm from the terminal tip. 5. Tiebar shown (if present) is a non-functional feature. 6. The configuration of the pin #1 identifier is optional, but must be located within the zone indicated. The pin #1 identifier may be either a mold or mark feature. 37 FN6891.4 June 21, 2011