AP6502 340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER Description Pin Assignments The AP6502 is a 340kHz switching frequency external compensated synchronous DCDC buck converter. It has integrated low RDSON high and low side MOSFETs. ( Top View ) The AP6502 features current mode control operation, which enables fast transient response times and easy loop stabilization. The AP6502 simplifies board layout and reduces space requirements with its high level of integration and minimal need for external components, making it ideal for distributed power architectures. BS 1 8 SS IN 2 7 EN SW 3 6 COMP GND 4 5 FB SO-8EP The AP6502 is available in a standard Green SO-8EP package with exposed PAD for improved thermal performance and is RoHS compliant. Figure 1. Package Pin Out Features Applications • VIN 4.75V to 23V • 2A continuous Output Current, 3A Peak • • • • • • • • • • VOUT adjustable to 0.925 to 20V • 340kHz switching frequency • Programmable Soft-Start • Enable pin • Protection • o OCP o Thermal Shutdown Gaming Consoles Flat Screen TV sets and Monitors Set Top Boxes Distributed power systems Home Audio Consumer electronics Network Systems FPGA, DSP and ASIC Supplies Green Electronics Lead Free Finish/ RoHS Compliant (Note 1) Note: 1. EU Directive 2002/95/EC (RoHS). All applicable RoHS exemptions applied. Please visit our website at http://www.diodes.com/products/lead_free.html. Typical Application Circuit 100 90 VIN = 5V EFFICIENCY (%) NEW PRODUCT The AP6502 enables continues load current of up to 2A with efficiency as high as 95%. VIN = 12V 80 70 60 50 VOUT = 3.3V L = 10µH 40 0 0.4 0.8 1.2 1.6 LOAD CURRENT (A) Efficiency vs. Load Current AP6502 Document Number: DS35423 Rev. 2 - 2 2 Figure 2. Typical Application Circuit 1 of 12 www.diodes.com September 2011 © Diodes Incorporated AP6502 340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER Pin Descriptions Name Description 1 BS High-Side Gate Drive Boost Input. BS supplies the drive for the high-side N-Channel MOSFET switch. Connect a 0.01µF or greater capacitor from SW to BS to power the high side switch. 2 IN Power Input. IN supplies the power to the IC, as well as the step-down converter switches. Drive IN with a 4.75V to 23V power source. Bypass IN to GND with a suitably large capacitor to eliminate noise on the input to the IC. See Input Capacitor. 3 SW Power Switching Output. SW is the switching node that supplies power to the output. Connect the output LC filter from SW to the output load. Note that a capacitor is required from SW to BS to power the high-side switch. 4 GND Ground (Connect the exposed pad to Pin 4). 5 FB Feedback Input. FB senses the output voltage and regulates it. Drive FB with a resistive voltage divider connected to it from the output voltage. The feedback threshold is 0.925V. See Setting the Output Voltage. 6 COMP Compensation Node. COMP is used to compensate the regulation control loop. Connect a series RC network from COMP to GND. In some cases, an additional capacitor from COMP to GND is required. See Compensation Components. 7 EN Enable Input. EN is a digital input that turns the regulator on or off. Drive EN high to turn on the regulator; low to turn it off. Attach to IN with a 100kΩ pull up resistor for automatic startup. 8 SS Soft-Start Control Input. SS controls the soft-start period. Connect a capacitor from SS to GND to set the soft-start period. A 0.1µF capacitor sets the soft-start period to 15ms. To disable the soft-start feature, leave SS floating. Functional Block Diagram Figure 3. Functional Block Diagram + OVP 1.1V OSCILLATOR E + 5 2 IN 1 BS - FB CURRENT SENSE AMPLIFIER RAMP - + NEW PRODUCT Pin # 100/340 KHz CLK Logic 0.3 V 100mΩ + SS 8 3 - + + 6uA ERROR AMPLIFIER 0.923 V CURRENT COMPARATOR 4 COMP SW 100mΩ GND 6 + 2.5V EN OK - disable LOCKOUT COMPARATOR IN < 4.10V IN EN + 7 0.9V AP6502 Document Number: DS35423 Rev. 2 - 2 SHUTDOWN COMPARATOR 2 of 12 www.diodes.com INTERNAL REGULATORS 5V September 2011 © Diodes Incorporated AP6502 340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER Absolute Maximum Ratings (Note 2) NEW PRODUCT Symbol Parameter VIN Supply Voltage VSW Switch Node Voltage VBS Bootstrap Voltage VFB Feedback Voltage VEN Enable/UVLO Voltage VCOMP Comp Voltage TST Storage Temperature TJ Junction Temperature TL Lead Temperature ESD Susceptibility (Note 3) HBM MM CDM Rating Unit -0.3 to 26 -1.0 to VIN+0.3 VSW-0.3 to VSW + 6 –0.3V to +6 –0.3V to +6 –0.3V to +6 -65 to +150 +150 +260 V V V V V V °C °C °C 4 400 1 kV V kV Human Body Model Machine Model Charged Device Model Notes: 2. Stresses greater than the 'Absolute Maximum Ratings' specified above, may cause permanent damage to the device. These are stress ratings only; functional operation of the device at these or any other conditions exceeding those indicated in this specification is not implied. Device reliability may be affected by exposure to absolute maximum rating conditions for extended periods of time. 3. Semiconductor devices are ESD sensitive and may be damaged by exposure to ESD events. Suitable ESD precautions should be taken when handling and transporting these devices. Thermal Resistance (Note 4) Symbol Note: Rating Unit θJA Junction to Ambient Parameter 74 °C/W θJC Junction to Case 16 °C/W 4. Test condition for SO-8EP: Measured on approximately 1” square of 1 oz copper Recommended Operating Conditions (Note 5) Symbol VIN TA Note: Parameter Supply Voltage Operating Ambient Temperature Range Min Max Unit 4.75 -40 23 +85 V °C 5. The device function is not guaranteed outside of the recommended operating conditions. AP6502 Document Number: DS35423 Rev. 2 - 2 3 of 12 www.diodes.com September 2011 © Diodes Incorporated AP6502 340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER Electrical Characteristics (VIN = 12V, TA = +25°C, unless otherwise noted) NEW PRODUCT Symbol Parameter Test Conditions Min Typ. Max Unit IIN Shutdown Supply Current VEN = 0V 0.3 3.0 µA IIN Supply Current (Quiescent) VEN = 2.0V, VFB = 1.0V 0.6 1.5 mA RDS(ON)1 High-Side Switch On-Resistance (Note 6) 100 mΩ RDS(ON)2 Low-Side Switch On-Resistance (Note 6) 100 mΩ Minimum duty cycle 4.4 A 0.9 A ILimit HS Current Limit ILimit LS Current Limit From Drain to Source High-Side Switch Leakage Current VEN = 0V, VSW = 0V, Vsw=12V AVEA Error Amplifier Voltage Gain (Note 5) GEA Error Amplifier Transconductance GCS COMP to Current Sense Transconductance FSW Oscillator Frequency VFB = 0.75V 0 ΔIC = ±10μA FFB Fold-back Frequency VFB = 0V DMAX Maximum Duty Cycle VFB = 800mV TON Minimum On Time VFB Feedback Voltage TA = -40°C to +85°C 300 900 Feedback Overvoltage Threshold VEN_Rising 800 V/V 1000 uA/V 2.8 A/V 340 380 kHz 0.30 fSW 90 % 200 ns 925 950 1.1 mV V 0.7 0.8 0.9 V EN Lockout Threshold Voltage 2.2 2.5 2.7 V 3.80 4.05 EN Lockout Hysteresis VIN Under Voltage Threshold Rising INUVHYS VIN Under Voltage Threshold Hysteresis TSD μA EN Rising Threshold INUVVth Note: 10 220 250 mV 4.40 V mV Soft-Start Current VSS = 0V 6 μA Soft-Start Period CSS = 0.1µF 15 ms 150 °C Thermal Shutdown 6. Guaranteed by design AP6502 Document Number: DS35423 Rev. 2 - 2 4 of 12 www.diodes.com September 2011 © Diodes Incorporated AP6502 340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER Typical Performance Graphs (VIN = 12V, VOUT=3.3V ,TA = +25°C, unless otherwise noted) 0.54 0.52 0.5 0.064 6 0.054 5.8 0.044 0.034 0.024 0.014 0.004 5 10 15 20 25 INPUT VOLTAGE (V) Quiescent Supply Current vs. Input Voltage 0.92 0.918 3.328 0.916 FEEDBACK VOLTAGE (V) 3.33 3.329 3.327 3.326 3.325 3.324 3.323 14.75 19.75 INPUT VOLTAGE (V) Line Regulation 0.906 0.9 -60 24.75 90 90 85 85 80 80 75 EFFICIENCY (%) VIN = 12V 65 60 55 -20 0 20 40 60 80 TEMPERATURE (°C) Feedback Voltage vs. Temperature 40 0 0.4 0.8 1.2 1.6 LOAD CURRENT (A) Efficiency vs. Load Current AP6502 Document Number: DS35423 Rev. 2 - 2 2 VIN = 5V VIN = 12V 65 60 40 0 0 20 40 60 TEMPERATURE (C) Current Limit vs. Temperature 80 100 0 20 40 60 80 TEMPERATURE (°C) Oscillator Frequency vs. Temperature 100 365 360 355 -40 -20 90 70 45 -20 100 50 VOUT = 1.2V L = 3.3µH -40 370 350 -60 100 55 50 45 -40 75 VIN = 5V 70 5.2 375 0.91 0.902 9.75 10 15 20 25 INPUT VOLTAGE (V) Shutdown Supply Current vs. Input Voltage 0.908 0.904 5.4 4.8 -60 5 0.912 3.321 3.32 4.75 0 0.914 3.322 5.6 5 OSCILLATOR FREQUENCY (Khz) 0 OUTPUT VOLTAGE (V) 6.2 CURRENT LIMIT (A) 0.56 0.074 EFFICIENCY (%) SHUTDOWN SUPPLY CURRENT (µA) QUIESCENT SUPPLY CURRENT (mA) 0.58 0.48 EFFICIENCY (%) NEW PRODUCT 0.6 80 70 60 50 VIN = 12V VOUT = 5V L = 10µH VOUT = 1.8V L = 3.3µH 0.4 0.8 1.2 1.6 LOAD CURRENT (A) Efficiency vs. Load Current 5 of 12 www.diodes.com 2 40 0 0.4 0.8 1.2 1.6 LOAD CURRENT (A) Efficiency vs. Load Current September 2011 © Diodes Incorporated 2 AP6502 340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER Typical Performance Characteristics (VIN = 12V, VOUT=3.3V ,L=3.3µH, C1=22uF, C2=47uF, TA = +25°C, unless otherwise noted) Steady State Test 2A Startup Through Enable_no load Time-2us/div Time-2us/div Time-10ms/div Startup Through Enable 2A Shutdown Through Enable_no load Shutdown Through Enable 2A Time-2ms/div Time-10ms/div Time-5ms/div Load Transient Test 1.0A to 2.0A Short Circuit Test Short Circuit Recovery Time-100us/div Time-20us/div Time-20us/div NEW PRODUCT Steady State Test no load AP6502 Document Number: DS35423 Rev. 2 - 2 6 of 12 www.diodes.com September 2011 © Diodes Incorporated AP6502 340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER NEW PRODUCT Applications Information Theory of Operation External Soft Start The AP6502 is a 2A current mode control, synchronous buck regulator with built in power MOSFETs. Current mode control assures excellent line and load regulation and a wide loop bandwidth for fast response to load transients. Figure 3 depicts the functional block diagram of AP6502. Soft start is traditionally implemented to prevent the excess inrush current. This in turn prevents the converter output voltage from overshooting when it reaches regulation. The AP6502 has an internal current source with a soft start capacitor to ramp the reference voltage from 0V to 0.925V. The soft start current is 6uA. The soft start sequence is reset when there is a Thermal Shutdown, Under Voltage Lockout (UVLO) or when the part is disabled using the EN pin. The operation of one switching cycle can be explained as follows. At the beginning of each cycle, HS (high-side) MOSFET is off. The EA output voltage is higher than the current sense amplifier output, and the current comparator’s output is low. The rising edge of the 340kHz oscillator clock signal sets the RS Flip-Flop. Its output turns on HS MOSFET. The current sense amplifier is reset for every switching cycle. When the HS MOSFET is on, inductor current starts to increase. The Current Sense Amplifier senses and amplifies the inductor current. Since the current mode control is subject to sub-harmonic oscillations that peak at half the switching frequency, Ramp slope compensation is utilized. This will help to stabilize the power supply. This Ramp compensation is summed to the Current Sense Amplifier output and compared to the Error Amplifier output by the PWM Comparator. When the sum of the Current Sense Amplifier output and the Slope Compensation signal exceeds the EA output voltage, the RS Flip-Flop is reset and HS MOSFET is turned off. For one whole cycle, if the sum of the Current Sense Amplifier output and the Slope Compensation signal does not exceed the EA output, then the falling edge of the oscillator clock resets the Flip-Flop. The output of the Error Amplifier increases when feedback voltage (VFB) is lower than the reference voltage of 0.925V. This also increases the inductor current as it is proportional to the EA voltage. If in one cycle the current in the power MOSFET does not reach the COMP set current value, the power MOSFET will be forced to turn off. When the HS MOSFET turns off, the synchronous LS MOSFET turns on until the next clock cycle begins. There is a “dead time” between the HS turn off and LS turn on that prevents the switches from “shooting through” from the input supply to ground. The voltage loop is compensated through an internal transconductance amplifier and can be adjusted through the external compensation components. Enable The enable (EN) input allows the user to control turning on or off the regulator. To enable the regulator EN must be pulled above the ‘EN Rising Threshold’ and to disable the regulator EN must be pulled below ‘EN falling Threshold’ (EN rising threshold – En threshold Hysteresis). AP6502 Document Number: DS35423 Rev. 2 - 2 External Soft Start can be calculated from the formula below: ISS = C * DV DT Where; Iss = Soft Start Current C = External Capacitor DV=change in feedback voltage from 0V to maximum voltage DT = Soft Start Time Current Limit Protection In order to reduce the total power dissipation and to protect the application, AP6502 has cycle-by-cycle current limiting implementation. The voltage drop across the internal high-side MOSFET is sensed and compared with the internally set current limit threshold. This voltage drop is sensed at about 30ns after the HS turns on. When the peak inductor current exceeds the set current limit threshold, current limit protection is activated. During this time the feedback voltage (VFB) drops down. When the voltage at the FB pin reaches 0.3V, the internal oscillator shifts the frequency from the normal operating frequency of 340Khz to a fold-back frequency of 102Khz. The current limit is reduced to 70% of nominal current limit when the part is operating at 102Khz. This low Foldback frequency prevents runaway current. Under Voltage Lockout (UVLO) Under Voltage Lockout is implemented to prevent the IC from insufficient input voltages. The AP6502 has a UVLO comparator that monitors the input voltage and the internal bandgap reference. If the input voltage falls below 4.0V, the AP6502 will latch an under voltage fault. In this event the output will be pulled low and power has to be re-cycled to reset the UVLO fault. Over Voltage Protection When the AP6502 FB pin exceeds 20% of the nominal regulation voltage of 0.925V, the over voltage comparator is tripped and the COMP pin and the SS pin are discharged to GND, forcing the high-side switch off. 7 of 12 www.diodes.com September 2011 © Diodes Incorporated AP6502 340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER Applications Information (cont.) NEW PRODUCT Thermal Shutdown The AP6502 has on-chip thermal protection that prevents damage to the IC when the die temperature exceeds safe margins. It implements a thermal sensing to monitor the operating junction temperature of the IC. Once the die temperature rises to approximately 150°C, the thermal protection feature gets activated .The internal thermal sense circuitry turns the IC off thus preventing the power switch from damage. A hysteresis in the thermal sense circuit allows the device to cool down to approximately 120°C before the IC is enabled again through soft start. This thermal hysteresis feature prevents undesirable oscillations of the thermal protection circuit. Setting the Output Voltage The output voltage can be adjusted from 0.925V to 18V using an external resistor divider. Table 1 shows a list of resistor selection for common output voltages. Resistor R1 is selected based on a design tradeoff between efficiency and output voltage accuracy. For high values of R1 there is less current consumption in the feedback network. However the trade off is output voltage accuracy due to the bias current in the error amplifier. R2 can be determined by the following equation: Compensation Components The AP6502 has an external COMP pin through which system stability and transient response can be controlled. COMP pin is the output of the internal trans-conductance error amplifier. A series capacitor-resistor combination sets a pole-zero combination to control the characteristics of the control system. The DC gain of the voltage feedback loop is given by: A VDC = RLOAD × GCS × A VEA × VFB VOUT Where VFB is the feedback voltage (0.925V), RLOAD is the load resistor value, GCS is the current sense transconductance and AVEA is the error amplifier voltage gain. The control loop transfer function incorporates two poles one is due to the compensation capacitor (C3) and the output resistor of error amplifier, and the other is due to the output capacitor and the load resistor. These poles are located at: fP1 = fP2 = ⎛V ⎞ R1 = R 2 ⋅ ⎜⎜ OUT − 1⎟⎟ ⎝ 0.925 ⎠ GEA 2π × C3 × A VEA 1 2π × C2 × RLOAD Where GEA is the error amplifier trans-conductance. One zero is present due to the compensation capacitor (C3) and the compensation resistor (R3). This zero is located at: fZ1 = Figure 4. Feedback Divider Network When output voltage is low, network as shown in Figure 4 is recommended. Vout(V) 5 3.3 2.5 1.8 1.2 R1(KΩ) 45.3 26.1 16.9 9.53 3 R2(KΩ) 10 10 10 10 10 1 2π × C3 × R3 The goal of compensation design is to shape the converter transfer function to get a desired loop gain. The system crossover frequency where the feedback loop has the unity gain is crucial. A rule of thumb is to set the crossover frequency to below one-tenth of the switching frequency. Use the following procedure to optimize the compensation components: 1. Choose the compensation resistor (R3) to set the desired crossover frequency. Determine the R3 value by the following equation: Table 1—Resistor Selection for Common Output Voltages R3 = 2π × C2 × fc VOUT 2π × C2 × 0.1× fs VOUT × < × GEA × GCS VFB GEA ×GCS VFB Where fC is the crossover frequency, which is typically less than one tenth of the switching frequency. AP6502 Document Number: DS35423 Rev. 2 - 2 10 of 12 www.diodes.com September 2011 © Diodes Incorporated AP6502 340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER Applications Information (cont.) Input Capacitor NEW PRODUCT Compensation Components (cont.) 2. Choose the compensation capacitor (C3) to achieve the desired phase margin set the compensation zero, fZ1, to below one fourth of the crossover frequency to provide sufficient phase margin. Determine the C3 value by the following equation: 2 C3 > π × R3 × fc Where R3 is the compensation resistor value. VOUT (V) Cin/C1 (µF) Cout/C2 (µF) Rc/R3 (kΩ) Cc/C3 (nF) L1 (µH) 1.2 1.8 2.5 3.3 5 12 22 22 22 22 22 22 47 47 47 47 47 47 3.24 6.8 6.8 6.8 6.8 6.8 6.8 6.8 6.8 6.8 6.8 6.8 3.3 3.3 10 10 10 15 Output Capacitor Inductor Calculating the inductor value is a critical factor in designing a buck converter. For most designs, the following equation can be used to calculate the inductor value; VOUT ⋅ (VIN − VOUT ) VIN ⋅ ΔIL ⋅ fSW f SW is the buck converter switching frequency. Choose the inductor ripple current to be 30% of the maximum load current. The maximum inductor peak current is calculated from: IL(MAX) = ILOAD + ΔIL 2 A 1µH to 10µH inductor with a DC current rating of at least 25% percent higher than the maximum load current is recommended for most applications. For highest efficiency, the inductor’s DC resistance should be less than 200mΩ. Use a larger inductance for improved efficiency under light load conditions. Document Number: DS35423 Rev. 2 - 2 Vout capacitor = ΔIinductor * ESR An output capacitor with ample capacitance and low ESR is the best option. For most applications, a 22µF ceramic capacitor will be sufficient. ΔIinductor 2 ) 2 Co = 2 (Δ V + Vout )2 − Vout L(Iout + Peak current determines the required saturation current rating, which influences the size of the inductor. Saturating the inductor decreases the converter efficiency while increasing the temperatures of the inductor and the internal MOSFETs. Hence choosing an inductor with appropriate saturation current rating is important. AP6502 The output capacitor keeps the output voltage ripple small, ensures feedback loop stability and reduces the overshoot of the output voltage. The output capacitor is a basic component for the fast response of the power supply. In fact, during load transient, for the first few microseconds it supplies the current to the load. The converter recognizes the load transient and sets the duty cycle to maximum, but the current slope is limited by the inductor value. Maximum capacitance required can be calculated from the following equation: ESR of the output capacitor dominates the output voltage ripple. The amount of ripple can be calculated from the equation below: Where ΔI L is the inductor ripple current. And The RMS current rating of the input capacitor is a critical parameter that must be higher than the RMS input current. As a rule of thumb, select an input capacitor which has RMs rating that is greater than half of the maximum load current. Due to large dI/dt through the input capacitors, electrolytic or ceramics should be used. If a tantalum must be used, it must be surge protected. Otherwise, capacitor failure could occur. For most applications, a 4.7µF ceramic capacitor is sufficient. Table 2—Resistor Component Selection L= The input capacitor reduces the surge current drawn from the input supply and the switching noise from the device. The input capacitor has to sustain the ripple current produced during the on time on the upper MOSFET. It must hence have a low ESR to minimize the losses. Where ΔV is the maximum output voltage overshoot. PC Board Layout This is a high switching frequency converter. Hence attention must be paid to the switching currents interference in the layout. Switching current from one power device to another can generate voltage transients across the impedances of the interconnecting bond wires and circuit traces. These interconnecting impedances should be minimized by using wide, short printed circuit traces. 9 of 12 www.diodes.com September 2011 © Diodes Incorporated AP6502 340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER NEW PRODUCT Applications Information (cont.) External feedback resistor dividers must be placed close to the FB i 34mm efficiency of the regulator. This solution is also applicable for D > 65%. The bootstrap diode can be a low cost one such as BAT54 or a schottky that has a low Vf. Input capacitor C1 must be placed as close as possible to the IC and to L1. 52mm AP6502 is exposed at the bottom of the package and must be soldered directly to a well designed thermal pad on the PCB. This will help to increase the power dissipation. External Bootstrap Diode It is recommended that an external bootstrap diode be added when the input voltage is no greater than 5V or the 5V rail is available in the system. This helps to improve the Figure 7—External Bootstrap Compensation Components Recommended Diodes: Voltage/Current Part Number Rating B130 30V, 1A SK13 30V, 1A Vendor Diodes Inc Diodes Inc Ordering Information Device AP6502SP-13 Note: 13” Tape and Reel Part Number Suffix Package Code Packaging (Note 7) Quantity SP SO-8EP 2500/Tape & Reel -13 7. Pad layout as shown on Diodes Inc. suggested pad layout document AP02001, which can be found on our website at http://www.diodes.com/datasheets/ap02001.pdf. AP6502 Document Number: DS35423 Rev. 2 - 2 10 of 12 www.diodes.com September 2011 © Diodes Incorporated AP6502 340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER NEW PRODUCT Marking Information Package Outline Dimensions (All Dimensions in mm) Detail "A" Exposed pad 2.4Ref. 3.70/4.10 45° 0.35max. 3.85/3.95 5.90/6.10 7°~9° 7°~9° 1 1 0.15/0.25 Bottom View 1.75max. 1.30/1.50 3.3Ref. 0/0.13 0.254 0.3/0.5 1.27typ 4.85/4.95 1 Gauge Plane Seating Plane 0.62/0.82 Detail "A" 8x-0.60 5.4 Exposed pad 8x-1.55 6x-1.27 Land Pattem Recommendation (Unit:mm) AP6502 Document Number: DS35423 Rev. 2 - 2 11 of 12 www.diodes.com September 2011 © Diodes Incorporated AP6502 340kHz 23V 2A SYNCHRONOUS DC/DC BUCK CONVERTER IMPORTANT NOTICE NEW PRODUCT DIODES INCORPORATED MAKES NO WARRANTY OF ANY KIND, EXPRESS OR IMPLIED, WITH REGARDS TO THIS DOCUMENT, INCLUDING, BUT NOT LIMITED TO, THE IMPLIED WARRANTIES OF MERCHANTABILITY AND FITNESS FOR A PARTICULAR PURPOSE (AND THEIR EQUIVALENTS UNDER THE LAWS OF ANY JURISDICTION). Diodes Incorporated and its subsidiaries reserve the right to make modifications, enhancements, improvements, corrections or other changes without further notice to this document and any product described herein. Diodes Incorporated does not assume any liability arising out of the application or use of this document or any product described herein; neither does Diodes Incorporated convey any license under its patent or trademark rights, nor the rights of others. Any Customer or user of this document or products described herein in such applications shall assume all risks of such use and will agree to hold Diodes Incorporated and all the companies whose products are represented on Diodes Incorporated website, harmless against all damages. Diodes Incorporated does not warrant or accept any liability whatsoever in respect of any products purchased through unauthorized sales channel. Should Customers purchase or use Diodes Incorporated products for any unintended or unauthorized application, Customers shall indemnify and hold Diodes Incorporated and its representatives harmless against all claims, damages, expenses, and attorney fees arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized application. Products described herein may be covered by one or more United States, international or foreign patents pending. Product names and markings noted herein may also be covered by one or more United States, international or foreign trademarks. LIFE SUPPORT Diodes Incorporated products are specifically not authorized for use as critical components in life support devices or systems without the express written approval of the Chief Executive Officer of Diodes Incorporated. As used herein: A. Life support devices or systems are devices or systems which: 1. are intended to implant into the body, or 2. support or sustain life and whose failure to perform when properly used in accordance with instructions for use provided in the labeling can be reasonably expected to result in significant injury to the user. B. A critical component is any component in a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or to affect its safety or effectiveness. Customers represent that they have all necessary expertise in the safety and regulatory ramifications of their life support devices or systems, and acknowledge and agree that they are solely responsible for all legal, regulatory and safety-related requirements concerning their products and any use of Diodes Incorporated products in such safety-critical, life support devices or systems, notwithstanding any devices- or systems-related information or support that may be provided by Diodes Incorporated. Further, Customers must fully indemnify Diodes Incorporated and its representatives against any damages arising out of the use of Diodes Incorporated products in such safety-critical, life support devices or systems. Copyright © 2011, Diodes Incorporated www.diodes.com AP6502 Document Number: DS35423 Rev. 2 - 2 12 of 12 www.diodes.com September 2011 © Diodes Incorporated